JP2000084484A - Apparatus for driving ultrasonic vibrator - Google Patents

Abstract

PROBLEM TO BE SOLVED: To drive an ultrasonic vibrator surely at a resonance frequency by connecting a capacitor for correction in series with an ultrasonic vibrator, installing a signal amplifying means and operating means for phase synthesis, and adjusting the amplification ratio of the signal amplifying means corresponding to the electrostatic capacity measured during the stop of the ultrasonic vibrator. SOLUTION: A capacitor 10a for correction is connected in series with an ultrasonic vibrator 1, and a signal amplifying means 20 and an operating means 21 for phase synthesis are installed. In this way, a feedback signal for driving the vibrator 1 at a resonance frequency fr can be detected. Moreover, the electrostatic capacity Cd is measured during the stop of the vibrator 1, and the amplification ratio A of the signal amplifying means 20 is adjusted corresponding to the electrostatic capacity Cd of the vibrator 1. In this way, the ultrasonic vibrator 1 can be driven surely at the resonance frequency fr. Therefore, easy response to diversifying surgical operations, a variety of ultrasonic welding machines, and others can be attained.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a driving device for an ultrasonic vibrator. More specifically, the present invention relates to a driving apparatus for a piezoelectric ultrasonic vibrator such as a Langevin type ultrasonic vibrator used in a surgical ultrasonic treatment apparatus or an ultrasonic welding machine.

[0002]

2. Description of the Related Art General Langevin type ultrasonic transducer 1
Are configured as shown in FIG. The structure is such that the piezoelectric elements 2 and the electrode plates 3 are alternately and successively overlapped, sandwiched between metal blocks 4, and entirely tightened with bolts 5.

[0003] Such an ultrasonic vibrator 1 has a resonance frequency fr that is physically determined from its structure and the like, and has the property that the ultrasonic vibrator 1 vibrates most efficiently when driven at this resonance frequency fr. It is known that an electrical equivalent circuit of the ultrasonic vibrator 1 near the resonance frequency fr is shown as in FIG. That is, a coil component (L) and a capacitor component (C) representing mechanical vibration characteristics (series resonance characteristics), and a resistance component (R) representing a mechanical load.
In addition, an equivalent circuit is formed in which a braking capacitor (Cd) composed of the piezoelectric element 2 and the electrode plate 3 is connected in parallel. Generally, the braking capacitor (Cd) is called the capacitance Cd of the ultrasonic vibrator 1.

When electric energy is applied to the ultrasonic vibrator 1, ultrasonic vibration having the same frequency as the electric energy is generated. Of the applied electric energy, only the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1 is converted to ultrasonic vibration. Therefore, when the impedance (Z) on the series resonance circuit side becomes minimum, the current (iZ) flowing on the series resonance circuit side of the ultrasonic vibrator 1 becomes maximum, and ultrasonic vibration can be generated most efficiently. Here, the impedance (Z) on the series resonance circuit side is Z = R +
j · ω · L + 1 / (j · ω · C).

Therefore, the condition for vibrating the ultrasonic vibrator 1 most efficiently is the coil (L) on the series resonance circuit side.
And the resonance frequency fr at which the capacitor (C) resonates in series,
That is, driving is performed at fr = 1 / (2π√ (LC)). In this case, the coil component (L) and the capacitor component (C) on the series resonance circuit side cancel each other, and the electric equivalent circuit becomes as shown in FIG. 19B, and the current (iZ) flowing through the resistance component (R) is reduced. Converted to ultrasonic vibration.

However, the coil component (L) and the capacitor component (C) change subtly according to load conditions applied to the tip of the ultrasonic vibrator 1, environmental conditions such as temperature, and temporal changes. For this reason, in order to always drive the ultrasonic vibrator 1 most efficiently, the ultrasonic vibrator 1 follows the change of the coil component (L) and the capacitor component (C), that is, the change of the resonance frequency fr by some method. 1 must be driven.

As a method of realizing this, attention is paid to the phase relationship between the voltage (V1) applied to the ultrasonic vibrator 1 and the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1, and PLL control for controlling the oscillation frequency fo of the oscillation source 31 based on a phase difference is known. This utilizes that when the ultrasonic vibrator 1 is driven at the resonance frequency fr, the impedance (Z) of the ultrasonic vibrator 1 on the series resonance circuit side is only the resistance (R). . That is,
The oscillation frequency fo of the oscillation source 31 is controlled so that the voltage (V1) applied to the ultrasonic vibrator 1 and the current (iZ) flowing on the series resonance circuit side of the ultrasonic vibrator 1 have the same phase.

FIG. 15 shows a principle diagram of a PLL control method in which the oscillation frequency fo of the oscillation source 31 follows the resonance frequency fr of the ultrasonic vibrator 1 by the PLL control circuit 30.
In this method, the current flowing through the ultrasonic transducer 1 can be detected by the current detecting means 34a, but the braking capacitor (C
Since the current (iCd) flowing through d) causes a phase shift, the current (iZ) flowing on the series resonance circuit side of the ultrasonic vibrator 1 cannot be accurately detected. That is, the actual resonance frequency f
Since the driving is performed at a frequency different from r, the efficiency of the ultrasonic vibration is reduced.

In order to solve this problem, as shown in FIG. 16, a correction coil 11 (L
A method for connecting d) in parallel is known. For example, Japanese Utility Model Laid-Open Publication No. 54-136943 discloses that the current (iCd) flowing through the braking capacitor (Cd) is corrected by the correction coil 11.
A method is disclosed in which the inductance of the correction coil 11 (Ld) is adjusted to be canceled by the current (iLd) flowing through (Ld). Here, the correction coil 11 (Ld)
Is: fr = 1 / (2π√ (Ld · C
d)) may be adjusted. That is, by causing the braking capacitor (Cd) of the ultrasonic vibrator 1 and the additional correction coil 11 (Ld) to resonate in parallel at the resonance frequency fr, the current (iCd) flowing through the braking capacitor (Cd), which has conventionally been a problem, ) Can be canceled. This can be easily explained from the fact that the current i1 flowing through the current detecting means 34a is expressed by the following equation. i1 = iZ + iCd + iLd = iZ + V · (j · ωr · Cd) + V / (j · ωr · Ld) = iZ + j · V · (ωr · Cd−1 / (ωr · Ld)) where ωr = 2πfr = 1 / ( By substituting √ (Ld · Cd)), the second and third terms on the right side of the equation are canceled, and i1 = iZ is obtained. That is, the current detecting means 34
With a, the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1 can be detected with high accuracy, thereby realizing tracking drive of the resonance frequency fr that can drive the ultrasonic vibrator 1 most efficiently. Such a method is disclosed in
No. 5,275, JP-A-2-277449, JP-A-2-283362, JP-A-2-283363, JP-A-2-283364, JP-A-8-11
No. 7687 and the like.

As another conventional technique, as shown in FIG. 17, an ultrasonic vibrator 1 and three correction capacitors 1 are provided.
There is a circuit in which 0s are connected in a bridge. This circuit converts a signal (Vf1) corresponding to a voltage (V1) applied to the ultrasonic vibrator 1 and a signal (Vf2) corresponding to a current (iZ) flowing on the series resonance circuit side of the ultrasonic vibrator 1 into a PLL. The feedback to the control circuit 30 causes the oscillation of the oscillation source 31 so that the voltage (V1) applied to the ultrasonic vibrator 1 and the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1 have the same phase. The frequency fo is controlled. Hereinafter, the first feedback signal (Vf1) and the second feedback signal (Vf2) are respectively applied to the voltage (V1) applied to the ultrasonic vibrator 1 and the ultrasonic vibrator 1
That the signal is equivalent to the current (iZ) flowing to the series resonance circuit side.

From the circuit diagram of FIG. 17, it is clear that the first feedback signal (Vf1) is the voltage (V1) applied to the ultrasonic vibrator 1 itself. That is, the first feedback signal (Vf1) is expressed by the following equation. Vf1 = V1 = V · Z / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) where the impedance (Z) of the ultrasonic vibrator 1 on the side of the series resonance circuit is Z = R + j · ω · L + 1 / (j · ω ·
C), the voltage applied to the entire ultrasonic transducer 1 and the correction capacitor 10 is V. Ω is the angular frequency.

The voltage (Va) applied to the correction capacitor 10a and the voltage (Vc) applied to the correction capacitor 10c are expressed by the following equations, respectively. Va = V · (Z · Cd / Cc + 1 / (j · ω · Cc)) / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) Vc = V · (Cd / (Cd / (Cd + Cc)))

The second feedback signal (Vf2) is
This is the difference between the voltage (Vc) applied to the correction capacitor 10c and the voltage (Va) applied to the correction capacitor 10a. That is, the second feedback signal (Vf
2) is represented by the following equation. Vf2 = Vc−Va = V · Cd / (Cd + Cc) −V · (Z · Cd / Cc + 1 / (j · ω · Cc)) / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) = J · V / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) / (ω · (Cd + Cc)) = j · V · Z / (Z · (1 + Cd / Cc) + 1 / (j) · Ω · Cc)) / Z / (ω · (Cd + Cc)) = j · V1 / Z / (ω · (Cd + Cc)) = j · iZ / (ω · (Cd + Cc)) ∽j · iZ where iZ is This is a current flowing on the series resonance circuit side of the ultrasonic vibrator 1 and is represented by iZ = V1 / Z. From the formula, it can be seen that the second feedback signal (Vf2) is proportional to the current (iZ) flowing on the series resonance circuit side of the ultrasonic transducer 1, and is a signal whose phase is advanced by 90 degrees.

As described above, the first feedback signal (Vf1) is the voltage (V1) applied to the ultrasonic vibrator 1.
And a second feedback signal (V
Since f2) is a signal (90 degrees advanced) corresponding to the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1, the voltage (V1) applied to the ultrasonic vibrator 1 and the ultrasonic vibration In order to control the current (iZ) flowing to the series resonance circuit side of the element 1 in the same phase, the phase difference between the first feedback signal (Vf1) and the second feedback signal (Vf2) is always 9
What is necessary is just to control so that it may become 0 degree. As a result, follow-up driving of the resonance frequency fr that can drive the ultrasonic transducer 1 most efficiently can be realized. Such a method is disclosed in
86149.

As another conventional technique, as shown in FIG. 18, a correction capacitor 10a (C
There is a circuit in which c) is connected in parallel. This circuit also has a signal (Vf1) corresponding to the voltage (V1) applied to the ultrasonic vibrator 1 and a signal (iZ) corresponding to the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1, as in FIG. (Vf2) is P
By feeding back to the LL control circuit 30, the voltage of the oscillation source 31 is adjusted so that the voltage (V1) applied to the ultrasonic vibrator 1 and the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1 have the same phase. The oscillation frequency fo is controlled. Hereinafter, the first feedback signal (Vf
1) and the second feedback signal (Vf2) are signals corresponding to the voltage (V1) applied to the ultrasonic vibrator 1 and the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1, respectively. Explain that

It is apparent from the circuit diagram of FIG. 18 that the first feedback signal (Vf1) is the voltage (V) applied to the ultrasonic transducer 1 itself. That is,
The first feedback signal (Vf1) is represented by the following equation. Vf1 = V

The second feedback signal (Vf2) is
This is the difference between the current (i1) flowing through the ultrasonic transducer 1 and the signal (A · i2) obtained by multiplying the current (i2) flowing through the correction capacitor 10a by A by the signal amplifying means 20. Assuming that a coefficient for converting the currents detected by the current detection units 34a and 34b into a voltage is k, each of the current detection units 34a and 34b
The output signals from a and 34b are k · i1 and k · i2. Therefore, the second feedback signal (Vf2)
Is represented by the following equation. Vf2 = k · i1−A · k · i2 = k · (iZ + iCd) −A · k · iCc = k · iZ + k (iCd−A · iCc) where A is the amplification factor of the signal amplifying means 20 and iCd
= A · iCc, the second feedback signal (Vf2) is expressed by the following equation. Vf2 = k · iZ ∽iZ From the equation, it can be seen that the second feedback signal (Vf2) is a signal proportional to the current (iZ) flowing to the series resonance circuit side of the ultrasonic transducer 1.

As described above, the first feedback signal (Vf1) is the voltage (V1) applied to the ultrasonic vibrator 1.
And a second feedback signal (V
f2) is a signal corresponding to (proportional to) the current (iZ) flowing on the series resonance circuit side of the ultrasonic vibrator 1, so that the voltage (V1) applied to the ultrasonic vibrator 1 and the voltage of the ultrasonic vibrator 1 The current (iZ) flowing to the series resonance circuit side may be controlled to have the same phase. As a result, follow-up driving of the resonance frequency fr that can drive the ultrasonic transducer 1 most efficiently can be realized. Such a method is disclosed in JP-A-8-117687.

In this method, it is important to set the amplification factor A of the signal amplification means 20 to an appropriate value. As a setting method, first, the oscillation source 31 generates a signal having a frequency f ′ different from the resonance frequency fr of the ultrasonic transducer 1. At this time, the impedance (Z) on the series resonance circuit side of the ultrasonic vibrator 1 has a very large value, and the impedance (Z)
Is open, that is, the impedance (Z) can be ignored. Therefore, as shown in FIG. 19C, the ultrasonic transducer 1 is equivalent to a capacitor having a capacitance of Cd. Therefore, at the frequency f ', the current (i1) detected by the current detecting means 34a becomes equal to the current (iCd) flowing through the braking capacitor (Cd). The correction capacitor 10a is detected by the current detecting means 34b.
The current (i2) flowing through (Cc) is detected, and the two detected currents are compared by the comparison circuit 40, and the amplification factor A of the signal amplifying means 20 is set so that iCd = A · iCc. afterwards,
In order to drive the ultrasonic vibrator 1 to follow the resonance frequency fr,
Start the PLL control.

[0020]

However, there are also the following points to be improved by the above three methods. The braking capacitor (Cd) is characterized in that it changes according to the temperature of the ultrasonic transducer 1 itself. When the ultrasonic vibrator 1 is driven for a long time, the temperature increases due to the heat generated by the ultrasonic vibrator 1 itself, and the capacitance changes (generally increases).
When the ultrasonic vibrator 1 is installed in a low-temperature atmosphere for a long time, the temperature of the ultrasonic vibrator 1 itself decreases, and the capacitance changes (generally decreases). Therefore, in the first method (FIG. 16), a correct feedback signal cannot be obtained because the inductance of the correction coil 11 (Ld) cannot follow the capacitance change of the braking capacitor (Cd), and as a result, the ultrasonic vibration The child 1 may not be able to be driven at the resonance frequency fr. Also, in the second method (FIG. 17),
The balance of the bridge is lost due to a change in the capacitance of the braking capacitor (Cd), so that a correct feedback signal cannot be obtained.
r may not be able to be driven. Also in the third method (FIG. 18), the capacitance of the braking capacitor (Cd) changes while the ultrasonic vibrator 1 is used only by setting the amplification factor A of the signal amplifying means 20 first. In such a case, a correct feedback signal cannot be obtained, and as a result, the ultrasonic transducer 1 cannot be driven at the resonance frequency fr.

The present invention has been made in view of the above-mentioned problems, and has as its object to drive the ultrasonic vibrator 1 reliably and always at its resonance frequency fr, thereby diversifying surgery and other types of surgery. It is an object of the present invention to provide a driving device of the ultrasonic vibrator 1 which can easily cope with the ultrasonic welding or the like which is becoming difficult.

[0022]

Means for Solving the Problems A first invention of the present invention is:
The ultrasonic vibrator 1 is connected to the ultrasonic vibrator 1 in series with the resonance frequency fr by connecting the correction capacitor 10a in series and installing the signal amplifying means 20 and the phase synthesizing arithmetic means 21.
And that the capacitance Cd is measured when the ultrasonic vibrator 1 is not driven, and the signal amplifying means 20 is operated in accordance with the capacitance Cd of the ultrasonic vibrator 1. The inventors have found that the ultrasonic transducer 1 can be reliably driven at the resonance frequency fr by adjusting the amplification factor A, and have reached the present invention.
Further, the second invention of the present invention provides the correction capacitors 10a, 10b, 10
c, and by installing the signal amplifying means 20 and the arithmetic means 21 for phase synthesis, it has been found that a feedback signal for driving the ultrasonic vibrator 1 at the resonance frequency fr can be detected. When the ultrasonic transducer 1 is not driven, the capacitance Cd is measured.
It has been found that the ultrasonic vibrator 1 can be reliably driven at the resonance frequency fr by adjusting the amplification factor A of the signal amplifying means 20 according to d, and the present invention has been reached.

The present invention relates to an ultrasonic vibrator driving apparatus for driving an ultrasonic vibrator at its resonance frequency, comprising: a. An oscillation source that outputs a high-frequency drive voltage signal; b. A power amplifier circuit for power amplifying an output signal from the oscillation source; c. A correction capacitor connected in series to the ultrasonic transducer; d. Signal amplification means for adjusting the magnitude of at least one of the voltage signals applied to the ultrasonic transducer or the correction capacitor; e. Calculating means for phase-synthesizing the output signal of the signal amplifying means and the voltage signal applied to the ultrasonic transducer or the correcting capacitor; f. A PLL control circuit for controlling the oscillation source to drive the ultrasonic transducer at the resonance frequency by using the output signal of the oscillation source or the voltage signal applied to the ultrasonic transducer and the output signal of the calculating means; g. Capacitance measuring means for measuring capacitance when the ultrasonic vibrator is not driven; h. And a gain adjusting means for adjusting the gain of the signal amplifying means in accordance with the capacitance of the ultrasonic vibrator.

[0024]

BEST MODE FOR CARRYING OUT THE INVENTION Hereinafter, the present invention will be described in detail. The present invention is a drive device having a resonance frequency tracking mechanism for reliably driving the ultrasonic transducer 1 at the resonance frequency fr. This is because the ultrasonic vibrator 1 is driven at the resonance frequency fr as described above.
And that the current (iZ) flowing to the series resonance circuit side of the ultrasonic transducer 1 has the same phase. That is, the voltage (V1) applied to the ultrasonic vibrator 1 and the current (iZ) flowing on the series resonance circuit side of the ultrasonic vibrator 1 are detected by some method, and the oscillation source 31 is controlled so that they have the same phase. A control circuit for adjusting the oscillation frequency fo may be incorporated.

FIG. 1 is a circuit diagram showing one embodiment of a driving device for an ultrasonic vibrator 1 according to the present invention. In FIG. 1, 1 is an ultrasonic transducer, 10a is a correction capacitor, and 33a is a first
Voltage detecting means (the voltage (V applied to the ultrasonic vibrator 1)
1)), 33b are second voltage detecting means (voltage (V2) applied to the correction capacitor 10a), 20 is a signal amplifying means, 21 is a phase synthesizing calculating means (calculating means), and 22 is an amplification factor adjustment. Means, 30 is a PLL control circuit, 31 is an oscillation source, and 32 is a power amplifier circuit.

This circuit includes a signal (Vf1) corresponding to the voltage (V1) applied to the ultrasonic vibrator 1 and a signal (Vf2) corresponding to the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1. ) Is fed back to the PLL control circuit 30 so that the voltage (V) applied to the ultrasonic vibrator 1 is
1) and the oscillation frequency f of the oscillation source 31 so that the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1 has the same phase.
o is controlled. Hereinafter, the first feedback signal (Vf1) and the second feedback signal (Vf2) are the voltage (V1) applied to the ultrasonic vibrator 1 and the current ( The fact that the signal corresponds to iZ) will be described.

According to the circuit diagram of FIG. 1, the first feedback signal (Vf1) is the voltage (Vf) applied to the ultrasonic vibrator 1.
1) It is obvious that it is. That is, the first feedback signal (Vf1) is expressed by the following equation. Vf1 = V1 = V · Z / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) Here, the combined impedance (Z) of the ultrasonic vibrator 1 on the series resonance circuit side is Z = R + j · ω.・ L + 1 / (j ・ ω ・
C), the voltage applied to the entire ultrasonic transducer 1 and the correction capacitor 10a is V. Ω is the angular frequency.

The voltage (V2) applied to the correction capacitor 10a is expressed by the following equation. V2 = V · (Z · Cd / Cc + 1 / (j · ω · Cc)) / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) 10

The second feedback signal (Vf2) is
It is determined by the difference between the voltage (V1) applied to the ultrasonic transducer 1 and the voltage (V2) applied to the correction capacitor 10a,
It is expressed by the following equation. Vf2 = V1−V2 = V · (Z−A · (Z · Cd / Cc + 1 / (j · ω · Cc))) / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) 11 Here Where A is the amplification factor of the signal amplifying means 20, and A = C
By setting c / Cd, the second feedback signal (Vf2) is represented by the following equation. Vf2 = j · V · Z / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) / Z / (ω · Cd) = j · V1 / Z / (ω · Cd) = j · iZ / (Ω ・ Cd) ∽j ・ iZ 1
2 where iZ is a current flowing on the series resonance circuit side of the ultrasonic vibrator 1 and is represented by iZ = V1 / Z. From equation (12), the second feedback signal (Vf2) is equal to the ultrasonic transducer 1
It can be seen that the signal is proportional to the current (iZ) flowing to the series resonance circuit side and has a phase advanced by 90 degrees.

As described above, the first feedback signal (Vf1) is the voltage (V1) applied to the ultrasonic vibrator 1.
And a second feedback signal (V
Since f2) is a signal (90 degrees advanced) corresponding to the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1, the voltage (V1) applied to the ultrasonic vibrator 1 and the ultrasonic vibration In order to control the current (iZ) flowing to the series resonance circuit side of the element 1 in the same phase, the phase difference between the first feedback signal (Vf1) and the second feedback signal (Vf2) is always 9
What is necessary is just to control so that it may become 0 degree. Of course, either the first feedback signal (Vf1) or the second feedback signal (Vf2) may be advanced by 90 degrees or delayed by 90 degrees, and controlled so that those signals have the same phase.

In the present system, by setting the amplification factor A of the signal amplifying means 20 to an appropriate value, the ultrasonic vibrator 1
Can be driven at the resonance frequency fr. That is, the capacitance of the ultrasonic transducer 1 (the braking capacitor (C
Even if d)) changes, it is possible to drive at the resonance frequency fr by adjusting the amplification factor A of the signal amplifying means 20. For example, it is assumed that the capacitance Cd becomes n times, that is, Cd ′ = n · Cd as a result of driving the ultrasonic transducer 1 for a long time. In this case, A = Cc / Cd ′ = Cc /
If it is adjusted to (n · Cd), it can be confirmed that the same relationship as in Equations (12) holds, and it can be seen that the resonance frequency fr can be followed. In other words, it can be said that even the ultrasonic transducer 1 having a slightly different capacitance Cd can be driven at the resonance frequency fr as long as the amplification factor A of the signal amplifying means 20 is adjusted.

FIG. 2 is a circuit diagram showing another embodiment. In FIG. 2, 1 is an ultrasonic transducer, 10a, 10b, 10
c is a correction capacitor, 33a is first voltage detecting means (voltage (V1) applied to the ultrasonic transducer 1), 33b
Is the second voltage detecting means (the voltage (V2) applied to the correcting capacitor 10a), and 33c is the third voltage detecting means (the voltage (V2 applied to the correcting capacitor 10c).
3)), 20 is a signal amplifying means, 21 is a phase synthesizing calculating means (calculating means), 22 is an amplification factor adjusting means, 30 is a PLL
A control circuit, 31 is an oscillation source, and 32 is a power amplifier circuit.
Further, m of the correction capacitors 10b and 10c is an arbitrary real number.

This circuit is an improved version of the circuit in which the ultrasonic transducer 1 and three correction capacitors 10 described in the prior art are connected in a bridge shape. Since the mechanism of the PLL control is the same as that of the conventional type, here, the first feedback signal (Vf1) and the second feedback signal (Vf1) are used.
2) is a voltage (V) applied to the ultrasonic vibrator 1 respectively.
1) and the fact that the signal is equivalent to the current (iZ) flowing to the series resonance circuit side of the ultrasonic transducer 1 will be described.

According to the circuit diagram of FIG. 2, the first feedback signal (Vf1) is the voltage (Vf) applied to the ultrasonic vibrator 1.
1) It is obvious that it is. That is, the first feedback signal (Vf1) is expressed by the following equation. Vf1 = V1 = V · Z / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) 13

The voltage (V2) applied to the correction capacitor 10a and the voltage (V3) applied to the correction capacitor 10c are expressed by the following equations. V2 = V · (Z · Cd / Cc + 1 / (j · ω · Cc)) / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) 14 V3 = V · (Cd / (Cd + Cc)) Fifteen

The second feedback signal (Vf2) is
The difference between the voltage (V3) applied to the correction capacitor 10c and the voltage (V2) applied to the correction capacitor 10a. That is, the second feedback signal (Vf
2) is represented by the following equation. Vf2 = V3−V2 = V · Cd / (Cd + Cc) −A · V · (Z · Cd / Cc + 1 / (j · ω · Cc)) / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc) )) 16 where A is the amplification factor of the signal amplification means 20, and A = 1
, The second feedback signal (V
f2) is represented by the following equation. Vf2 = j · V / (Z · (1 + Cd / Cc) + 1 / (j · ω · Cc)) / (ω · (Cd + Cc)) = j · V · Z / (Z · (1 + Cd / Cc) + 1 / ( j · ω · Cc)) / Z / (ω · (Cd + Cc)) = j · V1 / Z / (ω · (Cd + Cc)) = j · iZ / (ω · (Cd + Cc)) ∽j · iZ 17 iZ is a current flowing on the series resonance circuit side of the ultrasonic transducer 1, and is represented by iZ = V1 / Z. From equation (17), it can be seen that the second feedback signal (Vf2) is a signal that is proportional to the current (iZ) flowing on the series resonance circuit side of the ultrasonic vibrator 1 and whose phase is advanced by 90 degrees.

As described above, the first feedback signal (Vf1) is the voltage (V1) applied to the ultrasonic vibrator 1.
And a second feedback signal (V
Since f2) is a signal (90 degrees advanced) corresponding to the current (iZ) flowing to the series resonance circuit side of the ultrasonic vibrator 1, the voltage (V1) applied to the ultrasonic vibrator 1 and the ultrasonic vibration In order to control the current (iZ) flowing to the series resonance circuit side of the element 1 in the same phase, the phase difference between the first feedback signal (Vf1) and the second feedback signal (Vf2) is always 9
What is necessary is just to control so that it may become 0 degree. Of course, either the first feedback signal (Vf1) or the second feedback signal (Vf2) may be advanced by 90 degrees or delayed by 90 degrees, and controlled so that those signals have the same phase.

In the present system, by setting the amplification factor A of the signal amplifying means 20 to an appropriate value, the ultrasonic vibrator 1
Can be driven at the resonance frequency fr. In equation 17, A = 1 is set, but when the capacitance Cd of the ultrasonic transducer 1 changes, an optimal value of A exists. That is, even if the capacitance (braking capacitor (Cd)) of the ultrasonic transducer 1 changes, the signal amplifying means 2
By adjusting the amplification factor A of 0, it is possible to drive at the resonance frequency fr. For example, as a result of driving the ultrasonic vibrator 1 for a long time, the capacitance Cd becomes n times, that is, Cd ′ =
It is assumed that n · Cd has been reached. In this case, A = (n · Cd +
Cc) / (n · (Cd + Cc)), 13 to 17
It can be confirmed that a relationship similar to the formula is established, and it can be seen that the resonance frequency fr can be followed (a detailed calculation formula is complicated, and is omitted here). In other words, if the amplification factor A of the signal amplifying means 20 is adjusted even in the ultrasonic transducer 1 having a slightly different capacitance Cd, the resonance frequency f
It can be said that it can be driven by r.

FIG. 3 shows another embodiment (modification of FIG. 1). The difference from the embodiment of FIG. 1 is that a phase correction circuit 35 is provided and the output signal of the oscillation source 31 is used as a first feedback signal (Vf1). The first feedback signal (Vf1) needs to be a signal corresponding to the voltage signal (V1) applied to the ultrasonic transducer 1, but by appropriately setting the phase correction amount of the phase correction circuit 35, The output signal of the oscillation source 31 and the voltage signal (V1) applied to the ultrasonic transducer 1 can be in phase. Therefore, the operation characteristics are the same as those of the embodiment of FIG. 1, and the ultrasonic vibrator 1 can be driven at the resonance frequency fr. The voltage signal applied to the ultrasonic vibrator 1 is as large as several tens to several hundreds of volts, and to feed it back to the PLL control circuit 30, the voltage must be reduced by some method. Further, the voltage value of the voltage signal applied to the ultrasonic vibrator 1 changes depending on the driving state of the ultrasonic vibrator 1, and it is necessary to adjust the magnitude of the feedback signal each time. The circuit of FIG. 3 is an embodiment in which such annoyance is eliminated. That is, the output signal of the oscillation source 31 has a constant voltage value irrespective of the driving state of the ultrasonic transducer 1, and the output signal of the oscillation source 31 can be directly used as the first feedback signal (Vf1). FIG. 4 is a modification of FIG. 2, and the operation principle is the same as that of FIG.

FIG. 5 shows another embodiment (modification of FIG. 3). The difference from the embodiment of FIG. 3 is that a coil 36 (Ls) is installed in series with the ultrasonic transducer 1. The principle of feedback is the same as in the embodiment of FIG.
The ultrasonic vibrator 1 can be driven at its resonance frequency fr. The coil 36 (Ls) has an inductance that resonates in series with the combined capacitance of the ultrasonic oscillator 1 and the correction capacitor 10a at the resonance frequency fr of the ultrasonic oscillator 1. That is, the combined capacitance Cs of the ultrasonic vibrator 1 and the correction capacitor 10a is Cs = Cd · Cc
/ (Cd + Cc), the inductance of the coil 36 (Ls) is Ls = (Cd + Cc) / ((ωr) ^2.
Cd · Cc). Here, ωr = 2πfr.
The reason why the coil 36 (Ls) is connected is to make the voltage (V1) applied to the ultrasonic vibrator 1 proportional to the mechanical load (R) applied to the ultrasonic vibrator 1, that is, the mechanical load. This is for realizing the following mechanism.

In this case, even if the output voltage (Vs) of the power amplifying circuit 32 is constant, the voltage (V1) applied to the ultrasonic vibrator 1 is not affected by the mechanical load (R) applied to the ultrasonic vibrator 1. ). This can be explained by the fact that the voltage (V1) applied to the ultrasonic transducer 1 is expressed by the following equation. V1 = Vs ・ (R / (1 + j ・ ωr ・ Cd ・ R)) / (j ・ ωr ・ Ls + 1 / (j ・ ωr ・ Cc) + R / (1 + j ・ ωr ・ Cd ・ R)) = Vs ・ R / ( j · ωr · Ls + 1 / (j · ωr · Cc) + R · ((Cd + Cc) / Cc− (ωr) ² · Ls · Cd) = − j · ωr · Cd · Vs · R 18 where the ultrasonic vibrator 1 is driven at the resonance frequency fr, so that the combined impedance (Z) on the side of the series resonance circuit of the ultrasonic vibrator 1 is Z = R. Resonant frequency f of the acoustic transducer 1
r is almost constant because it is determined by r, and Cd has temperature dependence but does not fluctuate extremely, and the output voltage (Vs) of the power amplifier circuit 32 is constant. It can be seen that (V1) is almost proportional to the mechanical load (R). Since the voltage amplification rate of the general power amplification circuit 32 is constant, the output voltage (Vs) is constant unless the input voltage to the power amplification circuit 32 is changed. If this method is used, the voltage (V1) applied to the ultrasonic vibrator 1 can be maintained even if the output voltage (Vs) of the power amplifier circuit 32 is constant only by connecting the coil 36 (Ls) to the ultrasonic vibrator 1. ) Is proportional to the mechanical load (R) applied to the ultrasonic vibrator 1, so that a mechanical load following mechanism can be automatically realized. Also,
The circuit configuration is greatly simplified as compared with a method of detecting a current flowing in the ultrasonic vibrator 1 that fluctuates with respect to a mechanical load (R) and increasing or decreasing the voltage applied to the ultrasonic vibrator 1. There is also an advantage that you can do it. FIG. 6 is a modification of FIG. 4, and the operation principle is the same as that of FIG.

7 to 10 show another embodiment. These embodiments are effective when it is desired to drive the ultrasonic transducer 1 in a floating state. The principle of the feedback is the same as in the embodiment of FIGS. 1 and 2, and the ultrasonic vibrator 1 can be driven at its resonance frequency fr.

Next, a method for setting the amplification factor A of the signal amplifying means 20 according to the capacitance Cd of the ultrasonic transducer 1 will be described. The method is as follows. (1) When the ultrasonic vibrator 1 is not driven, the current capacitance Cd is measured by the capacitance measuring means 23, and (2) the measurement result of the current capacitance Cd and the known capacitance Cc From A = C
c / Cd is obtained, and (3) the gain A of the signal amplifying means 20 is adjusted so that A = Cc / Cd. FIG. 11 shows the configuration of this method. In FIG.
Is a capacitance measuring means, 24 is a switch means, and 28 is a capacitance measuring oscillation source. This method uses the ultrasonic vibrator 1
Since the time for measuring the capacitance Cd is required, it is very effective when the ultrasonic transducer 1 is driven by intermittent driving in which driving and stopping are repeated.

For measuring the capacitance Cd, the resonance frequency fr
And a capacitance measurement frequency fc different from the above. At the capacitance measurement frequency fc different from the resonance frequency fr, the impedance (Z) on the series resonance circuit side of the ultrasonic vibrator 1 has a very large value, and the impedance (Z) is released, that is, the impedance (Z) Z)
Can be ignored. Therefore, at the capacitance measurement frequency fc, as shown in FIG. 19C, the ultrasonic transducer 1 is equivalent to a capacitor having a capacitance of Cd. Therefore, the capacitance Cd can be easily measured. The resonance frequency fr of the general ultrasonic transducer 1 is 20
kHz or higher, the capacitance measuring frequency fc is 2
The frequency may be set sufficiently lower than 0 kHz, for example, 1 kHz. Although it is possible to select a frequency higher than the resonance frequency fr of the ultrasonic vibrator 1, it may be coincident with the sub-resonance point fr '. Therefore, it is better to select a lower frequency if possible.

The capacitance measuring means 23 of the ultrasonic transducer 1
Can be configured by a method of comparing a reference capacitance Ck and a measured capacitance Cx as shown in FIG. That is, it is a circuit for measuring how many times the measured capacitance Cx is larger than the reference capacitance Ck. In this circuit, the output signal Vo is Vo = (Cx / Ck)
Is represented by Vi. Input signal Vi and reference capacitance C
Since k is known, the measured capacitance Cx can be easily obtained by measuring the output voltage Vo.
In this circuit, the capacitance Cd of the ultrasonic vibrator 1 is replaced by replacing the reference capacitance Ck with the correction capacitor 10a (Cc) and replacing the measured capacitance Cx with the ultrasonic vibrator 1 (Cd). Can be measured.

On the other hand, as shown in FIG. 14 (b), the reference capacitance Ck is replaced by the ultrasonic transducer 1 (Cd), and the measured capacitance Cx is replaced by the correction capacitor 10a (Cc). By setting the input voltage Vi to 1 (V),
The output voltage Vo becomes Vo = Cc / Cd = A, and the amplification factor A of the signal amplifier 20 can be automatically obtained as a voltage value. In the configuration of FIG. 11, this method is adopted.

FIG. 11 shows a configuration when the ultrasonic transducer 1 is not driven. When the ultrasonic vibrator 1 is not driven, the ultrasonic vibrator 1 and the correction capacitor 10a are separated from the power amplifier circuit 32 by the switch 24a, and the capacitance measuring means 23 is connected by the switch 24b. Ultrasonic vibrator 1
And the output signal (measurement result) from the capacitance measuring means 23 is taken into a computer (CPU) by an A / D conversion circuit, and the amplification factor A of the signal amplifying means 20 is set to an appropriate value. Output an appropriate signal. The output signal from the computer (CPU) is output to the signal amplifying means 20 via the D / A conversion circuit. Signal amplification means 20
Is a voltage controlled amplifier (VCA: Voltage) in which the amplification factor A changes according to the output voltage from the D / A conversion circuit.
Control Amplifier).

The configuration when the ultrasonic vibrator 1 is driven is as shown in FIG. The amplification factor A of the signal amplifying means 20 is set to an appropriate value from the capacitance Cd of the ultrasonic transducer 1 measured when the ultrasonic transducer 1 is not driven. Therefore, the ultrasonic vibrator 1 has its resonance frequency fr
Can be driven. While the ultrasonic vibrator 1 is being driven, the amplification factor A of the signal amplifying means 20 is kept constant. When the driving of the ultrasonic transducer 1 is stopped, it is connected again to the capacitance measuring means 23, and the amplification factor A of the signal amplifying means 20 is set to an appropriate value again. In order to perform this efficiently, the drive switch means of the ultrasonic transducer 1 (not shown) and the switch means 24a and 24b may be linked. That is, when the drive switch is turned on, the switch 24a is also turned on and the switch 24b is turned off. Of course, when the drive switch is turned off, the switch 24a is also turned off and the switch 24b is turned on. Then, when the ultrasonic transducer 1 is not driven, the capacitance Cd is automatically measured, and the signal amplifying means 20
Is set to an optimal value each time the driving of the ultrasonic transducer 1 is stopped. The capacitance C when the ultrasonic transducer 1 is not driven
The reason why d is measured every time is that the capacitance Cd changes due to heat generation during driving and cooling / radiation during non-driving, and an appropriate value of the amplification factor A always changes.

As described above, when the ultrasonic transducer 1 is not driven, the amplification factor A of the signal amplifying means 20 is set to an optimum value, and when the ultrasonic transducer 1 is driven, the signal amplifying means is set. The operation is repeated to keep the amplification factor A of 20 constant. Therefore, since this method requires time to measure the capacitance Cd of the ultrasonic vibrator 1, it is very effective when the ultrasonic vibrator 1 is driven by intermittent driving in which driving and stopping are repeated. is there. Generally, the ultrasonic vibrator 1 is rarely continuously driven for a very long time, and is mostly an intermittent drive in which driving and stopping are repeated. FIG. 11 shows an embodiment in which this method is applied to the circuit of FIG.
As shown in FIG. 11, this method can be applied to the circuit of FIG. 2, and the operation principle is the same as that of FIG.

Further, as shown in FIG. 13, a capacitance measuring means 23 may be provided. When the capacitance changes to n · Cd, the amplification factor A is calculated as A = (n · Cd + Cc) / (n · Cd).
(Cd + Cc)) may be set as described above. In the method of FIG. 13, voltage signals (Va and Vc) applied to the correction capacitors 10a and 10c are measured. The measurement results are as follows: Va = n · Cd / (n · Cd
+ Cc), Vc = Cd / (Cd + Cc), and taking their ratio, Vc / Va = (n · Cd + Cc) / (n ·
(Cd + Cc)) = A is obtained. As a result, the amplification factor A
Can be easily obtained. Although the above equation regarding Vc / Va is complicated, actually, the effective value, average value or maximum value of Va and Vc are obtained, and
By taking these ratios, an appropriate value of the amplification factor A can be obtained.

A further advantage of these methods is that the capacitance Cd obtained by the capacitance measuring means 23 can be obtained by the ratio to the capacitance Cc of the correcting capacitor 10a.
Even if the capacitance Cc of the correction capacitor 10a changes to a value different from the conventional value due to a change over time or a change in temperature, the correct amplification factor A required to drive the ultrasonic transducer 1 at the resonance frequency fr is used. It is possible to obtain.
That is, it is possible to eliminate the influence of the change over time or the change in temperature of the capacitance Cc of the correction capacitor 10a.

According to the present invention, the ultrasonic vibrator can be reliably driven at the resonance frequency at all times, and can not only follow the constantly changing mechanical load, but also can change the temperature due to temperature change. Even if the capacitance of the ultrasonic transducer changes, the resonance frequency can be reliably followed. Further, it is possible to flexibly cope with ultrasonic vibrators having different capacitances, and to provide an ultrasonic vibrator driving device with high expandability and flexibility.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a circuit diagram showing an embodiment of the present invention.

FIG. 3 is a circuit diagram showing an embodiment of the present invention.

FIG. 4 is a circuit diagram showing an embodiment of the present invention.

FIG. 5 is a circuit diagram showing an embodiment of the present invention.

FIG. 6 is a circuit diagram showing an embodiment of the present invention.

FIG. 7 is a circuit diagram showing an embodiment of the present invention.

FIG. 8 is a circuit diagram showing an embodiment of the present invention.

FIG. 9 is a circuit diagram showing an embodiment of the present invention.

FIG. 10 is a circuit diagram showing an embodiment of the present invention.

FIG. 11 is a circuit diagram showing an embodiment of the present invention.

FIG. 12 is a circuit diagram showing an embodiment of the present invention.

FIG. 13 is a circuit diagram showing an embodiment of the present invention.

FIG. 14 is a circuit diagram showing a capacitance measuring means.

FIG. 15 is a principle diagram of a PLL control method.

FIG. 16 is a diagram illustrating the principle of a conventional PLL control method.

FIG. 17 is a principle diagram of a conventional PLL control method.

FIG. 18 is a principle diagram of a conventional PLL control method.

FIG. 19 is an electrical equivalent circuit diagram of the piezoelectric ultrasonic vibrator.

FIG. 20 is a structural diagram of a Langevin type ultrasonic transducer.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Ultrasonic vibrator 2 Piezoelectric element 3 Electrode plate 4 Metal block 5 Bolt 10 Correction capacitor 11 Correction coil 20 Signal amplifying means 21 Phase synthesis operation means 22 Amplification factor adjusting means 23 Capacitance measuring means 24 Switching means 28 Static Oscillation source for capacitance measurement 30 PLL (Phase Locked Loop) control circuit 31 Oscillation source 32 Power amplification circuit 33 Voltage detection means 34 Current detection means 35 Phase correction circuit 36 Coil 37 Insulation transformer 40 Comparison circuit

Claims (5)

[Claims] 1. An ultrasonic vibrator driving apparatus for driving an ultrasonic vibrator at its resonance frequency, comprising: a. An oscillation source that outputs a high-frequency drive voltage signal; b. A power amplifier circuit for power amplifying an output signal from the oscillation source; c. A correction capacitor connected in series to the ultrasonic transducer; d. Signal amplification means for adjusting the magnitude of at least one of the voltage signals applied to the ultrasonic transducer or the correction capacitor; e. Calculating means for phase-synthesizing the output signal of the signal amplifying means and the voltage signal applied to the ultrasonic transducer or the correcting capacitor; f. A PLL control circuit for controlling the oscillation source to drive the ultrasonic transducer at the resonance frequency by using the output signal of the oscillation source or the voltage signal applied to the ultrasonic transducer and the output signal of the calculating means; g. Capacitance measuring means for measuring capacitance when the ultrasonic vibrator is not driven; h. A driving apparatus for an ultrasonic vibrator, comprising: a gain adjusting means for adjusting an amplification factor of a signal amplifying means in accordance with a capacitance of the ultrasonic vibrator. 2. An ultrasonic transducer driving apparatus for driving an ultrasonic transducer at its resonance frequency, comprising: a. An oscillation source that outputs a high-frequency drive voltage signal; b. A power amplifier circuit for power amplifying an output signal from the oscillation source; c. A correction capacitor connected in a bridge shape with the ultrasonic vibrator on one side; d. Signal amplification means for adjusting the magnitude of at least one of the voltage signals applied to the ultrasonic transducer or the correction capacitor; e. Calculating means for phase-synthesizing the output signal of the signal amplifying means and the voltage signal applied to the ultrasonic transducer or the correcting capacitor; f. A PLL control circuit for controlling the oscillation source to drive the ultrasonic transducer at the resonance frequency by using the output signal of the oscillation source or the voltage signal applied to the ultrasonic transducer and the output signal of the calculating means; g. Capacitance measuring means for measuring capacitance when the ultrasonic vibrator is not driven; h. A driving apparatus for an ultrasonic vibrator, comprising: a gain adjusting means for adjusting an amplification factor of a signal amplifying means in accordance with a capacitance of the ultrasonic vibrator. 3. A phase correction circuit for correcting the phase of the output signal of the oscillation source so that the output signal of the oscillation source has substantially the same phase as the voltage signal applied to the ultrasonic transducer.
3. A driving device for an ultrasonic transducer according to any one of claims 1 to 2. 4. A coil which is connected in series to the ultrasonic vibrator and has an inductance which resonates with the combined capacitance of the ultrasonic vibrator and the correction capacitor at the resonance frequency of the ultrasonic vibrator. 4. The driving device for the ultrasonic vibrator according to claim 3. 5. A switch device for driving an ultrasonic vibrator, and a switch device for connecting a capacitance measuring device to the ultrasonic vibrator, wherein the operations of both switch devices are linked. A driving device for an ultrasonic transducer according to any one of claims 1 to 4.