ITTO940159A1 - CONTROL CIRCUIT FOR A BRUSHLESS SYNCHRONOUS ELECTRIC MOTOR - Google Patents

Abstract

The control circuit comprises a TRIAC (T) in series with the stator winding (1) of the motor, between the terminals of an alternating voltage source (Vm). A fixed sensor (H) is associated with the rotor (R) to provide a signal indicative of the magnetic polarity of the portion of the rotor (R) facing it. A diode circuit (DC), comprising at least one zener diode, supplies the sensor (H) and supplies a signal indicative of the polarity of the alternating supply voltage (Vm). A pilot circuit (PC) is connected to the sensor (H) and to the diode circuit (DC) and drives the command input (g) of the TRIAC (T) according to preset modes, according to the instantaneous position of the rotor (R ) and the polarity of the supply voltage (Vm). (Figure 2).

Description

DESCRIPTION of the industrial invention entitled: "Control circuit for a synchronous electric motor - brushless type"

DESCRIPTION

The present invention relates to an electric circuit for a brushless synchronous electric motor including a permanent magnet motor rotatable between polar stator terminations shaped so as to define, with respect to the rotor, an air gap of uneven amplitude, so that the rotor at rest is it has a predetermined angular position, and at start-up it has a preferential direction of rotation.

a first object of the invention is to provide a circuit suitable for controlling the supply of current to the stator winding of such an electric motor, both in the starting phase, in which the motor is driven asynchronously, and in the following phase of steady rotation, in which the motor properly exhibits a synchronous behavior, that is, it rotates at a speed rigidly linked to the frequency of the alternating supply voltage. typically the mains voltage.

This object is achieved according to the invention by means of a control circuit characterized in that it comprises in combination:

a bidirectional static switch in series with the stator winding of the motor, between the terminals of an alternating voltage source;

signal generating circuit means adapted to supply a signal indicative of the polarity of said alternating voltage;

a fixed electrical sensor, associated with the rotor, for providing an electrical signal indicative of the polarity of the portion of the rotor facing it; And

a driving circuit connected to the sensor and to said signal generating circuit means, and arranged to supply a conduction control signal to a control terminal of said bidirectional switch according to predetermined methods, according to the position of the rotor and the polarity of the voltage Power supply.

Brushless synchronous electric motors of the type defined above are used in particular in household appliances, for example for controlling pumps in washing machines. For such applications it is convenient that the control circuit associated with the motor can be physically mounted in the same casing as the motor, or the pump with which the motor is associated. For this purpose it is therefore necessary that the control circuit has contained dimensions and weight, and a minimum thermal dissipation. It is also necessary that the control circuit be optimized, in order to reduce the number of components, and therefore the overall dimensions, and also the cost.

These objectives are achieved according to the invention by means of a control circuit of the type specified above, which according to a further aspect of the invention is characterized in that the aforementioned power supply means and the signal generating circuit means are constituted by a single diode circuit. coupled to said source of alternating voltage, including at least one zener diode and adapted to supply a voltage with a non-zero average value for supplying the sensor and a voltage signal in phase with the voltage of said source and indicative of its polarity.

Further characteristics and advantages of the invention will appear from the detailed description which follows purely by way of non-limiting example with reference to the attached drawings, in which: Figure 1 is a schematic illustration of a brushless synchronous electric motor with two-pole stator and with permanent magnet rotor;

Figure 2 is the circuit diagram of a first embodiment of a control circuit according to the invention,

Figure 3 is a series of graphs which show, as a function of the time t reported on the abscissa, the trend of some signals developed in operation in the circuit of Figure 2;

Figure 4 is a diagram showing an embodiment variant of the circuit shown in Figure 2;

Figure 5 is a partial diagram of a further variant of construction of a control circuit according to the invention;

figure 6 shows the diagram of a further control circuit according to the invention;

Figure 7 is a series of graphs which show, as a function of the time t reported on the abscissa, the trend of some signals developed during operation in the control circuit of Figure 6; figure 8 is the diagram of a further control circuit according to the invention;

Figure 9 is a series of graphs showing the trend, as a function of the time t reported on the abscissa, of some signals developed during operation in the circuit of Figure 8;

figure 10 is the diagram of a further control circuit according to the invention; And

Figure 11 is a series of graphs which show, as a function of the time reported in the abscissa, the trend of some signals developed during operation in the circuit shown in Figure 10.

Figure 1 schematically illustrates a synchronous electric motor of the brushless type including a stator S and a rotor R. In the embodiment illustrated by way of example, the stator comprises a winding 1, with two terminals indicated by 2 and 3, wound on a pack of laminations 4 substantially U-shaped. The ends of this pack of laminations form two pole ends 5 and 6 between which the rotor R is rotatably mounted. In the embodiment illustrated by way of example in Figure 1, the air gaps 7 have progressively decreasing amplitudes, proceeding clockwise around the axis of the rotor R. It follows that at rest the rotor R is arranged in a predetermined angular position, in which the axis of its north N and south S poles, indicated by A, forms an angle a with respect to the axis B of the pole pieces 5 and 6.

By virtue of the uneven amplitude of the air gaps, the rotor R has a preferential direction of rotation when starting. Thus, by making an alternating current flow in winding 1, if the first half wave (for example the positive one) determines the appearance of a north pole at polar expansion 5, the rotor R starts clockwise, for those who observe the figure 1.

A position sensor H is associated with the rotor R, for example a Hall effect sensor, arranged in a fixed position, for example at an angle β with respect to the rest position of axis A. This sensor provides during operation a signal indicative of the polarity. of the portion of the rotor R which is instantly facing it.

For piloting in the starting phase of the rotation, and therefore subsequently in the steady-state rotation of the engine, according to the invention, for example, the control circuit is used which will now be described with reference to Figure 2.

In figure 2 the stator winding 1 of the motor and its terminals have been again indicated with the references 1 to 3. This winding is arranged in series with a bidirectional static switch T, between the terminals M and N of a voltage source alternating power supply, such as for example the electrical distribution network. The alternating voltage applied between these terminals is indicated with VM.

The switch T is for example a TRIAC, preferably of the sensitive type. A protection varistor VR is connected in parallel to this TRIAC.

Terminal N is also connected to GND ground.

A biasing resistor R1 is interposed between the gate g of the TRIAC T and the ground.

DC indicates a diode circuit which in the embodiment according to Figure 2 comprises two zener diodes Z1 and Z2 having the cathode and, respectively, the anode connected to the terminal M through respective resistors R2 and R3. The anode of Z1 and the cathode of Z2 are connected to ground GND.

The voltages which in operation develop at the ends of the diodes Z1 and Z2 in Figures 2 and 3 have been indicated with V1 and V2.

The sensor H has its power supply terminals connected to the cathode of z1 and, respectively, to the anode of Z2.

As shown by the graphs of Figure 3, in operation at each positive half-wave of the alternating voltage VM the zener diode Z1 is inversely polarized, while the zener diode Z2 is directly polarized. Consequently, for each positive half-wave of VM the voltage V1 on Z1 is equal to the zener voltage (for example 12V), while the voltage V2 on Z2 is equal to the voltage drop in direct conduction of the diode Z2 (of the order of about IV or lower).

At each negative half wave of VM the diode Z1 is directly polarized, while the diode Z2 is inversely polarized.

It follows that the trend of the voltages V1 and V2 is substantially as shown in figure 3: each of these voltages is isofrequential and in phase with the mains voltage VM and furthermore the amplitude or level of each of these voltages is indicative of the polarity of VM.

The voltage between the power supply terminals of the sensor H, indicated by VH in figure 2 (in which it is not referred to ground GND, but to the potential of the anode of Z2) corresponds to the difference between Vi and V2, and therefore assumes the trend shown in figure 3. As can be seen from this figure, the voltage vH applied between the power supply terminals of the sensor H is a voltage with a non-zero mean value.

The output of the sensor H is indicated with h, and the voltage between this output and the anode of the zener diode Z2 is indicated with vh. The signal Vh is a logic type signal, which assumes for example a "high" level when the sensor faces a north pole of the rotor R and a "low" level when a south pole faces the sensor.

The gate g of the TRIAC T is connected to a driving circuit indicated as a whole by PC.

In the embodiment shown in Figure 2, the driving circuit PC comprises a pair of diodes DI, D2, connected in antiparallel between the output h of the sensor H and the gate g of the TRIAC T. A resistor R4 is connected between the anode of DI and the cathode of Z1. A resistor R5 is connected in series with diode D2.

The current flowing in operation in the motor has been indicated with IM in Figures 2 and 3. This current is typically out of phase with a delay with respect to the alternating supply voltage VM, since the motor represents an impedance with inductive reactance. The current flowing in the motor is substantially identical to the current flowing in the TRIAC T, so that in the following this current will be indifferently defined as current flowing in the motor or current flowing in said TRIAC.

The control circuit described above with reference to Figure 2 operates in the following manner.

At start-up, that is when the alternating voltage vM is applied between the terminals M and N, the sensor H faces a very precise polarity of the rotor R (for example the north pole as shown in Figure 1). Consequently at the output h of the sensor H there is a signal Vh at the logic level corresponding to this polarity, for example at the "high" level, as shown in Figure 3.

If the first half wave of the alternating voltage VM is positive, as in the graph of Figure 3, during this half wave a current flows through the resistors R2, R4 and the diode DI to the gate g of the TRIAC T. Therefore the TRIAC T in this first half-wave is conductive, and in the motor and in this TRIAC a current iM flows with a delay compared to the half-wave of the voltage VM, as shown in figure 3. In these conditions the TRIAC is driven in the first quadrant (current entering the gate, positive voltage on the TRIAC) - The passage of current in the motor determines the application of a drive torque impulse to the rotor R which starts to start, accelerating depending on the mechanical load associated with it and its inertia.

During the first negative half-wave of VM the current lM passes through zero, and the conduction of current through the TRIAC is interrupted. In fact, if, as in the situation illustrated in the graphs of Figure 3, during this first negative half-wave the sensor H still sees the same polarity of the rotor R that it previously saw, the sensor output h still remains at a "high" level and both diodes D1 and D2 are effectively disabled, so that no conduction trigger command signal is applied to the gate g of the TRIAC T.

Operation continues substantially as described above, a torque pulse being applied to the rotor R at each half-wave of the current IM, and this until the polarity of the rotor R which occurs in front of the sensor H changes. indicated by the instant in Figure 3, the output voltage vh of the sensor H goes to a "low" level. In the graphs of Figure 3 it was assumed that the instant t1 falls within a positive half-wave, subsequently to the beginning of the passage of the current IM in the TRIAC and in the motor. Passing Vh at a "low" level does not interrupt the flow of current through the TRIAC and the motor. Rather, when the next negative half-wave of VM arrives, the output h of H (at low level) determines the reverse bias of D1 and the forward bias of D2. Consequently, a current can flow, exiting the gate g of T, passing through R5, D2, the sensor H and the resistor R3. In this situation, the TRIAC T is driven in the third quadrant (outgoing current, negative voltage) and the current IM after the instant t1 continues as shown in figure 3. In the IM graph of figure 3 the hatched area corresponds to the period d time in which the torque applied to the rotor R due to the effect of the current flowing in the winding 1 is not driving but resistant, i.e. in contrast with the direction of rotation of the rotor R.

After the instant t1 the TRIAC T continues to be piloted periodically in the third quadrant, until the output of the sensor H remains at a low level. Each negative half-wave of the current corresponds to the application of a drive torque impulse to the rotor, which continues its acceleration.

Operation essentially continues in the manner described above until, following the progressive acceleration, the rotor R moves into synchronism, that is, until the output of the sensor h becomes isofrequential and in phase with the current flowing in the motor.

Once synchronism is reached, the rotor rotation speed R stabilizes at a value strictly linked to the frequency of the supply voltage VH, as well as to the number of motor poles. In the case of a synchronous motor with two stator poles and two rotor poles supplied with mains voltage at a frequency of 50 Hz, the synchronism speed corresponds to 3000 rpm.

As the previous description allows to appreciate, the driving circuit PC supplies to the gate g of the TRIAC T a control signal which is a function of the output level of the sensor H and therefore of the angular position of the rotor R, as well as of the polarity of the voltage. VM power supply. In the control circuit according to Figure 2, the diode circuit DC performs at the same time the function of generating the supply voltage necessary for the sensor H, and of supplying the information about the polarity of the supply voltage VM to the driving circuit PC. This functional synergism allows to advantageously reduce the number of components and therefore the overall dimensions of the entire control circuit.

Hall effect sensors currently available typically have an "open collector" output. The driver circuit PC shown in Figure 2 takes this circumstance into account.

In the circuit according to Figure 2, the currents flowing in the resistors R2 and R3 are generally different from each other, the current flowing in R3 being in particular approximately double that flowing in R2.

Wanting to reduce the dissipation in the resistors R2 and R3, so that the current flowing in operation in R3 becomes substantially equal to that flowing in R2, it is possible to adopt a driver circuit PC having the structure shown in figure 4. In this figure a parts and components already described have been assigned the same reference numbers again. The driver circuit PC of Figure 4 comprises a transistor Q0 of the npn type, with the base connected to the output h of the sensor H, the emitter connected to the gate g of the TRIAC T through a resistor R6, and the collector connected to the cathode of Z1. A diode D2 has the anode and cathode respectively connected to the emitter and base of Q0. This diode is connected in antiparallel to the base-emitter junction of Q0.

A resistor R'4 is connected between the base and the collector of Q0 which conveniently has a much higher resistance than that of the resistor R4 of Figure 2. In particular, the resistance of R '4 is approximately β times the resistance of R4, β being the gain of Q0.

The operation of the driver circuit PC of Figure 4 is substantially similar to that of the corresponding circuit shown in Figure 2: the function of the diode Di of Figure 2 is in fact performed by the base-emitter junction of the transistor Q0. Moreover, since the resistance of R'4 is much higher than that of R4, there is a reduction in dissipation and one can tend to equalize the intensity of the currents that flow in the resistors R2 and R3 during operation, with an overall reduction of the dissipation in the circuit. control associated with the engine.

In order to reduce the dissipation, a further measure can consist, for example, in realizing the DC diode circuit as illustrated in Figure 5, in which the circuit branches comprising R2 and Z1 and, respectively, R3 and Z2, are coupled to the source of alternating voltage VM through a transformer TR which reduces the level of the alternating voltage which is applied to these circuit branches. This solution, which can be applied to the circuit according to figure 2 and to that according to figure 4, allows to effectively reduce the thermal dissipation, but implies a higher cost, weight and bulk due to the presence of the transformer TR.

A further solution which allows to reduce the thermal dissipation in the resistors R2 and R3 is illustrated in figure 6, in which the same reference symbols have been attributed again to parts and components already described.

This solution is based on driving the gate g of the TRIAC T in an impulsive rather than continuous way.

For this purpose, the driving circuit PC comprises a capacitor C connected to the output h of the sensor H by means of a resistor R7, and to the junction between the zener diodes z and Z2. A resistor R8 is connected between the output of the sensor H and the cathode of Z1. In parallel to the capacitor C is connected a resistor R9 which has a significantly higher resistance than R7 and R8.

In operation, in the starting phase of the rotation of the motor, when the alternating supply voltage VM has a positive half-wave and the sensor output is at a "high" level, the capacitor C charges very quickly.

Through a diode D3 the capacitor C is connected to a threshold switch formed by two transistors Q1 and Q2, respectively of the pnp and npn type, connected together as shown in Figure 6. In particular, the collector of Q1 and the base of Q2 they are connected to the junction between Z1 and Z2 through a resistor R10. The base of Q1 and the collector of Q2 are connected to the junction between two resistors R11 and R12, in series with the collector-emitter path of a transistor Q3 of the npn type. This transistor has the emitter connected to ground GND, and the base connected through a resistor R13 to the ungrounded terminal of the TRIAC T.

The emitter of Q2 is connected to the gate g of T through a resistor R14.

As previously mentioned, when the voltage VH has a positive half-wave and the signal VH is at a "high" level, the capacitor C charges rapidly and therefore, as soon as the voltage across it is such as to determine the passage into conduction of Q1 and Q2, this capacitor discharges on the gate g of the TRIAC T through the diode D3, the transistors Q1 and Q2, and the resistor R14. As shown in Figure 7, the pulse applied to the gate of the TRIAC τ {indicated with i) is delayed with respect to the beginning of the positive half-wave of VM. This pulse determines the passage in conduction of T, through which a current IM can flow, which crosses the stator winding of the motor.

If during the next negative half-wave of vM the signal Vh remains at a "high" level (as shown in the second half-period of VM in Figure 7), the current conduction in the TRIAC T is extinguished as soon as the current IM is canceled. When the passage of the current in T ceases, the transistor Q3, which was previously blocked, now becomes conductive. The resistors R11 and R12 then form a voltage divider, to which the collector of Q2 is connected. Consequently, the threshold of passage in conduction of Q1 and Q2 is raised. In addition to this raising of the threshold, during the duration of said negative half-wave of VM, while remaining VH at a "high" level, the capacitor C charges at a decidedly low voltage (about IV or lower) and therefore insufficient to determine the passage conducting Q1 and Q2. Therefore, in the negative half-wave of VM the capacitor C does not determine the application of any pulse to the gate g of the TRIAC T.

At the next positive half-wave of VM (as in the third half-period of VM in Figure 7) if vh is still at a "high" level, capacitor C charges again very quickly to a voltage value such as to bring Q1 and Q2 back into conduction, and then this capacitor discharges on the gate g of T, as previously described. The corresponding pulse i applied to the gate of the TRIAC τ triggers the passage of the current IM again. The transistor Q3 remains off until there is a passage of current in the TRIAC T.

The operation of the circuit proceeds as described above until the polarity of the rotor R presented to the sensor H is such as to determine the passage of the signal vh at a "low" level, as shown at the instant t3 in figure 7. This figure refers by way of example, to a situation in which the instant t3 occurs during a positive half-wave of VH.

After the instant t3 the current IM continues to flow in the motor and in the TRIAC T, even after the instant t2 of the beginning of the next negative half-wave of VM, until this current is canceled out at the instant t3.

Starting from the instant t2 there is simultaneously in the presence of a negative half wave of VM and of Vh at a "low" level. Consequently, starting from the instant t2, the capacitor C charges rapidly to a high voltage value of opposite sign with respect to the voltage at which it was charged in correspondence with the positive half-waves of VM.

The capacitor C is connected through a diode D103 to a further threshold switch formed by two transistors Q101 and Q102 with an essentially mirror arrangement with respect to that of the threshold switch formed by the transistors Q1 and Q2. The base of Q102 and the collector of Q101 are connected to ground GND through a resistor R110. The base of Q101 and the collector of Q102 are connected to the junction between two resistors R111 and R112. Resistor R111 is connected to the anode of Z2. Connected in series with R112 is the collector-emitter path of a transistor Q103 which has the emitter connected to ground GND, and the base connected to that of transistor Q3.

The circuit part just described, comprising the diode D103, the transistors Q101, Q102 and Q103 and the resistors R110, R111 and R112 is essentially specular to the circuit part formed by the diode D3, by the transistors Q1, Q2 and Q3 and by the resistors R10, R11 and R12. This last circuit part intervenes in the operation of the entire control circuit when the signal Vh is at a "high" level, as described above, while the underlying circuit part, which mirrors it, intervenes when the signal vh is at a "low" level. ".

In particular, when the signal vh is at a "low" level and the alternating supply voltage VH is in a negative half-wave (as in the fourth half-period of VM in Figure 7), the capacitor C charges rapidly to a high voltage of opposite sign to the one to which it loads when vh is "high" and VM is in positive half-wave. The voltage on the capacitor C increases in particular until the transistors Q101 and Q102 become conductive, thus allowing the discharge of the capacitor C on the gate g of the TRIAC T, which is then made conductive.

The pulse applied to the gate g of T has a delay with respect to the beginning of the negative half-wave of the supply voltage VM, linked to the charging time constant of the capacitor C.

This delay could be such that, immediately after the switching of Vh from the "high" level to the "low" level, the first negative pulse could be applied to the gate of the TRIAC T, while the latter is still conductive, i.e. between the instants t2 and t3 of Figure 7, as indicated by the dashed pulse in that figure. This situation must be avoided, since the negative impulse in question would be completely ineffective and useless, and it would then not be possible to re-trigger the current conduction in the TRIAC T after the instant t3 during the negative half-wave of VM. This situation is avoided with the circuit of Figure 6 thanks to the mechanism for varying the level of the voltage threshold for which the switch Q101-Q102 passes into conduction.This threshold is in fact controlled by the transistor Q103, which is disabled when T is in conduction. Therefore, with reference to Figure 7, between the instants t2 and t3 the transistor Q103 is off, and the voltage on the capacitor C is unable to make the transistors Q101 and Q102 pass into conduction. After the instant t3 the TRIAC no longer conducts current, and the transistor Q103 becomes conductive, lowering the threshold associated with the transistors Q101 and Q102 to a relatively lower voltage value, determined by the voltage divider formed by the resistors R111 and R112. After the instant t3, the voltage on the capacitor C then manages to exceed this lowered threshold, making the transistors Q101 and Q102 pass in conduction, which in turn allow the discharge of the capacitor C on the gate of the TRIAC T in an instant indicated with t4 in the figure 7 in correspondence with which the passage of current through the TRIAC T resumes.

The transistor Q103 and the associated resistors R111 and R112 on the one hand, and the transistor Q3 and the associated resistors R11 and R12 on the other, therefore enable the application to the gate of the TRIAC T of the first pulse following a change in the level of the signal vh only after the passage of current in the TRIAC T.

It should be noted that with the threshold variation mechanism described above, the first pulse applied to the gate of the TRIAC T after a level transition of the signal vh has a greater amplitude than that of the others, since the delay that is imposed on the discharge of the capacitor allows that capacitor to reach a higher voltage level.

In the control circuit according to Figure 6 the thermal dissipation in the resistors! R2 and R3 is practically limited to the charging times of the capacitor C, and therefore to extremely short times.

Finally, the circuit of Figure 6 comprises a pair of capacitors C2, C102, arranged specularly, for the suppression of disturbances.

Figure 8 illustrates a further variant embodiment of the control circuit according to the invention. In this figure, parts and components already described have again been attributed the same reference numbers.

In the diagram according to Figure 8, the DC diode circuit comprises a single zener diode Z0 which has the anode connected to the terminal M through a resistor R20 and the cathode connected to ground GND. The power supply terminals of the sensor H are connected to the cathode of the zener diode Z0, and to the anode of the latter through a diode D5. A capacitor CO is connected essentially in parallel with the zener diode Z0, to smooth the voltage that is applied between the power supply terminals of the sensor H.

As shown in figure 9, the voltage VO which in operation is located at the ends of the zener diode ZO is isofrequential and in phase with the alternating power supply voltage VM.The voltage VO has a non-zero average value, therefore it allows to power the sensor H and furthermore its level is indicative of the polarity of the VM voltage.

The anode of the zener diode ZO is connected to the pc input of the driver circuit PC. This circuit comprises a transistor Q5 of the npn type, which has the collector connected to the gate g of the TRIAC T through a resistor R21, and the emitter connected to the anode of the diode D5. The emitter of this transistor is also connected to its base through a resistor R22.

The pc input of the driver circuit PC is connected to the base of the transistor Q5 through a resistor R23 and two diodes D6 and D7, connected as shown in figure 8.

A diode D8 has the anode connected to the resistor R23 and the cathode connected to the output of the sensor H.

The input pc of the driver circuit PC is connected to the base of a further transistor Q6 by means of a resistor R24. The transistor Q6 is of the pnp type, and has the collector connected to the anode of D7 by means of a resistor R25, and the emitter connected to the collector of a transistor Q7, also of the pnp type. This last transistor has the emitter connected to the cathode of the zener diode ZO and the base connected to the junction of two resistors R26 and R27 connected as a voltage divider between the cathode of the zener diode ZO and the output h of the sensor

H.

The circuit according to the scheme of Figure 8 functions in the following way.

Suppose that the first half-wave of the alternating supply voltage VM is positive, and the signal-Vh at the output of the sensor H is at a "high" level, as shown in the graphs of figure 9. In these conditions the voltage VO on the zener diode ZO, relative to the ground GND, is equal to the small voltage that is localized on this diode which is directly biased. The transistors Q6 and Q7 are off, since the output of the sensor H is at a "high" level. The transistor Q5 is instead driven in conduction due to the effect of the current that reaches it in the base through R23, D6 and D7. The collector P of this transistor therefore moves to a potential substantially corresponding to that of its emitter, determining the triggering of the TRIAC T. The voltage between the collector P of Q5 and the ground GND is indicated by the wave form VP in figure 9 Following the triggering of the TRIAC T, a current IM circulates in the motor, delayed with respect to the positive half-wave of vM, as shown in Figure 9.

At the first negative half wave of vM the voltage VO on the zener diode ZO becomes negative, and substantially equal to the zener voltage of this diode. The current IH continues to flow in the motor until it is canceled, after which it is extinguished, as the TRIAC is cut off. If, as shown in the illustrative graphs of Figure 9, the voltage vh at the output of the sensor H remains at a "high" level, the <'> transistors Q6 and Q7 remain in the interdiction state. The transistor Q5 is also off, and the TRIAC T is not brought into conduction during the negative half-wave of VM.

Operation continues as described above until, following the progressive rotation of the motor rotor, the signal Vh at the output of the sensor H passes to a "low" level as shown at instant tl in figure 9. In this figure it has been assumed that said instant occurs while the alternating voltage VM has a positive half-wave.

The "low" level transition of Vh has the effect of biasing the transistor Q7. the transistor Q6 goes into conduction as soon as the next negative half-wave of VM begins, and then a current reaches the base of Q5 through the transistors Q7, Q6, the resistor R25 and the diode D7. The transistor Q5 is kept conductive for the entire negative half-wave of VM, so that the current IH continues to flow in the TRIAC and in the motor even after the zero crossing, becoming negative, as shown in the lower graph of figure 9. In this figure the hatched area in the graph of the current IM corresponds to the period of time in the current lM determines the application to the rotor of a resisting torque.

Subsequently, as long as Va remains at a "low" level, there is a passage of negative half-waves of current in the motor, delayed with respect to the negative half-waves of VM. When the Vh signal returns to the "high" level, positive half-waves of current pass through the motor, out of phase with respect to the positive half-waves of VM.

Operation then continues as described above, until, due to the subsequent accelerations, the rotor of the motor moves into synchronism with the frequency of the mains voltage vM.

In synchronism conditions the signal Vh is perfectly isofrequential and in phase with the current flowing in the motor.

In the variant embodiment shown in Figure 10, the DC diode circuit comprises two zener diodes Z3 and Z4. The anode of the diode Z3 is connected to the input pc of the driving circuit PC, which has a structure corresponding to that described above with reference to Figure 8. The anode of Z3 is also connected to the terminal M through a resistor R30.

The zener diode Z4 has the cathode connected to the ground GND, and the anode connected to the terminal M through a capacitor C5 and a resistor R31 connected together in series. The anode of Z4 is also connected to the cathode of diode D5.

In the circuit according to Figure 10, the power supply of the sensor H is ensured through the resistor R31, the capacitor C5 and the zener diode Z4. The resistor R31 in particular protects the zener diode Z4 (and the diode D5) from current peaks. The voltage that is localized on the zener diode Z4, smoothed by the capacitor CO thus ensures the power supply to the sensor H. It is however noted that due to the presence of the capacitor C5 the potential of the anode of Z4 with respect to the ground GND is out of phase in advance with respect to the alternating supply voltage vM.

On the other hand, the potential of the anode of the zener diode Z3 with respect to the ground GND is instead exactly in phase with the voltage vM, and can therefore drive the input pc of the driver circuit PC providing to this input an isofrequency signal and in phase with VM and indicative of the polarity of VM. This signal has been indicated with V3 in Figures 10 and 11.

Apart from the structural differences highlighted above, the circuit of Figure 10 substantially corresponds to that of Figure 8 and its operation is completely analogous to that of this circuit.

Figure 11 shows exemplary trends of VM, V3, vh, of the gate current lg of the TRIAC T and of the current IM flowing in the motor and in the TRIAC in the operation of the circuit of figure 10. These graphs are self-explanatory, and do not require therefore of further comments.

The circuit according to Figure 10 has, compared to that of Figure 8, the advantage of a lower thermal dissipation. It is noted that the resistor R31, intended to protect the zener diode Z4 from current peaks, could also consist of a PCT resistor.

Naturally, the principle of the invention remaining the same, the embodiments and details of construction may be varied widely with respect to what has been described and illustrated purely by way of non-limiting example, without thereby departing from the scope of the present invention.

Claims (16)

CLAIMS 1. Control circuit for a brushless synchronous electric motor including a permanent magnet rotor (R) rotatable between stator pole ends (5, 6) shaped so as to define with respect to the rotor (R) an air gap (7) of uneven amplitude so that the rotor (R) at rest is arranged in a predetermined angular position (a) and at start-up has a preferential direction of rotation; the control circuit being characterized in that it comprises in combination a bidirectional static switch (T) in series with the stator winding (1), between the terminals (M, N) of an alternating voltage source (VM); signal generating circuit means (Z1, Z2; ZO; Z3) adapted to supply a signal indicative of the polarity of said alternating voltage (vM); a fixed electrical sensor (H), associated with the rotor (R), to provide a signal (Vh) indicative of the polarity of the portion of the rotor (R) facing it; And a driving circuit (PC) connected to the sensor (H) and to said signal generating circuit means (Z1, Z2; ZO; Z3), and arranged to supply a conduction control signal to a control terminal (g) of said switch (T) according to predetermined modes, according to the position of the rotor (R) and the polarity of the alternating supply voltage (VM). 2. Control circuit according to claim 1, characterized in that said power supply means and said signal generating circuit means consist of a single diode circuit (DC) coupled to said source of alternating voltage (VM), including at least one diode zener (Z1, Z2; ZO; Z3, Z4) and able to supply a voltage (VH; VO) for the power supply of the sensor (H), and a voltage signal (vi, V2; VO; V3) in phase with the voltage of said alternating voltage source (VM) and indicative of its polarity. 3. Control circuit according to claim 2, characterized in that said diode circuit (DC) comprises: a first zener diode (Z1) having the cathode coupled through a first resistor (R2) to a first terminal (M) of the alternating voltage source (vM) to which the motor (1) is connected, and the anode connected to the second terminal (N) of said source to which the bidirectional switch (T) is connected; and a second zener diode (Z2) having the cathode connected to the anode of the first zener diode (Z1) and the anode coupled to said first terminal (M) of the source of alternating voltage (vM) through a second resistor (R3); the power supply terminals of said sensor (H) being connected to the cathode of the first zener diode (Z1) and to the anode of the second zener diode (Z2). 4. Control circuit according to claim 3, characterized in that said first and second resistor (R2, R3) are coupled to the alternating voltage source (VM) by means of a voltage reducing transformer (TR). 5. Control circuit according to claim 3 or 4, characterized in that the sensor is a Hall effect sensor (H) and the driving circuit (PC) comprises a pair of diodes (Di, D2) connected in antiparallel between the 'sensor output (h) and the control input (g) of the bidirectional switch (T); the output (h) of the sensor (H) being also connected to the cathode of the first zener diode (zi), 6. Control circuit according to claim 3 or 4, characterized in that the sensor is a Hall effect sensor (H) and the driving circuit (PC) comprises a transistor (QO) having the base connected to the output (h ) of the sensor (H) and the collector-emitter path connected between the cathode of said first zener diode (21) and the control input (g) of the bidirectional switch (T), and a diode (D2) having the anode and cathode connected between the emitter and the base of said transistor (QO), in antiparallel with respect to the base-emitter junction of said transistor (QO). 7. Control circuit according to claim 6, characterized in that a third resistor (R'4) is interposed between the base and the collector of said transistor (QO). 8. control circuit according to claim 3 or 4, characterized in that said sensor is a Hall effect sensor (H) and the driving circuit (PC) comprises pulse generating means (C; Q1-Q3, Q101-Q103 ; R11, R12; R111, R112; R14) having the input connected to the output (h) of the sensor (H), and the output connected to the control input (g) of the bidirectional switch (T); said pulse generating means being arranged to emit a positive pulse at each positive half wave of the voltage (VM) between said first and second terminals (M, N) of the alternating voltage source (VM) when the output of the sensor (h) is at a high level, and a negative pulse at each negative half-wave of the voltage (vM) between said first and said second terminals (IVI, N) when the sensor output (H) is at a low level. 9. Control circuit according to claim 8, characterized in that said pulse generating means {C, Q1-Q3, Q101-Q103, R11, R12; R111, R112, R14) are arranged to generate said positive and respectively negative pulses with a predetermined delay with respect to the beginning of each positive and respectively negative half-wave of the alternating voltage (VM) of said source. 10. Control circuit according to claim 8 or 9, characterized in that said pulse generating means comprise: a capacitor (C) connected between the output (h) of the sensor (H) and the junction (GND) of said first and second zener diodes (Z1, Z2); and a first and a second threshold switch circuit (Q1-Q3; R11, R12; Q101-Q103, R111, R112) connected between said capacitor (C) and the control input (g) of said bidirectional switch (T). 11. Control circuit according to any one of claims 8 to 10, characterized in that said pulse generating means {C; Q1-Q3, R11, Ri2; Q101-Q103, R111, R112) comprise enabling circuit means (Q3; Q103) connected to said bidirectional switch (T) and arranged to enable the emission of the first pulse following a level change of the output (h) of the sensor (H) only after the passage of current in said bidirectional switch (T) has ceased. 12. Control circuit according to claims 10 or 11, characterized in that said enabling circuit means (Q3; Q103) are connected to said threshold switches (Q1, Q2; Q101, Q102) and are arranged to determine the raising the threshold of said switches {Q1, Q2; Q101, Q102) when there is current flowing in the bidirectional switch {T). 13. Control circuit according to claim 2, characterized in that said diode circuit (DC) comprises a (first) zener diode (Z0; Z3) having the anode coupled through a first resistor (R20; R30) to a first terminal (M) of the alternating voltage source (VM) to which the motor (1) is connected, and the cathode connected to the second terminal (N) of said source to which the bidirectional switch (T) is connected; the anode of said (first) zener diode (Z0; Z3) being connected to the input of the driving circuit (PC) to provide it in operation with a signal indicative of the polarity of the aforementioned alternating voltage (VM). 14. Control circuit according to claim 13, characterized in that said diode circuit (DC) comprises a single zener diode (Z0). 15. Control circuit according to claim 13, characterized in that said diode circuit (DC) comprises a second zener diode (Z4) having the anode coupled through a second resistor (R31) to said first terminal (M) of the source of alternating voltage (VM) and the cathode connected to that of said first zener diode (Z3); the sensor terminals (H) being connected to the cathode and, respectively, to the anode of said second zener diode (Z4). Control circuit according to one of claims 13 to 15, characterized in that the driving circuit (PC) comprises a first transistor (Q5) having its base connected to the anode of said (first) zener diode (Z0; Z3) and the collector-emitter path connected between the control input (g) of the bidirectional switch (T) and the anode of said (first) zener diode (Z0; Z3), and a switching circuit comprises a second and a third transistor (Q7, Q6) having the base connected to the sensor output (H) and, respectively, to the anode of said (first) zener diode (Z0; Z3) and the collector-emitter paths in cascade, between the cathode of said (first) zener diode (Z0; Z3) and the base of said first transistor (Q5). All substantially as described and illustrated and for the specified purposes.