IT202000019711A1 - METALLIZED FOLDED BROADBAND DIELECTRIC WAVEGUIDE FILTERS - Google Patents

Description

Italian patent application entitled:

?HIGH BANDWIDTH FOLDED METALLIC DIELECTRIC WAVEGUIDE FILTERS?

DESCRIPTION

FIELD

The present invention ? relating in general to systems of communications and pi? in particular to filters that are indicated for use in cellular communication systems.

BACKGROUND

Filters are electronic devices that selectively pass signals based on the frequency of the signal. Several different filter types are used in cellular communications systems. Furthermore, as new generations of cellular communications services have been introduced, typically without the elimination of existing cellular communications services, both the number and types of filters used have expanded significantly. Filters can be used for example to allow radio frequency (?RF?) signals in different frequency bands to share selected components of a cellular communications system and/or to separate RF data signals from power signals and/or control. With the number of filters used in a typical cellular communications system proliferating, ? increased the need? of filters more? small, more light and/or less expensive.

Conventionally, cavity filters? resonant metal have been used to implement many of the filters used in cellular communications systems. As shown in figure 1A, in its most complete form simple, a cavity filter? metal resonant 10 pu? consist of a metal housing 12 which has walls 14 formed therein which define a row of cavities from 18-1 to 18-4. Although the exemplary filter 10 shown in FIG. 1A includes a total of four cavities 18, will be appreciated? what can? be expected any appropriate number of cavities? 18 as necessary to provide a filter having desired filtering characteristics. It is pointed out that herein, when multiple equal elements or structures are provided, they may be referred to in some cases using two-part reference numerals, wherein the two parts are separated by a dash. Herein, such elements may be referred to individually by their full reference numeral (e.g. cavities 18-2) and may be referred to collectively by the first part of the applicable reference numeral (e.g. cavities 18).

Referring again to FIG. 1A, a resonant element or ?resonator? coaxial 20-1 to 20-4 pu? be provided in each of the respective cavities? from 18-1 to 18-4. The walls 14 may include openings or ?windows? 16 which allow the resonators 20 in the adjacent ones of the cavities? 18 to couple together along a main coupling path extending from an input 22 to an output 24 of the filter 10. These coupled resonances can form a filter having a passband response without transmit zeros and a moderate fractional bandwidth (eg a bandwidth of up to 10-20% of the center frequency of the passband, depending on the specific geometry and size of the cavities and resonators).

When are higher bandwidths needed? It is possible to reverse the orientation of one coaxial resonator 20 every two. A filter 30 having this configuration? shown in figure 1B. In the filter 30, the electric and magnetic components of the couplings between adjacent resonators 20 are added in phase and therefore ? is it possible to increase the quantity? coupling total. Given that the? Bandwidth of a filter ? proportional to the quantity? total coupling, the filter 30 of figure 1B can? have an increased bandwidth compared to the filter 10 of Figure 1A .

The answer? of a filter refers to the energy that passes from a first orifice (for example, an input orifice) of the filter to a second orifice (for example, an output orifice) of the filter as a function of frequency. A filter response will typically include? one or more pass bands, which are frequency ranges in which the filter passes signals with quantity? relatively small attenuation. A filter response typically also includes one or more? rejection bands. A reject band refers to a frequency range where the filter will essentially not do? pass the signals, usually why? the filter ? designed to reflect back any signals that are incident on the filter in this frequency range. In some applications can? be it desirable that the filter response has a high degree of ?selectivity? local?, meaning that the transition from one passband to an adjacent rejection band occurs over a narrow frequency range. A technique to increase selectivity? local consists in adding transmission zeros in the filter response. One ?transmission zero ? refers to a portion of a filter frequency response in which the amount? of passing signal energy ? very low. The transmission zeros are pi? typically achieved using cross couplings.

The cross coupling, which ? the best technique common used to increase the selectivity? local in a cavity filter? resonant, refers to the intentional coupling between the resonant elements of cavities? not adjacent. Depending on the relative position of the transmit zero to the passband, the sign of the required cross-match may vary. When cross couplings are used to create drive zeros, the cavities? are often arranged in some form of planar grid as opposed to the single row of cavities? included in filters 10 and 30 of figures 1A-1B. This two-dimensional distribution of cavities? facilitates the coupling between cavities? non-adjacent (i.e. cross-mates). US patent No. 5,812,036 (?patent ?036?), the contents of which ? incorporated here by reference, describes various cavity filters? resonant that have such arrangements of cavities? two-dimensional including cross coupling.

Figure 2 of this application ? a sectional view from? top of a cavity filter? resonant two-dimensional 40 that ? described in the ?036 patent. As shown in FIG. 2, the filter 40 includes a total of six cavities. 18-1 to 18-6 which each have a respective coaxial resonator 20-1 to 20-6 disposed therein. There are 16-1 to 16-5 mating windows that allow for ?major? matings. between the adjacent ones of the six coaxial resonators 20-1 to 20-6 along the main transmission path through the filter 40 (i.e., between cavities 18-1 and 18-2, between cavities 18-2 and 18-3, between between cavities 18-3 and 18-4, between cavities 18-4 and 18-5 and between cavities 18-5 and 18-6). Additionally, the filter 40 includes two bypass coupling windows 26-1, 26-2 which permit cross-coupling between two pairs of nonadjacent resonators (i.e., between cavities 18-1 and 18-6 and between cavities 18-6). -2 and 18-5). The principal couplings between the five sequential pairs of resonators 20 and the two cross couplings between the two pairs of non-adjacent resonators 20 contribute to the overall transfer function of the filter 40.

Can cross couplings also be achieved in a cavity filter? in-line resonant (i.e. one-dimensional) by including some form of distributed coupling elements to implement cross-coupling. Figure 3 illustrates a filter 50 which ? implemented using this approach. As shown in figure 3, the filter 50 ? an in-line filter having four cavities? 18-1 to 18-4 which have respective coaxial resonators 20-1 to 20-4 mounted therein. Coupling windows 16 are provided which allow ?main? couplings? between the adjacent ones of the four coaxial resonators 20. ? further provided is a distributed coupling element 60 in the form of a direct ohmic connection between the coaxial resonator 20-1 and the coaxial resonator 20-4. The direct ohmic connection 60 pu? physically and electrically connect resonator 20-1 to resonator 20-4 without physical or electrical connection to any of the intermediate resonators (i.e. resonators 20-2 or 20-3 in this example). The use of the distributed coupling element 60 can? however, have various disadvantages including greater size, complexity? and cost of the filter, susceptibility? to damage, increased losses, and/or reduced out-of-band attenuation.

The cavity filters? resonator having cross-coupling can also be accomplished without the use of a distributed coupling element by providing some form of controlled mixed coupling between adjacent resonators so that spurious (cross-)coupling between non-adjacent resonators can be controlled in some measure. Such an approach? disclosed in U.S. Pat. No.

10,236,550 issued on March 19, 2019 (?the patent ?550?), the entire content of which ? incorporated herein by reference. Figure 4 ? a schematic cross-sectional view of a filter 70 which is one of the filters disclosed in the ?550 patent.

As shown in FIG. 4, the filter 70 includes a metal housing 12 which has a single cavity in the filter. 18 formed inside it. A plurality? of coaxial resonators 20 ? arranged in a row inside the cavity? 18. The top surfaces? 72 and bottom 74 of housing 12 form respective ground planes. A plurality? of adjusting screws 76 ? provided in the surfaces of the top? and bottom 72, 74 of the housing 12 which extend into the cavity? 18. Filter 70 further includes four conductive connectors 84, each of which provides a physical (ohmic) connection between respective adjacent pairs of resonators 20. of the resonators 20 and the absence of shield walls can result in non-negligible couplings between both adjacent and non-adjacent resonators 20. The couplings will include both capacitive couplings and inductive couplings. The quantity? of capacitive and inductive coupling ? a function, among other things, of the distance between the resonators 20. The quantity? of capacitive coupling can? can also be controlled by adjusting the length and/or width of the top of each resonator 20 to generate more? or less capacitive coupling between 20 different resonators. The capacitive coupling between adjacent resonators 20 will determine? negative coupling values. The inductive coupling can? be controlled by changing the distance between the resonators 20 and/or by adjusting the length of the underside of each resonator 20 which connects to the bottom surface 74 of the housing 12. Inductive coupling results in positive coupling between both adjacent and non-adjacent resonators 20 adjacent. Given that the filter 70 ? designed to have non-negligible inductive coupling between non-adjacent resonators 20, a cross-coupling can? be achieved in filter 70 without employing discrete bypass connectors that ohmicly connect nonadjacent resonators 20. The sign of the main pairs can? be positive or negative based on the quantity? of capacitive versus inductive coupling, while the signs of cross-couplings are always positive.

SUMMARY

In accordance with embodiments of the present invention, metallized dielectric waveguide filters are provided which include an upper metallized dielectric waveguide having a plurality of metalized waveguides. of cavities? resonant coils, the upper metallized dielectric waveguide comprising an upper dielectric block having metallized outer walls, and a lower metallized dielectric waveguide having a plurality of of cavities? resonant sensors, the lower metallized dielectric waveguide comprising a lower dielectric block having metallized outer walls. A first of the cavities? higher resonants ? operatively connected to a first of the cavities? lower resonants through at least one coupling window. A first fissure having metallized walls ? provided in a portion of the upper dielectric block which is part of the first of the cavities? higher resonants.

In some embodiments, the first slit may extend to one end? distal of the first of the cavities? higher resonants. In some embodiments, the first slit may extending longitudinally along a longitudinal axis of the upper metallized dielectric waveguide.

In some embodiments, the upper metallized dielectric waveguide is mounted on the lower metallized dielectric waveguide to form a folded filter having a generally U-shaped main transmission path. include a first cross coupling between a second of the cavities? upper resonants and a second of the cavities? lower resonants.

In some embodiments, the metallized dielectric waveguide filter can further include a cross coupling window which includes a non-metallized portion of a bottom surface of the second of the cavities? upper resonant and a non-metallized portion of a top surface? of the second of the cavities? lower resonant that at least partially overlaps the non-metallized portion of the bottom surface of the second of the cavities? higher resonants. The metallized dielectric waveguide filter can also include a second cross coupling between a third of the cavities? upper resonants and a third of the cavities? lower resonant, where the second cross-coupling passes through the cross-coupling window. The first cross mate can? have a negative sign and the second cross coupling pu? have a positive sign. The cross-coupling window can be placed in the center of a region in which the bottom surface of the second of the cavities? higher resonants ? in contact with the top surface? of the second of the cavities? lower resonants.

In some embodiments, a portion of the upper dielectric block which is part of the second of the cavities? higher resonants pu? include a hole having metallized walls.

The metallized dielectric waveguide filter can also include a first orifice of inlet/outlet that ? connected to a third of the cavities? resonant higher and a second orifice of input / output that ? connected to a third of the cavities? lower resonants. The first of the cavities? higher resonants pu? be the cavity? resonant of the superior metallized dielectric waveguide that ? the "P? far from the third of the cavities? higher resonants.

In some embodiments, the mating window can include a non-metallized portion of a bottom surface of the first of the cavities? upper resonant and a non-metallized portion of a top surface? of the first of the cavities? lower resonant that at least partially overlaps the non-metallized portion of the bottom surface of the first of the cavities? higher resonants.

In some embodiments, the top metallized dielectric waveguide can also include metallized vias that extend through the upper dielectric block to define the cavities? upper resonant, and the lower metallized dielectric waveguide also can? include metallized vias that extend through the lower dielectric block to define the cavities? lower resonants.

In some embodiments, a second slot having metallized walls may be formed in a portion of the lower dielectric block which is part of the first of the cavities? lower resonants.

In some embodiments, the upper metallized dielectric waveguide and the lower metallized dielectric waveguide may be identical.

In some embodiments, the at least one coupling window may include a first coupling window extending generally transversely at an interface between the first of the cavities? upper resonants and the first of the cavities? lower resonant cavities, and second and third coupling windows extending generally longitudinally at or near opposite lateral edges of the interface between the first of the cavities? upper resonants and the first of the cavities? lower resonants.

In some embodiments, the at least one coupling window may comprise a first mating window extending generally transversely across a bottom surface of the first of the recesses? higher resonants at or near one end? distal of the first of the cavities? higher resonants.

In some embodiments, the metallized dielectric waveguide filter can have less than 15 dB return loss throughout the 3.4-3.8 GHz band and have at least 50 dB out-of-band rejection in the 4.4-5.0 GHz band.

In some embodiments, the metallized dielectric waveguide filter can be configured so that radio frequency (RF) signals propagate through the metallized dielectric waveguide filter in the TE1,0,1 mode.

In some embodiments, a ratio of a top metallized dielectric waveguide height to a top metallized dielectric waveguide width can be between 0.42 and 0.48.

In accordance with further embodiments of the present invention, metallized dielectric waveguide filters are provided which include an upper metallized dielectric waveguide having a first cavity in a metalized waveguide. superior resonant, a? last cavity? upper resonant, and at least one cavity? upper intermediate resonant that ? positioned between the first cavity? upper resonant and the last cavity? resonant cavity, the upper metallized dielectric waveguide comprising an upper dielectric block having metallized outer walls and a lower metallized dielectric waveguide having a first resonant cavity. lower resonant, a? last cavity? lower resonant and at least one cavity? lower intermediate resonant that ? positioned between the first cavity? lower resonant and the last cavity? lower resonant, the lower metallized dielectric waveguide comprising a lower dielectric block having metallized outer relatives. The last cavity? superior resonant ? operationally connected to the first cavity? lower resonant through a first coupling window, a second coupling window and a third coupling window.

BRIEF DESCRIPTION OF THE DRAWINGS

Figure 1A ? a schematic side sectional view of a cavity filter? conventional line resonant.

Figure 1B ? a schematic side sectional view of another cavity filter? conventional in-line resonator in which every other resonator is flipped.

Figure 2 ? a schematic top sectional view of a cavity filter? conventional resonant that has a cross coupling between cavities? selected.

Figure 3 ? a schematic side sectional view of a cavity filter? conventional line resonant having an external cross-coupling element.

Figure 4 ? a schematic side sectional view of a cavity filter? conventional in-line resonant having a filter response with transmit zeros.

Figs. 5A and 5B are a perspective view and an exploded perspective view, respectively, of a metallized dielectric waveguide filter according to embodiments of the present invention.

Figure 5C ? a shaded perspective view of the metallized dielectric waveguide filter of Figs. 5A-5B .

Figure 6 ? a schematic diagram illustrating the dimensions of a cavity? rectangular resonant element of a metallized dielectric waveguide filter.

Figure 7A ? is a schematic side view of the metallized dielectric waveguide filter of FIGS. 5A-5C illustrating the different fits between its cavities. resonant.

Figure 7B ? a table showing the sign and direction of couplings included in the metallized dielectric waveguide filter of Figs. 5A-5C .

Figure 8A ? a top view of two cavities? adjacent resonant cells of the metallized dielectric waveguide filter of FIGS. 5A-5C illustrating a coupling window defining the two cavities? resonant.

Figure 8B ? a cross-sectional view illustrating the magnetic fields generated in the two cavities? resonators of figure 8A.

Figure 9A ? a graph that illustrates how? the positive horizontal coupling increases as a function of the width of the coupling window that connects two cavities? adjacent resonants.

Figure 9B ? a graph that illustrates the variation of the central frequency of the first mode of spurious transmission for the two cavities? resonant frequencies shown in Figure 9A as a function of coupling window width.

Figure 10 ? a horizontal cross-sectional view of the fourth cavity? resonant of the metallized dielectric waveguide filter of figures 5A-5C taken near the bottom of the cavity? resonant that illustrates the? intensity? and the direction of the magnetic field.

Figure 11A ? a graph that shows how the? coupling between the fourth and the fifth cavity? resonant of the filter of figures 5A-5C varies as a function of the width of the coupling windows between them.

Figure 11B ? a graph that illustrates the variation of the central frequency of the first mode of spurious transmission for the two cavities? shown in figure 11A as a function of the width of the coupling windows.

Figure 12 ? a graph illustrating the variation of the central frequency of the first spurious mode as a function of the ratio between height and width for the cavities? filter resonators of Figures 5A-5C.

Figure 13A ? a shaded perspective view of a cavity resonator that includes a blind hole in its upper portion.

Figure 13B ? a graph that illustrates the variation of the central frequency of the first spurious mode as a function of the depth? of a blind hole in the cavity? resonant of figure 13A.

Figure 14A ? a shaded perspective view of a cavity resonant that includes a slot in its upper portion.

Figure 14B ? a graph that illustrates the variation of the central frequency of the first spurious mode as a function of the depth? of a crack in the cavity? resonant of figure 14A.

Figure 15A ? a horizontal cross-sectional view of the third cavity? resonant of the metallized dielectric waveguide filter of figures 5A-5C taken near the bottom of the cavity? resonant that illustrates the? intensity? and the direction of the magnetic fields as well as? the size and location of a circular cross-coupling window that ? formed by omitting the metallization in the center of the bottom surface of the cavity? resonant.

Figure 15B ? a graph that shows how the? cross coupling between the third and the sixth cavity? resonant filter of Figures 5A-5C varies as a function of the radius of the cross-coupling window of Figure 15A .

Figure 15C ? a graph illustrating the variation of the center frequency of the first spurious transmission mode as a function of the radius of the cross-coupling window shown in Fig. 15A .

Figure 16A ? a graph that shows how the? cross coupling between the third and the sixth cavity? resonant of the filter of the figures 5A-5C varies according to the ratio between the height and the width of the cavities? resonant. Figure 16B ? a graph that illustrates the variation of the central frequency of the first spurious transmission mode as a function of the ratio between the height and the width of the cavities? resonant.

Figure 17A ? a shaded perspective view of a cavity that includes both a blind hole and a cross-coupling window.

Figure 17B ? a graph that shows how the? cross coupling between the third and the sixth cavity? resonant of the filter of the figures 5A-5C varies according to the depth? of the blind holes formed in the third and sixth cavities? resonant.

Figure 17C ? a graph that illustrates the variation of the central frequency of the first spurious transmission mode as a function of the depth? of blind hole shown in Figure 17B.

Figure 18 ? a perspective view of a filter according to certain embodiments of the present invention.

Figure 19A ? a graph of the simulated return loss for the filter in figure 18.

Figure 19B ? a graph of the simulated insertion loss for the filter in figure 18.

Figure 19C ? an insertion loss graph for a higher frequency range showing the simulated out-of-band rejection performance of the filter of figure 18.

Figure 20 ? a perspective view of a filter according to further embodiments of the present invention.

Figure 21A ? a graph of the simulated return loss for the filter in figure 20.

Figure 21B ? a graph of the simulated insertion loss for the filter in figure 20.

Figure 21C ? an insertion loss graph for a higher frequency range showing the simulated out-of-band rejection performance of the filter of figure 20.

DETAILED DESCRIPTION

A waveguide? a metal duct that can? be used to confine and route RF signals. A waveguide filter? a filter that ? formed using waveguide components. A main transmission path ? defined between an input and an output of a waveguide filter. The waveguide filter can? include a plurality? of cavities? resonants that form the main transmission path, in which each cavity? resonant ? implemented as a short length of guide? wave that ? locked at both ends? Openings that are referred to as ?mating windows? are provided at the ends? ?blocked? of each section of guide? wave so that a certain portion of the electromagnetic wave is made to pass out of the cavity? resonant and in a cavity? adjacent resonant along the main transmission path (or completely outside the filter). Un? electromagnetic wave that ? trapped inside a cavity? resonant ? reflected back and forth between its two ends, and will resonate? at a characteristic frequency based on a given geometry of the cavity? resonant. The resonance effect can be used to selectively pass certain frequencies through the coupling window in the cavity? adjacent resonant. Additional openings that are called ?cross-coupling windows? can be provided between cavities? resonators that are not adjacent to each other along the main transmission path. These cross-couplings can be used to generate transmit zeros in the frequency response of the filter, as explained above. A metallized dielectric waveguide filter? a waveguide filter that ? format using one or more? blocks of dielectric material which have metal provided on their outer surfaces. Metallized dielectric waveguide filters can be thought of as a conventional waveguide filter where the air inside the waveguide is replaced with a solid dielectric material. Metallized dielectric waveguide filters can be formed by metallizing the exterior of one or more filters. dielectric blocks. For example, metal can be plated onto the dielectric block (and into holes and slots in the dielectric block) by a metallization process such as screen printing, spray coating, dip coating or thin film metallization process. Typically, each dielectric block comprises a block of material which has a high dielectric constant such as a ceramic block having a dielectric constant of 10 or more. Suitable ceramic materials include for example barium titanate and zirconium titanate materials. In general terms, major ? the dielectric constant of the dielectric material, pi? filter size can be reduced. Vertically extending metal plated holes (acting as metal struts) or other structures are formed within the dielectric block to form the individual cavities. resonating within the metallized dielectric block. Metallized dielectric waveguide filters can exhibit a very high Q-to-volume ratio, have low insertion losses, and can easily handle 10-20 watts of power without generating unacceptable levels of passive intermodulation products. Therefore, metallized dielectric waveguide filters can be very suitable for many cellular applications. However, metallized dielectric waveguide filters can be relatively heavy, and therefore are generally only used at higher frequencies where the shorter wavelength of RF signals reduces the overall size (and therefore weight) of the filter. An additional limitation associated with metallized dielectric waveguide filters? which tend to have relatively small bandwidths compared to other filter technologies such as cavity filters. coaxial die castings. The bandwidth of a filter ? typically defined as the ratio of the size of the passband (ie, the frequencies at which the filter passes a signal without significant loss) to the center frequency of the passband. Metallized dielectric waveguide filters are available that operate in the 20 GHz frequency band. However, these filters have passbands that are typically only about 100-200 MHz (or less), and therefore the bandwidth of these filters ? only about 1-2%. Many cellular applications require filters having significantly higher bandwidths. large, such as bandwidths exceeding 10% of the center frequency of the bandwidth. As an example, the frequency band of 3400-3800 MHz ? been recently designated for various cellular applications. Filters designed to pass signals anywhere in this operating frequency band therefore require a bandwidth of at least 400 MHz, which is ? over 11% of the 3.6 GHz center frequency of the operating frequency band. Typical requirements for bandwidth are less than 15 dB return loss.

Another problem related to the use of metallized dielectric waveguide filters in cellular systems? that these filters tend to generate unwanted or ?spurious? at frequencies that are close to the passband. Waveguide filters can be designed to transmit an electromagnetic wave in a transverse electric (TE) mode or in a transverse magnetic (TM) mode, such as ? well understood by ordinary skilled in the art. In waveguide transmission systems, including waveguide filters, other unwanted transmission modes can arise which can adversely affect filter response. These unwanted modes are called ?spurious modes?. Spurious ways can lead to the reduction of the quantity? of rejection in a frequency range that ? above the passband frequency range. In many cases, cellular operators may require filters used in base station antennas to have extremely high degrees of rejection at frequencies that are close to the passband. If the spurious modes fall into frequency ranges where ? need this degree of rejection high, pu? be difficult to meet the attenuation specifications.

The requirements for a large passband and high rejection at frequencies close to the passband can be particularly difficult to achieve with respect to metallized dielectric waveguide filters, because the techniques that are used to improve any of these parameters tend to degrade the ?other parameter. For example, to increase the bandwidth of the passband, ? generally necessary to generate a strong coupling between the cavities? resonant, since filter bandwidth generally increases with increased coupling resistance. The typical way to increase the coupling is to increase the size of the coupling windows between the cavities. resonant. Unfortunately, since the coupling windows are enlarged, spurious modes that resonate close to the passband are generally close (in terms of frequency) to the passband. When we? happens, can? be difficult to achieve high levels of attenuation at frequencies close to the passband.

According to embodiments of the present invention, metallized dielectric waveguide filters are provided which have a high bandwidth and which also provide good out-of-band rejection at frequencies close to the passband. These filters can have a very high Q factor, can be low loss, can have high capacitance? of power handling, and can be small and reasonably light. These filters can also be more? cheap and more easier to manufacture than cavity filters? coaxial conventional die castings.

In some embodiments, metallized dielectric waveguide filters are provided which include an upper metallized dielectric waveguide having a plurality of metalized waveguides. of cavities? upper resonant and a lower metallized dielectric waveguide having a plurality of of cavities? lower resonants. The upper and lower metallized dielectric waveguides may comprise respective upper and lower dielectric blocks which each have metallized outer walls. The upper and lower metallized dielectric waveguides may include metallized vias that extend through the respective dielectric blocks to define the cavities. upper and lower resonants. A first of the cavities? higher resonants ? operatively connected to a first of the cavities? lower resonants through at least one coupling window. The pairing window can include a non-metallized portion of a bottom surface of the first of the cavities? upper resonant and an overlapping non-metallic portion of a top surface? of the first of the cavities? lower resonants. A first fissure having metallized walls ? provided in a portion of the upper dielectric block which is part of the first of the cavities? higher resonants. The first fissure can extend to one end? distal of the first of the cavities? superior resonants and pu? extending longitudinally along a longitudinal axis of the upper metallized dielectric waveguide. The first fissure can be in the cavity? superior resonant that ? more away from a filter inlet port.

The superior metallized dielectric waveguide can be mounted on the lower metallized dielectric waveguide to form a folded filter having a generally U-shaped main transmission path. include a cross-coupling window comprising a non-metallized portion of a bottom surface of a second of the cavities? upper resonant and an overlapping non-metallic portion of a top surface? of a second of the cavities? lower resonants. The cross-coupling window can be placed in the center of a region in which the second of the cavities? higher resonants ? batting with the second of the cavities? lower resonants. A first cross coupling between the second of the cavities? upper resonants and the second of the cavities? lower resonants? generated through this mating window. A second cross fit between a third of the cavities? upper resonants and a third of the cavities? lower resonants pu? also be generated through the cross-coupling window. The first cross mate can? have a negative sign and the second cross coupling pu? have a positive sign.

The top and bottom metallized dielectric waveguides may be identical in some embodiments, which can reduce manufacturing costs.

In other embodiments, metallized dielectric waveguide filters are provided which include upper and lower metallized dielectric waveguides. Does the upper dielectric waveguide have a first cavity? superior resonant, a? last cavity? upper resonant, and at least one cavity? upper intermediate resonant that ? positioned between them, and the lower dielectric waveguide similarly has a first cavity? lower resonant, a? last cavity? lower resonant, and at least one cavity? lower intermediate resonant that ? positioned between them. The upper and lower metallized dielectric waveguides each include a dielectric block having metallized outer walls. The last cavity? superior resonant ? operationally connected to the? last cavity? lower resonant through first, second and third coupling windows.

Each of the first through third mating windows can? include a respective non-metallized portion of a bottom surface of the last cavity? upper resonant and a respective non-metallized portion of a top surface? of? the last cavity? lower resonant which at least partially overlaps the respective non-metallized portion of the bottom surface of the last cavity? superior resonant. The first pairing window can? generally extend transversely on an interface between the last cavity? upper resonant and the last cavity? resonant cavity, and the second and third coupling windows may extend generally longitudinally at or near opposite lateral edges of the interface between the last cavity and the lower resonant cavity. upper resonant and the last cavity? lower resonant. The first pairing window can? be positioned at or near the end? distals of the interface between the last cavity? upper resonant and the last cavity? lower resonant. Also, the first pairing window can not overlap the second coupling window or the third coupling window along any axis extending parallel to a longitudinal axis of the metallized dielectric waveguide filter. Figures 5A-5C are various views of a metallized dielectric waveguide filter 100 according to certain embodiments of the present invention. The 100 filter can? be, for example, designed to operate in the 3.4-3.8 GHz band, with a passband return loss of less than 15dB, a passband insertion loss of less than 1.4dB, a rejection out bandwidth of at least 35 dB in the 4.4-5.0 GHz frequency range, and significant rejection for frequencies up to 19 GHz. Filter 100 has a ?bent ? two-layer having a first (upper) row of cavities? resonant from 130-1 to 130-4 and a second row (lower) of cavit? resonating from 130-5 to 130-8. In the embodiment shown, the filter 100 includes a total of eight cavities. resonant 130 and works in TE1,0,1 mode. Will you appreciate it? however that in other embodiments the filter 100 can? have different topologies (for example, two folds to provide three rows of cavities? resonant or as a single row filter) or pu? be designed to operate in a different transmission mode.

The metallized dielectric waveguide filter 100 includes an upper metallized dielectric waveguide 110-1 and a lower metallized dielectric waveguide 110-2. The upper dielectric waveguide 110-1 includes an upper dielectric block 112-1 which has a (mostly) metallized outer surface 114-1. The lower dielectric waveguide 110-2 includes a lower dielectric block 112-2 which has a (mostly) metallized outer surface 114-2. In some embodiments, the upper and lower metallized dielectric waveguides 110-1, 110-2 may have identical designs so that two instances of a single part can be used to implement the upper and lower metallized dielectric waveguides. bottom 110-1, 110-2. This arrangement can reduce manufacturing costs.

As highlighted above, the cavities resonators from 130-1 to 130-4 are formed in the upper dielectric waveguide 110-1 and the cavities? resonants 130-5 to 130-8 are formed in the lower metallized dielectric waveguide 110-2. Pairs of spaced vertically extending holes extend through the upper metallized dielectric block 112-1. These holes are metallized to provide metallized holes 116. Each pair of metallized holes 116 forms a respective mating window 140. resonant 130-1 to the cavity? resonant 130-2, the coupling window 140-2 connects the cavity? resonant 130-2 to the cavity? resonant 130-3, and the coupling window 140-3 connects the cavity? resonant 130-3 to the cavity? resonant 130-4. Similarly, pairs of spaced vertically extending holes extend through the lower dielectric block 112-2. These holes are metallized to provide metallized holes 116. Each pair of metallized holes 116 forms a respective mating window 140. resonant 130-5 to the cavity? resonant 130-6, the coupling window 140-5 connects the cavity? resonant 130-6 to the cavity? resonant 130-7, and the coupling window 140-6 connects the cavity? resonant 130-7 to the cavity? resonant 130-8.

A 102-1 inlet orifice ? coupled to the cavity? resonant 130-1 and an outlet orifice 102-2 (not shown in FIGS. 5A and 5B, and only partially shown in FIG. 5C)? coupled to the cavity? resonant 130-8.

Three regions on the bottom surface of the cavity resonant 130-4 are non-metallized, and three regions on the top surface? of the cavity? resonant 130-5 are similarly non-metallic. These non-metallized regions may be vertically aligned (or at least vertically overlapping) to form three coupling windows 142-1 to 142-3 which connect the cavity resonant 130-4 to the cavity? resonant 130-5. In addition, a region on the bottom surface of the cavity resonant 130-3 and a corresponding region on the top surface? of the cavity? resonant 130-6 are not metallized to provide a cross coupling window 150 between the cavities? resonant 130-3 and 130-6. A main transmission path through the filter 100 extends from the inlet port 102-1, through the eight cavities resonating from 130-1 to 130-8 in numerical order and then through output orifice 102-2.

As further shown in Figs. 5A-5C , metallized circular holes 118-1, 118-2 are formed in the top surfaces of the cavities? upper resonant resonators 130-2, 130-3, respectively, and metallized circular holes 118-3, 118-4 are formed in the bottom surfaces of the cavities. lower resonants 130-7, 130-6, respectively. These holes are referred to herein as ?blind holes? and can be used to shape the electromagnetic field to increase coupling across a coupling window and/or to increase the center frequency of the spurious first mode, as will be the case? discussed in detail herein. Finally, the first and second longitudinally extending slots 120-1, 120-2 are formed in the top surface? of the cavity? resonant upper 130-4 and in the bottom surface of the cavity? lower resonant 130-5, respectively. The slots 120-1, 120-2 each extend longitudinally along a central axis of the respective recesses. resonant 130-4, 130-5, and in the embodiment shown extend the full distance to the extreme? distal of the respective cavities? resonant cavities 130-4, 130-5 (ie the ends of the resonant cavities 130-4, 130-5 which are the most distant from the inlet and outlet orifices 102-1, 102-2).

In the embodiment shown, the upper and lower metallized dielectric waveguides 110-1, 110-2 are identical so that the filter 100 can be constructed by welding two identical pieces together. There? can? reduce manufacturing costs. The metallization on the bottom surface of the upper 110-1 metallized dielectric waveguide and on the top surface is of the lower metallized dielectric waveguide 110-2 allows the two metallized dielectric waveguides 110-1, 110-2 to be welded together. In other embodiments, however, the upper and lower metallized dielectric waveguides 110-1, 110-2 may have different structures to further fine tune the performance of the filter 100. In one exemplary embodiment, the dielectric material may ? understand a pottery. A dielectric constant of the dielectric material can? be for example between 15 and 40.

Figure 6 ? a schematic view of one of the cavities? resonant cells 130 of the metallized dielectric waveguide filter 100. As shown in FIG. resonant 130 has a length l, a width w and a height h. The length dimension ? the longitudinal direction of the dielectric block 112 in which ? formed the cavity? resonant 130. The dimensions of width and height are transverse to the dimension of length and perpendicular to each other. All three dimensions are also shown in figure 5C in relation to the cavity resonant 130-1.

each cavity? resonant 130 of the filter 100 has a resonant frequency. For a cavity? resonant 130 which has a rectangular shape, the resonant frequency can? be determined based on the size of the cavity? and to the dielectric constant (?) of the dielectric material such as

follows:

in which ? ? the permeability? magnetic of the dielectric material. Typically each cavity? resonant 130 ? designed to have approximately the same resonant frequency, which can? be the center frequency of the passband.

How ? evident from? Equation (1), a desired resonance frequency for a cavity? resonant 130 pu? be obtained by manipulating the length (l) and width (w) of the cavity? resonant 130 and the dielectric constant of the dielectric block 112. However, the length l and the width w (as well as the height h) heavily affect the electric and magnetic field distributions inside the cavity? resonant 130. Consequently, the length l, the width w and the height h must also be selected to take into account the couplings that are necessary between cavities? adjacent and non-adjacent resonant 130 to achieve a desired filter response.

The coupling between cavities? adjacent resonant chambers 130 includes both horizontal couplings, which refer to couplings between two cavities? resonant 130 which are part of the same metallized dielectric waveguide 110, both vertical couplings, which refer to couplings between two cavities? resonant sensors 130 which are adjacent along the main transmission path through the filter 100, but which are part of different metallized dielectric waveguides 110-1, 110-2. The horizontal and vertical couplings can be collectively defined as ?in line? as they are couplings between cavities? resonant channels 130 which are adjacent to each other along the main transmission path. In addition one or more Can cross couplings be expected between two cavities? resonators 130 which are not adjacent to each other along the main transmission path through the filter 100 and which are part of different metallized dielectric waveguides 110-1, 110-2. Fig. 7A shows the couplings ki,j that are present in the filter 100. As shown in Fig. 7A , there are seven couplings in line (k1,2, k2,3, k3,4, k4,5, k5,6, k6 ,7, k7,8) and two cross mates (k2,7, k3,6). The value of the percentage of

coupling ki,j between two cavities? resonant ? defined by equation (2): where ? the central frequency of the first transmission mode e

? the center frequency of the second transmission mode.

Yup ? found that, in order to configure the filter 100 to have a high bandwidth, the values of many or even all of the line couplings may need to be greater than 5% (e.g. in the range of 5-10% or even more?). Table 1 below provides the values of each of the fits for one embodiment of filter 100.

TABLE 1

k1.2 = 9.524% k2.3 = 6.772% k3.4 = 5.8474%

k4.5 = 8.243% k5.6 = 5.8474% k6.7 = 6.772%

k7.8 = 9.524% k2.7 = 0.082% k3.6 = -2.321%

Can couplings be characterized by their polarity? (positive or negative). This ? shown schematically in Figure 7B. Filter 100 includes only three of the four types of couplings shown in Fig. 7B . In accordance with embodiments of the present invention, techniques have been developed for generating strong matches for each of the three types of matches included in the filter 100.

Positive horizontal couplings (for example the couplings k1,2, k2,3, k3,4, k5,6, k6,7, k7,8) can be easily generated by doing s? that the magnetic field distributions in the cavities? resonant 130 overlap in the vicinity? of the coupling windows 140 that connect cavities? adjacent resonant 130. Ci? ? shown for example in FIGS. 8A and 8B, wherein FIG. 8A is a view from? top of the two cavities? adjacent resonant cells (here the resonant cavities 130-2 and 130-3) and figure 8B ? a cross-sectional view illustrating the magnetic fields generated in the two cavities? resonators 130 of figure 8A. The greatness of? coupling pu? be controlled by the size of the coupling window 140-2 (which in this embodiment is a window having a width w1). As discussed above, if the coupling is to be greater than about 5%, the size of the coupling window 140-2 (here w1) must be large. Unfortunately, since the size of the 140-2 coupling window is ? increased, the center frequency of the first spurious mode moves towards the passband. There? can? be problematic when, as here, ? It is necessary to have a high level of rejection at frequencies that are close to the passband. In addition, if the size of the coupling window 140-2 becomes too large, the transmission mode of the RF energy may change. switch from mode TE1,0,1 to mode TE1,0,2, and also this ? problematic.

Figure 9A ? a graph illustrating how the positive horizontal coupling increases as a function of the width w1 of the coupling window 140-2. The coupling percentage k2,3 increases from about 0.5% to almost 6% as the width w1 of the coupling window 140 ? increased from 2.5mm to 6.0mm. Figure 9B ? a graph that illustrates the variation of the central frequency of the first mode of spurious transmission for the two cavities? resonant channels 130-2, 130-3 shown in Fig. 9A as a function of the width of the coupling window 140-2 with each other (wherein each curve in Fig. 9B corresponds to a different width w1). As shown in Fig. 9B , the center frequency of the first spurious mode decreases as the coupling window width increases, and drops below 5 GHz to a width of about 5 mm. Together, Figures 9A and 9B show that as the coupling window width is increased from 2.5mm to 6mm, the coupling increases from 0.5% to 6%, while the center frequency of the first spurious mode drops from 5.36GHz to 4.83GHz.

To keep the size (here the width w1) of the coupling window 140 small (to prevent the central frequency of the first spurious mode from getting too close to the passband), the length l, the width w and the height h of the cavities? resonators 130 can be adjusted to increase the level of magnetic coupling in the proximity? of the 140-2 mating window. Change the height h of the cavities? resonant 130 pu? not be particularly effective due to the increase in the resistance of the magnetic coupling in the vicinity? of the coupling window 140. However, change the width w of each cavity? resonant 130 actually has a significant impact on the magnetic field distribution. As discussed above, the resonant frequency of each cavity? resonant 130 ? as a function of both the width w and the length l of the cavity? resonant 130. Therefore, to maintain the? appropriate resonance frequency, pu? be necessary to adjust the length of the cavity? resonant 130 as well as? the width w to increase the coupling in the vicinity? of the coupling window 140 while maintaining the proper resonant frequency. In other embodiments, cavity? different can be formed with dielectric materials having different dielectric constants instead? or in addition to changing the length and width of the cavities?. In practice, this solution pu? be more difficult to implement and/or more? expensive, but can? be advantageous in certain applications, particularly where the filter size must fit within tight constraints.

Table 2 below gives an example of how the width w and length l of a cavity resonant 130 can be modified while maintaining the resonant frequency of the cavity? resonant 130 to a desired value (which in the present example is 3.6 GHz). As shown in Table 2, the exemplary combinations of w and h provide quantities widely different mating. Given that the? objective in the present example ? have a strong coupling while maintaining the frequency of the first spurious mode above 5.0 GHz, set the width of the cavities? resonant 130 to 13 mm and the length of the cavities? resonant 130 to 11.5mm appears to be the best candidate of the four exemplary combinations.

TABLE 2

Width Length ki,j (%) Frequency of 1? (mm) (mm) spurious mode (MHz) 11 16 1.4 4870

12 13.15 4 5170

13 11.5 8 5090

14 10.4 13 4730

The second type of filter coupling 100 ? a positive vertical coupling. Given that this type of coupling ? positive, ? generated once again by? coupling between the magnetic fields in the two cavities? adjacent resonants 130. Figure 10 ? a horizontal cross-sectional view of the fourth cavity? upper resonant 130-4, in which the horizontal cross section is ? taken near the bottom of the cavity? resonant 130-4. The arrows in figure 10 illustrate the intensity? and the direction of the magnetic field, the lines of greater weight indicating more magnetic fields? intense. As shown in Figure 10, there are four regions where the magnetic field is the most intense, which are indicated by the boxes labeled 160-1 through 160-4 in Figure 10.

To provide strong positive vertical coupling between cavities? resonant 130-4 and 130-5, the metallization in the bottom of the cavity? resonant 130-4 and the metallization in the top? of the cavity? resonant 130-5 may be removed (or never be formed) in regions 160 where magnetic fields are strong. This, advantageously, also minimizes the electric field coupling between cavities? resonant 130-4 and 130-5 (which is the negative coupling that effectively offsets the magnetic coupling), since the electric field is? more intense in the center of each cavity? resonant 130-4, 130-5 where the magnetic field ? more weak.

In one exemplary embodiment, three mating windows 142-1, 142-2, 142-3 (i.e., non-metallized areas) were formed between the cavities? resonators 130-4 and 130-5 in the positions corresponding to the regions from 160-1 to 160-3 in figure 10. The smallest dimension? long w2 of each coupling window 142-1, 142-2, 142-3 (ie the width or length, based on the orientation of the coupling window 140) ? been changed from 4mm to 6.5mm. Figure 11A ? a graph that shows how the resistance of? positive vertical coupling between the cavities? resonant 130-4 and 130-5 varies with the variation of the dimension pi? long w2 of the coupling windows from 160-1 to 160-3. As shown in figure 11A, the positive vertical coupling between the cavities? resonators 130-4 and 130-5 varies from about 2.5% at w2 = 4 mm to about 13% at w2 = 6.5 mm. Figure 11B shows how the center frequency of the first spurious mode changes as a function of the pi? long w2 of the coupling windows 142-1 to 142-3, ranging from 5.68 GHz for w2 = 4 mm to 4.63 GHz for w2 = 6.5 mm. Thus, similar to horizontal positive couplings, where increasing the width of the coupling windows from 140-1 to 140-6 provided both improved (i.e., stronger) coupling and degraded (i.e., lower frequency) spurious mode rejection. ), here increasing the width of the coupling windows 142-1, 142-2, 142-3 favors only one of the two parameters (i.e. provides stronger coupling) while degrading the efforts to maintain the frequency of the first mode spurious more away from the bandwidth.

In an exemplary embodiment, the first coupling window 142-1 extends generally transversely over an interface between the cavity upper resonant 130-4 and the cavity? lower resonant 130-5. The second and third mating windows 142-2, 142-3 each extend generally longitudinally at or near opposite lateral edges of the interface between the cavity and upper resonant 130-4 and the cavity? lower resonant 130-5. The first mating window 142-1 ? located at or near one end? distal of the interface between the cavities? resonant 130-4 and 130-5.

In accordance with embodiments of the present invention, several additional techniques may be employed to obtain sufficient levels of positive vertical coupling between the cavities. resonant 130-4 and 130-5 while maintaining the center frequency of the first spurious mode above 5 GHz. In the first of these techniques, ? is it possible to adjust the height h of the cavities? resonant 130. Reducing the height h pu? advantageously push more? above the central frequency of the first spurious mode. Unfortunately, however, the relationship between the height h of the cavity? resonant and the width w of the cavity? resonant ? another key performance factor of the 100 filter, since the Q factor of the 100 filter can be optimized when the h/w ratio = 0.5. Given that the width w pu? be set on the basis of other considerations (as discussed above), reduce the? height h pu? degrade the performance of the filter 100. In some cases, however, pu? be tolerated a certain amount? of deviation from the optimal height-to-width ratio (e.g. h/w = 0.45) to move the center frequency of the first spurious mode away from the passband. Figure 12 illustrates how the frequency of the first distant spurious mode changes as a function of height-to-width ratio for the particular design of filter 100 discussed herein. In exemplary embodiments, a ratio of a height of one or more of the cavities? resonant 130 and the width of the cavity? resonant can? be between 0.42 and 0.48.

The use of so-called blind holes ? known in metallized dielectric waveguide filters. A blind hole refers to a hole that ? formed in the top portion? of the dielectric block in a cavity? resonant 130, and the outside of the hole ? metallic. Figure 13A ? a shaded perspective view of the cavity? resonant 130-4 when implemented with a blind hole 118 in its upper portion. In the particular example here, add a blind hole 118 and implement the blind hole 118 in order to have a depth? d2 pushed the central frequency of the first spurious way pi? top in terms of frequency. As shown in the graph of figure 13B , this technique has a limited capacity to change the central frequency of the first spurious mode, and ? been achieved a variation of only about 40 MHz even when you ? used a relatively deep 118 blind hole. Does the use of a blind hole 118 also change the resonance frequency of the cavity? resonant 130-4, but this pu? be compensated by changing the length l of the cavity? resonant 130-4.

Yup ? discovered that the formation of a longitudinally extending slot 120-1 in the upper dielectric block 112-1 at the top of a cavity? resonant 130 pu? have a greater impact on the center frequency of the first spurious mode than a blind hole 118, and pu? be used to raise the center frequency of the first spurious mode by more than 100 MHz. The 120-1 slot can be a longitudinally extending slot 120-1 (i.e., the slot extends along the main transmission path of the upper metallized dielectric waveguide 110-1). Figure 14A ? a shaded perspective view of the cavity? resonant 130-4 when implemented with a slot 120-1 in its upper portion. The central frequency of the spurious mode generally increases with the increase of the width w3 and of the depth? d3 of crack. A length l3 of the slot 120-1 can? be greater than a width w3 of the slot 120-1. Does adding the 120-1 slot reduce the resonance frequency of the cavity? resonant 130-4 with the reduction of the resonance frequency that increases as the depth increases? of slit d3. This reduction of the resonant frequency pu? be compensated, however, by reducing the length of the cavity? resonant 130-4. Increase the width w3 of the slot 120-1 pu? push even more at the top, the central frequency of the spurious mode but, as the width w3 and/or the depth? d3 of the slot 120-1 become too large, the Q factor of the filter 100 pu? deteriorate unacceptably. Figure 14B ? a graph that illustrates the central frequency of the first spurious mode as a function of the depth? of slit d3, assuming that the slit 120-1 has a length l3 of 2 mm. In this particular example, adding the 120-1 slot (with a depth of 2mm) could make it easier for you to add the 120-1 slot (with a depth of 2mm). increase the center frequency of the first spurious mode by about 130 MHz.

In the 100 filter, the 120-1 slot? provided in the cavity? resonant 130-4 and the slot 120-2 ? provided in the cavity? resonant 130-5. Embodiments of the present invention, however, are not limited thereto. In other embodiments, any of the remaining cavities resonants that include one or more of the cavities? resonating from 130-1 to 130-3 and from 130-6 to 130-8 pu? include 120 similar or identical slots. Can the slots 120 extend the full length of their respective cavities? resonating 130 or less over their entire length. The slots 120 may or may not extend to the outer end. distal of their respective cavities? resonant cells 130. The slots 120 may extend along the length direction 1 of the filter 100.

The presence of the cracks 120-1, 120-2 pu? significantly reduce the length of each of the cavities? resonant cavities 130-4 and 130-5, since the presence of the slots 120 changes the resonant frequency in a way that necessitates the reduction of the length of each cavity? resonant 130-4, 130-5 to obtain the desired resonant frequency. Therefore, the presence of the fissure can? reduce the size of the filter 100, thereby reducing its cost and weight.

The metallized dielectric waveguide filter 100 further includes two cross-fits between respective pairs of cavities. resonant sensors 130 which are not adjacent to each other along the main transmission path. Particularly, ? expected a negative cross coupling between the cavities? resonant 130-3 and 130-6 and ? expected a positive cross coupling between the cavities? resonant 130-2 and 130-7. The negative cross coupling ? generated by mixing the electric fields for the two cavities? resounding 130-3, 130-6 in question. As discussed above, the electric field distribution ? more intense in the center of the cavities? resounding 130-3, 130-6. Therefore, a negative cross match can be easily generated by omitting the metallization in the center of the bottom of the cavity? resonant 130-3 and omitting the metallization in the center of the top? of the cavity? resonant 130-6. These non-metallized regions may be vertically aligned or may be at least vertically overlapping (i.e., a vertical axis extending through the filter 100 passes through the non-metallized regions in both the bottom of the resonant cavity 130-3 and the top of the resonant cavity 130-3). resounding 130-6). Here, circular openings in the metallization may be preferred to maximize the amount of coupling for an opening of a given size. Figure 15A ? a horizontal cross-sectional view of the cavity? resonant 130-3 taken near its bottom which illustrates the? intensity? and the direction of the magnetic fields as well as? the size and position of a circular cross coupling window 150 which ? formed by omitting the metallization in the center of the bottom surface of the cavity? resonant 130-3 and in the center of the top surface? of the cavity? resonant 130-6.

Figures 15B-15C show the impact of the size of the cross-coupling window 150 on both the magnitude of the negative cross-coupling and the center frequency of the first spurious mode. As shown in FIG. 15B , increasing the radius r of the cross coupling window 150 from 1.1 mm to 2.6 mm increases the negative cross coupling from 0.2% to 2.7%. Unfortunately, however, this change results in a corresponding decrease in the first spurious mode center frequency from 5.6 GHz to 5.06 GHz or a decrease of 540 MHz. It should be noted that even with a large 150 cross-coupling window ? difficult to get quantity? high levels of negative cross-coupling.

The negative cross coupling pu? be increased by reducing the height h of the cavities? resonant 130-3 and 130-6 or by increasing the width w of the cavities? resonant 130-3 and 130-6. Therefore, reductions in the h/w ratio result in increased coupling. Figure 16A illustrates how the h/w ratio affects negative cross coupling. Fig. 16B illustrates the center frequency of the first spurious mode for several of the h/w values shown in Fig. 16A . In particular, reducing the h/w ratio advantageously increases the magnitude of the negative cross-coupling and also increases the center frequency of the first spurious mode. From a loss point of view, ? better in general to increase the width w instead of? reduce the height h. However, as discussed above, increasing the width w has a negative impact on the generation of positive vertical coupling. Therefore, there ? a compromise involved in increasing the width w of the cavities? resonant.

Provide a blind hole 118-2 in the top? of the cavity? resonant 130-3 and/or in the bottom of the cavity? resonant 130-6 pu? help to increase the size of the negative cross coupling and/or pu? further move the center frequency of the first spurious mode away from the passband. Figures 17A and 17B show these respective effects for the filter 100 in which the height h of each of the cavities resonant 130-3, 130-6 ? equal to 0.45 times the width w of the respective cavities? resonators 130-3, 130-6 (which are identical cavities in this embodiment). As shown in figures 17A and 17B, as the depth? of the blind holes 118-2 increases, increases the negative cross coupling and similarly increases the center frequency of the first spurious mode. The length l of the cavities? resonators 130-3, 130-6 must be reduced to compensate for the loading effect of the blind holes 118-2. Furthermore, the Q factor of the 100 filter is also ? reduced by the presence of the blind hole 118-2, and the reduction of the Q factor increases with increasing depth? for blind holes 118-2. Therefore, the blind holes 118-2 can be expected to generate sufficient negative cross-fit, but can? be necessary to limit the depth? of blind holes 118-2 to maintain a minimum required Q factor for filter 100.

Two metallized dielectric waveguide filters were designed to study the trade-off between insertion loss and drift of the first spurious mode frequency from the passband. The first design? a rectangular waveguide and the design focuses on minimizing insertion loss. The second design? a single bump waveguide and focuses on providing superior out-of-band rejection.

Figure 18 ? a perspective view of the filter according to the first design. Filter dimensions are 44.65mm x 12.6mm x 10.73mm (without 102 input and output connectors). The total volume (without connectors) ? of 6.04 mL. Figure 19A ? a graph of the simulated return loss for the first filter design. As shown in FIG. 19A, the return loss ? greater than -18 dB throughout the entire 3.4-3.8 GHz passband, with at least 4 MHz headroom on each side of the passband. Figure 19B ? a graph of simulated insertion loss for the first filter design. As shown, the insertion loss ? less than 1 dB across the entire 3.4-3.8 GHz bandwidth. Figure 19C ? an insertion loss graph for a higher frequency range showing the simulated out-of-band rejection performance of the first filter design. As shown, the out-of-band rejection ? of at least 35 dB at all frequencies between the passband and 5 GHz. An additional low pass filter can? be used to obtain the additional 50dB of rejection needed in this frequency range.

Figure 20 ? a perspective view of the filter according to the second design. This filter can be the filter 100 of FIGS. 5A-5C. Filter dimensions are 37.62mm x 13.0mm x 11.73mm (without connectors). The total volume (without connectors) ? of 5.74 mL. Figure 21A ? a graph of the simulated return loss for the first filter design. As shown in figure 21A, the return loss ? greater than -20 dB throughout the entire 3.4-3.8 GHz passband, with at least 4 MHz headroom on each side of the passband. There? represents excellent return loss performance. Figure 21B ? a graph of simulated insertion loss for the first filter design. As shown, the insertion loss ? less than 1 dB throughout the 3.4-3.8 GHz bandwidth. Figure 21C ? an insertion loss graph for a higher frequency range showing the simulated out-of-band rejection performance of the first filter design. As shown, the out-of-band rejection ? of at least 66 dB at all frequencies between the passband and 5 GHz. An additional low pass filter can? be used to obtain the additional 20dB of rejection needed in this frequency range, and to provide the necessary level of rejection at frequencies above 5GHz. have less than 15 dB return loss throughout a 3.4-3.8 GHz band and have at least 50 dB out-of-band rejection in the 4.4-5.0 GHz band.

Filters according to embodiments of the present invention may for example be suitable for use in time division duplex adaptive antennas operating in the frequency range of 2.3-6 GHz.

Embodiments of the present invention have been described above with reference to the accompanying drawings, in which embodiments of the invention are shown. These invention can? however, be made in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure is thorough and complete and fully conveys the scope of the invention to those skilled in the art. Like numerals refer to like elements throughout the text.

Will you understand? which, even if the terms first, second, etc. may be used here to describe various items, these items should not be limited by these terms. These terms are used only to distinguish one item from another. For example, a first element could be referred to as a second element and similarly a second element could be referred to as a first element, without departing from the scope of the present invention. As used herein, the term ?and/or? includes any and all combinations of one or more? of the associated listed items.

Will you understand? that when it is indicated that an element ? ?on? another element, it pu? be directly on the other element or there may also be intermediate elements. By contrast, when is it indicated that an element is ? ?directly on? another element, there are no intermediate elements present. Will you understand? moreover that when it is indicated that an element ? ?connected? or ?coupled? to another element, it pu? be directly connected or coupled to the other element or there may be intermediate elements. By contrast, when is it indicated that an element is ? ?directly connected? or ?directly coupled? to another element, there are no intermediate elements present. Other terms used to describe the relationship between elements should be interpreted similarly (ie ?between? versus ?directly between?, ?adjacent? versus ?directly adjacent?, etc.).

Related terms such as ?below? or ?above? or ?upper? or ?inferior? or ?horizontal? or ?vertical? may be used herein to describe a relationship of one element, layer or region to another element, layer or region as illustrated in the figures. Will you understand? that these terms are intended to include different orientations of the device in addition to the orientation shown in the figures.

The terminology used herein is only intended to describe particular embodiments and is not intended to be limiting of the invention. As used here, the singular forms ?un?, ?uno?, ?una? and ?the?, ?the?, ?the? intend to include plural forms as well, unless the context clearly indicates otherwise. Will you understand? moreover that the terms ?includes? ?comprising?, ?includes? and/or ?including?, when used herein, specify the presence of the indicated characteristics, operations, elements and/or components but do not exclude the presence or addition of one or more? further characteristics, operations, elements, components and/or assemblies thereof.

Aspects and elements of all of the embodiments described above may be combined in any way and/or combination with aspects or elements of other embodiments to provide a plurality of functions. of additional embodiments.

Claims (11)

1. Metallized dielectric waveguide filter, comprising: a superior metallized dielectric waveguide having a plurality? of cavities? upper resonant elements, the upper metallized dielectric waveguide comprising an upper dielectric block having metallized outer walls; a lower metallized dielectric waveguide having a plurality? of cavities? resonant resonators, the lower metallized dielectric waveguide comprising a lower dielectric block having metallized outer walls, in which a first of the cavities? higher resonants ? operatively connected to a first of the cavities? lower resonants through at least one coupling window, e wherein a first crack having metallized walls ? provided in a portion of the upper dielectric block which is part of the first of the cavities? higher resonants. The metallized dielectric waveguide filter according to claim 1, wherein the first slit extends: ? up to one end? distal of the first of the cavities? higher resonants; or ? longitudinally along a longitudinal axis of the upper metallized dielectric waveguide. The metallized dielectric waveguide filter according to claim 1, wherein the upper metallized dielectric waveguide is? mounted on the lower metallized dielectric waveguide to form a pleated filter having a generally U-shaped main transmission path, and wherein the metallized dielectric waveguide filter includes a first cross coupling between a second of the cavities; upper resonants and a second of the cavities? lower resonants; further preferably comprising a cross coupling window comprising a non-metallized portion of a bottom surface of the second of the recesses? upper resonant and a non-metallized portion of a top surface? of the second of the cavities? lower resonant that at least partially overlaps the non-metallized portion of the bottom surface of the second of the cavities? higher resonants; more Preferably does the metallized dielectric waveguide filter include a second cross-fit between a third of the cavities? upper resonants and a third of the cavities? lower resonant, where the second cross-coupling passes through the cross-coupling window, most preferably ? the first cross mate has a negative sign and the second cross mate has a positive sign; or ? the cross-coupling window ? located in the center of a region in which the bottom surface of the second of the cavities? higher resonants ? abutting on the top surface? of the second of the cavities? lower resonants. The metallized dielectric waveguide filter according to claim 1, wherein the upper metallized dielectric waveguide is? mounted on the lower metallized dielectric waveguide to form a pleated filter having a generally U-shaped main transmission path, and wherein the metallized dielectric waveguide filter includes a first cross coupling between a second of the cavities; upper resonants and a second of the cavities? lower resonant, and a portion of the upper dielectric block which is part of the second of the cavities? higher resonators includes a hole having metallized walls. The metallized dielectric waveguide filter according to claim 1, wherein: the dielectric further comprises a first input/output orifice which ? connected to a fourth of the cavities? higher resonants; a second orifice of entry/exit that ? connected to a fourth of the cavities? lower resonant, in which the first of the cavities? higher resonants ? the cavity? resonant of the superior metallized dielectric waveguide that ? the "P? far from the fourth of the cavities? higher resonants; does the mating window include a non-metallized portion of a bottom surface of the first of the cavities? upper resonant and a non-metallized portion of a top surface? of the first of the cavities? lower resonant that at least partially overlaps the non-metallized portion of the bottom surface of the first of the cavities? higher resonants; the upper metallized dielectric waveguide further includes metallized vias which extend through the upper dielectric block to define the cavities resonant resonators, and the lower metallized dielectric waveguide further includes metallized vias which extend through the lower dielectric block to define the cavities in the lower dielectric block. lower resonants; a second fissure having metallized walls ? formed in a portion of the lower dielectric block which is part of the first of the cavities? lower resonants; the upper metallized dielectric waveguide and the lower metallized dielectric waveguide are identical; the at least one coupling window comprises a first coupling window which extends generally transversely over an interface between the first of the cavities? upper resonants and the first of the cavities? lower resonant cavities, and second and third coupling windows extending generally longitudinally at or near opposite lateral edges of the interface between the first and lower resonant cavities. upper resonants and the first of the cavities? lower resonants; the at least one coupling window comprises a first coupling window which extends generally transversely on a bottom surface of the first of the cavities? higher resonants at or near one end? distal of the first of the cavities? higher resonants; the metallized dielectric waveguide filter has less than 15 dB return loss throughout the 3.4-3.8 GHz band and has at least 50 dB out-of-band rejection in the 4.4-5 band, 0GHz; the metallized dielectric waveguide filter? configured so that radio frequency (RF) signals propagate through the metallized dielectric waveguide filter in TE1,0,1 mode; or a ratio between a height of the upper metallized dielectric waveguide and a width of the upper metallized dielectric waveguide ? between 0.42 and 0.48. 6. Metallized dielectric waveguide filter, comprising: a superior metallized dielectric waveguide having a first cavity? superior resonant, a? last cavity? superior resonant and at least one cavity? upper intermediate resonant that ? positioned between the first cavity? upper resonant and the last cavity? upper resonant, the upper metallized dielectric waveguide comprising an upper dielectric block having metallized outer walls; a lower metallized dielectric waveguide having a first cavity? lower resonant, a? last cavity? lower resonant, and at least one cavity? lower intermediate resonant that ? positioned between the first cavity? lower resonant and the last cavity? bottom resonant, the bottom metallized dielectric waveguide comprising a bottom dielectric block having metallized outer walls, in which the last cavity? superior resonant ? operationally connected to the first cavity? lower resonant through a first coupling window, a second coupling window and a third coupling window. The metallized dielectric waveguide filter according to claim 6, wherein each of the first to third coupling windows comprises a respective non-metallized portion of a bottom surface of the last cavity. upper resonant and a respective non-metallized portion of a top surface? of? the last cavity? lower resonant which at least partially overlaps the respective non-metallized portion of the bottom surface of the last cavity? superior resonant. The metallized dielectric waveguide filter according to claim 6, wherein the first coupling window generally extends transversely through an interface between the last cavity? upper resonant and the last cavity? resonant cavity, and the second and third coupling windows extend generally longitudinally at or near opposite lateral edges of an interface between the last cavity and the lower resonant cavity. upper resonant and the last cavity? lower resonant; preferably the first coupling window generally extends transversely on a bottom surface of the last cavity? upper resonant at or near one end? distal of the last cavity? upper resonant; or first mate window doesn't overlap n? to the second mating window n? to the third coupling window along any axis extending parallel to a longitudinal axis of the metallized dielectric waveguide filter. The metallized dielectric waveguide filter according to claim 6, wherein the metallized dielectric waveguide filter further comprises a first slot having metallized walls in a portion of the upper dielectric block forming part of the last cavity. upper resonant; preferably the first crack extends to one end? distal of the last cavity? upper resonant; more preferably the first slit extends longitudinally along a longitudinal axis of the upper metallized dielectric waveguide; most preferably a second fissure having metallized walls ? formed in a portion of the lower dielectric block which is part of the last cavity? lower resonant. The metallized dielectric waveguide filter according to claim 6, wherein the upper metallized dielectric waveguide is? mounted on the lower metallized dielectric waveguide to form a folded filter having a generally U-shaped main transmission path, and wherein the metallized dielectric waveguide filter includes a first cross coupling between a first of the cavities; intermediate upper resonants and a first of the cavities? intermediate lower resonants; preferably further comprising a cross coupling window comprising a non-metallized portion of a bottom surface of the first of the cavities? intermediate upper resonant and a non-metallized portion of a top surface? of the first of the cavities? lower intermediate resonant which at least partially overlaps the non-metallized portion of the bottom surface of the first of the cavities? intermediate upper resonants; more preferably the cross-coupling window ? located in the center of the bottom surface of the first of the cavities? intermediate upper resonant abutting on the surface of the top? of the first of the cavities? intermediate lower resonants. The metallized dielectric waveguide filter according to claim 6, wherein a portion of the upper dielectric block which is part of the first of the cavities? intermediate upper resonant includes a bore having metallized walls; the metallized dielectric waveguide filter further comprises a first inlet/outlet orifice which ? connected to the first cavity? upper resonant; and a second orifice of entry/exit that ? connected to the first cavity? lower resonant; The upper metallized dielectric waveguide further includes metallized vias which extend through the upper dielectric block to define the first cavity. superior resonant, the last cavity? resonant superior, and l?at least one cavity? intermediate upper resonant cavity, and the lower metallized dielectric waveguide further includes metallized vias which extend through the lower dielectric block to define the first cavity. lower resonant, the last cavity? lower resonant, and the? At least one cavity? lower intermediate resonant; the upper metallized dielectric waveguide and the lower metallized dielectric waveguide are identical; the metallized dielectric waveguide filter has less than 15 dB return loss throughout the 3.4-3.8 GHz band and has at least 50 dB out-of-band rejection in the 4.4-5 band, 0GHz; the metallized dielectric waveguide filter? configured so that radio frequency (RF) signals propagate through the metallized dielectric waveguide filter in TE1,0,1 mode; a ratio between a height of the upper metallized dielectric waveguide and a width of the upper metallized dielectric waveguide ? between 0.42 and 0.48.