GB2547449A

GB2547449A - Converter apparatus

Info

Publication number: GB2547449A
Application number: GB1602847.4A
Authority: GB
Inventors: Whitehouse Robert; Taghizadeh Esfanjani Hassan
Original assignee: General Electric Technology GmbH
Current assignee: General Electric Technology GmbH
Priority date: 2016-02-18
Filing date: 2016-02-18
Publication date: 2017-08-23
Also published as: GB201602847D0

Abstract

A converter, particularly a DC-DC converter 100, comprises first and second pairs of resonant circuits 110, a plurality of switches 106, and has first and second commutation states. In the first commutation state, the first resonant circuit of the second pair 111b1 is connected in series to a low voltage terminal 102 and in parallel to the second resonant circuit of the second pair 111b2, and the first resonant circuit of the first pair 111a1 is connected in parallel to the low voltage terminal, and a high voltage terminal 104is connected to ground via the second resonant circuit of the first pair 111a2 in series with the low voltage terminal. In the second commutation state, the first resonant circuit of the first pair is connected in series to the low voltage terminal and in parallel to the second resonant circuit of the first pair, and the first resonant circuit of the second pair is connected in parallel to the low voltage terminal, and the high voltage terminal is connected to ground via the second resonant circuit in series with the low voltage terminal. A number of converters could form units of a multistage converter.

Description

CONVERTER APPARATUS

FIELD OF THE INVENTION

The invention relates to converter apparatus, and in particular but not exclusively to Direct Current to Direct Current (DC- DC) converter apparatus.

BACKGROUND OF THE INVENTION

There are various known designs of DC-DC converters. One class, known as Switched Capacitor (SC) DC-DC converters, may have one of a number of topologies. Both non-resonant and resonant circuits are known. Resonant switched capacitor converters are sometimes preferred as they can achieve lower losses with lower switching frequencies and/or smaller capacitors but, as further discussed below, tuning of the resonant circuits can be problematic.

Resonant switched capacitor converters have been proposed for various systems, for example in relation to stepping a voltage up or down between a generator (such as a wind farm) and a distribution network. In some examples, such as in US2013/0163202 in the name of ABB, modular designs are considered. Modular structures are attractive for their scalability: differing numbers of modules, or stages can provide different step up and step down ratios. In examples, each subsequent stage in the converter design increases (usually doubles) the voltage. This also generally requires that the voltage rating for switches increases in each stage. In such designs, faulty components (or stages) cannot be easily bypassed to allow the converter to continue normal operation, and the conversion ratio is usually fixed.

Therefore, in particular, it may be the case that different devices or designs are required for each of the stages/modules and for the switches within a stage/module, increasing the complexity of design. As will be familiar to the skilled person, in some examples, a switch may comprise a plurality of switching elements (usually semiconductor switching elements such as as Integrated Gate Bipolar Transistors (IGBTs), metal-oxide-semiconductor field-effect transistor (MOSFETs), Thyristors or the like) which are connected in series to provide a switch with a higher voltage rating than the individual switching elements. While such a series connection allows reuse of a single switching element type within a design, the switching elements may be controlled to operate substantially simultaneously to ensure that the voltage supported is shared over the series connection (and not by the switching element, or subset of switching elements, which switches first) and that voltages remain balanced. This can add to the complexity of controlling such circuitry and/or to the complexity of circuit design. In addition, providing a large number of switching elements in series can result in increased switching losses. In some examples, diodes may be connected antiparallel with the switches.

Furthermore energy storage elements (for example, capacitors) in each module will be subjected to a different voltage and therefore may have a different voltage rating depending upon their position within the structure. This means that, to allow operation following a fault within the converter, a redundant module rated for the highest anticipated voltage may be employed regardless of where the failed module is located.

Moreover, in some such converter designs, charge is transferred to an energy storage element such as a capacitor during one half cycle which may lead to a large ripple on the output voltage.

Known DC-DC converters based on resonant circuits often have a disadvantage resulting from coupling of resonant circuits. Tuning such interdependent circuits is complex. This creates a practical problem whereby the tolerances on each of these components means that the actual resonant frequency is not easily determined. As a consequence the switching frequency and the resonant frequency can be different and it is difficult to minimise converter losses.

SUMMARY OF THE INVENTION

According to a first aspect of the invention, there is provided a direct-current to direct-current (DC-DC) converter unit for converting an input voltage to an output voltage, the converter unit comprising a controller; a first and second pair of resonant circuits, wherein each pair of resonant circuits comprises a first and a second resonant circuit; and a plurality of switches, the switches being configured to selectively connect the resonant circuits with a low voltage terminal and a high voltage terminal; wherein the controller is arranged to control the switches such that: in a first commutation state of the converter unit, the first resonant circuit of the second pair of resonant circuits is connected in series to the low voltage terminal and in parallel to the second resonant circuit of the second pair of resonant circuits, and the first resonant circuit of the first pair of resonant circuits is connected in parallel to the low voltage terminal, and the high voltage terminal is connected to ground via the second resonant circuit of the first pair of resonant circuits in series with the low voltage terminal and, in a second commutation state of the converter unit the first resonant circuit of the first pair of resonant circuits is connected in series to the low voltage terminal and in parallel to the second resonant circuit of the first pair of resonant circuits, and the first resonant circuit of the second pair of resonant circuits is connected in parallel to the low voltage terminal, and the high voltage terminal is connected to ground via the second resonant circuit of the second pair of resonant circuits in series with the low voltage terminal.

The proposed topology allows charge transfer paths to be decoupled, in the sense that there may be no common branch or components between these paths. Therefore tuning the resonant frequency of the resonant circuits may be simplified. In particular, in some examples, each resonant circuit may be tuned to approximately the same frequency. The combined (series) resonance of two resonant circuits in parallel is also the same frequency. A small deviation in either resonant frequency will result in a small difference in the charge and discharge voltage at each cycle. However this will eventually transfer to the input/output terminals. This can be contrasted, for example, with circuits such as the circuit presented in US2013/0163202, in which case both charge transfer paths are coupled through a common component (for example, a switch). Therefore two charging paths have interaction and tuning of one path affect the resonant frequency of other paths.

As will be familiar to the skilled person, in relation to the terminals of a converter, the terms ‘high’ and ‘low’ have a relative, rather than absolute, meaning. The terminals could alternatively be referred to as ‘higher1 and ‘lower’ voltage terminals, for example, with the high voltage terminal operating at a higher voltage than the low voltage terminal.

In addition, as there are two charging paths between the high and low voltage terminals of the converter which may reduce the voltage ripple at either or both terminals.

The resonant circuits may comprise a series connection of at least one capacitive device and at least one inductive device.

The converter unit may be a step up converter unit, in which case, in one commutation state, an energy storage device (e.g. a capacitor) of first resonant circuit of a pair is arranged to receive charge from the low voltage terminal and to transfer charge to an energy storage device of the second resonant circuit of the pair, and in another commutation state, the energy storage device of the second resonant circuit of the pair is arranged to receive charge from the low voltage terminal and the energy storage device of the first resonant circuit of the pair, and to output charge to the high voltage terminal.

In other examples, the converter unit may be a step down converter unit, in which case, in one commutation state, an energy storage device of a second resonant circuit may be arranged to receive charge from the high voltage terminal and to transfer charge to the low voltage terminal and an energy storage device of first resonant circuit of the same pair, and, in the other commutation state, the energy storage device of the first resonant circuit of the pair is arranged to receive charge from the energy storage device of the second resonant circuit of the pair and to transfer charge to the low voltage terminal.

In some examples, each connection between at least one resonant circuit and a terminal comprises a number of switches which is proportional to the number of resonant circuits (or to the capacitance thereof) connected with the terminal. In such examples, the switches may have a common voltage rating. This is advantageous as it reduces complexity in manufacturing and repair, as the same type of switch may be used throughout the converter unit. Indeed, in some examples, all the switches are of substantially similar, or the same, current and/or voltage ratings. In some examples, each switch may comprise more than one switching element, for example connected in series and/or in parallel, to provide the desired switching capabilities.

In some examples, the switches are arranged in submodules, each submodule comprising (i) four switches in an H-bridge configuration, the controller being arranged to control two of the switches to be closed in the first commutation state and open in the second commutation state, and to control the other two switches to be open in the first commutation state and closed in the second commutation state or (ii) a pair of switches, and the controller is arranged to control one of the switches of the pair to be closed in the first commutation state and open in the second commutation state, and to control the other switch of the pair to be open in the first commutation state and closed in the second commutation state. Such submodules may comprise inductive elements.

In such a converter unit, each submodule has the same number of switches in a given commutation state at a given time (i.e. the number of switches in a submodule which are in a given switching state in a given commutation state may be the same for all submodules), and therefore the submodules may have the same voltage and current rating.

In one example, at least two parallel submodules interconnect the low voltage terminal and the resonant circuits, and at least two series connected submodules interconnect the high voltage terminal and the resonant circuits. This is a convenient layout which allows the voltage and current rating of the submodules to be the same.

In some examples, the submodules may be provided as an integrated (e.g. preformed, or prefabricated) component. As such, when a switch fails, the submodule may be replaced. In examples where the rating of each submodule is the same, any failed submodule may be replaced with a single spare submodule: there is no need to provide different spare parts for different failed submodules. Such an integrated component may exclude some or all passive components. In particular examples, such an integrated component may exclude any capacitors, for example capacitors which comprises a component of a resonant circuit. Passive components tend to fail less frequently and therefore are replaced less often. Capacitors in particular tend to be bulky and expensive, and therefore an integrated component excluding capacitors will be relatively small and less expensive. In addition, in a multistage converter such as is described in greater detail below, the capacitors of each stage may vary between stages, thus this component is not interchangeable in the same way as submodules as described above.

According to a second aspect of the invention, there is provided a multistage converter comprising at least two converter units according to the first aspect of the invention wherein the high voltage terminal of one converter unit is connected to the low voltage terminal of a second converter unit.

In such a multistage converter, energy storage devices of two adjacent stages transfer charge from one to another and provide a DC voltage difference between the two stages.

In examples where the converter units comprise switches arranged in submodules, the submodules of each stage may have the same current and voltage rating and therefore complexity in manufacturing and repair is reduced.

According to a third aspect of the invention, there is provided a method of DC-DC conversion comprising: providing a first and second pair of resonant circuits, each pair comprising a first resonant circuit and a second resonant circuit; controlling a plurality of switches into a first commutation state in which: the first resonant circuit of the second pair is connected in series to a first voltage terminal and in parallel to the second resonant circuit of the second pair, the first resonant circuit of the first pair is connected in parallel to the first voltage terminal, and a second voltage terminal is connected to ground via the second resonant circuit of the first pair in series with the first voltage terminal reconfiguring the plurality of switches into a second commutation state in which: the first resonant circuit of the first pair is connected in series to the first voltage terminal and in parallel to the second resonant circuit of the first pair, the first resonant circuit of the second pair is connected in parallel to the first voltage terminal, and the second voltage terminal is connected to ground via the second resonant circuit of the second pair in series with the first voltage terminal.

The method may be arranged to step up a voltage, or to step down a voltage.

The method may further comprise reconfiguring the plurality of switches repeatedly between the first and second commutation states to provide voltage conversion over a period of time. The first voltage terminal may be at a higher voltage than the second voltage terminal, or vice versa.

Features described in relation to one aspect of the invention may be combined with those of another aspect of the invention. In particular, the converter unit or multistage converter may be arranged to carry out at least some steps of the method of the third aspect of the invention.

The method, converter unit or multistage converter may be arranged for operation in a relatively high voltage environment (for example, in the kilovolt range and above).

The invention further comprises methods of use of the converter unit or multistage converter described above.

Embodiments of the invention are now described, byway of example only, with reference to the following Figures.

BRIEF DESCRIPTION OF THE DRAWINGS

Figures 1-3 show an example of a switched capacitor converter unit according to an embodiment of the invention;

Figures 4 and 5 show current in a non-resonant and resonant converter respectively; Figure 6 shows current waveforms for different switching frequency strategies;

Figure 7 shows a multistage converter according to an embodiment of the invention; Figure 8 shows another example of a switched capacitor converter unit according to an embodiment of the invention; and Figure 9 and 10 show examples of bi-pole converters.

DETAILED DESCRIPTION OF EMBODIMENTS

Figure 1 shows a resonant switched capacitor DC-DC converter unit 100 comprising a low voltage terminal 102 and a high voltage terminal 104.

The converter unit 100 further comprises a plurality of switches 106, which comprise switches 106a of a first switching group and switches 106b of a second switching group. The switches 106 are arranged in submodules 108, each submodule 108 comprising four switches 106, two switches 106 from each group. Although for simplicity the switches 106 are illustrated as solid state switches, in an example, they may each comprise one or a plurality of transistors, such as Insulated Gate Bipolar Transistors (IGBTs) or any other semiconductor switch, or indeed any other suitable switching device. As shown in the alternative representation of Figure 3, the switches 106 may be associated with an antiparallel diode.

The switches 106 of a submodule 108 are arranged in an H-bridge arrangement. In this example, each submodule 108 may comprise a preformed, i.e. integrated, component. In other examples, a pair of switches 106, comprising one of each group, could instead be provided as a preformed (i.e. integrated) component. In such examples, two such components may form a submodule, or each component may be viewed as a submodule.

The converter unit 100 further comprises four resonant circuits 110, which in this example comprise capacitors 111, comprising two capacitors 111 a1 and 111 a2 of a first pair and two capacitors 111 b1, 111 b2 of a second pair. Each pair comprises a first capacitor 111 a1, 111 b1, and a second capacitor 111a2, 111 b2. It will be noted that the capacitors 111, which may be relatively high voltage components, are outside the submodules 108. Capacitors 111 b1 and 111 a1 have a shared voltage rating. Capacitors 111 b2 and 111 a2 also have a shared voltage rating. However the capacitors within a pair (for example capacitors 111a2 and 111a1) do not have the same voltage rating. Capacitor 111a2 will have a voltage rating “one stage” greater than that of capacitor 111 a1. Each capacitor 111 may comprise one or more capacitive components. This means that capacitors at the same position in the converter unit 100 and which may be connected to the same submodule 108 have the same voltage rating as each other.

In series with each of the capacitors 111 is an inductor 112. This provides an LC resonant circuit 110 (also described as a resonant circuit, or in some cases a ‘resonant tank’ circuit). Each capacitor in the “tank circuit” provides a DC voltage difference between the two groups of submodules 108.

Two submodules 108 are arranged in parallel at the low voltage terminal 102, and a further two submodules 108 are connected in series between the low and high voltage terminals of the converter unit 100. The series connected submodules 108 are connected to the parallel connected submodules 108 via the resonant circuits 110 (i.e. via a series connection of capacitors 111 and inductors 112), which clamp the voltages between the submodules 108. The capacitors in the resonant circuits remove the stress of DC voltage stress between the submodules 108.

The inductors 112 may be provided as part of a submodule 108. In other examples, the inductors 112 (which have a relatively low failure rate compared to switches) may be external to the submodules 108 to reduce the cost, size and complexity thereof.

An example is now described in which the converter unit 100 is transferring power from a low voltage to a high voltage terminal although the skilled person will understand that the power could flow in reverse. The arrangement is such that, in use of the converter unit 100, all switches 106 are subject to substantially the same voltage and current.

Figure 2 shows a first 202, second 204 and third 206 charging path in a first commutation state, in which the switches 106a of the first group are closed, while the switches 106b of the second group are open. In an alternative representation of the converter unit 100 shown in Figure 3, two ‘sub-units’ 302a and 302b may be identified within the converter unit 100, the first 302a comprising two submodules 108 and the first capacitors 111a1, 210b1, and the second 302b comprising two submodules 108 and the second capacitors 111b2,111b2.

The two capacitors 111 b1, 111 b2 of the second pair are connected with the low voltage input 102. This allows the capacitor 111b1 to discharge, and capacitor 111 b2 to charge via the path indicated with the dashed line 202. The polarity of the capacitors 111b1 and 111 b2 will be opposite. This allows the second capacitor 111 b2 of the second pair to charge from the series connection of the input voltage and from the connected first capacitor 111 b1. It will be noted that route 202 is via two capacitors 111 and four switches 106a, each of the switches 106 being in a different submodule 108.

It may also be noted that, although, a simple inspection of the figure may suggest that the capacitors 111 b1 and 111b2 and the low voltage supply are in series, this is not viewed as a series connection: considering the voltages, from an electrical perspective, the summation of voltages (hence a series connection) of supply and capacitor 111 b1 are opposed to (hence in parallel) to that of capacitor 111 b2. Similarly, the capacitor 111a1 is in parallel to the low voltage terminal when the switches 206a of the first group are closed.

The second capacitor 111 a2 of the first pair is connected to the high voltage terminal 104, such that it can discharge thereto (see dashed line 204). Viewed another way, the high voltage terminal is connected to ground via the second capacitor 111 a2. The first capacitor 111 a1 of the first pair is connected to the low voltage terminal 102 (see line 206). Both of these charging routes comprise one capacitor 111 and two switches 106a, with each switch 106a being in a different submodule 108.

In the second commutation state, the second pair of capacitors 111b is connected in a manner similar to the first pair of capacitors 111a in the first commutation state, and vice versa. In particular, the second capacitor 111 b2 of the second pair is connected to the high voltage terminal 104, and can discharge thereto. The first capacitor 111 b1 of the second pair is connected to the low voltage terminal 102. The first pair of capacitors 111a is connected with the low voltage terminal, the second capacitor 111a2 of the second pair charging from the series connection of the input voltage and from the connected first capacitor 111a1.

Each of the first capacitors 111 a1, 111 b1 is connected to the low voltage terminal 102 with one polarity in the first commutation state and with a reversed polarity compared to the first commutation state in the second commutation state.

To consider power transfer from the low voltage terminal 102 to the high voltage terminal 104, the first capacitors (111 a1, 111 b1) cycle between charging from the low voltage terminal and discharging into the second capacitor (111a2, 111 b2) of the same pair. The second capacitors (111 a2, 111 b2) cycle between being charged from a first capacitor (111 a1, 111 b1) and discharging to the high voltage terminal (which therefore provides a load). The two pairs of capacitors operate in antiphase.

The effect of the inductors 112 is now discussed with reference to Figures 4 and 5. Absent the inductors 112, the current through a switch 106 will have a profile which depends on the switching frequency and the time constant of charging and discharging paths of capacitors.

If the capacitor is barely charged (relative to its capacity), the current profile will appear as shown in Figure 4a. Partial charging results in a profile as shown in Figure 4b and full charging results in the profile seen in 4c. The highest efficiency can be achieved in ‘no charging’ mode where the ratio of the root mean square current lrms to the average current lavg is lowest. However this operation mode requires very high switching frequencies and/or using large capacitors (‘large’ being understood in the context of the power level of the circuits). Whilst this may be practical and economical in relatively low power applications such as integrated circuits, as power increases, so does cost.

By employing inductors and operating in resonant mode, the charging and discharging current waveforms are reformed as shown in Figure 5. A low I rms / lavg ratio can be achieved at much lower switching frequencies (in some examples, around 10 to 20 times lower) and/or capacitance requirement compared to a non-resonant equivalent circuit, which makes it possible to economically implement these converters in high power applications, for example from tens of watts up to the MW range or higher. In some examples, switching frequency may be reduced to around 10 or 20 times lower by operating in the resonant mode while achieving a similar efficiency as a non-resonant equivalent circuit. However implementation of inductors in integrated circuits is difficult, so a non-resonant circuit is normally used in these applications.

For example when the switches of the second group 106b are on in the converter 100, capacitor 111b1 is charged directly from input source (path 1), capacitor 111a2 is charged through series connection of input source and capacitor 111a1 (path 2), finally capacitor 111 b2 is connected in series with input source and is discharged to the load (path 3). All these three mentioned paths are decoupled, and there is no common branch or components between these paths.

For efficient operation, all charging/discharging paths may have at least substantially the same resonant (tuned) frequency. The resonant frequency can be adjusted in each path by tuning the value of capacitor 111 and inductor 112 in a given group (i.e. in a particular resonant ‘tank’). If the resonant frequency differs between the various resonant circuits 110, the capacitors 111 will not be fully charged/discharged in a cycle. As can readily be seen, in this example, there is no interdependency between the various charge transfer paths, and therefore the circuits may be tuned with relative ease.

Figure 6 shows current waveforms when the path resonant frequency is less, equal or higher than the switching frequency.

If the switching frequency is equal to the resonant frequency (line 602), the switch operation coincides with zero current.

If, by way of contrast the resonant frequency is higher (line 608) or lower (line 604) than the switching frequency, switching will occur at a non-zero current with correspondingly increased losses.

Operation at a resonant frequency which is higher than the switching frequency and zero current switch turn off (line 608) removes the switching losses but increases the conduction losses and is therefore less efficient than the case where the switching frequency is equal to the resonant frequency (line 602).

Therefore, arranging the switching frequency to be equal to resonant frequency offers substantially zero switching losses.

Interdependencies between charge transfer paths can create issues. For example, if the resonant frequency in the charging path is not the same in the discharging path, tuning a resonant frequency in one path it will affect the resonant frequency in another path; tuning the resonant frequency of the affected path will change the resonant frequency of the previously tuned path, and so on.

In circuits where charge transfer paths are linked through a common branch (for example two charge transfer paths are linked though use of a common switch), the Kirchoff’s Voltage Law equations of the two paths are coupled through the stray inductance and the “on” resistance of the switch, and current in one path affects the current of the other path(s). Designing a converter having linked charge transfer paths so that all the paths have equal resonant frequencies is a complex task.

In the topology set out herein, however, there is no common branch between the charge transfer paths. Therefore each path operates independently from the other paths and the resonant frequency of each path can be simply calculated by 1/27tvTc, where L and C are equivalent inductance and capacitance of the charge transfer path respectively, and the switching frequency set accordingly.

It will also be noted that each group of switches 106 operates in anti-phase, with one group being set to open when the other group is set to close, and this cycle may continue repeatedly. Operation of all submodules 108 within the converter unit 100 is preferably well synchronised so that there is no short circuiting of the interconnecting capacitors 111.

It will also be noted that there are two charging paths between the low voltage and high voltage terminals, which may provide substantially continuous charge transfer, and may reduce the voltage ripple at either or both terminal. The two pairs of capacitors 111, inductors 112 and groups of switches 106 operate in the same manner but in different halves of a charge transfer cycle.

Each charging path 202, 204, 206 comprises a number of switches 106 which is proportional to the number of connected capacitors 111 (i.e. in this example, four switches 106 when there are two capacitors 111 and two switches 106 when there is a single capacitor 111). This allows the voltage rating of all the switches 106 to be the same. Moreover, the switches 106 in a given state are evenly distributed throughout the submodules 108, such that the same number of switches 106 is employed in both halves of the cycle, and half of the switches 106 in any one submodule 108 are closed and half are open.

This therefore allows the voltage and current rating of each submodule 108 to be the same. The current rating of each submodule 108 may be determined to be equivalent to the high voltage/low current terminal of the converter unit 100. The power in/out at the low voltage terminal is at a higher current via the parallel connected modules with charge transfer being, in this example, via the capacitors 111 which also provide the DC voltage difference between the submodules 108 and the terminals.

It will be noted that a common voltage and current rating would also be seen if a submodule 108 was defined as comprising just two switches 106, one from each group. For example, such a submodule could comprise the two capacitors on the left or right side of an H-bridge submodule, or the two switches 106 forming the top half or the bottom half of an H-bridge submodule. In such an example, eight submodules each comprising two switches 106 could be provided and the submodules the voltage and current rating of each submodule to be the same.

In both commutation states, the low voltage terminal current is split between a first and a second capacitor of a group. As noted above, and as shown in Figure 3, in effect, two ‘sub-units’ 302a, 302b are formed within the converter unit 100, the first comprising two submodules 108 and the first capacitors 111 a1, 111 b1, and the second comprising two submodules 108 and the second capacitors 111 a2, 111 b2.

As the skilled person will appreciate, a sub-unit 302 of Figure 3 could be a stage of a multistage converter 700 as shown in Figure 7. Each sub-unit 302 receives a voltage from a source such as a battery 702, or from a preceding sub-unit 302 of the converter 700 and outputs voltage to a further stage or to a load 704 as required. The voltages and current across each switch 106, and within each switching module will remain consistent throughout the stages.

In a multistage converter 700, capacitors 111 in each voltage stage are charged by previous capacitors 111 in series with the input source in one commutation state and may discharge to the next stage capacitor 111 in series connection with input source in the other commutation state. The DC voltage difference between the stages of a multistage converter 700 is provided by the difference voltage between the two converter unit capacitors 111.

The provision of multiple stages allows for a high ratio between input and output voltages.

The converter unit 100 shown in Figures 1-3 has the voltage conversion ratio 1:3. A complementary converter unit 800 can be developed as shown in Figure 8 which may be operated according to similar principles and which gives the same conversion ratio in an opposite direction -1:-3. The two converter units 100, 800 can be connected together to create a bi-pole configuration converter 900 as shown in Figure 9. A second configuration of a bi-pole converter 1000 is shown in Figure 10. In Figure 10, the parallel connected modules of the two complementary converter units 100, 800 are connected. This may reduce the number of stages required to construct a converter of a particular conversion ratio and may reduce the capacitance requirement of a converter with a given conversion ratio by a factor of about four. Furthermore, the two complementary circuits of the converter 1000 shown in Figure 10 can be controlled with a mutual phase shift to reduce the voltage ripple in both input and the output terminals.

While the invention has been illustrated and described in detail in the drawings and foregoing description, such illustration and description are to be considered illustrative or exemplary and not restrictive; the invention is not limited to the disclosed embodiments. Features from one embodiment may be combined with features from another embodiment.

The invention has been described with respect to various embodiments. Unless expressly stated otherwise the various features described may be combined together and features from one embodiment may be employed in other embodiments.

It should be noted that the above-mentioned embodiments illustrate rather than limit the invention, and that those skilled in the art will be able to design many alternative embodiments without departing from the scope of the appended claims.

The word “comprising” does not exclude the presence of elements or steps other than those listed in a claim, “a” or “an” does not exclude a plurality, and a single feature or other unit may fulfil the functions of several units recited in the claims. Any reference numerals or labels in the claims shall not be construed so as to limit their scope.

Claims

1. A direct-current to direct-current (DC-DC) converter unit for converting an input voltage to an output voltage, comprising a controller, a first and second pair of resonant circuits and a plurality of switches, wherein each pair of resonant circuits comprises a first and a second resonant circuit; the switches are arranged to selectively connect the resonant circuits with a low voltage terminal and a high voltage terminal; and the controller is arranged to control the switches such that, in a first commutation state of the converter unit, the first resonant circuit of the second pair of resonant circuits is connected in series to the low voltage terminal and in parallel to the second resonant circuit of the second pair of resonant circuits, and the first resonant circuit of the first pair of resonant circuits is connected in parallel to the low voltage terminal, and the high voltage terminal is connected to ground via the second resonant circuit of the first pair of resonant circuits in series with the low voltage terminal and, in a second commutation state of the converter unit the first resonant circuit of the first pair of resonant circuits is connected in series to the low voltage terminal and in parallel to the second resonant circuit of the first pair of resonant circuits, and the first resonant circuit of the second pair of resonant circuits is connected in parallel to the low voltage terminal, and the high voltage terminal is connected to ground via the second resonant circuit of the second pair of resonant circuits in series with the low voltage terminal.

2. A DC-DC converter unit according to claim 1 in which each connection between at least one resonant circuit and a terminal comprises a number of switches which is proportional to the number of resonant circuits connected with the terminal.

3. A DC-DC converter unit according to any preceding claim in which all the switches are of substantially similar current and voltage ratings.

4. A DC-DC converter unit according to any preceding claim in which the switches are provided as submodules comprising four switches in an H-bridge configuration, the controller being arranged to control two of the switches of the pair to be closed in the first commutation state and open in the second commutation state, and to control the other two switches to be open in the first commutation state and closed in the second commutation state.

5. A DC-DC converter unit according to any of claims 1 to 3 in which the switches are provided as submodules comprising a pair of switches, in which the controller is arranged to control one of the switches of the pair to be closed in the first commutation state and open in the second commutation state, and to control the other switch of the pair to be open in the first commutation state and closed in the second commutation state.

6. A DC-DC converter unit according to claim 4 or claim 5 in which at least two parallel connected submodules interconnect the low voltage terminal and the resonant circuits, and at least two series connected submodules interconnect the high voltage terminal and the resonant circuits.

7. A DC-DC converter unit according to any of claims 4 to 6 in which at least one of the submodules is provided as an integrated component.

8. A DC-DC converter unit according to claim 7 in which the integrated component comprises at least one inductor.

9. A DC-DC converter unit according to any of claims 4 to 8 in which, in use of the converter unit, the number of switches in a submodule which are in a given switching state in a given commutation state is the same for all submodules.

10. A DC-DC converter unit according to any preceding claim in which the switches comprise a first group of switches and a second group of switches, wherein, in use of the converter unit, the controller is arranged to control the switches of one group to be open when the switches of the other group are closed, and the first and second group comprise the same number of switches.

11. A DC-DC converter unit according to any preceding claim in which at least one resonant circuit comprises an energy storage device, for example comprising one or more capacitors.

12. A DC-DC converter unit according to any preceding claim in which each resonant circuit has a resonant frequency of l/2nVLC.

13. A DC-DC converter unit according to any preceding claim in which the controller is arranged to control the switching frequency to be the resonant frequency of the resonant circuits

14. A multistage converter comprising at least two converter units according to any preceding claim, wherein the high voltage terminal of one converter unit is connected to the low voltage terminal of a second converter unit.

15. A method of DC-DC conversion comprising: providing a first and second pair of resonant circuits, each pair comprising a first resonant circuit and a second resonant circuit; controlling a plurality of switches into a first commutation state in which: the first resonant circuit of the second pair is connected in series to a first voltage terminal and in parallel to the second resonant circuit of the second pair, the first resonant circuit of the first pair is connected in parallel to the first voltage terminal, and a second voltage terminal is connected to ground via the second resonant circuit of the first pair in series with the first voltage terminal reconfiguring the plurality of switches into a second commutation state in which: the first resonant circuit of the first pair is connected in series to the first voltage terminal and in parallel to the second resonant circuit of the first pair, the first resonant circuit of the second pair is connected in parallel to the first voltage terminal, and the second voltage terminal is connected to ground via the second resonant circuit of the second pair in series with the first voltage terminal.

16. A method of DC-DC conversion according to claim 15 comprising a method of stepping up a voltage, wherein each resonant circuit comprises an energy storage device and the energy storage device of a first resonant circuit of a pair receives charge from a first voltage terminal in one commutation state and transfers charge to the energy storage device of the second resonant circuit of a pair in another commutation state, and the energy storage device of the second resonant circuit of a pair receives charge from a first voltage terminal and the energy storage device of the first resonant circuit of the pair in one commutation state, and outputs charge to a second voltage terminal in another commutation state.

17. A method of DC-DC conversion according to claim 15 comprising a method of stepping down a voltage, wherein each resonant circuit comprises an energy storage device and the energy storage device of the second resonant circuit of a pair receives charge from a first voltage terminal in one commutation state and shares charge with the energy storage device of the first resonant circuit of the pair in another commutation state, and the energy storage device of the first resonant circuit of the pair receives from the energy storage device of the second resonant circuit in one commutation state, and outputs charges to a second voltage terminal in another commutation state.

18. A method of DC-DC conversion according to any of claims 15 to 17 further comprising substantially continuously transferring charge between the first and second terminals.

19. A method of DC-DC conversion according to any of claims 15 to 18 further comprising reconfiguring the plurality of switches repeatedly between the first and second commutation states.

20. A method of DC-DC conversion according to any of claims 15 to 18 further comprising switching between the first and second commutation states at the resonant frequency of the resonant circuits.

21. A bi-pole DC-DC converter comprising a first and a second converter unit according to any of claims 1 to 10.

22. A bi-pole DC-DC converter according to claim 21 in which the first and second converter units comprise a common low voltage input.

23. A DC-DC converter unit substantially as described herein with reference to any of Figures 1 to 6.

24. A multistage converter substantially as described herein with reference to Figure 7.

25. A bi-pole DC-DC converter substantially as described herein with reference to Figure 8, 9 or 10.