GB2510394A - Envelope tracking power supply with low power modes - Google Patents

Envelope tracking power supply with low power modes Download PDF

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Publication number
GB2510394A
GB2510394A GB1301853.6A GB201301853A GB2510394A GB 2510394 A GB2510394 A GB 2510394A GB 201301853 A GB201301853 A GB 201301853A GB 2510394 A GB2510394 A GB 2510394A
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United Kingdom
Prior art keywords
supply
frequency path
low frequency
high frequency
path
Prior art date
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GB1301853.6A
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GB201301853D0 (en
Inventor
Gerard Wimpenny
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Nujira Ltd
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Nujira Ltd
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Priority to GB1301853.6A priority Critical patent/GB2510394A/en
Publication of GB201301853D0 publication Critical patent/GB201301853D0/en
Priority to PCT/EP2014/051964 priority patent/WO2014118344A2/en
Publication of GB2510394A publication Critical patent/GB2510394A/en
Withdrawn legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H03F1/0216—Continuous control
    • H03F1/0233—Continuous control by using a signal derived from the output signal, e.g. bootstrapping the voltage supply
    • H03F1/0238—Continuous control by using a signal derived from the output signal, e.g. bootstrapping the voltage supply using supply converters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H03F1/0216—Continuous control
    • H03F1/0222—Continuous control by using a signal derived from the input signal
    • H03F1/0227—Continuous control by using a signal derived from the input signal using supply converters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/211—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/217—Class D power amplifiers; Switching amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
    • H03F3/245—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/72—Gated amplifiers, i.e. amplifiers which are rendered operative or inoperative by means of a control signal
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0045—Converters combining the concepts of switch-mode regulation and linear regulation, e.g. linear pre-regulator to switching converter, linear and switching converter in parallel, same converter or same transistor operating either in linear or switching mode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/06—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider
    • H02M3/07—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider using capacitors charged and discharged alternately by semiconductor devices with control electrode, e.g. charge pumps
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/375—Circuitry to compensate the offset being present in an amplifier
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/391—Indexing scheme relating to amplifiers the output circuit of an amplifying stage comprising an LC-network
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/432—Two or more amplifiers of different type are coupled in parallel at the input or output, e.g. a class D and a linear amplifier, a class B and a class A amplifier
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2203/00—Indexing scheme relating to amplifiers with only discharge tubes or only semiconductor devices as amplifying elements covered by H03F3/00
    • H03F2203/20—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F2203/21—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F2203/211—Indexing scheme relating to power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
    • H03F2203/21142—Output signals of a plurality of power amplifiers are parallel combined to a common output

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Amplifiers (AREA)
  • Radar Systems Or Details Thereof (AREA)

Abstract

An envelope tracking modulated supply provides a supply voltage tracking a reference signal, and comprises a low frequency path for tracking low frequency variations in the reference signal and a high frequency path for tracking high frequency variations in the reference signal. The supply voltage is provided based on either the low frequency path alone or on a combination of the low and high frequency paths. The supply voltage can be received by a RF power amplifier for a wireless communication system or mobile device. Combining elements such as inductor 28a and capacitor 30c can block high and low frequencies for the low frequency path and the high frequency path respectively. When the supply is based on the low frequency path the low frequency blocking element 30c can be connected to ground via switch 140. The high frequency path can be disabled in an average power tracking mode, and by opening switch 108 to remove a supply voltage from a switched mode supply, which tracks the reference signal, to linear amplifier 24. Feedback of linear amplifier 24 can be connected between its input and output. When the high frequency path is disabled the power in the low frequency path can be determined and the power amplifier connected to a DC supply voltage if the power exceeds a threshold. The high frequency path is selectively enabled according to a mode such as 2G (GSM/EDGE) or 3G/4G.

Description

LOW PONER MODES FOR 3G/4G ENVELOPE TRACKING MODULADDE
BACKGROUND TO THE INVENTION:
Field of the Invention:
The 3.nventicn relates to envelope tracking modulated power supplies suitable for radio frequency power amplifier applications. The invention is particularly conceited with such power supplies in which a reference signal is used as an input to a low frequency path and a high frequency path, and in which each path generates separate outputs which are combined to form a supply voltage.
Description of the Related Artt
Envelope tracking power supplies for radio frequency power arr1 if iers are well--known in the art. Typically a reference signal is generated based on an envelope of an input signal to be amplified. An envelope tracking power supply generates a power supply for the power amplifier which tracks the reference signal.
Figure 1 shows a prior art envelope tracking (lET) modulator architecture in. which. a frequency splitter 12 i.e used to divide an incoming envelope reference signal on line 10 into a high frequency (HF) path signal on line 14 and a low frequency (LF) path signal on line 16. The frequency splitter 12 may include a low pass filter is in the low frequency path-and a high pass filter 20 in the high frequency path. The signal in the LF path on line 16 is amplified by an efficient switched mode amplifier 22, and the signal in the HF path on line 14 is amplified by a widehand linear amplifier 24 requency selective combiner 26 is used to combine the signals in the LF and HF paths after their respective amplification In Figure 1 the combiner 25 is illustrated as including a low frequency combining element 28 in the low frequency path, and a high frequency combining element 30 in the big. frequency path. A combined signal from the combiner 26 on line. .32 provides a feed to a load 34 which for purposes of description is illustrated as a resistor. In a typical application the load is a power amplifier (PA) , and the reference signal is derived from an input siqnal to he amplified by the cower amplifier.
An example of a power amplifier system incorporating a supply architecture such as illustrated in Figure 1 can be found in "Band Separation and Efficiency Opt imisation in Linear--Assisted Switchxng cower Amplt iers", ouser zadeh et ad., [IEEE Power Electronics Specialists Conference 20061 Figure 2 shows an alternative prior art arrangement in which the frequency select e cnihiner 26 is an inductor-capacitor (LC) combiner. The low frequency combining element is an inductor 28a, and the high frequency corining element is a capacitor 30a, En this arrangement a feedback path 36 takes a signal from the combiner (or modulator) output on line 32, to the input of the linear amplifier 24. The signal on-the feedback path 36 is combined in a combiner 38 with the signal in the high frequency path on Line 14, to provine a feeooack adjusted input to the linear amplifier 24. The inclusion of this feedback path 36 achieves improved tracking accuracy comnared to the arrangement 01. Figure 1, An example of a power amplifier system incorporating a supni.y architecture such as illustrated in Figure 2 can be found in "Efficiency Optimisation in Linear-Assisted Switching Power Converters for Envelope Tracking in RF Power Amplifiers", Yousefzadeh et al.. [IEEE Symposium on Circuits and Systems 20051 Envelope tracking modulated power supplies such as described in the prior art as discussed above offer particular efficiency advantages. However in typical applications envelope tracking offers efficiency improvements only at the high operating powers which require significant modulation of the power amplifier supply voltage, In certain appi..ications when operating at significant back-off from maximum power, operation in envelope tracking mode may result in efficiency degradation rather than efficiency improvement.
It is an aim of ti.e invention to provide an improved power supply in which. an efficiency improvement is offered over a wide range of output powers.
It is an aim of the invention to provide an improved envelope trackig modulated power supply which addresses one or more of the above-stated problems.
SUMMARY OF THE INVENTION
The invention provides an envelope tracking modulated supply for provid.ina a supply voltage tracking a reference sicinal and contrising a low frequency path for tracking low frecuency variations in the reference signal and a high frequency path for tracking high frequency variations in the reference signal, wherein a power amolifier selectively receives a voltage suoply based on either the low frequency path alone or the comhinaton of the low frequency patn and the high frequency path.
There may be provided a combining element for providing the power supply by combining the low frequency and high frequency paths, comprising a high frequency blocking element or elements for the low frequency path and a low frequency blocking element or elements for the high frequency path.
When the power amplifier receives a voltage supply based on. the low frequency path the connection between the high frequency path and the low frequency blocking element be connected to electrical ground.
The low frequency path may be disabled when the connection between the high frequency path-and the low frequency blocking element is connected to electrical ground.
The low frequency path may comprise a switched mode power supp].y for generating an output voltage tracking the reference signal and a supply voltage for a J.inear amplifier in the high frequency path.
The supply voltage may he enabled when tb.e high frequency path is enabled The envelope tracking modulated supply may further comprise a comtnner for combining the outputs of the low frequency path and the high frequency path to generate a modulated supply voltage. The combiner may include a high frequency blocking element for the output of the low frequency path and a low frequency blocking element for the outpu.t of the high frequency path.
The low frequency blocking element may provide a path if or decoupling the output when the high frequency path is disabled. The low frequency blocking element may be a capacitor.
The high frequency path may he disabled in an average power tracking mode of operation.
The high frequency path may be disabled, and there may be determined the power in the low frequency path. The power in the low frequency path may exceed a threshold, and. the supply to the power arilifier may be switched to a DC supply voltage. The power in the low frequency path may he below a threshold, and the supply to the power amplifier may not be swItched to a supolv voltage.
There cray be provided a feedback path from the output of the linear amplifier to the input of the linear amplifier, such that the linear amplifier in the correction path amplifies a siqnal comprising the full spectrum of the frequencies in the reference signal.
An PB amplifier may include a voltage supply stage. A wireless communication system may include a voltage supply stage. A wireless mobile device may include a voltage supply stage.
The invention also provides a method in an envelope tracking modulated supply for providing a supply voltage tracking a reference signal, and comprising a low frequency path for tracking low frequency variations-in the reference signal and a high frequency path for tracking high frequency variations in the reference signal, wherein the method comprises providing a voltage supply based on either the low frequency path alone or the combination of the low frequency path and the high frequency path.
The method may further comprise connectina the connection between the combiner and the output of the high frequency path to electrical ground when the high frequency path is disabled, The method may further comprise disabling the high frequency path in an average power tracking mode of operation.
The method may further comprise determining the power in the low frequency Dath when the high frequency path is disabled, and selectively basing a voltage supply on either the low frequency path alone or a DC supply.
BRIEF DESCRIPTION OF THE FIGUPES
The inventior is now described by way of example with reference to the accompanying Figures, in which: Figure 1 illustrates a prior art envelope tracking modulated supply with high and low f»=requency paths; Figure 2 illustrates a prior art envelope tracking modulated supply incornorating feedback in the high frequency path; Figure 3 illustrates an improved envelooe tracking modulated supply ncororating fecanack ir hi nich:equency pat:.
switcher ripple current elimination in the low frequency path, and a preferred implementation of a switched mode supply, in which embodiments of the invention may he advantageously incorporated; Figures 4(a) and 4(h) illustrate a buck and boost switch mode voltage supply in accordance with a preferred arrangement; Figure 4 (c) illustrates a voltage waveform generated by the switched mode voltage supply of F...ic*res 4 (a) and 4 (b) Fiures 5 (a) and 5(b) illustrate a buck and boost switched mode voltage supply in accordance with a further preferred arrangement; Figures 5 (c) and 5 id.) illustrate voltage waveforms which may he simultaneously generated in a dual*--output buck and boost switched mode voltage supply according to the arrangement of Figure 5(a); Figure 6 illustrates the implementation of a dualoutput buck and boost-sitched t-icde voltage supply according to the arrangement of Figure 5 (a) in an envelope tracking modulated supply architecture according to Figure 3; Figure 7 illustrates the arrangement of Figure 6 modified in accordance with an embodiment of the invention to facilitate an average power tracking mode of operation; arid Figure 8 illustrates an embodiment in which an improvemen.t S is provided for 2G imoletrentaflon DESCRPTION OF THE PREFERRED MODMFNT8: in the following description the invention is described with reference to exenrlary enthodiments and implementations. The invention is not limited to the specific details of any arrangements as set out, which are provided for the purposes of understanding the invention.
Embodiments of the invention are described in the following description in the context of application to particular feedback architectures for the linear amplifier in the high frecuency correction nac.h. The invention and its embodiments are however not necessarily limited to the particular feedback arrangements in the high frequency correction path as showru With reference to Figure.3 there is illustrated the envelope tracking architecture as illustrated in Figure 2, with the addition of an exemplary implementation of the switched mode amplifier 22.
The switched mode amplifier of Figure 3 preferably also includes an arrangement to address a triangular ripple current flowing in the inductor 28a of Figure 2 as a result of the switching of the switched mode amplifier 22. In the arrangement of Figure 2 this ripple current must be shunted through the output stage of the linear amplifier 24 via the capacitor 30a in order to avoid the creation of unwanted voltage errors at the combiner output. The consequential ripple current flowing through the output of the linear amplifier 24 reduces its efficiency.
Figure 2 thus shows a preferable arrangement in which the frequency combiner 26 of Figure 2 is adapted to include an additional capacitor 28c and inductor 2Gb. The magnitude of the coupling factor between. inductors 28a and 2Gb may range between 0 and 1. The inductor 2Gb is connected between the output of the switched mode amplifier 22 and the inductor 28a.
The cazacitor 28c is connected between the common connection of the inductors 28a and 28b and electrical, around.
The ripple current due to the switched mode amplifier 22 flowing in the.inductor 2Gb and is now shunted to ground via the capacitor 28c. The loss associated cgj the ripple current flowing in inductor 28a oassinq through the linear output; stage as occurs in the Figure 2 envelope tracked modulated supply is thus avoided.
As regards the exemplary implementation of the amplifier, in a preferred arrangement th.e LF path switched mode amplifier 22 is preferably implemented as a peakcurrent--mode buck'-
converter which is a known prior art technique for
implementing Ii gh bandwidth switched mode power supplies.
As illustrated in Figure 3, the switched mode amplifier 22 includes a pulse width modulator (PWM) 50 which receives a control signal on line 56, and which controls a pair of switches 52a and 52b. Switch 52a is connected between a supply voltage and a cotrnon node 54, and switch 52b is connected between the common node and electrical ground. Tile supply voltage is provided by a battery1 and denoted Vbat. The pulse width modulator 50 controls the switches 52a and 52b to provide the low frequency path output to the combiner 26 in dependence on the control sIgnal. on line 56. The arrangement of a pulse width modulator in combination with a switched supply is known in the art.
The switched mode amplifier 22 includes an irner current control feedback loop and an outer voltage control feedback loop.
The inner current control feedback. loop senses the inductor current either directly or indirectly by sensing current in switch 52a or swItch 52b, and provide a feedback path 58 to a combiner 61. The combiner El combines tha feedback siqnal with a compensation ramp on line 63. lbs output of the conDiner 61 provides an Input to the inverting input of an amplifier 59.
The amplifier 59 receives at its non-inverting input an output from an amplifier 60. The amplifier 59 generates the control signal on line 56 The outer voltage control feedback loop provides a voltage feedback path 62 from the second terminal of the inductor 28b, where it connects to the inductor 28a and capacitor 28c. The feedback path provides a feedback signal to an inverting input of the amplifier 60. The amolifier 60 receives the low fxLuencv path sagncC on I we 16 at its non nvert:nq in3t Inductor 28b behaves as a current source due to the action of the inner current feedback loop provided by feedback path 58, A cornpensatior' -ramp is provided on Line 63 in this inner current feedback loop, and is used to prevent frequency halving at high duty cycles.
The outer voltage feedback loop provided by feedback path 62 is used to control the voltage at the junction of inductor 28b, inductor 28a, and canacitor 25c, The peak-current-mode buck--converter as illustrated in Figure 3 operates, in general9 as follows. rn
The low pass filter 18 generates a signal representing low frequency variation in the reference signal. This signal. on line 16 then comprises a control signal for the pulse signal for the buck switcher, comprising switches 52a and 52b which has a duty cycle determined by the control signal, such that the -voltage at the output of the buck switcher tracks the signal on line 16, i.e. the low frequency variation in the reference signal.
In addition, however, this control signal on line 15 is modified by the inner feedback control loop and the outer feedback voltage control loop.
The outer feedback voltage control loop firstly adjusts the control signal in amplifier SO. The control signal (i.e. the low frequency reference signal) has the feedback signal on feedback path 62 removed therefrom. The feedback voltage on feedback path 62 represents the voltage at the output of the low frequency paths and the rennval of this voltage from the low frequency signal on line 15 provides a signal representing the error between the outout voltage and the reference voltage The inner feedback control loop secondly adjusts the control signal in amplifier 59. The second adjusted control. signal (output-from amplifier 59) has the signal on feedback path 58 rej-rcved therefrom. The feedback signal on feedback path 58 reçresents th-e output current, The output voltage of the switch node amplifier 22 is provided by a buck switcher formed of the switches 52a, 52h connected to a battery supply voltage Vbat, The linear correction path is added to the buck switcher output, to provide high frequency correction to the low frequency switched voltager via the AC coupling capacitor 30a, As a result of combining with the correction voltage, the modulated supply is hence capable of providing short term output voltages on line 32 which are higher than the supply voltage Vbat. However the average output voltage on line 32 can be no S larger than Vbat.
There are son circumstances in which having an average output voltage which cannot exceed the supply (battery) voltage may be a problem. For example, this may be a problem when operating with a depleted battery with a low peak-to-average-power ratio (PAPR) signal, as the average output voltage may then need to be higher than the battery voltage.
Hence it is desirable for the switched mode power supply 22 to be capable of both buck and boost operation, to boost the average output voltage to a level above the battery voltage Vbat.
It is well knot'at in the art that conventional boost mode converters are difficult to stabilise on account of a right-half-plane (RHP) zero in their response characteristic. This results in such converters exhibiting a much lower closed loop bandwidth for a given switching frequency than a buck converter. Most prior art converters incorporating boost capability suffer from this disadvantage.
This arrangement addresses prior art problems by providing a voltage supply stage comprising an input supply voltage. A first and a second switch are connected in series the first and second series connected switches being connected in parallel with the input voltage source. A third switch and capacitor are connected in parallel with the first switch. A fourth switch is connected between the connection of the third switch and the capacitor and an output. A fifth switch is connected between the output and electrical ground. In a first phase ci operation, the first and fourth switches are closed, and the second, third and fifth switches are open. In a second phase of operation the second, third and fifth switches are closed, and the first and fourth switches are open. The duty cycle of operating phases is controlled such that the average voltage on the output varies between 0 volts and twice the input supply voltage. This is now described more fully with reference to the followinc Figures, Figures 4(a) and 6 (b) illustrate a switched capacitor voltage doubler cascaded with a buck output stage in which all switches are synchronously driven, in accordance with an advantageous arrangement. This embodiment shares the same control characteristics as a conventional buck converter but does not suffer from. the bandwidth limitations suffered by most boost and huck-boost converter topologies. The exemplary arrangements include a battery for providing the input voltage scrurce.
The buck output stage in Figures 4(a) and 4(b) comprises a battery 100, switches 102, 104, 106, 108, 110, and a capacitor 112, The battery 100 is connected between nodes 101 and 105.
The switch 102 is connected between nodes 101 and 103. The switch 1.04 i-connected between node 103 arid node 105. The switch 106 is connected between nodes 101 and 107. The capacitor 112 is connected between nodes 103 and 107. The switch 108 is connected between node 107 and node 111. The switch 110 is connected between node 105 and node 111. Node is connected to electrical ground. Node Ill is connected to an output line 114 on which the output voltage is generated. I)
Figure 4(a) shows the operation in a first phase (phase I) of the switching cycle, and Fig 4 (b) shows the operation in a second phase (phase 2) of the switching cycle-In the first phase of operation, as shown in Figure 4 (a) the switches 102 and 108 are closed, and the switches 104, 106 and 110 are open. The-arrow 202 denotes current flow in the arrangement of Figure 4 (a) In the second phase of operation, as shown in Figure 4 (b) the switches 104, 106 and 110 are closed, and the switches 102 and 108 are open-The arrows 204 and 206 denote current flow in the arrangement of Figure 4 (b) A controller, which is not shown in Figures 4 (a) and 4 (h) controls the switching between the first and second phases of operation. By controlling the swnching between tne firsz ann second phases of operation, and the duration for which each phase is active (Le. the duty cycle), the supply voltage can vary between zero volts and twice the battery voltageS The supply rail to the output buck switches 108, 110 at node 107 varies between voltages Vbat and 2.xvbat, hut the average output voltage of this stage can be set to any value between 0V and 2Vbat depending on the waveform duty cycle As shown in Figure 4 (c) the output voltage on line 114 comprises a pulse whicn switches cetween 0V and 4xVba1 The duty cycle of switching between the first and second phases can be varied to provide a desired average voltage between 0 volts and 2xvbat.
The topoloqy of. Figures 4 (a) and 4 (h) does not exhibit a rght-hai.t--plane zero and hence does not suffer the proh.ierns of the prior art and is capable of high closed loop bandwidth Figure 5 (a) shows an extension of the principle described with reference to Figures 4 (a) and 4 (b) to provide a two output buckboost converter capable of outputting two output voltages each having values between 0 volts and 2xvbat.
As illustrated in Figure 5 (a) * the circuit of Figures 4 (a) and 4(b) is extended to include further switches 116 and 118.
Switch 116 is connected between node 105 and a node 113÷ switch 118 is connected between nodes 107 and 113 Node 113 is connected. to an output line 115 on which a second output voltage is generated, the output voltage on line 114 now being referred to as a first output voltage.
In a buck and boost operation the circuit of Figure 5 may he controlled similar to the control of the circuit in Figures 4 (a) and 4 (h) . Figure 5 shows the switches in a first phase of operation, consistent with Fiqure 4 (a) . In a second nhase ot operation the swltcnes oL Figure. may Us switchea to tne positions shown in Fiure 4 (b) , with switch liE open and switch 116 closed. Different voltages are achieved for the first and second voltages by controlling the duty cycle of the switch oairs 108/110 and 118/116 independently A lower voltage output is produced by curtailing the pulse width of the lower voltage buck output stage.
The arrangement of the switches in Figure 5 (b) illustrates a buck only mode of operation, in which output voltage may only vary between DV and Vhat. In this mode switches 106 and 104 are permanently closed., and switch 102 is permanently open, switches 108 and 110 are toggled in first and second phases of operation to vary the duty cycle of the output waveform and achieve an average voltage between 0 volts and Vbat.
Thus if a boost operation is not required the switched capacitor doubler can be set to a fixed throuqh' mode as shown in Figure 5 (b) , with only switching between C) and Vbat occurring in tile buck output stage, thereby reducing losses associated with both stages If a peak-current--mode controi switcher is used as the switched mode amplifier 22 in the low frequency path, an exemplary implementation of which is illustrated in Figure 3, the loop dynamics are unaffected by the sudden change of supply rail voltage feeding the buck output stage, as the action of the current feedback is to trake the inductor behave as an ideal current source.
Figures 5 (c) and 5 (d) illustrate the generation of two supply voltages in buck-boost operation-As illustrated in Figure 5(c) , for the first output voltage Vouti the pulse width modulator controls the switches to maintain a high average voltage, such that in this example the first output voltage Vouti has an average value higher than at.
As illustrated in Figure 5(d), for the second output voltage Vout2 the pulse width modulator controls the switches to maintain a lower average voltage, such that in this example the second output--o].tage Vout2 has an.. average value lower than Vbat Figure 6 shows the dual-output buck-boost architecture of (a) applied in the advantageous context of an exemplary envelope tracking modulato.. To simplify the illustration, the low frecçuency path.. including the pulse width with modulator 50 for controlling the switching of the switcher is not shown in Figure 6, The reference numeral 123 denotes the boost-buck switched supply stage of Figure 5 (a) , which replaces the switches 52a, 52b of the Figure 3 arrangement. The arrow 125 denotes the control signal for the switches of the boost-buck switched S supply stage, which are provided by a pulse width modulator (such as tulse width modulator SO of Figure 3) , operating under the control of a signal representing the low frequency variation in the reference signaL A main supply is provided on the line 115 corresponding to the second output voltage in Figure 5 (a) and is used to provide the low frequency part of the modulator output..
The low frequency voltage output, or switched output voltage, on line 115 is applied to the node 54 as in Figure 3, and provides the low frequency input to the-low frequency combining element comprised of inductor 28a.
A lower power auxiliary supply is provided on line 114 corresponding to the first output voltage in Figure 5 (a) , and is used to provide the supply rail to the correction path linear output amplifier 24 through an inductor-capacitor filter arrangement provided by inductor 120 an-d capacitor 122.
This mirrors the inductor-capacitor filter arrangement provided by inductor 28b and capacitor 28c in the low frequency path -As illustrated further in Figure 6, two switch controllers are provided: a first PWM teak curren.t Trode controller 124 and a second PWM peak current mode controller 126.
With reference to Figure 6 there is illustrated an advantageous arrangement in the correction path. a feedback path for the linear amplifier 24 is taken directly from the output of the linear amplifier; rc. ther than the output of the combiner, in addition the high pass filter 20 of the Figure 3 arrangement is eliminated. As a result a full--spectrum representation of the reference signal is provided on tb-c path 14 rather than a signal with low frequency components removed, as i.n the arrangements of Figures 1 and 2 -Such an airangement offers efficiency improvements over the prior art, as it allows the peakto-peak supply voltage of the linear atrDlifi.er 24 to be minimised.
Embodiments of the invent ion are preferably implemented in such an arrangement, although the invention and embodiments are not limited to such an advantageous arrangement -The invention is advantageously applied in such an architecture.
Each-of the controllers 124 and 126 receive the low frequency reference siqnal (or enveloDe signal) as an input, such as the siyna on:me IS in F:gure 3 (oi a sgnal derd therefrom) . The first PWM peak current mode controller 124 controls the switches 118 and 116 which are used to produce the switcher output voltage on line 115, and the second PWM peak current controller 126 synchronises in frequency and phase with the first controller and controls the switches lOB and. 110 which are used to produce the voltage supply for the linear amplifier on line 114. Thus each of PWM peak mode controllers 124 and 126 is shown to provide general control signals l25a and 125h, which form part of the control signals 125 to the switched suoply stage 123.
Voltage doubler switches are controlled by the PWM waveform of the first or second controller, whichever has the larger duty cyci e, to ensure the input to both h.alf-bridcte stages (switches 108, 110 and 118, 1]. 6) is 2vbat when switches 108 or 1.:L8 are made (closed) . Equivalently, the PWM waveform controlling switches 102, 104 and 106 is a loqical OR' function of the PWM waveforms of controllers 1 and 2 (i.e. conL-roilers 124 and 126) The main output supply on line 115 is modulated, whereas the auxiliary output supply -namely the supply voltage to the linear amplifier on line 200 -may hea fixed voltage, or a voltage which is set according to the average the average power of the RF signal on a slot--by--slot basis in a communication system which is time---slot based.
Activation., of the boost mode to increase the output voltage to up to double the battery voltage can be controlled directly by a baseband controller, for example on a slot-by-slot basis, depending, for example, on any one or combination of the HF power level, the peak--to--average power ratio, and the battery voltage in a time-slot. The baseband controller can. control th.e PWM peak current mode controllers 124 and 126.
With reference to the linear amplifier feedback arrangement of earlier Figures, the signal at the output of the linear amplifier 24. in the high frequency path is not a full-spectrum signal because it does not contain any:10w frequency components. As a consequence the peak--to-peak amplitude of the signal-at this point is greater ti-an the-peak-to-peak amnlitude would be if the full spectrum of the envelope signal were present This reduces the efficiency of the linear amplifier 24, as its supply rails must be set to allow linear amplification of this larger peak-to-peak signal.
In an alternative improved arrangement as shown in Figure 6, the feedback path is taken from the output of the linear amplifier 24 itself rather than the output of the continer, and tFrus provides a signal which has a f till spectrum envelope signal. Hence the feedback signal has lower peak-to-peak amplitude than the signal at the output of the linear amplifier in the prior art linear amplifier feedback arrangement of earlier Figures.
With reference to Figure 6 there is illustrated an advantageous arrangement in which a feedback path for the linear amplifier is taken directly from the output. of the linear amplifier rather than the output of the combiner.
Alternatively to Figure 6, autonomous control of the boost setting may be possible by comparing the switcher output voltage or a scaled version of the input reference voltage, for example the signal on line 15 of Figure 3, with a t.hreshoJ..d voltage which may be defined as a percentage of the current battery voltage as shown in Figure 7. This reduces the firmware burden on the baseband controllerS With reference to Figure 7, a comparator 128 is introduced which generates a control signal on line 130 for enabling/disabling the voltage doubling circuitry provided by the switches 104, 102, L 06 and the capacitor 112, generally denoted by ref e-rence numeral 132.
Tne comparator 128 is arrangea to compare the outpuc voltage at the switched output, detected at the node at the junction o.f inductors 28a and 28b and provided as a first input to the comparator 128, with a th.rcshold value at the second input to the comparator 128. The threshoi.d voltage is provided at the junction of resistors 134 and 136, the other terminal of resistor ±34 conneotea to Vbat, ano the otner terminal of resistor hG connected to electrica± ground.
In Figure 7it is shown that the comparator 128 compares a threshold voltage to the output voltage of the low frequency path. However the output voltage of the low freqLency path is derived from the low frecuency cart of the reference voltage f or example the signal on line 16 of Figure 3, and the threshold voltage may be compared to any signa.l which is derived from the low frequency part of the reference voltage.
In the arrangement of Figure 7, the threshold voltage may be compared to the signal on line 16 of Figure 3, rather than the output of the low f reqtency path, for example If the voltage doubling circuitry 132 is disabled, the output voltages are generated by the respective output stages comprising switched pairs 108/110 and 116/118 as conventional buck stages. This allows the respective output voltages to switch between OV and Vbat. When enabled1 the voltage doubt,: rig circuitry 132 allows the respective output voltages to switch between 017 and 2xVbat.
In dependence on the comparison in the comparator 128, the voltage doubler circuitry 132 is enabled or disabled.
in many applications, envelope tracking only offers an efficiency improvement at high power amplifier operating powers-Typically, if the power amplifier average output power is less than one--tenth of its maximum value, it i.e more efficient to operate in average power tracking (APT) mode, in which the modulator output voltage is held at a constant value over each slot, but adapted according to the average power level of each slot.
The term slot refers to a time-slot as defined in 3G and 4G (e.g. LTE) standards.
Figure 7 further shows h-ow the envelope tracking architecture of Figure 6 may be re--configured as an APT power supply in which only the main switcher of the low frequency path is active.
The linear amplifier 24 is disabled for example by opening switch 108 to disconnect its power supply. A switch 140 is used to connect the capacit*or 30a of thei nductor-capacitor coirbiner to electrical ground, to provide a well-decoupled 2! output for feeding the power amplifier load. In this way the power dissipation associated with the quiescent current of the linear amplifier 24 is saved.
When the high frequency correction path including the linear amplifier is disabled, switch 140 is turned ton' and the capacitor 30a serves to provide additional de-coupl.ing No additional circuitry is required other than the switch 140 and its associated control.
When the high frequency correction path is disabled, two inductor-capacitOr sections, capacitor 28c and inductor 28b, and capacitor 30a and inductor 28a orovide 4 order filtering of the sw.i tched mode suoply output. Hence there is nrovided significantly greater switcher ripple rejection than a conventicnnl 2 oraer tlter, This is partcularly neneficiai for discontinuous mode operation of the PWN controller, as the low frequency ripple may be higher than in continuous mode operat. ion.
The increased output filter rejection provided by the two inductorcapacitor sections when the high frequency path is disabled as discussed above makes it possible to operate in a discontinuous conduction mode (Dafl, such as pulse frequency node (PFM) or burst mode ooerat.ion, without exceedinq stringent requirements for maximum voltage ripple. This reduces switching losses and enables high efficiency operation to he maintained at much lower output powers than are possible using PWM-only solutions.
Conventional APT solutions with only one inductor-capacitor section, such as provided by capacitor 28c and inductor 28h, typicall.y cannot achieve sufficiently low rinple i.n PG-I and so may he recuired to implement an additional low power mode in which a switch is used to connect the power amplifier directly to the battery, bypassing the PWM converter, but sacrificing efficiency When operating in 2G (GSM/EDGE) mode, a physically large inductor may need to be provided at the output of: the low frequency path to allow for the potentially large currents which are required in this mace. Currents as high as 2..SA, for example, may need to be handled by the low frequency path.
The presence of potentially large currents such as this dictates that the indxtctors at the output of the PWM 50 need to be large to handle such currents Figure 8 illustrates a modified arrangement which supports both 2G and 3G/4G modes in accordance with embodiments of the invent. ion, but which does not require the low frequency path to handle large currents and thereby implement a large inductor in accordance with a further embodiment of the invent ion.
With reference to Figure 8 port ions of earlier drawings are shown, and reference nume.x'a le corresponding to earlier drawings are shown where appropriate. Only those portions of earlier drawings are shown which are required to implement this embodiment.
As sho%m in the embodiment, an output power amplifier 100 is provided having an RF input on line 102 and an RF output on line 104. The RF power amplifier 100 receives a modulated supply voltage on line 106.
In accordance with the foregoinq arrangements, the modulated power sunplv on line 106 is provided either by the low frequency oath 11, or by the low freauency path in combination with the high frequency path When the power supply is provided by only the low frequency path, the switch 140 is closed so that the output of the linear amplifier 24 is connected to ground and the linear amplifier 24 is disabled. fin
Further in accordance with this embodiment, the power supply on line 106 may also be orovided directly from the supply voltage, by use of the switch 105.
The power amplifl2w:00 ptovicns owet amplificato:. in ay one OL 2G, 3G. or 4G tmodes of operation. As discussed anove, in a 2G tticde of operation at medium and low output powers in sthich the linear amplifier 24 is disabled and switch 140 is closed, the inductors in the low frequency path are reiired to handle the currents necessary for the. voltage supply for the 2G operation.
As 2G operation may require the handling of large currents at high output power, the inductors at the output of the low frecuency path need to be capable of handling large currents.
However, in accordance with this embodiment, at high outut power the low frequency path can he further bypassed, and the power supply to the amplifier 100 can he provided directly from the supply voltage V, by closing -witch 105.
At low or medium power, the arrangement of Figure 3 operates in APT mode as described hereinabove, Low or medium power may be defined according to the maximum current ratings of the inductor in the low frequency path, and therefore may be implementation dependent. The supply to the amplifier 100 is provided purely by the low frequency path.
At high frequency powers the arrangement of Figure 3 may operate in accordance with the preferred embodiment, and is switched to a. battery bypass mode in which the power supply for the amplifier 100 is provided directly from the sunply voitage Vb by nosing swLch 105.
Thus the arrangement of Figure 8 may operate i.n one of three modes, namely envelope tracking (ET) * in which low frequency and high frequency paths are enabled, average power tracking (APT) in which the high frequency path is disabled and switch is closed, and bypass mode, in which the high frequency path is disabled and switch 140 is closed, and also switch 105 is closed.
The invention and its embodiments relates to the application of envelope tracking (ET) to radio frequency (RF) power amplifiers, and is applicable to a broad range of implementations including cellular handsets, wireless infrastructure, and military power amplifier applications at high frequencies to microwave frequencies.
tO The invention has been described herein by way of example with reference to embodiments. The invention is not limited to the described embodiments, nor to specific combinations of features in embodiments. Modifications may be made to the embodiments within the scope of the invention. The scope of the invention is defined by the appended claims.

Claims (20)

  1. CLATJ'1S: I. An envelope tracking modulated supply for providing a supply voltage tracking a reference signal, and comprising a low frequency path for tracking low frequency variations in the reference signal. and a high frequency path for tracking high frequency variations in the reference signal, wherein a power amplifier selectively receives a voltage supply based on either the low frequency path alone or the corrination of the low frequency path and the high frequency path.
  2. 2. The envelope tracking powers upply of claim 1 wherein there is provided a combining element for providing the power supply by combining the low frequency and high frequency paths, comprising a high frequency blocking element or elements for th.e low frequency path and a low frequency blocking element or elements for the high frequency path.
  3. 3. The envelope tracking modulated supply of claim 1 or claim 2 wherein when the power amplifier receives a voltage suoply based on the low frecuency path the connection between the high frequency path and the low frequency blocking element is connected to electrical ground.
  4. 4. The envelope tracking modulated supply of claim 3 wherei.n the low frequency path is disabled w1en the connection between the high frequency path and the low frequency blocking element is connected to electrical ground.
  5. 5. The envelope tracked power supply of any one of claims 2 to 4 wherein the iow frequency block ng element is a capacitor.
  6. 6. The envelope tracked power supply of any one of claims 2 to 5 wherein the wherein the high frequency blocking element is an inductor.
  7. 7. The envelope tracked supply of any one of claims 1 to 6 wherein the high frequency path is disabled in an average power tracking mode of operation.
  8. 8. The envelope tracked supply of any one of claims 1 to 7, wherein when the high frequency path is disabled the power in the low frequency path is determined.
  9. 9. The envelope tracked power supply of claim 7 wherein the power amplifier is connected to the additional DC supply when it is determined that the average supply power exceeds a threshold, when the supply voltage is based on the low frequency path alone.
  10. 10. The envelope tracked power supply of claim B wherein if the power in the low frequency path is below a threshold, the supply to the power amplifier is not connected to the additional DC supply voltage.
  11. 11. The envelope tracked supply of claims 2. to 6 wherein the low frequency path comprises a switched mode power supply for generating an output voltage tracking the reference signal and a supply voltage for a linear amplifier in the high frequency path.
  12. 12. The envelope tracking modulated supply according to claim 7 wherein the supply voltage is enabled when the high frequency path is enabled.
  13. 13. The envelope tracking modulated supply of any preceding claim wherein there is provided a feedback path from the output of the linear amplifier to the input of the linear amplifier, such that the linear amplifier in the correction path amplifies a signal comprising the full spectrum of the frequencies in the reference signal.
  14. 14. An 1W amplifier including a voltage supply stage of any one of claims 1 to 13. I0
  15. 15. A wireless communication system including a voltage supply stage of any one of claims 1 to 13.
  16. 16. A wireless mobile device including a voltage supply stage of any one of claims 1 to 13.
  17. 17. A method in an envelope tracking modulated supply for providing a supply voltage tracking a reference signal, and comprising a low frequency path for tracking low frequency variations in the reference signal and a high frequency path for tracking high frequency variations in the reference signal, wherein the method comprises providing a voltage supply based on either the low frequency path alone or the combination of the low frequency path and the high frequency path.
  18. 18. The method according to claim 14 further comprises connecting the connection between the combiner and the output of the high frequency path to electrical ground when the high frequency path is disabled.
  19. 19. The method of claim 18 further comprising disAbling the high frequency path in an average power tracking mode of operation.
  20. 20. The method of claims 17 to 19 further comprising determining the power in the low frequency path when the high frequency path is disabled, and selectively basing a voltage supply on either the low frequency path alone or a DC supply.
GB1301853.6A 2013-02-01 2013-02-01 Envelope tracking power supply with low power modes Withdrawn GB2510394A (en)

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PCT/EP2014/051964 WO2014118344A2 (en) 2013-02-01 2014-01-31 Low power modes for 3g/4g envelope tracking modulator

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