GB2502992A

GB2502992A - A constant current switched mode power supply controller

Info

Publication number: GB2502992A
Application number: GB1210378.4A
Authority: GB
Inventors: Conor Mcauliffe; Paul Grogan; Marcus Oduinn
Original assignee: IKON SEMICONDUCTOR Ltd
Current assignee: IKON SEMICONDUCTOR Ltd
Priority date: 2012-06-12
Filing date: 2012-06-12
Publication date: 2013-12-18
Anticipated expiration: 2032-06-12
Also published as: GB201210378D0; GB2502992B; WO2013186227A3; WO2013186227A2

Abstract

A method of controlling a mains powered switched mode power supply e.g. used for driving an LED array with a constant current, in which a constant current converter is controlled using estimates of output current from the secondary side determined from measurements performed on the primary side. The method of control operates during a current half mains cycle based on an average of the estimates of output current determined during a previous one or more half mains cycles.

Description

TITLE

A CONSTANT CURRENT SWITCHED MODE POWER SUPPLY CONTROLLER

Field of the application

The present application relates generally to power conversion and more specifically to Switched Mode Power Supply (SMPS) controllers and the methods used therein.

Background

A switched mode power supply (SMPS) is an electronic power supply that incorporates switching and energy storage elements so as to achieve efficient conversion of electrical power from a power source to a load. A SMPS may convert DC-DC, AC-DC or DC-AC. There are three basic configurations or topologies of dual switch, single storage element SMPS; Buck, Boost and Buck-Boost as illustrated in Figure 1. In each case the power delivered to the load is controlled by the duty cycle of the control signal applied to the control terminal of the active switching elements (i.e. the gate of the MOSFET in Figure 1 (b)). In each case, a first (input) switch when switched on, transfers energy from an input source to an energy storage device (e.g. an inductor). The first switch may be a MOSFET or similar semiconductor switching device. When the first switch is turned off, a second switch is employed to transfer the energy stored in the inductor to the load. In its simplest form, the second switch is a diode or similar semiconductor device. It will be appreciated that in the exemplary circuits illustrated in Figure 1(b) a capacitor is shown as part of the load however this is for convenience of explanation and generally the capacitor is part of the SMPS.

Broadly speaking a SMPS operates so as to transfer energy from an input power source to an output load via an energy storage element. This is achieved through the operation of the switching elements so that during the first portion of the switching cycle, energy is transferred from the input source to the energy storage element and during the second portion of the switching cycle energy is transferred from the energy storage element to the output load. The ratio of the first portion of the switching cycle to the total switching cycle is referred to as the duty cycle of the SMPS. The power delivered to the load is controlled by the duty cycle.

A SMPS may operate in one of three different modes; namely DCM (Discontinuous Conduction Mode), BCM (Boundary Conduction Mode) and CCM (Continuous Conduction Mode). In DCM, the energy storage element is reset before the end of each switching cycle. In CCM, the energy storage element is storing some energy (non zero value) at the end/start of each switching cycle. In BCM, the energy storage element is emptied (reset to zero) precisely at the end of each switching cycle. i.e. the SMPS is operating at the boundary between DCM and CCM.

An SMPS controller is a device whose purpose is to control the output quantity (typically voltage or current) delivered by the SMPS to the load by observing various quantities (typically voltages or currents) within the SMPS which may or may not include the output quantities themselves and adjusting the on and off times of one or more switches within the SMPS according to the desired mode of operation (CCM, BCM or DCM) of the SMPS.

As well as providing control for the SMPS, the controller itself might also implement other ancillary functions such as, but not limited to; controlled start-up, fault protection (over voltage, over current, over temperature), standby and sleep modes as well as any other functionality that may be required for the specific application in which the power supply is being used.

An offline AC-DC SMPS is an electronic power supply that converts an incoming AC supply voltage into a DC output. An AC-DC SMPS typically incorporates a multiplicity of switching and energy storage elements configured so as to achieve efficient conversion of electrical power from the AC mains voltage supply to a load. Generally, the AC voltage is first converted by means of a rectifier or similar circuit to a rectified form. The rectified AC voltage is used as the input to the switching stage of the SMPS.

The rectifier very often is a diode bridge rectifier. A bulk capacitor is typically used to smooth the rectifier output before the SMPS. If the bulk capacitor is sufficiently large, the input to the SMPS is a pseudo-DC voltage with a ripple voltage present at twice the mains frequency. The amount of ripple depends both on the size of the bulk capacitor and the power drawn by the SMPS. In some cases it is desirable to have a small value of bulk capacitance at the output of the bridge rectifier so that the fully or near fully rectified AC mains voltage appears at the input to the SM PS.

Figure 2 below outlines a typical arrangement for an offline AC-DC SMPS which includes a diode bridge rectifier, a bulk capacitor for smoothing the rectified mains. The rectified mains is provided as an input voltage to the SMPS which in turn converts it for delivery to a load.

In offline AC-DC SMPS converter applications, for safety reasons it is often required to have isolation between the AC mains and the load of the SMPS converter. This may be achieved in an SMPS using a transformer as the energy storage device. In lower power applications, one of the most utilised isolated SMPS converter topologies is the Flyback converter, shown in Figure 3 which is essentially an isolated version of the Buck-Boost topology. A flyback transformer achieves the required isolation between the primary and secondary sides; the transformer also allows for voltage scaling by appropriate selection of the turns ratio of the transformer, which is the ratio between the number of turns on the primary winding and the number of turns on the secondary winding (N:N).

Additional circuitry may be included within an offline AC-DC SMPS converter for purposes other than that of actual power conversion. Thus for example, in the exemplary arrangement of Figure 4, an EMI filter is provided to limit the transfer of switching noise onto the mains. Similarly, protection circuits or devices may be provided, for example, to protect against over-voltage (surge protection) and over-current protection.

Typically, one or either or both the EMI filter and protection devices are placed at the front end before the diode bridge rectifier in order to meet the various regulatory compliance standards required when interfacing a circuit to the AC mains supply.

As with the general offline SMPS converters described above offline SMPS converters generally include an EMI filter and some form of over-voltage/over-current protection placed at the front end, followed by a diode bridge rectifier and bulk capacitor after which is some form of SMPS converter (typically isolated), controlled so as to deliver a constant voltage or current to the load which may also have some form of filtering across it(possibly in the form of one or multiple capacitors in parallel with the load).

One difficulty with offline SMPS control is that since isolation is generally required, the observation and feedback of secondary side quantities to the primary side are more complex due to the requirement to maintain the isolation barrier. One solution is to employ an isolated feedback device

S

such as either an optical, capacitive or inductive based couplers circuits to provide isolated feedback to the controller of measurements from the secondary side, for example output voltage, output current or both.

The use of isolated feedback devices is not ideal. For example, opto-couplers are a known weakness in isolated SMPS systems as they age badly; especially at higher temperatures, thus leading to a degradation of performance and accuracy over time as well as reducing the useful lifetime of the system. They also complicate the system stability due to the addition of extra poles and zeros in the overall control loop of the SMPS which make designing systems that use opto-coupler circuits more complicated, more costly and physically larger. Similarly, inductive coupling may be used in place of opto-couplers and whilst more reliable and accurate are costly in terms of area/volume and can also suffer significantly from interference.

Accordingly, to avoid the need for isolated feedback devices and other reasons, some switched mode power supply configurations have emerged that use what is termed primary side regulation. In primary side regulation, only quantities that are available on the primary side are measured. From these measurements, an inference or estimate of the output quantities is made. Primary side regulation removes the requirement for an opto-coupler to feed back secondary side quantities across the isolation barrier to the primary side while still maintaining the galvanic isolation. A disadvantage of these systems is that they may require complex manipulations of the primary side quantities (resulting in a commensurately complex implementation) and/or rely on a specific mode of operation (DCM) to work properly as they generally require that the transformer is reset on each cycle (de-energised). Moreover, they can be inaccurate in estimating the output quantities due to the complexity of the calculations required leading to a wide variability in performance between individual realisations of the same implementation.

One use for an offline SMPS is try provide drive currents to LED lighting. It is estimated that around 19% of total worldwide energy production is used for lighting. Given the ever increasing cost of energy and the desire to reduce harmful emissions of greenhouse gases there is a general movement away from incandescent lighting to more efficient sources of light of which LED's are one example.

Typically, at the heart of a LED based luminaire is a HB LED (High Brightness LED]. These HB LED's require a constant current drive to operate. The input power source (typically AC mains or DC supply bus] needs to be converted to a precise DC current that can support a range of output voltages dependant on the exact forward voltage (VP) of the LED and the number of LEDs that are driven in series. It will be appreciated that this differs from what is normally required of efficient Switched Mode Power Supplies (SMPS] which are generally designed to supply an accurate DC voltage across a wide range of load currents with good transient response.

One of the advantages of LEDs as a source of illumination is their relatively long lifetime (typically 50,000 -100,000 hours is reported for the LEDs themselves although this is heavily dependent on the operating junction temperature of the LED). When taken in the context of a LED light bulb or luminaire, this figure reduces (typically 25,000 -50,000 hours) due to the thermal conditions under which the light bulb or luminaire operates although it is still significantly better in comparison to incandescent lights (typically 750 -1,000 hours) or CFL (typically 5,000 -15,000 hours) lights.

Indeed one of the key issues with LED lighting is insuring the operating temperature of both the LEDs and the drive electronics components are kept as low as possible to insure maximum lifetime as all these components suffer lifetime degradation at elevated temperatures. This is important in minimising the total cost of ownership of LED based luminaires to insure they are an economically viable replacement technology as LED lighting typically has a higher up-front cost associated. Accordingly to be economically viable, they must make up the difference in up-front costs by lower running costs and less frequent replacements due to longer lifetime.

The SMPS selected for a LED based light may take many forms where various performance metrics are prioritised over others depending on the end goals of the design. One typical issue is to design the SMPS so that it adequately meets desirable characteristics (design requirements) without overly sacrificing any one characteristic over another.

Characteristics that might be considered include but are not limited to a) cost bJ regulation of output current c) Power Factor Correction (PFC) dJ dimmer compatibility e) Reliability & lifetime, and 1) Isolation A number of methods exist for the driving of a constant output current for applications such as LED lighting. It will be appreciated that certain solutions may be goods at ensuring some characteristics are met but have deficiencies in others.

These will now briefly be discussed with reference to four configurations commonly employed, namely Secondary Side Constant Current Converters, Secondary Side Active PFC converters, Constant power converters and Primary Side Constant Current Control converters.

In Secondary Side Constant Current Converters, a measurement is made of the output current, for example using a sense resistor, on the secondary side. The sensed output current or more generally the difference from a desired set point is fed back to a controller operating on the primary side typically using an opto-coupler. Constant current converters provide excellent output current regulation. However, opto-couplers are a known reliability hazard, add to overall cost and add to the complexity of loop stabilisation. Additionally, constant current converters using an opto-coupler isolated feedback generally require a large bulk capacitance as they operate from a heavily filtered rectified AC-mains input (pseudo DC input) as their main care-about is maintaining accurate output current regulation.

If the mains is not filtered sufficiently, excessive output current ripple at twice the AC-mains frequency can appear on the secondary side which may necessitate very large output capacitors for smoothing. Furthermore, TRIAC based leading edge dimmers are difficult to support as these generally will not work with the type of capacitive load presented by the front end of the converter which has a large bulk capacitor after the diode bridge to provide a pseudo dc bulk voltage.

Because the bulk capacitors employed tend to be large, the capacitor tends to be peak charged at the peak of the AC-mains voltage which results in a large harmonic current content drawn from the AC-mains which in turn results in relatively poor PFC.

A second configuration employed is that of a Secondary Side Active PFC converter. In this configuration, as with Secondary Side Constant Current converters a measurement is made of the output current, for example using a sense resistor, on the secondary side, which in turn is fed back to a controller operating on the primary side typically using an opto-coupler.

However, the controller operating on the primary side also employs a measurement of the primary side supply voltage in an effort to ensure that the current drawn from the supply is generally reflective of the point in the cycle of the supply voltage rather than at the peak of the cycle.

Unfortunately, PFC converters are generally more complex and in a single stage converter lead to a high output current ripple at twice the AC-mains frequency. Such a ripple is highly undesirable in offline LED lighting as it can result in visible flicker. To alleviate the problem of low frequency high output current ripple, large electrolytic capacitors are typically employed on the output side. These capacitors are notoriously unreliable and present problems because of their size in terms of fitting a complete solution into the form factor of the bulb. Additionally, this type of converter is not inherently dimmer compatible as the control loop of the controller is driven by the error signal fed back by the opto-isolator and as such will try to maintain the desired output current set-point irrespective of the input voltage.

A further configuration is that of a primary side constant power controller using fixed peak current. In this configuration, the output current is determined by the output power divided by the output voltage. The configuration is implemented by measuring the current on the primary side typically using a sense resistor and switching off the primary switch (MOSFET) when the current reaches a set point. However, as the output power is estimated based on a primary side measurement and is thus operating open ioop, design tolerances may result in significant variations in actual output from that predicted. Moreover, it will be appreciated that this configuration has the inherent limitation of only being able to indirectly control the output current as the exact operating point on the constant power current-voltage curve is dependent on the LED employed which in turn has an exponential IV characteristic. For any given converter, the output current can show significant variation depending on the LED.

Even with LED's selected from the same wafer there can be significant variations. As a result of these factors, constant power converters while simple to implement and resulting in a low cost converter, also have a relatively wide spread in output current that is not desirable. Additionally, the output will also vary with the AC-mains voltage level.

Furthermore, TRIAC based leading edge dimmers are difficult to support as these generally will not work with the type of capacitive load presented by the front end of the converter which has a large bulk capacitor after the diode bridge to present a pseudo dc bulk voltage. Additionally, because the bulk capacitor is large it tends to be peak charged at the peak of the AC-mains voltage which results in a large harmonic current content drawn from the AC-mains which leads to poor PFC.

A further configuration employed is that in Primary side constant current control converters an example of which is illustrated in Figure 6, in which a flyback topology converter is shown in which an input AC voltage (VAC) is rectified to a pseudo DC voltage V1-ii<. This pseudo DC voltage is switched through a transformer TRX1 and thus converted to an output voltage (V0J and current The exemplary converter shown comprises a transformer (TRX1J with primary and secondary windings. The primary winding is switchably connected by a switch SW1, which is typically a MOSFET or similar semiconductor switch, to the input voltage Vbulk.

Depending on whether the switch is "ON" or not the input voltage is effectively connected or disconnected across the primary winding. The switch SW1 in turn is controlled by a Pulse Width Modulator (PWM) circuit.

In the case where the switch is a MOSFET, the PWM provides a drive signal to the gate of the MOSFET. A current sensing circuit is employed to provide a measure 1seIse of the primary (input) current pri. The current sensing circuit may for example comprise a small sense resistor in series with the switch (Swi). The voltage across the sense resistor may be measured to provide a measure 1seIse of primary current. The secondary winding provides a secondary output (Vsec, see) which is connected through a switch suitably a diode (Di) to the load, which may for example be one or more LEDs (LED1-II). A capacitor (Ci) may be provided to smooth the output to the load. A sense (auxiliary) winding is provided on the primary side of transformer. The voltage on the sense winding (Va) provides an indication Vsense of the state of the secondary winding, i.e. whether it is energised or not. The dotted lines represent the isolation harrier provided by the transformer (TRX1). A less desirable alternative to using an auxiliary winding is to measure the voltage at a point between the switch and the primary winding.

In this configuration of Figure 6, the auxiliary winding on the primary side is used to provide a measure (V55) of the secondary side voltage either directly or as shown by means of a voltage divider comprising two resistors. The current on the primary side is also measured for example using a sense resistor. These two measurements (Vsense, Isense) are then employed in an estimator which is typically integrated into a controller IC to provide a estimate of OUT. The controller compares the estimate with a desired set point (IouT,sET). Based on this comparison, the controller adjusts the operating point of a pulse width modulation circuit which in turn is used to control the primary side switch SW1. Using the estimate of output current eliminates the requirement for an opto-isolator to feed back a measurement of the output current In this respect it is similar in operation to the secondary side constant current converter with the same limitations apart from the requirement for an opto-isolator. An advantage of having an auxiliary winding is that it may also be employed to provide a supply voltage to the controller IC using a diode and a capacitor.

Most converters employ a large bulk capacitor after the diode bridge to provide a pseudo dc bulk voltage. These large bulk capacitors tend to be large electrolytic capacitors. Unfortunately, electrolytic capacitors are typically unreliable and prone to failure. Additionally, they occupy considerable space in a circuit. It will be understood by those skilled in the art that the purpose of a bulk capacitor is to hold charge throughout a cycle so that the voltage supplied to the switching circuit appears as DC voltage with a ripple. It is also known to employ a filter capacitor at the input to the switching circuit. The function of this filter capacitor, alone or in combination with other components, is to prevent high frequency switching noise passing to the mains. Such filter capacitors are not electrolytic capacitors.

Additionally, as the bulk capacitor is large it tends to be peak charged at the peak of the AC-mains voltage which results in a large harmonic current content drawn from the AC-mains. This large harmonic current means that such converters have generally poor PFC performance.

Accordingly, there is a need for an alternative method of primary side control.

Summary

The present application employs a method of control which does not require the use of a bulk capacitor. The method is based on providing a constant current converter. The constant current converter is controlled using estimates of output current from the secondary side determined from measurements performed on the primary side. The method of control operates during a current half mains cycle based on an average of the estimates of output current determined during a previous one or more half mains cycles.

Accordingly, the present application provides a controller, a switch mode power supply, method and lighting device in accordance with the claims which follow.

Description of Drawings

The present application will now be described with reference to the drawings in which: The present application will now be described with reference to the drawings in which: Figure 1 illustrates exemplary SMPS circuit topologies known in the art, specifically an example of a Buck, a Boost and the Buck-Boost topologies; Figure 2 is a schematic representation of an offline AC-DC SMPS known in the art; Figure 3 is an exemplary schematic illustrating a Flyback converter topology known in the art; Figure 4 is an exemplary schematic illustrating a Flyback converter based offline AC-DC SMPS converter known in the art; Figure 5 is an exemplary constant power control scheme known in the art;; Figure 6 is an exemplary constant current converter known in the art with primary side sensing and control; Figure 7 is an exemplary arrangement according to a first aspect of the present application; Figure 8 is an exemplary controller for use in the arrangement of Figure 7; Figure 9 is another exemplary controller for use in the arrangement of Figure 7; Figure 10 is the exemplary controller of Figure 8 with dimmer functionality implemented; Figure 11 is the exemplary controller of Figure 8 with active PFC implemented; Figure 12 are results obtained from the arrangement of Figure lii illustrating the resulting performance; Figure 13 is a graph illustrating the PFC and output ripple performance of the present application; Figure 14 is a luminaire in which the arrangement of the present application may be employed; and Figure 15 is an equivalent circuit diagram for the luminaire of Figure 14.

Detailed Description

The present application provides a SMPS controller suitable for controlling a mains powered switch mode power supply, which will now be described with reference to Figure 7. The SMPS controller 70 is suitably implemented as a single integrated circuit. Although it will be appreciated that certain features may be implemented off-chip. Equally, the entire circuit may be implemented using a plurality of discrete components. Similarly, it will be understood that the circuits may be implemented on or off chip both digitally and in analog form with appropriate conversion of signals/measurements between analog and digital form using ADC's and DAC's as required. For ease of illustration, Figure 7 omits showing ADC's and DAC's.

As with the prior art generally, the power supply comprises a rectifier circuit for rectifying the mains supply to provide a rectified mains supply.

Suitably, the rectifier circuit is a full wave rectifier, although less desirably half wave rectification may be employed.

In contrast to the prior art, the present application does not require a bulk (smoothing) capacitor on the rectified input supply. Accordingly problems associated with using large bulk electrolytic capacitors are obviated. As with the prior art, a filter capacitor may be employed on the output of the rectifier circuit. The purpose of such a filter capacitor is to reduce noise and thus improve the converters EMI performance. Whilst, the specific value of the filter capacitor will vary based on the design of the power supply, the filter capacitor will suitably have a value less than 300nF and typically about lOOnF. The rectified mains is switchably connected by a switch Swi to the primary side of a transformer. The output on the secondary side of the transformer is provided through a conventional diode and capacitor arrangement to a load, which may for example be one or more LEDs.

The switch Swi is operated by a switching circuit 72 which in turn is controlled by a controller 76. The switching circuit suitably comprises a PWM circuit. In the exemplary switching circuit shown, the controller provides a desired setting (VSETISENSE) for primary side reference level to provide a desired output current on the secondary side. In the exemplary switching circuit, the switching circuit is switched at a switching frequency F. This switching frequency may be generated from an on-board clock.

Whilst the switching frequency may remain constant, it may also be variable. Thus for example a jitter circuit (not shown) may be included to cause the switching frequency to change. The use of such a jitter circuit improves the EMI performance of the switching circuit by spreading the switching frequency out over a period of time during which EMI measurements are made.

In one exemplary arrangement, the PWM circuit comprises a comparator comparing the value of Iseiise with the value of Vstseise. The output from the comparator is provided to a flip flop, whose output is reset upon the current reaching the required value. The flip flop is set at the start of switching cycle determined from the switching frequency signal F. The output of the flip flop is used as a switching cycle for switch SW1.

Suitably, the measurement of primary current is provided by means of a sense resistor R1 as would be familiar to those skilled in the art.

The controller 76 determines the desired value of primary side reference level VSETISENSE based on an estimate of secondary side current determined by an estimator 74. The estimator relies upon primary side measurements to provide the estimate of secondary side current. More specifically, the estimator uses values of T011 (secondary side on' time), in combination with the transformer ratio (Np/NS) of the transformer to determine an estimate of the output current during a switching cycle.

More specifically, the output current on the secondary side of the transformer I for an individual switching cycle may be taken generally to be proportional to VSET-ISENSE (Toff/ Ts) (1) Where VSETISENSE is the setting of primary side current and Ts is the duration of the switching cycle and the proportionality is dependent on the transformer turns ratio Np/NS and the value of the current sense resistor Ri.

This equation will be familiar to those skilled in the art as being applicable to discontinuous mode of operation but it will also be appreciated that a corresponding equation exists and may be used for continuous operation.

It will be appreciated that the values of VSETISENSE and Ts are known or at least measurable on the primary side. T0nis determined from a primary side measurement representative of the secondary side and in the exemplary circuit is determined from a measurement Vseiise made from an auxiliary winding on the primary side. A zero crossing detector (not shown) or similar circuit may used to determine the start and end points of secondary side conduction. A timer or similar circuit may be used to measure the duration between the start and end points. An alternative but less preferred technique to using an auxiliary winding to determine the secondary side conduction time is to employ a measure of the drain voltage of Switch Swi.

The estimator suitably provides an estimate of the output current after each switching cycle. These estimates are produced on a continuous basis.

The estimates of output current are provided to the controller. The controller employs these estimates on an averaged basis to determine a required setting for primary current VSETISENSE. More specifically, the controller may employ an average value of the estimates of output current obtained over a half mains cycle. Thus for example, whilst the converter may be operating at an exemplary switching frequency of 100KHz, or 100,000 switch cycles. The controller operates on the averaged basis of 1000 of these cycles that occur during a half mains cycle (where mains is 50Hz). It will be understood that a measurement need not be obtained\employed by the controller for each switching cycle for the averaging operation to be effective. The purpose of the averaging is to obtain an average measurement over a half mains cycle. It will be appreciated that a representative average may be obtained without using a value for each individual switching cycle during a half main cycle. Thus for example, the value for each second switching cycle could be ignored or not calculated and a reasonable average for the half mains cycle still obtained.

Equally, the averaging is not restricted to a single half mains cycle and may be extended to more than one half mains cycle. It will be appreciated that the averaging may readily be implemented digitally as a mathematical function.

It will be appreciated that a zero crossing or similar circuit (not shown) may be employed on the mains input to determine the start and end of each mains cycle.

Less desirably, a low pass filter having a bandwidth less than that of twice mains frequency (for full wave rectification) may also be used.

The average value of output current obtained over at least one half mains cycle is used by the controller in controlling the switching circuit over a subsequent half mains cycle.

Thus, for example with reference to Figure 8, once a value for the average output current has been estimated this value is compared by a comparator 82 to a desired output current (set-point IouT.SET) to generate an error value representing the difference. This error value may then be applied as an input to a compensator 84 (e.g. a PID compensator) whose output in turn is used to set the VSETISENSE value for the next averaging time-period (e.g. a half mains cycle). As the averaging function is applied over a pre-defined period of time (e.g. half mains cycle) the value of VSET.JSENSE remains constant over the same pre-defined period of time. Whilst a half mains cycle would be a convenient time-period the controller is not restricted to this.

It will be appreciated that the averaging need not necessarily be implemented before the comparator 82. Equally, the averaging may be performed by an average 80 on the output of the comparator as shown in Figure 9. Moreover, it will be appreciated that averager may be integrated with the compensator 84 (particularly in the digital domain).

Where dimming functionality is required, the value of OUT, SET may be manipulated to implement dimming as shown in Figure 10, in which lOUTSET is multiplied (gain scaled) by a dimming value DIM indicating the amount of dimming to be applied. The value of DIM is suitably between 0 and 1, where 0 is complete dimming and 1 is no dimming with intermediate dimming there between. The dimming value is suitably obtained from a dimming detection circuit (not shown) which may operate for example by measuring the duty cycle of the incoming mains supply, since this is indicative of the measure of dimming applied by a conventional triac based mains dimmer circuit. Dimming functionality is not restricted to implementation in the controller 76. Dimming may for example be implemented in the switching circuit 72 by, for example, sequentially gating the switching signal with a second switching signal based on a measure of dimming.

The use of averaging allows for measures to improve PFC performance by virtue of the elimination of the bulk capacitor. An arrangement with a measure for improving PFC performance is shown in Figure iii. In this arrangement which corresponds to the previously described controller of Figure 8, the value of VSETISENSE is modulated by a signal representative of the incoming mains supply. This representative signal may for example be the mains rectified signal \Jfjlt. In this way, the current drawn on the primary side follows the mains voltage.

The representative signal of the mains may be modified by a non-linear transfer function. Thus as shown in the dotted outline, the representative signal may be passed through a saturator, which limits the maximum value of the representative value. It will be appreciated that whilst in the analog domain a saturator function may readily be implemented using a clipping circuit, the digital domain allows for more complex non-linear functions to be implemented. Whilst, the use of a non-linear function may result in poorer PFC performance, it does result in improved ripple performance on the output. Accordingly, a designer can by careful selection of the non-linear transfer function, provide an optimum trade off between PFC performance and output ripple.

In the exemplary arrangement of Figure 11, a multiplier is used to modulate the value of VSETISENSE to force it to follow the rectified mains. The saturation function has the result of producing a fixed VSETISENSE over a section of the half mains cycle. The amount of the mains cycle over which the peak current remains constant depends on the level of the saturation.

When the mains voltage is below the level of this saturation then the peak current will follow the mains voltage over that portion of the waveform.

Clearly when the saturation is not enabled at all then the peak current follows the mains voltage shape over the full mains cycle. In this scenario, maximum PFC performance is obtained at the expense of highest output ripple.

It will be appreciated that the PFC and dimming functionality described above may be readily combined together. For example, the arrangement of Figure 11 may be configured to have OUT, SET varied in response to the level of dimming required or as described previously sequentially gating the switching signal with a second switching signal based on a measure of dimming.

Whilst the above examples, equations and circuits have been explained with reference to a flyback converter, the techniques, methods and circuits described are not so restricted. The methods may also be used in other topologies including non-isolated ones. It will be appreciated that depending on the topology, that a scaling factor or other modification may be required when implementing the method in a circuit.

Similarly, it will be appreciated that whilst the present application has been described generally with respect to providing drive current to a load comprising one or more LED's, that the present application is not restricted to any one particular application and is suitable for a wide variety of loads.

Nonetheless, the above described methods and circuits are particularly suited to the needs of LED lights and where used improve the reliability and efficiency without significant additional cost or components.

An example of such a LED light or Luminaire which may employ the previously described circuits and method is illustrated in Figure 14 with an equivalent circuit diagram in Figure uS. The luminaire comprises a base section (fitting) for engaging with a corresponding light socket in a light fitting. The fitting provides electrical contacts to connect the circuitry of the fitting to the mains electricity. The main body 502 of the light is affixed to the fitting and houses the circuitry of the light. The mains voltage provided by the fitting may initially be connected through a protection device such a thermal overload device or fuse and\or an EMI filter circuit 510. The rectifier circuit 512 is then employed to provide a rectified mains voltage to an SMPS circuit 514. The SMPS circuit which is suitably of the type generally described above provides a drive current to the load, which in this case may be a combination of LED's typically arranged in a series configuration. The LED's are housed in a top section (head) of the light which may also include an arrangement, for example, of lenses and/or reflectors to direct light from the LEDs into the space to be lit.

It will be appreciated that whilst several different embodiments have been described herein, that the features of each may be advantageously combined together in a variety of forms to achieve advantage.

In the foregoing specification, the application has been described with reference to specific examples of embodiments. It will, however, be evident that various modifications and changes may be made therein without departing from the broader spirit and scope of the invention as set forth in the appended claims. For example, the connections may be any type of connection suitable to transfer signals from or to the respective nodes, units or devices, for example via intermediate devices. Accordingly, unless implied or stated otherwise the connections may for example be direct connections or indirect connections.

Because the apparatus implementing the present invention is, for the most part, composed of electronic components and circuits known to those skilled in the art, circuit details will not be explained in any greater extent than that considered necessary as illustrated above, for the understanding and appreciation of the underlying concepts of the present invention and in order not to obfuscate or distract from the teachings of the present application.

Thus, it is to be understood that the architectures depicted herein are merely exemplary, and that in fact many other architectures can be implemented which achieve the same functionality. In an abstract, but still definite sense, any arrangement of components to achieve the same functionality is effectively "associated" such that the desired functionality is achieved. Hence, any two components herein combined to achieve a particular functionality can be seen as "associated with" each other such that the desired functionality is achieved, irrespective of architectures or intermedial components. Likewise, any two components so associated can also be viewed as being "operably connected," or "operably coupled," to each other to achieve the desired functionality. Thus for example references to a controller may be taken to include situations in which the control function is provided by a plurality of discrete elements as well as situations where it is provided as a single device such as an integrated circuit or as part of such an integrated circuit.

Furthermore, those skilled in the art will recognize that boundaries between the functionality of the above described operations are merely illustrative. The functionality of multiple operations may be combined into a single operation, and/or the functionality of a single operation may be distributed in additional operations. Moreover, alternative embodiments may include multiple instances of a particular operation, and the order of operations may be altered in various other embodiments.

However, other modifications, variations and alternatives are also possible.

The specifications and drawings are, accordingly, to be regarded in an illustrative rather than in a restrictive sense.

In the claims, any reference signs placed between parentheses shall not be construed as limiting the claim. The word comprising' does not exclude the presence of other elements or steps than those listed in a claim.

Furthermore, the terms "a" or "an," as used herein, are defined as one or more than one. Also, the use of introductory phrases such as "at least one" and "one or more" in the claims should not be construed to imply that the introduction of another claim element by the indefinite articles "a" or "an" limits any particular claim containing such introduced claim element to inventions containing only one such element, even when the same claim includes the introductory phrases "one or more" or "at least one" and indefinite articles such as "a" or "an." The same holds true for the use of definite articles. Unless stated otherwise, terms such as "first" and "second" are used to arbitrarily distinguish between the elements such terms describe. Thus, these terms are not necessarily intended to indicate temporal or other prioritization of such elements. The mere fact that certain measures are recited in mutually different claims does not indicate that a combination of these measures cannot be used to advantage.

Claims

Claims 1. A primary side method of controlling a mains powered switch mode power supply, the switch mode power supply comprising a transformer with a primary side and a secondary side and a switch for switching the primary side to a supply voltage, the method comprising: a] calculating a plurality of estimates of output current on the secondary side; a] comparing the determined plurality of estimates with a desired secondary side output current, where said comparison is performed on an averaged basis in setting a signal to operate the switch.
2. A method according to claim 1, wherein the averaged basis is performed by averaging the calculated plurality of estimates of output current prior to comparison with the desired secondary side output current.
3. A method according to claim 1, wherein the averaged basis is performed by comparing each of the plurality of estimates with the desired secondary side output current and averaging the results of the comparisons.
4. A method according to claim 1, wherein the plurality of estimates are determined over at least one half mains cycle.
5. A method according to any preceding claim, wherein an estimate of output current is determined for each switching period.
6. A method according to any preceding claim, wherein the method includes modifying the desired secondary side output current in response to a dimming signal.
7. A method according to claim 6, wherein the dimming signal is determined by measuring the duty cycle of the supply voltage.
8. A method according to any preceding claim, wherein the output of the comparison is provided as an error signal to a compensator which provides the setting to operate the switch.
9. A method according to any preceding claim, wherein the setting is modulated by the supply voltage.
10. A method according to claim 9, wherein the modulation is non-linear.
11. A method according to claim 9 or claim 10, wherein the modulation is implemented as a saturation function limiting the supply voltage employed in the modulation.
12. A method according to any preceding claim, wherein the signal set to operate the switch is a current and where the switch is switched off upon reaching this current.
13. A method according to claim 12, wherein the set current employed during one half mains cycle to operate the switch are used in the determination of the currents in a subsequent half mains switching cycle.
14. A method according to any preceding claim, wherein a primary side measure of secondary side voltage is employed to determine a measure of the secondary side off-time which in is used in each calculation of secondary side current.
15. A method according to any preceding claim, wherein the switch is switched on at a switching frequency and at least one of the following applies: a] the period of the switching frequency is employed in the estimate, and b] an estimate of output current is obtained for each switching cycle.
16. A method according to claim 15, wherein jitter is introduced to the switching frequency.
17. A method according to any preceding claim, wherein there is no bulk capacitor on the primary side of the power supply.
18. A method according to any preceding claim, wherein the mains voltage is rectified by a rectifier circuit and a filter capacitor is provided.
19. A method according to claim 18, wherein the filter capacitor has a value less than 300nF.
20. An SMPS controller for controlling a switch mode power supply comprising a transformer with a primary side and a secondary side and a switch for switching the primary side, the controller comprising: an estimator for estimating secondary side current values using primary side measurements; a switching circuit for operating the switch; a controller providing a control signal to the switching circuit, the controller being configured to compare said estimates of secondary side current with a desired value of secondary side current on an averaged basis and being further configured to use the result of the comparison in providing the control signal to the switching circuit.
21. An SMPS controller according to claim 20, wherein the controller comprises an averager for averaging the estimates of secondary side current prior to comparison with the desired value secondary side output current.
22. An SMPS controller according to claim 21, wherein the controller comprises a diffrencer for subtracting the estimates of secondary side current from the desired value of secondary side current to provide error values and an averager for averaging the error values.
23. A SMPS controller according to any one of claims 21 to 22, wherein estimates of secondary side current values or the error values are averaged in the controller over at least one half mains cycle.
24. A SMPS controller according to any one of claims 20 to 23, wherein an estimate of output current is determined for each switching cycle.
25. A SMPS controller according to any one of claims 20 to 23, wherein the controller is configured to modify the desired value of secondary side output current in response to a dimming signal.
26. A SMPS controller according to claim 25, further comprising a dimmer detection circuit for measuring the duty cycle of the supply voltage.
27. A SMPS controller according to any one of claims 20 to 26, further comprising a compensator wherein the comparison of said estimates of secondary side current with a desired value of secondary side current is provided as an error signal to the compensator which provides the control signal to the switching circuit.
28. A SMPS controller according to any one of claims 20 to 27, the SMPS controller including a modulator for modulating the control signal with a measure of supply voltage to provide PFC.
29. A SMPS controller according to claim 28, wherein the modulation is non-linear.
30. A SMPS controller according to claim 28 or claim 29, further comprising a saturator for limiting the measure of supply voltage employed in the modulation.
31. A SMPS for converting an input to an output, the SMPS comprising: a transformer having a primary side and a secondary side, a switch for switchably connecting the primary side of the transformer to the input; a controller according to any one of claims 20 to 30 for controlling the switching of the switch.
32. A SMPS according to claim 31, wherein the SMPS does not have a bulk capacitor on the primary side.
33. The SMPS of claim 31 or claim 32, wherein the SMPS is configured as a flyback converter.
34. A power supply comprising the SMPS of any one of claims 31 to 33, further comprising a rectifier circuit for rectifying a mains input voltage and providing this mains rectified voltage as the input to the SM PS.
35. A light comprising at least one LED, the light further comprising the SMPS of claim 31, wherein the at least one LED is driven by the output from the SMPS.
36. A luminaire having a connector for engaging with a light socket providing a mains voltage, the luminaire light fitting further comprising a light according to claim 35.