GB2499852A - Voltage converter with voltage regulator - Google Patents

Abstract

A voltage converter comprises a circuit which includes a pair of inputs 101, 102, an inductor L1, a flyback diode D1, a Zener diode D2, a switch Q1, a resistor R1,, a capacitor C1 and a pair of outputs 103, 104. The first input 101 is coupled is coupled by inductor L1 to node A. Node A is coupled to the second input 102 by switch Q1 and to the first output 103 by the flyback diode. The flyback diode is arranged such that its anode is coupled to node A and its cathode is coupled to the first output. Node A is also coupled to a node B by the Zener diode. Node B is coupled to the second input and the second output by the resistor. The first output and the second output are coupled by the capacitor. The circuit may be operated in a continuous or a discontinuous switching mode. Once the desired output voltage has been reached it is maintained by the Zener diode and the resistor acting as a voltage regulator. Any voltage across the regulator in excess of the regulation voltage is discharged through the Zener diode and the resistor to the second output.

Description

1

Voltage Converter

The present invention relates to a voltage converter. Particularly, but not exclusively, the present invention relates to a voltage converter which may be used to generate a 5 voltage for use by a sensor which forms part of a turbocharger.

Turbochargers are well known devices for supplying air to the intake of an internal combustion engine at pressures above atmospheric pressure. A conventional turbocharger comprises an exhaust gas driven turbine wheel mounted on a rotatable 10 shaft within a turbine housing. Rotation of the turbine wheel rotates a compressor wheel mounted on the other end of the shaft within a compressor housing. The compressor wheel delivers compressed air to the intake of the engine, thereby increasing engine power. The sensor may be used to measure the speed of rotation of a wheel of the turbocharger, for example the compressor wheel.

15

The speed of rotation of a compressor wheel of a turbocharger may be measured using a sensor which generates an electric field that is perturbated by blades of the compressor wheel as they pass through the electric field. The sensor may monitor the perturbations and thereby determine the speed of rotation of the compressor wheel. A 20 sensor of this type may have a particular set of power supply requirements. In the first instance the sensor may require a large and stable voltage. Secondly, the sensor may require only a small current.

It is an object of the present invention to provide a voltage converter which overcomes

I . • •. 25 or mitigates one or more disadvantages associated with the prior art.

• • • •

According to a first aspect of the present invention, there is provided a voltage converter comprising first and second inputs arranged to receive an input DC voltage,

: first and second outputs arranged to provide an output DC voltage, the second output

• •

30 being coupled to the second input, an inductor coupled between the first input and a node A, a switch connected between node A and the second input, a diode with anode and cathode, the anode coupled to node A and the cathode coupled to the first output, a voltage regulator coupled between node A and the second input arranged to discharge any applied voltage in excess of a regulation voltage, and a capacitor 35 coupled between the first and second outputs.

The voltage regulator may have an output, node B, which is configured to provide an output pulse which indicates that the regulation voltage has been exceeded.

The voltage regulator may comprise a Zener diode with anode and cathode, the anode coupled to the second input and the cathode coupled to the node A.

The voltage converter may further comprise a resistor, the resistor being coupled between the anode of the Zener diode and the second input.

The voltage converter may further comprise a control circuit configured to provide a control signal to the switch which selectively turns the switch on and off.

The node B may be coupled to an input of the control circuit.

The node B may be coupled to an input of the control circuit via a signal conditioning circuit which is arranged to extend the duration of the output pulse.

The signal conditioning circuit may comprise a bipolar transistor with its emitter coupled to the second output, a second resistor coupled between the base of the bipolar transistor and the node B, a pull up resistor coupled between the collector of the bipolar transistor and the first input, and a capacitor coupled between the collector of the bipolar transistor and the second output.

The collector of the bipolar transistor may be coupled to an input of the control circuit.

The inductor may be coupled to the first input by a current limiting resistor, a node C being located between the current limiting resistor and the inductor.

A reservoir capacitor may be coupled between the node C and the second input.

The switch may be a transistor.

A battery may be connected to the first and second inputs.

The second input may be coupled to ground and/or the second output may be coupled to ground.

The input DC voltage may be less than 30V.

According to a second aspect of the invention there is provided a sensor assembly comprising the voltage converter of any preceding aspect of the invention, and further comprising an electrode and a current detection circuit.

The voltage converter may be configured to provide a voltage greater than 100V to the electrode.

The current supplied by the first and second outputs of the voltage converter may be less than 1 mA.

The current detection circuit may comprise a current detector for detecting current flow between the voltage convertor and the electrode due to perturbation of the electric field by the passage of a blade of the compressor wheel through the electric field as the compressor wheel rotates, the current detector being configured to output a first signal modulated at a frequency corresponding to the frequency of perturbation of the electric field, and an amplifier circuit comprising a signal amplifier for amplifying the first signal and outputting a second signal modulated at a frequency corresponding to the frequency of perturbation of the electric field.

According to a third aspect of the present invention, there is provided a turbocharger comprising a turbine connected to a compressor, the compressor comprising a compressor wheel and a sensor assembly configured to measure rotation of the compressor wheel, wherein the sensor assembly comprises an electrode, a current detection circuit and a voltage converter according to any preceding aspect of the invention.

According to a fourth aspect of the invention there is provided a method of controlling a voltage converter, the voltage convertor comprising first and second inputs arranged to receive an input DC voltage, first and second outputs arranged to provide an output DC voltage, a diode with anode and cathode, the cathode coupled to the first output, a

4

node A coupled to the anode the diode, a voltage regulator coupled between node A and the second output arranged to discharge any applied voltage in excess of a regulation voltage, a capacitor coupled between the first and second outputs, and a switch configured such that the capacitor is charged when the switch is open wherein 5 the method comprises switching the switch at a switching frequency which is controlled by a control loop to be a minimum frequency required to maintain a substantially constant output DC voltage between the first and second outputs.

The control loop may monitor an output of the voltage regulator after a switching pulse

10 has been triggered, a pulse on the output of the voltage regulator being indicative of a switching pulse having a magnitude greater than the regulation voltage, and the control loop may reduce the switching frequency if it is greater than a predetermined first threshold.

15 The control loop may monitor an output of the voltage regulator after a switching pulse has been triggered, a pulse on the output of the voltage regulator being indicative of a switching pulse having a magnitude less than the regulation voltage, and the control loop may increase the switching frequency if it is less than a predetermined second threshold.

20

Specific embodiments of the invention will now be described, by way of example only, with reference to the accompanying drawings in which:

25 Figure 1 is an axial cross-section through a turbocharger with a fixed geometry turbine

• •

»' •' which illustrates the basic components of a turbocharger;

Figure 2 is a perspective view of a turbocharger compressor housing of a turbocharger in accordance with an embodiment of the invention;

:***•• 30

Figure 3 is a cross-sectional view of a noise baffle insert of the turbocharger compressor of figure 2;

Figure 4 is a cross-section through a portion of the compressor of figure 3;

35

Figure 5 shows schematic features of a sensor in accordance with an embodiment of the invention;

Figure 6 is a schematic circuit diagram of an embodiment of a sensor circuit in accordance with the present invention;

Figure 7 depicts a voltage converter according to an embodiment of the invention;

Figure 8 depicts a flyback voltage pulse generated as part of the operation of the voltage converter of figure 6;

Figure 9 depicts the voltage converter of figure 6 with input current modulation damping according to an embodiment of the invention;

Figure 10 depicts a voltage converter which corresponds generally with the voltage converter of figure 6 but which includes more detail; and

Figure 11 depicts a microcontroller unit (MCU) process flow sequence to be run by an MCU which forms part of figure 9.

Referring first to figure 1, this is an axial cross-section through a typical turbocharger with a fixed geometry turbine which illustrates the basic components of a turbocharger. The turbocharger comprises a turbine 1 joined to a compressor 2 via a central bearing housing 3. The turbine 1 comprises a turbine housing 4 which houses a turbine wheel 5. Similarly, the compressor 2 comprises a compressor housing 6 which houses a compressor wheel 7. The turbine wheel 5 and compressor wheel 7 are mounted on opposite ends of a common turbo shaft 8 which is supported on bearing assemblies 9 within the bearing housing 3.

The turbine housing 4 is provided with an exhaust gas inlet 10 and an exhaust gas outlet 11. The inlet 10 directs incoming exhaust gas to an annular inlet chamber, i.e. volute 12, surrounding the turbine wheel 5 and communicating therewith via a radially extending annular inlet passageway 13. Rotation of the turbine wheel 5 rotates the compressor wheel 7 which draws in air through an axial inlet 14, defined in part by an

annular inlet wall, and delivers compressed air to the engine intake (not shown) via an outlet 15.

Figures 2 to 4 illustrate a turbocharger compressor of a turbocharger according to an embodiment of the invention, including a sensor assembly according to an embodiment of the invention. Although features of the compressor housing 6 differ in detail from those of the turbocharger compressor of figure 1, the construction is generally the same and where appropriate the same reference numbers are used to identify corresponding features. Referring first to figure 2, the compressor includes a noise baffle 35 in the form of a plastic insert which is installed in the compressor inlet 14. Provision of a compressor noise baffle as an insert separate to the compressor housing is known in the art and the function of the noise baffle will not be described in detail. However, as can be seen in figure 6 the noise baffle 35 is an annular structure which includes a tubular portion 36.

The compressor includes a sensor assembly 16 which includes an electrode 17 which is embedded in the plastic noise baffle 35 as shown in figures 3 and 4. The electrode 17 is generally L-shaped, having an axially extending portion 17a and a radially extending portion 17b which projects into a bore 37 provided in the noise baffle 35 for connection to other components of the sensor assembly 16 as illustrated in figure 4.

Referring to figure 4, the compressor housing 6 is provided with a radial connector aperture 38 in the inlet wall 6a which aligns with the bore 37 of the noise baffle 35 to facilitate electrical connection to the radial portion 17b of the electrode 17. The tubular portion 36 of the noise baffle 35 extends towards the compressor wheel (not shown) so that the end of the axial portion 17a of the electrode 17 lies in close proximity to the compressor wheel.

The sensor assembly 16 further comprises a PCB 22 housed in an electrically insulating sensor casing 23, the PCB being connected to an electrically conductive socket 39. The socket 39 is located in a tubular portion 40 of the casing 23 which is inserted into the baffle bore so that the radial portion 17b of the electrode extends into the socket 39. The tubular portion 40 may be an interference fit in the bore 37 and the electrode radial portion 17b may be an interference fit in the socket 39. The end of the electrode radial portion 17b is tapered to aid the insertion into the socket 39. An O-ring

seal 28 seals the sensor casing 23 with respect to the aperture 38. The sensor assembly 16 is secured to the housing 6 by a bolt 25 which extends through an aperture 26 in the casing 23 and into a threaded bore 27 in the housing wall 69.

The PCB 22 is connected to an output pin 41 which may be used to connect the sensor assembly 16 to an engine ECU or other controller. As shown in figure 5, the sensor assembly may include a circuit 29 comprising a voltage source 31, a current detection circuit 32, an amplifier 33 and a frequency counter 34. The frequency counter 34 may comprise a frequency divider (not shown). The speed of rotation of the compressor wheel 7 is measured using the electrode 17 of the sensing assembly. An electric field is provided at the electrode 17, the electric field being perturbated by blades of the compressor wheel 7 as they pass through the electric field. The current detection circuit 32 monitors resulting current perturbations, thereby allowing the speed of rotation of the compressor wheel to be determined by the frequency counter 34. The frequency counter 34 outputs a signal representative of the rotational speed of the compressor wheel which may be for instance be supplied, via connecting cable 30, to an engine ECU or other controller.

Embedding the electrode 17 within a removable insert 35 ensures that integrating the sensor into the turbocharger is simple and can be done at very low cost. The electrode 17 may for instance be a simple conductive (e.g. copper) wire. Preferably it is of a durable metal. Different compressor inlet configurations can be readily be accommodated by appropriately configuring the baffle insert 35, without the need to change other details of the sensor assembly 16 or compressor housing 6. The electrode can also be replaced if necessary simply by replacing the removable baffle insert 35.

The voltage source 31 may be a voltage converter according to an embodiment of the invention. The voltage converter may convert a voltage supplied from a voltage source such as a vehicle battery to a voltage suitable for use by the electrode 17.

Referring to Figure 6, this shows a schematic circuit illustrating the operation principles of a portion of the sensor circuit 29. A voltage source 31 is electrically connected via a resistor 43 to the electrode 17. The voltage source 31 is also electrically connected to a

common ground 44. The common ground 44 may be a conductive component or may be bodywork or chassis of a vehicle in which the turbocharger is located, or may be earth (particular in the case of a turbocharger fitted to a stationary engine such as a power generator). The compressor wheel is also at the common ground potential indicated by 45. This establishes an electric field between the electrode and the compressor wheel. As discussed above, a modulated current will flow between the voltage source 31 and the electrode 17 as the compressor wheel rotates. This current will cause the voltage at a node 46 intermediate the electrode 17 and resistor 43 to change. The node 46 is electrically connected to the common ground 44 via a capacitor 47 and resistor 48. The change in voltage at node 46 will cause a current to flow through the coupling capacitor 47, which will result in a modulation of the voltage at a second node 49 intermediate the capacitor 47 and the resistor 48. The modulation of voltage at node 49 is amplified by the connected amplifier 33, the output 50 of which is then supplied to the frequency counter 34. The frequency counter (not shown in Figure 6) may be entirely conventional.

US2010/0148760, which is herein incorporated by reference, discloses other features of the turbocharger and the sensor assembly.

Referring to figure 7, a voltage converter 31 according to an embodiment of the invention is shown in a circuit diagram. The voltage converter comprises a circuit which includes a pair of inputs 101, 102, an inductor L1, a flyback diode D1, a Zener diode D2, a switch Q1, a resistor R1 a capacitor C1 and a pair of outputs 103, 104. The first input 101 is coupled by inductor L1 to node A. Node A is coupled to the second input 102 by switch Q1 and to the first output 103 by the flyback diode D1. The flyback diode D1 is arranged such that its anode is coupled to node A and its cathode is coupled to the first output 103. Node A is also coupled to a node B by the Zener diode D2. The Zener diode D2 is arranged such that its anode is coupled to node B and its cathode is coupled to node A. Node B is coupled to the second input 102 and the second output 104 by the resistor R1. The first output 3 and the second output 104 are coupled by the capacitor C1.

Under normal operation, a DC supply voltage of 5V is applied across the first and second inputs 101, 102 and a DC output voltage of 200V is extracted across the first and second output 103, 104.

9

5

10

15

20

• • • • • • • • •

• • • • •

* • • oc » • • •

• •

• * • ")

When the switch Q1 is closed the circuit may be considered to be in an on-state and a current is developed in the inductor L1. The current depends upon the applied DC voltage and the duration for which the switch Q1 is closed according to the equation:

t*-4 <1>

At L

where:

lL is the current in the inductor L1;

At is the duration for which the switch Q1 is closed;

V| is the input voltage applied across the first and second inputs 101, 102; and L is the inductance of the inductor L1.

The on-state duration can be optimized to maximize inductor L1 efficiency. If the on-state is too short then optimum inductor flux density is not achieved. However, if the on-state is too long then the inductor L1 will saturate at its maximum flux density, above which any additional current will be wasted. An on-state duration timed to allow the inductor flux density to reach 66% of its maximum or saturation flux density may be considered optimal.

When the switch Q1 is subsequently opened the circuit may be considered to be in an off-state and the current developed during the on-state will continue to flow, discharging energy stored in the inductor L1 through the flyback diode D1 to be stored in the capacitor C1. The current is in the form of a pulse, known as a flyback pulse. The current passing through flyback diode D1 upon opening of the switch Q1 is described by the equation:

_ K ~ Vq (2)

At L

30

where:

Vo is the output voltage seen across the first and second outputs 103, 104.

10

The energy stored in the capacitor C1 during the off-state is prevented from leaking away by the presence of the flyback diode D1, which, during the on-state, is reverse biased.

• • • • • • •

• • •

• • * 25 ••• •

5 Opening and closing of the switch Q1 is repeated a multiplicity of times. Each time the switch is closed, forward conduction through the flyback diode D1 will not occur until the flyback voltage spike exceeds the output voltage. When this condition is met, the flyback diode D1 becomes forward biased, allowing energy to be transferred to the capacitor C1. As a result of the charge stored in the capacitor C1 the voltage across 10 the first and second output 103, 104 will increase with each consecutive switching pulse by an amount equivalent to the energy which was built up in the inductor L1 during the on-state. The output voltage will approach a desired output voltage according to the relationship:

15 V0=& (3)

where:

Q is the charge stored in the capacitor C1; and C is the capacitance of the capacitor C1.

Considering an application requiring a 200V output voltage to be generated from a 5V input, a voltage gain of 40 is required. If an 80% efficient inductor is assumed, and a 5% cost to overall efficiency of control signals, then an overall voltage gain of 50 should be targeted.

20

It will be appreciated that while an input voltage of 5 V is mentioned here this is merely an example. The method described will be operable with a broad range of input voltages from as low as below 1 V to an input in excess of 100 V. Likewise, the output voltage required by the example discussed is 200 V, but the method will function as 30 well for a variety of output voltages, with the only limitation being that the output voltage is greater than the input voltage.

The switching process may be conducted in one of 2 modes: in continuous mode, or in discontinuous mode. In continuous mode, the switching time is such that inductor L1

11

will not have fully discharged (while switch Q1 (when switch Q1 is closed).

is open) before it begins charging again

During the on-state, the current in the inductor L1 is given by equation (1). At the end of 5 the on-state, the increase in current during the on-state is therefore given by:

1 rDT DT

U.p.'jh V,dt = i^-V, (4)

where:

10 T is the switching period; and

D is the proportion of the switching period where the switch is in the on-state, the duty cycle (0 £ D £ 1).

During the off-state, the current in the inductor L1 is given by equation (2). The change 15 in current during the off-state is therefore given by:

a(5)

If the converter is operated in steady-state conditions, where the amount of energy 20 stored in each component must be the same at the beginning and end of each . switching cycle, then the net current change in inductor L1 during one switching cycle

• must be zero. Therefore:

(6)

(7)

30 which can be reduced to a simple expression for the voltage gain of the converter:

• 14

'■+ • e •

^L fin + \joff ~ 0 .

25

and substituting from equations (3) and (4) yields:

• ••O

DT (1 - D)T AIL,o„ + A/w =^-v, + -V0) = 0

12

— = —!— (8)

Vj l-D

Alternatively, if the switching mode is discontinuous, the inductor L1 current at the beginning of each cycle is zero, and will reach a maximum value which is the same as the change seen in each on-state pulse in continuous mode. This is given by equation (4).

In the off-state, the inductor L1 is again discharged:

1 fDT+sr ST

W=TL (V,-V0)dt = —(V,-V0) (9)

where

5T is the proportion of the time period where lL is discharging.

Once again equating the current increase and decrease in the on-state and off-state, arrive at the expression:

lL„„+h.F.„=^V,+Y{V,-V0) = 0 (10)

which can be simplified to:

5=^^ in)

(v0-v,)

If it is assumed that there are no losses in the system, the current passing through the flyback diode D1 can be expressed as the average current discharged from the inductor L1 during the discharge phase (6T):

- I,

I = ID =-!**-S (12)

2

where l0 is the output current; and

ID is the diode current.

Now substituting in from equations (4) and (11)

7 1 VjDT V,D V'tfT 0 2 L (Vj -V0) 2L{V,-V0)

Which can be rearranged to give an expression for the converter voltage gain in discontinuous mode:

V V D2T

L^ = \ + LI^-L (14)

V, 2 LI0

By comparing this expression with equation (8), which expresses the voltage gain for operation in the continuous mode, it is clear that discontinuous mode allows more flexibility in control of the output voltage, by manipulation of duty cycle, time period and inductor size. The use of a discontinuous mode of operation allows a fixed duration on-state to be used, during which time the inductor L1 can be optimally energised.

Once the desired output voltage has been reached, the output voltage will be maintained at this level by the Zener diode D2 and the resistor R1, which act as a voltage regulator coupled between the node A and the second output 104. Any voltage across the voltage regulator in excess of the target voltage will cause the Zener diode D2 to be reverse biased above its Zener breakdown voltage, and cause current to flow to the second input 102 and output 104, through the Zener diode D2 and the resistor R1, until the voltage is no longer higher than the target voltage. A voltage pulse will be observed at the node B, which can be used to determine that a discharge has taken place.

A target output voltage will be maintained across the outputs 103, 104, but will be reduced by any current flow away from the capacitor C1. The output voltage will be increased each time a flyback pulse is caused to flow through the flyback diode D1 by the opening of the switch Q1. In this way, by periodically charging the capacitor C1 through flyback pulses, it is possible to maintain a substantially higher output voltage than the input voltage supplied to the inputs 101,102.

14

It will be appreciated that the components described above could be exchanged for any suitable alternatives, such as, for example, an avalanche breakdown diode in the place of the Zener diode D2.

5

Furthermore, it will be understood that the inductor L1 could be replaced by a suitably arranged transformer. For example, a primary winding of the transformer, having first and second terminals, could be coupled to the DC power supply through the switch Q1, becoming the primary side. A secondary winding of the transformer, having first and 10 second terminals, could be coupled between node A and the second output, becoming the secondary side. The transformer windings may be coupled to the primary side and to the secondary side in opposing directions, such that when a positive potential is applied to the first terminal of the primary winding, a positive potential is developed on the first terminal of the secondary winding but current is blocked from flowing in the 15 secondary side by the flyback diode D1. Energy stored in the magnetic core of the transformer during the on-state would be released through the flyback diode D1 during the off-state, in a mode of operation similar to that described above.

Voltage regulation could be performed on either the primary side of the transformer, or 20 on the secondary side. If regulation was performed on the secondary side, with the voltage regulator coupled so as to be in parallel with the secondary winding of the transformer an additional blocking diode may be necessary to prevent any current from flowing through the voltage regulator during the on-state.

25 Now referring to figure 8, a flyback pulse 105 is illustrated. The flyback pulse 105 illustrates the voltage at node A immediately after the switch Q1 is opened. This may be referred to as the flyback voltage 106. Any voltage in excess of the Zener breakdown voltage 107 will be discharged through the Zener diode D2. This voltage may be referred to as a control voltage 108. The control voltage 108 will be observed 30 as a voltage pulse across the resistor R1 at the node B.

It will be appreciated that any energy discharged through the Zener diode D2 is not stored in the capacitor C1 and is consequently lost. The control voltage 108 can be used to control the opening and closing of the switch Q1, to maintain a switching rate •»

which is optimal to maintain the target voltage output, while also minimising the energy cost associated with the control voltage 108.

The circuit shown in figure 7 has the advantage that for low current loads, extremely low ripple in the output voltage can be achieved.

It will be appreciated that by placing the voltage regulator (here the Zener diode D2 and the resister R1) in advance of the flyback diode D1, they are not exposed to the output voltage for the full duration of a switching cycle. If the voltage regulator was after the flyback diode D1, then in normal operation the Zener diode D2 would be subject to a reverse bias of at least approximately the Zener breakdown voltage 107 for the full switching cycle and would introduce considerable leakage current. Whether the voltage regulator is in advance of the flyback diode D1 or after it, the Zener diode D2 is subject to a reverse bias exceeding the Zener breakdown voltage 107 during each flyback pulse 105. However, by exposing the Zener diode D2 to no potential difference during the on-state and to the flyback pulse 105 during the off-state, due the closing of the switch Q1, the current leaked through the Zener diode D2 is minimised.

In an embodiment the output voltage may be 200 V, and the current requirement of the load may be in the pA range (e.g. 10's of pA). By the use of low leakage components for the flyback diode D1 and the capacitor C1, total current leakage from the capacitor C1 can be restricted to a total of 200 pA. In contrast to this, if the voltage regulator was connected across the outputs 103 and 104 then a leakage current of 2 pA would be expected if it was constantly biased just below its Zener breakdown voltage 107 of 200 V. This current drawn by the voltage regulator far exceeds the current required by the containment field sensor, and as such, is an inefficient mode of supply.

An advantage of the reduced leakage current is that the switching cycle frequency requirement is reduced. If it is considered that the output voltage must be maintained within a predefined range of the target voltage, and the capacitor C1 is discharged as described above, then the switch must be cycled at a frequency sufficient to charge the capacitor C1 each time it falls to the lower limit of permitted voltage. This variation in output voltage is called the ripple voltage, and can be calculated as follows:

16

Vrr=~L (15)

A-

where:

Vpp is the peak-peak ripple voltage;

I is the leakage current from capacitor C1; and f is the switching cycle frequency.

It will be seen that for a given ripple voltage and capacitance, the required switching frequency is directly proportional to the leakage current. For the exemplary leakage current values discussed above, the present invention permits a reduction in switching frequency of 10000 times.

Considering the leakage current values discussed above, a circuit suffering a constant leakage of 2 pA, with a maximum allowed ripple of 100 mV, and an output capacitor C1 of 10 nF, a switching frequency of 2 kHz is required.

Alternatively, if a leakage current of 200 pA was encountered, the required switching frequency to ensure maximum output voltage ripple of 100 mV using an output capacitor C1 of 10 nF is just 0.2 Hz. Instead, if the switching frequency was set at 10 Hz, the output ripple voltage would be reduced to just 2 mV.

A further advantage brought about by the reduction of leakage current is the reduction in circuit power consumption. It will be appreciated that this advantage is two-fold. Firstly, the reduced leakage current will consume less power. A leakage current of 2 pA across a potential of 200 V will consume 400 pW of power. However, a leakage current of 200 pA across the same 200 V potential will consume just 40 nW.

A second aspect of power saving is brought about by the reduced switching frequency. Each flyback pulse 105 which exceeds the Zener breakdown voltage 107 will cause some energy to be wasted. If the frequency of these flyback pulses is reduced, then the power associated with them will be reduced in proportion to the reduction in switching frequency.

17

It is a requirement of a circuit such as this that any negative voltage excursion at the node A be prevented. In a conventional circuit a steering diode would be required to perform this function, ensuring that node A could never be more negative than the second input 102. A further advantage of this invention is that it allows the Zener diode 5 D2 to serve both as a voltage regulator and as a steering diode, reducing the component count by one.

Referring to figure 9, a converter circuit is schematically illustrated with input current modulation damping. In addition to the components discussed in relation to figure 7, a 10 current limiting resistor R2 is coupled between the first input 101 and the inductor L1. A reservoir capacitor C2 is also coupled between a node C between the current limiting resistor R2 and the inductor L1 and the second input 102.

Considering first the on-state inductor energising phase of circuit operation in the 15 absence of the reservoir capacitor C2, it will be understood that while the switch Q1 is closed, there will be a gradual increase in current drawn from the DC supply, as described by equation (1). At the instant that the switch Q1 is opened, current will continue to flow through the inductor L1, but will be abruptly stopped from the DC supply.

20

The introduction of the reservoir capacitor C2 allows the current drawn from the DC supply to be amortised over the entire switching period, with the reservoir capacitor C2 acting as an intermediate power storage device. The current limiting resistor R2 performs the function of limiting the current drawn by the circuit when it is first 25 connected, before the reservoir capacitor C2 has been fully charged.

If an energising pulse of 1|js is used, with a supply voltage of 5V and an inductor of 100 pH, then the maximum current drawn is 50 mA.

If a pulse rate of 100ms is used, then the equivalent continuous current drawn from the supply will be just 2.5 pA, assuming a linear inductor current increase during the on-state from zero to 50 mA, and also assuming that no-losses are incurred.

The introduction of reservoir capacitor C2, if sized correctly, allows a steady current to 35 be drawn throughout the switching cycle. If a capacitance of 10 nF is selected in

conjunction of a 100 Q limiting resistor, then a maximum current of 30 mA will be drawn after a 1 ps pulse. If a 1 mF capacitor is used, then current modulation will be negligible.

Referring to figure 10, an exemplary practical circuit is schematically illustrated. The circuit operation is similar to that described about in relation to figures 7 and 9, however some components have been replaced with practical components which can be used to implement an embodiment of the invention.

The switch Q1 has been implemented as a field effect transistor (FET). A FET with a blocking voltage capability exceeding the Zener breakdown voltage 107 is required. A suitable component may be a MOSFET ZVN4525 made by Zetex Semiconductors. The FET's gate is coupled to an input VIN which is driven by a microcontroller unit (MCU) 109. Alternatively, any suitable form of switch may be used, for example a bipolar transistor.

A signal conditioning portion of the circuit has been added to ensure the flyback pulse 105, as seen at node B as a control voltage 106 pulse, can be captured by the MCU 109. This functions by storing the control voltage 106 pulse seen at node B and extending the duration for which it can be captured by the MCU 9.

In this embodiment, the signal conditioning portion of the circuit comprises a resistor R3, a bipolar transistor Q2, a pull up resistor R4 and a capacitor C3. The resistor R3 is coupled to the node B, allowing any control voltage 106 pulse to propagate to the base of a bipolar transistor Q2. A general purpose switching transistor such as the DNBT8105 made by Diodes Incorporated may be used for this purpose. The emitter of the transistor Q2 is coupled to ground, which in this circuit is also coupled to the second input 102 and second output 104. The collector of the transistor Q2 is coupled, via a pull-up resistor R4 to the first input 101. The collector of the transistor Q2 is also coupled to an output VREF which is connected to an input of the MCU 109. The output VREF is also coupled by a capacitor C3 to ground.

In operation, when a flyback pulse 105 exceeds the Zener breakdown voltage 107, a voltage is developed across the resistor R1 at node B. This voltage pulse causes a current to flow towards the base of the transistor Q2, the magnitude of which is

19

determined by the size of the resistor R3 and the magnitude of the flyback pulse 105. In response to the current flowing to the base of the transistor Q2 the transistor Q2 is switched on, connecting the output VREF to ground and discharging the capacitor C3. Once the Zener pulse is removed, the current to the base of the transistor Q2 is 5 stopped, and the transistor Q2 no longer conducts. The pull-up resistor R4 now recharges the capacitor C3.

The use of the capacitor C3 prolongs the time for which the voltage at the output VREF is held low. Without the capacitor C3 in place, it would be possible for the MCU to miss 10 a Zener pulse signal if sampling of the output VREF was not performed at the correct moment in time.

The size of resistor R1 may be chosen to ensure sufficient voltage is developed across it, as seen at node B, during a flyback pulse 105 to cause the transistor Q2 to switch 15 on. The resistor minimum value can be calculated from the Zener diode D2 minimum stable operating current as shown in equation 16:

V

R=—^~ IV

Z,min

(16)

20 Where:

R is the resistance of resistor R1;

Iz.min is the Zener diode D2 minimum stable operating current; and Vbe is the transistor switch on voltage.

£ • • •

» «

• • • o

25 For the components suggested above, Vbe may be 1 V, and lz,min 8 mA, yielding a suitable resistor choice of 120 Q.

A compromise may be reached in the selection of the resistor R1. A larger resistance value will lead to a beneficial increase in the output voltage observed at node B, 30 making detection of a flyback pulse 105 more reliable. However, a larger resistor R1 will also have the effect of limiting the current through the Zener diode D2. A low cost Zener diode D2 may have significant variation in switching voltage with a non-ideal linear 'knee' region in the switching characteristic. A larger resistor R1 will force the

20

Zener to be biased in this 'knee' region for a wider range of operating conditions, potentially introducing some uncertainty into switching the characteristic of the circuit.

The base current limiting resistor R3 may be selected to prevent excessive current 5 from flowing to the base of the transistor Q2. It will be understood that the maximum current through the base current limiting resistor R3 will be seen when the voltage at the node B is at a maximum, which will be seen when a flyback pulse 105 exceeds the Zener breakdown voltage 107. The largest voltage expected at the node B in the current configuration is around 2 V. If a maximum allowable current flowing into the 10 base of the transistor Q2 is 2 mA, then a 1 kQ resistor would be a suitable choice.

It will be appreciated that that voltage observed at node B during a flyback pulse 105 will depend on the specific switching characteristics of the Zener diode D2. A Zener diode with a slow response would lead to a delayed and enlarged control voltage 108, 15 for example 5 V, being observed at node B. However, a high quality Zener diode with a fast response time could result in a reduced control voltage 108, for example 0.8 V being observed at node B.

The pull-up resistor R4 should be selected to allow current from the voltage input to 20 charge the capacitor C3 in an acceptable time period. A suitable value is 10 kQ, which would allow a current of 0.4 mA to into capacitor C3 from a supply voltage of 5 V, when the capacitor C3 had been discharged to a voltage of 1 V.

I*.**. The capacitor C3 may be selected to provide a reliable duration for which the MCU can

; 25 sample the value of VREF. A flyback pulse 105 which exceeds the Zener breakdown

• •

voltage 107 will cause the transistor Q2 to discharge the capacitor C3. Once discharge has occurred, the capacitor C3 will be charged through the pull-up resistor R4. The selection of the component values of capacitor C3 and resistor R4 should be made together, to achieve a suitable time constant. For example, if the MCU sampling time 30 was 5 (js, then for the safe acquisition of a correct input value, the RC time constant *;•••* should be selected so that the input to the MCU was held below the input trigger level for at least 10 ps. The MCU input trigger level may be 0.8 V, with the supply voltage at 5 V.

35

The capacitor C3 will be charged through the resistor R4 according to the equation:

21

Fc(0 = ro(l-exp«c) (17)

Where:

R is the resistance of resistor R4;

C is the capacitance of the capacitor C3;

t is the charging time;

V0 is the supply voltage; and

Vc(t) is the time dependent voltage across the capacitor C3.

Rearranging equation 17 to allow the RC time constant to be calculated gives:

RC = (18)

'0

Now using the values described above for voltage and timing requirements, the RC time constant can be found by equation 18 to be ~ 57 (js. Using a resistor value of 10 kQ leads to a capacitance value of 5.7 nF. A component of 10 nF or similar may be used in this instance, providing a safe sampling window of approximately 17 |js.

It will be appreciated that the signal conditioning portion of the circuit could take any suitable form, making use of any suitable components. For example, a FET could be used in place of the bipolar transistor Q2 described above, with corresponding alterations made to the remainder of the circuit.

Referring to figure 11, an MCU process flow sequence (which may be referred to as a control loop) is illustrated. As previously described, each flyback voltage pulse 5 which exceeds the Zener breakdown voltage 107 will generate a control voltage 8 pulse across the resistor R1 at the node B (see figure 7, 9, 10). This voltage pulse can be used as an input to control circuitry, such as the MCU 109, to allow the control circuitry to optimize the switching cycle timing.

At step S1 a timer is loaded with an initial value from which to count down. At step S2 this timer is decremented before processing progresses to step S3. At step S3, if the

time value has reached zero processing progresses to step S4. However, if the timer value has not reached zero then processing returns to step S2. At step S4 a switching pulse is driven to open the switch Q1 for a fixed duration, before processing passes to step S5. At step S5 a run delay is introduced, to allow the switching pulse to propagate through to the voltage converter, before processing passes to step S6. At step S6 the input is tested to determine if the pulse caused a flyback pulse 5 which exceeded the Zener breakdown voltage 107. If the pulse has exceeded the Zener breakdown voltage 107, then a control voltage 108 will have been present at node B which will be detected by the input of the MCU 109. In this instance, processing progresses to step S7. In step S7, the timer start value is compared to a maximum setting (which may be referred to as a predetermined first threshold), which sets the minimum frequency at which the flyback pulse 105 will be generated. If the timer load value is at this setting then processing passes to step S1 and the timing sequence starts again. If the timer load value is not at the maximum setting then processing passes to step S8, where the timer load value is increased by 1, before processing again passes to step S1 where the timing sequence starts once more.

Returning now to step S6, i.e. where the MCU 9 tests whether a flyback pulse 105 which exceeds the Zener breakdown voltage 107, resulting in a control voltage 108, has occurred. If the pulse has not occurred, then processing passes to step S9. In step S9, the timer load value is compared to a minimum setting (which may be referred to as a predetermined second threshold), which sets the maximum frequency at which the flyback pulse 105 will be generated. If the timer load value is at this setting then processing passes to step S1 and the timing sequence starts again. If the timer load value is not at the minimum setting then processing passes to step S10, where the timer load value is decreased by 1, before processing again passes to step S1 of the control loop.

This control loop allows the MCU 109 to alter the elapsed time between successive openings of the switch Q1 at step S4 to achieve an efficient driving scheme. During the initial charging phase, the MCU 9 will cause the voltage converter to run at a high frequency, as the output voltage will be below the target voltage and no flyback pulses 105 will exceed the Zener breakdown voltage 107. However, once the Zener breakdown voltage 107 is reached, the pulse frequency will be reduced on each cycle until the control voltage 108 is no longer detected. At this point, the frequency will be

23

increased again, albeit only slightly, until a control voltage 108 is again detected. In this way, the control circuit can be tuned until the voltage converter is driven at the slowest rate required to maintain the desired output level. The switching frequency is controlled by a control loop to be a minimum frequency required to maintain a substantially 5 constant output DC voltage. Thus, the Zener current is maintained at a minimum detectable level, reducing the energy wasted.

A further advantage of the invention is the reduced need for output damping. By careful tuning of the switching frequency, the number of flyback pulses 105 can be reduced to 10 the minimum necessary to maintain a substantially constant output voltage at the capacitor C1 of, for example, 200 V. It will be understood that should the switching frequency be fixed at a value which is high enough to ensure that a high enough voltage is always maintained, the frequency will be higher than is necessary in atl but the most extreme operational conditions. As such, under normal operating conditions, 15 the output voltage will be at approximately the desired level, yet flyback pulses will be being regularly generated. This will have the effect of generating higher voltages at the peak flyback pulse voltage than would be the case if pulses were only generated when a depleted output capacitor C1 required charging. By only delivering flyback pulses at the minimum frequency necessary to replenish the output capacitor C1, flyback pulse 20 magnitude can be limited to the minimum level required.

The voltage converter described above may be operated in wide range of operating environments. For example, if the voltage converter forms part of a sensor assembly of a turbocharger, it may be expected to operate in a temperature range from - 40 °C to +

It will be appreciated that a Zener diode may have a negative temperature coefficient, and as such the switching voltage may reduce at elevated temperature. However, an internally compensated Zener diode may instead by used, negating the effect of 30 temperature on its switching voltage to a large extent.

Similarly, resistors and other components may be chosen to be internally compensated for the effects of temperature variation.

24

It should be noted that while a highly stable switching voltage may be desirable, a degree of variation in switching voltage may be acceptable in some applications. In fact, it is a further advantage of the voltage converter described above that the output voltage supplied between the output terminals 103 and 104 is relatively independent of 5 Zener switching voltage variation. The arrangement of the circuit so as to place the voltage regulator before the flyback diode D1 ensures that any reduction in switching voltage of the Zener diode is not immediately transmitted to the output voltage.

10

The voltage converter described above may be supplied with a DC voltage from a power supply such as a battery (e.g. a battery which forms part of a vehicle in which the sensor assembly is provided).

25

Claims (24)

CLAIMS:

1 A voltage converter comprising:

first and second inputs arranged to receive an input DC voltage;

first and second outputs arranged to provide an output DC voltage;

a diode with anode and cathode, the cathode coupled to the first output;

a node A at the anode of the diode;

a voltage regulator coupled between the node A and the second output arranged to discharge an applied voltage in excess of a regulation voltage; and a capacitor coupled between the first and second outputs.

2 The voltage convertor of claim 1 further comprising:

an inductor coupled between the first input and a node A;

a switch connected between node A and the second input; and wherein the second output is coupled to the second input.

3 The voltage converter of claim 1 or 2 wherein the voltage regulator has an output, node B, which is configured to provide an output pulse which indicates that the regulation voltage has been exceeded.

4 The voltage converter of any preceding claim wherein the voltage regulator comprises a Zener diode with anode and cathode, the anode coupled to the second output and the cathode coupled to the node A.

5 The voltage converter of claim 4 further comprising a resistor wherein the resistor is coupled between the anode of the Zener diode and the second output.

6 The voltage converter of any preceding claim further comprising a control circuit configured to provide a control signal to the switch which selectively turns the switch on and off.

7

The voltage converter of claim 6 as dependent from any of claims 3, 4 or 5 wherein the node B is coupled to an input of the control circuit.

26

8 The voltage converter of claim 7 wherein the node B is coupled to an input of the control circuit via a signal conditioning circuit which is arranged to extend the duration of the output pulse.

9 The voltage converter of claim 8 wherein the signal conditioning circuit comprises: a bipolar transistor with its emitter coupled to the second output;

a second resistor coupled between the base of the bipolar transistor and the node

B;

a pull up resistor coupled between the collector of the bipolar transistor and the first input; and a capacitor coupled between the collector of the bipolar transistor and the second output.

10 The voltage converter of claim 9 wherein the collector of the bipolar transistor is coupled to an input of the control circuit.

11 The voltage converter of any of claims 2 to 10 wherein the inductor is coupled to the first input by a current limiting resistor, a node C being located between the i

current limiting resistor and the inductor.

12 The voltage converter of claim 11 wherein a reservoir capacitor is coupled between the node C and the second input.

13 The voltage converter of any preceding claim wherein the switch is a transistor.

14 The voltage converter of any preceding claim in which a battery is connected to the first and second inputs.

15 The voltage converter of any preceding claim wherein the second input is coupled to ground and/or the second output is coupled to ground.

16 The voltage converter of any preceding claim wherein the input DC voltage is less than 30V.

17 A sensor assembly for use in measuring the speed of rotation of a compressor wheel, the sensor assembly comprising the voltage converter of any preceding claim and further comprising an electrode and a current detection circuit.

18 The sensor assembly of claim 17 wherein the voltage converter is configured to provide a voltage greater than 100V to the electrode.

19 The sensor assembly of claim 17 or 18 wherein the current supplied by the first and second outputs of the voltage converter is less than 1 mA.

20 The sensor assembly of claim 19 wherein the current detection circuit comprises:

a current detector for detecting current flow between the voltage convertor and the electrode due to perturbation of the electric field by the passage of a blade of the compressor wheel through the electric field as the compressor wheel rotates, the current detector being configured to output a first signal modulated at a frequency corresponding to the frequency of perturbation of the electric field; and an amplifier circuit comprising a signal amplifier for amplifying the first signal and outputting a second signal modulated at a frequency corresponding to the frequency of perturbation of the electric field.

21 A turbocharger comprising a turbine connected to a compressor, the compressor comprising a compressor wheel and a sensor assembly configured to measure rotation of the compressor wheel, wherein the sensor assembly comprises an electrode, a current detection circuit and a voltage converter according to any of claims 1 to 18.

22 A method of controlling a voltage converter, the voltage convertor comprising:

first and second inputs arranged to receive an input DC voltage;

first and second outputs arranged to provide an output DC voltage;

a diode with anode and cathode, the cathode coupled to the first output;

a node A coupled to the anode the diode;

a voltage regulator coupled between node A and the second output arranged to discharge any applied voltage in excess of a regulation voltage;

a capacitor coupled between the first and second outputs; and

a switch configured such that the capacitor is charged when the switch is open wherein:

the method comprises switching the switch at a switching frequency which is controlled by a control loop to be a minimum frequency required to maintain a substantially constant output DC voltage between the first and second outputs.

23 The method according to claim 22 wherein the control loop monitors an output of the voltage regulator after a switching pulse has been triggered, a pulse on the output of the voltage regulator being indicative of a switching pulse having a magnitude greater than the regulation voltage, and where the control loop reduces the switching frequency if it is greater than a predetermined first threshold.

24 The method according to claim 22 or 23 wherein the control loop monitors an output of the voltage regulator after a switching pulse has been triggered, a pulse on the output of the voltage regulator being indicative of a switching pulse having a magnitude less than the regulation voltage, and where the control loop increases the switching frequency if it is less than a predetermined second threshold.