GB2380073A

GB2380073A - Fly-back converter switched-mode power supply

Info

Publication number: GB2380073A
Application number: GB0213802A
Authority: GB
Inventors: Selim Goeksu
Original assignee: Vestel Elektronik Sanayi ve Ticaret AS
Current assignee: Vestel Elektronik Sanayi ve Ticaret AS
Priority date: 2001-06-15
Filing date: 2002-06-14
Publication date: 2003-03-26
Anticipated expiration: 2022-06-14
Also published as: GB2380073B; GB0213802D0

Abstract

A fly-back converter switched-mode power supply comprises: an AC input ACS, a rectifier, B, a transformer TR having separate first and second primary windings FPW, SPW and a secondary winding SW; a switching element K to chop the rectified dc voltage with associated control device L; and a charge pump comprising a capacitive element G, a rectifying element E and the second primary winding of the transformer. The "charge pump" technique is used to limit the mains current harmonics. In an first embodiment the number of turns in the second primary winding is selected such that the voltage Vd generated across the second primary winding is smaller than the peak value of the rectified DC input voltage such that a dead-time interval is caused in which no switching current flows through the capacitive element G. In a second embodiment the AC input is directly connected to the rectifier without interposing capacitive or inductive elements.

Description

FLY-BACK CONVERTER SWITCHED-MODE POWER SUPPLY WITH REDUCED POWER DISSIPATION AND LIMITED LINE HARMONICS FIELD OF THE INVENTION The invention relates to a fly-back converter switched-mode power supply for sinusoidal power consumption where harmonics of the AC input current (the mains) are to be kept below certain limits.

The invention also relates to a television set comprising such a fly-back converter switch-mode power supply. Such a power supply is to supply for example 200-300 W nominal power from the mains to a color TV (CTV) set. The invention is, however, applicable to Switched Mode Power Supplies (SMPS) in general Typically, such fly-back converter switched-mode power supplies comprise a switching element for the clocked application of a rectified and smoothed DC voltage to a primary winding of a transformer. The invention in particular addresses the problem how in a fly-back converter switch-mode power supply the power dissipation by such a switching element can be reduced while at the same time the requirement of the mains current harmonic suppression is fulfilled.

BACKGROUND OF THE INVENTION Fig. 1-1 shows a typical conventional fly-back converter switched-mode power supply SMPS in accordance with a first prior art. In principle, the fly-back converter converts an AC input voltage VACi of a certain voltage level into an AC output voltage VACO of a different lower or higher voltage level. The AC output voltage VACO can be further rectified by a downstream rectifier (not shown in Fig. 1-1). Depending on

the application, different voltage levels are used. If the fly-back converter switched-mode power supply SMPS is for example used as a power supply in a TV set, the AC input

voltage VACi will be in the order of 220 V whilst the AC output voltage VACo will be in the order of 24 V (for example between 8V to 150 V). In the application in a TV set typically the power supply must provide a large power of about 300 W for driving the TV set. Therefore, power dissipation is an important aspect in such switched-mode power supply circuits SMPS.

The fly-back converter switched-mode power supply SMPS typically comprises a rectifier B for rectifying the AC input voltage VACi into a rectified DC input voltage VDCi, a transformer having a first primary winding FPW with a first and second terminal Tll, T12 and a secondary winding SW from which the AC output voltage VACO is taken, a first capacitive element H connected between said second terminal T12 of said first primary winding FPW and a reference potential, for example ground, for smoothing the rectified DC input voltage VDCi, and a switching element K connected to the first terminal Tll of the first primary winding FPW. The switching element K is triggered by a control device L with a trigger signal TS of a predetermined switching frequency for a clocked application of the smoothed and rectified DC input voltage VDCi to the first primary winding. Typically, in a television set such a switching frequency can be in the order between 40 to 60 KHz.

Thus, the basic operation principle of a fly-back converter is to rectify the AC input voltage and then to apply with a predetermined frequency the rectified input voltage to the primary winding FPW by intermittently switching on and off the switching element K. The switched current and the primary winding FPW induces the necessary output voltage VACO in the secondary winding. Although not shown in Fig. 1-1 an additional feed-back circuitry may be provided for feeding

back to the control device L a voltage feed-back information from the secondary winding. The control device L generates the trigger signal TS with a switching frequency which is proportional to such a feed-back voltage information.

In such a switched mode of operation the current IDCi supplied by the bridge rectifier B has a typical waveform as shown in Fig. 7. As can be seen from Fig. 7, the current waveform IDCI comprises, due to the non-ideal rectifying process and the switching process of the switching element K, harmonic components in the current waveform. In principle, these harmonic components are always present in the rectifier output current Ici, aven independently of the power of the fly-back converter. However, it will be appreciated that such harmonic components are particularly severe in a fly-back converter for high power applications, for example in a 300 W television set. In such high power applications, the current peaks (harmonic components) shown in Fig. 7 can exceed the limits of the European EMC (Electro Magnetic Compatibility) Standards.

In order to comply with such European EMC Standards, conventional fly-back converters typically use a technique known as the"charge pump"technique, for reducing the harmonic levels of the current waveform shown in Fig. 7. Fig.

8 shows the output voltage VDCi and the output current IDCi using the"charge pump"technique.

Essentially, as shown in Fig. 1-1, for realizing the"charge pump"technique, a charge pump CP is provided for supplying to and extracting from said first primary winding FPW an harmonics suppression current ISUP. Such a charge pump CP, as typically shown in EP 0 598 197 Bl, comprises a capacitive element G connected between the output of the rectifier B and the first terminal Tll of the first primary winding FPW and a rectifying element E connected between the output of the rectifier B and the second terminal T12 of the first primary winding FPW. Preferably, for further harmonic suppression a

choke coil D can be connected between the output of the rectifier B and the node connecting the capacitive element G and the rectifying element E.

In principle, the"charge pump"technique with the charge pump CP operates as follows. As explained above, the switching element K is switched ON (i. e. connecting the first terminal Tll to ground) and OFF (i. e. opening the ground connection of the first terminal Tll) in accordance with the trigger signal TS shown in Fig. 9. If the switching element K is realized by a semiconductor switching element such as a MOS transistor, the trigger signal TS constitutes the voltage Vgate applied to the gate of the MOS transistor. During the OFF state of the switching element K a potential difference is generated across the capacitor G and the capacitor G will be charged with the rectified current IDCi provided by the bridge rectifier B. More particularly, since there is provided a rectifying element E, e. g. a diode whose anode is connected to the capacitor G and whose cathode is connected to the second terminal T12 of the first primary winding FPW, the capacitor G can be charged by a current as a result of the potential difference. As shown in Fig. 10, showing the voltage at the output of the switching element K, e. g. the drain voltage Vdrain in case the switching element K is realized by a MOS transistor, in the OFF periods of the switching signal TS a voltage will be applied to the capacitive element G.

On the other hand, during the ON state of the switching signal TS, the capacitor G is discharged over the switching element K to ground. The discharge and charge current of the capacitor G, as shown in Fig. 11, flows through the rectifier and changes the current waveform of the rectifier current IDCi in a way to reduce the harmonic levels as shown in Fig. 8. Since without the capacitor G the power factor would be less than 1/2 the capacitor G is generally known as a Power Factor Correction (PFC) capacitor G. For completion, Fig. 13 shows the current waveform Idrain of the switching element K when the conventional"charge pump"technique is used.

Thus, as explained above, one can say that the PFC capacitor G is connected in such a manner that it is discharged with a suppression current ISUP at a timing which is necessary for compensating the harmonics. The"charge pump"technique using a PFC capacitor and a rectifying element E can be used in various variants depending on the desired harmonic suppression characteristic and desired temperature characteristics of the switching element K. As may be appreciated, in a large power fly-back converter, large currents will have to be switched by the switching element K and therefore the power dissipation of the switching element K is an important aspect.

DESCRIPTION OF THE PUBLISHED PRIOR ART US 5,420, 776 and EP 0 598 197 Bl show fly-back converters with different harmonic limitation characteristics and different temperature values of the switching element K as in principle shown in Figs. 3-5 of the attached drawings. Common to all circuits is the constitution of a charge pump CP constituted by a PFC capacitor C (the capacitive element G) and a diode D having its anode connected to the capacitor C and having its cathode connected to the first terminal T12 of the first primary winding FPW.

In the fly-back converter of Fig. 3, the connection of the charge pump CP is made as explained in Fig. 1-1. Furthermore, a reverse coupled coil arrangement L1, L2 is used. The first coil LI is connected between the output of the rectifier B and the node 22 connecting the capacitor C and the diode D. The reverse coupled coil L2 is connected between ground and the rectifier B. For the fly-back converter in Fig. 3 the levels of the harmonics are reduced to acceptable values as required in the European EMC Standards, but the discharge current ISUP of the PFC capacitor C also flows through the switching element Tl and increases the turn-on losses. As shown in Fig.

13, showing the drain current IDrain in case the switching element K is constituted by a semiconductor MOS transistor,

and Fig. 15, showing the power dissipation of the switching element K, the discharge current leads to an additional first peak in the drain current IDrain and thus to an additional power dissipation, i. e. an excessive temperature rise.

In the fly-back converter SMPS of Fig. 4 the PFC capacitor C is not connected to the first terminal Tll of the first primary winding FPW but instead to a third terminal T13 of the first primary winding FPW which is situated between the first terminal T11 and the second terminal T12. The other end of the capacitor C is connected to the interconnection node X of the capacitor C and the diode D.

Alternatively, in the fly-back converter of Fig. 5 the PFC capacitor C is connected to a third terminal T13 of the first primary winding FPW and the interconnection point Y of the capacitor C and the diode D. In both fly-back converters of Fig. 4 and Fig. 5 there will be an additional inductance between the primary winding terminals T11 - T13 (Fig. 4) and Tll - T13 (Fig. 5) and this inductance essentially reduces the PFC capacitor discharge current such that there is a lower current flowing through the switching element K to ground when the switching element K is switched ON thus reducing the power dissipation. Since the reduction in the drain current IDrain of the switching transistor Tl is reduced in Fig. 4 and Fig. 5 of the prior art, of course also the compensating current or suppression current ISUP of the PFC capacitor C is reduced such that this of course counteracts the harmonic level suppression. Thus, even when using an additional inductance on the primary winding, i. e. using an additional terminal or tap on the first primary winding, the simultaneous effect of a superior harmonics suppression as well as a reduced power dissipation cannot be achieved.

Another alternative of a power source device for applying a high frequency AC voltage to a load is disclosed in US 5,644, 480. Fig. 1-2 shows the circuit diagram corresponding to

Fig. 11 of US 5,644, 480. This type of power source device is particularly adapted to improve the power factor and eliminate higher harmonic distortion in the input current from the AC source 11. As shown in Fig. 1-2, this power source device comprises an inductor LI between the rectifier B and the AC source 11, and an additional winding n connected through a capacitor Cl to the output of the rectifier B at a first terminal and connected to the other output of the rectifier B at a second terminal. A diode Dl is connected between an inverter circuit 12 and the output of the rectifier B. This type of power source device is particularly adapted for driving a load in the form of a discharge lamp 14. Thus, the voltage and power requirements for driving a discharge lamp are quite different ones to using such a power source device in a television set.

Thus, one can say that US 5.644. 480 has added to the first prior art in Fig. 1-1 the inductor LI and the second winding n together with the capacitor Cl and the diode Dl interconnected in the special manner as shown in Fig. 1-2. It is described in US 5,644, 480 that the inductor LI should always be present and that a resonance frequency fo of the resonance circuit formed by LI, Cl is considered to be

fo = 1/2 1/2 7c -VLICI whereby the inductance of the inductor LI and the capacitance of the capacitor Cl should be optimally set such that the resonance frequency fi will be substantially equal to the frequency f of the output high frequency AC voltage of the inverter circuit 12. In all embodiments described in US 5,644, 480 the solution for improving the power factor and eliminating a higher harmonic distortion includes the provision of the inductor LI.

Whilst the use of such a power source device as in Fig. 1-2 may be sufficient for driving a discharge lamp 14 together

with the provision of the inductor LI, such a power source device may have serious problems when being applied to a television receiver. First of all, considering the power consumption of about 150 W, the inductor LI is a quite costly component in the circuit. Furthermore, considering that the conductor LI is directly connected to the AC voltage source 11, the power loss in the inductor LI could be quite high if a product such as a TV receiver is switched to a kind of standby mode as is conventional in consumer electronics. Such a target value for some products in stand-by mode is between 1-3 Watt and therefore quite a considerable power loss could be caused in LI. However, in US 5,644, 480 the inductor LI is necessary for achieving the improvement in the power factor and the elimination of higher power distortion in the input current from the AC source.

Another disadvantage regarding the power dissipation is illustrated in the attached Fig. 1-3. In Fig. 1-3 Vd is the voltage over the additional winding n, VR is the rectified output voltage of the rectifier Band VEO is the voltage over the capacitor CO. As indicated in Fig. 1-3, the circuit in US 5,644, 480 operates with a level of Vd exceeding VEO. Therefore, there is a continuous current flow through the capacitor Cl during the full period of the rectified mains voltage signal VR. This presents a higher amount of load to the switching element and therefore increases the power dissipation and increases the operational temperature of the switching elements in the inverter circuit 12. Therefore, whilst the circuit in Fig. 1-2 shows some improvement of the power factor and the elimination of higher harmonic distortion by comparison to the circuit in Fig. 1-1, the arrangement of the inductor LI and the additional winding n still cause a quite high power dissipation.

SUMMARY OF THE INVENTION

As explained above, in the prior art the"charge pump" technique can be used in order to provide a compensating or suppression current for providing a harmonics suppression current. However, this leads to high temperatures in the switching element Tl. On the other hand, improvements in the prior art"charge pump"technique in order to reduce the power dissipation (temperature rise) in the switching element K on the other hand compromises the harmonic suppression effect.

Therefore, in the prior art no suggestion has been made to improve the power dissipation of the switching element whilst simultaneously providing a superior harmonics suppression.

Therefore, the object of the present invention is to provide a fly-back converter switched-mode power supply with an improved power dissipation of the switching element whilst simultaneously a superior harmonics suppression with a"charge pump"technique can be achieved.

This object is solved by a fly-back converter switched-mode power supply (claim 1) for sinusoidal power consumption, comprising a rectifier for rectifying an AC input voltage from an AC source into a rectified DC input voltage ; a transformer having a first primary winding with a first and second terminal and a secondary winding ; a first capacitive element connected between said second terminal of said first primary winding and a reference potential for smoothing said rectified DC input voltage ; a switching element connected to said first terminal of said first primary winding for a clocked application of the smoothed rectified DC input voltage to the first primary winding ; a control device for triggering said switching element with a trigger signal of a predetermined switching frequency ; and a charge or current pump for supplying to and extracting from said switching element an harmonics suppressing current ; wherein said transformer has at least one second primary winding separate from said first primary winding and having a first terminal connected to said reference potential and at least one second terminal ; and said

charge pump comprises said second primary winding, a second capacitive element connected between the output of the rectifier and the second terminal of said second primary winding and a rectifying element connected between the output of the rectifier and said second terminal of said first primary winding, and wherein the number of turns in the second primary winding is selected such that the voltage generated across the first and second terminal of the second primary winding is smaller than the peak value of the rectified DC input voltage for causing a dead-time interval in which no switching current flows through the second capacitive element.

Thus, also in the invention a charge pump is used but is constituted in connection with a second primary winding which is separate from the first primary winding, i. e. there is not only an additional terminal on the first primary winding but actually a separate primary winding. Since the capacitive element is connected to this second primary winding the flow of discharge current or suppression current from the PFC capacitor to the switching element is prevented such that the discharge current cannot make an additional contribution to the temperature rise in the current flowing through the switching element. On the other hand, the harmonic suppression current is effectively provided in a non-reduced manner for an effective harmonic suppression. Since in accordance with the invention the number of turns in the second primary winding is selected such that a dead-time interval is caused in which no switching current flows through the second capacitive element, the power dissipation can be reduced and the operational temperature of the switching element and the switch mode transformer can be decreased.

Furthermore, the object of the invention is solved by (claim 12) a fly-back converter switched-mode power supply for sinusoidal power consumption, comprising: a rectifier for rectifying an AC input voltage from an AC source into a rectified DC input voltage ; a transformer having a first

primary winding with a first and second terminal and a secondary winding ; a first capacitive element connected between said second terminal of said first primary winding and a reference potential for smoothing said rectified DC input voltage ; a switching element connected to said first terminal of said first primary winding for a clocked application of the smoothed rectified DC input voltage to the first primary winding ; a control device for triggering said switching element with a trigger signal of a predetermined switching frequency ; and a charge pump for supplying to and extracting from said switching element an harmonics suppressing current ; wherein said transformer has at least one second primary winding separate from said first primary winding and having a first terminal connected to said reference potential and at least one second terminal ; said charge pump comprises said second primary winding, a second capacitive element connected between the output of the rectifier and the second terminal of said second primary winding and a rectifying element connected between the output of the rectifier and said second terminal of said first primary winding ; and wherein said source is directly connected to said rectifier B without interposing capacitive or inductive elements.

PREFERRED ASPECTS OF THE INVENTION Preferably (claim 2), a feed-back circuit is provided for feeding back the said control device a voltage derived from said secondary winding wherein said control device generates the trigger signal with a switching frequency which is proportional to the feed-back voltage. Thus, the switching frequency of the switching element can be adapted to the output voltage level.

Preferably (claim 3), the second primary winding of the transformer can comprise two or more second terminals. This can be used for variable charge and discharge current for variable harmonic limitations.

Preferably (claim 4), a choke coil is connected between the output of the rectifier and the input of the rectifying element of the charge pump. The choke coil in principle helps to reduce the harmonics in addition to the"charge pump" technique.

Preferably (claim 5), said first and/or second capacitive element comprises a capacitor.

Preferably (claim 6), the rectifier is constituted by a bridge rectifier.

Preferably (claim 7), said rectifying element is constituted by a diode.

Preferably (claim 8), the switching element is constituted by a transistor, preferably a MOS transistor having its source connected to ground, having its drain connected to the first terminal of the first primary winding and having its gate connected to the control device for receiving the trigger signal.

Preferably (claim 9), the AC input voltage has a frequency between 50 to 60 Hz, wherein said predetermined switching frequency is between 40 to 120 KHz, preferably between 40 to 60 KHz.

Preferably (claim 10), the fly-back converter switched-mode power supply is used in a television set, preferably in a color television set.

Preferably (claim 11), the AC source is directly connected to said rectifier without interposing capacitive or inductive elements. Thus, whilst still a sufficient harmonics suppression and reduction of power dissipation can be achieved, there is no power loss on the path between the AC

source and the rectifier B which could cause an additional power dissipation, in particular when the converter is switched to a stand-by mode, for example in consumer electronics.

It should be noted that various modifications and variations of the invention can be made by a skilled person on the basis of the teachings contained herein. Moreover, the invention can comprise embodiments which include a combination of features and steps which have been separately described in the specification, the claims and the drawings.

Thus, the invention can also comprise other modifications and variations as will be apparent to a skilled person on the basis of the teachings contained herein. Therefore, what is described herein is only what the inventors currently conceive as the best mode of the invention and should in no way be seen as limiting the invention thereto.

Hereinafter, the invention will be explained with reference to its advantageous embodiments and with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS In the drawings: Fig. 1-1 shows a typical conventional fly-back converter switched-mode power supply SMPS using a charge pump CP for performing a harmonic suppression function ; Fig. 1-2 shows a block diagram of a power source device used for driving a discharge lamp ; Fig. 1-3 shows a diagram of the rectified output VR of the rectifier and the voltage Vd on the winding n in Fig. 1-2 ;

Fig. 2-1 shows a block diagram of a fly-back converter switched-mode power supply SMPS in accordance with the invention ; Fig. 2-2 shows the diagram of the rectified output voltage VDCi of the rectifier B and the voltage Vd on the second winding SPW in Fig. 2-1, in particular showing the dead-time Tde in which no switching current flows through the capacitor G ; Figs. 3-5 show block diagrams of further fly-back converter switched-mode power supplies SMPS in accordance with the prior art ; Fig. 6 shows the bridge rectifier output voltage waveform VDCi of the fly-back converters of Figs. 1-5; Fig. 7 shows the bridge rectifier output current waveform without using the charge pump technique in the prior art ; Fig. 8 shows the output voltage/output current of the rectifier when the charge pump technique is used, in accordance with the prior art and the invention ; Fig. 9 shows the trigger signal TS applied to the switching element K in accordance with the prior art and the invention ; Fig. 10 shows the drain voltage when the switching element K is realized by a MOS transistor, in accordance with the prior art and the invention ; Fig. 11 shows the current waveform of the suppression current (discharge current) ISUP of the capacitive element G of the charge pump CP in accordance with the prior art and the invention ;

Fig. 12 shows the current waveform of the first primary winding FPW, in accordance with the prior art and the invention ; Fig. 13 shows the switching element drain current waveform IDrain for Fig. 3, in accordance with the prior art ; Fig. 14 shows the switching element drain current waveform when using the fly-back converter of the invention in Fig. 2-1 ; Fig. 15 shows the power dissipation of the switching element K in accordance with the prior art ; Fig. 16 shows the power dissipation of the switching element K when using the fly-back converter in Fig. 2-1 in accordance with the invention ; and Fig. 17 shows an embodiment of the fly-back converter in Fig. 2-1, in accordance with the invention.

It should be noted that in the drawings the same or similar reference numerals denote the same or similar parts and steps throughout.

PRINCIPLE OF THE INVENTION Fig. 2-1 shows a block diagram of a fly-back converter switched-mode power supply SMPS in accordance with the invention. As can be seen from Fig. 2-1, one of the essential ideas of the invention for avoiding the undesired power dissipation shown in Fig. 15 of the prior art is to still use a charge pump CP as in the prior art configurations in Figs.

1, 3-5. The charge pump CP, which essentially provides a compensating current Isup for compensating harmonics generated by the rectifier B, is provided in such a manner that during

the switching of the switching element K the discharge and charge current of the capacitive element G of the charge pump CP does not flow through the switching element.

As in Fig. 1.1, the fly-back converter switched-mode power supply SMPS for sinusoidal power consumption comprises a rectifier B for rectifying an AC input voltage VACi into a rectified DC input voltage VDCi, a transformer TR having a first primary winding FPW with a first and second terminal Tll, T12 and a secondary winding SW, a first capacitive element H connected between the second terminal T12 of said first primary winding FPW and the reference potential, for example ground, for smoothing the rectified DC input voltage VDCi, a switching element K connected to said first terminal

Tll of said first primary winding FPW for a clocked application of the smoothed rectified DC input voltage VDCi to the first primary winding FPW, a control device L for triggering said switching element K with a trigger signal TS of a predetermined switching frequency, and a charge pump CP for supplying to and extracting from said first primary winding FPW, move precisely to/from said switching element K, an harmonics suppression current ISUP. In particular, the harmonic suppression current ISUP flows into the rectifier B in order to compensate mains harmonics. That is, the charge pump CP supplies to and extracts from the switching element K current a harmonics suppressing current Igup- As shown in Fig. 2-1 for the invention, the fly-back converter switched-mode power supply SMPS of the invention comprisessimilarly as the power source device in Fig. 1-2 of the prior art-a transformer TR with a first primary winding FPW and at least one second primary winding SPW which is separate from the first primary winding FPW and which has a first terminal T21 connected to the reference potential and at least one second terminal T22. The second primary winding or the several second primary windings SPW are positioned with respect to the first primary winding FPW in such a manner that a current flow

into the first terminal Til of the first primary winding FPW generates a voltage and a magnetic field on the second primary winding SPW being directed in the opposite direction to the voltage and the magnetic field generated on the first primary winding FPW.

A charge pump CP in Fig. 2-1 comprises said second primary winding SPW, a second capacitive element G connected between the output of the rectifier B and the second terminal T22 of said second primary winding SPW and a rectifying element E connected between the output of the rectifier B and said second terminal T12 of said first primary winding FPW.

That is, in Fig. 2-1 the power factor correction PFC capacitor G is not directly connected to the switching element K as for example in Fig. 1.1 but it is connected between the bridge rectifier B and the additional second primary winding SPW specifically wound in the switched mode transformer TR as explained above.

Furthermore, as can be seen from Fig. 2-1, preferably, the AC source ACS providing the input mains voltage VACi is directly connected to the rectifier B, i. e. by contrast to the circuit in Fig. 1-2 the harmonic suppression and the power loss dissipation reduction can be achieved without the inductor LI. Furthermore, as will be understood from a more detailed description of Fig. 2-2 below, the number of turns in the second primary winding SPW is selected such that the voltage Vd generated across the first and second terminal T12, T22 of the second primary winding SPW is smaller than the peak value

of the rectified DC input voltage VDCi output by the rectifier B. This causes a dead-time Tde in which no switching current flows through the second capacitive element G and this reduces the operational temperature of the switching element K and reduces the power dissipation.

In functional terms, the connection of the PFC capacitor G to the second primary winding SPW prevents the flow of discharge current ISUP through the switching element K whilst still the desired waveform in Fig. 8 is achieved in which the harmonics suppression of the mains input voltage VACi is achieved. Thus, the EMC (Electro Magnetic Compatibility) requirement can be fulfilled by the harmonic suppression to its full advantage whilst it is avoided that the PFC capacitor discharge current flows through the switching element K. Thus, whilst in the prior art in Fig. 1-1 the charge pump (charge pump) is achieved with the PFC capacitor G and the primary winding FPW, it can be said that one of the essential principles of the invention is to achieve the charge pump CP with the PFC capacitor G and an additional secondary winding SPW of the transformer TR.

The operation of the circuit in Fig. 2-1 can be understood more clearly firstly by considering the waveforms in Fig. 7, Fig. 8 and Fig. 14,16 versus Fig. 13, Fig. 15. In Fig. 2-1 a mains AC input voltage VACi with for example a sinusoidal waveform of 50 Hz and a voltage 220 V is rectified by the bridge rectifier B and the waveform shown in Fig. 6 is obtained as output from the rectifier B. A portion IDCi'of the rectified current IDCi is flowing through the rectifying element E. In accordance with another preferred embodiment of the invention a choke coil D can be connected between the output of the rectifier B and the input of the rectifying

element E. In this case the portion IDCi 1 of the rectified current IDCi is fed to the choke coil D and then to the rectifying element E, for example a diode. The choke coil D can preferably be provided in order to additionally reduce harmonics, i. e. in addition to the harmonic suppression by the charge pump CP. As in Fig. 1-1, the capacitor H is connected to the second terminal T12 of the first primary winding FPW for smoothing the rectified current IDCi.

The control device L supplies the trigger signal TS with pulses as shown in Fig. 9 to the switching element K and the switching process of the switching element K causes a switched current ILP to flow through the primary winding FPW of the transformer TR. The switched current ILP in the primary winding FPW is the excitation current of the transformer TR.

Without the presence of the capacitor G and the second primary winding SPW in the transformer TR the fly-back converter behaves as an ordinary fly-back converter, i. e. the current waveform is as shown in Fig. 7.

Connecting the PFC capacitor G between the output of the rectifier B and the second terminal T22 of the second primary winding SPW changes the current waveform from Fig. 7 to Fig. 8 as required by the European EMC Standards. However, whilst in the prior art in Fig. 1-1 the PFC capacitor G is charged with the voltage across the first primary winding FPW, in accordance with the invention the potential charging the capacitor G is generated across the first and second terminals T22, T21 of the additional second primary winding SPW.

On the other hand, during the OFF state of the switching element K, there is no current flowing through the first primary winding FPW and the voltage Vd on the additional second primary winding SPW is zero. Thus, in the OFF state, the PFC capacitor G is charged. During the charge period the compensating current ISUp flows from the node connecting the rectifier B and the choke coil D or, if not provided, the rectifying element E to the capacitor G. However, since in the OFF state of the switching element K no current can flow through the primary winding FPW, the compensating current +ISUp flows directly through the rectifier B.

Thus, during the ON state of the switching element K the primary current ILP flows through the first primary winding FPW and this excitation induces a positive voltage in the additional second primary winding SPW proportional to the

number of winding turns (the discharging voltage is denoted with"V"i Fig. 17). The positive voltage Vd directed from T22 to T21 is opposite to the voltage between T12 to Tll caused by the current IL- In instances when potential Vd is above the rectified voltage VDCi, the PFC capacitor G is discharged with the discharge current-ISUP flowing through the rectifier B in order to compensate the mains harmonics.

Thus, in Fig. 2-1 one of the advantages is that the PFC capacitor G can be charged and discharged by the second primary winding SPW and still the discharge current-sup does not flow through the switching element K. Thus, it can be avoided that the discharge current contributes to a temperature rise when it flows through the switching element K as would be the case in Fig. 1-1 and Figs. 3-5.

It should also be noted that the provision of an additional and separate second primary winding SPW is different to just adding an additional terminal like the terminals T13 in Fig. 4 and Fig. 5 of the prior art. As can be seen from Fig. 4 and Fig. 5, despite the provision of the additional terminals T13, there will always be a contribution of the discharge current ISUp which still flows through the switching element K during discharge. On the other hand, in the present invention the discharge current essentially does not flow through the switching element K. Therefore, as shown in Fig. 14, the first peak in the drain current IDrain is smaller than the one in Fig. 13 of the prior art which leads to a reduced power dissipation at this point in time, as shown in Fig. 16. On the other hand, the harmonic suppression is still obtained as seen in Fig. 8.

However, another important advantage of the circuit in Fig. 2- 1 can also be understood by considering the waveforms shown in Fig. 2-2 of the attached drawings. As shown in Fig. 2-2, in

accordance with the invention the level of Vd is adjusted to be below VH, i. e. the voltage across the capacitor H and preferably below the level of the rectified DC input voltage VDCi output by the rectifier B. When the voltage Vd is adjusted to be sufficiently lower than VDCi, then the deadtime interval Tde will occur in which no switching current is present in the capacitor G (except for a charging current at the beginning of the dead-time interval which is negligible). The switching current on the PFC capacitor G only flows during tpFCl and tpFC2 in which the voltage Vd is higher than the rectified mains voltage VDCi. A preferred value is that the number of turns of the second primary winding SPW is selected

in such a manner that the voltage Vd is about 10-20% smaller than the peak value of the rectified mains voltage VDCi. By reducing the number of turns in the winding SPW with respect to the number of turns of the first primary winding SPW allows to achieve the dead-time Tde while keeping the mains current harmonic values below certain limits. Since there is no switching current flowing through the capacitor G in the dead-time Tde, the operational temperature of the switching element K and the switch mode transformer TR can be reduced, in other words, the power dissipation can be reduced whilst still the mains current harmonic suppression is achieved. By contrast, in the circuit in Fig. 1-2 of the described prior art, there is a continuous current flowing through Cl during the full period rectified mains voltage. Thus, the power dissipation is much higher than in the embodiment of the invention in Fig. 2-1.

EMBODIMENTS OF THE INVENTION Fig. 17 shows an embodiment of the principle of the invention shown in Fig. 2. In Fig. 17 the rectifier B is constituted by a bridge rectifier, the rectifying element E is formed by a diode, the capacitive element G is formed by a capacitor and the other capacitive element H is formed by a large capacitor.

The switching element K is formed by an MOS transistor having its source connected to ground, having its gate connected to the control device L for receiving the trigger signal TS and having its drain connected to the first terminal Tl1 of the first primary winding FPW.

Such a fly-back converter switched-mode power supply can in particular be used to supply 200-300 W nominal power from the mains to a color TV (CTV) set.

Although not shown in Fig. 17, in accordance with another embodiment of the invention the second primary winding SPW can have two or more terminals for variable charge and discharge currents for variable harmonic current limitations. That is, in accordance with the EMC specifications in various countries different levels of charge and discharge currents for the harmonic suppression can be selected. This can also be performed automatically by connecting a respective switch between the output of the rectifier B and a respective terminal (second terminal) of the second primary winding SPW.

Alternatively or in addition to this, several second primary windings SPW can be provided.

INDUSTRIAL APPLICABILITY In fly-back converter switched-mode power supplies SMPS the "charge pump"technique is used to limit the mains current harmonics. Since the known charge pump technique used in flyback type converters SMPS has the problem of high power dissipation in the switching element, the invention suggests a new charge pump CP in which a PFC capacitor G is connected to an additional second primary winding SPW of the transformer TR. By means of the additional second primary winding SPW the stress on the switching element K is significantly reduced without sacrificing the limitation of mains harmonic suppression and without using any snubber or voltage clamping network.

The fly-back converter switched-mode power supply of the invention cannot only be used in TV sets. It may be used in various applications where a voltage conversion from an input AC voltage to an output AC/DC voltage is desired. It may be used for upward and downward conversion depending on the turn ratio of switching transformer TR.

Various modifications and variations of the invention can be carried out within the teachings of the invention as disclosed herein. In particular, the invention can comprise embodiments which comprise features and steps which have been separately described and illustrated in the description, the claims and the drawings.

Reference numerals in the claims only serve clarification purposes and do not limit the scope of these claims.

Claims

Claims 1. A fly-back converter switched-mode power supply (SMPS) for sinusoidal power consumption, comprising: a) a rectifier (B) for rectifying an AC input voltage (VACi) from an AC source (ACS) into a rectified DC input voltage (VDCi b) a transformer (TR) having a first primary winding (FPW) with a first and second terminal (ill, T12) and a secondary winding (SW) ; c) a first capacitive element (H) connected between said second terminal (T12) of said first primary winding (FPW) and a reference potential (GND) for smoothing said rectified DC input voltage (VDCi) ; d) a switching element (K) connected to said first terminal (ill) of said first primary winding (FPW) for a clocked application of the smoothed rectified DC input voltage (VDCi) to the first primary winding (FPW); e) a control device (L) for triggering said switching element (K) with a trigger signal (TS) of a predetermined switching frequency ; and f) a charge pump (CP) for supplying to and extracting from said switching element (K) an harmonics suppressing current (ISUP); wherein g) said transformer (TR) has at least one second primary winding (SPW) separate from said first primary winding (FPW) and having a first terminal

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(T21) connected to said reference potential (GND) and at least one second terminal (T22) ; h) said charge pump (CP) comprises said second primary winding (SPW), a second capacitive element (G) connected between the output of the rectifier (B) and the second terminal (T22) of said second primary winding (SPW) and a rectifying element (E) connected between the output of the rectifier (B) and said second terminal (T12) of said first primary winding (FPW) ; and wherein i) the number of turns in the second primary winding (SPW) is selected such that the voltage (Vd) generated across the first and second terminal (T21, T22) of the second primary winding (SPW) is smaller than the peak value of the rectified DC input

voltage (VDCi) for causing a dead-time interval (Tde) in which no switching current flows through the second capacitive element (G).
2. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that a feed-back circuit is provided for feeding back to said control device (L) a voltage derived from said secondary winding (SW), wherein said control device (L) generates said trigger signal (TS) with a switching frequency which is proportional to said feed-back voltage.
3. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that said second primary winding (SPW) comprises two or more second terminals.
4. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that

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a choke coil (D) is connected between the output of the rectifier (B) and the input of the rectifying element (E).
5. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that said first and/or second capacitive elements (H ; G) comprise a capacitor.
6. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that said rectifier (B) is constituted by a bridge rectifier.
7. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that said rectifying element (E) is constituted by a diode.
8. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that said switching element (K) is constituted by a MOS transistor.
9. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that said predetermined switching frequency is between 40 to 120 KHz.
10. A television set comprising a fly-back converter switched-mode power supply (SMPS) according to one or more of claim 1-9.
11. A fly-back converter switched-mode power supply (SMPS) according to claim 1, characterized in that said AC source (ACS) is directly connected to said rectifier (B) without interposing capacitive or inductive elements.

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12. A fly-back converter switched-mode power supply (SMPS) for sinusoidal power consumption, comprising: a) a rectifier (B) for rectifying an AC input voltage (VACi) from an AC source (ACS) into a rectified DC input voltage (VDCi b) a transformer (TR) having a first primary winding (FPW) with a first and second terminal (ill, T12) and a secondary winding (SW); c) a first capacitive element (H) connected between said second terminal (T12) of said first primary winding (FPW) and a reference potential (GND) for smoothing said rectified DC input voltage (VDCi) ; d) a switching element (K) connected to said first terminal (Tl1) of said first primary winding (FPW) for a clocked application of the smoothed rectified DC input voltage (VDCi) to the first primary winding (FPW) ; e) a control device (L) for triggering said switching element (K) with a trigger signal (TS) of a predetermined switching frequency ; and f) a charge pump (CP) for supplying to and extracting from said switching element (K) an harmonics suppressing current (ISUP) ; wherein g) said transformer (TR) has at least one second primary winding (SPW) separate from said first primary winding (FPW) and having a first terminal (T21) connected to said reference potential (GND) and at least one second terminal (T22) ;

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h) said charge pump (CP) comprises said second primary winding (SPW), a second capacitive element (G) connected between the output of the rectifier (B) and the second terminal (T22) of said second primary winding (SPW) and a rectifying element (E) connected between the output of the rectifier (B) and said second terminal (T12) of said first primary winding (FPW) ; and wherein i) said AC source (ACS) is directly connected to said rectifier (B) without interposing capacitive or inductive elements.
13. A fly-back converter switched-mode power supply (SMPS) according to claim 12, characterized in that the number of turns in the second primary winding (SPW) is selected such that the voltage (Vd) generated across the first and second terminal (T21, T22) of the second primary winding (SPW) is smaller than the peak value of the rectified DC input voltage (VDCi) for causing a dead- time interval (Tde) in which no switching current flows through the second capacitive element (G).