GB2270384A - A variable frequency oscillator system with ratio conversion properties - Google Patents

Abstract

The system converts an impedance circuit unbalance to a difference of two frequencies and thus, if desired, to a mark-to-space ratio. As shown the sinusoidal output from an oscillator circuit comprising an amplifier A1 with positive feedback is used to excite a bridge network B. The signal produced by any bridge unbalance is fed back in phase or antiphase with the oscillator output signal so as to vary the apparent reactance of the inductor Land capacitor C which determine the frequency of oscillation, thereby altering the frequency. Each time an integral number of cycles of oscillation have been completed, a change-over switching circuit S reverses between its position 1, in which L and C are unmodified, and position 2 in which a bridge unbalance voltage is fed to L and C to alter their effective values. The oscillation frequency thus alternates between two corresponding values. A circuit is described (Fig. 5 not shown) which generates a mark-to-space ratio which is directly related to, and is therefore a measure of, bridge unbalance. The oscillator system may thus effectively convert an impedance ratio to a time ratio. Simple analogue or digital circuits which measure the mark and space periods provide a readout of bridge unbalance. <IMAGE>

Description

A VARIABLE FREQUENCN OSCILLATOR SYSTEM WITH RATIO CONVERSION PROPERTIES THIS INVENTION relates to systems comprising an electronic variable frequency oscillator circuit interconnected with sensing elements whose impendance varies with change in some physical parameter, so that the oscillator frequency varies with and thereby can be used in measurement of the physical parameter.

It is an object of the present invention to provide an improved variable frequency oscillator system which effectively converts the ratio of two impendances in a bridge circuit to the ratio of two frequencies and, if required, to the ratio of two time periods, thus allowing measurements to be riade which are substantially unaffected by signal or power supply voltage levels. In systems designed to provide readout fran bridge circuits, there are well known advantages to be gained in using bridge unbalance to vary the frequency of the bridge excitation oscillator, so that the frequency change becanes a measure of the bridge unbalance.In many known circuits of this kind, the output of the bridge circuit is used to introduce a phase shift into the oscillator feedback loop or an auxiliary loop, thereby altering the frequency of oscillation by an amount dependent on the phase slope or Q-factor of the oscillator frequency-determining circuit. This requires the oscillator circuit to generate quadrature signals, and also means that the sensitivity of the system depends on the phase slope, which may not be well defined. The present invention avoids both of these disadvantages.

The present invention embodies a variable frequency oscillator system, comprising an oscillator consisting of an amplifier and positive feedback circuit where the amplifier is interconnected with a resonant circuit containing an inductor and a capacitor so that the gain of the amplifier has maximum amplitude and zero phase shift at the natural frequency of the resonant circuit and where the positive feedback circuit produces sinusoidal oscillations of controlled amplitude at the natural frequency of the resonant circuit, means for deviating the frequency of oscillation above and below the natural frequency of the resonant circuit by changing the effective inductance and capacitance of the resonant circuit caiponents this being achieved by applying to the terminals of the inductor and capacitor a modifying signal or signals in phase or antiphase with the oscillator output signal, and means for generating the modifying signal using a bridge circuit or other initially balanced circuit excited from the oscillator output signal so that any unbalance produces a signal whose amplitude is related to the degree of unbalance and whose phase is either the same as or the inverse of the oscillator output signal depending on the direction of unbalance.

A full understanding of the invention may be had from the description which follows with reference to the accompanying drawings, in which: Figure 1 is a block diagram circuit showing the basic elements of an oscillator system in accordance with the invention.

Figure 2 is a schematic diagram showing one version of the system embodying a parallel resonant circuit, with modifying signals consisting of two antiphase voltages.

Figure 3 is an alternative version of the system of Figure 2.

Figure 4 is a schematic diagram showing a system which is the electrical dual of the system of Figure 2, embodying a series resonant circuit with modifying signals consisting of three currents.

Figure 5 is a schematic diagram showing the addition to the basic oscillator circuit of a squarer and counter.

Figure 6 is a block diagram circuit showing an oscillator system with the addition of a fast response low ripple analogue output circuit.

Figure 7 is a graph showing the relationship between bridge unbalance x and asymmetry of the oscillator rectangular waveform.

Figure 8 is a block diagram circuit of the oscillator system with the addition of a digital readout circuit.

The operation of the oscillator may be understood from the block diagram of Figure 1, which shows an oscillator whose design is conventional apart from the means used to produce frequency variation. The amplifiers Al, A2 and A3 are voltage amplifiers designed using standard methods well known to those skilled in the field. The amplifier Al is interconnected with a resonant circuit in such a way that the combined circuit has maximum gain and zero phase shift at the natural frequency of the resonant circuit. The output voltage Vo from the resonant circuit is connected to a buffer amplifier A2, and a positive feedback loop is established by connecting the output of amplifierA2 to the input of amplifier Al. Oscillation therefore occurs at the natural frequency of the resonant circuit.The amplitude of oscillation is held constant by a conventional automatic gain control circuit AGC, wherein the oscillator output voltage is rectified to produce a control signal which is used to vary the gain of the amplifier Al appropriately.

The e voltage Vb fran the buffer amplifier A2 is also applied to a bridge circuit B, whose output is connected to a differential amplifier A3. Any unbalance in the bridge circuit, resulting for example fran a change in impendance of one of the arms, will produce at the output of amplifier A3 a voltage Vm = x.Vo, where x depends on the degree of unbalance of the bridge and on the gain of amplifiers A2 and A3. The variable x is arranged to lie within the range -1 through 0 to +1, so the voltage x.Vo will be zero when the bridge is balanced, and will be in phase or antiphase with the oscillator output voltage when the bridge is unbalanced.

The voltage Vin is applied to the resonant circuit in such a way as to alter the effective reactance of the circuit components and thus shift the resonant frequency. The methods of connection depend on the configuration of the resonant circuit, and are fully explained in the following sections by reference to the circuits of Figure 2, Figure 3 and Figure 4, which represent different realisations of the same basic principle.

Though Figure 1 shows a full bridge circuit, the system may also be used with any other initially balanced bridge or half bridge or differential transformer device, or the amplifier A2 may provide differential outputs for use with a half bridge circuit.

The circuit diagram of Figure 2 is similar to Figure 1 except that the resonant circuit is now shown as a parallel circuit containing inductor L and capacitor C. Amplifiers Al, A2 and A3 perform the same functions as in Figure 1, and the amplifier Al is now shown driving the parallel tuned circuit via a resistor R. The low potential sides of the inductor and capacitor are not connected to the common return line, but instead are connected to the outputs of two amplifiers A4A and A4B which have low output impendance, and voltage gains of unity and minus unity respectively. The input terminals of amplifiers A4A and A4B are connected together and to a two way switch S, which in practice would be an electronic switching device.In position 1, the switch S connects the amplifier inputs to the cannon line, and in this condition the resonant circuit action is unmodified and the system oscillates at the natural frequency1 However, if the switch S is moved to position 2, thereby connecting the modifying voltage Vm to the amplifier inputs, then the capacitor in the tuned circuit has the same voltage Vo on its upper terminal but now a voltage x.Vo on its lower terminal, so that the voltage across the capacitor is reduced to Vo.(l-x); the current through the capacitor faust fall proportionately, so the reactance of the capacitor appears to have increased, and the effective capacitance has changed fran C to a new value C. (l-x) .At the same time, the inductor has a voltage Vo on its upper terminal and a voltage -x.Vo on its lower terminal, giving an effective voltage of Vo.(l+x) across the inductor; the current through the inductor must rise proportionately, so the inductive reactance appears to have decreased, and the effective inductance has changed from L to a value L/(l+x). The net effect is to alter the resonant frequency of the circuit from the natural frequency fO, where

to a new value fm, where

and the frequency at which the circuit oscillates will change to this new value; a positive value of x will produce a frequency increase. The gain round the oscillator feedback loop is unchanged, so that the oscillation amplitude remains constant without the need for action from the amplitude control loop.It should be noted that if the bridge circuit is not exactly balanced reactively, so that the bridge output voltage contains a small component whose phase is at 90 degrees to the excitation voltage, then there will also be a change in loop gain in the positive feedback loop which will be carpensated for by the amplitude control circuit.

The circuit of Figure 3 shows a minor variant of the circuit of Figure 2, where the amplifier Al has been replaced by an transconductance amplifier A5 which has a high output impedince, so that the output of amplifier A5 is a current proportional to the amplifier input voltage. The resistor R is now connected in parallel with the tuned circuit rather than in series, otherwise the circuit action is exactly as described for Figure 2. In a further and unrelated change, the bridge network is shown with a differential transformer forming two of its arms, thereby illustrating an alternative mode of use.

The circuit of Figure 4 is an electrical dual version of the circuit of Figure 2, having a series tuned circuit in place of a parallel tuned circuit. The amplifiers Al, A2 and A3 have the same functions as in Figure 2, except that the input voltage to the amplifier A2 is now proportional to the current Io through the series resonant circuit. Each of three amplifiers A6A, A6B and A6C is a transconductance amplifiers, whose output is a current proportional to the input voltage. The input terminals of the three amplifiers are connected together and to a two way switch S, which in practice would be an electronic switching device. In position 1 of switch S, the amplifier inputs are connected to the common line, and in this condition the resonant circuit action is unmodified and the system oscillates at its natural frequency.In position 2 of switch S, the amplifier inputs are connected to the modifying voltage Vm, and now the amplifier A6B delivers a current 2.x.Io into the junction of the inductor and capacitor, while the amplifiers A6A and A6C each extract an antiphase current of x.Io from the outer end of the inductor and capacitor respectively. In this condition, the current through the inductor is reduced from a value Io to a value Io.(l-x); the voltage across the inductor must fall proportionately so the reactance of the inductor now appears smaller, and the effective inductance has changed from L to a new value L(1-x).At the same time, the current through the capacitor has increased from Io to Io.(l+x), so that the capacitive reactance appears to have increased, and the effective capacitance has changed fromC to C/(l+x).

The net effect is to alter the resonant frequency from the natural frequency fO to a new frequency fm, where

This equation is the same as that for the circuit of Figure 2. The main difference in the design requirements for the two circuits is that in Figure 2 the amplifiers A4A and A4B have large reactive currents flowing in their output circuits, whereas in Figure 4 the transconductance amplifier A6B has a large reactive voltage appearing on its output terminal. The design of amplifiers to cater for these easily calculable reactive currents and voltages is straightforward.

The circuit of Figure 5 is similar to that of Figure 2, but in addition the output Vb of the buffer amplifier A2 is connected to a squarer circuit whose output Vs changes state at each zero crossing of the oscillator output waveform. The squarer circuit is connected to a frequency divider, whose output Vd is used to control an electronic switch shown here as the two position switch S. The function of the amplifiers A4A and A4B of Figure 2 is here performed by the amplifiers A7A and A7B and associated feedback components, with values assigned using methods well known to those experienced in the field so that with the switch S in position 1, voltages of +x.Vo and -x.Vo appear respectively on the lower ends of the inductor and capacitor, and with the switch S in position 2 the sense of both voltages is reversed.The frequency divider output Vd is a rectangular waveform which changes state for every N complete cycles of oscillation so that, for a given bridge unbalance, the oscillator will perform N cycles of oscillation at a frequency fm below the natural frequency fo of the tuned circuit followed by N cycles at a frequency fs above the natural frequency. It is a characteristic of the invention that the transition fran one frequency to the other occurs without discontinuity other than the change of slope at the zero crossing of the sine waveform. In Figure 5, the waveform diagrams are for x = 0.33 and N = 2, but N may have any integer value fran 1 upwards. The time periods Tm and Ts for the mark and space states of the rectangular waveform Vd are related to the natural period To of the resonant circuit by the equations

so the difference between Rn and Ts is given by Tx, where

and the sum of Tm and Ts is given by Tt, where

T and the ratio of Tx to Tt is Tt Therefore measurement of the ratio of the time durations Tt and Tx gives a direct measure of any bridge unbalance.When the bridge is balanced, x=O and the mark and space periods Tm and Ts are exactly equal, so that Tax=0. The value of the time ratio Tx/Tt for a given bridge unbalance is independent of the oscillator natural period To, the division ratio N and the bridge excitation voltage Vb, but does depend on the gain of the amplifiers A2, A3, A4A and A4B; however, these can be accurately defined.

The presence of the switching facility in the circuit of Figure 5 thus confers important advantages. Firstly, the zero of the bridge circuit is effectively checked on each cycle of the rectangular waveform. Secondly, a direct voltage output whose value is proportional to the bridge unbalance x may be simply obtained, by arranging that the rectangular output waveform Vd swings between voltage levels of +Vd/2 and -Vd/2 relative to the common line, and applying the waveform to a low pass filter. An unbalance x will result in a direct output voltage of -x.Vd/2, and this output is a linear function of x.Thirdly, additional positions on switch S nay alternatively be used to scan round a number of bridge circuits all excited by the output of amplifier A2, with the switch S selecting each bridge output in turn thereby producing a timenmultiplexed output signal Vd. Fourthly, additional positions may be provided on switch S in which no modifying signal is connected, thereby producing an output waveform Vb consisting of bursts of frequencies fO, fm and fs, and if these frequencies lie in or are translated to the audible range and are applied to a suitable audio output device then the state of bridge unbalance may be monitored by listening to the audio output; this has applications for example in proximity detecting equipment.

Figure 6 shows an application of the oscillator system which provides a fast response low ripple output voltage Va proportional to the bridge unbalance variable x. The rectangular waveform Vd is used to switch an electronic integrator circuit which runs up during the time period Tm, and down at the same rate during the time period Ts; at the end of the period Ts, the value of the output integrator voltage is stored by a sample-and hold circuit, and the integrator is reset. The value stored by the sample-and-hold circuit is proportional to (TnrTd), and is related to the bridge unbalance x, the division ratio N and the oscillator natural period To by the equation for Tx derived previously.

The relation between x and Tx is plotted in Figure 7, and will be seen to be linear for small values of x with some non-linearity at higher values.

Figure 8 shows the oscillator system of Figure 5 with the addition of a digital readout facility which can be used to display the bridge unbalance quantity x in any desired units. The readout is performed during two successive cycles of the rectangular waveform Vd, identified here as cycle A and cycle B. During cycle A, the total time duration Tt is measured and stored. During cycle B, the time difference Tx is measured, and used with the stored value of Tt to produce a readout quantity proportional to the bridge unbalance variable x. The action during cycle B will be described first.

In cycle B, a variable period oscillator VPO running at a frequency Fv is connected to an up/down counter, for which the direction of count is controlled by the state of the rectangular waveform. During the mark period Tm the counter counts up fran zero to a value Fv.Tm, at which point the count direction reverses and the counter counts down for a further period Ts, reaching a final total (ThrTs) .Fv. At the start of the next rectangular waveform cycle the count is latched and displayed.

If the frequency Fv is fixed, then the displayed count will vary with bridge unbalance x as shown in Figure 7, with appreciable non-linearity for larger values of x. If this is not acceptable, then the non-linearity may be corrected as follows. During cycle A an integrator circuit is used to generate a ramp voltage waveform Vc which runs up fran zero to a final voltage Vcm proportional to the total period (TmtTs). This voltage level is then held for the whole of the following rectangular waveform cycle (cycle B), and is used to control the variable period oscillator VPo, so that the period of oscillation is directly proportional to the voltage Van. Thus the frequency Fv of the VPO output is inversely proportional to the voltage Van and thus to the value of (RntTs) for cycle A. The final count is therefore proportional to (Tn-Ts)/(Tin+Ts), which is a linear function of x as already shown. The initial value of the frequency Fv can be set to give any convenient full range value on the digital display.

Figure 8 also shows a pulse generator which produces the necessary latch and reset pulses for the integrator and display circuits. Each of the circuit elements shown as a block can be implemented in any of numerous ways well known to those skilled in the field.

Claims (1)

1. CTWMS

   cLAIM 1 A variable frequency oscillator system, comprising an oscillator consisting of an amplifier and positive feedback circuit where the amplifier is interconnected with a resonant circuit containing an inductor and a capacitor so that the gain of the amplifier has maximum amplitude and zero phase shift at the natural frequency of the resonant circuit and where the positive feedback circuit produces sinusoidal oscillations of controlled amplitude at the natural frequency of the resonant circuit, means for deviating the frequency of oscillation above and below the natural frequency of the resonant circuit by changing the effective inductance and capacitance of the resonant circuit components this being achieved by applying to the terminals of the inductor and capacitor a modifying signal or signals in phase or antiphase with the oscillator output signal, and means for generating the modifying signal using a bridge circuit or other initially balanced circuit excited fran the oscillator output signal so that any unbalance produces a signal whose amplitude is related to the degree of unbalance and whose phase is either the same as or the inverse of the oscillator output signal depending on the direction of unbalance.

   CLAIM 2 A system as claimed in claim 1, wherein the resonant circuit comprises an inductor and a capacitor connected in parallel and where the output signal fran the oscillator is derived from the voltage across the tuned circuit and where the modifying signals consist of mutually antiphase voltages applied to the low potential ends of the inductor and capacitor respectively so that the effective inductance and capacitance are increased or reduced together.

   CLAIM 3 A system as claimed in claim 1, wherein the resonant circuit comprises an inductor and a capacitor connected in series and where the output signal fran the oscillator is a voltage proportional to the current through the tuned circuit and where the modifying signals consist of a current injected into the common connection of the capacitor and inductor together with two other currents each half the amplitude and in opposite phase to the first current and connected to the outer ends of the inductor and capacitor respectively so that the effective inductance and capacitance are increased or reduced together.

   CLAIM 4 A system as claimed in claims 1, 2 or 3, incorporating switching means for disconnecting the modifying signal or connecting it in normal or reverse sense so that the oscillator frequency changes fran the natural frequency of the resonant circuit to frequencies above or below the natural frequency with the transition fran one frequency to another occurring without discontinuity.

   CLAIM 5 A system as claimed in claim 4, wherein the modifying signal is connected alternately in normal or inverse sense by a switching system which is actuated every time a predetermined integral number of cycles of oscillation has been completed thereby also generating a rectangular waveform for which the ratio of mark to space is equal to the ratio of the frequencies corresponding to normal and inverse connection of the modifying signal.

   CLAIM 6 A system as claimed in claim 4 and 5, wherein the oscillator output signal is applied simultaneously to a number of bridge circuits or other initially balanced circuits and where the modifying signal to the oscillator is switched to the output or inverted output of each bridge circuit in turn thus allowing monitoring of a number of different bridge circuits on a time multiplex basis.

   CLAIM 7 A system as claimed in claim 5, where an analogue voltage directly proportional to the degree of bridge unbalance is obtained at the output of a low pass filter connected to the rectangular wave generated as described but with the two voltage levels of the rectangular wave set equally above and below the common supply potential so that the output is zero when the bridge is balanced.

   CLAIM 8 A system as claimed in claim 5, wherein a fast response unsmoothed analogue measure of bridge unbalance is provided by using the rectangular waveform generated as described to control an integrating circuit which is reset at the start of each cycle and runs up at a constant rate during the mark period and down at the same rate during the space period and where a sample-and-hold circuit is used to store the value which the integrator output has reached at the end of each cycle thereby producing a continuously updated output signal which provides a measure of the bridge unbalance.

   CLAIM 9 A system as claimed in claim 5, wherein a digital readout of bridge unbalance is provided by using the rectangular waveform generated as described to control a reference oscillator connected to a bidirectional counter circuit which is zeroed at the start of a cycle of the rectangular waveform and counts up during the mark period of the rectangular waveform and then down during the space period of the rectangular waveform so that the count at the end of the space period is a measure of the deviation fran a one to one mark to space ratio and thereby also a measure of the bridge unbalance, and optionally means for varying the period of the reference oscillator in proportion to the total period of the rectangular waveform thereby providing a linear relationship between bridge unbalance and count.

   CLAIM 10 A modulated oscillator circuit substantially as described with reference to any one of the accompanying drawings.