GB2248522A - Slot antenna with dielectric coupling elements - Google Patents

Abstract

An antenna (10) incorporates a substrate (12) having a ground plane (14) formed with a radiative slot (16). A microstrip transmission line (18) provides a balanced signal food to the slot (16), and is located on the substrate surface (17). The ground plane is arranged against a dielectric coupling layer (20) of material with a dielectric constant of 10 and is a quarter wavelength thickness. An antireflection layer (22) of polymethylmethacrylate surmounts the coupling layer (20). Radiation coupling from the radiative slot (16) to free space is predominantly via the coupling and antireflection layers (20, 22), very little radiation being detectable at the substrate surface (17). The antenna device (10) may incorporate signal processing components on the substrate surface (17), most of which remains available for this purpose. The antenna (10) is intended as a building block for a phased array radar. <IMAGE>

Description

ANTENNA DEVICE This invention relates to an antenna device, and more particularly but not exclusively to an antenna device for use as a building block for constructing a phased array radar.

Phased array radars are mown. One such, known as "MESAR", was disclosed at a conference entitled RADAR-87, at London, England 19-21 October 1987.

It incorporated an array of nine hundred and eighteen radiating elements arranged in a square of side 1.8 metres. A viable phased array of like construction having four faces and fifteen hundred elements per face is estimated to cost in the order of two million pounds sterling. The complexity of MESAR demonstrates the difficulty and expense of constructing phased array radars.

For most applications it is important that a phased array should be efficient, and have a broad bandwidth (ie > 10%). In order to achieve this, each phased array element should ideally have a far field radiation pattern centred on a common forward direction. The element pattern shape should be close to a cosine curve over a range of scan angles to be covered by the complete array.

The cosine shape inhibits active mutual coupling between array elements.

Limiting the pattern to the forward direction prevents the formation of a second beam pointing backwards when the array elements are working together in the array. Furthermore, in a square array, it is important that the separation between elements be approximately Xo/2, where XO is the free space wavelength at the operating frequency. If the separation is greater than Xo/2, extra beams, known as grating lobes, will appear in the forward direction. Consequently each element and its associated circuitry should be realisable within a square of side X,/2 approximately.To clarify this, consider a phased array with an operating centre frequency fO and bandwidth 2af. The operating band is then from fO + 3f (high band limit) to fO - bf (low band limit).The spacing S between adjacent array element centres is chosen to be Xo/2, where XO is the free space wavelength at the centre frequency fO. At the high and low band limits, corresponding wavelengths X1 and X2 are equal approximately to XO - 6X and XO + 6X respectively, where 6X = XoAf/fo. The spacing S of Xo/2 will therefore be greater than that required at the high band limit and smaller at the low band limit. This prescribes the operating regime. Grating lobes restrict the scan angle at the high band limit, and losses arising from mutual coupling between array elements and associated impedence mismatching become important at the low band limit.

For convenience and cheapness, the radiating elements of a phased array should be simple, eg dipoles. The disadvantage of a dipole is that, when radiating in a uniform medium, it radiates equally in both forward and reverse directions. A phased array of dipoles would produce two output beams with respective centres on opposite sides of the array. This would waste power at least, and could produce damage to components behind the array. To eliminate the unwanted output beam a ground plane may be arranged behind the array dipoles. This is described by Kinzel, J.A., Edwards, B.J. and Rees, D., in Microwave Journal, January 1987, pages 89-102. It is also possible to arrange dipoles on a dielectric halfspace into which they radiate preferentially, as described by Brewitt-Taylor, C.R., Gunton, D.J., and Rees, H.D., Electronics Letters, vol.

17, pages 729-731, 1981. The disadvantage of these approaches is that they result in undesirably complex or bulky devices. A convenient compact type of microwave circuit is the microstrip variety, which is a form of waveguide comprising a metal strip separated from a ground plane by a dielectric substrate.

It would greatly simplify production and reduce costs if an individual radiating element could be formed on one side of a microstrip substrate and be arranged to radiate through the substrate. In the case of a metal dipole radiating element on one side of a substrate, this would require removal of a substantial region of the ground plane on the opposite side of the substrate to provide a window through which radiation could pass. Loss of this ground plane region reduces the area available for microwave circuitry. In consequence, a larger substrate area is required to accommodate a given amount of circuitry. This conflicts with the requirement to reduce area to enable construction of phased arrays with half-wavelength inter-element spacing.

It is an object of the present invention to provide an alternative form of antenna device.

The present invention provides an antenna device including: (1) a dielectric substrate sheet having first and second surfaces separated by a sheet thickness less than one eighth of a device operating centre wavelength in the substrate material, (2) a ground plane formed with a radiative slot on the first surface, (3) means on the second surface for feeding signals to and/or from the radiative slot, (4) a dielectric coupling element adjacent the first surface and having sufficiently great dielectric constant and thickness to provide for radiation coupling to or from the radiative slot to be predominantly through itself, and (5) a dielectric matching element having quarter wavelength antireflection characteristics and arranged to enhance radiation coupling between the coupling element and free space.

The invention provides the advantage that the radiating element, ie, the radiative slot, is located between the coupling element and substrate away from an air interface at the second substrate surface; in this location, since the substrate is thin, and by virtue of the coupling and antireflection elements, the slot is efficiently radiation coupled to free space on substantially one side only.

Furthermore, much of the substrate second surface remains available for mounting microwave components because the ground plane remains largely intact.

In a preferred embodiment, the coupling element has a dielectric constant greater than 9 and a thickness substantially equal to a quaner of the device operating centre wavelength in the coupling element material. The coupling element may be of greater thickness than this if the penalty of increased bulk is acceptable.

The antenna device may include a metal plate having through its thickness a slot which is larger than the radiative slot. This plate slot would incorporate within it at least part of the coupling element and be located over and against the radiative slot to provide antenna radiation coupling properties arising from the resonance characteristics of both slots. To facilitate this, the coupling element may be shaped as a sheet having an upstanding projection with formations complementary to those of the plate slot.

The antenna device may be assembled within conducting walls inhibiting radiation coupling to the radiative slot in directions parallel to the ground plane.

Signals may be fed to and from the radiative slot by means of a dual balanced feed in the form of a microstrip transmission line. Alternatively, a single feed may be employed. The antenna device may also include signal processing components mounted on the substrate second surface. Such components include isolators, mixers, amplifiers, and digital circuits.

In order that the invention might be more fully understood, embodiments of it will now be described with reference to the accompanying drawings, in which: Figure 1 is a perspective drawing of an antenna device of the invention shown in disassembled form; Figures 2 and 3 illustrate microstrip transmission line feeds for activating respective radiative slots as incorporated in the Figure 1 device; Figure 4 is a perspective view of an alternative embodiment of the invention in disassembled form and incorporating a slotted plate; Figure 5 is a graph of relative return signal strength against frequency for the Figure 4 device; and Figure 6 provides graphs of gain against angle for the Figure 4 device.

Referring to Figure 1, an antenna device 10 is shown in disassembled form; ie parts are shown withdrawn to the upper left and lower right respectively for ease of inspection. The device 10 incorporates a microstrip substrate 12 of alumina sheet material with a dielectric constant of 10 approximately. The substrate 12 is a square with 5.08 cm sides, and is 0.64 mm thick. A second substrate 12a of like material is cemented to the substrate 12 and extends perpendicularly from it. The substrates 12 and 12a have left facing and under surfaces respectively (not shown) ~ bearing metal ground planes indicated in position by respective arrows 14 and 14a. A central region of the ground plane is removed to define a rectangular slot 16 indicated by dotted lines. The slot 16 exposes the alumina substrate material but does not extend into it.The slot dimensions are 0.3 cm in width and 2.1 cm in length. The substrate 12 has a right-facing surface 17 bearing a microstrip transmission line 18, the line 18 having arms 18a and 18b and also extending over the second substrate 12a.

The antenna device 10 includes two rectangular dielectric blocks 20 and 22 for radiation coupling and antireflection purposes respectively, as will be described later. The blocks 20 and 22 are assembled in contact together within an open-ended aluminium casing 24, and are consequently indicated partly or wholly by chain lines. The coupling block 20 is for assembly flush against the slot 16, and is 0.854 cm in thickness. It is of PT1O, a proprietary material manufactured by Marconi Electronic Devices Ltd, a British Company. It is composed of a mixture of alumina and titanium dioxide ceramic materials bound by polystyrene, and has a dielectric constant of 10. The antireflection block 22 is of polymethylmethacrylate with a dielectric constant of 2.4, and is 1.73 cm in thickness. Both blocks have square surfaces such as 20a which are 4.2 cm on a side.

The casing 24 has a wall thickness of of 0.82 cm, and is closely fitting over the blocks 20 and 22. In sections (not shown) parallel to the block face 20a, the casing 24 is internally and externally square with sides 4.2cm and 5.84cm respectively. The casing 24 is 2.64 cm in length, this being the sum of the thicknesses of the blocks 20 and 22. Since the casing 24 is open-ended, radiation of appropriate wavelength incident on the exposed block face 20a near the perpendicular passes through both blocks to free space. The casing 24 has an end surface 24a which is flush with the coupling block surface 20a. When the antenna device 10 is assembled, the metal ground plane 14 on the substrate 12 is flush against the surfaces 20a and 24a, and the slot 16 is symmetrically located as regards its spacing from the sides of the casing 24. The casing 24 has screw holes 25 for assembly purposes.

The antenna device 10 also incorporates a terminal block 30 incorporating two coaxial sockets 32a and 32b. The terminal block 30 is hollow and open-ended, and has a wall thickness of 0.6 cm. It is of square section with internal and external sides of 4.64 cm and 5.84 cm respectively, and is 2.65 cm long. As indicated by recessed corner regions (not shown) which is recessed to accommodate and retain the substrate 12. It has slightly projecting side surfaces such as 30b which are sufficiently proud to be flush with the ground plane after assembly. It has screw holes 33 in corner locations equivalent to the like in the casing 24. The second substrate 1 2a is insertable within the terminal block 30.

To assemble the device 10, the casing 24 and terminal block 30 are brought together sandwiching the substrate 12. They are screwed together by screws (not shown) inserted in the screw holes 25 and 33. The substrate 12 is countersunk into the block 30 so that its ground plane 14 and the block 30 have surfaces in a common plane which is flush against the coupling block surface 20a and casing surface 24a. This electrically connects the ground plane 14 to the casing 24 and to the terminal block 30. The ground plane 14a of the second substrate 12a lies flush against an interior lower surface of the terminal block 30, and the transmission line arms 1 8a and 1 8b are connected to the central conductors (not shown) of the sockets 32a and 32b.

The antenna device 10 is designed for operation in the region of 2.8 GHz.

This corresponds to a free space radiation wavelength Xo of 10.7 cm. The radiation wavelength As in the substrate 12 is given by:

where E5 is the dielectric constant of alumina. Similarly, the wavelengths X1 and X2 of radiation in the media of the coupling and antireflection blocks 20 and 22 are given by:

where e2 is the dielectric constant of perspex; ie e2 = 2.4. -Equations (2) and (3) give X1 = 3.4 cm and X2 = 6.9 cm.

The thicknesses of the radiation coupling and antireflection blocks 20 and 22 are 0.854 cm and 1.73 cm, which correspond to X1/4 and X2/4 respectively. The blocks 20 and 22 are therefore each approximately a quarter wavelength in thickness at an operating frequency of 2.8 GHz.

The thickness of the substrate 12 corresponds to Xs/53 approximately, ie a lot smaller than Xs18.

The antenna device 10 operates as follows. A power source (not shown) is connected to the sockets 32a and 32b. This supplies power to the microstrip transmission line 18, which provides a balanced microwave feed to the slot 16.

The slot 16 in the metal ground plane 14 acts as a microwave radiating element.

It can be shown that a slotted ground plane at an interface between two semi-infinite media will have a resonant slot length L given by:

Where Ex and Ey are the dielectric constants of the media on respective sides of the ground plane and slot. The expression semi-infinite is employed to indicate that the medium is bounded by the ground plane, but extends infinitely to one side of it.

For PT1O material and alumina, the dielectric constants are both 10, whereas for air the value is unity. If L1 and L2 are the resonant lengths for a slot bounded by air and PT1O and alumina and PT1O respectively, these media being semi-infinite, then

where XO is 10.7 cm, the free space wavelength at 2.8 GHz.

In the case of the slot 16, there are media of finite thickness on either side, these being the substrate 14 of thickness Xs/53 and the coupling block 20 of thickness X1/4. Beyond these there is air and the antireflection block followed by air respectively. The effect of these non-infinite media is to produce a resonant slot length between L1 and L2 but closer to L1, this length being 2.1 cm as has been said. This resonant length is within 10% of L1, and the difference is attributed to the effect of the thin (us/53) substrate 14 distancing the slotted ground plane from air.

Power fed to the slot 16 by the microstrip transmission line 18 is radiated by the slot. The radiation passes predominantly into the coupling block 20 of PT1O material (X1/4 in thickness and E1 = 10). This is because the substrate 12, being less than Xs/8 in thickness, has very little effect on the radiation pattern of the slot 16. However, the coupling block 20 is A1 /4 in thickness, and exerts a substantial effect on the slot radiation pattern. In consequence, the slot radiation pattern is similar to that appropriate to a radiating element at the interface between semi-infinite PT1 0 and air media, since air is on one side of the substrate 12 and PT1 0 is on the other.A radiating element located at such an interface will exhibit radiation coupling which is predominantly via the material of higher dielectric constant, ie via the block 20 for which E1 = 10.

This effect is disclosed by C.R. Brewitt-Taylor, D.J. Gunton and H.D. Rees in Electronics Letters Vol. 17 pages 729-731, 1981. To achieve the effect, the coupling block 20 must be in the region of or greater than X1/4 in thickness.

An appropriate minimum value would be X1/5.

The antireflection block 22 inhibits reflection at the interface between the coupling block 20 and air. The criteria to be met to provide antireflection properties are that the layer should be substantially one quarter of a wavelength in thickness, and that its dielectric constant should be substantially equal to the geometric mean of the dielectric constants of the media on either side.In the present case, the media on either side are air (ca = 1) and PT10 (fl = 10), so ideally the dielectric constant eAR of an antireflection element would be given by:

In the present example, E2 = 2.4 is a sufficiently close approximation to 3.16 to achieve adequate antireflection properties; these properties are not very sensitive to departures from Equation 4, and r2 may acceptably lie anywhere in the range 2 to 4. The thickness of the antireflection block 22 is however more critical.

It should be within 10% of one quarter of the wavelength X2 in its material.

Tests on the antenna device 10 of Figure 1 have indicated that about 97% of the power input to the transmission line 18 at a frequency of 2.8 GHz from a network analyser is radiated from the slot 16 to free space via the blocks 20 and 22. Measurements in the magnetic H-plane indicate that the gain of the antenna device 10 is a maximum of +6 dB perpendicular to the plane of the slot 16 and to the left in Figure 1. The radiation pattern is close to the ideal cos8 curve over angular regions +70 in the E plane and +35 in the H plane.

Furthermore, the bandwidth of the antenna device 10 is particularly good. The device 10 radiates at least 95% of its input power over a 13% bandwidth. This bandwidth is defined as that over a frequency interval where the signal transmitted by the slot 16 is at least 14 dB above that reflected by the slot back to the microwave power source to which the line 18 is connected. This compares with a bandwidth of 3-4% for a patch antenna, which also incorporates a ground plane. It is comparable to the bandwidth of a metal dipole at the interface between two media, and which suffers from the disadvantage of not providing a usable ground plane.

In addition to the foregoing performance benefits, the antenna device 10 exhibits the important advantage that the ground planes 14 and 14a also serve as ground planes for microstrip circuitry on the reverse sides of the substrates 12 and 12a.

The microstrip transmission line conductor 18 operates in conjunction with the ground plane 14 and 14a, and the latter are also available to co-operate with other microwave components (not shown) mounted on substrate reverse sides.

The slot 16 accounts for less than 4% of the upper surface of the substrate 12.

Consequently, the substrate 12 retains almost all of the capacity for incorporating microwave circuitry and components which it would have had in the absence of the slot 16. Furthermore, the slot 16 is sandwiched between layers 12/20 of solid dielectric material, and the dielectric surface of the substrate 12 is exposed to air for convenient component mounting. The second substrate 12a adds to the space available for component mounting. If required, a third substrate may be cemented to the substrate 12 near the latter's upper end, the third substrate extending parallel to the second substrate 12a.

For comparison purposes, consider a metal dipole on the underside of a microstrip substrate radiating through the substrate thickness and through a ground plane on the other surface of the substrate. To avoid reflection at the ground plane, a window slot would be required to be opened through the ground plane. This would consume 25% or more of the available substrate area.

Circuitry would be required to be distributed outside the window region. The need for a window slot consequently increases the area necessary for a given microwave circuit as compared to that for an antenna device of the invention.

An important application of the invention is in the field of phased array radar, in which the spacing of adjacent antenna elements is required to be )o/2 for efficient radiation reception and transmission. This cannot be achieved if the microwave circuitry associated with each antenna in the array has too great an extent across the array plane. Ideally, each antenna and its associated circuitry in a phased array of antennas would occupy a square in the plane of the array less than or equal to Xo/2 in extent.

Referring now to Figure 2, in which parts equivalent to those previously described are like referenced, there is shown a plan view of a modified form of substrate 12. It is as shown in Figure 1 with the addition of a second microstrip transmission line 42 extending between two plated-through holes 44 in the substrate 12. The plating connects the line 42 to the ground plane 14, and the line length between the centre of the slot 16 and either hole 44 is one quarter of a wavelength at the device operating frequency. In consequence, the line 42 acts as an open circuit. It is illustrated to demonstrate that two microstrip lines may be electrically coupled to the slot 16.

The lines 18 and 42 may be arranged to define separate transmit and receive circuits. In this case, the line 42 would be extended as indicated by dotted lines 46 and over a third substrate indicated by chain lines 48 cemented to the substrate 12 and extending perpendicular to it. The plated holes 44 would be replaced by PIN diode switches so that a receive circuit connected to the line 42 would be isolatable from power fed to the slot 16 in transmit mode.

Referring now to Figure 3, in which parts equivalent to those previously described are like-referenced, there is shown a further alternative form of microwave feed to the slot 16. The substrate 12 is shown from the ground plane inside. The arrangement is as illustrated in Figure 1, except that the balanced feed line 18 has been replaced by a single ended feed line 50 (also of microstrip form). The line 50 is shown dotted because it is on the opposite side of the substrate 12 from the ground plane 14. It terminates in a plated-through hole 52 providing a short circuit to the ground plane 14 close to the slot 16.

A funher embodiment of the invention is shown in Figure 4, and is indicated generally by 100. It is similar to the device 10 of Figure 1, and equivalent parts are like referenced with the prefix 100. There is a single ended power feed line 150 and a shorted output line 142 as illustrated in Figures 2 and 3.

The length of the slot 116 is 1.9 cm, 10% less than that of the slot 16. It is however 0.3 cm wide as before. The most important difference between the devices 10 and 100 is that the length of the casing 124 is reduced by 0.3cm, and that the thickness of the coupling block 120 has also been reduced by 0.3 cm except for a central outwardly projecting finger 101 3.1 cm long and 0.8 cm wide. Necessarily, the finger 101 is 0.3 cm thick. The antenna device 100 also includes a metal plate 103 0.3 cm thick having a central slot 105 with dimensions and orientation as those of the finger 101.

When the device 100 is assembled, the plate slot 105 fits over the finger 101 of PATIO dielectric material. The plate 103 is sandwiched between the casing 124 and the ground plane 114 on the unseen surface of the substrate 112, and lies flush against both.

For testing purposes, the antenna device 100 was connected to a network analyser of commercially available kind. The analyser provides a signal of prearranged frequency and power to a device under test, and it measures the strength of the signal returning from that device. The measured strength is an indication of the signal transmission to free space, since the greater the transmitted power the less the return signal strength will be. The test results are shown in Figure 5, this being a graph of 10 log1 O(Pr/PO) against frequency in GHz. Here PO is the analyser output power coupled to the microwave transmission line 150 and thence to the radiative slot 116. Pr is the return signal reflected to the analyser from the radiative slot 116.

Figure 5 shows that the return signal Pr is at a power level at least 15 dB below that of the output signal PO in the frequency interval 2.8 to 3.35 GHz.

The -15 dB level is indicated by a chain line. Pr is therefore at least thirty times weaker than PO over an 18% bandwidth centred on 3.075 GHz. Very little power is absorbed in the antenna device 100, and in consequence it can be inferred that at least 97% of the source power PO in this bandwidth becomes radiated by the slot 116 to free space via the coupling and antireflection bocks 120 and 122. Tests show that power radiated in other directions is insignificant.

This demonstrates that the antenna device 100 is well matched to free space in the 3 0Hz region.

Figure 5 also indicates the collective effects produced by reduction in the radiative slot length to 1.9 cm and introduction of the slotted plate 103. As compared to the Figure 1 device 10, the Figure 4 device 100 has a centre band frequency shifted from 2.8 0Hz to 3.075 GHz, and a bandwidth increased from 13% to 18%. The increase in resonant frequency is due to reduction in radiative slot length. The bandwidth increase is due to the combination of the resonances of the radiative and plate slots.

Referring now to Figure 6, there are shown graphs of the gain (dB) of the antenna device 100 measured at 2.8 GHz and plotted as a function of angle relative to a boresight direction perpendicular to the radiative slot 116.

Measurements were made in the E and H planes respectively parallel and perpendicular to the width dimension of the radiative slot 116. Graphs 121 (chain line) and 123 (dotted line) represent the variation of gain with angle 6 relative to boresight for the E and H planes respectively. A solid line 125 indicates a calculated variation of gain proportional to cosy. An ideal radiating element for use in a phased array would have a cos6 dependence of gain. The element response would then be a maximum on boresight and would be zero in the plane of the phased array towards neighbouring elements. No practical radiating element exhibits this ideal gain curve. It can be seen that the gain of the device 100 in both the E and H planes shown by graphs 121 and 123 is close to the ideal cos 6 curve over the whole angular range from -90 to +90'.

Graph 121 is within +1 dB of graph 125 over at least -70 to +70', and the equivalent for graph 123 is -35 to +35 The antenna devices 10 and 100 of Figures 1 and 4 illustrate the characteristics obtainable with only two elements 20/22 or 120/122 between the respective radiative slot 16 or 116 and free space. At the expense of increased complexity, it is also possible to employ a structure of multiple elements in which the dielectric constants varies. Adjacent the radiative slot the dielectric constant would be substantial to produce good coupling, and the dielectric constants and thicknesses of successive layers would be appropriate in combination to provide antireflection properties. It is also possible to employ a plurality of plates similar to the plate 103, these having slots of differing sizes and arranged to provide further bandwidth broadening.

Claims (8)

1. An antenna device including: (1) a dielectric substrate sheet having first and second surfaces separated by a sheet thickness less than one eighth of a device operating centre wavelength in the substrate material, (2) a ground plane formed with a radiative slot on the first surface, (3) means on the second surface for feeding signals to and/or from the radiative slot, (4) a dielectric coupling element adjacent to the first surface and having sufficiently great dielectric constant and thickness to provide for radiation coupling to or from the radiative slot to be predominantly through itself, and (5) a dielectric matching element having antireflection characteristics and arranged to enhance radiation coupling between the coupling element and free space.

2. A device according to Claim 1 wherein the coupling element has a dielectric constant greater than 9 and a thickness substantially equal to a quarter of the device operating centre wavelength in the coupling element material.

3. A device according to Claim 1 or 2 including a metal plate having slot through its thickness, this slot incorporating within it at least part of the coupling element and being located over and against the radiative slot to provide antenna radiation coupling properties arising from the resonance characteristics of both slots.

4. A device according to Claim 3 wherein the coupling element is shaped as a sheet having an upstanding projection with formations complementary to those of the slot in the metal plate.

5. A device according to any preceding claim assembled within conducting walls which inhibit coupling of radiation to the radiative slot in directions parallel to the ground plane.

6. A device according to any preceding claim wherein the means for feeding signals to and/or from the radiative slot is a dual balanced feed.

7. A device according to Claim 6 including means for processing signals received from free space by the radiative slot, the processing means being disposed on the substrate second surface.

8. An antenna device substantially as herein described with reference to Figure 1 or 4.