GB2214742A - Noise-controlled frequency-modulation signal detector - Google Patents

Abstract

A noise-controlled frequency-modulation signal detector comprises a quadrature detector 4 turnable to resonate at noise frequencies within a specific band at an input thereto and to produce two 180 DEG out-of-phase outputs, a differential amplifier 22 to which d.c. components of the outputs are fed (via loop filter 21) and which is arranged for producing a control signal in response to a difference between said components, and means responsive to said control signal for retuning the quadrature detector 4 to a point of maximum energy distribution within the specified frequency band. The detector includes a parallel resonant circuit shunted by varactor diodes controlled by the feedback signal. This gives compensation for temperature drift and allows stable operation of the detector, since it remains locked to its input frequency even in the absence of received signal. If noise levels are sufficient to drive the discriminator, the control loop will cause it to settle at a frequency where the mean energy density on either side is equal. Manual adjustment 24 may be required if the energy is not uniformly distributed. <IMAGE>

Description

N < ;)ISE C: IROLIED FREOUZCY MODULAIION SIGNAL Dk-lECI^OR This invention relates to a noise controlled frequency modulation signal detector. It relates particularly to a tunable quadrature detector in which a means of compensation for drift, particularly temperature drift, is provided.

Quadrature detectors (also called discriminators) are used to recover the modulation frequency from a frequency modulated (f.m.) radio signal. Such detectors very often use as the phase sensitive network, a tunable circuit, the circuit being tuned so as to be at resonance with the intermediate frequency (i.f.) to be demodulated.

Unless special precautions are taken the resonant frequency of the circuit, after it has been tuned, may tend to drift, for example, with temperature changes. At the higher intermediate frequencies, 40MHz to 80MHz, the temperature sensitivity problem becomes more acute, especially where an ability to function effectively throughout a military temperature range is an operational requirement.

The drift compensation means which are available at the present time for tunable quadrature detectors, rely on the compensator effecting a change in the frequency of the local oscillator in order to maintain the correct intermediate frequency value. This compensation method relies on feeding back part of the output voltage in the form of a control voltage of the correct phase and magnitude in order to effect the desired compensated results.

Examples are disclosed in patent specifications No. EP 0079097, US 4127825 and GB 1421093.

Because of the relation of the feedback control voltage to the frequency to which the output circuit is tuned, the fed back control voltage forces the resonant frequency of the input circuit to approach the signal frequency to within very close limits. In the absence of the signal frequency, therefore, the feedback loop will be broken and with the passage of time the discrimination input circuit will tend to drift off frequency. Restoration of the input signal with its specific carrier frequency will cause the discrimination input circuit resonant frequency to be pulled back to the carrier frequency, provided of course that the original discriminator "free run" drift has not taken the loop control voltage outside the captive area. Outside this area, the control voltage will not be capable of pulling the input circuit resonant frequency back to the value of the signal frequency.

Particularly in a military application, the tendency for the discriminator to drift during a prolonged, or sometimes short, period of signal absence is undesirable. This tendency is also objectionable when frequency hopping is concerned and when wide band digital data is the mode of modulation. The time taken for the discriminator to pull in to the signal carrier can thus frequently corrupt the data signals that are received.

The present invention was devised to prevent the discriminator from drifting off a set frequency in the absence of a discrete radio frequency signal of a specific frequency.

According to the invention, there is provided a noise controlled frequency modulation signal detector, comprising a quadrature detector tunable to resonate at noise frequencies within a specific band at an input thereto and to produce two 1800 out-of-phase outputs, a differential amplifier to which d.c. components of said outputs are fed and which is arranged for producing a control signal in response to a difference between said d.c. components, and means responsive to said control signal for retuning the quadrature detector to a point of maximum energy distribution within the specified frequency band.

Preferably, the detector is balanced, and the d.c. components of the ouput, at the resonant frequency, are equal.

The detector may be arranged to detect the centre of the energy distribution of the input noise signals, the difference in the d.c. components of the outputs being indicative of a displacement of the tuned frequency from the point of maximum noise energy within the predetermined frequency band.

By way of example, a particular embodiment of the invention will now be described with reference to the accompanying drawing, in which: Figure 1 is a circuit diagram of a tuned f.m. discriminator showing the conventional arrangement for ensuring that the circuit remains in tune to its input frequency; Figure 2 shows the circuit of the present invention; Figure 3 shows one form of quadrature detector that can be used in the circuit of Figure 2; and, Figure 4 shows waveforms appearing at different points in the circuit of Figure 3 when in operation.

As depicted in Figure 1, the conventional tuned discriminator circuit comprises a mixer 1 to which the incoming radio frequency signal is fed on a line 2. An output from the mixer 1, at a specific frequency, is delivered through an i.f. amplifier 3 to a detector 4 and this produces an audio frequency output signal on the line 6.

In order to provide the required temperature compensation, the output signal is also fed to a low pass filter 7 and a zero crossover detector 8 to which is applied a reference voltage on the line 9. An output from the crossover detector 8 is applied to a loop gain amplifier 11 and then to a loop filter 12 and a voltage controlled oscillator 13. The output from the oscillator 13 provides a local oscillator signal which is fed with the incoming radio frequency signal on the line 2 to the mixer 1.

The basic operation of this circuit is as follows: The discriminator is temperature compensated over the operating range by the design and choice of components. The local oscillator represented by the oscillator 13 is continuously variable to enable choice of programme on domestic f.m. radios or channel on t.v. receivers. In the event of any local oscillator drift due to external influences, the intermediate frequency will attempt to move in sympathy. However, the frequency change will be detected by the crossover detector 8, which will then generate a control voltage back to the oscillator 13 forcing it to return to the correct frequency. In reverse, a drift in the "on tune" condition of the discriminator will result in the generation of a control voltage which in turn forces the oscillator 13 to a new frequency resulting in a new intermediate frequency.This new i.f. is now accepted by the discriminator as the "on tune" frequency.

Although the above described system works extremely well, there are some disadvantages which may manifest themselves at extended temperature ranges. For example: (1) There is no i.f. frequency without signal and the complete loop is broken, unless the remaining noise can be regarded an an intermediate frequency.

(2) Drift of the discriminator characteristics can only be compensated for by a corrective change of local oscillator frequency which in turn produces an offset i.f. This situation is perfectly acceptable provided the resulting intermediate frequency signal remains within the passband of the i.f. filters, bearing in mind that also the signal sidebands must be accommodated.

(3) This system is difficult to operate when the local oscillator signal is synthesized and it is locked to a master crystal controlled oscillator. Of course, such a synthesized local oscillator signal would eliminate local oscillator drift, but it does not solve the problem of discriminator drift, particularly when large temperature variations are to be considered. A phase locked loop f.m. detector would be able to cope with this situation, but this type of f.m. detector falls outside the scope of the above considerations as it is essentially an untuned detector.

The present invention utilises the principle of the automatic tuning of a quadrature detector. Details of this and an explanation of the relevance of the present invention now follow.

The circuit shown in Figure 2, comprises a mixer 1 to which the incoming radio frequency signal is fed on the line 2. The mixer 1 also has a further input to which a local oscillator 13 is connected.

An output from the mixer 1 is delivered to a first i.f. band pass filter 14 and from there to an i.f. amplifier 3.

From the amplifier 3, and via a second i.f. band pass filter 16 the signal is fed to a limiter and detector 4 which is a conventional balanced quadrature detector. The detector 4 has a first output line 17 upon which a signal consisting of the d.c. component with the detected audio frequency signal superimposed thereon is carried. On a second output line 18, the detector 4 provides a second output which is 1800 out of phase with the first output but which contains similarly the d.c. component with the detected audio frequency signal superimposed thereon. The second output line 18 is also connected to an output lead 19 which provides the audio frequency output signal from the circuit.

The first and second output lines (17, 18) are connected through a loop filter 21 to a differential amplifier 22. The output of the differential amplifier 22 provides a control voltage signal on a line 23 which is applied to the detector 4. A manual control voltage or current can be fed in through a line 24 to provide an additional voltage on the control line 23. This will allow the application of an off set control voltage, when necessary.

In operation of this circuit, since the detector 4 provides two outputs 1800 out of phase consisting of the d.c. component with the detected a.f. frequency superimposed, it is this differential d.c.

voltage which is the feature of interest. When the single tuned circuit is manually tuned to resonate with the i.f. frequency, the differential d.c. component is ideally zero. Subsequent deviation of resonance of the tuned circuit away from the i.f. frequency will produce a d.c. differential voltage change, with a polarity depending on whether the deviation of resonance is above or below the i.f.

frequency. The differential voltage filtered by the low pass filter 21 and after suitable amplification by the differential amplifier 22 is applied to the frequency sensitive components of the quadrature detector 4 as a corrective voltage. This forces the tuned circuit to return to the original resonant frequency, that is the intermediate frequency.

The method described above virtually locks the resonant frequency of the tuned circuit forming part of the quadrature components in this type of detector, to the frequency of the modulated input signals. This implies that the detector resonance will also follow the input frequency should this, for some reason, deviate from the original value.

The present invention thus ensures that, provided certain operating conditions are satisfied, the quadrature discriminator when connected in an f.m. receiver remains locked to its input frequency even in the absence of any signal at the aerial input. The input to the discriminator now consists of random noise frequencies within a total bandwidth determined by the pre-detector filter. Assuming that the energy density of all the noise frequencies is uniformly situated on either side of the passband centre frequency of this filter, the discriminator will lock itself to the centre frequency. This situation will be maintained as long as the levels of the random noise frequencies at its input are sufficient to drive the discriminator beyond the operating threshold level.Any deviation from a uniform energy density over the passband will cause the discriminator to settle at a frequency (not necessarily in the centre of the passband) where the mean energy density on either side is again equal. The manual control signal 24 enables the frequency adjustment of the discriminator to be set automatically anywhere within the passband of the i.f. filter 16. This feature is likely to be required if the filter used is not an "ideal" filter exhibiting uniform energy density over its working passband.

It can be seen from the above discussion that the tuned discriminator can be constructed without special precautions with regard to temperature coefficients in its tuned circuit components, and that this will provide a quadrature discriminator system which will remain on tune within the pre-detector of a i.f. passband over a wide temperature range.

Figure 3 of the accompanying drawing shows the circuit of the quadrature detector 4. In the present embodiment, the circuit was based on a MC1596 balanced modulator circuit chip but of course a similar circuit to carry out the necessary functions could be constructed in several alternative ways.

The circuit comprises two pairs of transistors Ql, Q2 and Q3 Q4 together with two further transistors Qs, Q6. The transistors are connected in a balanced circuit arrangement with a first reference input port 24, having radio frequency signal lines which are 0 and 1800 out of phase with one another. There is also a second reference (+900) input port 26, which is fed with radio frequency signals which are 900 out of phase with those at the first input port 24, and similarly 0 and 1800 out of phase with each other. The circuit also has two output ports 27 and 28, and a bias output lead 29.

The two sets of input signals at the input ports 24 and 26 are therefore balanced.

Consider the case where the signals at the two input ports are in phase. One cycle of signal frequency is shown in Figure 4, and for convenience it is depicted as a square wave, with the four possible kinds of signal A to D. Figure 4A shows the signal at transistors Q1 and Q4, Figure 4B shows the signal at transistors Q3 and Q2, Figure 4C the signal at Qs and Figure 4D the signal at Q6.

Consider first the signal output at output port 27 and with the signals in phase.

First half cycle Q2 off, Q4 on Q on, Q6 off Second half cycle Q on, 44 off Q off, Q6 on Over the whole of the cycle there is no current flowing out of output port 27, and the signal output is therefore at a minimum.

Consider the signal output at output port 28 with the signals in phase.

First half cycle Q1 on, Q3 off Q4 on, Q6 off Second half cycle Q1 off, Q3 on Q5 off, Q6 on There is always a current flow out of output port 28, and this current is therefore at a maximum.

It is seen that with in-phase input signals, the differential current at the two output ports is at a maximum. If the phase difference between the two input signals departs from zero degrees, the average current out of output port 27 rises due to the appearance of current pulses, and the average current out of output port 28 falls due to the appearance of current "holes".

With a 900 phase difference between the two input signals the mark space ratio of the current pulses at the two output ports is unity and their respective average current is equal. This condition results in a differential output current equal to zero. This consideration assumes that all the circuit parameters are perfectly matched and balanced.

As the input signals reach a phase difference of 1800 the differential output current is again at a maximum, but with a sign reversal.

It follows that when the phase difference of the input signals is made to be frequency dependent, having a phase difference of 900 at the wanted signal frequency, a differential output signal, about zero, is obtained when the wanted r.f. signal deviates from the centre frequency.

The output currents are converted into an output voltage by providing the collectors of the combinations Q1 and Q3, and Q2 and Q4 with suitable loads. The radio frequency component at the output is filtered, leaving the audio freqency voltage for further processing.

The quadrature circuit providing the 900 phase shift between the two input ports at the signal frequency contains a parallel resonant circuit, tuned to the wanted frequency. The resonant circuit tuning capacitor is shunted by varactor diodes. This allows the resonant frequency of the tuned circuit, and consequently the phase, to be controlled by a d.c. feed back voltage, derived from the loop amplifier via the loop filter.

The foregoing description of an embodiment of the invention has been given by way of example only and a number of modifications may be made without departing from the scope of the invention as defined in the appended claims. For instance, certain modifications could be made to the circuits as depicted in the drawing which will be well known to those skilled in the art. Thus, audio frequency filters could be fitted optionally in the circuit of Figure 2 before the differential amplifier 21, although the post amplifier loop filter 22 would tend to remove any audio frequency component from the wanted d.c. component. The inclusion of a preamplifier filter would be purely a practical consideration as otherwise the amplified audio frequency component at the output of the high gain loop amplifier might cause problems in adjacent circuits carrying out other functions.

Alternatively, the d.c. component of only one of the output ports 27 or 28 (Figure 3) can be utilised to provide the d.c. feedback control voltage by comparison with a fixed and stable d.c. potential generated externally, the resulting amplified differential d.c. voltage again forming the basis of a correcting feedback control voltage.

Claims (6)

CLATMS

1. A noise-controlled frequency modulation signal detector, comprising a quadrature detector tunable to resonate at noise frequencies within a specific band at an input thereto and to produce two 1800 out-of phase outputs, a differential amplifier to which d.c.

components of said outputs are fed and which is arranged for producing a control signal in response to a difference between said d.c. components, and means responsive to said control signal for retuning the quadrature detector to a point of maximum energy distribution within the specified frequency band.

2. A detector as claimed in Claim 1, in which the two outputs for the differential amplifier are balanced outputs.

3. A detector as claimed in Claim 1 or 2, in which, at the resonant frequency, the d.c. components in the two detector outputs are equal.

4. A detector as claimed in any one of Claims 1 to 3, in which the quadrature detector is arranged to detect the centre of energy distribution of the input noise signals within the predetermined pass band.

5. A frequency modulation signal detector substantially as hereinbefore described with reference to any one of Figures 2 to 4 of the accompanying drawing.

6. Radio communication apparatus including a frequency modulation signal detector as claimed in any one of Claims 1 to 5.