GB2199202A - Electric power regulator snubber circuit - Google Patents

Abstract

A switching voltage regulator such as a step-down (Fig. 3), flyback or boost regulator has a capacitor C1 and diode D1 snubber circuit to assist turn-off of a semiconductor switch S and utilises transformer coupling with the inductive element L to derive a snubber capacitor charging current in a charging circuit 20 without the losses associated with charging resistors. The inductor L is formed as a primary winding Ts1 of a transformer Ts and the charging circuit 20 has two opposed secondary windings Ts2, Ts3, of half the number of turns. Ts3 with diode D3 and capacitor C2 forms a voltage storage circuit and Ts3 is part of, or coupled to, an inductance L1 which with diode D2 effects voltage doubling in charging the snubber capacitor C2. Switch off generates voltage in the storage circuit of Vout/2 and in Ts2 a voltage of Vout/2 which opposes it. Switch-on generates a voltage of (Vin-Vout)/2 in Ts2 which adds to the storage voltage of Vout/2 and applies charging current to the snubber capacitor at Vin/2 which is doubled to Vin as required. The inductive element may be a coupling transformer in a transformer coupled regulator. <IMAGE>

Description

ELECTRIC POWER REGULATOR SNUBBER CIRCUIT This invention relates to electric point regulating or converting circuits and in particular to snubber circuits associated with semiconductor switching elements thereof.

Snubber circuits are well known n switched-mode power regulating or converting circuits wherein a semiconductor switching element controls delivery of current to a load by way of an inductor which, with a capacitor, comprises a low pass filter that removes high frequency variations in supply due to switch-interupted input.

Such circuits fall into two main categories which may be considered as a directly-coupled kind, wherein there is a direct current path between the switching element and output filtering elements including an inductive element and as an indirectly-coupled kind wherein the switching element is decoupled from the output components by a transformer or the like but which itself represents an inductive element.

There are a number of regulating or converting circuits based upon these components which have different functions and hence different names, depending largely upon the points at which current is supplied to, and extracted from the circuit and the position of the switching element in the circuit, the most common types of circuit being the so called step-down or buck switching regulator, which supplies current to a load at a voltage lower than the input voltage, the boost regulator which supplies current to a load at a voltage higher than the input voltage, and the flyback regulator which supplies current to a load at a voltage of opposite polarity to the input voltage.

Although these circuit types are known by various names such as 'regulator' or 'converter' in this specification all such circuit types, which are in themselves well known, are referred to as 'regulators'.

It is convenient to exemplify the problems associated with such regulators by consideration of the directly coupled step-down or buck switching regulator as shown in Figure 1, wherein the symbols L, C and D denote the regulator inductive element, capacitor and rectifier respectively, S the switching element and V. the input voltage and V t the output in out voltage applied to a load.

The known circuit does not require detailed description, but briefly the switching element S is repetitively opened and closed and energy stored in the inductive element due to load current passing therethrough whilst the switch is closed is extracted by the load whilst the switch is open, that is, the inductive element then effectively becomes a generator of load current, the rectifier D providing a return path for the load current by-passing the open switch, so-called freewheeling, effectively clamping the junction between the inductive element and switch to the potential of the common input terminal and preventing the voltage at the switch from swinging to a large negative value. The capacitor C forms a reservoir to help maintain the level of the output voltage. The inductive element L in this case provides a series inductor between input and output and is referred to accordingly.

The level to which capacitor C charges, the output voltage across the load, is a function of the proportion of time that the switch provides current to the inductor, that is, the duty cycle of the switch, and one convenience of this type of regulator circuit is that the output voltage level is so readily controlled by the switch duty cycle.

The switching element is in practice a semiconductor device, such as a transistor or gate controlled rectifier, and is conveniently referred by its functional role simply as a switch. Switching limitations inherent in a semiconductor device may cause performance limitations in the regulator circuit due to an overlap in the buildup of voltage across an opening switch before the current flow therethrough is stopped, the product of the switch voltage drop, current and switching time being a measure of the energy lost per switching cycle as heat in the switching element. The total power lost is the product of the cycle energy and the switching frequency and with high-current devices having low switching speeds and/or at the most useful switching speeds often causes trouble both with regard to energy lost and also the necessity to dissipate the heat generated in the switching element.

It is known, as shown in Figure 2, to pr'- a snubber network from components C1 and D1 which resolves the problem of on-load switching by producing, at the time of switch opening, a source of inductor current alternative to the switch from energy stored on the capacitor C1 enabling the current flow through the switch to fall rapidly and the voltage thereacross to rise at a controlled rate (cependent on the capacitor discharge) minimising the generation of heat in the semiconductor switch. However, the capacitor C1 has to be charged when the switching element is closed (conducting) and to this end a resistor Rl provides current to charge the capacitor C1 to the level of the input voltage.The resistor does of course present an undesirable power loss and although the heat generated therein by the charging current will usually be less troublesome than in the switching element steps still need to be taken to effect its dissipation.

In this respect of providing a snubber to mitigate the effects of switch opening upon the switching device such capacitor-rectifier switching networks are well known for the above described and other switched regulator circuits and such snubber networks do not per se require detailed description here. Resistive charging arrangements, as described above, are undesirable for the reasons given. Proposals have been made to reduce losses associated with snubbing by deriving charging currents by inductive coupling from inductive elements in the regulator but such proposals have relied upon components and configurations which severely restrict the range of switch duty cycles that can be used. Examples and discussions of lossless snubber circuits can be found in U.S. Patent Specification No.

4,365,171 and a paper entitled "An overview of Low-Loss Snubber Technology for transistor converters" by Angelo Ferraro in IEEE Power Electronics Specialists Conference PESC '82 Record. Such so-called lossless snubber circuits provide a starting point for the present invention of which it is an object to provide for a snubber network in a switching voltage regulator a snubber capacitor charging arrangement of simple construction which is substantially lossless and permits greater control range.

According to the present invention, a switching voltage regulator circuit includes an inductive element, a repetitive switching element connected to the inductive element and operable to control the flow of current therein, voltage level clamping means operable to limit the voltage level developed across the inductive element when the switching element is non-conductive, a snubber network, including a capacitor, connected to the switching element and inductive element to provide a temporary source of voltage and current for the inductive element, alternative to the switching element, to facilitate the switching element becoming non-conductive, and a snubber capacitor charging circuit operable when the switching element is conductive to change the charge voltage of the capacitor by not less than the change in limit voltage level at the junction between switching and inductive elements when the switching element becomes non-conductive, the snubber capacitor charging circuit comprising a reference point connected to a part of the regulator circuit, source means operable to derive a voltage, with respect to the reference point, related to the limit voltage prevailing across the inductive element when the switching element is non-conductive, and an inductive charging circuit including a rectifier and inductive means in series between the source means and the snubber capacitor, the inductive means being inductively coupled to the inductive element to develop, when the switching element is conductive, a voltage related to the limit voltage prevailing across the inductive element and provide with the source means charging current to the snubber capacitor at a voltage with respect to the reference point which is not less than half of the voltage change across the snubber capacitor required to charge it, and having a self inductance which effects, with the rectifier, doubling of the voltage at which charging current is supplied thereby to the snubber capacitor.

Embodiments of the invention will now be described by way of example with reference to the accompanying drawings, in which: Figure 1 is a circuit diagram of a known (prior art) basic form of direct-coupled step-down switching regulator, Figure 2 is a circuit diagram of the regulator of Figure 1 including a known (prior art) snubber network with energy-losing snubber capacitor charging resistor, Figure 3 is a circuit diagram of a direct-coupled voltage step-down switching regulator similar to Figure 2 but with the snubber capacitor charging resistor replaced by a lo-ssless charging circuit in accordance with the present invention, Figure 4 is a circuit diagram of a direct-coupled step-down switching regulator employing the components of Figure 3 interconnected differently, Figure 5 is a circuit diagram of a direct-coupled regulator employing the components of Figure 3 interconnected to define a flyback regulator in which the output voltage is of opposite polarity to the input voltage, Figure 6 is a circuit diagram of a direct-coupled regulator employing the components of Figure 3 interconnected to define a boost regulator in which the output voltage is of the same polarity as, but at a higher level than, the input voltage, Figure 7 is a circuit diagram of a direct-coupled step down switching regulator similar to Figure 3 but showing an alternative form of inductive coupling, Figures 8(a) and 8(b) are circuit diagrams of alternative interconnections of components which form direct coupled voltage step-down switching regulators, Figure 9 is a known (prior art) transformer-coupled voltage forward converter switching regulator, Figure 10 is a circuit diagram of a transformer-coupled forward converter of the type shown in Figure 9 including a capacitive snubber network and capacitor charging circuit in accordance with the present invention, Figure 11 is a circuit diagram of a modified form of the regulator of Figure 10 for which component values are optimised, Figure 12 is a circuit diagram of a transformer-coupled flyback regulator and employing a capacitive snubber network and capacitor charging circuit in accordance with the present invention, and Figure 13 is a circuit diagram a direct coupled voltage step down regulator in accordance with the present invention adapted to supply a variable voltage to a load, a Baxendall resonant current switching inverter, and modifications to the snubber capacitor charging circuit to accommodate such a variable voltage load.

Referring to Figure 3 a direct-coupled switching regulator 10 of the stepdown or buck type, described above with reference to Figure 2, comprises a pair of input terminals 11, 12 the terminal 11 being connected to a voltage source V. to in provide input current to a semiconductor switching element S which permits when conductive, that is, when the switch it comprises is closed, input current flow by way of an inductive element, a series inductor L, to an output terminal 13 and through a load circuit 10' operatively connected thereto. The other input terminal 12 is connected to a second output terminal 14 by way of a common line 15 which provided a return path for load current.A shunt capacitor C is connected between output terminals 13 and 14 to define an output voltage level Vout and a so-called freewheel rectifier diode D is connected between the junction 16 of switching elements and series inductor L and the common line 15, the diode being poled for conduction in the direction of current flow between common line 15 and inductor L. The freewheel rectifier diode represents voltage level clamping means which limits the voltage level developed across the inductive element when the switching element is non-conductive and which limit voltage prevails until the switch is made conductive again.

A snubber network 17 comprises snubber capacitor C1 and rectifier diode D1 in series between the common line 15 and junction 16 between switching element and inductor, effectively in parallel with the rectifier diode D and with the diodes D1 and D-poled to conduct in the same direction.

A junction point 18 between the snubber capacitor C1 and diode D1 provides a charging connection point for charging current to the snubber capacitor, enabling it to charge to a positive voltage (with respect to common line 15) and discharge by way of diode D1 when D1 is suitably biased by the voltage of the junction point 16.

The step down regulator thus far described is of known form and operation, the snubber capacitor being charged by undefined means to the voltage conducted by switch S when it is closed, that is, Vin, such that upon opening of the switch the supply of current to the inductor is temporarily satisfied by the snubber capacitor (and which current flows in the load circuit), which enables the switch to stop conducting and the voltage thereacross to change at a controlled rate.

In accordance with the present invention the regulator circuit is provided with a snubber capacitor charging circuit consisting only of essentially non-dissipative components indicated generally at 20.

The circuit 20 comprises a reference point 21 connected to a part of the regulator circuit connected to the non-switched input terminal 12, in this case the common line 15, and a delivery point 22 connected to the snubber capacitor C1 at circuit point 18. The charging circuit 20 comprises two functional parts connected in tandem between the reference and delivery points, namely voltage source means 23 and inductive charging circuit 24. The voltage source means, which is considered further hereinafter, provides a predetermined voltage with respect to the reference point for application to the inductive charging circuit 24 at intermediate point 25.The inductive charging circuit 24 comprises a serial combination of a rectifier diode D2 and inductive means 26, the inductance (11) of which is chosen such that with the snubber capacitor C1 and diode D2 it forms a resonant voltage doubler circuit, that is, a positive-going voltage change occurring between circuit point 25 and common line 15 results in the capacitor being charged towards twice the voltage change.

Thus in accordance with the above described snubber operation the application of a voltage change of V. /2 to the in intermediate circuit point 25 is required to charge snubber capacitor C1 to the level V. necessary for satisfactory in snubber operation.

Furthermore, the application has to be contemporaneous with the switch being closed, that is, the semiconductor switching element being conductive.

In accordance with the invention the charging current at the correct voltage is provided by inductive coupling between the inductive means 26 and the series inductor L whereby a changing current in the series inductor generates a current in the inductive means at a voltage related to the limit voltage prevailing across the series inductor, that is, after any changes due to switch operation have settled.

The inductive means 26 comprises two inductive components, a charging inductor L1 which has the appropriate self- inductance characteristics (11) for the voltage doubler and a secondary winding of a transformer for which the series inductor L forms a primary winding. As the primary winding is in series with the regulator load current it is convenient to refer to the transformer as a series transformer T . The turns ratio of the transformer T is at least 2:1 between S primary and secondary windings.

The transformer primary and secondary windings Tsl and Ts2 respectively are related in winding phase as shown by the conventional dot notation such that a positive voltage applied between circuit points 13 and 16 produces a corresponding polarity voltage between points 25 and charging inductor L1.

The predetermined voltage provided by voltage source means 23 is chosen to be equal to half of the output voltage (out2) In regulator operation, assuming the switch S has been closed for some time and a load current flows and by way of the conductive switch to point 16, that load current also flows through the transformer primary winding Tsl comprising the series inductor and a voltage (VOUt-Vi ) appears acro the in winding. The snubber capacitor is charged to the level of the input voltage V. and, because this is greater than the in voltage (out2) of the voltage storage circuit 23, diode D2 is reverse biased and no current flows from the snubber capacitor to the voltage source means.

If the switch S is opened,current continues to be provided to point 16 and to transformer primary winding T 1 by snubber capacitor C1 which discharges as current is drawn from it and the voltage at the point 16 falls with respect to the output voltage VoUt until it is clamped to 0 volts (the common line 15) by virtue of the negligable voltage drop across the freewheel diode D forward biased by the load current returning to the inductor, that is, the voltage across the transformer primary falls to (Vout -0) which prevails for all or a substantial part of the remaining time for which the switch is non-conductive and it behaves as a generator as energy is extracted therefrom by the load current.

A transformer secondary winding voltage of Vout/2 is generated between point 25 and L1 which cancels out the voltage provided by voltage source means 23 such that no current is drawn from that circuit and the voltage at circuit point 22 is equal to that of the common line 15. Consequently no current flows from supply point 22 to charging point 18 and by way of diode D1 to the load circuit.

When the switch S closes again the voltage across the transformer primary changes to a new limit of (V out -V. ) and the transformer secondary winding T2 generates a corresponding voltage (V out -V. )/2 between circuit points 25 and L1.

This generated voltage is additional to the source means voltage of Vout/2 with respect to the common line 15 such that the voltage applied for doubling and snubber capacitor charging is Vin/2 and the snubber capacitor is charged towards a voltage of V. as required.

in quired.

The charging current thus comprises that transformed from series inductor L and by the voltage source means 23, and it is necessary for the voltage source means to supply the charging current without significantly altering its voltage level.

The voltage source means 23 may take any convenient form but preferably comprises a voltage storage device in which snubber capacitor charging current is replaced by its own charging current also derived by inductive coupling with the series inductor.

The voltage storage device conveniently comprises a storage capacitor C2 having a large capacitance so that its charge voltage does not decrease significantly when snubber capacitor charging current is drawn therefrom. The storage capacitor is connected between the reference point 21, and the circuit point 25 and has connected in parallel therewith a serial combination of rectifier diode D3 and a tertiary winding T53 of the transformer T .The tertiary winding S T53 has half the number of turns of the primary winding and its winding phase relationship with the primary is, as shown by the dot notation, opposite to that of the secondary winding Tis2. The diode conduction polarity is such that a charging current is applied to the storage capacitor C2 to charge it to a voltage of +Vout/2 with respect to reference point 21 and common line 15 when the voltage across the transformer primary Tsl is VoUt, that is, when the switch S opens. When the switch S closes, the transformed voltage is of the opposite polarity but rectifier diode D3 prevents any effect on the charge state of the capacitor.

Thus the change of field in the core of series inductor L, and voltage across the transformer primary formed thereby, upon opening of the switch S reinforces the charge voltage on storage capacitor C2 by way of winding Ts3 and, by way of winding T52, opposes or backs-off this voltage from causing current to flow to the now discharged snubber capacitor C1 and diode D1. A further change of the field in series inductor L upon closing of the switch provides a current in T 1 with a voltage (Vout -V. ) across it and thus current in winding in Ts2 at a voltage in out )/2 between L1 and point 25 -v L1 which, additively combines with the supply at V /2 from storage capacitor C2 to provide a charging current at Vin/2 to charge snubber capacitor C1 to a voltage V.

in The energy for the charging current is provided by changes in current flow in the inductor by substantially loss less transformer coupling, insofar as any inevitable power losses in transformer coupling are significantly less than those resulting from use of a resistor.

Furthermore, the output voltage level Vout is conveniently changed with respect to V. by altering the time in that the switch remains conductive at any switching frequency, that is, its switching duty cycle, and the voltage source means 23 which is required to produce current at a voltage Vout/2 is, by virtue of the transformer coupling, automatically charged to this level irrespective of changes in the duty cycle.

A situation may arise in operation where the output voltage level is reduced by changing the switching duty cycle from a higher to a lower level. Initially the storage capacitor C2 is charged to a higher voltage than required for recharging the snubber capacitor but it will be appreciated that when the switch is opened any excess storage capacitor voltage will overcome that of the backing-off voltage developed across secondary winding T52 and current will flow by way of diode D2 inductor L1 and supplement the snubber capacitor discharge current. Thereafter the voltage generated on the secondary winding will be (V. -V t)/2 at the new value of in out VoUt and with the newly stabilised value of Vout/2 from the storage capacitor will continue to charge the snubber capacitor to a voltage Vin.

The switching step-down regulator described above is only one possible form of step-down regulator and the step-down regulator one of several different types of switching directcoupled regulator employing similar components and snubber network but defined as to type and function by interconnection. Such similar circuits are also suited to include a snubber capacitor charging circuit in accordance with the present invention.

An alternative configuration of a switching step-down regulate~ is shown in Figure 4. Circuit elements and points corresponding to those in Figure 3 are given like references.

This circuit differs from that of Figure 3 in that the rectifier diode D1 and snubber capacitor C1 are interchanged so that when switch S is closed, the snubber capacitor is charged negatively with respect to the voltage V. then appearing at in the circuit point 16. To effect such negative charging with respect to V. the reference point 21 of the snubber capacitor in charging circuit 20 is connected to regulator input terminal 11, the charging diodes D2 and D3 are of opposite conduction polarity from the Figure 3 arrangement and secondary and tertiary transformer windings are also oppositely phased with respect of primary winding to those of Figure 3.

Opening of switch S puts å voltage V Out across the transformer primary between points 13 and 16, which charges storage capacitor C2 of the source means 23 to a voltage of -Vout/2 with respect to V. , and when S closes the voltage generated across secondary winding T52 is (Vout-Vin)/2 between L1 and point 25 which added to the source means voltage gives a snubber capacitor charging voltage at L1 of -V. /2 with respect to V.

in Another well known form of snubbed regulator is the flyback regulator shown in Figure 5. The circuit will be seen also very similar to that of Figure 3 in respect of component polarities and differs therefrom in that the common line 15 of Figure 3 and the common output terminal 14 becomes a negative voltage output terminal 14' and the positive output voltage terminal 13 becomes a common output terminal 13'. The common input terminal 12 is connected to this common output terminal 13' as is the snubber capacitor C1.

The snubber capacitor requires to be charged positively with respect to the common input and output terminals 12 and 13' to provide current to the inductor when the switch S is opened and to continue supplying current until the point 16 feeding the series inductor reaches a negative level when diode D ;clamps the voltage to the output terminal 14'..The voltage across series inductor L between points 13 and 16, the primary winding l of the transformer formed thereby, swings between -V. when S in is closed and VoUt when S is open which is satisfied by 2 snubber capacitor charge voltage of (V. +VoUt) with respect in out to the output terminal 14' and a charging circuit voltage of (V. +V )/2 with respect to that terminal. The reference in out point 21 of the snubber charging circuit 20 is thus connected to the output terminal 14' but diode conduction polarities and transformer winding phases are as in Figure 3.

When the switch S opens the (positive) voltage Vout out between common terminal 13' and point 16 causes the tertiary winding Ts3 to genrate a charging voltage of V Ut/2 which is applied to storage capacitor C2 to charge it positively with respect to the reference point 21 and the secondary winding T 2 generates a voltage of Volt/2 between point 25 and L1, s2 out thus completely backing-off the storage capacitor voltage.

Closure of the switch S connects point 16 to source at V. and in a voltage across the primary winding of (0in) and generates current at voltage of -Vin/2 in the tertiary winding which is blocked by the diode D3 and voltage of Vin/2 between L1 and point 25. A combined voltage exists between L1 and reference point 21 of (Vin/2+Vout /2) so that the snubber capacitor is charged by way of D2 and L1 to a voltage of (V. +VoUt) with respect to the reference point 21 or output in out terminal 14'.

Yet another form of regulator, a so-called switching boost regulator, is shown in Figure 6 which again illustrates the similarity of components. There are some circuit similarities with that of Figure 5, the main departure being that the diode polarities in the snubber network and snubber charging circuit are reversed so that output terminal 14' gives a positive voltage with respect to common input and output terminals 12 and 13' and the switch S and series inductor L (transformer primary T 1) are interchanged in position. The junction between inductor, switch and freewheel diode D is still labelled as point 16. The input terminal 11 is connected directly and permanently to the end of the series inductor remote from the circuit point 16.

Operation of the regulator is such that when the switch S is closed the point 16 is at 0 volts and a voltage -V.

in appears across the inductor between point 16 and input terminal 11 whereas where the switch S is open the point 16 is clamped to V Out by freewheel diode D and a voltage (V -V. ) appears out in across the inductor between point 16 and input terminal 11.

The snubber capacitor is required to provide a current sink until the voltage rises across D sufficiently for it to conduct load current. Accordingly, the voltage at point 16 at switch opening, to which the snubber capacitor requires to be charged, is 0 with respect to common input and output terminals 12 and 13'.

The snubber capacitor charging circuit 20 is connected with the reference point 21 connected to the output terminal 14' so that charging voltages are measured relative thereto, that is, the snubber capacitor is charged to a potential of -VOUt with respect to the output terminal 14' (+tout) The voltage source means 23 is arranged such that opening of the switch charges the capacitor C2 to -(VOUt )/2 with respect to V out and the secondary winding T52 arranged to generate an opposite polarity voltage to back it off. Closure of the switch generates a secondary winding voltage of -V. /2 between L1 and point 25 which, with the source means voltage -(VO t-Vin)/2 between points 25 and 26, gives a snubber capacitor charging voltage -V /2 with respect to V at out out L1 and the snubber capacitor charges to a potential of .Vout with respect to V out at output terminal 14'.

The above described embodiments exemplify the use of the snubber capacitor charging circuit in relation to different types of direct-coupled switching regulator and for ease of description it has been assumed that circuit losses associated with the transforming and charging operations are absent, whereby the voltages generated across the secondary and tertiary windings of the series inductor/transformer are exactly equal in value and are exactly half the level of the voltages appearing across the transformer primary winding.

It will be appreciated that in practice some circuit losses will occur and in this respect the turns ratios between the primary and secondary winding and primary and tertiary windings may be increased to ensure that in operation the theoretically optimum voltage levels are approached.

It will also be appreciated that the above considered voltage level generated in the snubber capacitor charging circuit is the minimum level of snubber capacitor charging voltage, as charging of the snubber capacitor to a lesser voltage will introduce a step discontinuity in inductor voltage when the switch is initially opened. On the other hand, charging of the snubber capacitor to a higher voltage merely feeds current into the load, by way of rectifier diode D1, which has the effect of reducing the current drawn from the input supply but establishes a current 'loop' between the inductor/transformer, charging inductor L1 and diode D1 through which energy may be wasted due to any losses in the transformer operation.

Thus it is preferable for the transformer and snubber capacitor charging circuit in general to produce a snubber capacitor charging voltage which is as close to, but not less than, half of the voltage to which the snubbed capacitor has to be charged.

Similarly the ratio between the secondary and tertiary windings may be other than unity in response to modifications of the basic regulator circuits described.

For example, it is known in the above described regulator circuits that diode types used as the freewheel rectifier D exhibit capacitance-like effects when turned off rapidly by closure of switch S, the effect being to add a current spike to the leading edge of the current waveform in the switch and resulting in component damage and/or radiated electromagnetic emissions. To effect so-called spike suppression recovery for the rectifier it is known to include in series with the freewheel rectifier D a further inductor L2, such as shown ghosted in Figure 3, which buffers the rectifier from the switch and allows the current therein to be reduced at a controlled rate when the switch S closes.

When the switch S opens however, the rectifier D cannot conduct when the potential at the circuit point 16 (the output of the switch) reaches zero. The current in the further inductor is a function of time and the increasing potential across it as the snubber circuit allows the voltage at the circuit point 16 to swing negative and with it the voltage across snubber capacitor C1. As the current in the further inductor becomes equal to the load current the voltage across it collapses to zero but leaving the snubber capacitor correspondingly negatively charged.

To prevent the snubber network diode D2 conducting during this time the 'back-off' voltage generated across the secondary winding Ts2 upon switch opening may be increased to a value greater than that generated across the storage capacitor by tertiary winding T52 by an appropriate difference in their turns ratio, the correspondingly increased voltage across T 2 being used as required to provide adequate charging current for the snubber capacitor C1 from the initially negative to the desired positive voltage when switch S closes. By this means energy stored in the further inductor, and normally lost, is returned to the load circuit.

This and further modifications may be effected to the snubber circuit of the other regulator types described above.

It will be appreciated that as the switch S opens and closes continuously and often at high frequency the voltage changes across the transformer primary provides a consistent source of charging current to the storage capacitor. If desired the voltage storage device of the source means may be other than a capacitor and could, for instance, be a rechargable battery or the like, although such a storage device is not readily suited to regulator output voltage change by simple alteration of the regulator switching duty cycle and is useful only with a fixed output voltage level.

Also, the inductor L need not be formed as the primary winding of a series transformer but as shown in Figure 7 may comprise a simple inductor in parallel with which is connected a primary winding Tpl of a suitable parallel transformer T to p effect inductive coupling with the snubber capacitor charging circuit in which secondary and tertiary windings of the transformer, Tp2 and Tp3 are disposed. Unless the parallel transformer T is of a type designed to handle a continuous p polarising current in its primary without core saturation a blocking capacitor C3 is provided in series therewith to restrict current flow to the changes brought about by switch operation and prevent the passage of load current which would cause such core saturation.

The circuit of Figure 3 or as modified in Figure 7 may effect inductive coupling by a 'tightly' coupled transformer which has minimal losses but requires in the inductive means a discrete inductor L1 for its inductance properties. The circuits described may be simplified by arranging poor coupling between the primary and secondary windings of the series inductor-transformer or separate transformer so that the leakage inductance of the secondary winding functions as the snubber capacitor charging inductance L1 and enables a separate component to be omitted.

It will also be appreciated from Figure 3 that the components of the charging circuit 20, namely the diode D2, inductive means 26, separate inductor L1 and secondary winding Ts2 of part 24 and the whole source means 23 are all two-terminal components serially connected between the reference point 21 and the delivery point 22 and that it is possible to make the series connection with the elements in any order, provided of course that diode polarity and transformer phasing is observed.

As examples of further variations in component arrangements within direct coupled regulator circuits.

Figure 8(a) shows a step down regulator circuit of the form shown in Figure 4 but for a negative supply voltage -Vi , in' the only changes required being reversal of rectifier conduction polarities in the regulator and snubber capacitor charging circuits.

Figure 8(b) shows the circuit of Figure 8(a) simply 'inverted' as to polarity so that the supply voltage is +V.

in' as in Figure 4(a), but the switching element S is in the common line as often preferred for controlling semiconductor switching elements so that controlling voltages can be provided relative to the (more negative) common supply terminal 12.

All of the regulator configurations considered in the above description are of the direct-coupled type in which a series inductor between regulator input and output, in smoothing the output made intermittent by the repetitive switch operation, causes the switching element problems requiring snubbing.

However a common form of regulator is indirectly coupled and includes transformer coupling between the switching and the output current and voltage defining components including such a series inductor where appropriate.

Figure 9 shows a basic transformer-coupled switching voltage regulator also known as a forward converter forming prior art, the regulator configuration being well known and not requiring full discussion here.

A series inductor L' which defines the load current and voltage is in the secondary part of the circuit and as such is not a direct influence on, nor usable to mitigate, switching transients.

The coupling transformer T is c ' however, an inductive element and does in practice exhibit some leakage inductance in its primary winding which causes switching-effects similar to those of the series inductor in the direct coupled regulator which can be mitigated by the use of a snubber circuit.

Furthermore the energy stored in the core of the inductive element may be snubbed without losses by charging a snubber capacitor in accordance with the present invention.

In the known circuit of Figure 9 the coupling transformer T has a primary winding TCl which connects the c input terminals 11, 12 by way of series connected switch S. A secondary or, more conveniently, output, winding T is co coupled to rectifier and output regulation components shown generally at 27 which deliver current to a load 10' at an output voltage V t. The transformer T also has an energy Out c recovery, or clamp, winding TCc connected in series combination with a rectifier diode D across the input terminals.The transformer primary and clamp windings are connected with opposite winding phase, as shown by the dot notation, and a known turns ratio of n : 1 respectively such that when the switch is opened the clamp winding limits or clamps the voltage appearing across the primary winding to n times the input voltage V. appearing across the clamp winding in and also feeds current inductively coupled from the dissipating energy in the transformer primary winding back into the supply.

The voltage developed across the primary winding TCl is V. when the switch S is closed and n.V. for at least in in part of the time that switch S is open. In this form of regulator flux resetting of the transformer core between switch operations is of prime importance to ensure no flux accumulation and consequential core saturation, and the turns ratio between primary and clamp windings to ensure flux resetting is usually chosen on the basis of the maximum duty cycle of the switch S.

when V. is of high value, and particularly if n is in greater than unity, the n-times multiple of this appearing across the switch may aggravate problems with switching a semiconductor switching element, and the switch S may benefit from snubbing by means of a capacitor-rectifier snubber circuit of the type described above. A transformer coupled regulator circuit with a snubber capacitor and charging circuit in accordance with the present invention is shown in Figure 10 and parts thereof corresponding directly with those of earlier Figures are given like references.

In Figure 10, the regulator comprises in addition to the component arrangement shown in Figure 9 a snubber network 17, that is, capacitor C1 and rectifier diode D1 connected in series, connected across the primary winding Tcl of the transformer Tc between points 28 and 29 and a snubber capacitor charging circuit 20 having a reference point 21 connected to the common line terminal 12 and a supply point 22 connected to the junction 18 between the capacitor. and diode of the snubbcr circuit.

fhe snubber capacitor charging circuit is substantially as described above, comprising source means 23 in tandem with inductive charging means 24, but with the transformer windings being auxiliary windings of the transformer corresponding to the secondary and tertiary windings of Figure 3 and conveniently referenced Tc2 and Tic3, and related to each other in winding phase so as to provide assisting rather than opposing voltages for any given polarity of voltage on the primary winding. In conformity with the circuit of Figure 9, the clamp winding T cc is assumed to have a unit member of turns, shown as 1, in relation to which the primary winding TCl has n turns.

The tertiary winding Tc3 of the source means has x turns in relation to the clamp winding and the secondary winding Tc2 of the inductive charging circuit has y turns.

Considering operation of the regulator circuit, when the switch is closed the voltage appearing at the circuit part 29 is zero, that is, the level of the common terminal 12. When the switch is opened the voltage appearing across the primary winding tends to rise in magnitude in the opposite sense to that when the switch is closed but is clamped by the clamp winding T to n.V. , measured in relation to the voltage V. on cc in in the input terminal to which the transformer primary is connected; this represents an actual voltage of (1 + n).V.

in across the switch S.

The energy in the flux remaining in the transformer is, at this primary winding voltage n.V. extracted by charging in the capacitor to the same extent, thereby storing the energy rather than losing it by dissipation within components of the circuit. The voltage appearing between circuit points 29 and 18 is thus n.V. and the voltage at 18 is V. with respect in in to the common terminal 12, that is, the same as the positive supply line and the other end of the primary winding at 28.

When the switch is closed the capacitor terminal at point 29 is caused to fall from (1 + n).V. to 0 volts, in drawing the other capacitor terminal, such fall drawing the point 18 negatively by the same amount, that is, from a voltage of +V. to -n.V.

in in To effect snubbing the snubber capacitor charging circuit is required, whilst the switch is closed, to provide charging current to raise the voltage of the point 18 back to +V. for when the switch opens.

in As described hereinbefore the snubber capacitor charging circuit invokes voltage doubling as part of the charging which means that the source means and inductive charging winding (tic2) has to provide a charging voltage of (Vin+n.Vin)/2 from the most negative voltage reached by the in in point 18, that is, from -n.V. , or a charging voltage V. /2 in in - n.V. /2 with respect to the common input terminal 12.

in If the tertiary winding Tc3 in source means 23 has x = 1/2 turns with respect to the clamping winding T then cc upon switch opening, when the voltage prevailing across the clamp winding is Vin, the storage capacitor C2 will charge to Vin/2 with respect to the reference point and common input terminal, this voltage being available upon switch closure to supplement that of the inductive charging circuit.The transformer secondary winding Tc2 in this part of the circuit has y = n2/2 turns with respect to the clamping winding Tcc When the switch S closes a voltage Vin appears across the transformer primary winding between the circuit points 28 and 29 such that a corresponding voltage n.Vin/2 appears between charging circuit point 25 and inductor L1, this voltage being of opposite polarity to, and therefore substracted from, the voltage on capacitor C2 so that charging current is able to flow through L1 only to the extent that the capacitor terminal is more negative than the charging circuit diode D2 to charge snubber capacitor C1 at a voltage limited by the source and inductive charging means to Vin/2 - n.Vin/2 to charge it to the required level.

It will be appreciated that as with the aforedescribed embodiments the turns ratio between the auxillary windings in the snubber capacitor charging circuit may be other than the specific values given, providing the voltage criteria for snubber charging are met upon the switch opening and closing.

It will b appreciated that as n is chosen with respect to the unit turns of the clamp winding to ensure flux resetting of the transformer core there may or may not be latitude for variations ir its value.

One modification which can be made is the effective elimination of the clamp winding when the sum of the turns of the secondary and tertiary windings is equal to 1, the number of turns of the clamp winding, provided the point between secondary winding Tc2 and the charging inductor L1 is connected to the positive supply point 28 by diode D.

Such relationships must be satisfied in accordance with the snubbing requirements, namely a snubber charging circuit voltage of Vin. (l-n)/2.

in As stated, the transformer winding in the source circuit has x turns and the transformer winding in the charging circuit has y turns. The voltage developed across C2 in the source means when S is open is a fraction x/(x+y) of the voltage V. appearing between the snubber capacitor point 18 and the in common terminal 12, that is, V. and the voltage across winding y when S is closed, and Vin appears across the n turns of the primary, is V. .y/n., so that V. .x/(x+y) - V. y/n = V.. (ln)/2 ....... (1) in in in If the number of turns in the two windings in series is made equal to the unit turn of the clamp winding, that is, x + y = 1 (2), equations (1) and (2) may be solved together to give y = n/2 and x = 1-n/2 It will be seen from the above that when n=2 not only is the energy recovery winding replaced by, or equivalent to, the secondary and tertiary windings of the snubber capacitor charging circuit but also the winding x, and therefore the whole source means, can be dispensed with. Such a circuit arrangement as derived from Fig. 10 is shown in Figure 11, further illustrating elimination of energy recovery winding and source means, by the connection indicated by the broken line 23.

It will be appreciated that in such an optimised circuit where the auxiliary winding y also functions as the energy recovery winding the snubber capacitor charging inductor L1, cannot be formed by the secondary winding.

In other respects the auxiliary windings Tc2 and Tc3 may be windings of a separate transformer (not shown) connected across the primary winding TCl the coupling transformer hereby enabling a simpler transformer construction with fewer windings to be employed.

The use of snubber capacitor charging according to the present invention in transformer-coupled regulator circuits is not limited to the forward converter regulator described above but may also be employed, with modifications required in accordance with the operating philosophy, in other circuits such as the flyback regulator shown in Figure 12.

The flyback converter arrangement is generally similar to the step-down regulator of Figure 10 except that the coupling transformer of the regulator has no separate clamping winding to limit the voltage rise across the primary winding Tcl when the switch is opened but flyback converter the voltage is limited by the output winding T in accordance with the ratio of number co of turns in their windings. The output winding T is co considered as having unit number of turns (1) and the primary winding n times this so that the voltage prevailing across the primary winding when the switch is opened is clamped to n.Vout with respect to the positive line and the snubber capacitor is also charged to this voltage.

The source means also differs in comprising a feedback transformer Tf having a primary winding Tfl of unit number of turns connected to the output part of the circuit, one end of the winding being connected to the output common line and terminal and the other end of the winding being connected by rectifier diode D4 to the other output terminal, in effect to a source of voltage defined by the output capacitor C . The 0 feedback transformer has a secondary winding Tf2 of n/2 turns (with respect to the primary) connected between the charging circuit reference point 21 and the inductive charging portion at 25.

Closure of t" switch S connects the capacitor terminal at point 29 to 0 volts of the common input terminal and forces the other capacitor terminal, at point 18, to -nV t volts.

The snubber capacitor charging circuit is thus required, upon switch closure to charge the capacitor from -n.V to +V. in readiness for the next switch opening.

in As the charging involves voltage doubling a voltage of (V. + in n.Vout)/2 with respect to -n.VOut is required, that is, a voltage of V. /2 - n.V /2 with respect to the common input in out terminal.

It will be seen from the serial connection of the windings in the charging circuit and the conduction direction of the diode D2 that a current path exists between the common terminal 12 and the negatively charged capacitor terminal 18 which will tend to create a negative going voltage across the secondary winding of the feedback transformer as soon as the switch is closed. The increasingly negative voltage is however clamped by the voltage VOUt prevailing across the secondary winding Tf2 and the n/2:1 turns ratio to a voltage of n- /2.

The secondary winding Tc2 of the coupling transformer, which forms the inductive charging means has y = n/2 turns so that when the switch closes and a voltage V.

in prevails across the primary winding TCl (of n turns) a corresponding voltage of V. /2 is developed across the secondary winding Tc2 When the voltages across both windings Tf2 and Tc2 are established charging current is then able to flow into the capacitor by L1 and D2.

The winding polarities are, as shown by the dot notation, arranged such that the clamped voltage developed across the feedback transformer is additive with that developed across the secondary winding to give a desired voltage of (Vin/2 - n.Vout/2) at the inductor L1.

It will be seen that despite the apparent complexity caused by the additional feedback transformer of the source means, the use of the output capacitor as a source of stored voltage with which to clamp the voltage across the primary winding helps mitigate any increase in the number of components of the snubber capacitor-charging.circuit.

The transformer windings of the snubber capacitor charging circuit are employed solely to provide capacitor charging current and as outlined hereinbefore suitable leakage reactance designed into one of the windings may enable the inductor L1 to be eliminated as a separate component. As the coupling transformer T will in general be designed for c optimum transfer with minimal leakage reactance it is preferable that such be confined to the feedback transformer Tf, although the charging circuit winding Tc2 may be a secondary winding of a separate, and possibly loosely coupled, transformer connected in parallel with the primary winding of the main coupling transformer.

In demonstrating the development of snubber capacitor charging current at the correct voltage in the transformercoupled regulator circuits they have, like the direct-coupled regulator circuits, been described with assumed ideal turns ratios. It will also be appreciated that in practice departures from these ideal values may be effected to accommodate losses in other components such as rectifier diode voltage drops and transformer winding resistance and any leakage inductance.

The voltage regulators described above have all been concerned with substantially constant input and/or output voltages and the provision of snubber capacitor charge current at a voltage which is constant by the use, in the source means, of voltage storage means in the form of a capacitor having a large capacitance charged as a function of the input or output voltages.

It will be apparent that providing the relationship between the voltage produced by the source means and the regulator input or output voltage is maintained these voltages may vary with time independently of the switching frequency of the regulator.

As an sample of such a regulator circuit Figure 13 shows a step down regulator 10, to which a d.c. input voltage V. is applied by way of terminals 11, 12, which forms an in input to a resonant current switching inverter 30 of the type developed by P.J. Baxendall and commonly known as a Baxendall push pull oscillator. The oscillator includes two switches S1 and S2 which generally comprise semiconductor switch elements controlled to switch in synchronism from a centre tapped auxilliary feedback or switching winding (not shown) of the oscillator transformer T o A sinusoidal voltage is developed across the transformer T at a frequency determined by the primary 0 winding shunt capacitor C and the inductance of both series connected primary windings.As each switch alternately clamps its half of the primary winding to 0 volts (the common line) the voltage appearing across each switch when open takes the form of a half sine wave and a continuous train of such half sine waves at half the switch voltage appears at the centre tap of the transformer primary winding.

It is known to feed such an oscillator by way of a step-down voltage regulator of the type shown in Figure 2, the input inductor of the oscillator being formed by the series inductor of the regulator. A step-down regulator 10 in accordance with the present invention including a capacitor/rectifier snubber network 17 and snubber capacitor charging circuit 20 may be employed although the aforementioned varying voltage load at the transformer centre tap 31 complicates the provision of snubber capacitor charging current.

The regulator and charging circuit will be seen similar to that shown in Figure 3 and described in detail hereinbefore, series inductor L comprising a primary winding Tsl of m turns of a series transformer T of which a secondary winding Tis2, S of half the number of turns, comprises part of the inductive charging circuit 24 of the snubber capacitor charging circuit.

The source means 23 of the snubber capacitor charging circuit comprises a further centre tapped auxilliary winding Tol of the oscillator transformer To, the centre tap 32 providing the reference terminal 21 of the charging circuit which is connected to the common input line or terminal 12.

The windings are respectively connected by way of individual rectifier diodes D5, D6 as a bi-phase rectifier with the cathode terminals of the diodes commonly connected at 25 to the inductive charging circuit and providing the source voltage.

The voltage as provided by source means 23 is not constant and developed by charging a storage capacitor only upon regulator switch opening but now varies continuously in accordance with the state of the voltage at the oscillator transformer primary centre tap point 31 and the snubber capacitor C1 is automatically charged to the correct regulator output voltage.

Claims (28)

Claims:

1. A switching voltage regulator circuit including an inductive element, a repetitive switching element connected to the inductive element and operable to control the flow of current therein, voltage level clamping means operable to limit the voltage level developed across the inductive element when the switching element is non-conductive, a snubber network, including a capacitor, connected to the switching element and inductive element to provide a temporary source of voltage and current for the inductive element, alternative to the switching element, to facilitate the switching element becoming non-conductive, and a snubber capacitor charging circuit operable when the switching element is conductive to change the charge voltage of the capacitor by not less than the change in limit voltage level at the junction between switching and inductive elements when the switching element becomes non-conductive, the snubber capacitor charging circuit comprising a reference point connected to a part of the regulator circuit, source means operable to derive a voltage, with respect to the reference point, related to the limit voltage prevailing across the inductive element when the switching element is non-conductive, and an inductive charging circuit including a rectifier and inductive means in series between the source means and the snubber capacitor, the inductive means being inductively coupled to the inductive element to develop, when the switching element is conductive, a voltage related to the limit voltage prevailing across the inductive element and provide with the source means charging current to the snubber capacitor at a voltage with respect to the reference point which is not less than half of the voltage change across the snubber capacitor required to charge it, and having a self inductance which effects, with the rectifier, doubling of the voltage at which charging current is supplied thereby to the snubber capacitor.

2. A switching voltage regulator as claimed in claim 1 in which the inductive element comprises, or is connected in parallel with, a primary winding of a transformer.

3. A switching voltage regulator as claimed in claim 2 in which the Primary winding of the transformer is in parallel with the iaducti; element and has in series therewith a direct current blocking capacitor.

4. A switching voltage regulator as claimed in claim 2 or claim 3 in which the inductive means of the inductive charging circuit includes a secondary winding of the transformer.

5. A switching voltage regulator as claimed in claim 4 in which said secondary winding of the transformer has tight inductive coupling with the primary winding and the inductive means includes in series with the secondary winding an inductor having said self inductance to provide said voltage doubling characteristic.

6. A switching voltage regulator as claimed in claim 4 or claim 5 in which the source means provides half the limit voltage prevailing across the inductive element when the switching element is non-conductive and the ratio of turns between primary and secondary transformer windings is at least two to one.

7. A switching voltage regulator as claimed in claim 6 in which the ratio is substantially equal to two to one.

8. A switching voltage regulator as claimed in any one of claims 4 to 7 in which the secondary winding is arranged to produce when the switching element is non-conductive a voltage in a sense which opposes that of the source means to prevent current flow to the snubber capacitor when the switching element is non-conductive.

9. A switching voltage regulator as claimed in any one of the preceding claims in which the voltage level clamping means is operable to limit the voltage level developed across the inductive element as a function of the voltage at the regulator output and the voltage derived by the source means is a function of said regulator output voltage.

10. A switching voltage regulator as claimed in claim 9 in which the inductive element is a series inductor through which load current flows between regulator input and output, said regulator being operable to maintain the output voltage at a substantially constant output voltage whereby the limit voltage prevailing across the inductor while the switching element is in either state is substantially constant.

11. A switching voltage regulator as claimed in claim 9 in which the inductive element is a series inductor through which load current flows between regulator input and output, and which comprises a current source input to a load including a transformer operatively causing variation in the output voltage of the regulator and in which the source means comprises one or more auxilliary windings of said load transformer connected to the reference point and by way of rectifier means to provide a source voltage related to said regulator output voltage to the inductive means.

12. A switching voltage regulator as claimed in claim 9 comprising a flyback converter in which the inductive element is a primary winding of a coupling transformer between regulator input and output circuits and an output circuit winding of the transformer provides said clamping means to limit the voltage level across the primary winding when the switching element is non-conductive and in which the source means comprises a feedback transformer having a primary winding across which the output circuit voltage prevails and a secondary winding connected between the reference point and inductive means, said feedback transformer being operable to clamp the voltage developed across the source means when the switching element is conductive as a function of the voltage appearing across the primary winding of the inductive element when the switching element is non-conductive.

13. A switching voltage regulator as claimed in claim 9 in which the switching regulator comprises a forward converter in which the inductive element is a primary winding of a coupling transformer between regulator input and output circuits and said voltage level clamping means is provided by an auxilliary clamp winding of the coupling transformer in series with a rectifier diode and connected between the regulator input and output terminals.

14.. A switching voltage regulator as claimed in claims 9, 10 or 13 in which the voltage storage means comprises a storage capacity: connected between the reference point and the inductive charging circuit and in parallel with the capacitor a serial combination of rectifier and winding inductively coupled to said inductive element, said inductive coupling phase and rectifier polarity being such that the capacitor is charged by inductive coupling when the switching element is non-conductive and the inductive coupling is such as to define said predetermined level of storage voltage as a function of the limit voltage prevailing across the inductive element.

15. A switching voltage regulator as claimed in claim 14 when dependent on claim 4 in which the inductively coupled winding comprises a tertiary winding of the transformer formed by the inductive element.

16. A switching voltage regulator as claimed in claim 15 in which the ratio of turns between primary and tertiary transformer windings is at least two to one.

17. A switching voltage regulator as claimed in claim 16 in which the ratio is substantially equal to two to one.

18. A switching voltage regulator as claimed in claim 16 or claim 17 when dependent on claim 6 in which the turns ratio between secondary and tertiary windings differs from unity.

19. A switching voltage regulator as claimed in claim 18 when dependent on claim 10 or claim 13 in which voltage level clamping means includes an inductive element operable to effect a change in the charge polarity of the snubber capacitor between the different conduction states of the switching element and the ratio-of turns between secondary and tertiary windings is greater than unity to accommodate charging of the snubber capacitor from an initial voltage of one polarity to a required voltage of opposite polarity.

20. A switching regulator as claimed in any one of claims 14 to 19 when dependent on claim 13 in which the turns ratios of the secondary and tertiary windings are chosen to sum equal to, and replace, the clamp winding.

21. A switching regulator as claimed in claim 20 in which the turns ratio of the secondary and tertiary windings are chosen such that the inductive means satisfies the snubber capacitor charging voltage characteristics and the source means provides zero voltage with respect to the reference point.

22. A switching voltage regulator substantially as herein described with reference to any one of Figures 3 to 8 or 10 to 13 of the accompanying drawings.

23. A direct-coupled switching voltage step down regulator substantially as herein described with reference to any one of Figures 3, 4, 7 or 8 of the accompanying drawings.

24. A direct-coupled switching voltage step down regulator substantially as herein described with reference to Figure 13 of the accompanying drawings.

25. A direct-coupled switching voltage flyback regulator substantially as herein described with reference to Figure 5 of the accompanying drawings.

26. A direct-coupled switching voltage boost regulator substantially as herein described with reference to Figure 6 of the accompanying drawings.

27. A transformer-coupled switching voltage forward converter regulator substantially as herein described with reference to Figure 10 or 11 of the accompanying drawings.

28. A transformer-coupled switching voltage flyback converter regulator substantially as herein described with reference to Figure 12 of the accompanying drawings.