GB2176951A - High frequency chopper - Google Patents

Abstract

The present invention relates to a chopper including a switch (S1) in series with an inductor (L1) with associated free wheel diode (D1) further including one or more switching sub-assemblies each comprising a further inductor (L2, L3, LN) and a further switching device (S2, S3, SN) and together connected in parallel with the chopper and a further diode (D1, D2, DN) connected in parallel with the further inductor and the original load (V2). Each switching device and each switching sub-assembly are coupled to operate with a phase difference thereby producing a smoothing of the current flow through the load. <IMAGE>

Description

SPECIFICATION Improvements in and relating to high frequency switching This invention relatesto high frequency switching and has particular referenceto an inductively regulated power supply capable of being switched without any ofthe inherent reverse recovery problems associated with free-wheeling diodes.

One method is particularly suited to high voltage flyback regulators where the necessity of a high voltage, hence relatively slow free wheeling diodes can give rise to large reverse recovery current spikes. Protection against the spikes gives rise to added circuit complexity, reduced efficency and hence limiting the economic switching frequency attainable.

High frequency switching is desirable because ofthe speed of control avaiiable. It also cuts down the sizeof the inductors, which have a bearing on cost and efficiency figures due to less copper losses etc. It also allows a much morecompactsupplyto be attainable.

One circuit/method was originally conceived as a way of achieving very high efficient figures across a controllable constant voltage drop. Examples include a zenor diode or a plasma-discharge tube, (such as used in an inert gas laser), a variable current load can be catered for by maintaining the constantvoltagevia a feedback loop arrangement, as is the normal practice with D.C. to D.C. converters.

Atypical basic circuit configuration is a switching assembly comprising a load, an inductor in series with said load and a controllable switching device in series with said load and said inductor, and a "freewheeling" diode in parallel with the load and said inductorsothatifthe switching device is closed, the currentthroughthe inductor/load will rise proportional to the supply voltage minus load voltage across it with respect to time, thus storing energy in the inductor and if the switching device is open, the stored energy in the inductor is released and the current will decay through the inductor/load/free wheeling diode at a rate proportional to the load voltage plusfreewheeling diode forward voltage drop across the inductor.Such a system is generallywell known and will be referred to in this specification as a switching assembly of the type described.

According to the present invention, there is provided a switching assembly of the kind described further including one or more switching assemblies each comprising a furtherinductorand afurtherswitching device connected in series and together connected in parallel with said inductor and said switching device and a further diode connected in parallel with said further inductorandthe original load wherein the said switching device and the further switching device are coupled to operate with a phase difference thereby producing a smoothing of the currentflowthrough the load.

A plurality of switching assemblies may be appended to the original switching assembly in which casethe switching devices can be switched in sequence at spaced intervals to provide a smoothing of the currentflow through the load during switching. This can also be used to increase current handling capability with no loss in circuit efficiency.

Following is a description by way of example onlywith reference to the accompanying drawings of methods of carrying the invention into effect.

Figure lisa circuit diagram ofthe prior art switching assembly of the kind described.

Figure2 is a graph showing the load current against time for operation of the circuit forming the subject of Figure 1.

Figure 3 is a graph showing load cu rrent against time for the circuit shown in Figure 1 for a different set of perameter provisions.

Figure 4 is a simplified circuit in accordance with the present invention.

Figure 5is a series of diagrams of current in the conductor Ll,current in conductor L2 both against time, and the resultant current in the load againsttime showing the smoothing effect imparted by the present invention.

Figure 6is a circuit diagram of a more complex switching arrangement in accordance with the present invention.

Figure 7is an example of a switching system in accordance with the invention applied to a laser.

Turning now to the circuit diagram in Figure 1,V1 represents the supply voltage across the conductors, L1 is an inductor, D1 is a freewheeling diode, S1 is a switch which can any controllable switchable device such asa transistor or a MOS FET and V2 represents the voltage drop across the constant voltage drop device approximating to a constant voltage drop with varying load current.

Itwill be appreciated bythe man skilled in the artthatthis is a common flyback circuit arrangementoften used for current and/or voltage control across a known load variation.

In the circuit diagram shown in Figure 1, if the switch S1 is closed the current in the inductorwill rise proportional to the voltage across it with respect to time, this stores energy in the inductor.

Inductor/load current increases is proportional to supply voltage minus load voltage. wrttime (S1 closed).

If the switch S1 is opened the stored energy in the inductor will be released and the current will decay proportional to the voltage across it, through freewheeling diode D1.

current decrease in load in proportion to load voltage + free wheeling diode forward voltage drop.

wrttime (S1 open). (For a high current silicon diodeforward voltage drop is approximately IV).

Bywayofan example: If the switch S1 is used with a 50% duty cycle, i.e. equal time closed and open the following parameters apply: Case 7 -If V2 is less than orapproximatelyequalto V112 On closing the switch the load currentwill increase proportional to (V1 - V2)dt and on opening the current will decrease proportional to (V2 + D1V)dt. (Where D1V is thefreewheeling diode D1 forward voltagedrop).

Repetitive switching of the load current against time as shown in Figure 2 of the accompanying drawing.

Thecurrentandvoltage can be controlled in this manner, as is the normal case, by varying the mark, space ratio ofthe switch on offtimes, via a feed back loop arrangement.

It is neververy satisfactory at high frequency/current/voltage switching due a parasitic effect in diodes known as reverse recovery. Normally a diode will let current pass through it in only one direction, however, if a diode is conducting in this direction and a sudden voltage applied in the opposite direction the diodewill allow a surge of current in its blocking direction. This can have a catastrophic effect on semiconductor switching devices if not catered for, with snubber networks etc.

This method, therefore, becomes impracticle at high current/voltages, where large relatively slow diodes are used having poor reverse recovery characteristics and as a consequence make high frequency switching uneconomic due to high snubber network losses etc.

Case 2 - if V2 is greater than and approximately equal to V112 As with case 1 on closing the switch the current in the load will increase proportional to (V1 -V2)dtandon opening the current will decrease proportional to (V2 + D1V)dt.

For repetitive switching thiswill give riseto the load current/time configuration as shown in Figure 3 inthe accompanying drawings.

Letting the current fall to zero or approximately zero (there will be ringing duets the inductor and diode capacitance etc) the diode reverse recovery effect is eliminated so that very high frequency high efficiency switching is possible.

It should be noted that considerable increases in efficiency can also be obtained by switching when the current has reduced considerably as the diode reverse recovery current surge is roughly proportional to the currentflowing in the non-blocking direction.

Turning nowtothe embodimentofthe invention shown in Figure 4 of the accompanying drawings, itwill be appreciated that inductor L2 and switch S2 are now disposed in series with one another and in parallel with the original inductor L1 and the original switch Li. A second free wheeling diode is connected between the original bus and the junction between the switch S2 and the inductor L2 so that the free wheeling diode D2 is in parallel with the load V2 and inductor L2 when taken together.

It voltage V1 is selected so that V1 is smallerthan and approximately equal to twice the voltage V2 and a 50% duty on off cycle is selected for each switch with the switches being operated 1 80" out of phase. The current wave forms obtained through each inductor L1 and L2 are as shown in Figure Thus, the resultantcurrent obtained in the load is a smoothed current having a current ripple as shown in the bottom graph of Figure 5.

It can be shown thatthe current in the load at any instantwill be the sum ofall the currents flowing in the triangularcurrentwaveforms at the above instant (Kirchoff's law). This can give rise to a very desirable low ripple currentwave form with the correct choice of V1 with respect to V2.

To lowerthe ripple still further and atthe same time increase the current capability with substantially no detrement in efficiency, the above circuit can be extended to cover many switching assemblies having staggered switch on-off times, see, for example, the circuit diagram shown in Figure 6.

The total current in the load will be the sum of all the currents flowing in thetriangularcurrentwaveforms within the inductors L1 to LN at any instant.

Returning to the examples shown in Figure 1,where2 is greaterthan one half of V1, switching can still be catered for by the method in accordance with the invention by using a maximum reduced mark to space ratio (switch close/open time) than the 50% used in the example, where the following equation is satisfied:- (max %time switch closed) x (V1 -V2) < = (V2 + D1V) x (max %time switch open) The current in the load can be controlled by various methods, pulse width modulation,frequency modulation or a combination of the two etc.

The simpletwo switching element supply to be described is primarily intended for a small water cooled Hrgon Lasertubecapable of continuous operation at 25A 100 in Figure 7 and with a typical tube voltage drop of 200V. For such a tube the typical threshold lasing currentwould be of the order of 5 Amps, so the supply must be capable of regulated operation between say, 4 and 25 Amps.

The supply must also be capable of monitoring the cooling water flow and at the sametime be capable of reduced regulated current output with reduced cooling water flow. Overtemperature switches etc., must also be capable of system shut down.

With a typical tube voltage drop of 200V it can be seen thatfull wave rectified British single phase mains (or halfwave rectified 30 mains) satisfies the previously defined criteria for the proposed circuit/method, i.e.

tube voltage > supply vo Itag e 2 even with 15% high mains (worst design case).

With onlyatwo switching element supply the currentripplethrough the tube is likely to be higherthan with a multi-switching element supply with staggard on-offtimes in the previously described manner.

Current control using pulse width modulation techniques gives inherently a greater percentage of current ripple as the pulse width is reduced to the lower percentage on-off times, hence is more suited to muiti-switching element supples of the type previousiy described.

Frequency modulation techniques give proportionally the same percentage of current ripplethroughoutthe current range. The main disadvantage ofthis method is that to keep the inductors small, high frequencies at maximum current must be utilised, say 100 khg. Sothatto obtain the minimum current required, 4 Ampsin this example, a maximum switching frequency of: 25 4 x 100 Khz =625Khz This is very high frequency switching for a regulated output and requires very low switching propogation delays (the delay between required switching and actual switching) say less than 60 nsec (60 > c 1 10-9 sec)total.

Tube currentvariations above approximately 500 Khztend to be "smoothed", both by the gas discharge and the associated wiring capacitances/inductances. However, adding a small inductance in series with the tube, followed by a capacitance down to ground (see Figure 7) will tend to smooth out any associated ripple, howeverwill have a detrimental effect on the transient response times.

The seematic circuit shown in Figure 7 is a circuit having extremely low propagation delay times typically < 50 nsec total, so that switching in excess of 1 Mhz should be feasible without any of the normal associated pitfalls with free wheeling diode reverse recovery. As can be seen the proposed switching circuit is basically the one described by Figure 4, with the switches Si and S2 being replaced by proprietory MOS FETS. The addition of source follower resistors enables very efficient current measurements to be made and is used in thefeedbackarrangement.

The following is a brief mathematical exercise to determine the expected efficiency underworking conditions, though is not intended as a detailed design exercise.

The following design criteria are assumed:- 1. More average laser tube current 25 Amps at 100 Khz switching frequency.

2. Average lasertube voltage at 25 Amps = 200V (assumed constant over current range).

3. The inductors are ideal, i.e. contain a negligable resistive capacitive component (almosttrue due to the very small inductances required).

4. MOS FET "on resistance" = 0.55 (IRF 740 Siliconix).

5. MOS FETswitching time < 50nsec.

6. Supply voltage = 340 V (worst case assuming average mains with no regulation from smoothing networks).

From assumption 3) the current waveform through each MOS FET will be triangular and with a duty cycle of 50%.

rms current rating through each MOS FET 25 ~ 22 3 (duty cycle 50%) 7.2 Amps 12R losses through each MOS FET(assumption IV) (7.2)2.0.55 28.6 Watts each MOS FET Switching losses using assumptions i),v) and vi) average power dissipated per cycle 340 25 2 50 nsec: 340 . 2 . 50 x 10-9 H . 2 at 100 assuming MOS FET switch on is at zero current switching losses are approximately zero.

~340x 2 x50x10-9x 100 x 10 2 2 = ia 7 Watts each MOS FET Free wheeling diode losses assuming epitosual fast recovery, voltage drop < 1.2 Volts.

diode conduction time ~ (340 - 200) x 100% 200 (duetovoltage differences) total average losses -7.2 . 1401.2 200 -6.2 Watts each diode Current sense precision resistor losses 0.05, gives a signal of 1.25V at 24A.

12R losses - (7.2)2 x 0.05 2.6 Watts each resistor Mains rectifier losses. Assume voltage drop across rectifier = 2V at an rms current of 25A - will give a ball park figure of: ~ 50 Watts.

Other losses including smoothing/snubber and regulation/drive circuit losses, approximately:- ~ 50 Watts.

Total losses excluding focusing coil.

-2(28.6 + 10.7 + 6.1 + 2.6) + 50 + 50Watts.

196 Watts.

Regulated supplyefficiencyat25Amps -100 - 196 x 100% 25 x 200 ~ 96%.

Focusing coil and cathode heater losses have not been taken into account in this brieftheoretical analysis, these could easily be in excess of 500 Watts. Some lasertube manufacturers use the focusing coil resistance as a tube biasing resistance (i.e. in series with the tube) this is not required with the proposed circuit/method so that high efficiency low loss coils could be used.

Otherthan efficiency savings this type of power supply is also very cheap to produce. Component costs are low and due to the very low component count attainable, assembly and setting up costs are also reduced as againstthe existing systems in use.

The size ofthe supplywill be extremely small, so making an additional supply remote from the lazerhead unnecessary i.e. putting the laser head and supply into the same casing, giving a very desirable compact system.

Other advantages can include an improved meantime beforfailure (M.T.B.F.) with correct design due to the lower component count. This system can also accept a variety of existing laser heads with only slight modifications, such as gas tube pressure to obtain the correct running voltage.

The speed of response of this system might also have many uses, removing the need for lossy modulating systems used in laser light displays or data transmittion systems etc.

One speed of response can also be used with a better feed back system than with regulating the tube current, such as using the actual laser output as a tube current control, a highly stble system can be devised independent oftemperature and gas effects etc.

Claims (8)

1. A switching device ofthe kind described further including one or more switching sub-assemblies each comprising afurther inductor and a furtherswitching device and together connected in parallel with said inductorofthe said switching device ofthe kind described and a further diode connected in parallel with said further inductor and the original load whereby the said switching device and each switching sub-assembly are coupled to operate with a phase differencethereby producing a smoothing of the currentflowthrough the load.

2. A device as claimed in claim 1 wherein a plurality of further switching sub-assemblies are appended to the original assembly of the kind described whereby each of the further switching sub-assemblies are switched in sequence at spaced intervalsto provide a smoothing orthe currentflowthrough the said load during switching.

3. A device as claimed in any preceding claim wherein the current in the said load is controlled by pulse width modulation.

4. A device as claimed in any preceding claim wherein the current in the load is controlled byfrequency modulation.

5. Adevice as claimed in any preceding claim applied to awatercooled argon laser having athreshold lasing current ofthe order of 5 amps.

6. A device as claimed in claim 5 wherein the currentsupply is capable of monitoring cooling waterflowto the laser and includes means for reducing regulated current output with reduced cooling waterflow.

7. A device as claimed in claim 1 and substantially as herein described with reference to and as illustrated in the accompanying drawings.

8. Each and every novel embodiment herein set forth taken either separately or in combination.