GB2162344A - Wide bandwidth bandgap voltage reference circuit - Google Patents

Abstract

A bandgap voltage reference circuit includes an amplifier circuit (40,41,23,25,50,51) that detects imbalances in a bandgap cell (2) and transmits the imbalance signal through only high bandwidth NPN transistors to achieve very high, stable gain and very high bandwidth. The output voltage (VREF) produced thereby is sampled to provide feedback to an input (100) of the bandgap cell. Avoidance of active PNP transistors in the signal path results in a simple, yet very high bandwidth, bandgap voltage reference circuit and stable, highly predictable, highly reproducible circuit performance. <IMAGE>

Description

SPECIFICATION Wide bandgap voltage reference circuit The invention relates to bandgap voltage reference circuits, and more particularly to such circuits having much higher bandwidth than can be achieved by prior bandgap circuits.

Bandgap circuits are commonly used in integrated circuits to provide temperature-invarient reference voltages or biased voltages.

The various known bandgap voltage reference circuits have a veriety of shortcomings. Most of these circuits are quite complex, and occupy a large amount of semiconductor die ara. Some of the prior bandgap voltage reference circuits do not have adequate voltage gain and therefore are unduly sensitive to variations in load currents, which must be supplied by the bandgap circuit to a load circuit. Some of the prior bandgap voltage reference circuits do not have adequate voltage gain and therefore are unduly sensitive to variations in load currents, which must be supplied by the bandgap circuit to a load circuit. Some of the prior bandgap voltage reference circuits are capable of generating only a particular reference voltage, and cannot be adjusted to produce higher scale-up temperature-independent reference voltages. The state of the art is generally indicated in U.S.

Re-issue Patent Re-30,586 (Browkaw); 4,325,017 (Schade, Jr.); 4,249,122 (Widlar); 4,339,707 (Gorecki); and 4,064,448 (Eatock).

All of the closest prior bandgap circuits used lateral PNP transistors as active devices in the signal paths of gain stages that amplify differences between the differential inputs of a bandgap cell. Use of lateral PNP transistors severely limits the bandwidth of the gain stages, unless complex circuits techniques are used, since the bandwidth of lateral PNP transistors in present integrated circuit technology is limited to approximately one megahertz. This is an important limitation of prior bandgap circuits, becuase modern applications of bandgap circuits involved their use in integrated circuits that control system which are subject to high frequency transient currents.If the bandgap circuit has low bandwidth, transient voltages at a reference node voltage generated by the bandgap circuit and controlling other critical circuitry becomes filtered and tends to "ripple" or settle slowly, causing erroneous circuit performance in the controlled system. Bandgap reference circuits therefore need to have high bandwidths, for precisely the same reason that voltage regulator circuits generally need to have high bandwidth, i.e., so that they can have very fast recovery from transient load currents.

The lateral PNP gain stages used in conjunction with some prior bandgap voltage circuits are convenient to use with current bipolar integrated circuit technology, and some of the circuits must be carefully designed in order to achieve adequately high gains.

Thus, there remians a need for an improved, simple, high bandwidth bandgap voltage reference circuit providing a scaled-up temperature invariant reference voltage which is highly stable and has a very predictable response to transient load variations.

An object of the invention is to provide an improved bandgap voltage reference circuit having a much higher bandwidth then previous bandgap voltage reference circuits.

It is another object of the invention to provide a bandgap voltage reference circuit which is economical, has high bandwidth, and is highly predictable recovery to transient load variations.

According to the present invention there is provided a high bandwidth bandgap reference voltage circuit, comprising in combination: (a) a bandgap cell having first and second differential inputs, the first differential input being connected to a first reference voltage; (b) ratio circuit means for applying a thermal voltage to said second differential input of said bandgap cell; (c) load means in said bandgap cell for producing a first amplified signal representative of a current imbalance in said bandgap cell; (d) amplifying means responsive to said first amplified signal for producing an output reference voltage, said amplifying means including a plurality of active transistors through which said first amplified signal passes;; (e) temperature sensitive feedback means responsive to said output reference voltage for producing a feedback signal and applying said feedback signal to said ratio circuit means, said ratio circuit means operating in response to said feedback signal to compensate said thermal voltage so that said output reference voltage is substantially invarient with respect to the temperature sensitivity of said feedback means, all of said active transistors being NPN transistors, said amplifying means containing no transistors through which said first amplified signal passes which are not NPN transistors.

According to a further aspect of the present invention there is also provided a method of producing a bandgap reference voltage, comprising the steps of: (a) providing a bandgap cell having first and second differential inputs, and connecting the first differential input to a first reference voltage; (b) applying a thermal voltage to said second differential input of said bandgap cell by means of a resistive ratio circuit; (c) producing a first amplified signal representative of a current imbalance in said band gap cell at a first output node of said bandgap cell by means of a first load circuit coupled to said first output node; (d) amplifying said first amplified signal to produce an output reference voltage, by passing said amplified first signal through a plurality of active transistors;; (e) producing a feedback signal in response to said output reference voltage and applying said feedback signal to said resistive ratio circuit and by means of an active feedback transistor to compensate said thermal voltage so that said output reference voltage is substantially invarient with respect to temperature sensitivity of said feedback transistor, all of said active transistors through which said first amplified signal or said feedback signal passes, and the active transistors of said bandgap cell being vertical NPN transistors.

In accordance with one embodiment of the invention a high bandwidth bandgap reference circuit is provided including a bandgap cell having first and second differential inputs, the first one of which is connected to a reference voltage, a resistive ratio circuit for applying a "thermal voltage" to the second differential input of the bandgap cell, a load circuit in the bandgap cell for producing an amplified signal representative of a current imbalance in the bandgap cell and circuitry for further amplifying the signal to produce an output reference voltage, wherein the amplified signal passes only through vertical NPN transistors in the load circuitry and amplifying circuitry.The high bandwidth bandgap reference voltage circuit also including a feedback circuit responsive to the output reference voltage to produce a feedback signal and applying it to the resistive ratio circuit to establish and correct the thermal voltage. The described embodiment of the invention utilized a PNP current mirror circuit as a signal ended constant current load device for one side of the bandgap cell, an NPN emitter follower circuit to buffer the signal ended output of the bandgap cell and applies it to the input of an amplifier stage having an active NPN transistor and a PNP current mirror constant current load transistor. The amplified signal is buffered by an NPN emitter follower and applied to an NPN feedback transistor and a ratio circuit which resistively divides down the output reference voltage applied to the resistive ratio circuit means.The resistive ratio circuit is adjusted to produce precise compensation for the negative temperature coefficient of the VBE of the NPN feedback transistor.

Embodiments of the present invention will now be described with reference to the accompanying drawings, in which: Figure 1 is a simplified schematic drawing of a bandgap voltage reference circuit; Figure 2 is a detailed circuit schematic of an embodiment of the invention; Figure 3 is a graph useful in describing the operation of the circuits of Figs. 1 and 2.

Referring now to Fig. 1, bandgap reference voltage circuit 1 includes a bandgap cell outlined by dotted lines 2. Bandgap cell 2 includes an NPN transistor 33 having its base connected to conductor 101 and its emmiter connected to a constant current source circuit 3. The emitter of transistor 33 is also connected to the emitter of transistor 34, the base of which is connected to ground conductor 5. The four arrows symbolizing the emitter of transistor 34 indicate that its emitter area is four times that of transistor 33. The collector of transistor 33 is connected by means of conductor 6 and resistor 1 7 to a + V voltage supply conductor. Similarly, the collector of transistor 34 is connected by conductor 7 and resistor 1 8 to the + V conductor.

The base of transistor 33 is coupled by conductor 101 and resistor 21 to feedback conductor 103, and is also coupled by means of resistor 1 6 to ground conductor 5.

Conductors 6 and 7, which comprise the output terminals of bandgap cell 2, are connected to the positive and negative inputs, respectively, of high gain amplifier 8. The output of amplifier 8 is connected to conductor 37, at which the output voltage VREF is connected.

Resistor 1 9 is connected between conductor 37 and conductor 102, on which a bandgap voltage VBG is produced. Resistor 20 is connected between conductor 102 and ground conductor 5. Conductor 102 is also connected to the base of NPN feedback transistor 36, the collector of which is connected to + V and the emitter of which is connected to feedback conductor 103.

It should be noted that the foregoing structure shown in Fig. 1 is provided, not to precisely disclose the present invention, but to facilitate description of the basic operation of the bandgap circuit, an improved embodiment of which constitutes the present invention and is shown in Fig. 2. The operation of the circuit of Fig. 2 is similar to that of the simplified circuit of Fig. 1.

In Fig. 1, the basic operation is as follows.

Amplifier 8 senses any minute difference between the voltages of conductors 6 and 7, and causes VREF to change to a value that is fed back and forces VTH to have a value that causes conductors 6 and 7 to be at precisely the same voltage. It is assumed that the values of resistor 1 7 and 1 8 are equal. Therefore, the very high gain of amplifier 8 causes equal currents to be forced to flow through transistors 33 and 34.

Resistors 1 9 and 20 perform the function of "scaling up" the voltage VBG on conductor 102 to establish the value of VREF. Stated differently, VREF is divided down in accordance with the ratio of resistor 1 9 to resistor 20 to establish the value of bandgap voltage of VBG that is applied to the base of feedback transistor 36.

At this point, it should be borne in mind that an object of bandgap reference voltage circuit 1 is to produce a level of VREF which is temperature invariant.

The base-to-emitter voltage of feedback transistor 36, however, is not temperature invariant, and in fact decreases with inreasing temperature as indicated by curve 39 in Fig.

3. In Fig. 3, the desired zero variation of bandgap voltage VBG on conductor 102 is indicated by reference numeral 38. Reference numeral 44 designates the cross-hatched area in Fig. 3 representing the difference between curve 39 and curve 38, and represents the magnitude of the voltage at node 1 03.

Next, it will be appreciated that the voltage on conductor 101, namely the voltage VTH is equal to (kT/g)1 n(N), where N is the ratio between the area of the emitter of transistor 34 and the area of the emitter of transistor 33. It can be seen that the value of VTH increases with increasing temperature.

Those skilled in the art will easily be able to see that the ratio of resistor 21 to resistor 1 6 can be selected so that the variation of the feedback voltage on conductor 103 has a slope that is equal to but opposite in polarity to the slope of line 39 in Fig. 3 and therefore precisely compensates for the negative temperature coefficient of the base to emitter voltage of feedback transistor 36.

At this point, the detailed embodiment of the present invention will now be described with reference to Fig. 2. Reference numeral Fig. 1A designates the improved bandgap voltage reference circuit of the present invention. Where applicable, the same reference numerals have been used to indicate identical or similar components in Fig. 2 as in Fig. 1.

The structure of the bandgap cell 2 in Fig.

2 is somewhat different from that of the structure shown in Fig. 1 for certain practical reasons. First, the resistors 1 7 and 1 8 shown in Fig. 1 have been eliminated, and instead a leateral PNP current source transistor 9 has its emitter coupled by a degneration resistor 1 0A to + V and its collector connected to conductor 7. Lateral PNP transistor 9, in conjunction with lateral PNP transistors 11 and 13, comprise a PNP current mirror circuit, as those skilled in the art will readily recognize. The base of transistor 9 is connected to the emitter of transistor 13, the collector of which is connected to - V. The base of transistor 9 is also connected to the base of transistor 11, the emitter of which is coupled by degeneration resistor 1 OB to + V.The collector of transistor 11 is connected to the base of transistor 1 3 by means of conductor 14, and is also connected to the collector of NPN transistor 3B, which is one of the current source transistors of an NPN current mirror circuit including NPN transistors 3A, 3B, 3C, 3D, 3E and 3F. A current source 29 drives diode-connected transistor 3F, which establishes the base-to-emitter voltage of the current source transistors of this current mirror circuit.

If desired, regeneration resistors can be used in series with the emitters of each of the NPN current source transistors to improve matching thereof, but are omitted from Fig. 2 for convenience.

Those skilled in the art will readily recognize that the collector of PNP current source transistor 9 presents a much higher impedence to the collector of NPN transistor 34A than is achievable by a practical resistor such as 1 8 shown in Fig. 1, at the present state of the art of integrated circuit technology.

The resistor 1 7 of Fig. 1 is not used in the bandgap cell in Fig. 2. An NPN transistor 10 having its base connected to a reference voltage having a level 3 diode voltage drops above ground conductor 5 has its emitter connected to the collector of transistor 33A.

The function of transistor 10 is to equalize the collector-base voltages of transistors 33A and 34A.

In order to increase the nominal voltage across resistor 16, the expedient of adding a diode-connected transistor 33B identical to transistor 33A in series with the emitter of NPN transistor 33A has been utilized. Similarly, diode-connected transistor 34B is connected in series with the emitter of transistor 34A, the collector of which is connected to conductor 7. The emitters of transistor 33B and 34B are connected together and are also connected to current source transistor 3A, the emitter of which is connected to - V. In the preferred embodiment of the invention, the emitter area of transistor 33B is equal to that of transistor 33A, and the emitter area of transistor 34B is equal to the emitter area of transistor 34A.Note that the larger the ratio of emitter areas of transistors 34A and 34B to the corresponding emitter areas of transistors 33A and 33B, the less the feedback attenuation due to resistors 21 and 1 6 needs to be the smaller that attenuation is, the greater the closed loop bandwidth will be.

While the output of the bandgap cell shown in Fig. 1 is applied as a differential input signal to amplifier 8, in Fig. 2 the amplified output produced by the bandgap cell 2 on conductor 7 is taken as a "single ended" output, rather than a differential output, and is applied to the base of an emitter follower transistor 40, the collector of which is connected to + V.

The emitter of emitter-follower or buffer transistor 40 is connected to conductor 27, which in turn is connected to the collector of current source transistor 3C. The base of transistor 40 is also coupled to one terminal of compensation capcitor 1 2. The emitter of buffer transistor 40 is also coupled by conduc tor 27 to the base of a second stage amplifier transistor 41, which is an NPN transistor, in accordance with the present invention. The emitter of NPN transistor 41 is connected to ground conductor 5, and its collector is coupled by means of conductor 22 to the other terminal of compensation capacitor 1 2 and also to the collector of PNP current source transistor 23. Lateral PNP transistors 23 and 25 form another PNP current mirror.The emitters of these two PNP current transistors are connected by means of degneration resistors 10C and 1 OD, respectively, to + V. Their bases are both connected to the collector of transistor 25 by means of conductor 26.

The output of the second gain stage in Fig.

2 produced on conductor 22 and is coupled by conductor 22 to the base of NPN emitter follower transistor 50, the collector of which is connected to + V, and the emitter of which is connected to conductor 28.

The collector of PNP current source transistor 25 is connected by conductor 26 to the collector of NPN current source transistor 3D.

Conductor 28 connects the emitter of NPN emitter follower transistor 50 to the collecotr of NPN current mirror transistor 3E. Conductor 28 is also connected to the base of NPN transistor 51, the emitter of which drives conductor 37 on which the scaled up temperature invariant bandgap reference voltage VREF is produced. The collector of transistor 51 is connected to + V. Conductor 37 is connected to the collector of NPN feedback transistor 36, the emitter of which is connected to feedback conductor 103, as in Fig. 1. Resistors 1 9 and 20 and conductor 102 are connected in the same fashion as in Fig. 1, and perform the same function of scaling up VBG to produce VREF.

The DC operation of the single-ended amplification accomplished by bandgap cell in Fig.

2 differs from that shown in Fig. 1 in that the PNP current mirror including transistors 9, 11 and 1 3 allows the current 1 2 flowing through transistors 34A and 34B to the 50 microamperes, as determined by the current 11, which is also 50 microamperes. the later being produced by NPN current mirror source transistor 3B and current souce 29.

NPN current mirror transistor 3A causes the current 1 3 flowing through its collecotor to be 100 microamperes. Thus, the current 14 flowing through transistors 33A and 34A is forced to be 50 microamperes. Any imbalance between 12 and 14 caused by a change in the voltage VTH on conductor 101 is amplified at conductor 7 with a gain of approximately 500, and is further amplified at conductor 22 with an additional gain of approximately 2,000, and feedback is provided in a manner analogous to that described above with reference to Fig. 1 to provide the needed correction at conductor 101.

A novel aspect of the above invention is the exclusive use of NPN transistors as active devices in the gain producing portions of the bandgap circuit of Fig. 2, including transistors 40, 41, 50, 51 and 36. These NPN transistors have a much higher cutoff frequency than any lateral PNP transistors that are presently practically implementable at the present state of the art.

Exemplary values for the resistors and the capacitor of the embodiment of Fig. 2 are given in the following table.

TABLE 1 Resistor 10A 2 kilohms Resistor 10B 2 kilohms Resistor 1 OC 2kilohms Resistor 1 OD 2 kilohms Resistor 1 6 700 ohms Resistor 21 Skilohms Capacitor 12 2.5 picofarads The values of resistors 1 9 and 20 are not at all critical, and are chosen simply to provide an adeuate bias current through emitter follower transistor 51 and to provide the desired ratio of scaling up the bandgap voltage VBG to the desired temperature invariant value of VREF- The precise values of currents produced by NPN current mirror transistors 3C, 3D and 3E are not at all critical.

Those skilled in the art will realize that at room temperature, the value of VTH is approximately 71 millivolts if N, the ratio of the emitter areas of transistors 34A and 34B the emitter areas of transitions 33A and 33B, respectively, is 4.

Typically, the value of VBG that is desired is equal to the bandgap voltage of silicon, which is about 1.1 4 volts. It is desirable that the base to emitter voltage Vie(35) of transistor 36 plus the node voltage at conductor 103 adds to the bandgap voltage VBG. Fig. 3 illustrates these two components of voltage and their sum over a theoretical range of temperature.

Mathematically, the value of VBE(36) can bye extrapolated to VBG at a temperature = 0" Kelvin. At this same temperature the extrapo lated value of Vim is zero. The shaded section 44 of the drawing represents the voltage of node 103 as being equal to R21 + R16 VTH R16 where R,6 and R21 are the resistances of resistors 1 6 and 21. The proper selection of the resistor ratio and N allows node 103 to increase in magnitude with respect to temperature at such a rate that the bandgap node 102 remians nearly constant in the range of interest from 200 to 400e Kelvin.

It should be noted that in a precise analysis the Vie(38) component displays slight curvature and that the bandgap voltage varies slightly with temperature. These effects can be included in the total design of the reference circuit of Fig. 2.

It has been found that the above described embodiment of the invention has a very high bandwidth, approximately 30 megahertz. This is far higher than the amplifier bandwidth of any prior known bandgap voltage reference amplifier circuit, all of which use single gain stage amplfiers comprised of PNP current mirror circuits or boot strap circuits, sometimes called floating current mirror circuits. The later circuits typically have an amplifier bandwidth of less than one megahertz.

Yet, the circuit shown in Fig. 2 does not require more semidconductor chip area then the single stage PNP floating current mirror amplifiers, and is far simpler from a circuit analysis point of view. The improved bandwidth of the circuit shown in Fig. 2 actually has two gain stages rather than one.

The first gain stage comprises the right hand half of gain cell 2 and the PNP current source transistor 9, and the second stage includes NPN transistor 41 and PNP current source transistor 23. The vastly improved performance results because the amplified signal does not pass through any of the PNP current source transistors; they simply function as high impedence loads devices. The cutoff frequency of the NPN vertical bipolar integrated circuit transistors is typically over 500 megahertz. For the lateral PNP transistors, the cutoff frequency is typically about only one magahertz. As a result, high frequency lateral PNP transistor circuits, and amplfiers including them, have bandwidths limited to roughly one magahertz bandwidth, unless complex circuit techniques are used.

By way of comparison of the simplicity of design and analysis of the circuit of Fig. 2 with the above-mentioned prior art "floating PNP current mirror" amplification stages of prior art bandgap voltage circuits, the gain of my circuit with its two stages of NPN type amplification is simply gm,9m2R"R,2 9m1 being the transconductance of the right hand side of bandgap cell 2, R1, being the impedence at node 7, 9m2 being the transconductance of NPN transistor 41, and R12 being the impendence at node 22. This gain is highly predictable and stable, and has a value of approximately 1,000,000.As previously mentioned, the gain of the floating PNP current mirror circuit theoretically may be nearly infinite, but as a practical matter, when these devices are actually constructed, subtleties in the circuit operation frequently causes the actual gain to be far less than expected, often much less than 1,000,000. Furthermore, the bandwidth of the PNP current mirrors is always far less than the 30 megahertz bandwidth of the present circuit.

This extremely high bandwidth allows compensation capacitor 1 2 to be very small, typically only about 2-5 picofarads, compared to the 30 to 40 picofarads range of compensation capacitors typically required in prior bandgap voltage reference circuits.

Consequently, the recovery period of the bandgap reference circuits of the present invention in response to output transient load variations is reduced by an order of magnitude or more over prior bandgap reference voltage circuits employing active lateral PNP transistors in their gain stages. Accordingly, the bandgap circuit of Fig. 2 is considered to be a much safer, lower risk design that does not require any compromise in size, complexity, or performance, and always has far higher bandwidth and shorter recovery times than any known bandgap voltage reference circuit.

While the invention has been described with reference to a particular embodiment thereof, those skilled in the art will be able to make various modifications to the described embodiment of the invention without departing from the true scope thereof. It is intended that elements and steps of similar devices and processes which accomplish substantially the same function in substantially the same way to achieve substantially the same result are within the scope of the invention. For example, various ways other than the way described can be utilized to accomplish equal current densities in NPN transistors on each of the right and left sides of the bandgap cell.

Claims (16)

1. A high bandwidth bandgap reference voltage circuit, comprising in combination: (a) a bandgap cell having first and second differential inputs, the first differential input being connected to a first reference voltage; (b) ratio circuit means for applying a thermal voltage to said second differential input of said bandgap cell; (c) load means in said bandgap cell for producing a first amplified signal representative of a current imbalance in said bandgap cell; (d) amplifying means responsive to said first amplified signal for producing an output reference voltage, said amplifying means including a plurality of active transistors through which said first amplified signal passes;; (e) temperature sensitive feedback means responsive to said output reference voltage for producing a feedback signal and applying said feedback signal to said ratio circuit means, said ratio circuit means operating in response to said feedback signal to compensate said thermal voltage so that said output reference voltage is substantially invarient with respect to the temperature sensitivity of said feedback means, all of said active transistors being NPN transistors, said amplifying means containing no transistors through which said first amplified signal passes which are not NPN transis tors.

2. A high bandwidth bandgap reference voltage circuit as claimed in claim 1 wherein said bandgap cell includes first and second NPN transistors, having their bases coupled to said first and second differential inputs, respectively, and wherein said load means includes a first PNP current source transistor having its collector coupled to the collector of said first NPN transistor, said first amplified signal being produced at said collector of said first PNP current source transistor.

3. A high bandwidth bandgap reference voltage circuit as claimed in Claim 2 wherein said amplifying means include a third NPN transistor having its emitter coupled to said first reference voltage, and a second PNP current source transistor having its collector coupled to the collector of said third NPN transistor, and first coupling means for coupling said collecotr of said first PNP current source transistor to the base of said third NPN transistor to couple said first amplified signal to the base of said third NPN transistor.

4. A high bandwidth bandgap reference voltage circuit as claimed in Claim 3 wherein said temperature sensitive feedback means includes an NPN feedback transistor, the emitter-base voltage of which decreases with respect to temperature and wherein said ratio circuit means includes first and second resistors coupled in series between the emitter of said NPN feedback transistor and said first reference voltage, the junction between said first and second resistors being coupled to said second differential input of said bandgap cell.

5. A high bandwidth bandgap reference voltage circuit as claimed in Claim 4 wherein the ratio of said first and second resistors is selected to compensate for a negative temperature coefficient of said NPN feedback transistor.

6. A high bandwidth bandgap circuit as claimed in any of Claims 3 to 5 wherein said first coupling means includes a first NPN emitter follower transistor having its base connected to the collector of said first PNP current source transistor and its emitter connected to the base of said third NPN transistor.

7. A high bandwidth bandgap circuit as claimed in any of Claims 4 or 5 wherein said first coupling means includes a first NPN emitter follower transistor having its base connected to the collector of said first PNP current source transistor and its emitter connected to the base of said third NPN transistor, and wherein said amplifying means includes a second NPN emitter follower transistor having its base connected to the collector of said third NPN transistor, and a third NPN emitter follower transistor having its base connected to the emitter of said second emitter follower transistor, and its emitter coupled to said output reference voltage and also coupled to the base of said NPN feedback transistor.

8. A high bandwidth bandgap circuit as claimed in Claim 7 including a third resistor coupled between the emitter of said third emitter follower transistor and the base of said NPN feedback transistor, and a fourth resistor coupled between the base of said NPN feed- back transistor and said first reference voltage.

9. A high bandwidth bandgap reference voltage circuit as claimed in any of claims 3 to 8 including a compensation capacitor coupled between the collector of said third NPN transistor and the collector of said first PNP current source transistor.

10. A high bandwidth bandgap circuit as claimed in any of Claims 3 to 9 wherein the geometry of a transistor in said bandgap cell and the geometry of said first PNP current source transistor is selected to produce a gain of approximately 500 for said bandgap cell.

11. A high bandwidth bandgap cell as claimed in Claim 10 wherein the geometry of said third NPN transistor and the geometry of said second PNP current source transistor are selected to produce a second storage gain of approximately 2000 for said amplifying means.

1 2. A high bandwidth bandgap circuit as claimed in any of Claims 2 to 11, wherein said bandgap cell includes a first diode connected transistor having its base and collector coupled to the emitter of said first NPN transistor and a second diode connected transistor having its base and collector coupled to said second NPN transistor and having its emitter coupled to the emitter of said first diode coupled transistor.

1 3. A high bandwidth bandgap cell as claimed in any preceding Claim wherein all of said active NPN transistors are vertical NPN transistors in an integrated circuit.

14. A high bandwidth bandgap cell substantially as herein described with reference to Fig. 2 of the accompanying drawings.

1 5. A method of producing a bandgap reference voltage, comprising the steps of: (a) providing a bandgap cell having first and second differential inputs, and connecting the first differential input to a first reference voltage; (b) applying a thermal voltage to said second differential input of said bandgap cell by means of a resistive ratio circuit; (c) producing a first amplified signal representative of a current imbalance in said bandgap cell at a first output node of said bandgap cell by means of a first load circuit coupled to said first output node; (d) amplifying said first ampliffed signal to produce an output reference voltage, by passing said amplified first signal through a plurality of active transistors;; (e) producing a feedback signal in response to said output reference voltage and applying said feedback signal to said resistive ratio circuit and by means of an active feedback transistor to compensate said thermal voltage so that said output reference voltage is substantially invarient with respect to temperature sensitvity of said feedback transistor, all of said active transistors through which said first amplified signal or said feedback signal passes, and the active transistors of said bandgap cell being vertical NPN transistors.

16. A method as claimed in Claim 15 wherein said bandgap cell includes first and second NPN transistors, having their bases coupled to said first and second differential inputs, respectively, and wherein said first load includes a first PNP current source transistor having its collector coupled to the collector of said first NPN transistor, said first amplified signal being produced at said collector of said first PNP current source.

1 7. A method of producing a bandgap reference voltage substantially as herein described with reference to the accompanying drawings.