GB1595773A

GB1595773A - Method and apparatus for inserting synchronizing words in a digitalized television signal data stream

Info

Publication number: GB1595773A
Application number: GB1819180A
Authority: GB
Original assignee: Ampex Corp
Current assignee: Ampex Corp
Priority date: 1976-10-29
Filing date: 1977-10-24
Publication date: 1981-08-19

Description

(54) METHOD AND APPARATUS FOR INSERTING SYNCHRONIZING WORDS IN A DIGITALIZED TELEVISION SIGNAL DATA STREAM (71) We. AMPEX CORPORATION. a Corporation organised and existing under the laws of the State of California. United States of America, of 401 Broadway, Redwood City, State of California, United States of America, do hereby declare the invention, for which we pray that a patent may be granted to us, and the method by which it is to be performed, to be particularly described in and by the following statement: The present invention generally relates to recording and reproducing apparatus and, more particularly, to apparatus that is adapted to record and reproduce television signals, using digital techniques.

The continued advances in technology have resulted in many changes in the equipment that is currently being used in television broadcast stations. One of the more recent changes that has evolved is the shift away from photographic techniques toward the use of magnetic media in many phases of the operation of the commercial broadcast television station. For example, feature films being broadcast often originate from magnetic tape rather than film and television station news departments are increasingly converting to videotape recording systems rather than using film cameras to provide the visual coverage of the news stories. Moreover, many systems utilize travelling transmitters that can either broadcast on location coverage or transmit such coverage to the station which can either be broadcast "live" or videotaped, edited and broadcast at a later time. Some of the many benefits of these techniques are the ease of handling, flexibility and speed of processing compared to the use of photographic film, coupled with the ability to reuse the magnetic tape when the information that is recorded on them is no longer needed.

One of the last remaining film domains in the present day commercial television broadcasting station is the Telecine island which uses 35 millimeter film transparencies. The Telecine island is used to provide video still images that are used during programming, commercials, news and the like, i.e., wherever a still image may be used during operation. Their use is extensive as is evidenced by the fact that the average commercial broadcast television station maintains a total file on the order of about 2000 to 5000 35 millimeter transparency slides. The maintenance of the total file represents a laborious operation which requires introduction of new slides, the discarding of obsolete slides and the maintenance of an accurate index so that they can be readily obtained when needed. When slide program sequences are to be assembled, they must be manuallv carried to the Telecine island. cleaned and manually loaded. Even with the cleaning operation, dust particles and scratches and the like may easily result in an unsatisfactory end product even when the projectionist is careful. Moreover, following their use during broadcasting, the slides must be removed and returned to the file. The entire assembling, use and refiling of the slides represents a substantial labor investment because of the many manual operations that are required. The Telecine operation is considered to be one of the most antiquated operations in many modern broadcast stations and is basically incompatible with a fully automated station operation.

In contrast to the Telecine island or the use of opaque graphic material as the source for generating video still images, the present invention involves a recording and playback apparatus that will record and reproduce still images, with the still image video information being stored on magnetic media. The apparatus of the present invention utilizes generally standard computer disc drives (though modified in some respects as will be described) as the magnetic storage media and thereby eliminates the many problems that are associated with slide transparencies.

Since the still images are recorded on magnetic media, the problems of physical degradation during use, e.g., dust particles and scratches, are not experienced. Moreover, since the recorded information can be easily accessed, the same still image may be used by operators at different locations almost simultaneously.

While the apparatus of the present invention is suited to recording and reproducing still images and a preferred embodiment is described herein as arranged for such operations, the apparatus can be arranged to record and reproduce a sequence of images depicting motion with both altered and unaltered time base effects.

As will be described herein, the recording and reproducing apparatus described herein processes a composite video information signal for recording on disc packs of computer disc drives and, during the processing, removes the horizontal synchronization pulses from the signal, thereafter samples the analog video information signal and converts the samples to a number of digital data streams. The present invention is directed to apparatus for inserting a unique digital synchronizing word, preferably on alternate television lines, generally in the location of the previously removed horizontal synchronization pulse. The synchronization word is used as a reference for correcting for time base and skewing errors that can occur among the plurality of bits of data in the data streams that must be combined to derive the value of each sample. Since the synchronization word is a multibit word that is inserted in the data stream, it should be apparent that the bits defining the synchronization word may occasionally randomly occur during the active portion of the television line and, for this reason, the horizontal blanking interval is made to be devoid of information content in the data streams, except for the presence of the unique synchronization word. By doing so, the circuitry in the reproducing portion of the signal can reliably and accurately detect the synchronization word which is used as the timing reference. The apparatus described also removes all information content of the data streams during the approximately first half of the vertical blanking interval so that only the synchronization word occurs on alternate lines through about the first 10 to 12 lines of the vertical blanking interval. This provides sufficient time for the detecting circuitry to lock onto the synchronization word, in the event that the circuitry becomes disoriented for some reason.

The apparatus of the present invention inserts the synchronization word in the proper location during alternative lines in the horizontal blanking interval without stopping the data streams that are generally continuously moving. However, no active video information is lost, since the synchronization word is inserted during the horizontal blanking interval.

This invention will be described in conjunction with a complete recording and reproducing apparatus of which various features are the subjects of our copending Applications Nos.

8018055; 8018056; 8018192; 8018193; (Serial Nos. 1595771; 1595772; 1595774; 1595775) 8018385; 8018386; 8018387; 8018563 and 80/8564, (Serial Nos. 1595776; 1595777; 1595778; 1595779 and 1595780). likewise divided out of 45195/76, now abandoned.

DESCRIPTION OF THE DRAWINGS FIG. 1 is a perspective view of the apparatus embodying the present invention, illustrating its overall appearance. including the internal access station and two disc drive units; FIG. 2 is an enlarged perspective view illustrating a representative remote access station that an operator can use to control the operation of the apparatus of the present invention; FIG. 3 is an enlarged top view of a portion of the internal access station keyboard shown in HG. 1 particularly illustrating the various keys and bars that an operator uses during operation; FIG. 4 is a broad functional and simplified block diagram of the entire apparatus of the present invention; FIG. 5A illustrates a portion of a typical television signal illustrating the vertical interval thereof; FIG. 5B illustrates a portion of a color television signal, particularly illustrating the horizontal synchronization pulse and color burst signal; FIG. 6 is a functional block diagram broadly illustrating the signal flow path through the apparatus during a record operation; FIG. 7 is a functional block diagram broadly illustrating the signal flow path through the apparatus during a playback operation; FIG. 8 is a block diagram illustrating the internal computer control system that controls the operation of the signal system, disc drives and their associated control system and the access stations used by the operators; FIGS. 9A and 9B together comprise a block diagram illustrating the signal system for the apparatus of the present invention, including control interconnections between the various blocks; FIG. 9C is a timing diagram illustrating sampling of a television signal and phase relationships that occur at different locations of the signal system: FIG. 10 is a functional block diagram of the video input circuitry (substantially similar to the reference input circuitry) which is a portion of the signal system shown in FIG. 9A; FIG. 11A is a functional block diagram of the reference logic circuitry which is a portion of the signal system shown in FIG. 9A; FIG. 11B is a timing diagram for the PALE Flag generator included in the reference logic circuitry shown in FIG. 11A.

FIG. 12A is a functional block diagram of the reference clock generator circuitry which is a portion of the signal system shown in FIG. 9A; FIG. 12B is a timing diagram illustrating the operation of portions of the reference clock generator shown in FIG. 12A; FIG. 12C is a timing diagram illustrating the operation of portions of the reference clock generator shown in FIG. 12A; FIGS. 13A, 13B, 13C and 13D together comprise an electrical schematic diagram illustrating the encoder switch which is a portion of the signal system shown in FIG. 9A; FIG. 13E is a block diagram of the blinking cross delete signal generator included in encoder switch circuitry shown in FIGS. 13A through 13D; FIG. 13F is a diagrammatic representation of the blinking cross delete signal produced with two television fields upon playback; FIG. 14 is a functional block diagram of the encoder and syne word insertion circuitry which is a portion of the signal system shown in FIG. 9A; FIG. 15A is a functional block diagram of the data rate and time base corrector circuitry which is a portion of the signal system shown in FIG. 9A; FIGS. 15B and 15C are timing diagrams for the data rate and time base corrector circuitry shown in FIG. 15A; FIG. 16 is a functional block diagram of the data transfer circuitry which is a portion of the signal system shown in FIG. 9A; FIG. 17 is a block diagram of one embodiment of the chroma separator and processing circuitry of the signal system shown in FIG. 9A wherein the chrominance inverter portion is a digital transversal filter with odd symmetry; FIG. 18 is a more detailed block diagram of the chroma inverter portion of the circuitry shown in the block diagram of FIG. 17; FIGS. 19 and 20 are block diagrams of alternative embodiments of the chroma separator and processing circuitry of the signal system shown in FIG. 9A; FIG. 21 is a block diagram of an alternative embodiment of the circuitry employed to reconstitute four fields of color television signals from a single stored field; FIG. 22 is a functional block diagram of the blanking insertion and bit muting circuitry which is a portion of the signal system shown in FIG. 9A;-analog converting 9A; FIG. 23 is a functional block diagram of the digital-to-analog converting and burst and sync insertion circuitry which is a portion of the signal system shown in FIG. 9A; FIG. 24 is a block diagram of a playback circuit which includes the equalization circuit of the signal system; FIG. 25 is a block diagram of one embodiment of the equalization circuit shown in FIG. 24; FIG. 26 is a block diagram of another embodiment of the equalization circuit shown in FIG.

24; FIG. 27 is a graph showing the playback response of a conventional reproduce head and preamplifier combination circuit; FIG. 28 is a graph showing an equalization curve provided by the equalization circuit shown in FIG. 24 which curve compensates for the curve shown in FIG. 27; FIG. 29 is a functional block diagram of the central processing unit interface portion of the computer control system of the apparatus; FIG. 30 is a functional block diagram of the remote access station interface portion of the computer control system of the apparatus; FIG. 31 is a functional block diagram of the remote access station and internal access station portion of the computer control system of the apparatus; FIGS. 32A and 32B together comprise an electrical schematic diagram of the signal system interface portion of the computer control system of the apparatus; FIGS. 33A and 33B are functional block diagrams of the first data track interface portion of the computer control system of the apparatus; FIGS. 34A, 34B, 34C, 34D, 34E, 34F, 34G and 34H together comprise electrical schematic diagrams of the second data track interface portion of the computer control system of the apparatus; FIGS. 35A and 35B together comprise an electrical schematic circuit diagram of the disc drive interface portion of the computer control system of the apparatus; FIG. 36 is a functional block diagram of the disc drive servo phase lock circuitry of the disc drive portion of the apparatus: FIGS. 37A and 37B are electrical schematic circuit diagrams of the record play control circuitry for the disc drive portion of the apparatus; FIGS. 38A and 38B are electrical schematic circuit diagrams of the record timing circuitry for the disc drive portion of the apparatus; FIG. 39 is an electrical schematic circuit diagram of the timing generator circuitry for the disc drive portion of the apparatus; FIGS. 40A and 40B are electrical schematic circuit diagrams of the error check circuitry for the disc drive portion of the apparatus; FIGS. 41A and 41B together comprise an electrical schematic diagram of the disc phase lock control circuitry for the disc drive portion of the apparatus shown in the block diagram of FIG.

36; FIGS. 42A, 42B, 42C and 42D together comprise an electrical schematic diagram of the input circuitry of the signal system shown in the block diagram of FIG. 10; FIGS. 43A, 43B, 43C and 43D together comprise an electrical schematic diagram of the reference logic circuitry of the signal system shown in the block diagram of FIG. 11; FIGS. 44A, 44B, 44C and 44D together comprise electrical schematic diagrams of the reference clock generator of the signal system shown in the block diagram of FIG. 12A; FIGS. 45A, 45B, 45C and 45D together comprise an electrical schematic diagram of the encoder and sync word inserter circuitry of the signal system shown in the block diagram of FIG. 14; FIG. 45E is a timing diagram illustrating the operation of the data encoder circuitry shown in FIGS. 45A, 45B, 45C and 45D; FIGS. 46A, 46B, 46C and 46D together comprise an electrical schematic diagram of the data decoder and the data rate and time base corrector circuitry of the signal system shown in the block diagram of FIG. 15; FIG. 46E is a timing diagram illustrating the operation of the data decoder circuitry shown in FIGS. 46A and 46B; FIGS. 47A and 47B together comprise an electrical schematic diagram of the data transfer circuitry of the signal system shown in the block diagram of FIG. 16.

FIGS. 48A, 48B and 48C together comprise electrical schematic diagrams of the chroma separator of the chroma portion of the signal system shown in FIG. 17; FIGS. 49A and 49B together comprise electrical schematic diagrams of the chroma inverter circuitry for use in the chroma portion embodiment illustrated by the block diagram of FIG. 18 and timing control therefor; FIG. 49C is a functional block diagram of the timing control portion of the chroma inverter circuitry of the signal system schematically illustrated in FIGS. 48A, 48B, and 48C; FIG. 49D is a timing diagram illustrating the operation of the timing control portion of the chroma inverter shown in FIG. 49C; FIGS. 49E and 49F together comprise electrical schematic diagrams of the chroma inverter circuitry for use in the chroma portion embodiment illustrated by the block diagram of FIG. 20 and timing control therefor; FIGS. 50A and 50B together comprise electrical schematic diagrams of the chroma band pass filter circuitry of the chroma portion of the signal system shown in the block diagram of FIG. 17; FIGS. 51A and 51B together comprise an electrical schematic diagram of the blanking and bit muting circuitry of the signal system shown in the block diagram of FIG. 22; FIGS. 52A, 52B, 52C and 52D together comprise an electrical schematic diagram of the digital-to-analog converter and burst and sync insertion circuitry of the signal system shown in the block diagram of FIG. 23; FIGS. 53A and 53B together comprise an electrical schematic diagram of the equalizer circuits of the signal system shown in the block diagram of FIG. 24; and FIGS. 54A and 54B together comprise an electrical schematic diagram of the preamplifier circuits employed in the playback circuit shown in the block diagram of FIG. 24.

For a complete understanding of the system in which the present invention is preferably employed, reference may be made to the specification of our copending application No. 80/8055 (Serial No. 1595771) in the Figures of which: FIGS. SSA, 55B, 55C and 55D together comprise an electrical schematic diagram of the remote access station interface circuitry of the computer control system shown in the block diagram of FIG. 30: FIGS. 56A, 56B, 56C and 56D together comprise an electrical schematic diagram of the remote access station and internal access station keyboard circuitry of the computer control system shown in the block diagram of FIG. 31; FIGS. 57A and 57B together comprise an electrical schematic diagram of the first data track interface circuitry of the computer control system shown in the block diagram of FIG.

33; FIGS. 58A, 58B, 58C and 58D together comprise an electrical schematic diagram of the central processing unit interface portion of the computer control system of the apparatus; FIGS. 59A and 59B together comprise an electrical schematic diagram of the disc predriver portion of the disc drive portion of the apparatus shown in the block diagram of FIG. 36; FIGS. 60A and 60B together comprise an electrical schematic diagram of the disc drive data interface portion of the apparatus; FIG. 61 is a top view of a portion of the access assignment panel keyboard particularly illustrating the various keys and bars that an operator uses during operation; FIGS. 62A, 62B and 62C together comprise an electrical schematic diagram illustrating the access assignment station keyboard circuitry of the access assignment panel shown in FIG. 61; and FIGS. 63A through 63VV inclusive consist of sheets which together comprise a flow chart of .one control program for the computer control system illustrating the program steps followed in causing requested functional operations to be performed by the apparatus.

Broadly stated and referring to FIGS. 1-3, the present invention is directed to a recording and reproducing apparatus, indicated generally at 70 in FIG. 1 which includes two bays 71 and 72 containing electrical circuitry associated with the apparatus, together with the various monitoring and control hardware shown specifically in the upper portion of the bay 72. The system also includes a pair of disc drives 73 located adjacent the rightward bay 72 with each of the disc drives 73 having a disc pack 75 mounted thereon. While two disc drive units are specifically illustrated in FIG. 1, it should be understood that there may be additional disc drives used with the system to increase the on-line storage capacity of the apparatus. It should also be appreciated that a single disc drive may be used, but many desirable functional operations cannot be performed with a single disc drive as will be apparent from the ensuing description. Operational control of the apparatus is performed by one or more operators using either one of many remote access stations, such as the remote access station 76 shown in FIG. 2, or an internal access station 78 which is located in the bay 72. If desired, a video monitor 79, vector and "A" oscilloscopes 80 may be provided as shown in bay 72. Phase control switches 81 are provided above the internal access station 78.

The apparatus is controlled by an operator using either the internal access station 78 or a remote access station 76. both types of which have a keyboard with numerical and function keys and bars, a 32 character display 82, which provides a readout of information that is needed to carry out functional operations during use, as well as to display the information concerning the identity of certain stills being addressed and other information. It should be understood that the remote access station 76 shown in FIG. 2 is representative of each of the remote access stations and that in the preferred embodiment, up to seven remote access stations can be used to control the apparatus 70. The internal access station keyboard indicated generally at 83 in FIG. 1, as shown in the enlarged fragmentary view in FIG. 3, has more expanded operational capability than the remote access stations, whose keyboards have fewer function keys. As will be explained in detail hereinafter, the keyboard contains a large cluster of keys indicated generally at 84 and a smaller cluster of function keys 85 located on the left side of the keyboard. Additionally, a turn key controlled switch 86 may be provided to switch between normal and delete operations to safeguard against the possibility of inadvertent or unauthorized erasure of actively used stills.

Referring to the very simplified block diagram shown in FIG. 4 the apparatus receives a video input signal which is processed by record signal processing circuitry 88 and is then applied to record signal interface circuitry 89 which directs the signal to all of the disc drives 73. Gating circuitry located within a selected disc drive 73 is enabled to allow the signal to be recorded on a selected drive. More than one disc drive 73 can be simultaneously selected for recording the video signal provided by the record signal interface circuitry 89. Switcher circuitry can be substituted for the signal interface and associated gate circuitry so that the signal provided by the record signal processing circuitry 88 is coupled only to selected disc drives having the disc packs 75 upon which the signal is to be recorded. During playback, a signal originating from one of the disc drives is applied to the playback switching circuitry 90 which directs it to one of the playback channels 91, each of which provides a video output channel. A computer control system 92 is interfaced with the record processing circuitry, signal interface and switching circuitry and disc drives for controlling the overall operation of the various components of the apparatus and also interfaces the remote access stations and internal access station. As will be hereinafter more fully explained, an operator can select a particular disc in which to store a still, provided that the disc pack is on-line, i.e., it is physically loaded on one of the disc drives 73. In this regard, it should be understood that the apparatus addresses disc packs rather than disc drives for the reason that the apparatus is adapted to identify up to 64 separate disc packs, only one of which can be located on a disc drive at any one time. Thus, in the event the apparatus has two disc drives, only two disc packs can be on-line at one time. The operator can use an access station keyboard 83 to enter the address of a disc pack upon which he wishes to record a still and, through the interaction of the computer with the disc drive on which the selected disc pack is loaded, can carry out the recording operation on the selected on-line disc pack. Similarly, an operator can play back a still frame from the disc pack on one of the disc drives and can define the playback channel that he wishes the still frame to be layed through.

The apparatus has four major operating modes or conditions, i.e., (1) record/delete, (2) playback or reproduce, (3) sequence assembly and (4) sequence play. The record and play operations will be initially described, while referring to FIGS. 6 and 7 which illustrate somewhat simplified block diagrams of the signal flow paths during recording and playback, respectively, with respect to one of the disc drives 73.

Turning first to the record signal flow block diagram of FIG. 6 the composite video input signal is applied to the input stage circuitry 93 where clamping of the signal takes place and the synchronization and subcarrier components are stripped from the composite video signal. The input stage also regenerates the synchronization (hereafter often referred to merely as "sync") and subcarrier signals for later use during reproduction and, accordingly, the regenerated sync and subcarrier signals are directed to a clock generator 94 which also generates reference signals that are used by the downstream elements during operation.

The clamped analog video signal with the color burst component is then applied to an analog-to-digital converter (A/D) 95 which provides an output signal at a sample rate of 10.7 megasamples per second, with each of the samples comprising 8 bits of information.

The digital video signal is a non-return to zero code (NRZ) which means that it is a binary code defining a ONE as a high level and a ZERO as an equivalent low level. The digitized video signal appears on 8 parallel lines, i.e., one bit per line, which is applied to an encoder and sync word inserter 96 which converts the digitized video into a special recording code (referred to herein as a Miller code or a Miller squared code) that is particularly suitable for digital magnetic recording in that it minimizes DC content of a data stream. The circuitry also inserts a synchronizing word on alternate television lines with respect to a particular phase angle of the color subcarrier as represented by the color burst sync component. The sone word is used as a reference for correcting time base and skewing errors that occur during playback among the eight parallel bits of data that must be combined to define the value represented by each sample. The digital video information in the eight parallel lines is then applied to a recording amplifier circuitrv 153 and head switch circuitry 97 associated with the selected disc drive 73 which switches between two groups of eight recording heads for recording the digitized video signal by the disc drive. The disc drive is servo controlled so that its spindle rotational speed is locked to vertical sync, with the rotational disc speed being 3600 revolutions per minute. By locking the spindle drive to vertical sync, the apparatus records one television field per revolution of the disc pack and simultaneously records the eight data streams on eight disc surfaces. At the completion of recording one field, the recording amplifier circuitry 15 and head switch circuitry 97 is commanded to activate another set of heads for simultaneously recording the second field of a television frame on another set of eight disc surfaces so that a full picture (i.e. two interlaced fields) is recorded on two revolutions of the disc drive, using 16 heads. Each disc pack located on a disc drive preferably contains 815 cylinders, each of which has 19 recording surfaces and can therefore store 815 digital television frames. There is one read/write head for each of the 19 disc recording surfaces of a disc pack and all heads are mounted vertically aligned on a common carriage whose position is controlled by a linear motor. It should be understood that a cylinder is defined to comprise all recording surfaces that are located on the same radius of a disc pack. However, the term track, rather than cylinder, is preferred herein and, accordingly, a track is meant to include all recording surfaces on a same radius, i.e. all surfaces on a cylinder. Thus, an addressed track for recording or playing back a still actually referes to the 19 individual surfaces on the cylinder available at that radius, Of the 19 surfaces that are available for recording, one is used to record the address and other housekeeping information, rather than active video information, and it is specifically referred to as the '-data track". Two of the 19 surfaces are available for recording a parity bit and 16 surfaces are used to record the picture data, as will be explained further hereinbelow. Also one of the heads, generallv referred to as the servo head, travels on the 20th disc pack surface that contains only servo track information prerecorded by the pack manufacturer. The servo tracks carry out primarily determined by quantization noise rather than recording media and preamplifier noise as is the case with conventional videotape recorders. Thus, the present apparatus delivers a signal to noise ratio of about 58 dB and effects such as moiré and residual time base error do not exist, the digital random error of the storage channels being typically low enough to make occasional transmission errors virtually invisible.

By recording a digital data stream at a rate of 10.7 megabits per second on each of the eight disc surfaces, the linear packing density of the apparatus is about 6000 bits per inch which is about 60% greater than is used in conventional disc drive usage in data processing.

During playback and referring to FIG. 7, the heads read, i.e., reproduce the digital video information from the eight surfaces per field and obtain the recorded channel encoded digital video signal from the two fields of each frame. The reproduced signal is applied to a playback amplifier circuitry 155 and head switch circuitry 97 associated with the selected disc drive 73 which amplifies the data streams of digital video information carried by the eight data bit lines and applies the same to equalizer and data detector circuits 99. The equalizer compensates for phase and amplitude distortion introduced to the signal by the band limiting effects of the record and reproduce processes and insures that the zero crossings of the reproduced signal are distinct and accurately positioned. Following equalization, the channel encoded signals in each data bit line are processed as described hereinbelow for transmission to the playback circuitry of the signal system over a twisted pair line. The processed channel encoded signals are in the form of a pulse for each zero crossing or signal state transition of the channel encoded signal. The twisted pair lines for the eight data bits of the digital video information apply the processed channel encoded signals to the decoder and time base corrector circuitry 100 of one or more of the playback channels 91 of the apparatus. the decoder and time base corrector circuitry 100 reprocesses the received signals to place them in the channel encoded format, decodes the signal to the non-return to zero digital form and time base corrects the digital signal with respect to station reference to remove inter-data bit line time displacement errors (skew errors) and timing distortion within each of the data streams carried by the data bit linest To facilitate processing of the reproduced signals, phase continuous clock signals are used to time the operation of the decoder and time base corrector 100 and following circuitry. As will be explained in more detail hereinbelow, this prevents the time base corrector portion of the circuitry 100 correctly positioning the synchronization word in alternate reproductions of the picture frame. Thus. the time base corrector portion of the circuitry 100 serves to align the eight bits defining a single sample and remove timing distortion in each of the data bit lines relative to station reference. However, the aforementioned mispositioning of the synchronization word would lead to horizontal displacement of the picture frame upon alternate reproductions and resulting visible jitter in the displayed video image. It should be realised that each playback channel 91 is provided with decoder and time base corrector circuitry 100 and within each playback channel each of the eight data bit streams travels through a separate decoder and time base corrector. The output of the circuitry 100 is then applied to a comb filter and chroma inverter 101 which separates the chroma information and selectively inverts and recombines the signal for reconstruction of a four field NTSC sequence. This reconstructed digital signal is applied to circuitry 127, which adjusts the mispositioning of the synchronizing word in alternate reproductions of the recorded two fields of the video information and applies the adjusted video signal to a digital-to-analog converter 102 which provides an analog video signal. The new sync and burst are then added by a process amplifier 103 to produce a composite video analog output signal of the playback chammel 91 as is desired.

The general operation of the apparatus will now be described in the context of an operator using either the internal access station or remote access station keyboard for performing the various functions that can be carried out using the apparatus. The apparatus also can be operated from the keyboard of an auxiliary access panel 116 (see FIG. 8) connected to operate through either a remote access station 76 as shown in FIG. 8 or an internal access station 78. As previously mentioned, the remote access station shown in FIG. 2 has a leftward cluster of functional keys 85 as does the internal access station located in the apparatus bay 72 shown in FIG. 1. The remote access station left cluster 85 has only four functional keys as contrasted with nine keys on the internal access station, so that more functional operations can be carried out at the internal access station than a remote access station.

More specifically, the internal access station has a total of nine functional keys, one of which is a spare, the others of which can be pressed to place the apparatus in a particular condition, including the following: PLAY, for playing back or reproducing a still image; REC/DEL. for a recording or deleting operation; SEQ ASSY, for assembling a sequence of stills; and, SEQ PLAY, for playing a sequence of stills. These four operations can be carried out by an operator at the internal access station as well as at a remote access station.

However, in addition to these described functional operations, the internal access station also is adapted to carry out the following additional operations: E-to-E, for operating in an electronic to electronic operation where the video input signal is processed through the entire record circuitry up to the disc drive and is then applied to the playback switcher and processed back through the playback circuitry so that virtually everything is done to the video input signal except record it on the disc drives (essentially a testing operation); PACK IDENT, for checking the identification data that is recorded on the data tracks of a specified disc drive; PACK DEFINE, for entering the identification data onto all data tracks of a new pack that may become part of the disc pack library; PACK DUPE, for making an exact duplicate disc pack which contains the entire digital video information stored on a specified disc pack. Thus, the above eight functional operations broadly define the operation of the apparatus in terms of the modes or conditions of operation.

Each of the remote access stations 76 as well as the internal access station 78 has the rightward cluster of keys 84 and is identical for both stations. As can be seen from FIG. 3, the keyboard cluster 84 includes the numerals 0 through 9 for entering addresses, sequence list items and the like, "+1" and "-1" keys for incrementing and decrementing entered still addresses by one, respectively, keys A, B and C for entering video channel or sequence list letters, a LIST bar for loading sequence list items into memory, an EOL key for informing the computer system that the last item on a sequence list is to be entered, a KYBD RTN or keyboard return key for changing the origin of the display's message and an INITIATE bar for generating initiating commands to complete the entry of data into the computer system.

Additionally, the entire keyboard 83 has a DEL/EDIT ENABLE bar 104 which, when activated in conjunction with the previously mentioned key switch 86 allows individual still frames in bulk tracks or whole sequences in working tracks to be deleted. In this regard, there are 64 tracks of the 815 tracks on each disc pack that are defined as working tracks (addresses 1-64), and it is on these tracks that sequences of stills are assembled for play; except for two, the remaining tracks are defined as bulk tracks which provide the permanent library or file. Certain precautions and constraints are built into the apparatus to insure that stills are not easily, or at least inadvertently, erased or otherwise destroyed.

Thus. the DEL/EDIT ENABLE bar can be activated to allow edit of material on the working tracks, but must be used together with the key switch 86 to allow deletion of stills from the bulk storage memory or deletion of an entire sequence of stills in the working tracks.

As is shown in FIGS. 1 and 2, each of the access stations 76 and 78 have a display 82 that is used to present message origin codes, operator data entries, computer system requests and responses. The display unit is an alpha numeric dot matrix, self scanning display preferably having a capacity of 32 characters. The computer system 92 is preferably programmed so that display words and symbols appear which will indicate a status, request or identify an improper or illegal entry or other error. Moreover, as will be hereinafter described, when an operator enters a mode on the leftward cluster of one of the access stations, the mode identified requires the entry of data in a specific order. When a mode is selected, the display guides the operator through the entry sequence with a cursor symbol which advances through the display until all elements of the data message have been entered. The entry can be cleared and corrected at any time prior to the INITIATE bar being depressed which initiates operation by the computer system 92. On receipt of the data, the computer system 92 switches the display to a return message (RTN) which may either be a data validation or an equipment status response. If the data is validated the selected operation will be or has been performed.

Each mode of operation identified in the leftward cluster of keys 85 is performed in three basic steps. The operator first presses a mode select button, then enters data in the form of channel selects, storage addresses and instructions and finally depresses the INITIATE bar to request execution of the operation.

The various modes will now be described in terms of the interaction of the display and keyboard controlled by the operator.

In the PLAY mode, random access to any on-line rasher than still is provided, i.e. any still image that is located on a disc pack that is on a disc drive rather than being in storage. A still is selected by entering the video channel letter (A, B or C when three channels are provided) and a five digit still address number. The computer system 92 accesses a requested still when the INITIATE bar is pressed and the address is shown on the display. If an illegal or off-line still address is requested, the computer system will so advise the operator through the display. If the address read from the still's identification data differs from the entered address and the computer system cannot resolve the discrepancy (a true tally error), the output video will go to black and an error message will be shown on the display. If an adjacent still in a pack is to be addressed, the next or previous still can be addressed by pressing either the "+1" or "-1" key and then the INITIATE bar.

To compose a sequence list, the operator enters on-line or off-line five digit addresses into the memory which is divided into two sections of 64 items each. One suction is identified as LST A and the other list LST B. Each list includes item numbers 1 through 64.

To list an address, the still is first selected by entering the video channel letter and the five digit still address. The LIST bar on the keyboard is then pressed and the first item number and list identification will be requested of the operator. When these characters have been entered into the keyboard, the INITIATE bar is pressed and transfer into the memory occurs. The display switches to return (RTN) message and advises the operator of the completed transfer. The next still can then be selected for listing. The item (ITM) number is automatically incremented by one by the computer control system so that subsequent listings are made simply by entering the still address, pressing the LST and then the INITIATE bar. Following the last ITM number on a list, the END-of-LIST (EOL) button must be pressed before pressing the INITIATE bar. If it is desired to view a still before listing, the still's address is entered and the INITIATE bar is pressed before the LST bar.

Subsequent listing of the still is effected as described above. If the selected still is off line it can still be stored in the list, but the still obviously cannot be viewed. From the sequence list stored in memory, a sequence can be assembled as will be described hereinafter.

In the record/delete mode, recordings can be made on deleted bulk tracks and assembled sequences can be deleted as can occupied bulk tracks. It is also possible to over record individual stills of assembled sequences in this mode. The normal procedure for recording a still is to enter the letter of the video channel through which the recorded video is to be monitored, the two digit pack address, i.e., 1 through 64, in which the still is to be recorded and three zeros in lieu of a track address since the still is to be recorded on the next available track. Upon pressing of the INITIATE bar the computer system 92 automatically searches for the next available deleted track in the address pack and, following a true tally check, makes a recording. The search occurs in the computer system's status memory and does not require stepping through the tracks to find a deleted one. After recording, the RTN display is updated by the computer system to reflect the five digit address where the still was recorded. If a recording was not permitted, the operator is so notified. When an off line pack is placed on a drive to make a next available track recording, an initial search of all tracks using the pack identification mode is necessary to establish their status in the computer system memory. As long as the pack remains on line, subsequent next available track recordings do not require that this initial search be repeated.

If it is desired to record a still on a specified track, the operator must enter the video channel letter and the five digit address before pressing the INITIATE bar. If a recorded still is present on the specified track, the recording will not occur and the operator will be notified through the display that the track is occupied.

To delete a still, its 5 digit address must be entered and the delete function is initiated by actuating the NORM/DEL key switch and then pressing the DEL/EDIT ENABLE and the INITIATE bars simultaneously. If however, the INITIATE bar is pressed prior to pressing the DEL/EDIT ENABLE bar, the track's video is available for viewing. After viewing, the delete operation can be reinitiated by simultaneously pressing the DEL/EDIT ENABLE and initiate bars. This permits visual checking of a still before it is deleted.

To delete an entire assembled sequence of stills, or a portion of the sequence concluding with the last still of the sequence, the ITM number of the first still to be deleted in the sequence is entered in place of the normal track address entry. The computer system automatically recognizes that the entered number defines a working track and not a bulk track and initiates the deletion of the sequence. The NORM/DEL key is actuated and the DEL/EDIT ENABLE and INITIATE bars are simultaneously pressed to effect the deletion. Deletion will cease after the item identified as EOL (end of list) has been deleted.

In the event last minute editing of a completed assembled sequence is desired it is possible to over record assembled stills in this mode by addressing a working track by its five digit address and simultaneously pressing the DEL/EDIT ENABLE and INITIATE bars. It should be understood that the over record capability does not apply to the bulk tracks, only to the working tracks as previously mentioned.

To assemble a sequence of stills, the SEQ ASSY button is depressed. This mode causes a sequence of items in a sequence list to be automatically assembled in a defined pack. To assemble the sequence, the two digit address of the pack to receive the assembled sequence is entered, followed by the first ITM number in the sequence and the LST letter. When the INITIATE bar is pressed, the computer system automatically assembles the on-line stills onto the working tracks of the addressed pack. If an off-line still is encountered during the assembly operation, off-line status indication will appear on the display. When assembly of on-line items is completed, each off-line pack which includes list items is identified on the display. To add an off-line still to the assembly. a separate assembly must be performed while the disc pack containing that still is on4ine. It is necessary to perform assembly as many times as there are different off-line pack addresses in the sequence list. For each assembly operation, the previously assembled stills are not disturbed. On each disc pack, a working track is dedicated to each item number, providing a maximum assembled sequence length on each ack of 64 items. As each item in a sequence is assembled onto a working track it is recorded with an OCCUPIED status indication. The status prevents an item from another sequence being assembled onto the same working track.

In the sequence play (SEQ PLAY) mode of operation, access to assembled stills in the working track of a pack through their respective sequence item numbers is allowed. To play a sequence, the video channel letter, two digit pack address, initial ITM number must be entered. When the INITIATE bar is pressed, the still assigned to that ITM number is accessed. The RTN display is then updated to contain the accessed still's address, video channel letter and ITM number. Also, the keyboard ITM number is automatically incremented by one so that sequential items in a sequence list can be accessed merely by pressing the INITIATE bar with no new data entered. To skip a next item in a list, the "+1" key in the rightward cluster is pressed which increments the keyboard ITM number by two. Similarly, pressing the "-1" button will decrement the item number by one. When the last ITM in a sequence has been played, the end of list EOL is displayed. If the INITIATE bar is pressed after the EOL item is played, playback stays at the EOL item.

With the addition of an auxiliary access panel, stills stored in the working tracks of disc packs located on two drives can be accessed for a sequence play operation. In the apparatus described herein, the auxiliary access panel need only include two keys; and INITIATE bar and an auxiliary access panel select key. An access station is constructed and interfaced with the auxiliary access panel so that the functional keys of the access station are used to enter the data associated with operations to be performed through the auxiliary panel. Once the conditions for sequence play operation have been set for the access station and for the auxiliary access panel, stills from the disc pack on the drive controlled by the auxiliary access panel can be accessed by operating the INITIATE bar of that panel while stills from the disc pack on the drive controlled by the access station can be accessed by operating that access station's INITIATE bar. The display of the auxiliary access panel together with the display 82 of the associated access station are updated as described above to keep the operator or operators informed of the status of the sequence play operation.

The E-to-E mode is set up by pressing the E-to-E button in the leftward cluster in the internal access station and allows bypassing the disc pack to evaluate video performance on a channel independent of the recording and playback process. When selected, the digital video input to a drive is sent directly to the video playback channel and, in this mode, it is possible to select individual video channels for the signal path. To perform the operation, the video channel letter, disc drive number is entered and the INITIATE bar is pressed which causes E-to-E video to be available for monitoring. Pressing the INITIATE bar again returns the system to play mode and the disc video may be viewed. The E-to-E mode is useful in performing diagnostic and maintenance checks to determine performance characteristics.

To enter the pack identification mode, the PACK IDENT key is pressed. This mode provides a means for reading and storing in the computer control system memory the identification data recorded on all data tracks of a disc pack. When this mode is selected and the video channel designator and disc drive number are entered, the INITIATE bar pressed, then a check of each track in the disc pack is made. The number of erroneous pack addresses encountered in the check is also displayed.

The pack definition mode is entered by depressing the PACK DEFINE key and this mode facilitates entering new packs into the library. One of the disc drives, for example, drive No. 1 is designated as the defining disc drive so that any pack on this drive automatically becomes a new pack when this mode is initiated. The mode is initiated by entering the new two digit pack address, activating the NORM/DEL key switch 86, then pressing the DELIEDIT ENABLE key and the INITIATE bar simultaneously. At the disc drive, the pack's data tracks are recorded with new identification data and each track is recorded with a deleted indication. Completion of the definition operation is signalled by a FINISHED message on the display.

In the pack duplication mode, achieved by pressing the PACK DUPE key. an exact duplicate of the entire digital video information recorded on a disc pack can be made. In this mode, one of the disc drives, for example, drive No. 1 is defined as the source and another defined as the receptor for the duplication operation. To initiate the pack duplication mode, the operator enters the two digit pack address, actuates the NORM/DEL key switch 86 and then presses the DEL/EDIT ENABLE and INITIATE bars simultaneously. The apparatus automatically transfers the contents of each of the source packs' tracks to the corresponding tracks in the pack that is on the receptor disc drive. The receptor pack number becomes the pack number that was entered following selection of the duplication mode. Completion of the duplication operation is signalled by a FINISHED message on the display.

It should also be understood that the keys in the leftward cluster 85 which define the mode of operation that is to be carried out preferably are of the type which light when they have been placed in an active status. Thus, when the play operation is selected by pressing the PLAY key, it will light and will remain lighted until the apparatus is taken out of that mode of operation.

The flow chart of the control program that carries out the above functional description in terms of the operation of the access stations controlling the computer control system 92 is contained in FIGS. 63.

Referring more specifically to the computer control system 92 shown in the block diagram of FIG. 4, it is shown in more detail in the block diagram of FIG. 8. The computer system 92 is shown to comprise a central processing unit or CPU 106 and associated program storage memory unit 107 communicating with several interfacing devices to effect control of the various devices used in the operation of the apparatus. A single main bus 105 is provided for transferring both address and data information between the CPU 106, the memory unit 107 and the several interfaces, the address and data information being time multiplexed along the bus 105. Interrupt bus 143 comprised of several lines is provided to connect the CPU 106 to the access stations used by operators to direct the performance of functions by the apparatus. Whenever an access station requires the service of the CPU 106, the station causes an interrupt command to be sent by the remote access station interface 115 to the CPU over a line of the bus 143. This causes the CPU to interrupt its operation and service the calling station. In addition, a control bus 144 comprised of several control lines is provided to interconnect the units, and interfaces and access stations for transmitting control, timing and status information between them. Under the direction of the control program stored in the memory unit 107, the CPU 106 interprets a set of instructions received in response to the operation of an access station, panel or other system access device and executes the necessary routines and arithmetic functions to direct the computer system 92 to cause the requested functional operations to be performed by the apparatus. The manner in which the control program causes the CPU 106 to execute received instructions to perform the various functional operations capable of the apparatus is described in the flow chart contained in FIGS. 63. The control program described by the flow chart is arranged to operate with a CPU manufactured by Digital Equipment Corporation and identified hereinbelow.

To effect control of the apparatus, the CPU 106 and memory unit 107 are connected through the main bus 105 to a central processing unit interface 108 that includes an address decode unit 113 which identifies the system unit selected for either receiving information from or transmitting information to the CPU 106. For the Digital Equipment Corporation manufactured CPU, a 16 bit address is employed to identify the selected system unit. The 3 most significant bits of the 16 bit address identifies whether a peripheral device or one of the memory banks in the memory unit 107 is selected for interconnection with the CPU 106.

The next 13 most significant bits of the address forms the address word identifying the particular address location requested within the selected system unit. A byte organized addressing scheme is employed in the Digital Equipment Corporation manufactured CPU and the least significant bit of the address word identifies whether an odd or even byte address is received.

The CPU 106 operates asynchronously with the other system units comprising the apparatus. However, the other units operate synchronously with respect to a system clock.

The time interfacing of the asynchronously operated CPU 106 and the other synchronously operated system units is effected at the address time during the address/data multiplex cycle of the main bus 105 and is accomplished by a bus sync signal issued by the CPU 106 at the address time and transmitted over one of the control lines 144 to the CPU interface 108.

The CPU interface 108 is responsive to the bus sync signal to issue the appropriate device select signal determined by the address word at address time and, thereby, permit the selected system unit to be interfaced with the CPU 106.

In the apparatus described herein, several peripheral devices are employed for performing the various functional operations desired of the apparatus. When the 3 most significant bits of the 16 bit address identifies that a peripheral device has been requested for interfacing with the CPU 106, the address decode unit 113 is commanded by the CPU to decode the 13 bit address word and identify which of the 21 separate device select lines are to be activated for effecting the interface between the CPU and the requested peripheral device. Six of the device select lines are used to activate either a teletype interface device 109 for communicating with an external teletype keyboard 110, or a paper tape reader 111 or a read only memory 112 for receiving from or transmitting to the CPU 106. The 15 separate device select lines which go to the right in FIG. 8 as shown by the group of lines 114 are used to activate additional peripheral devices for interfacing with the CPU 106.

Control signals issued by the CPU 106 over control lines 144 to the requested peripheral unit determine in accordance with the functional operations to be performed by the apparatus whether the requested peripheral device is conditioned to receive from or transmit to the CPU 106. With respect to the remote access station interface 115, a UART clock timing signal is generated by the CPU interface 108 provided to this interface over line 1121 when it is requested for interfacing with the CPU 106.

With reference to the peripheral devices associated with the device select lines 114, a remote access station interface 115 interfaces the bus 105 with the remote access stations 76, an auxiliary access panel 116, via the remote access station or the internal access station 78 other interconnecting lines that are necessary for controlling the timing and synchronization of the circuitry represented by the various blocks. The interconnection of the signal system with the computer control system (described in conjunction with the block diagram of FIG.

8) is also shown, in that the input and output lines from the various blocks in FIGS. 9A and 9B which have an asterisk adjacent to them are lines which extend to the computer control system 92.

It should also be understood that the apparatus of the present invention will be described herein with respect to use in an NTSC system which has a television field comprised of 525 lines, horizontal synchronizing pulses occurring at a rate of about 15,734 Hz (often referred to herein as "H sync") which means that the period between successive H pulses is approximately 63.5 microseconds. Moreover, the vertical blanking rate in the NTSC system occurs at a 60 Hz frequency and the chrominance information is modulated on a subcarrier signal having a frequency of about 3.58 megahertz (MHz). Because of the relationship of the color sub carrier phase with respect to horizontal sync, NTSC color signals have a four field sequence, which is commonly referred to as a color frame. The subcarrier frequency of 3.58 MHz will often be referred to herein simply as SC which means 1 times the subcarrier frequency and, similarly, other commonly used clocking frequencies in the described apparatus include 1/2SC, 3SC and 6SC. The 3 times subcarrier frequency (3SC) often occurs for the reason that during sampling of the analog composite video signal for digitizing the signal, a sampling rate of 3 times the subcarrier frequency, i.e., 10.7 MHz is used. The composite video signal of an NTSC system is illustrated in FIGS. 5A and 5B.

Referring again to FIG. 9A, but before discussing the functions of each of the blocks shown therein, some broad general considerations should be understood with respect to the overall operation of the illustrated signal system. Firstly, the video input signal that is fed to the video input circuitry 93A is an analog signal which is processed and applied to an analog-to-digital converter 95. The output of the converter contains the video information in digital format and the digitized data is further processed and recorded on a disc pack in a digital format. Similarly, it is played back from the disc pack, time base corrected and chroma separated and processed using digital techniques and is not converted to an analog signal until one of the final steps where the digital-to-analog converter and sync and burst insertion circuitry 102, 103 provides the analog composite video output as shown.

In the analog-to-digital converter 95, the analog composite video signal is sampled three times per nominal subcarrier cycle, or at a sampling rate of 3SC (10.7 MHz), and each sample is digitally quantized into an 8 bit digital word. A sampling clock having a frequency of three times or any odd multiple of the NTSC subcarrier frequency is necessarily an odd multiple of one-half of the horizontal line frequency. If such a sampling clock is phase continuous from line to line, its phase at the start of consecutive lines changes. Using such line to line phase continuous sampling clocks will result in the instantaneous amplitude of the analog signal being sampled during consecutive lines at different times relative to the start of the consecutive lines at different times relative to the start of the consecutive lines.

Because of this, the quantized samples are not in vertical alignment from line to line.

Vertical alignment of the samples from line to line is desired to facilitate the use of a digital comb filter to obtain a separated chrominance component of a television signal by combining quantized samples from three consecutive (all odd or all even fields) television lines of a television field, which may be designated T (for top), M (for middle), and B (for bottom) in proportion to the formulae (Chrominance) C = M - 1/2 (T + B) (Luminance) Y = M + 1/2 (T + B).

It should be appreciated that if the samples of the NTSC television signal are taken at an even multiple of the subcarrier frequency, the comb filtering technique would be ideal because the phase of the sampling clock would not change from line to line. Hence, the digital code words or quantized samples would describe the instantaneous amplitudes of each line of the analog signal at the same times relative to the start of each line and all of the samples in three consecutive lines would be aligned vertically from the top to middle to bottom lines.

The lack of vertical alignment of the samples of consecutive lines when using a 3SC, line to line phase continuous sampling clock can be more readily appreciated with reference to FIG. 9C(1) which shows a number of cycles of subcarrier in television line 1 that are sampled by the positive transition of a 3SC sample clock (FIG. 9C(3)) wherein the upward transition has an arrow depicting an "X" sample point that is also placed on the subcarrier for television line 1 at every sample point (FIG. 9C(1)). As shown, there are three samples for each cycle of the subcarrier. However, during television line 2, i.e., the next consecutive line, the subcarrier has a reversed phase as shown in FIG. 9C(2) and similarly, the sampling clock 3SC is of opposite phase (FIG. 9C(4)) relative to its phase in line 1 (FIG. 9C(3)) so that during television line 2 the samples are taken where shown by the X's of the television line 2 subcarrier (FIG. 9C(2)) on the upward transitions and it is seen that the X samples from line 1 to line 2 are misplaced by 60 with reference to SC, which detrimentally affects the response of the comb filter, which utilizes the instantaneous amplitude of the analog signal in the above mentioned equations for properly deriving the chrominance information. It should be appreciated that the samples taken on all odd lines will be vertically aligned and that the samples taken on all even lines will be vertically aligned but that the samples taken on even lines will be displaced 60 with reference to SC relative to those samples on the odd lines.

To avoid the problem created by sampling at an odd multiple of subcarrier frequency, i.e., 3SC in the present apparatus described herein, vertical alignments of samples in all lines can be achieved by changing the phase of the sampling clock on alternate lines. In the examples shown in FIG. 9C, reference is made to FIG. 9C(5) which illustrates the 3SC sampling clock for television line 2 which has its phase reversed relative to what it would have been for television line 2, which is shown in FIG. 9C(4). By sampling on the upward transitions at the "0" sample points, samples marked by the "0" on the subcarrier for line 2 result as shown in FIG. 9C(2). Thus, the sample points in the subcarrier for television line 1 ("X's") are vertically aligned relative to the sample points ("0's") that are sampled using the alternated phase sample clock shown in FIG. 9C(5) rather than what would have normally occurred as shown by FIG. 9C(4). This technique is commonly referred to as phase alternate line encoding or PALE and the terms PALEd, PALEing and the like will commonly be referred to throughout the description of the apparatus described herein.

While the apparatus described herein utilizes comb filtering techniques together with a sampling rate of 3SC or 10.7MHz and requires the use of a PALE sampling clock, it should be appreciated that a 4SC sampling frequency would eliminate the need for PALEing. The use of a 4SC sampling frequency is within the contemplation of the apparatus described herein in the event that the frequency response of the recording media, i.e., the disc packs on the disc drives is sufficient to permit operation at the 4SC, 14.3 MHz frequency. In this regard, it is to be appreciated that standard disc drives used in data processing applications typically operate in the range of about 6-1/2 megabits and the recording at a rate of 10.7 megabits represents a significant increase in the packing density of the disc packs themselves.

Another important aspect of the operation of the present apparatus that is a result of the use of PALEing will also be described with respect to FIG. 9C. By changing the phase of the sampling clock on every consecutive line, a phase discontinuity necessarily occurs with respect to SC. It is more convenient during the channel encoding of the signal for use in subsequent recording to channel encode the digitally quantized samples with respect to a continuous phase clock, i.e., no phase discontinuities from line to line. For this reason during recording, the PALEd data that results at the output of the analog-to-digital converter 95 is clocked out of the channel encoder 96 with a clock that has a continuous (i.e., no discontinuities) 3SC phase from line to line. However, clocking the encoder with a line to line continuous phase clock shifts the data in time on alternate lines by 1/2 cycle of 3SC, which disturbs the line to line sample time alignment created by sampling with a PALE clock. Since during playback the chroma processing circuitry requires the samples of data to be vertically aligned from line to line, which was the reason that a PALE sample clock was used in the analog-to-digital converter in the first place, it is necessary to retime or reclock the data from the continuous phase clock back to the PALE clock so that the sample time disturbance is removed and the chroma processing comb filter can process the data without error. Succinctly stated, the converter 95 samples the analog signal using a PALE clock having line to line phase discontinuities. For recording, the encoder 96 encodes the PALE data with a line to line continuous phase clock, which requires, during playback and after decoding, the retiming of the NRZ information to a PALE clock for use bv the chroma processing circuitry. However, the latter retiming from a continuous to a PALE clock is not performed during transfer modes of operation when the video data recorded on one disc drive memory is played back to be transferred and recorded on another disc drive memory. In such cases, the line to line continuous phase data clocking of the played back video data is retained and the data is rerecorded without disturbing the data clocking.

The above considerations will now be described in conjunction with FIG. 9C where the PALE data for lines 1 and 2 are shown in FIGS. 9C(6) and 9C(7), respectively. The bits Al through El are consecutive bit cells that represent the instantaneous samples of the analog video signal that occur in line 1 corresponding to the X's shown in FIG. 9C(1), with each bit cell lasting a full clock cycle of the 3SC clock shown in FIG. 9C(3). Similarly. the line 2 bit cells A2 through E2 represent data that is derived by the sampling at the "O's" in FIG.

9C(2) using the PALE sample clock, which for television line 2 is shown in FIG. 9C(5). To clock the PALE data with a line to line continuous phase 3SC clock, arrows beneath the bit cells shown in FIGS. 9C(6) and 9C(7) depict the clocking points of the line to line continuous phase clock that produce the bit cells that are shifted and are in the relation shown in FIGS. 9C(8) and 9C(9). The start of each bit cell occurs at the clocking point and the level of the cell will be continuous through the bit cell interval so that the bit cells maintain their identity during the clocking.

To retime the data from the line to line continuous phase clock back to PALE clock so that the bit cells (samples) are vertically aligned as they should be, i.e., A2 is vertically aligned with Al, B2 with B1, etc., the retiming from the continuous phase clock to the PALE clock must be correctly done or misalignment of the bit cells will result. In this regard, the retiming or reclocking must be complementary, i.e., a bit cell that was clocked in the right portion thereof in a PALE-to-continuous reclocking must be left clocked in the continuous-to-PALE reclocking to insure proper playback. Thus, given the line to line continuous phase clocked data shown in FIGS. 9C(8) and 9C(9), the solid arrows illustrate the proper complementary clocking for the two television lines and produce the retiming of the data to the PALE clock having the Al and A2 bits vertically aligned as shown in FIGS.

9C(10) and 9C(11). It should be noted that where bit cells that were right clocked going from PALE-to-continuous reclocking, are left clocked in the opposite conversion as is evident from viewing any of the bit cells, e.g., Al, with their associated clocking arrows in FIGS. 9C(6) and 9C(8). In the event that complementary clocking is not performed, then the bits will not be properly aligned as is shown by the dotted clocking arrows in FIGS 9C(8) and 9C(9) which produce the relationship shown in FIGS. 9C(12) and 9C(13). The reclocking from either PALE to continuous or the converse is performed at various locations as will be evident from the ensuing description.

It should also be realized that the NTSC television signal does not have any specified, defined relationship between the horizontal sync pulse occurring at each line and the phase angle of the subcarrier signal with the exception that the phase of the subcarrier changes 1800 from line to line. In other words, the phase angle of the subcarrier signal relative to the H sync signal can vary from one video source to another and this variance makes the H sync an undesirable signal to control the operation of the apparatus. Accordingly, the apparatus herein uses the input signal's subcarrier as represented bv the color burst sync component as the basic timing reference for the system and defines a new H sync related signal that is used for timing purposes instead of the signal's H sync. The new H xync related signal is chosen to be at a frequency of 1/2 of the nominal horizontal line frequency because it represents a whole number of cycles of the subcarrier frequency, i.e., two complete horizontal lines of subcarrier frequency or 455 cycles. Moreover, the H sync related signal is given a definite relation to the subcarrier, i.e., it is synchronized with respect to the phase angle of the subcarrier. In the record portion of the signal system a synchronizing word is inserted in the video signal on alternate television lines at a location corresponding approximately to that of the video signal's H sync and phase coherent with respect to a particular phase angle of SC generated from the video signal color burst subcarrier synchronizing component. The location of the new H sync related signal is defined at the beginning of each picture and is maintained for the duration of the picture to provide the video signal with an H sync related signal accurately and consistently defined with respect to the phase of the video signal's subcarrier. For the playback portion of the signal system, an H sync related signal designated H/2 is provided that is redefined to be coherent with respect to a particular phase angle of the reference input subcarrier, which phase angle is selectable through the playback system phase control.

The redefined H sync related signal, H/2. is used as a basic timing reference signal for the system during playback operations.

By using the redefined H sync related signal as the horizontal sync reference for the system, processing signals for recording, playback and other operations of the system is facilitated because a consistent time relationship is established between the video signal's cubcarrier and redefined H sync related signal.

Additionally, the use of internal horizontal and subcarrier reference signals that can be varied in time relative to the television station reference sync, permits timing control that will enable the television signal to reach a remote location at the proper time after having experienced the usual propagation delays that occur.

Referring again to the block diagram of FIGS. 9A and 9B, the analog video input is applied to the input of input circuitry 93A where several operations occur in the processing of the analog video signal before it is applied to the analog-to-ditigal converter 95. More specifically, the input circuitry 93A amplifies the analog video signal, provides DC restoration. separates the sync components contained in the video signal for use in generating timing signals for the signal system, detects the level of the tip of the H sync and thereafter clips the same. Moreover, the H sync is separated using a precision sync circuit for use in producing a regenerated sync. The circuit also produces a regenerated SC signal that is derived from the burst of the video input or, in the absence of burst, from an H/2 reference signal that is generated and is derived from the video input H sync.

It should be understood that the video input circuitry 93A and the reference input circuitry 93B shown in the lower left of FIG. 9A perform similar functions, the video input circuitry primarily for the signal recording portion of the signal system and reference input circuitry primarily for the playback portion of the signal system. Therefore, for convenience of manufacturing and service, identical circuitry is used. However, the input circuits are connected in the apparatus to receive only the input signals required to perform their respective functions and while the same signals are produced by each circuit, they are not all utilized from each circuit. The reference input to the reference input circuitry is the station reference color black video signal which contains all components of a color television signal except that the active video portion of it is at a black level. Thus, the burst, H sync and the like are present at the reference input circuitry 93B as they are at the video input circuitry 93A. In addition, the reference input circuitry 93B uses an H phase position adjusting circuit that receives control signals from a thumb wheel switch or the like (such as switches 81) for adjusting the H phase position of the regenerated H sync used in the playback portion of the signal system.

As shown, many of the output signals provided by the input circuits 93A and 93B are applied to the reference logic circuits 125A and 125B associated with the respective input circuits. The reference logic circuit 125A during the record mode of operation uses the inputs from the video input circuitry 93A, the analog-to-digital converter 95 and the computer control system 92 and through precision phase lock loop circuitry, generates a number of recording clocks at frequencies of 6 SC, 3SC, 1/2SC and a PALE flag signal. The PALE flag and 3SC signals are used by the reference logic circuit 125A to generate a 3SC PALE sampling clock signal whose phase is set for each line of the video signal by the PALE flag, which is at a frequency of H/2. The PALE flag signal changes state at that rate although it does so asymmetrically, i.e., the two states of the PALE flag signal are of unequal time intervals. It is made asymmetrical so that the sampling clock phase for the color burst portion of the video signal is constant with the phase of the subcarrier and only the portion of the television line thereafter has a sampling phase which is alternated on consecutive lines. This PALE clock is coupled to the analog-to-digital converter 95 and is the sampling clock signal for deriving the samples at 3SC or 10.7 MHz.

The reference logic circuit 125B uses inputs from the reference input circuitry 93B and the computer control system 92 and generates a clock reference signal at a frequency of SC and various other timing control signals. These signals are used in the operation of the apparatus in modes other than that of recording input video signals.

During the record and playback modes of operation, the reference logic circuits also generate servo sync signals for each of the disc drives for properly operating the disc drives at the proper phase.

During playback and other modes of operation other than that of recording input video signals, a reference clock generator 98 generates various clocks and additional timing control signals required by the various parts of the signal system used in such modes. The reference clock generator uses the inputs from reference input circuitry 93B, reference logic 125B, the playback portion of the signal system, an operator's control switch and generates clock signals at frequencies of 6SC, 3SC, SC and 1/2SC and various other timing control signals. The reference logic circuitry 125A and 125B and the reference clock generator circuitry 98 together comprises the signal system's clock generator 94 that provides the system timing control signals.

The clamped and H sync stripped analog video signal from the video input board is applied to the analog-to-digital converter 95 which converts the signal to an 8 bit binary coded signal in a PALEd NRZ (non-return to zero) format which is applied to the encoder switch 126. The analog-to-digital converter 95 is not shown in detail herein as it is identical in its design and operation to the one incorporated in the Ampex Corporation digital time base corrector No. TBC-800. More specifically, the schematic diagrams of the analog-todigital converter 95 are shown in the catalog No. 7896382-02 issued October 1975. The specific circuitry for the analog-to-digital converter is shown in schematic drawing No.

1374256 appearing on page 3-31/32 of the catalog and in schematic drawing No. 1374259 appearing on page 3-37/3 & of the catalog. These schematics are incorporated by reference herein.

The output from the analog-to-digital converter is then fed to an encoder switch 126 which comprises switching circuitry that ordinarily receives either the 8 bit digitized video data from the converter or from data transfer circuitry 129. As will be described hereinafter, the data transfer circuitry 129 enables the video information to be transferred from one disc drive to another disc drive as previously discussed with respect to the operation of the apparatus using the remote or internal access stations. During the transfer mode of operation, the digitized information is read off of the disc drive, decoded to the NRZ digital format, time base corrected and is then applied to the encoder switch which can select either source of digitized video information for the encoder 96. Because the channel encoded data recorded on the disc drives 73 has been clocked with a continuous phase clock, the NRZ data received by the data transfer circuitry 129 also is timed with respect to the continuous phase clock. Ordinarily, the data transfer circuitry 129 is provided with a PALE flag signal that is used to effect retiming of the NRZ digital data with respect to a PALE clock signal so that the data provided to the chroma separator and processing circuitry 101 is in the correct PALEd format. During the transfer mode of operation, this retiming is not necessary. The encoder switch 126 has circuitry for interrupting the coupling of the PALE flag signal to the data transfer circuitry 129 and thereby preventing the retiming of the NRZ data with respect to the PALE clock during the data transfer mode.

The encoder switch 126 is controlled by the computer control system 92 to gate the video data from either the input video or data transfer paths. It also switches between video and reference 6SC and 1/2SC timing signals since the reference timing signals are used during the data transfer mode and the video timing signals during the record mode. The encoder switch is also adapted to generate a signal that will produce a blinking cross through the TV image which is a visual indication that the still location or address for a still is unoccupied and therefore available for recording and also to provide signals for performing diagnostic functions. With respect to the sync word inserter, the encoder switch 126 couples the 8 bit digital video data from the analog-to-digital converter 95 and the timing signals derived from the timing reference to the encoder 96.

The 8 bit data from the encoder switch 126 is then applied to the encoder 96 which initially generates a parity bit and then encodes the PALEd data into a Miller squared channel code format, which is self-clocking, DC free, non-return to zero type of code.

While PALEd data is applied to the encoder, the output of the encoder is a 9 bit data stream (if parity is included) that has a phase continuity with respect to 3SC. The continuous phase clocked data is easier to process, particularly, during the decoding operations. The DC free code avoids any possible DC component that could occur due to a preponderance of one logical state over a period of time which could have an effect of disturbing the data in the playback process.

In limited bandwidth information channels which do not transmit DC, binary waveforms suffer distortions of zero crossing location which cannot be removed by means of linear response compensation networks. These distortions are commonly referred to as base line wander and act to reduce the effective signal to noise ratio and modify the zero crossings of the signals and thus degrade the bit reliability of the decoded signals. A common transmission format or channel data code that is utilized in recording and reproducing systems is disclosed in Miller U.S. Patent No. 3,108,261 issued October 22, 1963. In the Miller code logical 1's are represented by signal transitions at a particular location, i.e., at mid-cell, and logical 0's are represented by signal transitions at a particular earlier location, i.e., near the leading edge of the bit cell. The Miller format involves the suppression of any transition occurring at the beginning of 1 bit interval following an interval containing a transition at its center. Asymmetry of the waveform generated by these rules can introduce DC into the coded signal and the so-called Miller "squared" code used in the present apparatus effectively eliminates the DC content of the original Miller format and does so without requiring either large memory or the necessity of a rate change in the encoding and decoding.

The encoder circuitry 96 also generates a unique sync word in the form of a 7 digit binary number and inserts the sync word on alternate lines in a precise location determined by the 6SC and 1/2SC clock signals. In the record mode of operation, clock signals generated from the synchronizing components of the input video signal by the reference logic circuitry 125A are provided to the encoder circuitry 96 by the encoder switch 126 and result in the sync word being inserted at a location that approximately corresponds to where the video signal's horizontal sync pulse was previously located. In other modes of operation, the 6SC and 1/2SC clock signals are generated from the synchronizing components of the station reference color black video signal by the cooperative action of the reference logic circuitry 125B and reference clock generator 98. The encoder gates the H sync related sync word into the data stream on alternate television lines at the proper time relative to the regenerated subcarrier phase.

Data track information to be recorded on the data track of the disc drives 73 is also encoded by the encoder 96 prior to recording. The data track information is provided by the computer control system 92 through its data track interface 120.

With reference to FIG. 9B, the ten data streams of encoded digital data appearing at the output of the encoder 96 is applied to an electronics data interface 89 which is merely signal splitting and buffering circuitry which couples the encoded data to the three disc drives 73 for selective recording on a disc pack 75. Each disc drive includes a disc drive interface 151 adapted to receive the encoded digital data from the electronics data interface 89 and send it to the record amplifier circuitry 153 and head switch circuitry 97 for recording on an associated disc pack 75 as well as to receive reproduced or detected data from the playback amplifier circuitry 155 and head switch circuitry 97 and send it to the data select switch 128.

In addition, the disc drive interface 151 receives the multiplex servo reference signal through the electronics data interface and sends it to the timing generator (FIG. 39) of the disc drive control circuitry. This signal is selected by the computer control system 92 from either reference logic circuitry 125A or 125B. The timing generator employs the multiplex servo reference signal application to the generator 98 which has outputs of 6SC, 1/2SC, 11/2 and other signals for timing playback operations. The clock and timing signals, including the reference H/2 signal, are synchronized to the reference color subcarrier to facilitate processing of the reproduced video signals. The reference H/2 signal is defined with respect to a particular phase of the reference color subcarrier in the first line of alternate fields of the reference color black video signal. The generator's outputs are applied to the corrector 100, circuitry 129 and the processor 101 and to a circuit 127 that inserts blanking, performs selective bit muting, and provides a selected video picture signal for output by the signal systems when the heads associated with a disc drive coupled to the playback channel are moved between track locations. Because of the use of the redefined reference 11/2 signal in the data decoder and time base corrector 100, the synchronizing word contained in alternate reproductions of the two field video signal is mispositioned relative to the station reference H sync. This would introduce a jitter in the displayed video image if it were not corrected. The aforementioned mispositioning is corrected in the circuitry 127 by the insertion of corrective delay in the signal path upon alternate reproductions of the two field video signal. The generator 98 identifies which reproduction of the two field video signal sequence requires the delay by examination of a color frame rate signal, H drive signal and field index signal (all provided by the reference logic circuitry 125B) and the reference color subcarrier signal. In response to the identification, the reference clock generator 98 generates a frame delay switch signal that is coupled to the blanking insertion and bit muting circuitry 127 for controlling the insertion of the corrective delay. The 8 bits of digital information are then applied to the circuitry 102 and 103. Moreover, during the transfer and diagnostic modes of operation, the generator 98 supplies the basic timing clocks for the encoder 96 through the encoder switch 126 as shown.

During playback, the 10 bit parallel data stream comprising 8 bits of video data, the parity bit and data from the data track reproduced from a disc pack is amplified, equalized and detected by circuitry shown in FIGS. 24 to 28, 53 and 54 and is then applied through the disc drive data interface circuitry 151 to a data select switch 128 which can switch any of the outputs of the three disc drives onto one or more of three channels. Thus, the data select switch can switch the information from disc drive No. 1 into channel A, or to two channels while simultaneously applying a data stream from another disc drive onto another channel.

While information from two drives can not be simultaneously applied to a single channel, the converse is possible. The data select switch 128 comprises conventional switching circuits which are not set forth in detail herein.

Each of the detected nine bit streams of video data and parity data from the data select switch 128 is then applied to nine individual data decoders and time base correctors 100 which decode the data and the;l independently time base correct each of the nine data streams with respect to a common H/2 reference, which is defined with respect to the phase of the regenerated reference subcarrier, to remove any timing errors that may be present among the nine lines of data, i.e., it aligns all sync words so that each 9 bit parallel byte comprises the correct 9 bits of data. The other bit stream from the data track is coupled by the data select switch 128 to only the decoder portion of the decoder and time base corrector circuitry 100 and the decoded data track information is coupled to the data track interface 120 for transmission to the CPU 106. The time base corrector does its correction using a continuous phase clock. However, the data is again retimed with respect to a PALE clock by the data transfer circuitry 129, i.e., the phase of the signal is alternated by reclocking it at every horizontal line, so that the 8 bit data stream that comes from the data transfer circuitrv is a true PALEd signal again. The data transfer circuitry 129 also performs a parity check of the off disc data and performs error masking of individual byte errors when they occur by substituting what is likely to be the most similar previously appearing byte for the byte that was detected as being in error. In this regard, the byte that is substituted is the third previous byte, which is the most recent sample that was taken with the same phase relation to SC.

The output of the data transfer circuitry is applied to the chroma separator and processing circuitry 101 in the event that the video information is desired for viewing, as opposed to being recorded on another disc drive (transfer), in which case the data from the data transfer circuitry 129 is coupled to the encoder switch 126. The chroma separation and processing circuitry 101 works in the digital domain and separates the chroma information from the luminance using comb filter techniques and inverts the chroma information on alternate frames to form a four field composite NTSC signal that is then applied to the blanking insertion and muting circuitry 127 which inserts a reference black level during the blanking period, inserts grey level signals during the interval between the playback of consecutive stills, and performs bit muting operations if desired. The bit muting effectively mutes any bit or bits of an 8 bit television signal by shutting down that data bit stream and by so doing, achieves unusual visual effects in the resulting television signal such as producing exaggerated tones, ghostlike images and the like. The output from the blanking insertion and bit muting circuitry 127 is then applied to the digital-to-analog converter 102.

The digital-to-analog converter receives clock signals from the circuitry 127 and converts the data to its analog form and also inserts the sync and burst components of the signal to produce a full composite analog television signal.

While the foregoing generally describes the overall operation of the signal system in a general manner, a more specific description of each of the blocks that are contained in FIGS. 9A and 9B will be given either with respect to the separate functional block diagrams or the specific electrical schematic diagrams for the circuits themselves. Also where functional block diagrams are used to describe the operation of the individual blocks of FIGS. 9A and 9B, the electrical schematic diagrams corresponding to those more detailed block diagrams are also included herein.

The video input and reference input circuitry 93A and 93B broadly described with respect to the block diagram of FIG. 9A contain substantially similar circuitry in both locations, although different inputs are received by each and all of the outputs that are available from each are not used. During record operations, the composite video input signal to be recorded is applied to the video input circuitry 93A which is used to obtain a regenerated subcarrier signal, and various vertical and horizontal sync rate related signals that are used by the apparatus in the performance of the record operations. The video input circuitry also provides an amplified and filtered video signal suitable for feeding the A/D converter 95.

During playback operations, a reference color black video signal is applied to the reference input circuitry 93B which provides similar signals for use by the apparatus in the performance of the playback operations.

Referring more specifically to the block diagram for the video and reference input circuits shown in FIG. 10, the video signal is applied on line 200 into a video amplifier 201 which amplifies the signal and restores the DC component through a clamp 202. The clamp 202 samples the output of the amplifier on line 203 and produces a DC component on line 204 that extends to the amplifier 201. The DC restored video signal on line 203 is then passed through a low pass filter 205, the output of which appears on line 206 extending to a video gain control amplifier 207. The amplifier 207 is connected to another video amplifier 208 where a second clamp circuit 209 assures that the blanking level of the signal is at ground level by the application of a DC control signal via the line 210 to the video amplifier 208.

The output of the video amplifier appears on line 211 and is coupled by one of the lines 218 extending therefrom to the sampling input of the clamp 209. Line 211 also extends to a gated sync clipping circuit 212 as well as to a precision sync separator 213. A tip of sync detector 214 detects the level of the tip of sync and provides a corresponding signal level on line 215 that extends to a comparator 216 and to the separator 213. In the video input circuitry 93A, a remote video gain control signal on line 217 is also applied to the comparator 216 for controlling the gain control amplifier 207 from a remote location. In the reference input circuitrv 93B, the gain of amplifier 207 is not controlled from a remote location. The output of the tip of sync detector 214, which may contain alternating current ripple, is applied to one input of the precision H sync separator 213 while the other input to the separator is provided by one of lines 218 that extends from the output of the video amplifier 208. The two inputs to the precision sync separator 213 will both have AC ripple thereon if present in the signal and, accordingly. they are common moded so that the separator produces an AC ripple free precision separated sync on line 220 that is applied to miscellaneous sync circuits 221 and to an input of a horizontal sync phase detector 222.

Another of the lines 218 from the output of the video amplifier 208 extends to a less precise sync separator 219 that produces a generally less precise separated sync signal which is applied to a gate pulse generator 223, outputs of which appear on lines 224 that extend to both clamps 202 and 209 as well as to the tip of sync detector 214. When the horizontal sync signal is detected and separated. a gate is produced by the pulse generator 223 which closes the clamps as well as the sync tip detector at the appropriate time during horizontal blanking.

The clamp 209 is closed during burst time for a whole, integral number of cycles, rather than an arbitrary period, so that the blanking level of the video signal can be accurately obtained using integration techniques as will now be described in detail. The burst appears on line 225 which is applied to a burst limiter circuit 226 that is in turn connected to an amplifier 227 providing complementary outputs of the limited burst input. The output of the limiter circuit 226 is also connected to a burst presence detector circuit 228 having an output on line 229 that extends to a precision gate generator 230 as well as an output on line 260 that extends to a phase detector 231. When the presence of burst is detected, the precision gate generator 230 generates a precision burst gate signal that is coupled to enable the amplifier 227 and permit it to pass the middle three cycles of burst to apply them to the phase detector 231. The phase detector responsively provides an error signal to a voltage controlled oscillator 232 that reflects the difference in phase between the output of the oscillator and the phase of the burst cycles from the amplifier 227. The effect of the phase detector circuit controlling the oscillator 232 is to correct for longer term changes and not short term changes in the phase of the three cycles of burst that are used on every line as the subcarrier reference. The output of the oscillator 232 appears on line 233 after having been buffered by a buffer 234 and is a continuous regenerated subcarrier signal SC phase locked to the color burst when burst is present. However, if the burst detector circuit 228 fails to detect burst, the phase detector 231 compares the phase of an 11/2 signal with the output of the oscillator 232, the H/2 signal being produced by a sync generator 235 from an oscillator 236 controlled by the detector 222. This continuously regenerated subcarrier signal SC is coupled to the reference logic circuit 125A and, as will be desctibed in detail hereinbelow, is employed in the apparatus described herein to generate the 3SC PALE clock used by the A/D converter 95 to effect digitization of the video signal.

A control 237, is provided for use in the circuitry 93B to adjust the horizontal positioning of the regenerated sync. An 8 bit binary number is loaded into latches 238 by an operator-controlled thumb wheel switch or the like (for example, control switches 81 located by the internal access station 78) to preset a counter 239 which is clocked by a 400H clock derived from the oscillator 236. When the counter reaches its terminal count, it triggers a ramp generator 240 having an output 241 which feeds a second input of the detector 222. Thus, by adjusting the latches, up to plus or minus 20 microseconds can be inserted in the feedback loop on line 241 and the phase of the regenerated sync signal can be adjusted for horizontal positioning of the image represented by the video information signal. Since a delay in the feedback loop means that the regenerated sync will be advanced, the horizontal position control can effectively advance the video information signal to compensate for delays in propagation of a signal through cabling in a television station. As will be explained hereinafter in the detailed description of the reference clock generator circuitry 98, this horizontal phase position control is operated in conjunction with a subcarrier phase control whereby the amount of delay can be controlled in small increments, which in the embodiment of the apparatus described herein is about + 0.8 nsec.

The output of the oscillator 236 also is used by the sync generator 235, which is of conventional design for television signal processing equipment, to generate the various vertical and horizontal sync rate related signals indicated in FIG. 10. These sync rate related signals are generated with respect to the phase of the precisely regenerated H sync as provided by the phase detector 222 and, therefore, will always have a phase related to the input signal.

An important aspect of the circuitry shown in FIG. 10 is that the H sync of the video signal is clipped at precisely 1/2 its value and the level of the blanking is precisely clamped to ground. the regenerated subcarrier is phase locked with the burst and a precision horizontal sync signal is regenerated utilizing the precision sync separator. This signal is used by the sync generator 235 to provide a reset pulse (30 Hz field index pulse) for resetting a line identification or sync word inserter that will be hereinafter described. Since the clamp circuitry 209 examines for a zero average level of video at burst time using a clamping pulse which lasts precisely a whole number of cycles of burst. there is no need for low pass filtering the video and rejecting the burst before clamping is performed. This is due to the fact that resulting integration of the burst is equal to zero and there is no H/2 ripple introduced by integrating a signal that does not contain complete cycles of burst.

The block diagram shown in FIG. 10 describes the functional operation of the input circuitry and specific circuitry which can be used to carry out the operation thereof is shown in FIGS. 42A through 42D which together illustrate the video input processing circuitry.

With respect to the operation of the clamp 209 (see FIG. 42C), the voltage at the output of the amplifier 208 appears on lines 211 and 218, one of the latter of which extends downwardly to the base of an emitter follower transistor 244 that provides a voltage drop.

Under equilibrium conditions, the blanking level of the video signal appearing on line 218 will be at ground potential. This signal is shifted by about 0.7V toward the negative as a result of the voltage drop through the emitter follower 244. A matching emitter follower transistor 245 with its emitter connected to the negative input of a differential amplifier 246 by line 247 shifts the comparison level (ground) toward the negative as does transistor 244.

The emitter of the transistor 244 is connected to the positive input of the differential amplifier 246 when a transmission gate or switch 248 is closed during and for a whole number of cycles of burst by a signal on the line 224 that is produced by redefined gate pulse generator 223 shown in FIG. 42D. Thus, during the burst time, switch 248 is closed charging a capacitor 249 to the average level of the burst. The switch is closed for an integral number of cycles of the subcarrier. This eliminates the need for low pass filtering the video to remove the burst before the clamping is performed. which is ordinarily done in the prior art in order to eliminate H/2 modulation of the clamping level. The charge on the capacitor 249 reflects exactly the average value of the burst and the differential amplifier 246 output represents an error that is applied to the video amplifier 208 through line 251, transistor 252 and line 210 which is connected to the emitter of the transistor 252. The blanking level of the signal on line 211 is thus held very close to ground due to the high DC gain of the differential amplifier 246. The operation of the clamp 202 is substantially similar to the operation of the clamp 209 and is shown in FIGS. 42A and 42B.

Referring again to FIG. 42C, the closing of the switch 248 gates burst through the switch into capacitor 249 and onto line 225 which extends leftwardly to FIG. 42A which is connected to the emitter of a transistor 254 and the burst therefore appears on the collector and on line 255 that extends to the burst limiter circuit 226. When burst is present, the burst presence detector circuit 228 provides a limited burst signal on its output line 229 that clocks the precision gate generator 230. A counter is employed as the precision gate generator and counts cycles of the limited burst signal and produces a precision burst gate during the middle three cycles of the nine to eleven cycle burst interval that is coupled by line 256 to enable the amplifier 227. Therefore, except for the middle three cycles of burst, the amplifier 227 is disabled by the output of the precision gate generator 230. When burst is present, the diode detector 257 and following latch circuit 258 of the detector circuit 228 provides a more negative level on line 260 extending to a switching transistor 259 (FIG.

42B) of the phase detector 231 When burst is present, switching transistor 259 is shut off and another switching transistor 261 of the detector 231 is turned on. When transistor 261 is on, the three cycles of burst from the amplifier 227 is applied by the driver 277 to a transformer 262 of the detector 231. The driver is in turn connected to the phase comparator 231a for comparing the phase of the burst with the phase of the output of the 3.58 MHz (SC) oscillator 232 that is present on line 233. When burst is not detected by the detector circuit 228, transistor 259 is switched on, which applies the signal H/2 to the other input of the driver 277 that is also connected to the transformer 262 and the phase of the oscillator output on line 233 is compared with the phase of the H/2 signal.

Turning now to the detailed circuitry for performing the precision H sync separation and referring to FIG. 42C, the sync is taken from the amplifier 208 on the line 218 extending to a low pass filter 264 whose output is coupled to the base of a transistor 265. The emitter of transistor 265 is connected to a transmission gate or switch 266 that is closed during the presence of sync by control line 224. The level of the sync is determined by charging a following capacitor 267 (FIG. 42D), which is buffered by a unity gain amplifier 268, and half the DC level of the tip of sync together with the full level of AC ripple present in the signal is then applied via line 215 to one input of sync separator 213, the other input is fed by line 269 from the emitter follower transistor 265. In the embodiment of the input circuitry 93A and 93B illustrated in FIGS. 42A-D, the precision H sync separator 213 is a comparator. The output on line 220 is a separated sync whose timing is not affected by AC ripple on the video, because any AC ripple will appear on both inputs of the comparator 213 and will be prevented from appearing in the output of the comparator because of common mode rejection. The sync appearing on line 220 is a precision sync that is used by other parts of the signal system to generate horizontal line related synchronizing signals redefined in relation to a particular phase angle of the subcarrier signal which serve as timing references in the signal system for processing the video signals. Also, the horizontal line related synchronizing signal used in the system is at a rate of 1/2H sync because there are a whole number of subcarrier cycles for every two horizontal lines (227.5 x 2 = 455) and this consideration becomes important in the operation of apparatus described herein as will be evident from the ensuing description.

A less precise separated sync is also developed by taking the sync from the low pass filter 264 via line 270 to the imprecise sync separator 219, the output of which appears on line 271 that is applied to the gate pulse generator 223 which includes a one shot serving as a sync presence detector 276. The upper circuit, indicated generally at 272, generates a gate for use by the switch 266 to close the switch during the presence of sync, a circuit 273 produces a backporch sample and a circuit 274 redefines with respect to SC phase a burst gate signal.

With respect to the generator 223, it should be appreciated that if no sync is present and therefore does not appear on line 271 from the imprecise sync detector 219, the sync presence detector 276 will through circuit 274 close the switch 248 in the clamp circuit 209 as well as a similar switch 275 in the clamp 202 so that all clamps operate on a DC feedback loop rather than permitting them to remain open. Thus, if sync is not present, the level on line 224 is placed high until sync returns and is detected. In addition, as a precautionary measure in the event the precision gate generator 230 does not receive the necessary number of burst cycles to clock it to its terminal state or count after its count cycle has been initiated, the detector 276 is coupled through circuit 274 to provide the burst gate signal to the precision gate generator 230 to assure termination of its count cycle and provision of the precision burst gate signal. This assures that the precision gate generator 230 will always properly respond to every input burst signal.

Because of the desirability of having a field index signal in the encoder switch 126 that is accurately related in phase to the input video signal's vertical sync, the output of the precision H sync separator 213 and an output of a vertical sync detector 278 (FIG. 42B) are provided to a NOR gate 279 (FIG. 42D) which provides the desired field index signal.

The reference logic circuitry 125A and 125B shown in the block diagram of FIG. 9A receive various signals from the input circuitry 93A or 93B relating to horizontal and vertical sync signals, regenerated subcarrier and the like and respectively generate a number of clock and timing control signals used in the operation of the apparatus. In addition, the computer control system 92 provides control signals to both logic circuitry 125A and 125B which cause the generation of servo sync signals which control the operating phase of the disc drives in accordance with the operation, viz, record, playback, transfer and the like, being performed by the apparatus. The reference logic circuitry is essentially duplicated so that one reference logic circuit is provided for use with the video input circuitry 93A and another for the reference input circuitgy 3B, with the function of the reference logic circuitry being somewhat different during different operations of the apparatus such as recording, playback, transfer and the like. Because the logic circuitry 125A and 125B perform different functions, different inputs are received by each and all outputs that are available from each are not used.

The operation of the reference logic circuitry will now be explained in further detail with reference to a functional block diagram shown in FIG. 11A that has a dotted line extending horizontally in approximately the middle of the drawing. As is shown thereon, the upper portion of the circuitry is used only during a recording operation, whereas the lower portion is used during recording. playback and other operations performed by the signal system.

The function of the upper portion of the circuitry is to generate various phase locked clock signals for recording operations using the regenerated subcarrier that was produced by the video input circuitry 93A from the color burst as has been previously described. The circuitry also generates a nonsymmetrical PALE flag signal at a rate of H/2 which is used within the circuitry to alternate the phase of the analog-to-digital converter sampling clock on consecutive horizontal lines for the reasons that have been hereinbefore described. The PALE flag is also available as an output from the reference logic circuitry 125B for use by other parts of the signal system, primarily those used in processing playback signals. The circuitry also generates a drive synchronization signal for operation of the servo control of the disc drive motors, providing a set of three pulses at a rate of 15 Hz which is multiplexed with H sync for use in controlling the disc drive servo. Other timing control signals are provided by the reference logic circuitry 125B as will be described in the following detailed description.

Referring to the upper portion of FIG. 11A, the subcarrier signal (SC) from either the video input circuitry 93A for the reference logic circuitry 125A or reference input circuitry 93B for the reference logic circuitry 125B is applied on line 300 and it is extended to a phase comparator 302, the output of which appears on line 303 to a summer 304 that has a second input on line 305 provided by an integrator 306. A precision digital burst phase decoder 307 receives the actual digitized video data taken from the output of the analog-to-digital converter 95 on line 308 and decodes whether the samples were taken at the proper phase of burst and produces a plus or minus error signal to the integrator 306 via line 309 for use in adjusting the phase of the sample clock so that the video signal is always correctly sampled.

The output of the summer 304 appears on line 310 which is applied to a loop amplifier and filter 311 that is connected to a voltage controlled oscillator 312 by line 313 which also extends to one of two trouble lamp drivers 314. The output of the oscillator 312 appears on line 315 at a frequency of 6SC which is applied to a divide by 6 counter 316 as well as to a divide by 2 counter 317 which produces a PALE clock output at a frequency of 3SC on line 318. The divide by 6 counter has an output on line 319 at a frequency of SC which is applied to a divide by 2 counter 320 as well as to the other input of the phase comparator 302. The output of the divide by 2 counter 320 is a 1/2SC signal on line 321 which also extends to a pulse former 322 that is used to set and reset the divide by 2 counter 317 on alternate lines, the control being supplied through line 323 at an H/2 rate that is supplied by a PALE flag generator 324 as will be discussed hereinafter.

The operation of the upper portion of the circuit is to generate a 6SC frequency signal at the output of the voltage controlled oscillator 312 that is precisely controlled so that sampling that is performed by the analog-to-digital converter 95 is done precisely at the same phase of the color burst synchronizing signal at all times. This is important when it is considered that the phase of the video that is sampled will ultimately determine the color that is produced by the apparatus. Thus, the phase comparator 302, of which one input is fed by the divided output of the VCO 312 through line 319, provides a phase-locked loop that will lock the phase of the output relatively close to the phase of the video or reference subcarrier sync on the line 300 which feeds the other input of the comparator 302. The divided output of the VCO 312 through the phase lock loop produces an SC signal that is generally within approximately 10 . However, the digitized video output from the analog-to-ditigal converter 95 is also applied through line 308 to the precision digital burst phase decoder 307 which is enabled by the precision burst sampling gate signal received from the video input circuitry 93A over line 307a to generate an error signal derived during the burst interval of the video that is integrated by integrator 306 to provide an average value that is applied to the summer 304. This causes the voltage level out of the loop amplifier 311 controlling the VCO 312 to be adjusted to correct variations in the sampling times of the video signal as reflected in the burst samples provided to the decoder 307. The burst samples will represent the same quantity values for all lines if no variation in sampling times occur. By examining the sampled data actually appearing at the output of the analog-to-digital converter 95, it can be precisely determined whether the samples were taken at the proper phase and in this manner, the VCO output on line 315 which is applied to the divide by 2 counter 317 produces a PALE 3SC clock on line 318 which controls the analog-to-digital converter 95 for keeping the sampling at the proper phase. The precision digital burst phase decoder 307 effectively corrects any errors that may be produced due to temperature drifting and the like which can be on the order of 5 to 10 . In this regard, the phase of the synchronising signal on line 300 provides the basic lockup for the VCO 312 and the precision correction that appears on line 305 in the reference logic circuitry 125B is arranged to change the phase by a few degrees, i.e., up to about 20 .

With respect to the lower portion of the block diagram of FIG. 11A, the PALE flag generator 324 produces a PALE flag signal at the H/2 rate for switching a switch 325 which steers 1/2SC pulses into the set or reset terminals of the divide by 2 counter 317 that produces the PALE clock on the output line 318. The PALE flag changes state every line as will be described herein with respect to FIG. 11B. The PALE flag signal is nonsymmetrical so that the phase of the 3SC PALE clock is never reversed during the burst interval of the video signal even through it is reversed during the active video of alternate lines. Thus, the net effect is that only the portion of the line after burst is sampled with a clock signal whose phase is reversed on alternate lines, i.e., a nonsymmetrical signal. As is shown in FIG. 11A, the PALE flag generator 324 has inputs from the video (or reference) input circuitry 93A or 93B of H drive applied on line 326. a field index pulse on line 327 and a burst flag on line 328. The burst flag keeps the PALE flag generator from producing the PALE flag signal on output line 323 until after burst has occurred, since the sampling phase of burst must not be altered for the operation of the burst phase decoder 307 in the upper portion of FIG. 11A.

The PALE flag generator 324 provides an H/2 rate transfer reset pulse which is sent over line 324a to the encoder switch 126 which emplovs it during data transfer operations to generate a signal that is used by the encoder 96 to reset its sync word inserter.

The H drive and field index signals are also applied to a drive servo sync generator 330 which has an output extending to a drive sync switcher 331 through line 332 and it provides the basic drive sync signals on line 334 for each of the disc drives 73 when commanded by the control line 333 from the computer control system 92. The sync signals are required for all operations in which the information is transferred between a disc pack 75 and the signal system. The computer system 92 differentiates whether a record or playback operation is desired. The sync information is in the form of a multiplex sync signal that appears on lines 334 that extend to the disc drive units and includes a set of three consecutive wide pulses to indicate the first field being recorded or played back at a 15 Hz set rate as well as horizontal sync pulses (at H rate) and is used for control of the spindle servo motor. Color frame and related sync signals also are provided for control of the servo drive and for use by the reference clock generator in generating control signals used during playback operations.

The color frame related sync signal is obtained from a color frame generator 301. which receives the 30 Hz field index pulse signal over line 327 and frequency divides it by 2 to obtain the 15 Hz color frame signal. The color frame signal is sent over line 329 to the disc drives 73 and the reference clock generator 98.

The specific circuitry that can be used to carry out the operation of the block diagram shown in FIG. 11A is illustrated in FIGS. 43A through 43D which together comprise an electrical schematic diagram of the reference logic circuitry. Since the operation of the circuitry shown in the detailed schematic diagram is carried out generally in the manner as has been previously described with respect to FIG. 11A. it will not be described in detail herein. However, with reference to the decoder 307, it is convenient to mention that the eight-bit video subcarrier synchronizing signal or color burst (derived from the output of the analog-to-digital converter 95) appears on lines 308 which are connected to arithmetic logic units 335 which in turn connect to shift registers 336. The shift registers 336 are clocked by the logic circuitry, indicated generally at 337. which is activated upon receipt of the precision burst sampling gate over line 307a and together with the arithmetic logic units 335 perform the arithmetic steps that are necessary to determine the sign of the phase of the digitized color burst on line 309. The error of any sampling is determined by examining the quadrature component of the samples which would be zero if the samples are taken at the proper phase of the color burst. More specifically, the quadrature component is proportional to the function X1 - 1/2(X2 + X3) where the samples X1, X2 and X3 are 1200 apart. The clocking logic 337 performs the sequence that enables arithmetic units 335 and shift registers 336 to make the arithmetic computation which will produce either a plus or minus signal on line 309 indicating an error in the phase of the actual samples.

Turning now to FIG. 43A which contains circuitry 324 for generating the PALE flag signal on line 323, the H drive signal is inverted by inverter 342 and is applied via line 338 into the clock input of an FF 339 which is a divide by 2 having output line 340 applied to the input of a second FF 341 that is clocked by the burst gate or flag signal on line 328. Line 340 also extends to a NAND gate 343 as does the output line 344 from the FF 341. The operation of the PALE flag generator 324 will now be described in connection with the timing diagrams shown in FIG. 11B which has the H drive signal (line 326) shown in FIG.

11B(1); the signal on line 340 shown in FIG. 11B(2), the signal on line 344 shown in FIG.

11B(3), the burst gate clock on line 328 shown in FIG. 11B(4) and the output of the NAND gate on line 345 appearing in FIG. 11B(5). The PALE flag signal on line 323 is the inverse of the signal on line 345 by virtue of inverter 346. While the PALE flag signal occurs at a rate of H/2, FIG. 11B(5) shows it to be nonsymmetrical because the output of FF 341 appearing on line 344 and applied to the NAND gate 343 is delayed with respect to the output from the first FF 339 because the FF 341 is clocked by the burst gate rather than by H drive.

The reference clock generator 98 produces the basic timing signals for the apparatus during playback, data transfer, diagnostic and other like operations during which input video signals are not recorded and uses as its input timing reference the regenerated SC (3.58 MHz) that is produced by the input circuitry 93B and passed through the reference logic circuitry 125B. The reference clock generator can shift the entire system's phase and includes a phase locked loop and assorted counters and logic circuits to generate timing signals accordingly. It also generates control signals used by the corrector 100 and the circuit 101. Also, the reference clock generator 98 identifies alternate reproductions of the recorded two field picture frame and issues a frame delay switch signal employed in the blanking insertion and bit muting circuitry 127 to prevent jittering in the display of the output video signal that would otherwise exist because of the use of an H sync related timing control signal synchronized with reference color subcarrier signal to control the processing of the reproduced video information.

* The operation of the generator 98 will be described with reference to FIG. 12A. The circuitry in the upper half of FIG. 12A produces various timing signals including several clock signals and that shown in the lower half uses reference synchronizing information, such as color frame from the reference logic circuitry 125B and field index and horizontal drive signals from the reference input circuitry 93B and generates the control signals used by the time base corrector 565 (FIG. 15A) and the circuitry 101 and 127. More specifically, the SC signal is applied to the generator 98 at input line 340', causing the generator to produce 1/2SC, SC, 3SC and 6SC clock timing signals and various time base corrector pulse timing signals as indicated at the right of FIG. 12A. The reference clock generator 98 includes circuitry that is controllable by an operator through, for example, a thumb wheel switch 349 so that the phase of the output signals can be adjusted relative to the phase of the regenerated SC signal on the input by introducing various amounts of phase shift into the circuit and thereby set the playback system phase. Using the horizontal sync position control included in the reference input circuitry 93B and the SC phase control together enables an operator to determine and control the delay introduced to the playback signal channel over a large range in small increments. To control the phase of SC, the input regenerated SC signal on line 340' is divided by 2 by a divider 343', the output of which appears on line 344' that extends to two locations, one being the programmable counter 345' while the other is another divide by 2 divider 346' which in turn is connected by line 347 to a phase comparator 348. The thumb wheel switch 349 introduces a ten bit BSD number, ranging from 0 to 399, into the programmable counter 345' which has the effect of varying the phase of the subcarrier over a range of 0 to 399" in 1" increments. The output of the programmable counter, which is a periodic signal whose duty cycle may be varied in increments of precisely 1/720 of its basic period by means of the thumb wheel switch 349, extends to a current switch 351a which modulates the current from a current source 351 of one of two matched current sources 351 and 353. This modulated current feeds a low pass filter 354a which develops a DC voltage proportional to the duty cycle of the signal on line 354.

A circuit of identical DC characteristics comprising the other matched current source 353, a current switch 353a and a low pass filter 355a, develops a DC voltage on line 355 which is proportional to the duty cycle of the output of the phase comparator 348. The voltages on lines 354 and 355 are applied to a differential amplifier 356, the output of which is extended by line 357 to the control input of a voltage controlled oscillator 358, which operates at a nominal frequency of 6SC. A number of dividers 360 (divide by 6), 363 (divide by 2) and 365 (divide by 2) sequentially operate on the output of the oscillator 358, producing a signal with a nominal frequency of 1/4SC on line 342' which extends to the second input of phase comparator 348, so that the duty cycle of the signal at the phase comparator output varies with the phase angle between its inputs. Under steady state conditions, the duty cycle of the signal on line 352 is forced to be equal to that of the signal on line 350 within a very small margin of error due to close matching of the current sources 351 and the DC impedance of filters 354a and 354b.

A change in the duty cycle of the signal at the phase comparator 348 output of 1/720 of its basic period requires a change of phase of 0.25 between its inputs, which have a frequency of 1/4SC, and this in turn requires a change of 1" between lines 340' and 361, where the frequency is SC. Thus, changing the value by one on the thumb wheel switch 349 causes a 1" change in phase of the SC signal on line 361. The total range of phase comparator 348 (180 at 1/4SC) corresponds to 720" at 1SC. For convenience, the thumb wheel switch is limited to 399 , which still insures adequate overrange capability with respect to the necessary 360".

The phase controlled oscillator 358 provides the phase continuous 6SC clock timing signal over its output line 341' and, through the cooperative operation of the chain of dividers 359, 360 and 363, causes phase continuous 3SC, SC and 1/2SC clock timing signals to be provided at the outputs as designated in FIG. 12A. The dividers also furnish 3SC and SC clock signals to the logic circuitrv 362 that produces phase continuous SC rate read/write (R/WR) mode, write enable (WR EN), demultiplex (DMPLX) clock and multiplex (MPLX) clock signals used by the time base corrector circuitry 565. The details of the logic circuitry are shown in FIGS. 44C and 44D and relationships between the signals provided by the logic circuitry can be found by reference to FIG. 12C. The schematic diagram illustrated by FIGS. 44A through 44D together with the timing diagram of FIG. 12B disclose the operation of one embodiment of logic circuitry 362 for providing phase continuous time base corrector clock signals with the desired timing relationships.

With respect to the lower portion of the circuitry shown in the block diagram of FIG.

12A, the circuitry redefines an H sync related, namely, 11/2, signal so that it is synchronous with the phase continuous 3SC signal that is produced by the upper portion of the circuitry and occurs at the first reference horizontal line following alternate reference vertical sync.

As will be apparent later from a description of the circuit 367 hereinbelow, maintaining H/2 synchronized with respect to the reference subcarrier and such that it occurs at the first line of the first field of every two reference field sequence (which corresponds to the placement of the sync word in the video signal),requires frame rate phase inversion of the subcarrier rate clock controlling the reclocking circuit 367 to redefine H/2 with respect to the phase of SC. Subsequent reclocking of the redefined H/2 signal with the phase continuous 3SC clock signal within the circuit 367 introduces a 46 ns (1/2 cycle of 3SC) picture to picture motion of redefined H/2 relative to reference H sync. Use of the redefined H/2 in the time base corrector circuitry 565 to correct a repetitively reproduced video signal transfers the 46 nsec picture frame to picture frame motion to the video signal output by the time base corrector.

This motion occurs because the reclocked and redefined H/2 is mispositioned relative to the reference H sync position on alternate picture frames and causes the time base corrector circuitry 565 to misposition the sync word a corresponding amount, or 1/2 cycle of 3SC,on alternate reproductions of a picture frame. As will be explained hereinbelow upon consideration of the sync word insertion circuitry portion of the encoder 96. the H/2 rate sync word is inserted on the video signal on alternate picture frames at a position that is displaced 1/2 cycle of SC from that corresponding to the reference H sync. This is because the sync word inserter is reset every frame and because the sync word is positioned on the first line of every picture frame, it being understood that the first line of successive picture frames have oppositely phased SC. The time base corrector circuitry 565 inherently removes all of this displacement except for the aforementioned 1/2 cycle of 3SC. A frame delay detector 368 of the reference clock generator 98 generates a frame delay switch signal used by the blanking insertion and bit muting circuitry 127 to correct for such motion. Also, it is not desirable to have the H/2 positive going transition of the unredefined 11/2 signal coinciding exactly with a subcarrier transition in the reclocking circuit 367 because an ambigously timed redefined H/2 pulse signal will be produced for use by the time base correctors 565 and errors in time base correction will result.

To produce an H/2 signal redefined with respect to the phase of the phase adjusted. phase continuous regenerated subcarrier signal, SC provided by divider 360 is coupled to one input of a phase inverter 393 formed by an exclusive OR gate circuit. The other input of the phase inverter is coupled through a NAND gate circuit 397 to receive the 15 Hz color frame pulse signal generated by the reference logic 125B and present on input line 396a. The level this signal at the input of the phase inverter 393 determines the phase of SC at the output of the inverter, a high level resulting in the inversion and a low level not. Inversion of the phase of SC is necessary because an H/2 signal is required that is phase coherent with H sync. (In the recorded video signal, a sync word is inserted in the same lines for all picture frames of the video signal, which in this apparatus is the odd numbered lines of the 525 lines forming an NTSC picture frame.) Without inversion of the phase of SC, the phase of the redefined H/2 signal would change at a 15 Hz rate with respect to H sync by one half of an SC cycle. Such an H/2 signal would not be suitable as a reference for use in processing reproduced video signals during playback operations. The SC signal output provided by the phase inverter 393 is fed to the reclocking circuitry 367 and is used together with the reference H drive signal on line 396 and the field index signal on line 395 (both signals being generated by the circuit 93B, FIG. 9A) to generate the H/2 signal. The reclocking circuitry 367 includes logic circuitry to ensure that an unambiguously timed H/2 signal is produced and defined with respect to the phase of SC.

The output of the reclocking circuitry 367 is then applied to the frame delay detector 368 which produces the frame delay switch signal on line 369 that identifies the first or second playing of a reproduced still, composed of two television fields or a picture frame, so that the clock generating circuitry for the circuitry 127 will know whether to insert an additional 1/2 period of 3SC offset for correcting the previously mentioned 46 ns picture frame to picture frame motion of H/2.

The redefined 11/2 pulse signal generated by the reclocking circuitry 367 appears on line 386 that is gated through gate circuitry 370 and 371 to appear on line 372 for use as the reference time base corrector circuitry during playback operations, signified by an enabling signal provided on line 373 by the encoder switch 126 (FIG. 9A) from control signals issued by the computer control system 92. During playback operations, a high level signal appears on line 373 and the playback H/2 on line 386 will satisfy AND gate circuitry 370 and will appear on line 372.

In other operations, such as E to E and transfer, involving the processing of video signals in a playback channel, the H/2 signal as ordinarily generated by the H/2 vs SC definition circuitry 367 is not used. In E to E operations, continuous time base correction is not necessary since the video signal does not experience a record and reproduce process.

Hence, the EE or PB command provided by the encoder switch 126 from control signals issued by the computer control system 92 is coupled over line 398 to the reference clock generator 98 associated with the playback channel selected for use to disable the phase alteration of SC. The phase alteration is disabled through the operation of the NAND gate circuit 397 placing a low level signal on the second input of the phase inverter 393.

Furthermore, the EE or PB command is coupled to logic circuitry 399 that responsively generates an EE TBC disable signal used to allow the time base corrector circuitry 565 to operate for approximately ten lines at the beginning of each color frame and, thereby, generate the proper timing correction for each color frame or every 15 Hz. This timing correction is required because during the sync word insertion process for E to E operations the sync word generator is reset ever two fields, i.e., every picture frame. This results in a discontinuity of one half SC cycle in the position of the sync word every other frame or every 15 Hz.

When the apparatus is performing a transfer operation through a playback channel, a low level signal is placed on line 373 of the reference clock generator 98 associated with that playback channel. This enables the AND gate circuit 374 to pass a transfer H/2 signal present on line 375 to the OR gate circuit 371 which gates the transfer H/2 to the output on line 372. The transfer H/2 is derived from the sync word inserter portion of the encoder 96 circuitry. An output pulse from the encoder 96 that is coincident with the sync word or line identification is produced and that pulse is used as the time base corrector reference. The pulse appears on line 376 and passes through a shift register delay circuit 377 which correctly positions the pulse that is present on line 376. The transfer H/2 signal is positioned so that the digitized video signal provided to the encoder 96 during a transfer operation has a correctly identified location for insertion of a new sync word.

Specific circuitry that can be used to carry out the operation of the block diagram shown in FIG. 12A is shown in FIGS. 44A through 44D. The operation of the specific schematic circuitry will not be described in detail since it carries out the operation as has been previously described with respect to FIG. 12A. However, with respect to the generation of the H/2 signal so that it is unambiguously redefined with respect to SC, the reclocking circuitry 367 includes an H/2 signal generator 378 comprising a divide by 2 counter and following pulse former respectively formed by an edge triggered flip-flop and following self resetting flip-flop. The counter receives at its clock input H drive signals present on input line 396 and provides an H/2 signal at its output. The H/2 signal is formed into a train of negative pulses, each at a positive going transition, by the H/2 generator's pulse former.

The 30 Hz field index signal resets the counter portion of the generator 378 at beginning of the first field of every picture frame so that the phase of the H/2 signal is the same at the time of the first line of the first field of every picture frame.

The SC signal provided by the phase inverter 393 is also formed into a train of negative pulses by a pulse former 393 a. A pulse coincidence detector circuit 378a formed by a low level AND gate and following D latch examines for a coincidence of the SC transition related pulses received from the pulse former 393a and the H/2 transition related pulses provided by a timing selection circuit 379 in response to each negative pulse provided by the pulse former portion of the generator 378. If the positive transition of the H/2 signal provided by the generator 378 becomes too close in time to the positive transition of the SC signal, the transition related pulses will overlap in time at the coincidence detector circuit 378a resulting in the toggling of the latch of the detector circuit. Toggling of the latch changes the level at an input of an exclusive OR gate 379a included in the timing selection circuit 379 to change it between its inverting and non-inverting mode. The timing selection circuit 379 includes a self resetting, edge triggered flip-flop 379b having its clock input coupled to the output of the exclusive OR gate 379a. By selectively inverting and not inverting the negative pulses provided by the H/2 signal generator 378, the positive edge of the pulse output of the exclusive OR gate is moved relative to SC. The timing selection circuit 379 cooperates with the coincidence detector circuit 378a to position the positive edge of the pulse output of the exclusive OR gate 379a so that unambiguous redefinition of H/2 will always result.

Redefinition of H/2 is performed by the reclocking edge triggered flip-flop 367a having its reset input coupled to an output of the timing selection circuit 379 and its clock input coupled to receive the SC signal provided by the phase inverter 393. Each H/2 transition related pulse resets the flip-flop 367a and the immediately following positive transition of the SC signal received at the clock input changes its state to thereby generate the redefined H/2 transition. A following latch 367b couples the redefined H/2 transition signal to a delay means 391 composed of a counter and following shift register operated to provide a properly timed H/2 signal on the line 380 extending to the frame delay detector circuit 368.

The redefined H/2 transition signal output by the latch 367b is coupled to reset the delay means 391 and an SC signal, opposite in phase to that utilized in the reclocking circuitry 367, provided over line 392 clocks the delay means to effect issuance of the redefined H/2 transition signal to the detector 368.

With reference to the frame delay switch signal that appears on line 369 in FIG. 44D, it is a signal that changes level on alternate picture frames and is used in the blanking and bit muting circuitry 127 for adjusting the half cycle of 3SC mispositioning as previously discussed. The operation of this portion of the circuitry will now be discussed in connection with FIG. 12C. The signal appearing on line 380 is an H/2 rate pulse signal which has been unambiguously redefined with respect to the phase of the regenerated SC. which is inverted on alternate frames so as to ensure that the SC redefined H/2 transition signal is stationary with respect to H sync reference. This transition signal is clocked into a shift register 381 by a phase continuous 3SC signal appearing on line 394 and appears on the first output line 385 delayed and synchronized to the 3SC signal. Because the continuous phase 3SC clock is an odd multiple of one half the picture frame frequency, its phase during a first picture frame is 1800 different with respect to H sync reference than its phase at the same time during the next picture frame and. hence, is also 1800 different frame to frame with respect to the redefined H/2 pulse. Because of this 1800 phase relationship difference, the positive transition of the 3SC clock shifts one half cycle picture frame to picture frame relative to the redefined H/2 pulse and consequently the clocking of the shift register 381 relative to the occurrence of the stationary H/2 pulse will change frame to frame by one half of the 3SC clock period. To detect the relationship between the redefined H/2 signal and the phase continuous 3SC clock signal, a stationary pulse is generated from the positive transition of the redefined H/2 signal and is used by the frame delay detector latch or D type flip-flop 368a to determine the phase of the 3SC clock at the beginning of alternate picture frames and provide the phase indicative frame delay switch on line 369 as shown in FIG. 12C. More specifically, the pulse forming circuitry comprised of inverter 382. resistor 388. capacitor 387 and NAND gate 389 generates a stationary pulse from the leading edge of the H/2 pulse signal present on line 380 at the input of the shift reg the signals are in phase, the H12 pulse signal appears on line 385 one cycle of 3SC after its presence on the input line 380. When the signals are out of phase, the H/2 pulse signal appears on line 385 1/2 cycle of 3SC earlier. The signal level on line 385 is strobed into the D flip-flop 368a by the positive going transition of the stationary pulse on line 384, which occurs 3/4 of a cycle of 3SC after the occurrence of the redefined H/2 pulse signal at the input of the shift register. The output of latch 368a on line 369 indicates whether the 11/2 pulse was present on line 385 after a delay of 3/4 period thereby determining if the time delay between the positive going signals on lines 394 and 385 is 1/2 period or 1 period of 3SC. This signal on line 369 is fed to the circuitry 127 to insert a 1/2 3SC period offset selectively in the clocking of the video data, whereby to compensate for the 46 ns picture frame to picture frame motion of the redefined H/2.

With reference to the frame phase inverter switch signal that appears on line 356a in FIG.

44D, it is a signal that changes levels on alternate picture frames and is used in the chroma separator and processing circuitry 101 to effect inversion of the chrominance component included in the reproduced video signal on alternative reproductions of the two field color video signal. The playback burst is provided on input lines 361a by the data transfer circuitry 129 and is phase compared with the phase continuous SC by the exclusive OR gate 362a. SC and playback burst alternate between in phase and out of phase conditions with alternate reproductions of the two field color video signal, causing the level of the output of the exclusive OR gate 362a to change at a 15 Hz rate with the change occurring at the time of playback burst. The frame phase inverter switch signal is obtained by clocking the output of exclusive OR gate 362a through a latch 363a with one properly timed clock at every burst flag. The latch 364a receives at its D input the burst flag signal provided on line 360a by the reference input circuitry 93B and is clocked by the phase continuous SC provided at its clock input by the divider 360. Each time a burst flag signal is present on input line 360a, the latch 364a issues a pulse to the latch 363a defined with respect to the phase of SC. This pulse is used to clock the level at the input of the latch 363a to its output. Because the level at the input of the latch 363a changes with alternate reproductions of the two field color video signal, the level at the output of the latch 363a also changes with alternate reproductions to produce the 15 Hz frame phase inverter switch signal on line 356a that defines when the chrominance should be inverted or not in the chroma separator and processing circuitry 101.

The encoder switch 126 described with respect to the block diagram of FIG. 9A is interconnected with the computer control system 92 and, upon receiving the appropriate commands, performs the principal function of selecting either the video data streams from the analog-to-digital converter 95 when in the record operating mode or the data streams that originate at the data transfer circuitry 129 when a transfer operating mode occurs. In the transfer mode, the recorded picture frame is transferred from one disc drive to another so that the video information does not go through the chroma separator and processing circuitry 101. Instead, it is directed to the encoder switch 126 to be thereafter encoded and recorded on another one of the disc drives. The encoder switch 126 also switches between the appropriate clock signals, i.e., 6SC and 1/2SC. It switches to clock signals generated by the reference logic circuitry 125A which are used when the video information from the analog-to-digital converter 95 is being recorded. During the transfer mode, it switches to the 6SC and 1/7SC signals provided by the reference clock generator 98 and are used as the basic reference clock signals during the recording ot the transferred video signal, all of which is generally shown in the block diagram of FIG. 9A.

The encoder switch also performs functions in addition to switching the proper reference signals. depending upon whether the regular record or transfer modes are being performed.

Circuitry is included for generating a blinking cross picture display signal, one diagonal line of which is supplied by one field and the other by the second field which provides an indication that the track has been deleted and is available to receive a still in that particular location. The encoder switch also includes circuitry that generates a PALE switch signal which terminates PALEing during the transfer process, the PALE switch (or flag) signal extending to the data transfer circuitry 129 which normally PALEs the data going to the chroma circuitry 101. The PALEing by the transfer circuitry is stopped because there is no need of aligning the samples line to line during a transfer mode of operation. The encoder switch also includes circuitry for performing diagnostic testing, which circuitry selectively generates a recurring sequence of digital information, as well as a random word for use in such testing.

More specifically, and referring to FIGS. 13A through 13D which together form a composite electrical schematic diagram of the encode switch 126, the bits of data appear on sets of input lines 400 or 401, there being eight lines for each of the sets corresponding to the eight bits in the data stream from either the analog-to-digital converter 95, or from the data transfer circuitry 129. The lines 400 comprise the eight data lines from the converter 95, while the input lines 401 together represent the eight bits of video information from the data transfer circuitry 129. The input lines are connected to a number of multiplexer switchers 402 which are commanded by signals on lines 403 and pass either the information from the lines 400 and 401 onto output line 404. The multiplexer switchers 402 also pass, on command, the bits forming the blinking cross signal or bits forming the data used for diagnostic testing. To delete information from a track on a disc pack, the blinking cross signal is recorded in the track over the information being deleted. Thus, either the blinking cross signal, diagnostic testing data or the transfer or original digitized video information is provided at the output of the encoder circuitry 96.

As is best shown in FIG. 13B, the 6SC signal provided by the reference logic circuitry 125A is available on line 405 and a similar reference related 6SC signal is applied to line 406 from the reference clock generator 94. Similarly, 1/2SC signals from reference logic circuitry 125A appears on line 407 while reference related 1/2SC signals from the reference clock generator 94 appears on line 408. A number of AND gates 409 are provided to selectively gate through either the 6SC and 1/2SC signals from either reference logic circuitry 125A or the clock generator 94 to the output lines 410 and 411 which respectively provide the 1/2SC and 6SC signals that are used by the encoder 96. The logic level of a reference select signal provided by the computer control system 92 on input line 445 and set in the latch 446 by a strobe control signal also provided by the computer control system determine which of the 6SC and 1/2SC signals are provided to the encoder 96. During the record mode of operation, a reference select signal is provided that enables the AND gates 409 associated with the 6SC and 1/2SC signals received on lines 405 and 407 from the reference logic circuitry 125. In the other operating modes, i.e., delete data, diagnostic test and data transfer, the provided reference select signal enables the AND gates 409 associated with the 6SC and 1/2SC signals received on lines 406 and 408 from the reference clock generator 94.

Referring to FIG. 13C, reset pulses from the video input circuitry 93A and the reference logic circuitry 125B are applied to lines 412 and 413, respectively. and one of the lines is gated through AND gate 414 to provide a reset pulse on line 415. the line being gated through determined by the reference select signal latched into latch 446. The reset pulse is used for resetting the sync word generating circuitry in the encoder 96. The input frame signal provided by the video input circuitry 93A and the transfer ID reset signal provided by the reference logic circuitry 125B are used as the reset pulses. Similarly, strobe or V drive, No. 2 and No. 1 pulses generated by sync generating circuit of the video and reference input circuitry 93A and 93B, respectively, are applied to input lines 416 and 417, respectively, one of which is selectively gated through AND gate 418 onto line 419 by the latched reference select signal for use by the sync word generating circuitry in the encoder 96. The gated strobe or V drive pulses are also applied to line 435 for controlling the generation of the deleted track blinking cross signal by the circuitry 420 shown within the dotted line enclosure on FIGS. 13C and 13D. which will be described below.

A number of jumpers, indicated generally in FIG. 13D at 421. are adapted to be connected to an external input, a repeating data word generator 427 or a random word generator 429 controlled by the sync word gate signal provided over line 429a by the sync word generating circuitry included in the encoder 96.

Data select signals on input lines 422 from the computer control system 92 form a two digit command that conditions the command lines 403 for setting the multiplexer switches in the correct state for the operating mode of the apparatus. A strobe control signal on line 448, which also originates from the computer control system, strobes a pair of latches 449 to place the two digit command on the command lines 403 that extend to the multiplexer switches 402. The latched command is also detected by a NAND gate 423 which provides a signal on line 424 identifying whether the apparatus is operating in a transfer mode or normal recording mode. In a transfer mode, a NAND gate 425 (FIG. 13D) provides the PALE switch signal to shut off the PALEing in the transfer circuitry 129, the other input of NAND gate 425 being supplied by a PALE flag control signal on line 426 extending from reference logic circuitry 125B.

The circuitry 420 for generating the deleted track blinking cross signal will now be described with reference to the block diagram of FIG. 13E and the front view of two fields of a television picture in FIG. 13F which exemplifies a visual display of the deleted data signal. A horizontal (H) counter 430 counts down from a preset count in response to an 80H clock which is obtained, for example, from the sync generating circuit of the reference input circuitry 93B, and which inputs on line 431. The 80H clock is at a frequency of 80 times the reference H sync rate. H counter 430 is preset every horizontal line by an H drive signal which is received from the reference input circuitry 93B over line 428 (FIG. 13D) and is input to the counter on line 432. A vertical (V) counter 433 counts up and down in response to the H drive signal steered onto one of the input lines 434 to the counter 433 by steering gate circuitry 434a (FIG. 13C). The V counter 433 is preset after every other field by a preset command placed on line 447 by a divide-by-two flip-flop 436. The preset command is generated from the V drive signals provided by the reference input circuitry 93B over line 417 and placed on the line 435 extending to the input of the flip-flop 436 by one of the AND gates 418 that is enabled by the latch 446 as described above (FIGS. 13C and 13D).

For purposes of forming a preferred geometrical blinking cross shape, a divide-by-four (+ 4) means 438 is coupled to the input of the V counter 433 so that four H drive signals must be received to change the count state of the V counter. The divide-by-four means 438 is conveniently formed by shifting the output connections of the V counter 433 by two bit positions with the V counter 433 connected in this manner, the state of its output will change upon receipt of every fourth H drive signal whether counting up or down. digital comparator 437 is coupled to the H counter 430, and also to the V counter 433 positions so that four H drive signals must be received to change the count state of the V counter whether counting up or down. A line width generator 439 is coupled to the comparator, and the output therefrom comprises the digital signal introduced to the multiplexer switches 402 of previous mention.

In operation, referring also to FIG. 13F, the television picture is divided into an X and Y matrix. For example, the horizontal direction is divided into 80 counts, and the vertical direction is divided into a number of counts corresponding to the lines contained in a field.

When the corresponding H and V counters increment equal counts, the comparator 437 detects a point of coincidence and generates a "1" output pulse corresponding to a white level. When no coincidence is detected, the comparator generates a "0", or black level.

Both counters 430, 433 are preset via their respective preset inputs, H drive and V drive, the latter being divided down by two so that the V counter 433 is preset every other field. In the first television line of the first field, the H counter 430 is clocked from zero through 80 by the 80H clock. The preset divide-by-four means 438 and V counter 433 are incremented by the first clock pulse of H drive indicating line 1, and as the H counter clocks the first clock pulse, coincidence between the outputs of the counters occurs. This defines point 1-1, which corresponds to the upper, left corner of the raster or horizontal lines forming the displayed field.

The diagonal line width generator 439 includes a counter 443 which counts to a pre-determined number to cause a logical "1" to be placed on the data bit lines, which corresponds to white level. This causes a short line segment from the initial point of coincidence, as depicted at 440 of FIG. 13F. Since the V counter input is divided by four, after the counter 433 is clocked out of its preset state by the first clock or H drive pulse, it takes four H drive pulses to increment the V counter 433 again. Thus, four television lines are scanned on the same output pulse of the V counter 433, whereby lines one through four are in coincidence with the first clock pulse of the 80H clock as it counts the duration of the horizontal line. Thus, four short line segments 440 are generated side-by-side on four adjacent television lines. When the fifth H drive pulse is received, the V drive counter 433 is incremented one count and, for the next four H clock pulses, causes short line segments for another four horizontal lines to be generated, but one count later in the H counter 430 count cycle.

This leads to the generation of the diagonally extending series of diamond shapes which define a diagonal line 441. After every fourth horizontal line occurs, H drive increments the output of V counter one count (to count 2 and so on) for the interval of the next four television lines. The 80H clock increments the H counter 430, whereby coincidence occurs at point 2 during lines 5-8, at point 3 during lines 9-12, etc. The process continues with the point of coincidence moving along the diagonal until the diagonal line 441 ends at the lower right corner of field 1 with television lines 261-262. At this point, no outputs are generated since this is the time period corresponding to the vertical interval.

On the second field, data corresponding to the oppositely directed diagonal line 442 is inserted into the signal. The diagonal lines 441, 442 are purposely drawn on alternative fields to provide a readily visible flicker or blink of 30 Hertz upon playback and display of the recorded blinking cross signal.

To this end, the V counter 433 is maintained at its count at end of the first field.

However, the steering gate circuitry 434a causes the H drive signal related pulses to be steered onto the other of the input lines 434 to the divide-by-four means 438 and V counter 433 as a result of clocking the flip-flop 436 by another V drive related signal placed on line 435 at the end of the first field (FIGS. 13C and 13D). The V counter 433 now counts back down from its coincidence point count. For the second field, this corresponds to the upper right corner of the field (which corresponds to its last point at the lower right corner of field 1), as it would be displayed on a monitor. Since the field raster scan always begins at the top of the displayed raster of horizontal lines and scans the lines sequentially to the bottom, the first full horizontal line scanned will be across the top of the displayed raster of lines forming field 2 as shown in FIG. 13F. The H counter 430 operation is unaffected by the V drive signal and continues counting the received 80H signal.

The divide-by-four divider 438 and counter 433 are not incremented until receipt of the H drive or clock pulse at the beginning of the fourth full line, i.e., line 267 in the second field.

During this line, the H counter 430 counts until it reaches a count of 79, at which time a coincidence with the V counter 433 occurs. Thereupon, the comparator 437 generates logical "1" bits to be inserted in all the data bit lines via multiplexer switches 402 (white level) whose duration or line segment length is determined as before by the line width generator 439. The diamond shapes are drawn on the picture along the opposite diagonal, forming the diagonal line 442 from right to left since the V counter counts down whereby each subsequent coincidence is earlier than the previous one. Since the V counter 433 is not incremented until the fourth full line, the diagonal line 442 is actually shifted slightly to the left of the true diagonal. However, this slight shift is insignificant to the purpose served by the blinking cross and is not noticeable to the observer except upon very close inspection of its display.

The +4 means 438 and the line width generator 439 are employed herein to properly determine the angle of the diagonal and to form the diamond shapes which define a thicker.

more symmetrical line. However, the above conditions were imposed by the limited input signals available; viz, the 80H signal. If a signal of, for example, 262H were available, the matrix would constitute a 262 by 262 grid and the H and V counters 430, 433 could be arranged to increment together while drawing the diagonal lines 441, 442 from corner to corner, without need for mathematical compensation such as the + 4 means 438.

The two fields are herein illustrated one below the other to facilitate understanding the manner in which the diagonal line 442 of the field 2 is formed to cause a display to be formed from top-to-bottom and right-to-left. The fields are, in fact, actually interlaced on the same television display device, and thus the diagonals are superimposed from field to field to define the blinking cross.

The output from the line width generator 439 is introduced to the multiplexer switchers 402 of FIGS. 13A-B, which generate a digital deleted data word, which, in the apparatus described herein, is a timed series of "1" bits corresponding to the white level. An output line 444 of the line width generator 439 is selected for parallel passage through the switchers 402 on output lines 404 thereof by the 2-digit command introduced thereto via the latch command lines 403, as previously described. The delete word on lines 404 provides an input to the video recording system and is processed for recording as is the video signal received from the analog-to-digital converter 95 over lines 400 or 401. The blinking cross signal is recorded on a track when a previous record is deleted and provides visual means to indicate a track is available to receive a video signal for storage. If a request to play back data from a thusly identified track is initiated, the delete signal, and thus the blinking cross. is interrogated and reproduced in the same manner as when reproducing a field or frame of stored video.

The various components 430-439 and 444 of the block diagram, FIG. 13E. are shown in schematic diagram in FIGS. 13A-D, wherein like components are similarly numbered.

Thus the H and V counters 430, 433 comprise a pair of four bit counters, wherein the output connections of the V counter 433 are shifted two bit positions to provide the divide-by-four function depicted at block diagram 438 of FIG. 1A. The digital comparator 437 provides a carryout pulse upon detecting a point of coincidence of the counters 430, 433. while the line width generator 439 provides the white level output via line 444 thereof, for a selected time period corresponding to the preset number of counts made by the counter 443 thereof representative of the desired length of the short line segments 440 (FIG. 13F) forming the diamond shapes of the diagonal lines 441, 442. The (eight bit) digital delete word is formed herein via the (eight) output lines 404 of the multiplexer switchers 402, as previously mentioned.

Although the circuitry 420 is described with reference to the generation of a blinking cross geometry, it is understood that other geometrical arrangements, pictures, etc.. may be used as the deleted data signal, wherein a portion of the display may be generated in one field, with the remaining portion being generated in the alternate field, to cause a readily visible flicker at the 30 Hertz rate. The entire display may be generated in one field if the flickering effect is not desired. However, encoding the deleted data signal to cause a visible flicker upon its display facilitates generation of a distinctive deleted track signal since displays of visual information typically do not flicker. For the video frame storage apparatus, the flicker effect is convenientlv obtained by separating parts of the deleted data signal in each of the two television fields recorded in the track of deleted data. It should be appreciated that other techniques can be employed to reduce the repetition rate of the deleted data signal and produce the flicker effect, such as by encoding one of the two television fields with the complete cross while leaving the other field devoid of information.

In addition, a display other than visual may be generated. For example, in a video storage system where audio signals may be recorded in video tracks, the playback deleted data signal would be "displayed" as an audible tone. In a computer oriented, data storage system employing a file of deleted data regions, the computer could be arranged to electronically interrogate and detect the deleted data signals recorded in the deleted tracks, with an electronically detectable display provided if desired. In the system illustrated herein, the display is visual, and the deleted data word is interrogated in playback.

The encoder 96 shown in the block diagram of FIG. 9A of the video signal system contains circuitry which performs functions in addition to channel encoding the digitized data on each of the eight video data bit lines, the parity bit and the data track sequence as described hereinbelow. One of the additional functions involves the use of a parity generator to perform a parity check to verify that the data is correct on all of the eight data itlines. The parity bit Is optional and requires an extra data bit line such as is available in the apparatus described herein. The encoder 96 also generates and inserts the sync word (also referred to herein as the line identification or line ID). The sync word is in the form of a 7-digit binary number which is placed in alternate television lines, generally where the horizontal sync pulse had been previously located, it being understood that the horizontal sync had been stripped from the composite video signal by the video input circuitry 93. The sync word is inserted within one cycle of SC of the location previously occupied by the horizontal sync pulse, and the encoder 96 inserts the sync word into each of the eight video data lines, the parity bit line and the data track line before the channel encoding is performed so that the output of the encoder 96 which is connected to the electronics data interface 89 contains the sync word in each of the 10 data streams recorded on a disc pack 75 (or sent to the playback channel 91 during E to E operations).

The encoder 96 operation will now be described in conjunction with a block diagram shown in FIG. 14 and schematic circuit diagrams of FIGS. 45A-D. NRZ-L data from the encoder switch 126 enters on input line 450 and exits on output line 451 of each data bit line after having been (i) checked for parity, (ii) had the sync word inserted in alternate (odd) lines and (iii) channel encoded in a format that is conductive to magnetic recording and reproducing the digitized information with respect to one of the disc packs 75. The input data on each data bit line is applied to one input of a data input AND gate 452 which is connected to a channel encoder 453, which may be switched between two channel encoding formats, both of which will be described hereinafter. In the schematic circuit diagram of FIGS. 45 A-D, identical channel encoders for two video data bit lines are shown in their entirety. Identical channel encoders for the other video, parity and data track data lines are contained in dotted line enclosures below the encoders shown in their entirety. A sync word input AND gate 454 in each of the 10 bit lines is used to gate the sync word into the encoder at the proper time. These AND gates are also arranged to insert a test signal in the 10 bit lines i desired, the test signal being provided on line 450a (FIGS. 45A and 45B) by a suitable test signal source, such as digital test pattern generator. A first clock generator 455 has input signals 6SC and 1/2SC applied thereto by the encoder switch 126 and provides various SC and 3SC outputs as shown. Two of the 3SC outputs are applied by lines 472 and 473 to a second clock generator 456 which provides two time displaced 3SC clock signals on the two lines 474 and 475 that are extended to the channel encoder 453 for clocking the same. The clock signal on line 475 is a 1 clock and is displaced one-half cycle of 3SC from the clock signal on line 474, which is a 2 clock. During recording operations, these time displaced clocks are derived from the continuous phase 6SC, 1/2SC signals generated by the reference logic circuitry 125A and provided to the encoder 96 by the encoder switch 126.

During other operations, such as recording the blinking cross delete signal, the reference clock generator 98 provides the clock signals. The 1 and 2 3SC clock signals are used to drive the channel encoder 453 so that a continuous channel encoded digital signal without phase discontinuities is provided at the output on line 451.

The clock generator 455 has an SC clock output 471a driving a + 455 divider 457 which can also be reset by a reset pulse provided by the encoder switch 126 on line 463 at a 30Hz rate. The divider 457 sets a flip-flop (FF) 458 through the start line 464 and subsequently resets the FF 458 when a pulse appears on the stop line 465 extending to the reset pin. The START and STOP pulses define a window during which a single 7-digit binary sync word provided at the output of a sync word generator 459 can be inserted in all data bit lines simultaneously.

During the vertical blanking period. a pulse is applied to a monostable multivibrator 460.

The multivibrator is active for a period of about 10 lines of the vertical blanking period by switch vertical signal provided on line 466 by the encoder switch 126 and its output is applied to one side of gate 461 (shown in this block to be a NAND gate), the other side of which is supplied by the output of the window generating FF 458. The output of the NAND gate 461 extends to the other input of the AND gate 454 as well as through an inverter 462 to one side of the AND gate 452.

During the operation of the encoder circuitry 96, it is desired that the data stream for each bit be applied on an input such as input 450 which is representative of the eight separate data input lines, each of which is connected to a separate encoder 453 and the associated data and sync word input AND gates 452, 454 and inverter 462 so that a data output line 451 exists for each of the data bits and each of the data streams is properly channel encoded and has a sync word inserted therein. Since it is desired that the sync word occur close to the former location of the horizontal sync pulse and since it is also desired that it not be confused with data of the data stream, the data bit lines input to the channel encoders 453 are disabled by the data input gates 452 when the sync word is inserted during a sync word gate window that is generated by the divider 457 and FF 458. More specifically, the divider 457 provides a START pulse for setting the FF 458 and this enables one input of each AND gate 454 while simultaneously disabling each AND gate 452 thereby blocking the data entering on lines 450. The divider 457 issues a pulse to the sync word generator 459 over line 467 twelve data bit intervals after the generation of the START pulse and the sync word generator 459 then generates the 7-digit binary word which is applied to the upper input of all AND gates 454 which have previously been enabled. The AND gates 454 pass the sync word into each channel encoder 453 where it is encoded onto the data stream.

After the sync word has been generated, the divider 457 issues a STOP pulse 29 data bits later which resets the FF 458, disabling all AND gates 454 and simultaneously enabling all AND gates 452 so that the data on lines 450 will be passed into the channel encoders. It should be understood that the data stream line 450 is continuous in its flow and that disabling the AND gates 452 merely blocks it from passing. Hence, the information is onls discarded in a sense during the insertion of the sync word. However, since the sync word is inserted approximately at the previous location of the horizontal sync pulse, no active video informational data is lost.

During the vertical blanking interval. the multi-vibrator 460 provides an output to the NAND gate 461 which occurs for an interval of about 10 lines. This disables the data input AND gate 452 during the 10-line interval of the blanking period so that the received data is prevented from passing to the channel encoder during this interval. Thus. the only data or logical 1 bits that appear on the output data line 451 during the 10 line interval of the vertical blanking period are those in the sync words that appear every other line, as previously described, and pass through the sync word gate 454. This insures that the decoder and time base corrector circuitry 100 will be locked on the actual sync word during playback rather than some randomly occurring sync word bit pattern that might be contained in the active video information during the flow of the data stream.

Another aspect of the operation of the encoder 96 will be more clearly understood bv referring to FIGS. 9A and 9B. The electronics data interface 89, disc drive data interface 151 and data select switch 128 couple the encoder 96, disc drive 73 and decoder and time base corrector circuitry 100. It should be appreciated that during a seek operation when the heads in the disc drive 73 are moving between tracks, it is desirable to prevent the introduction of perturbances in the signal system. Ordinarily, the record signal processing system 88 will provide at the output of its encoder 96 digitized data even in the absence of an input video signal. While this signal will represent noise information, the digital signal processing electronics of the apparatus cannot distinguish between digitized noise and digitized video information. This factor is taken advantage of when the apparatus is performing a seek operation. During seek operation, the transducing heads create noise signals that do not conform to the channel encoded formate of the digital data ordinarily present in the signal system. Such noise signals if permitted to enter the playback channel 91, undesirably perturb the phase lock loops of the decoder and time base corrector circuitry 100. To avoid such perturbances, the disc drive data interface 151 is switched (as in an E-to-E operation) to reroute the output provided by the encoder 96 to the decoder and time base corrector circuitry 100. In this manner. the decoder and time base corrector circuitry 100 receives channel encoded digital signals that maintain the respective phase lock loops in the circuitry 100 within their normal operation range. Hence, when the heads of the disc drive 73 are properly positioned and playback data provided to playback channel 91, the decoder and time base corrector circuitry 100 are prepared to immediately provide the output decoded and time base corrected signals.

In addition, the encoder 96 also serves to cause black level data to be generated for use in recording on the disc surfaces as previously described during the first two revolutions of the disc pack 75 prior to the recording of the video signal information on the subsequent two revolutions of the disc pack. Accordingly, the prerecord line 470 (FIG. 45A) extending from the electronics data interface 89 is activated as a result of signals provided by the disc drive data interface 151 and causes NAND gate 461 to block any logical "l's'' as may be present on the input lines 450 thereby producing the black level at the input of the channel encoder circuitry 453. It should be noted, however, that the encoder 96 still functions to insert the sync word in the black level signal.

Each of the ten parallel data bit lines recorded on a disc pack 75 is encoded selectively by a channel encoder 453 into one of two DC free self clocking channel codes. As will be described further hereinbelow, the two position code selection switch 480 selects between the two channel codes. In both codes, the NRZ-L data bit stream on a data bit line is broken into discrete bit times commonly designated as data bit cell times.

When the switch 480 is in a first position (POS.1) the code (known as Miller code) followed result in logical first bits, e.g., logical 1's to be represented by signal transitions at a particular location in the respective bit cells, specifically at mid-cell, and logical second bits or logical 0's to be represented by signal transitions at a particular earlier location in the respective cells, specifically at the beginning or leading edge of each bit cell. Any transition occurring at the beginning of one bit interval following an interval containing a transition at its center is suppressed.

When the selection switch is in a second position (POS.2) the input data stream in each data bit line may be viewed as the concatenation of variable length sequences of three types: (a) sequences of the form 1111---111, any number of logical 1's but no logical 0's; (b) sequences of the form 0111---1110, any odd number of consecutive 1's or no 1's, with 0's in the first and last positions; (c) sequences of the form 0111---111, any even number of consecutive 1's preceded by a 0. A sequence is of type (c) only if the first bit of the next following sequence is a zero. Sequences of types (a) and (b) are encoded according to the Miller code. A sequence of type (c) is similarly encoded for all bits except the last logical 1, and for this 1 the transition is simply suppressed. By this means, the type (c) sequence, viewed in isolation, is made to appear the same as a type (b) sequence, that is, the final logical 1 looks like a logical 0.

By definition, the type (c) sequence is followed immediately by a logical 0 at the beginning of the next sequence. No transition is allowed to separate the type (c) sequence from the following 0. Therefore the special coding is distinctive for decoding purposes. The decoder must merely recognize that when a normally encoded logical 1 is followed by two bit intervals with no transitions, then a logical 1 and logical 0 should be output successively during those intervals. Other transition sequences are decoded as for the Miller code.

The encoding procedure for this code (known as Miller squared code) requires that a modulo-2 count be maintained of the number of logical 1's output by the encoder since the last previous 0 which was not the final bit of a type (b) sequence. If the count is 1 (odd number of 1's) and the next two bits to be encoded are 1 and 0 in that order, then no transitions are output during the next two bit intervals. If the next subsequent bit is another 0, then it is separated from its predecessor by a transition in the usual manner of the Miller code. The Miller squared code provides for the transmission of data in binary form over an information channel such as a magnetic record/playback system, incapable of transmitting DC, the information being transmitted in self-clocking fashion.

With respect to the channel code it makes no difference which binary state is considered logical 1 and which binary state is considered logical 0. In the foregoing and following descriptions the state normally marked by mid-cell transitions is considered the 1 state, whereas the state normally indicated by cell edge transitions is considered the 0 state.

The channel encoders 453 illustrated by the FIG. 45A through 45D operate in accordance with the aforedescribed code rules. FIG. 45E is a timing diagram depicting the operation of the channel encoder 453 included in one of the data bit lines 450, the switch 480 shown in FIG. 45B being in the indicated position.

The channel encoder will now be described with the code selection switch 480 set as shown in FIG. 45B to effect channel encoding. A description of the difference in the operation of the encoder when the switch 480 is set in its other position to effect channel encoding will follow.

As described above, data encoded when switch 480 is in its second position requires examining two successive data bits to be encoded whenever the modulo-2 count of logical l's previously encoded is odd. For this purpose, each channel encoder 453 includes a pair of serially connected input latches 481 and 482 clocked by the trailing positive edge of the 2 3SC clock signal (FIG. 45E - (2)) on line 474a, which is coupled to line 474 by an inverter 483. The input latches provide a two bit cell delay from the input of latch 481 to the output of latch 482. At each trailing positive edge of the 2 clock, latch 481 is operated to latch the present data level of the bit stream at its input so that it appears at its output (FIG. 45E (3)) and latch 482 is operated to latch the preceding data level of the bit stream contained in latch 481 so that it appears at its output (FIG. 45E - (2), (3) and (4)). Therefore, the outputs of the latches 481 and 482 contain the data bits of two consecutive bit cells that are to be encoded.

The outputs of the latches extend to the inputs of three NAND gates 486, 487 and 488 for separately gating through pulses corresponding to logical 1's and 0's in the data bit stream.

NAND gate 486 receives three inputs; one from the output of latch 481, one from the output of latch 482 and 01 clock pulses (FIG. 45E - (1)) placed on line 475 by an inverter 484 connected to the output line 475a of the clock generator 456. This NAND gate is enabled to provide an output pulse 489 (FIG. 45E - (6)) upon receipt of a 1 clock whenever its other two inputs are at a low level, which occurs only when successively received data bits are logical 0's. Consequently, NAND gate 486 issues logical 0 related pulses that are marked by transitions in the channel encoded format of the data stream output by the channel encoder 453. A logical 0 bit that immediately follows a logical 1 bit is blocked from passage by the NAND gate because the latch 482 will be high when, for example, the 1 clock pulse 490 (FIG. 45E - (1)) occurs. Hence, the channel encoder 453 follows the code rules as if the switch 480 was in its first position for successively occurring logical 0 data bits.

On the other hand, the NAND gate 487 has two inputs and is enabled to provide an output pulse (FIG. 45E - (5)) upon receipt of a 1 clock for all logical 0 data bits. Because the output of latch 482 enables the NAND gate 487, the logical 0 related pulses are provided one data cell time after the data has been latched into the channel encoder 453.

NAND gate 488 has three inputs and is enabled by the inverter output of the latch 482 to provide an output pulse. (FIG. 45E - (7)) upon receipt of a 2 clock for all logical 1 data bits, unless a high level bit suppression command 491 (FIG. 45E - (10)) is placed on the input of the NAND gate by a line 492 extending from a bit suppression NAND gate 493 as will be described hereinbelow. NAND gate 488 generates the logical 1 related pulses during the interval of the 2 clock, hence, before the latch 482 is clocked by the trailing positive edge of the 2 clock. The logical 1 related pulses are provided by the NAND gate 487 one data cell time after the data has been latched into the channel encoder 453 at latch 481.

An OR gate 494 has two inputs connected to receive the logical 0 pulses 489 (FIG. 45E (6)) provided by NAND gate 486 and the logical 1 pulses 515 (FIG. 45E - (7)) provided by the NAND gate 488. The output of the OR gate 494, which appears on the encoder output line 451, will, therefore. be a train of pulses (FIG. 45E - (14)) that occur according to the code rules for the channel encoder. Hence, the NAND gates 486 and 488 together with the OR gate 494 serve to encode the incoming NRZ-L data stored by the latches 481 - 482 into the selected channel code format. The NAND gate 487 operates with bit suppression logic circuitry 500 described below to control the selective suppression of logical 1 data bit related transition in the channel encoded data. By disabling the bit suppression logic circuitry 500, as would occur bv changing the position of the switch 480 from that shown in FIG. 45C, the NAND gates 486 and 488 will encode the data according to the 3,108,261 Patent rules.

To encode the data bit stream according to the Miller squared code rules, the bit suppression logic circuitry 500 includes two modulo-2 counters 495 and 496 for counting encoded logical l's and 0's and, together with cooperating gate circuitry, effecting the generation of the bit suppression command on line 492 that suppresses selective logical 1 bit related transitions in the channel encoded data appearing on line 451. The modulo-2 counter 495 counts the logical 0 related pulses coupled to its clock input by the NAND gate 487. Logical 1 related pulses provided by NAND gate 488 are coupled to the clock input for counting by the modulo-2 counter 496. Counter 495 recognizes the beginning of each sequence by toggling in response to logical 0 pulses each time a logical 0 is encoded and being cleared each time a logical 1 related transition is suppressed. As can be seen from the aforedescribed code rules, counter 495 toggles twice during a type (b) sequence and never changes state during a type (a) sequence, and therefore is in its cleared state before the start of any sequence. The bit suppression logic circuitry 500 must recognize the end of a type (c) sequence. Modulo-2 counter 496 is employed in the performance of this function by toggling in response to logical 1 pulses each time a logical 1 is encoded and being cleared in response to logical 0 pulses each time a logical 0 is encoded. Waveforms (8) and (9) of FIG.

45E illustrate the respective operations of the modulo-2 counters 495 and 496 if their outputs are not connected together at the wired-OR 501. Waveform (13) of FIG. 45E illustrates the actual state at the wire-ORed connection 501. As should be appreciated from the foregoing, if counter 496 is not in its cleared state, the counter 495 is in its cleared state, the present bit to be encoded is a logical 1 and the next following bit is a logical 0, the bit suppression command is provided by NAND gate 493 on line 492 to disable the NAND gate 488 and thereby suppress the encoding of the present logical 1 bit.

Considering the cooperating gate circuitry for controlling the clearing of the two modulo-2 counters 495 and 496, counter 496 has its set terminal coupled to the NAND gate 487 so that its output is set high each time a logical 0 related pulse is output by the NAND gate 487. The counter 495 has its set terminal coupled to the output of a NAND gate 497 so that its output is set high each time a logical 1 related transition is suppressed in the channel encoding of the data bit stream. For reasons that will become apparent from the following description, a pair of capacitors 498 and 499 are connected in the output circuits of the modulo-2 counter 495 and NAND gate 493, respectively, to delay the set logic level of counter 495 appearing at the wired-OR 501 and removal of the bit suppression command from NAND gate 488.

The bit suppression command is generated by the NAND gate 493 that examines the first of consecutive data bits to be encoded and which is present in inverted form at the output of the latch 482, the next following of the consecutive data bits to be encoded and which is present at the output of the latch 481 and the counter states of the modulo-2 counters 495 and 496. If either one of the counter outputs at the wire-OR 501 is high, the NAND gate is disabled. However, whenever the beginning of a type (c) sequence occurs, both counters 495 and 496 will be low, thereby placing an enabling signal at the input of the NAND gate 493. If the next two bits to be encoded are a logical 1 followed by a logical 0, the bit suppression command 491 will be generated and placed on line 492 upon the occurrence of the 2 clock pulse 502 (FIG. 45E - (2)) immediately preceding the 1 clock pulse 490 that would effect the generation of the logical 1 related pulse through NAND gate 493. Hence, when the 1, clock pulse 490 (FIG. 45E (2)) occurs on line 474 that would cause the NAND gate 488 to generate a logical 1 bit pulse, the NAND gate 488 is disabled by the bit suppression command on line 492 and the logical 1 bit pulse is suppressed as represented by the pulses 512 shown in phantom at line (14) of FIG. 45E. The bit suppression command is terminated upon setting the counter 495. The set pulse 505 (FIG. 45E - (12)) is provided by the NAND gate 497 in response to the bit suppression command 491 (FIG. 45E - (10)) on line 510 and the aforementioned 1 clock pulse 490, which occurs 1/2 cycle of 3SC after the 2 clock pulse or about 47 nanoseconds. To insure that the counter 495 is not set and the bit suppression command not removed until after the prl clock pulse 490 has ended, the delay capacitors 498 and 499 are provided to delay the return of the counter 495 to its high set state, hence, disabling of the NAND gate 493 and to delay the return of NAND gate 493 to its low disabled state, hence, extending the duration of the bit suppression command 491.

The delay's effect is exhibited at the rounded parts 508 and 509 of the waveforms (10) and (13) of FIG. 45E.

To disable the bit suppression logic circuitry 500, switch 480 is placed in the position that places a high level signal (ground in the channel encoder 453 of this apparatus) on the set line 510 for the counter 495. This places the counter permanently in its set state, thereby placing a disabling high level signal permanently at the wire-OR input of the NAND gate 493. Hence, bit suppression commands 491 can not be generated and bits will not be suppressed.

Commonly, self clocking channel encoded data code formats carry data and clock information as particularly placed transitions between two signal levels. When such encoded data is sent through a transmission channel, it usually experiences some timing distortion because of the non-linear characteristics of most transmission channels. If the timing distortion is significant, errors may result because of the inability of the channel decoder to determine the correct location of the transmitted transitions. Furthermore, at high data rates, such as found in the apparatus described herein, the timing distortion may result in unacceptable errors in the transmitted data. This is particularly the case where, as in the case of the channel codes selected for use in the apparatus herein, oppositely directed transitions carry the data and timing information. Non-linear transmission channels will alter the positively and negatively going transitions in a non-linear manner with respect to time. Hence, level sensitive data detectors commonly used at the terminal of a transmission channel to restore the transmitted data so that it has properly positioned transitions will position the positive and negative transitions differently. Different positioning occurs because a positive transition with substantial timing distortion will reach the level selected for sensing the presence of transitions at a time after its nominal position that is different from that required by a similarly distorted negative transition.

To enhance the reliability of transmission of channel encoded data in which oppositely directed transitions carry the data and clock information, each of the channel encoders 453 encodes the data bit stream at its input by providing pulses in accordance with the rules of the selected channel code at the transition locations of the channel encoded format. In the particular channel encoder used in the apparatus described herein, logical 1 data bit pulses 515 (FIG. 45E-(7) and (14)) are provided at the data cell boundaries to define logical 1 bit related transitions that appear in the channel encoded data and logical 0 data bit pulses 489 (FIG. 45E-(6) and (14)) are provided at center of a data cell to define logical 0 bit related transitions that appear in the channel encoded data. The transition-related pulses are generated by the clock generator 456 to have a precisely defined edge, the leading edge being selected. The second clock generator 456 includes two one-shot multivibrators that are clocked by the oppositely phased 3SC clock signals provided by the first clock generator 455 over lines 472 and 473. Since the leading edges of the positive pulses generated by each of the one shot multivibrators are defined by rapidly switching the multivibrators from its stable state to its quasi-stable state (there being no significant time constant determining components involved), each leading edge will be identical to all others and occur at a precise time following the occurrence of the positive clocking transition of the clocking signal. The two multivibrators of the second clock generator 456 provide 1 and 02 clock pulse trains, which in the embodiment described herein have a pulse width of about 17 nsec, with the leading edges of the pulses of each train precisely defined with respect to each other and those of the other train. As described hereinbefore, the 1 clock pulses provided on line 475 are gated through the NAND gate 488 as logical 1 data bit transition related pulses that appear in the channel encoded data and the 2 clock pulses provided on line 474 are gated through NAND gate 486 as logical 0 data bit transition related pulses that appear in the channel encoded data. Since the NAND gates 488 and 486 are in an enabled condition at the times the 1 and 2 are received for transmission as transition related pulses (FIG. 45E-(4), (7) and (14) for logical 1 bit pulses and FIG. 45E-(3), (4), (5), (6) and (14) for logical 0 bit pulses), their respective leading edges will not be noticeably affected by the transmission through the NAND gates. Because the transmission channel over which the pulses are sent will act on identical pulse edges the same, the precise locations of the transition-related positive pulse edges, hence, data signal transitions themselves, are not lost as a result of any distortion that may be introduced to the pulses by the action of the transmission channel.

The channel encoded transition related pulses output by the encoder 96 over lines 451 are coupled by the electronics data interface 89 to the transmission line 152 extending to the disc drive data interfaces 151 associated with the disc drives 73. The electronics data interface 89 includes conventional logic converters which convert the TTL logic on lines 451 to emitter coupled logic levels which provide complementary level pulses on two lines in a manner that is described hereinafter with reference to FIGS. 53A and 53B. The interface 151 or the disc drive selected for recording the video data passes the data to the selected drive's record amplifier and head switch circuitry (FIGs. 54A and 54B). A divide by two JK flip flop 1070 included in each data bit line receives the transition related pulses and is responsive to the leading edges of the transmitted pulses to be rapidly switched between its two stable conduction states. This converts the transmitted pulse form of the channel encoded data to the level transition form for recording as transitions between two signal states. Prior to being converted by the JK flip-flop 1070, the transmitted pulses in each data bit line are passed through a differential amplifier line receiver 20201 included in the disc drive data interface (FIG. 60A) of the kind described hereinafter with respect to the decoder portion 525 (FIG. 46A) included in the data decoder and time base corrector circuitry 100 to regenerate the transmitted pulses with precisely defined leading edges after passage through the associated transmission line of the transmission line bus 152 (FIG. 9B).

The 10 data bit streams of channel encoded data, comprising 8 video data bit streams, 1 parity bit stream (if a parity bit is added) and 1 data track bit stream, transmitted by a disc drive 73 (FIG. 9B) over a transmission line bus 154 are received by one or more of the playback channels 91 (FIG. 4) selected by the data select switch 128. At the input of each playback channel, each of the 10 transmitted data bit streams is received by a separate data decoder and time base corrector included in the circuitry 100 for decoding the channel encoded data back to the NRZ-L form of digital code and then time base correcting the NRZ-L data to remove any intra channel and inter channel bit time displacement errors that may be present in the received data streams. Bit time displacement errors result from the data transmission channel acting on the transmitted data to introduce intersymbol interference and reflections caused by impedance discontinuities in the transmission channel. This disturbs the timing of the data transmitted in the channel. In a video recorded data transmission channel, bit time displacement errors commonly are a result of changes in record medium dimensions, usually caused by environmental changes, of differences in the relative head to medium record and reproduce velocities of the relatively transported head and record medium and of machine to machine mechanical variations resulting in geometric differences between the heads and record medium. Video disc recorders utilizing rigid record media, such as the disc packs 75 used in the apparatus described herein, ordinarily do not cause large time displacement errors in the transmitted apparatus, particularly, at the data rates common for analog type video disc recorders that are in wide use today. The rigid record media used in such recorders are dimensionally stable and the servo mechanisms used are able to maintain the relative transport of the heads and rigid record media within sufficient tolerances so that time displacement errors are kept small. In some applications of video disc recorders, the time displacement errors are so small as to be insignificant and time base correction is not necessary.

However. as described herein, the present apparatus in which the time base corrector circuitry is used employs (with little modification) highly reliable disc drives that have been specifically designed and manufactured for computer data processing. Unfortunately, the computer disc drives do not maintain the relative head to disc velocity stable enough to avoid the introduction of intolerable bit time displacement errors into the data bit streams when such disc drives are used in the present apparatus to process video data. This is because the disc pack spindle in the drive is not servoed but instead is driven by a common three phase AC motor referenced to a relatively unstable line voltage and the rotational position of its disc pack is not controllable with respect to an external reference. The resulting positional errors and bit time displacement errors are particularly detrimental at the high data bit rates, i.e., 10.7 MHz, required to faithfully process broadcast quality video data without reduction in the quality of the video information. Thereofre, to take advantage of the mechanical reliability of the existing computer disc drive design, the apparatus described herein is provided with a positional servo for the AC motor and time base corrector circuitry to remove any unacceptable time displacement errors introduced into the data bit streams rather than altering the reliable design of the computer disc drives.

As described above, before the received data bit streams are time base corrected, each channel encoded data bit stream is decoded back to its original NRZ-L digital form. For this purpose, and with reference FIGs. 46A and 46B, the data decoder and time base corrector circuitry 100 includes for each data bit line a channel decoder circuitry portion 525 having a pair of input terminals 526 coupled to the data select switch 128 (FIGs. 9A and 9B) for receiving channel encoded data, which as described hereinbefore, is in the form of channel encoded transition related pulses, such as pulses 515 and 489 shown in FIG. 45E-(14). The pair of input terminals 526 are coupled to a differential amplifier line receiver circuit 527 connected to reject common mode noise in the pair of complementary transition related pulses received from the transmission line pair included in the transmission line bus 154 after passage through the data select switch 128 (FIG. 9B) In addition, the differential amplifier line receiver circuit 527 regenerates a single transition related pulse from each transmitted pair of complementary transition related pulses so that the regenerated pulse has a well defined leading edge properly positioned according to the code rules of the channel code selected for originally encoding the video NRZ-L data. More specifically, the differential amplifier line receiver circuitry 527 provides a single regenerated transition pulse with leading and trailing edges provided when the levels of the edges of the received complementary pulses are the same. By examining the edges of the transmitted complementary pulses in this manner, the leading edges of all regenerated pulses will be properly positioned according to the channel encoding rules because the same sense, i.e., leading positive going and leading negative going, edges of each pair of the complementary pulses are employed to define the occurrence of the leading edge of each regenerated transition related pulse. Because the transmission channel through which the transition related pulses are sent to the decoder circuitry 525 effect identical pulse edges the same, any time distortion introduced to the pulse edges will not affect the regneration of the transition related pulses.

Following the regeneration of the transition related pulses, they are coupled over line 528 to clock a one shot multivibrator 529 at each occurrence of a regenerated pulse, using the defined leading edge to effect clocking. The one shot 529 is rapidly switched from its stable conduction state to its quasi-stable conduction state to provide the precisely defined leading edge of the transition related pulses. The one shot 529 has one of its outputs connected to line 530a that extends to the clock input of a divide by two flip flop 531. Upon the occurrence of each regenerated transition related pulse, the flip flop 531 is rapidly switched between its two stable conduction states by the leading edges of the regenerated pulses and thereby converts the pulse form of the channel encoded data to the level form for subsequent decoding of the data back to its original NRZ-L digital form as will be described hereinbelow.

The one shot 529 provides complementary outputs of the channel encoded data on line 530a and 530b. The complementary outputs are coupled to a 6 SC clock generator 532 which provides complementary 6 SC clock signals on its output lines 533 and 534 for use by the data decoder circuitry 525 for decoding the received data. The clock generator includes a 6 SC voltage controlled oscillator 537 which is locked by an operatively associated phase detector 535 to the phase of the data clock carried by the channel encoded data. The complementary transition related data pulses output by the one shot 529 on lines 530a and 530b are coupled to the input of the phase detector 535, which has its output on line 536 coupled to the control input of the 6 SC voltage controlled oscillator 537. The phase detector 535 examines the phase of the 6 SC clock provided by the oscillator 537 with respect to the received and regenerated transition related data pulses and provides an error correction signal to the oscillator via the phase error smoothing capacitor 538. A change in the phase of the received data causes the phase detector 535 to change the average voltage level on the capacitor 538 by a corresponding amount and thereby cause the phase of the 6 SC clock provided by the voltage controlled oscillator 537 to be adjusted to clock carried in the channel encoded data.

The phase detection operation is performed by a pair of matched current sources 540 and 541, each having an output line 542 and 543 respectively connected to the line 536 coupled to the error averaging capacitor 538. In the absence of a transition related data pulse, the line 530b extending from the one shot 529 is high, which enables the current source 541.

Because the base electrodes of each transistor of the differential pair forming a current switch 545 at the output of the current source 541 are grounded, the current provided by the current source 541 divides equally in the two current paths defined by the current switch 545. Current in the path defined by the current switch 545 connected to the output line 543 flows onto line 536 to charge the error smoothing capacitor 538 to a level which, when a data stream is not input to the decoder circuitry 525, will cause the voltage controlled oscillator 537 to provide a 6SC clock at a nominal frequency and phase. Thus, even in the absence of a data bit stream at the input of the decoder circuitry 525, a 6SC clock is provided at its nominal frequency. This facilitates rapid synchronization of the oscillator 537 to data clock when a data bit stream is initially received and proper decoding of the channel encoded data.

When a transition related data pulse is received on the input line 526, the one shot responsively provides a high level signal on line 530a and a low level signal on line 530b, for an interval determined by its time constant circuit 529a, which in the decoder circuit described herein is about 17 nsec. The low level signal on line 530b disables the current source 541, thereby terminating the provision of charging current through the current switch 545 to the error smoothing capacitor 538. However, the high level signal on line 530a enables the other current source 540, which provides charging current to the error detection capacitor 538 in accordance with the relative conduction periods of the halves 544a and 544b of a current switch 544 formed by the transistors arranged in circuit as a differential pair. The transistors forming the two halves 544a and 544b of the current switch have their respective base electrodes coupled to receive the 6SC clock provided over line 533. When the clock is at a low level, transistor 544a is disabled. However, the other transistor 544b is allowed to conduct because the long time constant RC circuit 547 holds the voltage at its base electrode at an average voltage level which is more positive than the low level of the 6SC clock. Consequently, all of the current furnished by the current source 540 will flow through the one enabled transistor 544b to the output line 542 of the current source 540.

When the 6SC clock goes high the base of the transistor 544a goes more positive than the base of the transistor 544b. Therefore, transistor 544a is enabled and transistor 544b disabled. This removes the current flow to the error smoothing capacitor 538. If the transition related data pulse received by the current source 540 is positioned in time relative to the 6SC clock provided to the current switch 544 so that low to high level transitions in the 6SC clock occur at the center of the transition related data pulses, each transistor 544a and 544b of the current switch will be enabled for equal intervals and the voltage on the error detection capacitor 538 will be maintained at an average level corresponding to a correctly phased 6SC clock. Any change in the data bit rate of the received channel encoded data bit stream changes the position of the transition related pulses at the input to the current source 540 relative to the low to high level transitions of the 6SC clock at the input to the current switch 544. If this occurs one of the transistors of the current switch 544 will be enabled during the period that the current source 540 is enabled (by the transition related pulse) for a longer interval than the other transistor, with one of the transistors enabled for a longer interval depending upon whether the data bit rate increased or decreased. This causes a corresponding change in the current provided to the error smoothing capacitor 538 and a corresponding corrective change in the average voltage level on the capacitor. A change in the voltage level on the capacitor causes the voltage controlled oscillator 537 to change its phase and frequency until the transition related pulses are centered with respect to the low-to-high level change in the 6SC clock provided to the current source 540. With the low to high level change in the 6SC clock adjusted to be centered with respect to the duration of the transmission related pulses, the two halves, 544a and 544b, of the current switch will individually pass current from the current source 540 for equal intervals. Hence, the average voltage on the capacitor 538 will be maintained at the level required to lock the frequency and phase of the 6SC oscillator 537 to the data clock rate of the received channel encoded data.

If the 6SC voltage controlled oscillator 537 fails to lock to the received data or data is not received by one of the decoder and time base correctors 100 included in one of the 10 bit lines of a playback channel, a frequency unlock signal is provided on an output line 550 that extends to the reference clock generator circuitry 98. All of the lines 550 from the 10 decoder and time base correctors of the playback channel are ORed in the reference clock generator circuitry 98 for coupling a frequency unlock command to the computer control system 92 via the signal system interface 119 (FIGS. 8, 32A and 32B) in the event that one or more frequency unlock signals are generated in a playback channel. The computer control system 92 responds to the frequency unlock command by providing via the signal system interface a video mute command to the blanking insertion and bit muting circuitry (FIGS. 51A and 51B) that blocks the sending of data to the requesting station. In the channel decoder 525, the frequency unlock signal is generated by detecting the failure of the channel decoder to provide a data bit for 16 cycles of 6SC. The frequency unlock signal is provided by a divide by two circuit 546 that has its clock input coupled to receive a clock pulse provided on line 548 each time the channel decoder 525 fails to detect a data bit for an interval of four cycles of 3SC, hence, 8 cycles of 6SC. If a second clock pulse appears on line 548 before the divide by two circuit 546 is reset by the NAND gate 549, the divide by two circuit 546 provides the frequency unlock signal on line 550. The NAND gate 549 resets the divide by two circuit 546 each time a coincidence occurs between a low level of the 6SC clock provided by the oscillator 537 and a low level on line 530b, which occurs when a transition related data pulse is received at the input 526 of the channel decoder.

After the divide by two flip flop 531 converts the channel encoded data from the transition related pulse form to the channel encoded NRZ-L form, the data is coupled by line 531a to a pair of latches 551 and 552 at the input of the decoding circuitry 525a. The decoding circuitry is able to decode data that is channel encoded according to both the aforementioned code rules. FIGS. 46E (1) and (2) illustrate the resulting encoded data for the first and second positions respectively of the switch 480. The latches are clocked by 1 and 2 3SC clocks, respectively, derived from the 6SC clock generated by the oscillator 537.

The 6SC clock on line 534 is coupled to one input of each of the NAND gates 553a and 553b. The other input of each of the NAND gates receives complementary 3SC square waves generated by the divide by two flip flop 534a from the 6SC clock on line 534. The NAND gates are enabled when their inputs are low to issue the positive 1 (FIG. 46E - (4)) clock pulses to clock the latch 552 and positive 2 (FIG. 46E - (3)) clock pulses to clock the latch 551. The 1 and 2 clock pulses are displaced in time by one half cycle of 3SC. Hence, the time that the level of the channel encoded NRZ-L data on line 531a is latched by latch 551 is displaced one half cycle of 3SC from the time the level is latched by latch 552 (FIG.

46E - (5) and (6)). Both latches are coupled to the two inputs of an exclusive OR gate 554a.

The exclusive OR gate serves to detect the occurrence of a change in state in the level of the channel encoded NRZ-L data at the input of latches 551 and 552 between the times they are clocked by the displaced 1 and 2 clocks (FIG. 46E - (7)). To determine if the change in state at the input of latches represented a logical 1 bit, the output of the exclusive OR gate 554a is coupled to one input of a NAND gate 555. The other input of the NAND gate receives inverted 1 3SC clock pulses coupled from the NAND gate 553a by the inverter 555a. If the change in state at the input of the latches represents a logical 1 bit, the output of the exclusive OR gate 554a will be low at the occurrence of an inverted 1 3SC clock pulse.

The NAND gate 555 will be enabled, placing a high level on its output. To assure safe latching of the detected logical 1 bit pulse at the output of the NAND gate 555, a delay circuit 556 is connected to the input of the NAND gate 555 receiving the inverted 1 clock so that the output of the NAND will be maintained high for an interval longer than the 1 3SC clock pulse (FIG. 46E - (8)). This permits the following latch 557 to be clocked with the positive trailing edge of the l 3SC clock to latch the delayed high level provided by the NAND gate 555 (FIG. 46E - (9)). If the input data is channel encoded according to the code described in U.S. Patent No. 3108261 the output of latch 557 will be the channel decoded NRZ-L data. This is represented by the doted lines in the timing diagram shown by FIG.

46E. In the decoder shown by FIGS. 46A and 46B, however, an additional latch 558 is needed to permit decoding. However, for the Miller code the additional latch 558 only delays the output of the decoded data by one cycle of 3SC.

When data is encoded according to the Miller squared code, specified logical 1 bit related transitions are suppressed. If a logical 1 bit related transition has been suppressed, there will be an absence of data transitions for an interval greater than 1-1/2 cycles of 3SC. This is detected by a modulo-4 counter 559 having its clock input coupled to receive 0 clock pulses provided by the NAND gate 553b and its reset input to the output of the edge detecting exclusive OR gate 554a. The exclusive OR gate 554a provides a reset pulse to clear the counter 559 each time a transition occurs in the channel encoded data FIG. 46E - (10)).

The output of the modulo-4 counter 559 is coupled to one input of a AND gate 560 which also receives 0 clock pulses at its other input. Both inputs are low 1/2 cycle of 3SC after the modulo-4 counter has counted four prl 3SC clock pulses without being reset, which corresponds to an absence of data transitions for an interval of 2-1/2 cycles of 3SC (FIG.

46E - (11), (12) and (13)). Ordinarily, this signifies that a logical 1 bit has bit suppressed in the channel encoded data. To make certain that no errors have been introduced to the data stream, a following NAND gate 561 examines an output of the latch 558 at the time when AND gate 560 provides the low state signal representing a suppressed logical 1 bit. If the examined output of the latch 558 is also low, it verifies that a logical 1 bit has been suppressed and a pulse is fed out on line 562 (FIG. 46E - (14)) by NAND gate 561 that is wire ORed with the output of latch 557. Line (14) of FIG. 46E represents the state of NAND gate 561 as if it was not wire ORed with the output of latch 557. The second pulse 563 (FIG. 46E (14)) provided by the gate 561 occurs at the time of and is latched into the latch 558 by the 1 3SC clock. This prevents the output of the latch 558 from being returned low, thereby, inserting the suppressed logical 1 bit into the decoded NRZ-L data (FIG. 46E - (15)) appearing on line 566. In the data track bit line, the decoded data is coupled by line 566 to the data track interface 120 (FIG. 8). The decoded data clock provided by the flip flop 534a on line 574 and the line 1D or sync word from the first shift register and sync word detector circuitry 572 are also coupled to the data track interface.

If the phase of the 3SC decode clock provided by the flip-flop 534a is incorrect, a one-shot multivibrator 534b is enabled by the coincidence of the 6SC clock on line 534 and a pulse provided on line 564. This pulse will be generated 3 cycles of 3SC before the line ID is first detected by sync word detector portion of the circuitry 572 if the level of the decoded data at that time is low, therefore, incorrect. A counter 590 (FIGS. 15A and 46C) receives 3SC decoded data clock and, as will be described hereinbelow, provides an advanced end of count pulse at H/2 rate on line 591. Because of the known data bit pattern of the sync word interval, which interval ordinarily occurs when the advanced end of count pulse is generated, the decoded data level can be examined at the shift register portion of the circuitry 572 to determine if decoding is performed correctly. The gating circuitry 592 issues a pulse on line 564 when the examined decoded data level is low that enables the one-shot 534b to provide a disabling signal at the clock input of the flip-flop 534a for one cycle of 6SC. This results in a shift in the phases of the prl and 2 clocks by 1/2 cycle of 3SC, thereby establishing the right phase for correct decoding of the channel encoded NRZ-L data.

During playback operations, each bit stream of channel decoded NRZ-L data provided at the output line 566 of the decoder circuitry 525 will contain time base errors in the form of bit time displacement errors as previously described. Furthermore, bit line to bit line or skew time displacement errors will be present in the 9 data bit streams that carry the 8 parallel bits of digitized video and 1 parity bit, if included. To remove these errors from the NRZ-L data, a time base corrector 565 is provided in each data bit stream and corrects such errors by electronically adjusting a variable delay through which the NRZ-L data is passed.

Each corrector contains circuitry which processes the received data so that the data bit rates in all video data and parity bit lines are frequency and phase coherent with respect to the reference 3SC provided by the reference clock generator 98 for the playback channel 91.

Furthermore, each corrector 565 also aligns the data bits in the data bit lines with respect to a common redefined 11/2 reference provided by the playback channel's reference clock generator 98. As a result of these combined functions, any relative time displacement errors between the data bits in the 9 bit lines are removed, i.e., line to line or skew errors removed and any bit time displacement errors within a bit line corrected. However. as described hereinbefore, the redefined H/2 signal, while being synchronized to a particular phase of SC and thereby facilitating processing of the reproduced video data. is not stationary with respect to reference H sync. For this reason, use of the 11/2 signal by the time base corrector 565 results in a mispositioning of the sync word in the video data that is output by the time base corrector for alternate reproductions of the picture frame of video data.

The operation of the time base corrector 565 included in each data bit line will be described in connection with the block diagram shown in FIG. 15A and the timing diagrams of FIGS. 15B and C. Specific circuitry which can be used to carry out the operation of the time base corrector is shown in FIGS. 46B, 46C and 46D. The decoded data in each data bit line received from the decoder 525 over line 566 is time base corrected independently of the other 8 data bit lines by using a periodically occurring time reference common to all of the data bit lines and defined in terms of the frequency and phase of a higher rate clock used to encode the data. In video recording and reproducing apparatus such as described herein, horizontal line related H/2 signals derived from the periodically occurring sync words synchronously inserted in each data bit stream in the horizontal blanking interval as hereinbefore described is defined in terms of the frequency and phase of the higher rate (455 times H/2) signal color subcarrier component and the 3SC data clock (1365 times H/2) and is available for use as the periodically occurring timing reference.

To effect time base correction of the reproduced and channel decoded data, the data in each data bit line is retimed to a common reference 3SC clock, being directed through a phaser 567. All the phasers in the data bit lines are clocked by a common stable reference 3SC clock provided by the reference clock generator 98 (FIG. 9A) to retime the data to a stable clock signal. In the illustrated embodiment, a multiple port shift register 568 performs the retiming by having data written into addresses determined by the write address generator 569 clocked by the decoded 3SC data clock provided by the channel decoder 525 on line 574. The data is read out of the register 568 under the control of the read address generator 570 clocked by the reference 3SC clock provided on line 571.

Because all of the phaser read address generators 570 in the 9 data bit lines are clocked by the same reference 3SC clock, the data in all the data bit lines are retimed to the desired stable 3SC reference clock, which for an NTSC television signal standard is 10.7 MHz.

The write and read address generators 569 and 570 are preset and reset respectively to their starting addresses by the sync word included in the data being corrected, with the starting write address in advance of the starting read address by four addresses. Each time a sync word is detected in the received decoded data by the first shift register and sync word circuitry 572, a reset signal resets the read address generator 570. The decoded data on line 566 enters a seven bit shift register in circuitry 572 and is examined for the occurrences of the seven bit sync word pattern by logic circuits forming the sync word detector portion of the circuitry 572. After passage through the shift register, the data is clocked into the multiple port shift register 568, which has an 8 bit capacity and is initially operated to read an address four 3SC cycles after writing of data at the address. Because the write address generator 569 and the read address generator 570 are clocked by the 3SC data clock and the reference 3SC clock respectively, data bit displacement errors in the received data will change the time of writing into an address relative to the time of reading from that address.

This change results in retiming of the received data to the 3SC reference. The phaser 567 will properly retime the received data to the stable 3SC reference even if the sync word is not detected by the first sync word detector 572 provided that unanticipated large time displacement errors, greater than the storage capacity of the register 568, do not occur; if they do, the video data emerging from phaser 567 will be at the proper rate but incorrectly phased.

The sync word detector 572 provides a first input to the gating circuitry 592 (FIG. 46C) each time a sync word is detected in the decoded data. The seven bit shift register is clocked by the decoded data clock on line 574 to enter the decoded data received over line 566 for examination by the logic circuitry. The sync word detector 572 is enabled for sync word detection by the sync word enable pulse generator 600. This generator is enabled by a divide by 1364 counter 590 clocked by the 3SC data clock on line 574. The generator 600 provides a sync word detection enable pulse on line 601 (FIG. 15B-(3)) which is initiated by the advanced EOC pulse (FIG. 15B-(2)) issued by the counter 590 over line 591 three counts in advance of the expected occurrence of a sync word at the first sync word detector circuitry 572 (FIG. 15By(6)0. This advanced EOC pulse also is coupled by line 591 to the gating circuitry 592 that responsively examines the output of the shift register to determine the data logic level and, hence, the phase of the decoded data clock. Upon the detection of a sync word by the second detector 575 (FIG. 15B-(6)), a reset signal is issued over line 608 to the generator 600. The reset signal terminates the enable pulse on line 601 before the counter 590 reaches a count of fifteen. The counter position 15 in the counter 590 terminates the enable pulse if a sync word is not detected by the second detector 575 (FIG.

15B-(7)). The shift register 604 provides the automatic EOC reset pulse to counter 590 over line 610 upon the occurrence of the third 6SC clock pulse after the advanced EOC reset pulse (FIG. 15C (2) and (5)). The shift register 604 and the pulse generator 605 cooperate to allow the sync word enable pulse to follow changes in the time of the occurrence of consecutive sync words in the amount of + 1 cycle of 3 SC. The pulse generator 605 simultaneously examines three outputs of the shift register 604 and generates a gating waveform (FIG. 15B-(4)) that prevents the sync word enable pulse from resetting the counter if it occurs within 1 clock time of the occurrence of the automatic EOC reset pulse generated by the shift register 604. If the reset enable pulse derived from a sync word arrives one count before the automatic EOC reset pulse, the counter 590 will not be reset (FIG. 15B-(4) and (8)). If the reset enable pulse is provided one count after the occurrence of the EOC reset pulse, the counter 590 will not be reset again (coincidence with the second positive pulse of the gating waveform provided by the pulse generator 605). If a sync word is not detected during the interval of the sync word enable pulse, the counter 590 will continuously reset itself through shift register 604 and line 610 (FIG. 15B-(5)) and, thereby, with generator 600 will retain, as a memory, knowledge of when to provide sync word enable pulses until a sync word is detected. Provided that the detected sync word is not in coincidence with the positive gating waveform (FIG. 15B-(4)) provided by the generator 605, NAND gate 612 is enabled to permit the sync word to be placed on line 613 for resetting the counter 590.

The vertical blanking signal on line 606 (FIG. 15B-(1)) is coupled to place the sync word enable pulse generator 600 in the enabled state for an interval of ten horizontal lines by disabling gate 611 and preventing coupling of the 15 count position of counter 590 to generator 600. This enables the decoder/time base corrector circuitry to lock to the sync word detectors 572 and 575 and thereby to be enabled at sync word time and set the phase 568 and error gate 582 for proper operation.

The data is read from the multiple port register 568 with the 3 SC reference clock into the shift register portion of the second sync word detector circuitry 575 (FIG. 46B). The shift register portion has three of its output lines 576 coupled to the data input of a serial to parallel converter 577. The multiplex clock provided on line 578 by the reference clock generator 98 is at the SC rate and latches the data in blocks of three data bit cells from the shift register portion of the circuitry 575 into the converter 577 of which the contents are transferred to a following RAM 579 each cycle of SC. The three output lines 580 of the converter 577 extend to the input of a RAM 579. The final time base correction is performed in RAM 579 whose write address generator 614 is clocked at reference SC, since the data rate at the input operation of the RAM is SC. The read address generator 615 and the latch and subtractor circuitry 623 are also clocked at reference SC to cause the reading of the RAM addresses. Read/write mode signals and write enable signals from the reference clock generator 98 of FIGS. 44A-D control the reading and writing of the RAM addresses so that a read cycle occurs during one part of a subcarrier sycle and a write cycle at a different part of the cycle (see FIG. 12B).

The amount of time displacement error required to be corrected is determined by the error gate 582. Upon the detection of the sync word by the second sync word detector 575, a signal placed on line 608 opens the error gate and allows reference 3SC clock pulses placed on line 571 by the reference clock generator 98 to pass to a divide by three counter 583. One output of the counter 583 extends to the read error address generator 623 to provide SC rate clock pulses to the generator. When the reference H/2 is received on line 581 and from the reference clock generator 98, the error gate 582 is dosed, terminating the coupling of reference 3SC clock pulses to the counter 583. Consequently the SC rate clock pulses are no longer provided to the rear error address generator 623 and the number being provided at such time represents the time displacement between the video signal's sync word and reference 11/2 in a whole number cycles of SC. Also, a delayed pulse is generated by the delay and pulse former 621 in response to closure of the error gate 582. The delayed pulse is coupled to the read address generator 623 and latches the error count therein.

Subsequently, a reset pulse is generated from the latch pulse to reset the + 3 binary counter 583 and read error address generator 623. The counter sets the read address in accordance with the timing difference between reference H/2 and the sync word detected by the second sync word detector 575 measured in cycles of 3SC divided by three. The measured value of the timing difference is coupled to a latch and subtractor 624 and is subtracted from the write address to generate the correct read address. Because the clocks representing error are divide-by-three, the RAM 579 will adjust for errors of integral numbers of subcarrier cycles. A 3-bit shift register 617, error latch 618 and gates 619 provide correction in fractions of one cycle of 3SC of any residual error remaining after the data has passed through the RAM 579. The parallel to serial converter 620 at the output of the RAM 579 receives the demultiplex clock from the reference clock generator 98 and converts the data rate back to 3SC at the input of the shift register 617. FIG. 15C shows the typical correction performed by the phaser 567 and following time base correction by the RAM 579 and shift register 617. The corrected output of the time base corrector 565 appears at terminal 622.

However, the use of the reference H/2 signal, which is redefined with respect to a particular phase of subcarrier, in measuring the time displacement error through the operation of the error gate 582 results in the 46 nsec 15hz jitter in the video signal provided by the time base corrector 565.

The 9-bit parallel output of the time base corrector 565 is coupled to the data transfer circuitry 129.

During playback, after the data has been decoded and time base corrected by the data decoder and time base corrector circuitry 100, the eight bit lines of video data plus the added single parity bit line, if parity protection is included, are applied to the data transfer circuitry 129 shown in block diagram of FIG 9A and the output of the data transfer circuitry is applied to the chroma processing circuitry 101 in the event that a normal playback mode is being used or to the encoder switch 126 in the event the data is being transferred to another disc drive memory using the apparatus in the transfer mode.

The data transfer circuitry performs a parity check of the data coming from the time base correctors and initiates an error masking function in the event that errors are detected during the parity check. The data is appearing at a 3SC rate and every three samples of the NRZ data will represent approximately the same video information. The error masking portion of the circuitry clocks the data streams through a series of flip-flops which provide a three bit memory so that if the parity check detects an error, the third previous data word is reinserted into the position where the error was detected. Reinsertion of the third previous data word masks the error, the assumption being that the third prior data word would be more representative of the correct video information than the data word that contains the error. Every third sample is reinserted in place of the detected error sample because the 0 sample (for example) from the previous cycle of 3SC would probably be more accurate than a sample taken at either 1200 or 240 , assuming the level of the signal which is sampled contains chrominance information that remains approximately constant for a period of a few samples.

The data received at the input to the output using a 3SC PALE clock to reposition the samples into the desired vertically aligned positions that was achieved by the original PALEing during sampling in the analog-to-digital converter 95. When the signal was channel encoded, the alignment was changed due to the fact that a line to line continuous phase 3SC clock was used to channel encode the NRZ data. The data emerging from the time base corrector circuitry 565 is thus aligned the same way as the encoded data at the output of the encoder 96. Accordingly, the data transfer circuitry 129 again PALEs the data to realign the samples in the manner as shown in FIGS. 9C(10) and 9C(11).

Referring to the block diagram of the data transfer circuitry 129 shown in FIG. 16, time base corrected data provided by the decoder and time base corrector circuitry 100 over nine bit lines, i.e., eight bit lines containing video information and one parity line, are applied at nine input lines of the data transfer circuitry. The line 625 in FIG. 16 represents the most significant bit line and is representative of each of the nine input lines provided for each bit stream. The data is clocked into FF 626 and FF 627 using a 3SC PALE clock signal which appears on lines 628 and 629. The PALE clock is generated by a PALE clock generator shown at the lower portion of the block diagram from 6SC and 1/2SC signals received from the reference clock generator 98 on lines 630 and 631, respectively, and a PALE flag signal received from the reference logic circuitry 125B via the encoder switch 126 on line 632. The PALE flag signal is applied through an inverter 633 and line 634 to one input of an AND gate 635. The line 634 also connects to a second inverter 636 which extends to one input of another AND gate 637 via line 638. The 1/2SC signal on line 631 passes through a pulse former 639 and clocks a divide by 2 FF 640 which produces 3SC output signals of opposite phases on output lines 641 and 642 which respectively extend to the other inputs of the AND gates 635 and 637. The outputs of the AND gates are connected to line 643 and extend to complementary dual output buffer 645 which clocks FF 626 and FF 627. The PALE flag signal on line 632 is a two state or level signal that changes state at an 11/2 rate and, upon changing levels, alternately disables AND gate 635 and enables the AND gate 637 to gate one of the 3SC signals from lines 641 and 642 onto the output line 643. Thus, in effect the PALE flag alternately changes the phase of the 3SC signal that is used to clock the data on line 625 and FF 627 so that consecutive horizontal lines of video data are clocked with opposite phased 3SC signals. This retimes the video data bits from the continuous phase clock back to the PALE clock so that the vertical alignment of samples of consecutive lines is re-established for subsequent chroma separation and processing. As previously described, the video data bits are not to be retimed in the transfer mode of operation. To prevent the retiming, the encoder switch 126 blocks the coupling of the PALE flag from the reference logic circuitry 125B to the data transfer circuitry 129 and instead places a low level signal on line 632. This places an enabling signal on the input of the AND gate 635 and a disabling signal on the input of the AND gate 637, where a line to line continuous phase 3SC clock signal is provided on line 643 through the AND gate 635.

The data on the output of FF 627 extends to an AND gate 647 via line 648 and AND gate 647 has output line 649 connected to the first of three FFs 651, 652 and 653 which serve to shift the serial bits to the output of the last FF which appears on line 654. Line 654 also extends to one input of another AND gate 655. A parity tree error detecting circuit 656 is coupled to receive the data bits of the nine bit streams as described below and has two outputs lines 657 and 658 which extend to AND gates 655 and 647, respectively. When an error is detected, it disables AND gate 647 to block the bit containing the error and enables AND gate 655 so that the output data on line 654 can be clocked through AND gate 655 onto line 649. This has the effect of replacing the incorrect bit with the third previously occurring bit in the data stream and effectively masks the error with the bit that is approximately correct for the reasons that had been previously discussed.

Five bits, i.e., bits 2 through 6, or the next most significant bit through the sixth most significant bit are also sampled through a resistor ladder network 659 having weighted values to produce an analog conversion of the digital information which approximates digitally encoded analog information and is used to detect if chroma phase needs to be inverted. The output on line 660 extends to the reference clock generator 98 and is compared with the phase of the burst of the station reference video signal to determine if the chroma phase needs to be inverted. The digital-to-analog conversion occurring in the data transfer circuitry is gated to reject all but the burst and produces an imprecise, but sufficiently accurate determination of the burst phase for use by the reference clock generator.

Specific circuitry that can be used to carry out the functions of the block diagram shown in FIG. 16 is illustrated in FIGS. 47A and 4713. Since the operation is essentially the same as that described with respect to the block diagram, a detailed description of the circuitry will not be described herein. Referring to FIG. 47A, the parity tree error detection circuitry 656 comprises a number of exclusive OR gates 661 which are interconnected and associated with the eight data bit lines containing the video information. The outputs of the interconnected exclusive OR gates 661 are connected to one of the inputs of another exclusive OR gate 662 whose other input is coupled to receive the parity bit on channel 9. The exclusive OR gate 662 controls an FF 663 that has output lines 657 and 658 for controlling the AND gates as previously described to either pass the video data bits received on input lines 625 or replace an erroneous byte of 8 bits with the third previously occurring byte of 8 bits. The operation of the remainder of the circuitry shown in the schematic diagram shown in FIGS. 47A and 47B are essentially the same as described with respect to the block diagram of FIG. 16.

A television picture with a region of saturated color bounded along the bottom by a region of no color defines a vertical color transition along the horizontal boundary or color edge. Given three successive television lines A, B and C of a field, wherein the lines are within the saturated color region immediately above the color edge, a conventional comb filter generates the vectors representing chrominance in accordance with the relationship, - 1/4A + 1/2B - 1/4C.

However, the color subcarrier of an NTSC television signal has a 1800 phase shift between alternate lines A, B and C. Thus 1800 inversion of, for example, lines A and C and subsequent summation of the vectors +1/4A +1/2B +1/4C generates a full chrominance vector, herein termed 1B or simply, +B, the chrominance on line B. When this chrominance vector +B is subtracted from the wideband signal (which also contains the chrominance vector +B), the chrominance vectors cancel. The comb filter has effected complete chrominance and luminance separation, i.e., all chrominance is in the chrominance channel.

However, in a second case, if lines A and B are in the saturated color region, with line C in the region of no color, line A provides a chrominance vector equal to B in the negative direction and line B a vector equal to B in the positive direction. But line C provides a zero chrominance vector since it lies in a region of no color. When combining the vectors in accordance with the previous relationship, -1/4 of vector A is inverted and added to + 1/2 of vector B, thereby providing a sum of +3/4 of a full vector B. It follows that when the chrominance +3/4B is subtracted from the wideband signal, i.e., line B, there is a residual of +1/4 of the chrominance vector left in the luminance channel, while only +3/4 of the chrominance vector is extracted in the chrominance channel.

A third case exists wherein only line A is within the saturated color region, and lines B and C are in the region of no color. The third case is similar to the second case above, wherein however, the signs are opposite.

The consequence of the second (and third) case given above, wherein line C (or B and C) lies in a region of no color, prove disadvantageous when attempting to reconstitute a composite NTSC color television signal from a single stored color field, or picture frame. As is well known, when reproducing the composite video signal from a single stored picture frame, in one frame the chrominance is added directly back to the luminance previously separated therefrom, whereas in the second picture frame the chrominance component is first inverted and then is added to the luminance. Therefore, in the second case mentioned above wherein line C is in a region of no color, in the non-inverting frame, the + 114 chrominance vector which remained in the luminance channel due to incomplete separation is added to the +314 chrominance vector separated into the chrominance channel. Thus the full vector B, i.e., the full chrominance signal, is recovered to define a correctly reconstituted color television signal for the non-inverted frame. However, when reconstituting the second picture frame of color video from the single stored picture frame, the chrominance (+314B) is first inverted, providing a -3/4 chrominance vector, which when subsequently added to the +1/4 vector in the luminance channel provides only a -1/2 chrominance vector for the inverted frame. Thus, in the non-inverted frame, the chrominance is reproduced with full saturation, whereas in the alternate, inverted frame the chrominance is reproduced at 1/2 saturation. Thus the color saturation defining the color edge between the region of full color and that of no color will flicker at a 15 Hz rate between 1/2 saturation and full saturation. This visible flicker is objectionable when reproducing the composite NTSC four-field color coded television signal.

The chrominance separating and processing system provides various embodiments of digital circuits which perform the inversion process digitally in combination with a digital comb filter and digital bandpass filter, while further providing a conditioned chrominance signal which, when digitally re-combined to form the composite NTSC color television signal, minimizes or cancels completely the objectionable 15 Hz flicker at the vertical transitions.

Although the combination is hereinafter particularly described utilizing a three times subcarrier (10.7) megaHertz phase alternating line encoding (PALE) sampling technique with a PCM encoded NTSC video signal, it is to be understood that other encoding techniques, sampling techniques, frequencies, etc., may be employed. Furthermore, the single lines depicting the inputs and outputs of the block diagram components are representative of words of selected numbers of bits, as exemplified in the detailed schematics of FIGURES 48, 49 and 50.

Referring to FIGURE 17, there is shown a digital chrominance separating and processing system wherein a 10.7 megaHertz (MHz) PALE PCM color video signal is introduced via input terminal 700 to digital comb filter means 701. The filter means 701 is per se generally typical of digital comb filters presently utilized in various television signal processing systems, but herein is adapted via a specific clocking technique further described hereinbelow, to separate the chrominance from the digital wideband color signal. The outputs from filter means 701 and the associated clocking techniques, include a 1H delayed wideband signal (delayed by onehorizontal-line delay period) on line (terminal) 702, and an extracted chrominance signal (with low frequency components still included) on line (terminal) 703a. The term "extracted" is herein used to define the chrominance signal which is separated into a chrominance channel, whether the separation is complete, or incomplete, as previously described with respect to case two (and three) hereinabove.

* The extracted chrominance signal is fed to bandpass filter means 704 which removes vertical resolution losses due to the comb filter means, by passing only that frequency band occupied by the chrominance information. The bandpass filter means 704 is centered at 3.58 MHz (the NTSC subcarrier frequency) and has a bandwidth of, for example, 1.5 MHz.

The resulting combed chrominance signal is fed via line (terminal) 703b to a digital circuit for inverting the chrominance signal on alternate frames at the frame rate. In FIGURE 1 the inverting circuit comprises a digital transversal filter with odd symmetry 705, which herein may be further identified as a modified, digital, "Hilbert" transformer. It is understood that the transversal filter 705 provides one form of inversion; i.e., employs what is basically known as the Hilbert transformer, but which is herein modified to provide a specific form of transversal filter with odd symmetry, while further providing a digital rather than analog inversion implementation. The transversal filter of interest has the property of rotating the phase 90" of all frequencies of a selected range which herein, for example, may be two to four MHz.

Thus the term inversion, or inverting is employed to define the circuitry and process of digitally conditioning the chrominance at the frame rate (or field rate if one field is used to reconstitute the four field color coded NTSC color television signal) as by phase shifting, rotating, inverting or otherwise handling the phase. Further, the successive playbacks of either a single stored field or picture frame is referred to generically as "alternate repretitive reproductions".

The chrominance signal is also fed to a negative input of digital adder (subtractor) means 706.

The 1H delayed wideband video signal of terminal 702 is fed to the positive input of the adder means 706. The transversal filter 705 includes a control input at 707 which determines the conditioning of the chrominance signal phase. In one embodiment, for example, the transversal filter may provide a plus and then a minus 90" phase rotation of the chrominance with respect. to the luminance signal in alternate repetitive reproductions. The chrominance and luminance signals are then summed in digital adder means 708 to provide the composite color television signal on output terminal 728.

Control means 709 includes various timing and clock inputs thereto which, for example, relate to the overall apparatus timing and thus originate upstream in the apparatus. In turn, the control means 709 generates specific control signals for the comb filter means 701, for the transversal filter control input 707, for the bandpass filter means 704, etc., which control signals include a PALE clock and 1H delay line, four-phase clocks, inter alia. The control means 709 and the various inputs and outputs are further shown and described in detail in FIGS. 48A, 48B 49A, B and C and thus are not further described here.

Briefly, in FIG. 17, the comb filter means 701 combines the three adjacent television lines A, B, C of previous mention, and includes a pair of digital, one-horizontal-line (1H) delay lines 710, 711 and a pair of adder means 712, 713. The 10.7 MHz PALE video signal is fed to delay line 710 as well as to adder means 712. The 1H delayed signal is fed to 1H delay means 711, and to the adder means 713. The 2H delayed signal is fed to the other input of adder means 712, whose output in turn is fed to the negative input of the (subtractor) adder means 713.

The digital comb filter means 701 and digital bandpass filter means 704 exemplified in block diagram herein, generate (eight bit) digital words corresponding to the separated chrominance and 1H delayed wideband signals, and are further depicted in the schematic diagrams of FIGS.

48A-B and 50A-B.

The combed chrominance signal is subtracted from the 1H delayed wideband video signal via the digital adder means 706, wherein the resulting combed luminance signal is fed to the digital adder means 708.

FIG. 18 shows the digital transversal filter 705 wherein the digital combed chrominance signal is fed to a series of one-sample-period delays 714a-714c, and also to the positive input of an adder means 715b. The negative input of adder means 715b is coupled to the output of the last delay 714c. The positive and negative inputs of an adder means 715a are coupled to the input and output respectively of the delay 714b. The outputs of adder means 715a, 715b are coupled to respective multiplier programmed read-only memories (PROMs) 716a, 716b, and thence to adder means 717. The latter is coupled via an inverter stage 718 to the adder means 708 of previous mention, along with the combed luminance signal from adder means 706, whereby means 708 generates the composite color television signal. The control input 707 is coupled to the inverter stage 718.

In operation, the transversal filter 705 provides digital circuits for conditioning the phase of the chrominance signal with respect to the luminance signal; i.e., for providing the digital implementation of phase inversion of the chrominance on alternate color picture frames. To this end, the 1H delayed wideband signal, and the chrominance signal are introduced to the adder means 706 via terminals 702, 703b respectively, whereupon the resulting luminance signal is introduced to adder means 708. The chrominance signal is delayed one-sample period (e.g., 93 nanoseconds) in each of the delays 714a-714c, whereby the undelayed chrominance and the three-sample delayed chrominance are introduced to the adder means 715b, and the one-sample and two-sample delayed chrominance signals are introduced to adder means 715a.

The delays 714a-714c may comprise a single stage of a shift register. The adder means 715a, 715b provide signals to multiplier PROMs 716a, 716b respectively, which perform a multiplication of the respective signals by .575 and .096 in a digital approximation of a conventional convolution operation. The resulting signals are then summed via adder means 717, and the summed signal has all of its frequency components advanced 90" with respect to the luminance signal, to define the conditioned chrominance signal of previous mention. The output of adder means 717 is delivered to the adder means 708 via the inverter stage 718.

During one color picture frame the inverter means 718 has a high, or "1", introduced thereto via the control input 707 from control means 709, whereby the (eight) bits of the output word are delivered unchanged to the adder means 708. On alternate video color picture frames the control input 707 is a low (or "0") invert enable signal (see FIG. 49). Data is represented in this device in the signed two's complement negative system where negative numbers have a "1" in the sign bit position and the magnitude is the 2's complement of its absolute value. Therefore, inversion amounts to changing the sign and forming the 2's complement, via the "0" invert enable input 707. Thus the conditioned chrominance signal (which is rotated +90 ) is directly added to the luminance in one frame, and is inverted and then added to the luminance in the alternate frame, to provide the composite color television signal on output terminal 728.

Alternately, the chroma first may be rotated -90" on each frame by reversing the inputs to the adder means 715a, b and then adding directly in one frame and inverting 1800 and adding in the next.

In another alternative, the transversal filter 705 may be implemented whereby during one color picture frame the phase of the chrominance signal is advanced by 90 , and during the alternate color picture frame is retarded by 90 , to provide in essence the 1800 inversion of the frequency components between frames.

FIGS. 48A-C, 50A-B and 49A-B illustrate one schematic implementation of the embodiment of FIGS. 17 and 18 utilizing the digital transversal filter with odd symmetry 705. FIGS. 48A-C illustrate one implementation of the digital comb filter means 701, and part of the control means 709 of FIG. 17; FIGS. 50A-B illustrate one implementation of the digital bandpass filter 704; and FIGS. 49A-B illustrate one implementation of the digital transversal filter 705, signal re-combining adder means 706, 708, and the remaining circuits of the control means 809. In all figures, like components of FIGS. 17 and 18 are indicated by similar numerals.

Thus, in FIG. 48A the 10.7 MHz PALE video signal is introduced via the input terminal 700 to the digital comb filter means 701. The output thereof (FIG. 48C) comprises the separated chrominance and 1H delayed wideband signals on terminals 703a and 702 respectively. The inputs at terminal 719, 725 comprise a group A and B control signals and a symmetrical PALE clock, generated in the respective portion of control means 709 of FIG. 49B further described below. The terminals 719, 725 are coupled to a four-phase clock generator 720 of control means 709 depicted in FIG. 48A. The clock generator 720 forms part of the timing circuits for clocking the shift registers which comprise the 1H digital delay lines 710, 711. The delay lines 710, 711, adder means 712, 713 and terminals 702, 703a, are interconnected via integral latching circuits 712a, 713a and 721 which conventionally temporarily store the respective digital products of the preceding shift registers. adders, etc. Terminal 703a provides the input to the succeeding digital bandpass filter means 704 of FIGS. 50A-B, and the terminal 702 provides the input to the adder means 706 of succeeding FIG. 4913.

The delay lines 710, 711 further each include a series of two-phase shift registers 750, 751 respectively, employing two-phase clocks, wherein the register stages are further arranged into groups 750A, 750B of delay line 710, and 751A, 751B of delay line 711. Shift register stage selectors 752A, 752B select portions of the digital word corresponding to specific clock phases of groups A, B of delay line 710, and shift register stage selectors 753A, 753B do the same for delay line 711. Wideband signal selectors 754, 755 of delay lines 710, 711 respectively, then provide selection of the digital words corresponding to the 1H and 2H delayed wideband signals, respectively.

The wideband video signal word is split, and is clocked into four bit stages of the shift registers 750A, 750B by the four-phase clocks, which are in effect, four phases of the symmetrical PALE clock. The stage selector 752A receives and loads the pairs of four bits in response to PALE clock, alternately from different pairs of stages of shift register 750A. Stage selector 752B does the same with shift register 750B stages. The group A stage selectors 752A unload into one (four bit) wideband signal selector 754, while the group B stage selectors 752B unload into the other (four bit) selector 754, in response to timed PALE clocks respectively. At selected times, the group B selectors are clocked whereby the combined group A and B registers provide a total of 680 bits per television line. One NTSC horizontal television line sampled at three times subcarrier rate will contain 682-1/2 samples. However, as will be described in more detail hereinbelow, the clocks for the shift registers are generated and applied to the registers so that the total bits per television line output by the register for each bit line is equal to an integral number of samples. In the embodiments described herein, 680 samples per television line are clocked through the registers. The clocking of the registers is arranged so that the discarded interval of 2-1/2 sample intervals occurs outside the active video information portion of the television line during the horizontal blanking interval.

The control circuits 720 of FIG. 48A, which provide the four-phase clocks for the shift registers 750A, 750B and 751A, 751B and which receive a symmetrical PALE clock, are further described in operation in the block diagram and clock waveforms of the combined control means 709 in FIGS. 49C-D infra, with one implementation thereof illustrated in the schematic diagrams of FIGS. 48A, 49A-B.

FIGS. 50A-B depict the bandpass filter means 704 with terminal 703a providing the incoming extracted chrominance signal from the comb filter 701 output, FIG. 4813. The combed chrominance signal from the bandpass filter means 704 is provided at terminal 703b of FIG.

50B, which forms the input to the transversal filter with odd symmetry 705 of succeeding FIGS.

49A-B. Immediately preceding the terminal 703b is an adder/latch stage 756, wherein the latches are clocked by a chroma invert enable signal via a terminal 757. In the embodiment employing the transversal filter 705 (FIGS. 17, 18, 49), the chroma invert enable signal does not enable the clear input of the latches, and the signal into the adder/latch stage 756 appears at the terminal 703b. The PALE clock of terminal 725 couples to various inverters (FiG. 50B) to provide a plurality of clocks for the adders and latches that comprise the bandpass filter means 704. The latches are thus clocked by the PALE clock to deliver the digital output from the preceding logic processor component (viz, the adders) to the succeeding logical processor components (also adders).

The final adder/latch stage 756 of the bandpass filter means 704 delivers the combed chrominance signal.

One-horizontal-line delay lines are required to provide the comb filtering process of chrominance signal separation from a wideband signal. The delay lines, and thus the comb filter 701, must be in synchronism with the overall system timing, which inter alia is represented by the input termed PALE flag. As discussed herein with reference to the video signal system of FIG. 9A and the reference logic circuit 125B of FIG. 11A in particular, the PALE flag signal is an 11/2 rate signal generated from H sync and a field index signal indicative of the first field of every two field sequence of a television signal and asymmetrical, i.e., has one phase during the active video signal portion of one line and the alternate phase during the horizontal blanking interval and the active video signal portion of the next line. The phase of the PALE clock changes coherently with the asymmetrical PALE flag. However, the PALE clock utilized by the instant chrominance separating and processing circuit utilizes a symmetrical PALE clock, i.e., one in which clock has alternate phases for the same time duration.

A paramount problem when attempting to reconstitute the composite color television signal from a single stored color field or frame, stems from the fact that each line of a field is of a duration equal to 227-1/2 cycles of subcarrier fsc. That is, it is equal to an integral number of cycles plus one-half cycle of subcarrier time. It follows that a required condition of 1H delay lines, when they are formed of digital shift registers such as, for example, those in the comb filter means 701, is that there is an integral number of samples per line of television and thus one horizontal line of delay.

Accordingly, the present invention provides the control means 709 which inter alia generates the symmetrical PALE clock from the overall apparatus asymmetrical PALE flag, and which, during the horizontal blanking period deletes an integral number plus one-half of subcarrier cycles. to shift by 1800 with respect to previous samples at the line rate. The PALE clock thus is in the proper phase relationship with the subcarrier frequency as required to reconstitute the four fields required to color encode the television signal, while also being in proper timing relationship with the overall apparatus.

Accordingly, FIG. 49C depicts in block diagram form the digital control means 709 shown in one schematic implementation in FIGS. 48A-B and 49A-B. FIG. 49D is a timing diagram of the waveforms generated at various points along the circuit of FIG. 49C, as well as FIGS. 48A-B and 49A-B. Inputs from the overall system include the asymmetrical PALE flag from reference logic circuitry 1258, a x6 phase continuous subcarrier frequency (6fsc) and a xl/2 phase continuous subcarrier frequency (1/2 fsc) from the reference clock generator 98 and a field index pulse from reference input circuitry 93B, on respective terminals 758, 759, 760 and generally indicated at 762, which in turn is coupled to the four-phase clock generator 720 of that portion of control means 709 in FIG. 48A. The latter provide the four-phase clocking of the shift registers 750A-B and 751A-B, as further described below.

The PALE clock generator 762 receives the PALE flag via terminal 758, and feeds it to an exclusive OR 763. The latter is coupled to a D-type flip-flop 764, together with the 1/2 fsc clock from terminal 760. The exclusive OR 763 and flip-flop 764 define a gated phase detector. A D-type flip-flop 765 is coupled to flip-flop 764 and is clocked by a correction pulse on line 766 corresponding to the group A control signal provided by a count decoder 772, further described infra. A JK-type flip-flop 767 is coupled at pin K thereof to flip-flop 765, and is clocked by the 6 fsc clock from terminal 759. The flip-flop 767 is coupled to an AND gate 768 and back to the clear pin of the flip-flop 765. The flip-flops 765, 767 and the AND gate 768 together define a gated phase corrector. AND gate 768 also received the 6 fsc clock, and is coupled in turn to a divide-by-two (+ 2) JK-type flip-flop 769 and to a divide-by-1365 (+ 1365) counter 770. The + 1365 counter 770 receives the field index pulse from terminal 761, and is coupled to the + 2 flip-flop 769 via a reset pulse generator means 771. As shown in FIG. 49B, the field index pulse first is reclocked to inverted 2 fsc via a flip-flop stage. The counter 770 is also coupled to a count decoder 772 which generates the group A and B control signals on terminal 719. The group A control signal defines the correction pulse 766 which clocks the flip-flop 765. The output of the t 2 flip-flop 769 comprises the symmetrical PALE clock which is fed back to the second input of the exclusive OR 763 to define a closed loop in the PALE clock generator 762. The PALE clock is also fed via terminal 725 to the four-phase clock generator 720 of FIGS. 48A-B and 49C, which as shown, only generates group A four-phase clocks.

In operation, referring to FIGS. 49C and 49D, when the chrominance separating and processing system is turned on, the counter 770 is not properly set and accordingly is reset via the reclocked field index pulse. The latter is a 30 Hz pulse which occurs on a selected field wherein sync pulses coincide with vertical interval. After reset, the PALE clock generator starts generating an initial PALE clock which resembles true PALE clock. However, the PALE clock must be in phase with the apparatus PALE flag, during the active part of a television line. That is, when PALE flag is up, the rising edge of 1/2 fsc is supposed to coincide with a rising edge of PALE clock, and vice versa. To this end, the (initial) PALE clock, which may resemble the waveform of either FIG. 49D - 17 or 18 when the circuit is turned on, is fed back to the exclusive OR 763 together with the PALE flag. When PALE flag is high. the exclusive OR output is low when PALE clock is low. When PALE flag is low, the exclusive OR output is low when PALE clock is high. Thus the PALE clock is de-PALEd to provide 3 fsc which is fed to the flip-flop 764 together with 1/2 fsc. The flip-flop 764 compares the de-PALED signal and the 1/2 fsc signal waveforms 49D - 16, 17 and 18). If flip-flop 764 takes the data the PALE clock is not in phase with the PALE flag, and vice versa. Thus the exclusive OR and the flip-flop 764 provide the gated phase detection.

If the PALE clock is not in proper phase, the gated phase corrector formed of flip-flops 765, 767 and AND gate 768, deletes one cycle of the 6 fsc clock to shift the phase by 1800 and bring PALE clock into the proper phase relative to PALE flag. The correction pulse 766 delays the time that the detection and correction is made, i.e., during the active part of the television line where it is known that the phase is the same. Because PALEing of the sampling clock used in the video signal system does not occur during the horizontal blanking interval as decribed hereinbefore with reference to FIGS. 9 and 11, detection of the proper phase of the symmetrical PALE clock cannot occur during the horizontal interval. However, once the proper symmetrical PALE clock phase is detected, the PALE clock phase thereafter changes during the horizontal blanking interval in the chrominance separator and processing circuitry 101.

The counter 770 counts down 1365 counts of 6 fsc (FIG. 49D - 1) corresponding to one television line, and delivers a carryout (FIG. 49D - 3) to the reset pulse generator 771 on a rising edge of 2 fsc (FIG. 49D - 2). The latter includes a series of D-type flip-flops which provide six counts after carryout goes low, and thus the succession of highs depicted in FIG. 49D - 4 through 8. The inverse output signals corresponding to the waveforms of FIG. 49D - 6, 8 provide the start and end of a low state to the + 2 JK-type flip-flip 769 (FIG. 49D - 9), which in turn generates the symmetrical PALE clock at 3 fsc (FIG. 49D - 10) which appears at terminal 725.

As may be seen by comparing FIG. 49D - 10, 11, the phase of the PALE clock is shifted by 1800 by deleting 2-1/2 cycles of the phase continuous 3 fsc signal. To this end, after the input to the t 2 flip-flop 769 goes low, the rising edge of the PALE clock corresponding to the next rising edge of 6 fsc stays low, as do the two following rising edges of PALE clock. On the following rising edge of 6 fsc after the input to flip-flop 769 goes high, the PALE clock goes high, but with 1800 phase shift relative to its phase during the prior line (FIG. 49D - 11), thus the requirement of deleting the 1/2 cycle of subcarrier each television line is accomplished.

The count decoder 772 is coupled to the counter 770 and generates the group A and B control signals after a selected count, the signals being introduced via terminal 719 to the four-phase clock generator 720. The group A control signal is also fed to the gated phase corrector as a pulse on line 766 as previously mentioned.

The four-phase clock generator 720 provides for selected timing control of the comb filter shift registers 750A-B and 751-B, whereby the outputs thereof fulfill the requirement of generating an integral number of samples per television line, e.g., 680, utilizing the symmetrical PALE 3 fsc sample clock. This circumvents a further problem caused by the integral number of subcarrier cycles plus one-half cycle per line, wherein the one-half cycle prevents proper sampling from line-to-line and must be deleted, or otherwise compensated for. To this end, the four-phase clock generator 720 includes a divide-by-four ( . 4) binary counter 773 coupled to the PALE clock via terminal 725, and thence to a one-out-of-four binary decoder 774, and to the shift register stage selector 752A (and selector 753A) of previous mention in FIGS. 48A-B.

The binary decoder 774 data input is connected to a high, wherein the selected output equals a low, and the unselected outputs equal highs. The shift register stage selectors 752A and 752B are coupled to the wideband selector 754 (FIG. 48A) which selects digital words from shift register group A or B in response to the group A and B control signals from the count decoder 772. Binary decoder 774 is coupled to a latch 775 and thence to four D-type flip-flops 776a-d.

The latch 775, whose output follows its input, is coupled to PALE clock and flip-flops 776a-d are also coupled thereto via an inverter 777. The four phase clocks are generated on outputs 1, 2, 3 and 4 of the flip-flops 776a-d via inverter stages, and are shown in FIG. 49D - 12 through 15. The clocks 1, 2, 3 and 4 are introduced to the shift registers 750A of delay line 710, as well as to shift registers 751A of delay line 711 (FIGS. 48A-B). The video input signal is introduced to the shift registers at terminal 700.

In operation, the overlapping four-phase clocks 1 - 4 (of the order of 150 nanoseconds) are applied to the multi-stage, two-phase shift registers 750A (751A) to clock successive four bit pairs into alternate stages to provide the clocking rate required, which rate the shift registers could not handle without employing the four-phase clocking into alternate stages. Note that the four-phase clocks are disabled, FIG. 49D - 12-15, during the 2-1/2 cycles of PALE clock FIG.

49D - 10, to provide the exact 1H delay. In addition, since shift registers having a capacity of 512 bits are readily available, they are employed to provide the 680 bits corresponding to one-horizontal-line delay.

Although only the group A shift registers 750A, 751A and timing controls therefor of only the delay line 710 and 711 are shown in FIG. 49C, it is understood that the PALE clock line 725 and group B control signal line 719 also are introduced to the group B shift registers of delay line 710 (FIG. 48A). Furthermore, the delay line 711 (FIG. 48C) is identical to delay 710 and similarly employs the PALE clock and group A and B control signals.

FIGS. 49A-B depict one digital implementation of the control means 709 of FIG. 49C, and also of the transversal filter with odd symmetry 705 of FIG. 18, the latter including terminals 703b and 702 for receiving the combed chrominance and 1H delayed wideband signals, respectively.

The various components 714-718 of the filter 705 are shown in schematic, and define means for rotating the phase of the chrominance signal +90 , whereupon inverter means 718 inverts the signal 1800 in response to the control input 707. A -90" rotation may be generated by corresponding sign changes. i.e, by clocking the latches of adders 715a, 715b, with inputs that are opposite in sign to those shown in FIG. 18. Inverter means 718 is defined herein as a plurality of exclusive ORs which essentially perform the 1800 inversion.

The bandpass filter inherently has a gain of 27/32, hence the gain of the wideband signal must match this gain. Therefore, in FIGS. 49A-B, the 1H delayed wideband signal is coupled to a 27/32 multiplier PROM 722 which multiplies the wideband signal by 27/32, to provide an overall gain of unity. The wideband signal is then fed through an (eight stage) delay 723, which equalizes the delays in the wideband channel with the delays in the chrominance channel caused by the bandpass filter means 704, and thence to the adder means 706. Various latches 724 are provided between the adder means 706 and 708, which provide a temporary store of the intermediate signal while clocking the luminance signal from adder means 706. The composite color television signal is provided on output terminal 728 via the adder means 708 of FIG. 49B by combining alternate repetitive reproductions of the stored video signal.

The block diagram of the PALE clock generator 762 of FIG. 49C is shown in schematic in FIGS. 49A-B while the four-phase clock generator 720 of FIG. 49C is shown in schematic in FIGS. 48A-B. Since the operation of the generators 762 and 720 were described in FIG. 49C, no further explanation is required in the schematic diagrams of FIGS. 49A-B, wherein like components are identified by similar numerals.

However, in addition, FIG. 49A includes a terminal 778 for receiving a chroma switch and a frame switch input, which are provided by the computer control system 92 via the blanking insertion and bit muting circuitry 127 and the reference clock generator 94, respectively. The frame switch input is a chrominance inversion enable signal generated by comparing the phases of the system playback color burst and the continuous phase SC. When the phases differ, the frame switch input conditions the chroma circuitry for inversion. Thus the frame switch input generates the control input 707 to the transversal filter 705 in the form of the chroma invert enable signal, as further described below, which is the same chroma invert enable signal which is fed to the adder/latch stage 756 (FIG. 50B) on the terminal 757 of FIGS. 49B and 50A. As previously described in FIG. 18, the chroma invert enable is high during one picture frame to pass the input unchanged through the inverting ORs 718. In the alternate frame, the invert enable is low to change the sign and form the 2's complement to thus invert the chrominance.

The chroma switch input of terminal 778 couples to the frame switch input via AND gate 779 and prevents the frame switch signal from enabling inversions when the apparatus is not receiving signals from the (disc/tape) storage, e.g., when the apparatus is in electronics-toelectronics mode and chrominance inversion is not desired.

Referring still to FIG. 49A-B, the PALE clock generator also provides the PALE clock on lines 781, 782 via the inverse pin of the + 2 JK flip-flop 769 and inverters 780. The PALE clock is used conventionally to clock the various latches associated with the adder means 715a, b, the multiplier PROMs 716a, b, the one-sample delay lines 714a, b, c and the delay 723.

Referring now to FIGURE 19, there is shown an alternative embodiment of the chrominance separating and processing system, wherein like components are similarly numbered as in FIGURE 17. The transversal filter 705 of FIGURES 17, 18, 49 is replaced by digital inverting means 705a, which is selectively enabled via a control input 707a thereto. In one frame the inverting means passes the incoming signal from the bandpass filter means 704 without changing it, whereas in the alternate frame the control input 707a provides an invert enable signal to the inverting means to shift the bits of the incoming digital word by 180C prior to introducing them to the adder means 708. The luminance signal derived from adder means 706 is delivered to the adder means 708, which latter means generates the composite color television signal on terminal 728, as previously described.

FIG. 20 depicts a modification of the alternative embodiment of FIGURE 19, wherein adder means 706 is deleted and the inverting means 705a is replaced by inverting means 705b. Like components in the block of FIG. 20 are also similarly numbered. The inverting means 705b constitutes a digital multiply-by-two (x 2) stage 756a coupled to the bandpass filter 704, and thence to a negative input of an adder means 708a adapted to perform a subtraction process. As shown in FIGURES 49E-F, the x 2 stage 756a is actually disposed at the output of the bandpass filter means 704, and corresponds to the adder/latch stage 756 of FIGURE 505. The 1H delayed wideband signal on terminal 702 is introduced to the positive input of the adder means 708a.

In operation the x 2 stage 756a is controlled via the control input 707bow i.e., the chroma invert enable signal, whereby in one frame the stage provides a zero output such that the adder means 708a reconstitutes the composite color television signal from only the 1H delayed wideband signal. On alternate frames the chroma invert enable (707b) disables the x 2 stage 756a to allow passage of the digital signal to the negative input of the adder means 708a, together with the wideband signal from the comb filter means 701. The multiply-by-two process is actually performed by shifting the lines one bit, whereby subtraction of the doubled chrominance signal from the wideband signal via the adder means 708a sums the alternate repetitive reproductions to define the composite color television signal on terminal 728.

It may be seen that the system of FIG. 20 is simplified in that the adder means 706 is deleted.

In any event, the systems of FIGS. 19 and 20 provide a lesser degree of conditioning of the chrominance signal on repetitive playbacks than does the system of FIGS. 17, 18, 49. Thus the systems of FIGS. 19, 20 provide full saturation of the chrominance in the non-inverted frame, with of the order of 1/2 saturation in the inverted frame. However, the stability improvement provided by the all-digital processing, including the inversion process, correspondingly visually improves the color edges.

FIGS. 49E-F depict in schematic the inversion means and control means therefor, for the digital chrominance separating and processing system shown in FIGURE 20. To this end, the lH delayed wideband signal is introduced from the comb filter means 701 (FIG. 48B) via terminal 702, and the bandpass filter means 704 output of the combed chrominance signal is introduced via the digital X2 stage 756a (which herein forms part of the inverting means) from the terminal 703b of FIGURE 50B. To simplify the specification, the portion of the inverting means 705b corresponding to the adder/latch stage 756 of FIGURE 50B, is depicted in FIGURE 49E hereof, by the dashed block 756a inserted after the terminal 703b. The control inputs 707b, corresponds to the chroma invert enable signal of terminal 757 as previously described. Thus the latter enable signal enables the clear input of the latches of stage 756a on the non-inverting frame, to prevent passage therethrough of the signal and provide in effect the zero input from the bandpass filter to the adder means 708a. On the inverting frame, the chroma invert enable signal disables the clear input of the latches of stage 756a to pass the chrominance signal. The multiply-by-two process is conducted by shifting the wire connections to provide a bit shift of the digital word to double the chrominance signal.

The 1H delayed wideband signal is introduced to a delay 723a (FIGURE 49E) similar to delay 723 of FIGURE 49A, which equalizes the delays in the wideband signal with those of the chrominance signal introduced via the bandpass filter means 704. The wideband signal is then introduced to a 27/32 multiplier, 722a (FIGS. 49E-F), which performs a gain adjusting function.

The wideband signal from the 27/32 multiplier 722a is introduced to the adder means 708a, along with the output from the digital X2 stage 756a. The composite video signal is provided on terminal 728 via the subtraction process conducted on alternate frames, i.e., on alternate repetitive reproductions, by adder means 208a.

As in the circuit of FIGS. 49A-B, FIGURES 49E-F include the control means 709 having the inputs 758, 759, 760 and 761, the PALE clock generator 762, and the count decoder 772, as well as the group A, B control signals on terminal 719, and the PALE clock on terminal 725. As previously mentioned, the chroma invert enable on terminal 757 is introduced to the digital X2 state 756a. The PALE clock provided by the JK flip-flop 769 via inverters 780, is introduced via lines 781. 782 to the varous latches associated with the delay 732a, the 27/32 multiplier 722a, and adder means 708a, to clock the digital signals from the preceeding logical processor component to the succeeding logical processor component, as well known in the art. The various logical elements of FIGS. 49E-F are thus essentially similar to those of FIGURES 49A-B.

FIG. 21 illustrates in block diagram a digital chrominance separating and processing system which generally functions as those previously described, but which reconstitutes the composite color television signal by repetitive reproductions of a single stored color field. As in the previous figures, like components are similarly numbered. Thus the chrominance signal is separated from the field wideband signal via comb filter means 701, and is introduced to bandpass filter means 704 via terminal 703a. The 1H delayed wideband signal is introduced to the adder means 706 via terminal 702. The combed chrominance signal is introduced via terminal 703b to an inverting means 705c, and more particularly to: a transversal filter with odd symmetry 705 similar to that of FIGS. li, 18, 49, a third input to an electronic switch means 737; and a first input to a second electronic switch means 738. The number of the inputs of the switches corresponds to the playback number of the single field used to reconstitute the four fields of the composite color television signal. Accordingly, the output from the transversal filter 705 is coupled to a second input to the switch means 737, and to a fourth input to the switch means 738. The output from switch means 737 is coupled to an inverting means similar to 705b of FIGS. 20, 49E-F (or inverting means 705a of FIG. 19), which in turn is coupled to second and third inputs of switch means 738. The output of the latter is coupled to one input of the adder means 708. and the output of adder means 706 is coupled to the other input of adder means 708.

Control means 709 provides switching signals via control input 707c to step the switch means 737 and 738 through the inputs thereof at the field rate. to enable the transversal filter 705 and inverting means 705b. and to control the filter means 701, 704, adder means 706, 708, etc., as described above.

As is well known, a 90" phase rotation is required between fields since there is an integer number plus three-fourths cycles of subcarrier in a field. Thus the inverting means 705c provides shifting of the single stored field by 90"on each of four successive plays thereof, to reconstitute the four fields of the composite color television signal. To this end, on first playback of the stored field, the switch means 738 is stepped to the first input thereof, to deliver the combed chrominance signal from the bandpass filter means 704 directly to adder means 708 through switch means 738. together with the incoming luminance signal from adder means 706. The first field at Phase shift is thus delivered to terminal 728.

On the

On the fourth playback, the switch means 738 is stepped to the fourth input, whereby the chrominance signal is delivered to adder means 708 via the transversal filter 705 only, to provide a +90 rotation of the chrominance signal. The four fields are combined on successive playbacks via adder means 708 to generate the composite color television signal on terminal 728.

The sign of the phase shifting may be changed, and the circuit connections and clocks thereto adapted correspondingly, whereby on the second playback of the field the transversal filter 705 rotates the chrominance -90" and is then coupled to the adder means 708. On the third playback the inverting means 705b rotates the chrominance -180 , and on the fourth playback the transversal filter 705 provides -90" rotation, and is coupled to the inverting means 705b which provides -180" rotation, wherein the combination shifts the chrominance -270 , thus providing the 90" phase shift between playbacks.

The control means 709 provides the PALE clock, the four-phase clocks, the chroma invert enable signal, etc., to the varous components of the inverting means 705c, the filter means 701, 704 and to the adder means 706, 708, as described and shown in the embodiments of the previous figures.

As well known, when a composite color television signal is reconstituted from a single field, the horizontal sync pulses are not aligned on successive playbacks without the addition of one-half horizontal line delay on alternate fields. Although the chrominance processor of FIG.

21 is not directly concerned with this problem and will deliver the desired succession of fields, the use thereof would require adjunct means for detecting the vertical interval and for inserting the one-half line delay in response thereto as required, and as conventionally known in the art.

Although a 3 fsc sampling rate is employed in the description above, other sampling rates may be used. For example, 4 fsc 16/5 fisc. etc., may be employed. A sampling rate which provides an integral number of samples per television line is advantageous since PALE clock is not required; i.e., the PALE clock generator 762 may be omitted. Thus, the PALE clock per se is not necessary to provide the chrominance separating and processing functions herein. In addition, components such as the 27/32 multiplier and multiplier PROMs may be deleted from the systems, in the event a bandpass filter of unity gain is employed.

The functions carried out by the blanking insertion and bit muting circuitry are primarily those of inserting a black level during the blanking period as well as inserting a grey level during the time in which one picture or still image has been played and another has been addressed for playback. The disc drive head movement may take from one to four fields of time duration in which to change from one still to another, the time increasing the greater the radial movement.

Thus, if a track on the outside of a disc pack was being played and the next picture frame that was addressed happened to be on an inside track of the same disc pack, then almost four full fields of time would be required for the heads to move to the new position. Since it is aesthetically pleasing not to have a black picture during this time period, a grey level is inserted.

The circuitry also is adapted to perform bit muting operations which essentially enables one or more of the bits defining samples of a field to be set to the logical zero state for the purpose of performing special effects during playback. The circuitry shown in block 127 of FIG. 9A also generates a PALEd 3SC clock signal from a PALE flag signal for use by the digital-to-analog converter circuitry 102 and it also generates a continuous subcarrier sine wave signal that can be phase adjusted from the continuous phase 6SC and 1/2SC square wave signals that are applied to the circuitry by the reference clock generator circuitry 98. Moreover, the circuitry is also adapted to adjust the 1/2 cycle of 3SC that is present during the second playback of a picture frame that was detected in the reference clock generator circuitry 98 as previously discussed.

The chroma inversion enable signal that enables the chroma separator and processing circuitry 101 to invert the phase of chrominance of alternate frames of the receive television signal during playback operations is also generated by the circuitry 127 and is output over line 874 (FIG. 22).

The operation of the blanking insertion and bit muting circuitry 127 will now be described in connection with a block diagram shown in FIG. 22. The frame delay signal from the reference clock generator 98 is input on line 857 to one input of an exclusive OR gate 872, the other input of which is supplied by line 878 carrying the PALE flag signal received from the reference logic circuitry 125B. The output of the gate 872 appears on line 878' extending to steering logic 876.

The frame delay signal serves to invert the PALE flag signal at a picture frame rate. thereby superimposing a frame-to-frame 1/2 6SC clock period offset onto the PALE clock. which is used at the output of the blanking insertion and bit muting circuitry 127 and following digital-to-analog converter circuitry 102 to effect the repositioning of the final output video.

In order to ensure reliable repositioning of the video data and strobing of the data within the digital-to-analog converter 102 by the PALE digital-to-analog converter clock that is modified by the frame delay switch signal through the EXCL OR gate 872, the video data itself is selectively delayed by a 1/2 clock period, so that strobing of the data does not occur during a transition between bits. This is accomplished by the upper portion of the circuitry shown in FIG.

22 as follows. The video data from the chroma processing circuitry 101 is applied on lines 850 which extends to an 8 bit latch 851, the output of which appears on lines 852 which extend to another 8-bit latch 853 as well as to a 4 to 1 by 8 bit data multiplexer 854. The latches 851 and 853 are clocked by the continuous phase 6SC clock on line 855 and the output of the 8 bit latch 853 is also applied to the multiplexer 854 on lines 856. Each of the latches effectively clocks the data from lines 850 through with a 1/2 cycle of 3SC delay, so that the data appearing on line 852 is delayed 1/2 cycle of 3SC, whereas the data on line 856 has a full cycle of 3SC delay, by virtue of having been clocked through the two latches. While the same data is applied to the multiplexer 854 by lines 852 and 856, the data on line 856 is 1/2 cycle of 3SC delayed relative to the data on line 852.

The frame delay signal from the reference clock generator circuitry 98 on line 857 also extends to address control logic, indicated generally at 858, which controls the multiplexer 854 through lines 859. During alternate frames, the frame delay signal commands the address control logic to alternately pass the data from lines 852 and 856 to correct for the 1/2 cycle of 3SC offset that is present on the second playing of the picture frame as previously described.

When the black mute, or grey mute commands provided by the computer control system 92 via the signal system interface 119 are applied on lines 860 and 861 respectively, they are strobed into a latch 862 by the V drive (strobe 1) generated by the reference input circuit 93A and provided on line 862'. The latch 862 controls address control logic 858 in accordance with its stored command to cause the logic to provide the appropriate levels on lines 859 to insert the black or grey level digital information on lines 863 and 864, respectively, so that the black level or grey level data inserted in the video data stream appears on output lines 865. The black and grey levels are produced by setting switches 866 and 867 with the appropriate 8 bit word digitally defining the black or grey levels. When selective bits are to be muted, bit mute control lines 868 are applied to the multiplexer via line 869, assuming that gates 870 are enabled by a bit mute enable signal on line 871 that originates at the address control logic 858. Bit muting is inhibited during the blanking interval so as not to change the setup level of the video. The inhibiting is accomplished by the H and V gated blanking signal provided to the address control logic 858 by the D/A converter and sync insertion circuitry 102 and 103 over line 858'.

With respect to the generation of the PALE SC signal, the continuous phase 1/2 SC and 6SC inputs appear on lines 873 and 855, respectively, with the 1/2 SC signal being applied to a pulse former 875 that forms 1/2 SC pulses that extend to steering logic 876 via line 877. A PALE flag signal appearing on line 878 steers the 1/2 SC pulses to either the set (879) or reset (880) inputs of a divide by two divider 881 that is clocked by the 6SC signal on line 855. The output is a 3SC signal on line 882 that is changed in its phase by the 1/2 SC pulses appropriately steered by steering logic 876 in accordance with the level of the PALE flag signal on line 878.

The 6SC and 1/2 SC signals are also applied to a coarse burst phase circuit 884, the output of which appears on line 885 into a 6 bit shift register 886 that is clocked by 6SC and has 6 lines to permit the picking up of every 60 of burst phase and the application of the selected burst phase signal over a line 887 into a voltage variable capacitor network 888 which permits fine burst phase adjusting with control 889. The outlet is a SC square wave signal on line 890 that is applied to a limiter and filter 891 to produce a continuous sine wave SC signal on output line 892 for use in generating the burst for the composite analog television signal.

Specific circuitry that can be used to carry out the operation of the block diagram of FIG. 22 is illustrated by the detailed electrical schematic diagrams of FIGS. 51A and 51B. Since the operation of the circuitry shown in FIGS. 51A and 51B operates substantially as does the circuitry exemplified in the block diagram of FIG. 22, it will not be explained in detail.

However, with respect to the address control logic 858, it provides appropriate commands on lines 859, 871 and 874 to operate the blanking insertion and bit muting circuitry 127 to pass data to the following D/A converter and sync insertion circuitry 102 in accordance with the control inputs at lines 860, 861', 862' and 874'. The EE/PB signal provided by the encoder switch 126 over line 874 from control signals provided by the computer control system 92 is strobed into the latch 862 by the V drive signal on line 862'. When playback operations are performed, the latch 862 places the chroma invert enable command on line 874 which extends to enable two circuits.

One of the circuits is the chroma separator and processing circuitrv 101 as previously mentioned. The other is a NAND gate 857a in the frame delay switch line 857. The NAND gate 857a is enabled by the command to pass the frame delay switch to the address control logic 858 for use as previously described. During E to E operations, the chrominance of the video signal is not inverted and the previously mentioned frame to frame 46 nanosecond jitter does not occur in the video signal processed by the playback system 91 because a continuous four field color encoded television signal is provided to the electronics of the playback system 91. The EE/PB signal latched into latch 862 disables the NAND gate 857a and removes the chroma inversion enable signal status from line 874.

The address control logic 858 includes NAND gates 883a, 883b and 883c and a multiplexer 858a for directing the commands provided by NAND gates 883a and 883b onto the appropriate multiplexer control lines 859. NAND gate 883c inhibits bit muting during blanking for reasons discussed above and is provided with three inputs connected to receive the gated blanking signal over line 858' and the black and grey mute commands from the latch 862. Should any of these three functions become active, associated inputs of 883c will go low forcing line 871 high, disabling the bit muting circuitry. Consequently, NAND gate 883c provides a bit mute enable signal on line 871 except during blankmg intervals and grey and black mute operations.

NAND gates 883a and 883b have their inputs connected so that during normal playback operations NAND gate 883b provides a low level signal output and NAND gate 883a provides a high level signal output. The multiplexer 858a switches these output signals at the two lines 859 each frame in response to the frame delay switch signal 857 to cause the 4x1 multiplexer 854 to alternately pass the data received from the two latches 851 and 853 as previously described.

When a grey mute command is placed on line 861. latch 862 provides a low disabling signal to one of the inputs of the NAND gate 883c. thereby removing the bit mute enable signal from line 871. However, the inverter 861a inverts the low level provided by the latch 862, causing the output of the NAND gate 883a to be low. The multiplexer 858a activates lines 859 to cause the 4x1 multiplexer 854 to couple grey level digital information from lines 864 to the data output lines 865.

Black level mute operations are selected by the switch 860a being placed in a condition to couple the black mute command output of the latch 862 to one input of each of the NAND gates 883a. b and c. The black mute command causes all of these gates to issue high level signals. Hence, the bit mute enable signal is removed from line 871. Also. the multiplexer 858a activates lines 859 to cause the 4x1 multiplexer 854 to place black level digital information from lines 863 on data output lines 865.

The final playback processes that are performed in the signal system shown in the block diagram of FIGS. 9A and 9B involve the converting of the digitized video signals to an analog signal in a proper manner as well as generating and inserting the color burst and the compsite sync signals. However, before these processes are performed. the video data. delayed on alternate picture frames bv 1/2 cycle of 3SC and present at the output of data multiplexer 854 (FIG. 22) is clocked into a latch 901 (FIG. 23) with the PALE 3SC clock provided by the blanking insertion and bit muting circuitry 127 on line 902, which effects the reclocking to correctly reposition video data. The functions that are carried out will be described in connection with the block diagram of FIG. 23 which has the digitized video information on the eight bit lines 900 that extend from the circuitry 127 to latches 901. The latches serve to fix the repositioning of the video data to remove the aforementioned 46 nsec picture frame-to-picture frame jitter and. also. latch each of the bits on the bit lines to align them so that the digital-to-analog conversion can be made. The 3SC PALE clock generated bv the circuitry 127 misapplied on line 902 which clocks the latches 901 as well as the following timing circuits, including a second latch 903 and a resampling gate 904. The output of the latches C)()1 containing the digitized video information is clocked through output lines 905 into current switches 906 that have reference current generators connected thereto. The current switches 906 are connected via lines 907 to a resistor ladder network 908 that provides a weighted analog value of each eight bit digital word, thus providing an analog value of each eight bit digital word, thus providing an analog value having 256 possible levels.

The analog output signal from the ladder network appears on line 909 that splits into two paths. an upper path 910 and a lower path 911, the upper path 910 of which represents the normal path during which the video information is passed into a switch 912. The lower path 911 extends to a blanking filter 913 which is switched during the blanking time for the purpose of shaping the blanking pulse so it has the proper transition rate.

If the reshaping filter is not utilized, then the rapid video to blanking transition time can cause ringing in many television receivers. Accordingly, the output of the filter 913 appears on line 914 into the switch 912 which is controlled bv line 915 that comes from the latch 903 which is clocked by the 3SC PALE clock on line 902. During operation, the analog signal on line 909 extends through both paths 910 and 911 and the switch 912 is in the upper position passing the video information except during the blanking period, during which the switch 912 is switched to the lower position which connects to the resample gate 904 from the signal that has been filtered by the blanking filter 913.

The signal from the switch 912 appears on line 916 that is connected to the resampling gate 904 which operates to sample the level of the signal immediately before a level transition at a point where all transients from the previous transition have disappeared. For example. in the eight bit digital word, a change in value may result in up to seven or eight changes between logical states, i.e., from 1 to 0, each of which will produce a transient condition in the switch.

The resampling gate 904 provides a sample and hold operation while blocking the transients so that they do not affect the analog information that is present on line 917 that extends to the buffer and low pass filter 918.

The output of the low pass filter is connected to an amplifier and equalizer 919 via line 92() which performs a sine x/x roll off compensation. The compensated signal is applied to a black clipper circuit 921 which clips any luminance components of the video signal which appear below black level. The output 922 of the equalizer 919 is also part of a DC restoration loop comprising a switch 923, and a loop amplifier 924 which produces a feedback signal to the low pass filter. The switch 923 is controlled by a clamp pulse on line 925, effecting DC restoration of the video signal on line 922. The clamp pulse is contained in the blanking and composite sync signals provided on a pair of lines 933 by the reference input circuitry 93B.

The output of the black clipping circuit 921 appears on line 927 that extends to the sync and burst adder 928. Burst is added to the signal by line 929 and sync is added by line 930 so that a complete composite analog television signal appears on line 931 to output amplifiers 932. The sync signal is generated by a sync shaping circuit 934 that utilizes a sync pulse contained in the blanking and composite sync signals appearing on line 933, with the sync shaper providing the proper 140 nanosecond rise time and proper shaping. The burst is produced by a burst envelope generator 936 in response to a burst flag signal provided by the reference input circuitry 93B on line 935. The burst ag signal triggers the burst envelope generator 936 to modulate the SC sine wave generated in the bit muting and blanking insertion circuitry 127 previously described. The SC sine wave is coupled to a multipler 938 to be modulated by the output of the burst envelope generator 936 present on line 937. The output on line 929 contains the burst envelope with the 9 to 11 cycles of burst therein which are added in the sync/burst adder 928 to the analog video signal supplied on line 927.

One embodiment of specific circuitry that can be used to carry out the operation of the block diagram of FIG. 23 is shown in FIGS. 52A through 52D which operates in the manner as described with respect to the block diagram of FIG. 23 and therefore will not be described in detail. However, referring to FIGS. 52A and 52B, a blanking signal is applied to line 950 which extends to the latch 903 and produces an output which extends via lines 915 to a number of switching transistors 953. These, together with two transistors 954 and 955, respectively comprise the switch 912 that selects either the signal on the upper path 910 or on the lower path 914 from the filter 913. When blanking occurs, the transistors 953 effectively cut off transistor 954 while placing transistor 955 into conduction and during all other times, the reverse switching occurs.

With respect to the resampling gate 904, a clock appearing on line 902 extends to a number of inverters 957 and 958 which have the effect of providing a small amount of propagation delay to the signal so that the clock signal on line 902 that extends to transistors 961 and 959 are out of step with one another which has the effect of providing a positive transition in the primary side of a transformer 960, the secondary of which is connected to a diode bridge 904 that blocks signal flow during the period of the pulse to prohibit passing of the transients or spikes during switching of the digital-to-analog converter switches 906.

Fig. 24 shows a portion of the data detector and equalizer 99 of the record/playback channel, including a reproduce heat 1008 coupled to a preamplifier 1009, the combination of elements 1008 and 1009 being designated as block 1001. The magnetic flux patterns recorded on a disc surface are picked-up by the reproduce head 1008 and amplified by the preamplifier 1009. Due to the differentiating action of the reproduce head, which is well nown in the magnetic recording art, the output signal of block 1001 at terminal 1006 is a voltage proportional to the time-derivative of the recorded flux. Hence, the transfer function of block 1001 in the conventional symbolic notation of the Laplace transformation is G1 = kls (1) where G1 is a complex transfer function kl is a gain constant, and s is the complex Laplace variable.

Note: With respect to the above-indicated symbolic notations G; k; s; these will be maintained throughout the specification while only the indexes thereof will be changed, indicating specific circuits to which the notations pertain. In the following equations symbolic notations R,C with indexes attached thereto indicate respective resistance and capacitance values pertaining to corresponding circuit elements indicated by identical notations and indexes in the specification and drawings.

To the output of block 1001 of Fig. 24 an equalization circuit 1000 is coupled, the later circuit being shown in an idealized form suitable for theoretical explanation of the equalization operation which follows. The equalization circuit 1000 has an input terminal 1006, to which the output signal of block 1001 is fed. To the input terminal 1006, inputs of an integrating circuit 1002 and a differentiating circuit 1003 are coupled, respectively. The transfer function of the integrating circuit is G2 = k2/s (2) and the transfer function of the differentiating circuit is G3 = k3s (3) In the differentiating signal path, a variable gain control circuit 1004 is shown which enables to change linearly the high frequency boost effected by the differentiating circuit 1003, as it will be explained later in more detail. The difference of the respective output signals of the integrating and differentiating circuit is taken, as it is schematically shown by a subtraction circuit 1005. The resulting difference signal at output terminal 1007 of the equalization circuit 1000 is the required amplitude and phase-equalized signal with respect to the input signal at terminal 1006. The resulting record/playback channel has an overall flat amplitude response and linear phase response for all transmitted signal frequencies, as will be seen from the more detailed description below.

The overall transfer function of the portion of the record/playback channel shown in Fig.

24 comprising block 1001 and the equalization circuit 1000 coupled thereto is Govern = g1 (G2 - G3) (4) and after substituting for G1, G2 and G3 from (1), (2) and (3) Goveraii = kls (k2/s - k3s) = kirk2 (1 k s2) 2) (5) When substituting s = jw, the following is obtained GoVeral, (jw) = klk2 (1 + t W2) (6) The overall phase shift introduced by the portion of the record/playback channel shown in Fig. 24 is determined by phase of G(jw) = arctan YmeGiww (7) Since the expression on the right side of equation (6) is a real number (the imaginary part being zero), the overall phase shift determined by equation (7) is zero. At zero phase shift, the requirement of a linear phase response for all frequencies transmitted through the channel is satisfied.

It is essential for the equalization circuit to provide a difference signal at the output terminal 1007, rather than a sum of the respective output signals of the integrating and differentiating circuit. Each of the latter circuits introduces an equal phase shift of 90" but opposite in sense, lagging in the integrator and leading in the differentiator. Thus, the respective output signals of circuits 1002 and 1003 in Fig. 24 are out of phase by exactly 1800 with respect to each other and a difference signal yields a resulting signal combination, for which the respective signal amplitudes are added together, rather than subtracted from each other. Besides that, a -90 . Besides that, a -90" phase shift of the integrator output signal combined with the +90 phase shift of the differentiating action of the reproduce head yields an 0 overall phase shift. On the other hand, the to90" phase shift of the differentiator output signal combined with the +90 phase shift of the differentiating head yields a 1800 overall phase shift which is simply an inversion. Whether the resulting overall phase shift of the record/reproduce channel is 0 or 1800, that is, whether the output signal at the terminal 1007 is in phase or inverted with respect to the polarity of the recorded flux, depends on the sense of the 90" phase shift introduced by the equalizer 1000 as it will be described later in more detail.

Besides providing a linear phase response for all the frequencies transmitted through the channel, the equalization circuit also compensates for the non-constant amplitudefrequency response of the reproduce head, as it will be disclosed below. As it is well known in the art, the output voltage of the reproduce head 1008 and preamplifier 1009 combination of Fig. 24 rises at low frequencies at a rate of 6dB/octave, levels off at mid-band frequencies and falls at high frequencies. Such an amplitude response is shown as an example at GR in Fig. 27. Consequently, if an overall flat amplitude response of the record/playback channel is to be obtained, it is necessary for the equalizer to boost the amplitude at both low and high frequencies. This required equalizer characteristic is obtained by the circuit of Fig. 24 in a following manner. As an example, Fig. 28 shows a graph representing the gain G2 of the integrating circuit 1002 and the gain G3 of the differentiating circuit 1003 in dB, respectively, as dependent on frequency, the frequency values being plotted on a logarithmical scale. The characteristic G2 falls and the characteristic G3 rises with frequency at a rate of 6dB/octave. There are also shown diagrams of two other transfer functions G3, and G3" of the differentiating circuit, representing linear variation of these functions with variation of the gain control circuit 1004 output signal, as it will be described in more detail later. At GE a resulting transfer function of the equalization circuit 1000 is shown, obtained by adding the linear magnitudes G2 and G3. It can be seen that the transfer characteristic GE of the equalization circuit 1000 is complementary to the transfer characteristic GR of When comparing equation (11) with (3) we obtain k3 = -R2C2 (12) The subtraction circuit of Fig. 24 is implemented in the circuit of Fig. 25 by a differential amplifier 1005. The output of the inverting integrating circuit 1002 is coupled to an inverting input of the differential amplifier 1005 while the output of the inverting differentiating circuit 1003 is coupled to a non-inverting input of amplifier 1005. The output signal at terminal 1007 is the difference signal which also represents the equalized signal of the recording/reproducing channel. The resulting equalized signal has 0 phase difference with respect to the signal recorded on the magnetic medium, that is, it is in phase therewith.

Thus, the phase response of the overall channel becomes linear when the equalization circuit 1000 is utilized.

However, the circuit of Fig. 25 is still considered idealized to the extent that exact implementation of the above transfer functions (9) and (11) would require unlimited gain in the integrating amplifier circuit 1002 at low frequencies and in the differentiating amplifier circuit 1003 at high frequencies. In practical applications both these extremities are avoided, for example, by adding a shunt resistor R" to C1 and a series resistor R' to C2 as shown in Fig. 25, to truncate the respective integrating and differentiating approximations at selected frequencies below and above the frequency range of interest. Considering the presence of the respective resistors R', R" in the circuit of Fig. 25, the respective transfer functions G2, G3 will be

where RI, R2, R', R", C1 and C2 are component values pertaining to corresponding circuit elements.

When considering in equation (13) R"C1s 1 5 1 RlC (15) R"C1 We obtain G s2 (16) which is identical to the transfer function of (2).

When considering in equation (14) R'C2s((l 1 R'C2s(( 1 < s (( R'l (17) we obtain G3- k3s (18) which is identical to the transfer function of (3).

It follows from the above discussion that when substituting for s=jw, the respective transfer functions of the integrating and differentiating circuit of the equalization circuit 1000 shown in Fig. 25 will approach that of an ideal integrator and differentiator in the frequency range 1 1 R"CI ((w(( R'C2 (l9) In Fig. 26 still another example of practical implementation of the equalization circuit is shown. The integrating circuit of Fig. 24 is here implemented by a passive integrating network 1002 comprising series resistor RA and parallel capacitor CA followed by a non-inverting amplifier 1012 providing the necessary amplification in the integrating signal path. Analogously, the differentiating circuit of Fig. 24 is implemented in Fig. 26 by a passive differentiating network 1003 comprising a series capacitor CB and a parallel resistor followed by a non-inverting amplifier 1013 providing the necessary amplification in the differentiating signal path. Similarly as in the circuit of Fig. 25 the subtraction circuit is implemented by a differential amplifier 1005. In the circuit of Fig. 26 the integrated and subsequently amplified signal at the output of amplifier 1012 is fed to a non-inverting input of the differential amplifier 1005, while the differentiated and subsequently amplified signal at the output of amplifier 1013 is fed to an inverting input of amplifier 1005 The output signal at terminal 1007 of the circuit in Fig. 26 is the resulting difference signal which represents the equalized signal of the record/playback channel. The resulting equalized signal has a 0 phase difference with respect to the signal recorded on the magnetic disc.

Thus, the phase difference caused by the presently described equalization circuit does not introduce non-linearities in the phase response of the overall channel, but to the contrary, it yields an overall linear phase response.

The respective transfer functions of the integrating and differentiating circuit of Fig. 26 are

where A2 is the gain of amplifier 1012 and A3 is the gain of amplifier 1013. 1 When comparing equation (20) with (2) we obtain for w RACA w > > RACA k2 = (22) D d RACA when comparing equation (21) with (3) we obtain for w < < 1 RnCn k3 = A3RBCB (23) A potentiometer 1014 in Fig. 26 connected to the amplifier 1013 in the differentiated signal path represents a variable gain control circuit. By adjusting the gain A3 of amplifier 1013, the gain constant k3 expressed by (23) and the corner frequency of the boost changes as it has been described above in connection with the description of Fig. 28 and equation (8).

A detailed electrical circuit diagram of the data detector and equalizer 99 is shown as an example in consecutive Figs. 53A and 53B and will be now described. In the video frame storage recording and reproducing system, a color television signal is encoded in digital form and recorded on a magnetic disc. The digital code utilized is the DC free self clocking channel code described above with reference to FIG. 45. Upon playback, the digital data is reproduced by a reproduce head and amplified by a reproduce preamplifier 1009 (reproduce head and pre-amplifier are shown in Fig. 54B). Figs. 53A and 53B show two identical playback equalizer and data detector circuits utilized for the ten separate data streams received from the disc drive data interface 151. However, only one of these circuits will be described. In the circuit of Figs. 53A and 53B the preamplified playback data in the channel encoded format, for example, of the type described in the foregoing, is equalized by an equalization circuit 1000 corresponding to the above-described equalization circuit with reference to Figs. 24 to 26. The equalized signal is filtered in a low pass filter circuit 1018, and thereafter amplified and amplitude limited to produce a rectangular pulse sequence in an amplifier-limiter circuit 1019. The pulse sequence from the limiter is fed through a pulse former circuit 1020 which forms output pulses for each detected signal transition. The pulses from circuit 1020 are fed to the data decoder and time base corrector circuitry 100 which decodes and removes timing errors from the playback data from which the original color television signal is recovered.

As shown in Figs. 53A and 53B, the playback data from the preamplifier is applied to differential input terminals 1021 and 1022 of a differential amplifier 1033, such as manufactured by RCA, type CA 3004. This type of amplifier contains open-collector differential output transistors connected to output terminals 1034 and 1035. Resistor 1036 is the load resistor for the non-inverting output terminal 1034. The gain of the amplifier 1033 to output terminal 1034 is constant throughout the frequency range of interest. The non-inverted signal is buffered by emitter follower 1037 and then applied ta a differentiating network 1003 comprising capacitor 1038 and resistor 1039. This network 1003 performs differentiation for signal frequencies below 60 MHz. Its transfer function is G3 - (R1039)(C1038)s 1 + (R1039)(C1038)s (22) and for w ( (R1039)(c1038) G3 - (R1039)(C1038) 5 (23) Equation (23) corresponds to previously discussed equation (3) related to the block diagram of Fig. 24 where k3 = (1039) (1038). Since signals of interest in this particular embodiment extend only to about 10 MHz, this network 1003 may be viewed as a true differentiator. The output of the differentiator 1003 is applied to input terminal 1040 of differential amplifier-multiplier circuit 1041, such as manufactured by Motorola, type MC1496. Input terminals 1040 and 1042 of the circuit 1041 are differential input terminals biased by connection to +7.5V. The amplifier-multiplier 1041 receives a second input signal at differential input terminals 1043 and 1044 and at output terminal 1045 an output current is provided proprtional to the negative of the product of the input signals at terminals 1040, 1042 and 1043, 1044. In the present circuit a direct current gain control voltage is applied to input terminal 1043, while terminal 1044 is grounded. The control voltage at 1043 corresponds to an output voltage from a remote variable gain control circuit (not shown on FIG. 53), such as previously described in connection with circuit 1014 of Fig. 26. In the presently described embodiment of the frequency equalizer, the gain of the circuit 1041 in the differentiated signal path is remotely and automatically controlled by a digital-to-analog converter to obtain desired gain variations dependent on the variations of the recording track length of the magnetic disc. A particular track number (corresponding to a specific track length) from which a particular data is being reproduced is decoded in a digital decoder and converted in the digital-analog converter to a direct current voltage level which is then applied as a gain control signal to input terminal 1043 of circuit 1041. As it has been mentioned before, the variable gain adjustment in the differentiated signal path is designed to compensate for higher pulse density on inner tracks of the disc while linearity of the high frequency boost of the equalized signal is maintained for the entire frequency band transmitted.

The magnitude of the current at output terminal 1045 of the amplifier-multiplier circuit 1041 is proportional to the input signal at input terminal 1040 and to the gain value determined by the control voltage at terminal 1043. The output current from terminal 1045 of the circuit 1041 is applied as an input current to the emitter of a common-base transistor amplifier serving as the subtraction circuit 1005 which has been previously described and shown in Figs. 24, 25 and 26. This input current produces an output voltage at the collector of the amplifier which is proportional to both the input current and resistance of a collector load resistor 1047. Thus, the above-indicated part of the transistor 1005 output voltage is proportional to the negative of the signal derivative amplified by the amplifier-multipler circuit 1041.

The inverting output terminal 1035 of the differential amplifier 1033 has a load resistor 1048 and a parallel load capacitor 1049. The direct current gain of the amplifier 1033 to output terminal 1035 is higher than the gain to the non-inverting output terminal 1034 by the ratio of the respective load resistances R1048/R1036, that is, by the factor of about 3.

For signal frequencies above 80 KHz the gain to output terminal 1035 is determined by C1049 and is inversely proportional to the frequency. Thus, the output circuit R1()48 C1049, connected to terminal 1035 functions as an integrating network for frequencies above 80 KHz and throughout the frequency range of interest which is approximately from 0.3MHz to 10 MHz. The transfer function of the amplifier 1033 to the output terminal 1035 is G1033 = - 3A1033 f 1 R1048) (C1049)s+1 (24) where A1033 is the gain of the differential amplifier 1033 to output terminal 1034.

For w ) 1 (R1048) (C1049) G1033~ - 3Alo33 1 (R1048)(C1049)s (25) Equation (25) corresponds to previously discussed equation (2) related to the block diagram of Fig. 24, where k2 = -3A1033 (R1048)(C1049) The inverted and subsequently integrated signal from the output terminal 1035 of amplifier .1033 is applied to the common emitter transistor amplifier 1005 Transistor 1005 inverts this input signal and multiplies it by the ratio of its respective collector and emitter load resistances R1047/R1050. The transistor 1005 operates as a common emitter amplifier in the integrating signal path and as a common base amplifier in the differentiating signal path. The resulting output signal at the collector of transistor 1005 is the sum of two input signal contributions, one proportional to the integral of the playback signal from the reproduce head and preamplifier combination, the other one proportional to the negative of the derivative of the playback signal. Thus, the resulting output signal at the collector of transistor 1005 corresponds to a difference signal, such as previously described with reference to the output signal at the output terminal 1007 of the previously described embodiments of the equalization circuit shown in FIGS. 24, 25 and 26. Thus, the output signal of the equalization circuit 1000 of FIGS. 53A and 53B corresponds to the equalized signal of the record/playback channel as previously disclosed with respect to the embodiments of FIGS. 24, 25 and 26.

Now the remaining part of the detailed circuit diagram shown in FIGS. 53A and 53B will be described. The equalizer 1000 converts the voltage peaks of the playback signal provided by the playback preamplifier 1009 (FIG. 54B), which represent zero crossings of the recorded flux, back into properly positioned zero crossings at the output of the equalizer.

This equalized output signal is present at the collector of transistor 1005 of the equalizer and is filtered by a low pass filter circuit 1018 and thereafter fed through a first buffer amplifier 1051, such as type MC10116P and arranged to provide complementary outputs, of an amplifier-limiter circuit 1019. The output signal from the buffer amplifier is fed through a series of five amplitude-limiting amplifiers, preferably of the same type as the buffer amplifier. The equalized playback signal provided at the input of the amplitude-limiting circuit 1019 is in the channel encoded form with the transitions properly positioned.

Amplitude limiting the playback signal serves to restore the rectangular shape of the playback data signal which has been considerably distorted by the record and reproduce processes. Furthermore, the buffer amplifiers of the amplitude limiting circuit 1019 also serve to provide opposite phased waveforms of the restored data signal which are subsequently used to generate a pulse for each transition of the rectangularly shaped channel encoded playback data signal. As previously described herein with reference to the channel encoding of the data signals by the encoder 96 and subsequent recording of such signals, the transition-related pulses are generated so that a precisely defined edge, the leading edge being selected in this embodiment, can be sent through a transmission channel without introducing errors to the data although the data signal may be distorted by the channel. As described hereinbefore, the high bit rate data streams, such as processed by the apparatus described herein, are particularly susceptible to the introduction of error because of the differential response to different sensed transitions of transmission lines such as those which couple channel encoded data between disc drives and the signal system.

To generate a pulse for each transition of the playback data signal so that only leading, positive edges of the pulses identify the data signal transitions, the amplifier-limiter circuit 1019 provides two opposite phased waveforms of the data signal. First, a sequence of transitions between signal levels of non-inverted polarity is provided at the output terminal 1052 of the last amplifier 1053 of the series of amplitude-limiting amplifiers and second, an identical sequence of inverted polarity is provided at the output terminal 1054 of the same amplifier 1053. Both these sequences have their transitions positioned according to the code rules of the channel code selected for originally encoding the video data and are applied respectively to clock two identical one-shot multivibrators 1055 and 1056, such as type M(v10131L of the pulse former circuit 1020. Each multivibrator forms a positive pulse, respectively, for each positive going transition of the playback data signal received at its clock input. Consequently, the one-shot multivibrator 1055 receiving the non-inverted form of the playback data signal provides a positive pulse at each positive going transition in the data signal. On the other hand, the other one-shot mutlivibrator 1056 receiving the inverted form of the playback data signal provides a positive pulse at the location of each negative going transition in the data signal. Since the leading edges of the positive pulses generated by the multivibrators 1055 and 1056 are defined by rapidly switching the multivibrators from its stable state to its quasi-stable state (there being no significant time constant determining components involved), each leading edge will be identical to all others and occur at a precise time following the occurrence of the positive clocking transition of the playback data signal. Because the transmission channel over which the pulses are sent will act on identical pulse edges the same, the locations of the transition-related positive pulse edges, hence, data signal transitions themselves, are not lost as a result of any distortion that may be introduced to the pulses by the action of the transmission channel. If necessary, an amplitude level sensitive detector means can be coupled to the output of the transmission channel, such as is used at the input of the decoder circuitry portion of the previously described decoder and time base corrector 100, to accurately redefine the relative locations of the playback data signal transitions.

For transmission of the transition related pulses to the signal system, the output pulses of both one shot multi-vibrators 1055 and 1056 are applied to separate inputs of a positive OR-gate 1057 which forms an output pulse for each input pulse. The output pulses of the OR-gate 1057 are applied to the disc drive data interface 151 (FIG. 9B) for transmission over lines 154 to the data select switch 128, which couples the transmitted pulses to the input of the data decoder portion of the decoder and time base corrector circuitry 100 of the selected playback channel 91 for decoding of the playback data and subsequent processing to recover the original color television signal.

The disc drive interface 151 includes a conventional complementary output buffer amplifier arranged to receive a single input signal and generate coincident complementary output signal forms of the single input signal. The complementary buffer amplifier converts each transition related pulse provided by the OR-gate 1057 to a pair of coincident complementary level pulses, which are coupled to the data select switch 128 for transmission to the selected playback channel 91.

FIGS. 54A and 54B show consecutive parts of a detailed electrical circuit diagram including the record driver and playback preamplifier circuits of four identical data record and playback channels, designated 1058, 1059, 1060 and 1061 utilized in the video frame storage record and playback system. A fifth channel designated 1062 includes a servo track head permanently connected to a servo playback preamplifier and it also includes a data track record and playback channel. In the video frame storage record and playback system, five more data record and playback channels (not shown) identical with the above-indicated data record and playback channels shown in FIGS. 54A and 54B are utilized. A relay 1063 in channel 1058 is shown having its contacts in a position connecting one of the heads 1008a and 1008b for recording as occurs when a record command is received from the disc drive control circuitry on line 1066, as described hereinbefore. In absence of a record command, the relay 1063 is in the playback position. In this position, the contacts of relay 1063 are in their alternative positions. Heads 1008a and 1008b are utilized for both recording and playback and are switched alternatively for odd and even television fields. Switching of these heads 1008a and 1088b is controlled by the 30 Hz head switch signal continuously provided on line 1067 provided by the record timing circuit of FIG. 38A located in the disc drive electronics. The playback data received alternately from the heads 1008a of the respective channels 1058, 1059, 1060 and 1061 is fed into the playback equalizer and data detector circuits associated with the respective channels such as shown in previously described FIGS. 53A and 53B. The record/playbaek heads utilized in the video frame storage recording and reproducing system are conventional heads such as manufactured by Applied Magnetic Corporation or Information Magnetics Corporation, for digital recording on disc packs of the kind employed in the apparatus.

WHAT WE CLAIM IS: 1. Apparatus arranged for inserting a digital synchronization word in a continuously clocked digitally encoded data stream comprising at least one digital component of a video signal having a horizontal blanking interval separating successive lines of video information and a vertical blanking interval separating successive fields of video information, said horizontal blanking interval having horizontal synchronization signals removed therefrom, said apparatus having an input for receiving said data stream and an output upon which said data stream selectively appears with said synchronization word inserted therein, said

**WARNING** end of DESC field may overlap start of CLMS **.

Claims

**WARNING** start of CLMS field may overlap end of DESC **. respectively to clock two identical one-shot multivibrators 1055 and 1056, such as type M(v10131L of the pulse former circuit 1020. Each multivibrator forms a positive pulse, respectively, for each positive going transition of the playback data signal received at its clock input. Consequently, the one-shot multivibrator 1055 receiving the non-inverted form of the playback data signal provides a positive pulse at each positive going transition in the data signal. On the other hand, the other one-shot mutlivibrator 1056 receiving the inverted form of the playback data signal provides a positive pulse at the location of each negative going transition in the data signal. Since the leading edges of the positive pulses generated by the multivibrators 1055 and 1056 are defined by rapidly switching the multivibrators from its stable state to its quasi-stable state (there being no significant time constant determining components involved), each leading edge will be identical to all others and occur at a precise time following the occurrence of the positive clocking transition of the playback data signal. Because the transmission channel over which the pulses are sent will act on identical pulse edges the same, the locations of the transition-related positive pulse edges, hence, data signal transitions themselves, are not lost as a result of any distortion that may be introduced to the pulses by the action of the transmission channel. If necessary, an amplitude level sensitive detector means can be coupled to the output of the transmission channel, such as is used at the input of the decoder circuitry portion of the previously described decoder and time base corrector 100, to accurately redefine the relative locations of the playback data signal transitions. For transmission of the transition related pulses to the signal system, the output pulses of both one shot multi-vibrators 1055 and 1056 are applied to separate inputs of a positive OR-gate 1057 which forms an output pulse for each input pulse. The output pulses of the OR-gate 1057 are applied to the disc drive data interface 151 (FIG. 9B) for transmission over lines 154 to the data select switch 128, which couples the transmitted pulses to the input of the data decoder portion of the decoder and time base corrector circuitry 100 of the selected playback channel 91 for decoding of the playback data and subsequent processing to recover the original color television signal. The disc drive interface 151 includes a conventional complementary output buffer amplifier arranged to receive a single input signal and generate coincident complementary output signal forms of the single input signal. The complementary buffer amplifier converts each transition related pulse provided by the OR-gate 1057 to a pair of coincident complementary level pulses, which are coupled to the data select switch 128 for transmission to the selected playback channel 91. FIGS. 54A and 54B show consecutive parts of a detailed electrical circuit diagram including the record driver and playback preamplifier circuits of four identical data record and playback channels, designated 1058, 1059, 1060 and 1061 utilized in the video frame storage record and playback system. A fifth channel designated 1062 includes a servo track head permanently connected to a servo playback preamplifier and it also includes a data track record and playback channel. In the video frame storage record and playback system, five more data record and playback channels (not shown) identical with the above-indicated data record and playback channels shown in FIGS. 54A and 54B are utilized. A relay 1063 in channel 1058 is shown having its contacts in a position connecting one of the heads 1008a and 1008b for recording as occurs when a record command is received from the disc drive control circuitry on line 1066, as described hereinbefore. In absence of a record command, the relay 1063 is in the playback position. In this position, the contacts of relay 1063 are in their alternative positions. Heads 1008a and 1008b are utilized for both recording and playback and are switched alternatively for odd and even television fields. Switching of these heads 1008a and 1088b is controlled by the 30 Hz head switch signal continuously provided on line 1067 provided by the record timing circuit of FIG. 38A located in the disc drive electronics. The playback data received alternately from the heads 1008a of the respective channels 1058, 1059, 1060 and 1061 is fed into the playback equalizer and data detector circuits associated with the respective channels such as shown in previously described FIGS. 53A and 53B. The record/playbaek heads utilized in the video frame storage recording and reproducing system are conventional heads such as manufactured by Applied Magnetic Corporation or Information Magnetics Corporation, for digital recording on disc packs of the kind employed in the apparatus. WHAT WE CLAIM IS:

1. Apparatus arranged for inserting a digital synchronization word in a continuously clocked digitally encoded data stream comprising at least one digital component of a video signal having a horizontal blanking interval separating successive lines of video information and a vertical blanking interval separating successive fields of video information, said horizontal blanking interval having horizontal synchronization signals removed therefrom, said apparatus having an input for receiving said data stream and an output upon which said data stream selectively appears with said synchronization word inserted therein, said

apparatus including means for generating said digital synchronization word; means for selectively switching said data stream so that said data stream is blocked from said output, the information contained in said data stream being discarded when said switching means is so switched, and means for operating said switching means so that said data stream is blocked from appearing at said output at least during alternate ones of successive horizontal blanking intervals, these being provided means for controlling said synchronization word generating means for activating the same to inject said synchronization word onto said output during alternate ones of successive horizontal blanking intervals, so that said synchronization word appears in said data stream on alternate successive horizontal blanking intervals throughout each field of video information.

2. Apparatus according to claim 1 wherein said switching means blocks the data stream from appearing at the output during the vertical blanking intervals and the synchronization word is injected onto the output during those intervals.

3. Apparatus according to any foregoing claim wherein at least two data streams are present, said apparatus being arranged to insert said synchronization word in each of said data streams.

4. A machine for recording a still image, the machine including means for converting an analogue television signal into a multiplicity of digitally encoded data streams, means for recording said streams on respective magnetic surfaces simultaneously, and apparatus according to any foregoing claim for inserting digital synchronization words in the data streams.