GB1594989A - Phase shifting microstrip transmission lines - Google Patents

Description

(54) PHASE SHIFTING MICROSTRIP TRANSMISSION LINES (71) We, HAZELTINE CORPORA TION, a corporation organised and existing under the laws of the State of Delaware, United States of America, of Greenlawn, New York 11740, United States of America, do hereby declare the invention, for which we pray that a patent may be granted to us and the method by which it is to be performed, to be particularly described in and by the following statement: This invention relates to phase adjustable microstrip transmission lines.

Our prior British Patent No. 1538565 (7461/76) discloses an array antenna system wherein a coupling network interconnects groups of array antenna elements. Wave energy signals supplied at the input of any element group are coupled directly to the elements of that group and are also supplied through the coupling network to selected elements in the remaining element groups of the array. As a result, the array aperture is provided with an excitation, which closely approximates an ideal excitation to produce an effective element pattern wherein substantial radiation occurs only in a desired region of space.

Figure 16 of the prior application discloses a technique for shifting the angular location of the effective element pattern of the array by providing phase shifters, which may be switchable, between the antenna elements and the coupling networks. As illustrated in Figure 15 of that prior application, the effective element pattern can be displaced, for example, to one side of the broadside axis of the array. This prior technique for shifting the effective element pattern also angularly shifts the radiated array pattern by the same amount, since the phase adjustments are provided immediately adjacent to the radiating elements. As a result, if the phase adjustments illustrated in Figure 16 of the prior application are utilized in an array antenna such as shown in Figure 6 of that application, both the antenna element pattern and main beam of the antenna are shifted in space. If phase shifters 13 of the antenna are set to radiate a beam in the broadside direction, the phase adjusting line lengths 75 will cause a shift in the direction of the antenna beam off the broadside axis by the same angular displacement as is given element pattern 77.

A similar effect results when the phase adjustment line lengths 75 are provided in an antenna having an input commutation switch, such as is shown in Figure 7 of the prior application. In this case, the antenna radiates a pattern wherein the radiated frequency varies as a function of angle from the broadside axis of the array. The phase adjustments 75 will shift not only the effective element pattern, but also the frequency coding of the radiated signal.

Figure 2 illustrates a microwave landing system environment wherein the present invention is particularly useful. A navigation antenna 52 of the type described in the referenced prior application is located adjacent an airport runway 54. Near the approach of runway 54, there is located uneven terrain 56. When an aircraft 58 is approaching runway 54, it may receive a signal 66 directly from antenna 52, and may also receive a signal 64 which has been reflected off the uneven terrain 56. In such an installation, it is particularly desirable to shift the location of the effective element pattern 60 of antenna 52 such that the radiation in the angular direction of the uneven terrain 56 is reduced, thereby to reduce navigation error resulting from multipath signal 64. In the event angular shifting of element pattern 60 is achieved by the method shown in Figure 16 of the prior application, there will also be a shift in the direction of the antenna beam 62. If antenna 52 is used in a "scanning beam" landing system wherein a narrow antenna beam is moved through space at regular time intervals, the shift of antenna beam 62 will be manifested by an angular change in the direction of the antenna beam at any particular instant of time. In the event antenna 52 is used in a "Doppler" landing system, making use of a commutator arrangement such as shown in Figure 7 of the prior application, antenna beam 62 represents the signal which is detected by a narrow bandwidth receiver, since antenna 52 radiates into the entire angular region defined by element pattern 60 with a radiation pattern wherein radiated frequency varies with angular direction. In a Doppler system, the prior art pattern shifting technique will result in a change in the angular frequency coding, thereby causing a frequency change in the radiated signal at any particular angle.

Since the prior art technique of changing the angular position of the effective element pattern results in a change in the frequency or time coding of the radiated signal, such a modification to the antenna system to accommodate uneven terrain at a particular installation location results in additional complexity in the navigation equipment.

Either the receiver in aircraft 58 must be advised of, and perform a correction calculation for, the resulting change in navigation coding or the coding mechanism of antenna 52 must be adjusted to correct for the change in the frequency or time coding of the radiated signal.

Another problem with the prior art technique of providing a phase shift adjustment at the inputs of the particular antenna elements is that such a phase adjustment eliminates the possibility of having uniform antenna element groups, each group consisting of elements, power divider, interconnecting transmission lines, couplers, and interconnecting networks, which could be produced as a modular unit. The element pattern steering technique of the prior application required different phase adjustment for each element. This eliminated the possibility of uniform modular construction.

Further, the amount of phase adjustment could be very large for large array.

According to the present invention there is provided a phase shifting microstrip two conductor transmission line comprising a dielectric substrate, a ground plane and a conductive strip, said ground plane and said conductive strip forming said two conductors of the transmission line and being attached to opposite sides of said dielectric substrate, and a conductive plate located at the side of said strip away from said substrate and separated from said strip by a selected spacing whereby in use a wave signal propagating along said line has its field affected by said plate to shift the phase of said signal.

Embodiments of the invention will now be described, by way of example, with reference to the accompanying drawings, in which: Figure I is a schematic diagram of an antenna system in accordance with the invention the subject of the parent Patent Application.

Figure 2 illustrates a microwave landing system installation using the Figure 1 antenna.

Figure 3 is a graph showing the element pattern and array pattern of a prior art antenna.

Figure 4 is a graph showing the element pattern and array pattern of the Figure 1 antenna.

Figure 5 is a graph illustrating the amplitude of the element aperture excitation in the Figure 1 antenna.

Figure 6 is a graph illustrating the phase of the element aperture excitation of the Figure 1 antenna.

Figure 7 is a cross-sectional perspective view of a microstrip transmission line.

Figure 8 is a cross-sectional view of the Figure 7 transmission line.

Figure 9 is a cross-sectional view of a phase adjustable transmission line in accordance with the invention.

Figure 10 is a planar view of the transmission line of Figure 9.

Figure 11 is a graph showing phase as a function of separation (d) for the Figure 9 transmission line.

Figure 1 is a schematic diagram of an antenna system, which closely corresponds to the schematic diagram of Figure 6 in the above-referenced prior application. The Figure 1 antenna includes a plurality of element groups with their associated coupling networks. Each element group 20 of the antenna system includes two antenna elements 21 and 23 which are connected to an element group input terminal 27 by hybrid power divider 22 and transmission lines 24 and 26. The difference terminal of hybrid 22 is terminated in a resistor 25.

Transmission lines 24 and 26 interconnect the colinear terminals of hybrid 22 with elements 21 and 23, respectively.

In accordance with the teachings of the prior application, transmission lines 24 and 26 of each of element groups 20 are inter- connected by coupling means comprising transmission lines 28 and 30. Transmission line 28 is coupled within each group 20 to transmission line 26 by coupler 34. Transmission line 30 is similarly coupled within each group 20 to transmission line 24 by coupler 32. Also in accordance with the teachings of the prior application, the ends of transmission lines 28 and 30 are terminated in resistors 46. The transmission lines include resistive loads 36 and 38 which are arranged between the points at which transmission lines 28 and 30 are coupled to transmission lines 24 and 26 in each of the adjacent element groups 20.

In accordance with the understanding of the prior application, hybrid power divider 22 and its associated output transmission lines 24 and 26 comprise a first coupling means, one for each element group 20, for coupling wave energy signals supplied at the input 27 to antenna elements 21 and 23 of each group 20. Also in accordance with the prior application, transmission lines 28 and 30 comprise second coupling means interconnecting the first coupling means so that signals supplied at the input 27 to any of the first coupling means are also supplied to selected elements in the remaining element groups of the array.

Alternate networks for coupling wave energy signals to the array are shown in Figures 6 and 7 of the prior application. The network shown in Figure 1, comprising oscillator 50, power divider 48, and phase shifters 44 corresponds to the network shown in Figure 6 of the prior application.

The network shown in Figure 7 of the prior application includes an oscillator and a commutating switch for sequentially supplying wave energy signals to the inputs 27 of the element groups 20. The present invention is equally applicable to each of these alternative networks, which provide either radiation of a scanning narrow antenna beam or a broad radiation pattern wherein the frequency of radiation varies as a function of angular direction with respect to the array of antenna elements.

One object of the parent invention is to provide a spacial movement of the effective element pattern associated with each of the inputs 27 to the antenna groups of the Figure 1 antenna system. Accordingly, there are provided in the Figure 1 system phase shifters 40, 42 and 100 in transmission lines 28, 30 and 26 associated with each of the element groups 20. In accordance with that invention, the phase shifters in the transmission line 28 are of opposite sense to those in transmission lines 30 and 26. The selection of which phase shifters will be positive is in accordance with the desired direction of element pattern shift. In the drawing of Figure 1, phase shifters 40, 42 and 100 are schematically illustrated as additional lengths of transmission line, but it should be understood that this can represent either a positive, or a negative phase shift. In order to illustrate the operation of the parent invention, it will be assumed that phase shifter 40 is negative, that is decreased transmission line length, while shifters 42 and 100 are positive. The magnitudes of shifters 40 and 42 are equal and twice that of phase shifter 100.

In accordance with the prior application, wave energy signals supplied to the input 27c causes the antenna aperture to have the amplitude excitation 70 illustrated in Figure 5, which approximates the ideal amplitude excitation 72, also shown in Figure 5. In accordance with the prior application, transmission lines 28 and 30 have a transmission line length which is an odd multiple of a halfwave between couplers 32 and 34 in adjacent element groups. The effect of this selected transmission line length is to provide a 1800 shift in the phase of wave energy signals coupled to elements in alternate element groups.

Without phase shifter 100 signals supplied to the input 27c are supplied with equal amplitude and phase to elements 21c and 23c. A portion of the signal is also coupled from transmission line 26c onto transmission line 28 in an upward going direction in Figure 1. The signal on transmission line 28 is coupled with reduced amplitude to element 23b. Without phase shifter 40, the signal supplied to element 23b has the same phase as the signal supplied to elements 21c and 23c, since the 1800C phase shift of transmission line 28 between groups 20c and 20b is effectively removed by the 90" phase shift of each of the couplers 34 through which the signal passes to reach element 23b.

The signal on transmission line 28 is also coupled to element 23a. Without phase shifter 40, there is an additional 1800 phase shift on transmission line 28 between module 20b and 20a, and the signal at element 23a will be 1800 out of phase with the signals at elements 23b, 21c, and 23c. This phase relation is indicated by negative polarity of the excitation signal in Figure 5.

Signals in transmission line 24c are similarly coupled by transmission line 30 to elements 21d and 21e to complete the opposite side of the aperture excitation illustrated in Figure 5.

In accordance with the parent invention, it is desired that the effective element excitation illustrated in Figure 5 be provided with the same linear phase variation along the aperture. It is also desired that this phase variation be provided in a manner which maintains the same absolute phase of the array excitation which is formed from the composite of the signals provided at the various inputs 27. Phase shifters 40,42 and 100, see Figure 1, provide the necessary linear phase variation of the element aperture excitation without affecting the composite excitation in any other way, and therefore provide an angular shifting of the element pattern without changing the phase characteristics of the composite pattern resulting from the combination of all the excitations provided to the inputs 27. As a result, if the antenna system is used in a scanning beam operation, the direction of the main beam is unchanged, but the amplitude of the main beam is modified for any angular direction in accordance with the change in the element pattern in that direction. Likewise, if the antenna is one which radiates a frequency coded pattern, the frequency coding remains unchanged, but the amplitude of radiation in any particular direction is modified in accordance with changes in the element pattern. Since phase shifters 40 are negative, corresponding to decreased line lengths 6 between corresponding portions of groups 20, the phase at elements 23b and 23a will lead the phase at elements 23c by 6 and 26, respectively.

Since the phase shifters 42 in transmission line lengths 6, the phase at elements 21d and 21e will lag the phase at element 21c by 6 and 26, respectively. The result will be an element pattern shift in the + 0 direction shown in Figure 1. Phase shifter 100 provides an appropriate 6/2 phase adjustment between elements 21c and 23c. The resulting phase of the aperture excitation 70 is illustrated in Figure 6 and is an exact linear phase slope 74. Each of the phase shifters 40 and 42 has magnitude 6, which is twice that of shifter 100 and the slope of line 74 therefore corresponds to a phase variation of 6 for each distance S along the array, which corresponds to the spacings of element groups 20. Those skilled in the art can easily compute the required value of 6 in accordance with the desired angular movement of the antenna element pattern. When pattern shape requirements are not critical phase shifter 100 may be dispensed with while maintaining an approximation to the linear phase slope.

A typical element pattern movement is shown in Figures 3 and 4. The figures show the element pattern 68 which is a function of the angle 0 from the broadside axis 67 of the array. An angular region 69 corresponding to elevation angle 0 is shown. Within angular region 69, there may be structures or terrain which will cause undesired multipath signals. The composite array pattern for the directional beam antenna shown in Figure 1 is illustrated by narrow beam pattern 71. In accordance with the understanding of those skilled in the art, the relative amplitude of pattern 71 at any particular angle corresponds to the amplitude of element pattern 68. Figure 4 illustrates the effect of phase shifter 40,42 and 100 on element pattern 68. The element pattern has been removed by a desired amount in the positive direction of angle 0 so that the amplitude of element pattern 68' is substantially reduced in the region 69 between broadside axis 67 and angle 0.

This shifting of the element pattern does not affect the angular location of array pattern 71, but merely reduces the amplitude of pattern 71 when scanned to region 69 wherein multipath radiation may occur.

When the antenna system is used to radiate a frequency coded pattern, phase shifters 40, 42 and 100 likewise cause an angular shift in the radiated amplitude pattern without affecting the angularfrequency coding. Those skilled in the art will recognize that the present invention can be used to advantage in any of the alternate antenna network configurations shown in Figures 10, 13, and 14 of the referenced prior application.

Microstrip embodiment The coupling networks of the Figure 1 antenna, particularly interconnecting transmission lines 28 and 30, are advantageously formed using microstrip transmission line which is shown in Figure 7. This transmission line includes a ground plane 76 over which there is a slab 78 of dielectric material. On the opposite side of dielectric slab 78 from ground plane 76, there is provided a conductive strip 80. Typically, ground plane 76 is a thin copper cladding on dielectric 78 and strip 80 is the remains of a similar cladding which had been largely removed by photoetching. Strip 80 and ground plane 76 form a two conductor transmission line whose impedance is determined by the thickness (t) and dielectric constant(k) of slab 78 and the width(w) of conductive strip 80. A typical 50 ohm transmission line may be formed using Teflon-glass dielectric with a (k) of 2.2, a thickness (t) of 0.020.inches and having a conductive strip with a width (w) of 0.050 inches. Teflon is a Registered Trade Mark. Figure 8 is a cross-sectional view of the transmission line shown in Figure 7 and illustrates the electric fields associated with a typical wave energy signal.

A small fringing portion of the field 82 passes through the air adjacent the conductive strip before entering the dielectric material.

The inventor has discovered that by providing a structure that acts upon and alters the fringing electric field 82, it is possible to shift the phase of wave energy signals on the microscope transmission line. In accordance with the invention, both positive and negative phase shifts can be achieved depending on the type of field affecting structure used.

The cross-sectional view of Figure 9 shows a field altering structure comprising conductive plate 84 which is arranged to be spaced a distance (d) from conductive strip 80. In order to accurately regulate spacing (d), conductive plate 84 has a cross-sectional configuration which includes a groove whose depth is selected in accordance with the required spacing (d). Screws 85 are provided to electrically connect conductive plate 84 to ground plane 76 of the transmission line.

Those skilled in the art will recognize that conductive plate 84 will draw some of the electric field emanating from conductive strip 80 through the region of air formed by the spacing (d) between conductive strip 80 and conductive plate 84. Since a major portion of the electric field will then be passing through air dielectric, the effective dielectric constant, and hence the propagation constant of the microstrip transmission line will be lower. It will also be recognized that as conductive plate 84 is arranged closer to conductive strip 80, the phase shifting effect will be increased. Figure 11 is a graph showing an estimate of the phase shift at 5 GHz which might be realized by a conductive plate of the type shown in Figure 9 with a length (L) of a half wave at the propagation constant of the transmission line. Figure 10 is a planar view of such a conductive plate indicating the location of grounding screws 85 and the length (L) of the conductive plate.

It will be evident to those familiar with such transmission lines that it is advantageous to select the length (L) of the field altering structure to be equal to an integral number of half wave length(s), so that the signal reflections occurring at each end of the field altering structure will be approximately self-cancelling.

Those familiar with microwave circuits will recognize that the phase shifting structure of Figures 9 and 10 may be used in circuits other than that shown in Figure 1.

The structures are advantageously used in complex microstrip networks to trim out phase errors which may result from manufacturing tolerances and variations in dielectric materials or components.

WHAT WE CLAIM IS: 1. A phase shifting microstrip two conductor transmission line comprising a dielectric substrate, a ground plane and a conductive strip, said ground plane and said conductive strip forming said two conductors of the transmission line and being attached to opposite sides of said dielectric substrate, and a conductive plate located at the side of said strip away from said substrate and separated from said strip by a selected spacing whereby in use a wave signal propagating along said line has its field affected by said plate to shift the phase of said signal.

2. A transmission line as claimed in claim 1 wherein said plate has a length in the direction of said conductive strip which is shorter than the length of said strip whereby in use signal reflections at each end of said plate are substantially self-cancelling when said plate length equals an integral number of half wave-length(s) of a said signal.

3. A transmission line as claimed in claim 1 or claim 2 wherein said conductive plate is electrically connected to said ground plane.

4. A transmission line as claimed in any one of claims 1 to 3 wherein said conductive plate comprises a grooved plate bridging said conductive strip and having a groove depth selected so as to provide said selected spacing between the conductive plate and said conductive strip.

5. A phase shifting microstrip two conductor transmission line substantially as described herein with reference to Figures 9 to 11 of the accompanying drawings.

6. An antenna system including a transmission line as claimed in any one of claims 1 to 5.

**WARNING** end of DESC field may overlap start of CLMS **.

Claims (6)

**WARNING** start of CLMS field may overlap end of DESC **. provided to electrically connect conductive plate 84 to ground plane 76 of the transmission line. Those skilled in the art will recognize that conductive plate 84 will draw some of the electric field emanating from conductive strip 80 through the region of air formed by the spacing (d) between conductive strip 80 and conductive plate 84. Since a major portion of the electric field will then be passing through air dielectric, the effective dielectric constant, and hence the propagation constant of the microstrip transmission line will be lower. It will also be recognized that as conductive plate 84 is arranged closer to conductive strip 80, the phase shifting effect will be increased. Figure 11 is a graph showing an estimate of the phase shift at 5 GHz which might be realized by a conductive plate of the type shown in Figure 9 with a length (L) of a half wave at the propagation constant of the transmission line. Figure 10 is a planar view of such a conductive plate indicating the location of grounding screws 85 and the length (L) of the conductive plate. It will be evident to those familiar with such transmission lines that it is advantageous to select the length (L) of the field altering structure to be equal to an integral number of half wave length(s), so that the signal reflections occurring at each end of the field altering structure will be approximately self-cancelling. Those familiar with microwave circuits will recognize that the phase shifting structure of Figures 9 and 10 may be used in circuits other than that shown in Figure 1. The structures are advantageously used in complex microstrip networks to trim out phase errors which may result from manufacturing tolerances and variations in dielectric materials or components. WHAT WE CLAIM IS:

1. A phase shifting microstrip two conductor transmission line comprising a dielectric substrate, a ground plane and a conductive strip, said ground plane and said conductive strip forming said two conductors of the transmission line and being attached to opposite sides of said dielectric substrate, and a conductive plate located at the side of said strip away from said substrate and separated from said strip by a selected spacing whereby in use a wave signal propagating along said line has its field affected by said plate to shift the phase of said signal.

2. A transmission line as claimed in claim 1 wherein said plate has a length in the direction of said conductive strip which is shorter than the length of said strip whereby in use signal reflections at each end of said plate are substantially self-cancelling when said plate length equals an integral number of half wave-length(s) of a said signal.

3. A transmission line as claimed in claim 1 or claim 2 wherein said conductive plate is electrically connected to said ground plane.

4. A transmission line as claimed in any one of claims 1 to 3 wherein said conductive plate comprises a grooved plate bridging said conductive strip and having a groove depth selected so as to provide said selected spacing between the conductive plate and said conductive strip.

5. A phase shifting microstrip two conductor transmission line substantially as described herein with reference to Figures 9 to 11 of the accompanying drawings.

6. An antenna system including a transmission line as claimed in any one of claims 1 to 5.