GB1579583A - Method and system of conveying information - Google Patents

Description

(54) METHOD AND SYSTEM OF CONVEYING INFORMATION (71) We, SPACETIME SYSTEMS CORPORATION, A Corporation organized according to the laws of the State of California, United States of America, of 11980 San Vicente Boulevard, Brentwood, California 90049, United States of America, do hereby declare the invention,'tfor which we pray that a patent may be granted to us, and the method by which it is to be performed, to be particularly described in and by the following statement: Summary of the invention The present invention relates to methods and systems of intelligence transmission, and more particularly to improvements in the transmission of information or intelligence.

Background of the invention Present communication or intelligence transmission techniques use a carrier or carrier waves, usually referred to as a sine wave carrier. The carrier is modulated by intelligence or information to be transmitted. There are three principal types of modulation used, and these include amplitude modulation, frequency modulation, and phase modulation. All of these modulation techniques manipulate a carrier wave generated by a tuned circuit, and the modulating information is transmitted by changing the carrier amplitude, frequency or phase.

There are other techniques for conveying information including, for example, the generation of pulse bursts, the generation of pulses which have a characteristic (e.g., pulse amplitude, pulse duration and so forth) directly related to input analog or digital information, and so on. In each case, a complex wave of some type is generated and in turn transmitted, recorded or otherwise utilized.

All such complex waves contain sidebands which are used fo convey the information or intelligence. This means, of course, that a given bandwidth is required. In conventional amplitude, modulation (AM) systems, an oscillator circuit is used which is tuned to a carrier frequency, fc. When modulated (amplitude modulated) by an audio frequency fm, the circuit generates sideband frequencies of values fc + fm and fc - frn, all of which are transmitted to the receiving station. A minimum bandwidth of 2fm thus is required to transmit a single tone of frequency frn, even though the tone itself may contain no information.

A narrow band frequency modulation (FM) system involving, of course, frequency modulation of a carrier as distinguished frorn amplitude modulation thereof) is similar to an AM system, except that the first sideband frequencies fe + fm and fc - fm are almost always out of phase with the carrier frequency, and do not appreciably affect its amplitude unless the modulation index is changed. The bandwidth normally must be greater than 2fry, where fm is the highest audio frequency transmitted. Wideband FM systems, on the other hand, use bandwidths which are several times as wide as the highest audio frequency, thereby transmitting the sideband frequencies of fc + turn, where n typically is 4 or more.

The information capacities of various communication systems have been determined by Nyquist, Hartly, Shannon and others. It is customary to define information in terms of discrete components or elements thereof, such as "bits," and system capacity in terms of bits per second. From definition, a character with n discrete signal levels (e.g., voltage levels) can be represented by a binary number of log2 n bits. If each character is transmitted at intervals of T seconds, the system capacity, C (bits per second) is obtained by the following equation: C = T log2 n. (1) For conventional communications systems in which a carrier wave is modulated in amplitude (AM), frequency (FM) or in pulses (PCM), the time response T of the system is inversely proportional to the bandwidth B.The number of discrete signal levels depends on the signal-to-noise ratio (S/N) as follows: n+l+N5. (2) From equations (1) and (2), 1 S C = T log2 (1 + N ) (3) Shannon therefore, assuming both random signal and noise distribution, obtained an equation for the information capacity of a communications system which shows the capacity dependent on the bandwidth B and the signal-to-noise ratio, S/N as, S C (1 B log2 (1 + N ) (4) In the conventional AM systems discussed above, the maximum amount of information which can be transmitted by amplitude modulation in the bandwidth B-2fm is obtained from equation (4).

Since conventional AM or FM transmission systems are based on the random distribution of intelligence and are normally non-synchronous, the modulation and the carrier must produce sidebands relative to the phase variation between the vector components or the frequencies being combined.

The present invention provides a method of conveying information including the steps of dividing the information into increments, the time duration of each increment varying as a function of the amplitide of the information during the respective increment, generating substantially pure sine wave cycles, the cycling time duration of each sine wave cycle being proportional to the time duration of the corresponding increment to form a sine wave signal train, and conveying the signal train.

The invention further provides a method for conveying or retrieving information contained in a waveform wherein the cycle time duration of the waveform cycles is proportional to the amplitude of the information, including the steps of determining the cycle time duration of each cycle of the waveform, generating substantially pure sine wave cycles of uniform amplitude as a function of said information, each cycle commencing near the 90 degree positive peak point, a?d each sine wave cycle having a duration of cycling time proportional to the waveform cyclic time duration to form a sine wave signal train, and either: (i) conveying the signal train formed from the sine wave cycles or (ii) decoding the sine wave cycles to retrieve the information.

In this aspect, the invention is particularly applicable to the conveying of information contained in a frequency modulated waveform A system for carrying out a conveying method in accordance with the invention comprises first means for receiving input information, modulator means coupled with said input means for converting said input information into a frequency modulated signal, first cycle time sampler means coupled with said modulator means and responsive thereto for producing as an output a train of substantially pure sine wave cycles, the duration of cycling time of each sine wave cycle varying as a function of the cycling time duration of each corresponding cycle of the frequency modulated signal, output means connected to the first cycle time sampler means for conveying the sine wave signal train, second means for receiving said sine wave signal train, second cycle time sampler means coupled with said second means for generating respective new substantially pure sine wave cycles, the duration of cycling time of each new sine wave cycle corresponding to the cycling time duration of each cycle of the received sine wave signal train, and decoding means coupled with said second sampler means for retrieving said information.

Embodiments of the invention described herein, for conveying or communicating information or intelligence differ from methods and systems used in the past because a carrier is not modulated in the same manner as is done in usual systems, and widely spaced sidebands are not necessary to convey information. The information to be communicated radiated, induced or conducted as by through the air, water, wire, or other medium, is sampled and briefly stored to allow for individual sine waves to be generated as a function of each sample. Each sine wave has a time duration representing the amplitude and rate of change of the information sampled. There results a series of individual sine waves forming a carrier which contains therein the information to be conveyed.Thus, a new form of modulated carrier wave is created or generated similar to a phase or frequency modulated carrier, which is not a modulated wave in the usual sense, but rather, a wave wherein each cycle is basically a sine wave having a particular and unique time duration (note Figure 3b).

In this method and system, the sampling rate is very high compared to the information rate and the phase deviation is very small compared to the carrier frequency producing only low frequency sidebands very close to either side of the carrier. With this method and system then, bandwidth can be substantially reduced, thereby enabling different channels of information to be transmitted or otherwise conveyed significantly closer together in the frequency spectrum. Additionally, transmitted power can be reduced since the encoding and decoding process along with the narrower bandwidth requirement significantly improves the signal-to-noise ratio.

In the new system described herein, the information transmission capacity cannot be obtained directly from equations above since the reference equations all assume a natural or random information distribution. The combining or vector addition of the frequency information or rate of change is synchronously sampled at the zero crossovers of the carrier (note Figure 3c) where the noise amplitude is minimum, stored and then precisely projected onto the individual cycles making up the carrier when the amplitude is maximum and the rate of change is minimum. This provides a carrier which contains the same information as carrier using previous modulation methods without producing the wide sidebands associated with non-synchronous or random information distribution.All information can be transmitted in a bandwidth of approximately 100 hertz, or less, with the carrier frequency modulated in the range of fc + 50 hertz. If the transmitter signal is not filtered, sideband amplitudes outside the 100 Hz bandwidth, due to the encoding process, on the order of two to three percent of the carrier exist in frequency fc + fm; however, these sidebands are not necessary to the retrieval process and the receiver can be sharply tuned to the carrier to increase the sensitivity and further improve the signal-to-noise ratio. The signal-to-noise ratio referenced in equations (2) and (3) is very high, because the receiver accepts noise only within the sampling time period, and there the components in the receiver itself do not contribute any appreciable noise.The maximum noise in the retrieved information is essentially dependent upon the thermal noise of the final stage or stages thereof.

Briefly, and in summary, the devices, methods and/or systems discussed herein for transmitting intelligence do not rely upon sidebands to convey frequency information. In converting the intelligence or information into the transmitted wave, the information is sampled at a given rate, and individual approximated sine waves, each of which has a cycle duration which represents the amplitude of the intelligence, are generated to form the transmitted wave. The change of the approximated sine waves from cycle to cycle represents frequency of the intelligence. This transmitted wave is a continuous series of individual almost pure sine waves rather than a continuous complex wave. A receiver receives the transmitted wave, and a cycle time sampling operation similar to that performed in the transmitter is performed thereon.A new resulting waveform of individual sine waves is recreated or generated (similar to those generated by the transmitter), and the sine waves have cycle durations corresponding to the respective cycle durations of the transmitted wave (and, thus, directly correspond to each sample of the original intelligence). These sine waves are each decoded in order to recover the original intelligence supplied to the transmitter.

Through use of discrete level detecting in performing the cycle time sampling in both the transmitter and receiver, both internal (e.g. thermal noise) (and external (e.g., static and local interference) noise effects are generally disregarded, and through the use of high Q filter devices in the receiver, the information can be conveyed essentially within the generated carrier wave and with a very high signal to noise ratio. Also, no AVC is necessary in the receiver and the cycle time sampling portion thereof provides an output which is not dependent on the signal strength of the received signal. Therefore usual fading of the output signal does not occur, but the signal is substantially constant until the input signal to the final IF stage falls below a predetermined value.

By way of example, embodiments of the invention will now be described with reference to the accompanying drawings in which: Figure I is a block diagram of an exemplary transmitter according to the present invention.

Figure 2 is an exemplary block diagram of a receiver according to the present invention.

Figures 3a and 3b respectively illustrate the conventional method of modulation of a carrier and the improved method described herein of transmitting information.

Figure 3c is a diagram of waveforms representing cycle time sampling and sine wave generation within a transmitter and receiver.

Figure 4 is a circuit diagram of an exemplary cycle time modulator used in a transmitter according to the present invention.

Figure 5 is a circuit diagram of an exemplary cycle time sampler and sine wave generator used in a transmitter and a receiver in accordance with the present invention.

Figure 6 is a diagram illustrating phase relationships of waveforms within a cycle time sampler.

Figure 7a is a circuit diagram of an exemplary decoder used in a receiver according to the present invention and Figure 7b illustrates the characteristic of the filter used in the decoder.

Figures 8a-8c illustrate a component structure for minimizing inductance and a waveform pertaining thereto Figures 9a-9c are portions of a circuit diagram of a commercial transmitter-receiver device modified in accordance with the present invention.

Although an exemplary embodiment of the present invention will be described and discussed in the environment of a particular Citizen's Band transmitter-receiver system, it is to be understood that this exemplary embodiment is for illustrative purposes only and for facilitating a full under-standing of the methods, circuits, and system aspects of the present invention. The concepts, circuits, systems, and devices described herein are readily applicable to other methods and systems wherein intelligence is transmitted, or conveyed or measured, in any portion of communications channels of the electromagnetic spectrum, from direct current or very low frequency transmissions and above. For example, embodiments of the present invention are fully applicable to the transmission of various types of information over conventional audio, video, microwave, and other communication links.Likewise, embodiments of this invention are useful in conveying various types of information, including information in either analog (e.g., audio information such as music, video information, and so on) or digital form. Because of the lack of dependence on sidebands or bandwidth for conveyance of intelligence, the rate of transmission may be increased (more information per unit time) appreciably with no expansion of bandwidth.

DETAILED DISCUSSION Overall method and system Turning now to the drawings, and first to Figures 3a and 3b, there are shown exemplary waveforms which illustrate differences between the conventional method of modulation and the method of the present invention in transmitting information or intelligence. As can be seen from Figure 3a, in the conventional method of modulation there is a random insertion of information as indicated by symbols Xl through Xs during three cycles of the carrier. As will be apparent from Figure 3a, there is a slope change at each insertion point and the slope changes are distributed randomly throughout the cycle or period.

Additionally, there is not intelligence hold or storage time as in the case of the present method and system as will be described below.

Figure 3b is a waveform illustrating both the sampling and insertion points of what may be considered a carrier wave in accordance with the present invention. One waveform is shown in Figure 3b but this waveform represents a composite for illustrating (1) the sampling points (at or near zero degrees) of an input information wave, and (2) the insertion points (near ninety degrees) of a generated substantially pure sine wave which serves as the carrier. As will be seen therefrom, sampling of the information or intelligence to be conveyed occurs at or near the zero degree point in the information wave (which, as will be explained later, is a narrow band FM signal). Insertion of the intelligence occurs near the ninety degree point, or the positive peak, of the resulting carrier. In practice, this insertion occurs slightly offset, to the left or to the right, from the ninety degree point. The letter Y refers to the insertion point and represents the maximum modulation displacement as either increasing (positive) or decreasing (negative) time. The designations Yl, Y2 and Y3 indicate the first half cycle displacement proportional to the intelligence sampled (wherein Yl is negative, Y2 is zero and Y3 is positive). The shaded bottom half cycle is not changed in any way.

It will be apparent from Figure 3b tht there is no slope change associated with increasing or decreasing the time period at the top of the half cycle in the insertion process.

Additionally, it should be noted that the period Z indicates the maximum storage time (from sampling to insertion), and this intelligence storage, or hold, time is less than the period of the carrier divided by four.

Thus, as noted earlier, a new form of modulated carrier wave is created or generated similar to a phase or frequency modulated carrier, but which is not a modulated wave in the usual sense, but rather, a wave wherein each cycle is basically a sine wave having a particular and unique time duration (360 + X). Additionally, there is no random information distribution, but both information sampling and information insertion occur at precise points. With the present concepts the sampling occurs where the noise amplitude is minimum and storage of the information occurs precisely on the individual cycles making up the carrier when the amplitude is maximum and the rate of change thereof is minimum.

Inasmuch as insertion occurs at or near the peak, there is no slope change such as occurs in conventional modulation methods. This method, therefore, produces a carrier which contains information but without producing the wide sidebands associated with nonsynchronous or random information distribution as noted earlier.

Turning now to Figures 1 and 2, there is shown a block diagram of an exemplary transmitter and receiver incorporating the present invention. The block diagrams in Figures 1 and 2 represent a commercially available five channel Citizen's Band Johnson Messenger III transmitter/receiver, which has been modified to incorporate several circuit changes and additions which will be described in greater detail later. Although an exemplary embodiment in the form of a modified conventional Citizen's Band transceiver is illustrated and discussed, the present invention is applicable to a wide range of intelligence and communications conveying the measuring systems as noted earlier.

Transmitter The transmitter shown in Figure 1 includes an information source 10, such as a microphone for receiving voice information, and an amplifier 11 for amplifying the audio signals from the microphone. Other information sources can be used as will be apparent to those skilled in the art. The output of the amplifier 11 is applied to a cycle time modulator 13. The cycle time modulator 13 is an added circuit, and its output is coupled to the standard crystal oscillator 14 of the transmitter/receiver unit. The modulator 13 and oscillator 14 convert the original information to a narrow band FM signal. The amplifier 11 of the conventional circuit is essentially an audio amplifier feeding a transformer, the secondary of which is coupled to the cycle time modulator 13.The resulting signal voltage applied to the input of the cycle time modulator, which voltage is a function of the audio input of the transmitter, varies a capacitor within the cycle time modulator proportional to the incoming signal. This operation in turn varies the capacitance of the crystal of the crystal oscillator 14 thereby affecting its oscillating frequency. This action causes the oscillator 14 to have a small complex frequency change in its output proportional to the amplitude of the audio information, which frequency will vary over a small range (e.g., 25 to 50 hertz) depending on the modulation drive within the cycle time modulator 14. If desired, a mixer and heterodyne oscillator may be used before or after the oscillator 14 in a conventional manner.If the original information is a narrow band FM signal (or any other suitable signal with a frequency or time deviation proportional to amplitude samples of the intelligence and which can be sampled at a precise point, such as a single sideband signal), the modulator 13 and oscillator 14 can be .eliminated and the signal applied directly to a sampler 15 to be described below.

Inasmuch as the frequencies of an oscillator are not sine waves, but contain harmonics plus the slight frequency changes noted above, a cycle time sampler is used to remove wave distortions and any unwanted amplitude variations added to the signal by the oscillator 14 (or by a mixer if used). The narrow band FM signal from the oscillator 14 is not transmitted, but is converted into individual sine waves as will be made apparent subsequently. In the transmitter of Figure 1, two stages of cycle time sampling are employed, and are represented by cycle time samplers 15 and 16. The cycle time samplers 15 and 16 sample each cycle of the signal from the oscillator 14 and generate respective clean or substantially pure sine waves, each sine wave having a cycle time duration proportional to the amplitude of a respective discrete sample of th original audio information from the amplifier 11.In a preferred embodiment, when the output signal of the oscillator 14 exceeds 0.05 volts a sine wave output is produced by the cycle time sampler 15. Inasmuch as the output from the oscillator 14 can be relatively large, but may not be as high as this voltage level, two stages, samplers 15 and 16, are used. As will appear subsequently, the receiver typically uses more stages of these cycle time samplers because of the lower magnitude of the input signal thereto and because of the desire to maximize the signal-to-noise ratio.

Considering the operation of the cycle time samplers 15 and 16 further, Figure 3c illustrates waveforms for five stages of samplers in the receiver an the waveforms for stages 3 and 4 are representative of the operation of the respective cycle time samplers 15 and 16 of the transmitter. Curve 20 of stage 3 can be considered representative of a single cycle output of the oscillator 14 which is applied to the input of the sampler 15. The sampler 15 includes transistor stages that operate between saturation and cut-off, and function to produce an output sine wave cycle through a resonant tank circuit output as will be explained in greater detail in connection with a discussion of Figure 5. In an exemplary embodiment, each sampler 15 includes two transistor stages, each having a gain of 10.

Waveform 21 of stage 3 in Figure 3c represents the output of the first transistor stage and input to the second stage of sampler 15, and waveform 22 of Figure 3c represents the output waveform of the sampler 15. This signal is applied as an input to the second sampler 16 and is clipped as indicated by waveform 23 of stage 4 shown in Figure 3 by the first stage of the sampler 16. Typically, the clipping level of the waveform 23 is 0.05 volts as indicated on waveform 24 of Figure 6. Similarly, the clipping level of the waveform 25 of Figure 3 (representing the output of the first transistor stage and input to the second transistor stage of the sampler 16) is 0.5 volts as indicated on waveform 26 in Figure 6. The cycle time sampler 16 then provides an output sine wave as indicated at 27 in Figure 3c.

The sine wave output 27 of the second sampler 16 is applied to a driver amplifier 30. The output of the driver amplifier 30 is further amplified by a power output, or final, stage 31 which drives the transmitter antenna 32.

In the exemplary embodiment of the transmitter in Figure 1 the information source 10, amplifier 11, oscillator 14, driver 30, power output stage 31 and antenna 32 represent portions of the coventional transceiver and the cycle time modulator 13 and samplers 15-16 are added circuits. Additionally, in the conventional transceiver, the oscillator 14 is connected to the driver 30, and the amplifier 11 serves as a modulator to drive the driver 30 and power output stage 31, but these latter connections are eliminated in the modification.

Briefly, in summary, the audio input to the transmitter is used to vary the frequency of the oscillator 14, or modulate the frequency thereof. This operation results in the oscillator 14 providing an output wave of desired average frequency (such as 27.07 Mhz, representing Channel 10 of the Citizen's Band) with the audio information impressed thereon. Then, each cycle of the signal from the oscillator 14 is sampled by the cycle time sampler 15, the output of which is similarly sampled by the sampler 16 to ultimately produce a series of sine waves, each having a time duration proportional to the amplitude (and rate of change proportional to frequency) of discrete samples of the input audio, and wherein the sampling rate is determined by the frequency of the oscillator 14.

Assuming that each sampler 15 and 16 has a 40 db voltage gain, if the signal input to the first sampler 15 is less than approximately 0.05 volts then the second transistor stage of the first sampler 15 will not go into saturation and into a constant current mode and, thus, will operate as a voltage amplifier with an input-to-output voltage gain of 100, such as represented by the waveforms 20, 21 and 22 of stage 3 in Figure 3c. Hence, the wave 20 from the oscillator 14 will be amplified as indicated at 22. As many stages as are necessary than will operate as a small signal amplifier until an input to a subsequent stage reaches a voltage of approximately 0.05 volts.It has been determined that in the transmitter of Figure 1 only two stages of the samplers 15 and 16 are necessary in order to achieve a sine wave output from the output stage of the second sampler 16, wherein the output transistor stage thereof operates as a constant current course for the resonant tank circuit thereof, and the output of this stage of the second sr;;ipler 16 will have a fixed amplitude (i.e., a fixed amplitude sine wave whose duration is proportional to the input information sample). The output of the second sampler 16, and thus the signal applied to the transmitter antenna 32, comprises a signal wave formed of a series of discrete sine waves wherein each sine wave has a time duration directly proportional to the amplitude of a sampled portion of the input audio.This waveform is similar to, but different from, a frequency modulated wave in as much as the latter is created by modulating a carrier rather than taking discrete samples of the audio input at the oscillator or carrier frequency rate.

As an alternative, the oscillator may be amplitude modulated and the sampling accomplished at a predetermined voltage level related to the peak of the unmodulated wave of the oscillator, rather than at or near the zero line as in the case of the previously described frequency modulated oscillator. Thus, when a fixed voltage level is reached, sampling occurs. Assuming, for example, a continuous oscillator output of one volt peak (from zero line to peak) and a sampling reference at 0.7 volt, then the wave cycle time would occur at the same point on each wave; that is, when the peak of the wave reaches 0.7 volt. This would then provide a constant carrier of sine waves of equal duration. Now assuming the oscillator is amplitude modulated, for example, ten percent so the peaks would be 1.1 volts maximum, when the modulated wave reaches 0.7 volt sampling likewise will occur.In this case if the instantaneous sampled wave maximum or peak voltage is 1.1 volts, then its 0.7 volt point arrives earlier in time than it would with a 1 volt peak and, therefore, would appear to make the cycle a different duration. If the second wave sampled had a peak at 1.09 volts, then the sampling point would occur later on the time scale and create a longer duration sine wave. In this manner the amplitude variations are converted to respective cycle time durations. This also is true for phase modulation since when the phase leads the original wave it occurs earlier as amplitude and thus a variable phase will produce variable sine wave durations.

Receiver The receiver shown in Figure 2 includes an antenna 32a coupled to a crystal tuner 36. As with conventional transceivers, the antenna 32 of the transmitter and the antenna 32a of the receiver may comprise a single, common antenna. The output of the tuner is applied to a mixer 37 which has connected thereto a local oscillator 38. The output of the mixer 37 is coupled through a first IF stage 39 to a second mixer 40 having a local oscillator 41 coupled thereto. The foregoing components of the receiver are conventional ones in the exemplary transceiver described herein; however, the local oscillator 41 is modified to be variable as will become apparent when considering Figure 9c later.With the unit tuned to Channel 10, the frequency of the crystal tuner 36 is 27.075 Mhz, and the local oscillator 38 operates at 31.375 Mhz, 4.3 Mhz above the incoming frequency. Thus, a 4.3 Mhz signal is applied through the first IF 39 to the mixer 40. The local oscillator 41 can be a hand tuned oscillator and is not necessarily a crystal oscillator, and the frequency thereof is 4.755 Mhz to provide a 455 Khz second IF signal to the first cycle time sampler 42 of the receiver.

The cycle time samplers of the receiver comprise five samplers 42 through 46. Each of these samplers is similar to the samplers 15 and 16 of the transmitter; however, more stages are generally needed in the receiver inasmuch as it is necessary for at least two samplers (e.g. 45 and 46) to operate in a saturated mode in order to ensure a maximum signal-to-noise-ratio. The output of the fifth sampler 46 is applied to a decoder 48 which retrieves, from the sine wave train output of sampler 46, the original audio information fed to the transmitter. The decoder includes a high Q filter. In the present embodiment, the output of the decoder 48 is coupled to an audio output stage 49 which may include a loudspeaker or any other suitable reproduction device.

The cycle time samplers 42 through 46 are components added to the receiver bypassing the usual second IF of the conventional receiver described herein. The receiver as shown in Figure 2 acts much like another transmitter which takes information from the mixer 40 and generates new clean sine waves (each having a cycle time duration which is a function of the mixer output signal) and these sine waves then are decoded by the added decoder 48 to retrieve the original audio informatin. The waveforms shown in Figure 3c particularly illustrate the operation of each of the five samplers 42 through 46 of Figure 2.

Waveform 50 of Figure 3c represents the input to the first sampler 42 from the mixer 40.

Waveform 51 represents the output of the first transistor stage, and thus the input to the second transistor stage, of the first sampler 42 as will be described in more detail in connection with a discussion of Figure 5. Waveform 52 of the first sampler stage 42 will be a complex wave because the magnitude of the input signal to the first sampler is too low to cause saturation therein. The first sampler stage 42 operates as small signal amplifier and the output 52 will be a sine wave modulated by signal noise and thermal noise of the components of this stage thereby resulting in an output complex wave 52. The second sampler stage 43 operates in a similar manner to produce waveforms similar to 50, 51 and 52 of Figure 3c but the waves are of increased magnitude as indicated in Figure 3c because of the gain of the first stage.The number of stages shown in Figure 2, and the waveforms in Figure 3c assumes that the third sampler stage 44 input voltage is the first stage whose input voltage reaches the 0.05 volt level (or other predetermined voltage level) wherein the second transistor stage thereof will saturate and operate as a constant current source for the resonant tank circuit thereof to thereby produce the output sine wave 22 shown in Figure 3c. When the input signal 20 to the third sampler stage 44 just reaches 0.05 volts, the first transistor stage thereof will still operate as a small signal amplifier inasmuch as the input is not yet sufficiently high to cause this transistor stage to saturate; however, the second transistor stage will saturate and produce a sine wave 22. The wave 22 from the output of the third sampler stage 44 is applied as an input to the fourth sampler stage 45.Inasmuch as each stage has gain and raises the amplitude of preceding output signals ultimately to the 0.05 voltage level and above (such as up to one or two volts or more), the input wave to the first transistor stage of the fourth sampler 45 will be clipped as indicated at 23 in Figure 3c as will be the input to the second transistor stage of the fourth sampler 45 as indicated at 25 at Figure 3. The output of the fourth sampler stage 45 than is a clean pure sine wave 27. Each succeeding stage after the third stage 44 recreates a new sine wave, but the rise time (note, for example, leading edge 25a of waveform 25 as compared to leading edge 21a of waveform 21 in Figure 3c is reduced to one per cent of the preceding stage.This factor, coupled with the fact that each succeeding sampler stage essentially operates as a free running single cycle generator, enables the thermal noise and static or disturbing noise in the system to be substantially minimized, such as constantly at 96 db down from the input signal. The only thermal noise generated is in the resonant tank circuit of the final sampler 46. Then, the resultant output of the fifth sampler 46 is a series of sine waves each having a slightly different time duration respectively proportional to the different amplitudes of the input audio. When the time duration of any cycle is changed by 10-14 seconds or more, the decoder 48 will operate to retrieve the original information.

From the foregoing, it will be apparent to those skilled in the art that a method of information or intelligence transmission and reception, and system therefor, is described and illustrated wherein discrete samples of the amplitude of input information are taken and respective discrete and almost pure individual sine waves are generated as a function thereof, thereby producing a resultant waveform composed of these sine waves. The input information is converted into a frequency modulation signal; however, such signal is not composed of clean or substantially pure sine waves but it is a complex wave. Thus the FM signal is sampled to create clean or substantially pure sine waves. This resulting waveform may be transmitted (through the air, water or wires, or stored for later use, as by recording the same).As is known, transmitted information may become distorted in amplitude by noise, static and other interference, but the same is discriminated against in the receiver.

This waveform is received by the receiver and may be operated upon in conventional manners (as by heterodyning), and then sampled much in the same way as the original information was sampled in the transmitter to generate a series of sine waves having a cycle time duration representing the original information. These sine waves then are decoded in a decoder including a high Q filter to retrieve the original information, and the resulting signal may then be amplified and applied to a suitable transducer (e.g., a loudspeaker) or otherwise utilized.The system thus does not transmit the frequency spectrum of the transmitter oscillator, inasmuch as the latter only determines the time duration of each sample cycle, and then the cycle time samplers cause the generation of discrete sine waves having the same time duration as the output of the oscillator and thus proportional to amplitude samples of the original information. This also is true in the receiver wherein the input waveform thereto, which may have distortions on it as a result of interference and the like, is not amplified in a traditional sense but is sampled and sine waves produced in what may be considered as the second or final IF stage of the receiver. This IF stage requires no AVC inasmuch as it functions as a constant current amplifier and thereby generates a current sine wave of constant amplitude from its final stage (e.g., sampler 46 of Figure 2).

Thus, the various amplitude signals, or "hash," resulting from thermal component noise and interference are discriminated against, and the final result is a fixed signal-to-noise ratio to large value (e.g., 65-95 db down) regardless of the gain in this IF stage. Since no AVC is employed. there is no gain increase when the carrier wave (i.e., sine wave train) is absent and therefore any audio output from the receiver is relatively low. Tests have indicated the noise ratio reduces to 96 db down with no carrier.

Additionally, since no AVC is necessary, and the IF amplifier stage functions in a constant current mode, the amplifier volume output (e.g., from output stage 49 of the receiver) is not dependent upon signal strength and essentially no fading exists. That is, the volume output will be constant until the input signal to a given IF stage (e.g., to the input of the first cycle time sampler 42) falls below a predetermined voltage (such as, 0.05 volts), and then the volume level drops to zero. This feature of the present system allows the same to be used as a proximity detector or the like by virtue of the fact that at a predetermined distance of separation between transmitter and receiver, a predictable and controlled increase (as the two are moved together) or decrease (as the two are moved away) in output from the output stage of the receiver can be ascertained. A detector (i.e., a voltage detector) can be used in an IF stage to sense a given signal level and to turn on or off the output of the receiver when this signal level occurs. Thus, the receiver may be set to have a predetermined signal to noise ratio or maximum reception distance Exemplary circuits Considering now the transmitter of Figure 1 along with the circuit diagram of the cycle time modulator 13 as shown in Figure 4, the output of the amplifier 11 of Figure 1 is connected to input terminal 61 and 62 of the cycle time modulator 13 shown in Figure 4.A terminating resistor 63 is connected across the input terminals 61 and 62, and a coupling capacitor 64 is connected between the terminal 61 and 62, and a coupling capacitor 64 is connected between the terminal 61 and a junction 65 between a voltage variable capacitor 66 and a radio frequency (RF) bypass capacitor 67. A resistor 68 provides a d.c. path for the coupling capacitor 64. An adjustable resistor 69 forms a voltage dividing resistor to supply reference voltage to the voltage variable capacitor 66. This resistor is adjustable to provide this reference voltage at the frequency of interest (e.g., the average frequency of the oscillator 14). A capacitor 70 and inductor 71 form a filter for the frequencies of the crystal oscillator 14.The resistor 69, with inductor 71 in series and resistor 72 thus form a voltage divider to provide the d.c. voltage operating point for the voltage variable capacitor 66, with the adjustable resistor 69 supplying the reference voltage, and the resistor 72 forming a portion of the voltage divider and serving as a stabilization resistor for the capacitor 66. The Junction point 73 at the upper end of the capacitor 66 is connected to the crystal 75 of the crystal oscillator 14.

The signal voltage across input terminals 61 and 62 varies the ground voltage and the voltage across the capacitor 66, thus varying its capacitance proportional to the magnitude of the incoming signal. This operation in turn varies the equivalent circuit capacitance of the crystal 75 thereby shifting its oscillating frequency proportional to the change in circuit capacitance by the incoming signal. The crystal oscillator 14 is a standard component of the transceiver described herein and is shown in greater detail in Figure 9a. The crystal 75 of the oscillator 14, as is known, has capacity which would result in some amplitude modulation on the output wave and, thus, this output wave is not a true, or substantially pure, sine wave.

The left-hand terminal of the crystal 75 in the conventional transmitter/receiver unit is normally connected to ground (at 62). However, with the connection as shown in Figure 4, the capacitor 66 isolates the amplitude voltages created by the modulation signal from the crystal 75 and, therefore, removes any amplitude drive to the base of the transistor 76 in the oscillator 14, thus removing the amplitude effect of modulation. The oscillator 14 thus will have a small complex frequency change in its output and the frequency will vary by only several cycles (such as through 25 to 50 cycles) depending on the modulation drive of the crystal 75. Desired roll-off of the transmitter can be achieved by changing the values of capacitor 67 and resistor 68 (and/or with the addition of a resistor in the line from capacitor 64 to resistor 68).

Inasmuch as the frequencies produced by the oscillator 14 are not clean or pure sine waves, but the output signal thereof contains harmonics plus the slight frequency changes in the oscillator frequencies, the cycle time samplers 15 and 16 shown in Figure 1 are used in the transmitter to remove essentially all wave distortions and amplitude effects caused in stages preceeding the first sampler 15. The combination of the modulator 13 and sampler stages 15 and 16 essentially supply a Fourier analysis of the input complex information wave at the clock rate of the oscillator 14. As has been noted before, a mixer and heterodyne oscillator may be used preceeding the first sampler stage 15 to provide a desired carrier frequency.The cycle time samplers 15 and 16 of Figure 1 then remove the distortion or complex wave from the carrier applied to the input of the first sampler 15, and generate the output sine waves as described earlier.

Considering the cycle time samplers 15 and 16 in more detail, a circuit of an exemplary sampler is illustrated in Figure 5. Input terminals 80 and 81 thereof are coupled with like terminals connected with the secondary of transformer 79 at the output of the crystal oscillator 14 in Figure 4. This, of course, is the manner of connection of the first cycle time sampler 15; however, the input of the second cycle time sampler 16 is connected to the output of the crystal oscillator 14. The cycle time sampler (which also is a sine wave generator) of Figure 5 includes two transistor stages comprising transistors 82 and 83 which generate the waveforms briefly discussed earlier in connectin with Figure 3c. A current limiting resistor 84 is connected from input terminal 80 to the base of the transistor 82.A bypass capacitor 85 is connected between the input terminal 81 and a common supply line 86. The capacitor 85 is of the non-inductive type. A voltage divider comprising resistor 87 and 88 is provided to supply bias voltage to the transistor 82. A diode 89 connected from junction 90 to the base of the transistor 82 serves as a bias limiting diode. A load resistor 91 is connected between the collector of the transistor 82 and a voltage supply line 92 (lines 86 and 92 provide the voltage supply for the sampler). This resistor 91 also serves as a base current supply for the second stage transistor 83 when the first stage transistor 82 is in a cut-off state. Resistor 93, in conjunction with by-pass capacitor 94, forms a current limiting network for the transistor 82, and resistor 95 and by-pass capacitor 96 provide a source voltage for the bias voltage for the transistors 82 and 83.Variable resistor 97 functions as a current limiting resistor for the constant current mode of the second stage transistor 83, and adjustment of this resistor determines the current through the collector-emitter path of the transistor 83 when the same is in its constant current (saturated) state. This potentiometer is adjusted for a given output voltage (such as, five volts peak-to-peak for 34 db gain per stage when in class A operation) from transformer winding 102. A pair of diodes 98 and 99 are connected from the left hand terminal of the variable potentiometer 97 to the junction of the base of transistor 83 and the collector of transistor 82. This diode network provides a reference voltage for the transistor 83, when the transistor 83 operates in the constant current, or saturated, mode.This diode network is of a noninductive type, and is constructed in a manner similar to that shown in Figures 8a-8b and will be described in more detail later. However, it should be noted that eight of the diodes 98 and 99 are used in each of the cycle time samplers 15 and 16 of the transmitter; whereas, two of these diodes are used in each of the samplers 42-46 of the receiver in the exemplary embodiment illustrated and described herein. The number of diodes used depends upon the overshoot (or spike generated) of the transistor 83 (see Figure 8c), and upon the frequency of operation of the circuit. More diodes are utilized where this overshoot is high and significant and/or in higher frequency applications. It is preferable to use diodes having a very sharp knee.

The collector of transistor 83 is connected to a resonant tank circuit 100. This tank circuit includes a primary winding 101 and a secondary winding 102, the latter having output terminals 103 and 104 (which, in the case of the first cycle time sampler 15 of Figure 1 are connected to input terminals 80-81 of the second time sampler 16). A resistor 105 is connected to a tap on the primary 101 and to the line 92 and serves as a wave shaper and current control for the tank circuit 100. The tank circuit 100 also includes a capacitor 106 in parallel with the primary 101. In the case of the transmitter, the tank circuit 100 is specially constructed; whereas, in the case of the receiver a suitable tank circuit for 455 Khz IF, such as a Miller coil No. 2032 or equivalent, can be used.The Q of the capacitor 106 must be equal to the Q of the tank coil for bandpass symmetry, and the effective reactance of both the capacitor 106 and the inductor 101 of the tank circuit 100 must be equal and 1800 out of phase at the desired frequency in order to generate clean or substantially pure sine waves.

In the case of an exemplary embodiment of the transmitter, the winding 101 comprises seven and one half turns on a one-fourth inch core, and the winding 102 comprises two and a half turns on this core. The core is approximately two inches long, with the longitudinal spacing at A (between the coils 101 and 102) being approximately three-fourths inch. Two iron slugs are disposed within the core and are adjustable. The capacitor 106 may be 51 pf, and in this transmitter embodiment a 250 pf capacitor may be added across the winding 102.

The resistor may be 4700 ohms. This arrangement has been used in a 27 Mhz transceiver as illustrated and described herein. For other frequencies, the windings 101 and 102 are wound and the slugs positioned until the tank circuit 100 resonates with a Q of greater than 150 at the desired carrier frequency. Examples of other components used in the sampler of Figure 5 will be given later herein.

As explained earlier, typically an exemplary transmitter as shown in Figure 1 need use only two of the samplers 15 and 16 illustrated in Figure 5, because of the relatively high signal voltage output of the oscillator 14; whereas, generally more stages are necessary in the receiver. In order to most suitably illustrate the operation of the cycle time sampler of Figure 5, the same will be discussed in a combination of several stages (e.g., five) and with reference to the waveforms of Figure 3c. Also, as noted earlier, the circuit of Figure 5 is designed such that with 0.05 volts input to the base of the transistor 82, the transistor 83 will saturate.Assuming then that several stages are cascaded (as samplers 42-46 in the receiver of Figure 2) and each sampler stage is designed to have a 40 db gain, and the signal input of terminals 80, 81 is less than approximately 0.05 volts, the second stage transistor 83 will not go into saturation, or its constant current mode, and the first sampler stage (i.e., 42) will operate as a voltage amplifier with an input-to-output voltage output gain of 100 and in a manner illustrated by waveforms 50, 51 and 52. Waveform 50 represents the complex wave input to the base of the transistor 82 (from oscillator 14 or mixer 40 of Figures 1 and 2), and waveform 51 illustrates the collectcr voltage of transistor 82 and the base voltage of the transistor 83.Waveform 52, which in tl;e first sampler stage is an amplifier version of input waveform 50, and thus is a complex wave, represents the output current of the transistor 83.

Sampler stages are added as necessary to operate as small signal amplifiers until the output of a stage reaches a voltage of approximately 0.05 volts. A second sampler stage, such as sampler 43 of Figure 2, is used for this purpose and has waveforms similar to 50, 51 and 52 but of increased magnitude as indicated in Figure 3c because of the gain of this second sampler. When the input at the first stage (i.e., sampler stage 44 in Figure 2) is very small, several stages may be necessary to reach the 0.05 output (and consequent input to the next succeeding stage). Assuming that the third stage (i.e., sampler stage 44 in Figure 2) and "Stage 3" of Figure 3c input voltage is the first stage whose input voltage exceeds the 0.05, or other predetermined input threshold voltage, then the second transistor 83 of this third stage will operate as a constant current source for the resonant tank circuit 100, and the output of this third stage will be a fixed amplitude substantially pure sine wave, all as indicated by waveforms 20, 21 and 22 of Figure 3c. The waveform 22 of Figure 3, from the third sampler stage, now is a sine wave cycle as distinguished from the complex waveform 52 of the first sampler stage shown in Figure 3c, inasmuch as this first sampler stage operates as a small signal amplifier.The output waveform 52 is a complex wave which represents a sine wave which is modulated by signal noise and thermal noise of the components of the first stage.

Thus, when the input signal 20 (Figure 3c) to the base of the transistor 82 of the cycle time sampler in Figure 5 (representing the third stage 44) has reached 0.05 volts, transistor 82 will operate as a small signal amplifier in as much as it has not yet reached its saturation point. However, the transistor 83 is caused to saturate (when its base input reaches 0.5 volts) and operates in a constant current mode, since the voltage applied to the base of the transistor 83 will be limited by the diodes 98-99. Therefore, the noise signal available to the tank circuit 100 will only exist during the rise time of the diodes 98-99. The balance of the output sine wave cycle will be free running and will not contain the complex signals of the modulated noise.The tank circuit 100, whose current will be zero upon the completion of the non-conducting cycle, will be 90" out of phase with the base voltage of the transistor 83.

When the transistor 83 conducts current, the current in the tank circuit will rise immediately and output voltage will exist after the flux of the tank circuit decreases. Instantly, the tank circuit 100 will have the maximum current of the transistor 83, and the load current caused by resistor 105 will reverse voltage there across to discharge the circuit. As the voltage in the tank circuit increases, the current in the resistor 105 will decrease until this current passes through the normal zero point of the sine wave, and then the voltage across the resistor 105 will reverse, and the current in the tank circuit will decrease inasmuch as it will share the current produced by transistor 83 when conducting. This operation will be more apparent by examining the waveform 27 of Figure 3c and the lower waveform of Figure 6.

Figure 6 shows the phase relationship of the cycle time sampling in one stage of Figure 5, and the half cycles A and B shown with respect to stage 4 in Figure 3c illustrate the phase relationship between the input to the base of transistor 83 and the output current of the transistor 83. The current waveform in the winding 101 thus leads the voltage 90" upon excitation (at the zero point of output current wave where it is beginning to go down), and the current and voltage are in phase from the bottom peak (at 1800) back up to the upper peak (360 ). The diodes 98 and 99, and the variable resistance 97, form a constant current circuit for the transistor 83 to supply a constant current source for the tank circuit to each half cycle.

When the input voltage of a preceeding stage has decreased to 0.05 volts, the first transistor 82 (such as in the third stage sampler if the input reaches or exceeds 0.05) will commence to shut off the conducting signal to the second transistor 83 which, in turn, turns off the transistor 83. Thus, the transistor 83 ceases to conduct current, and this occurs after it has been conducting for approximately 173 of the sine wave (as more fully illustrated in the waveform 27 of Figure 3), and the tank circuit will then reverse its polarity and produce current in its reverse cycle thereby forming a sine wave current cycle illustrated at 27 in Figure 3c. The resistance of the windings of the tank circuit and the dissipation factor of the capacitor 106 must be made equal to produce a pure or substantially pure sine wave.

Each succeeding stage (e.g., the fourth and fifth samplers 45 and 46 of Figure 2) therefore will reduce the rise time to one per cent of the preceeding stage, and since the sine wave generator output (i.e., transistor 83 circuit and tank circuit 100) is generally in a free running state, the thermal noise and static or disturbing noise in the system can be maintained significantly down at a constant value, such as 96 db down from the signal. The only thermal noise generated in the cycle time sampler stages is in the final tank circuit and in the output amplifiers.Large db gain amplifiers can be used because the samplers are not directly dependent on signal-to-noise ratios since the noise is virtually eliminated by the sampling process except in the final .age. The rise time of each succeeding sampler stage is one hundred times faster because of the gain of each stage; thus, any hash that occurs on the leading and trailing edges of the signal is significantly reduced. Because of this, the receiver accepts noise only within the sampling time period which is minimal.

It should be noted that the network in the lower portion of Figure 5 comprising capacitors and resistors 85, 87 and 93-96 forms a temperature compensating network to maintain the saturation point of the transistors 82 and 83 at a constant level with respect to the predetermined voltage levels (e.g., 0.05 and 0.5) applied to their respective bases. The reactance of the capacitor 96 with respect to the resistor 95 (and like-wise 94 with respect to 93, and 85 with respect to 87) is less than 10 percent of the resistance of resistor 95.In the transmitter, because of operation at higher frequencies (e.g., 27 Mhz and vs. 455 Khz in the receiver) higher frequency response transistors are used, and higher frequency response transistors are used where more sampler stages are employed so as to maintain the fast rise time (e.g., 25a of Figure 3c) of the waveforms on the transistors 82 and 83. The resistor 84 functions as a current limiting resistor when the first transistor 82 saturates, and the diode 89 keeps the impedance of the preceding coil (of the preceding stage) coupled to input terminals 80 and 81 constant, to thereby protect the transistor 82 from negative voltages, and minimizes the danger of burning out the transistor 82.Since the resistance 91 supplies base current to drive the second transistor 83 to saturation, the value of this resistance should be increased if the supply voltage to the sampler is increased.

In summary, the cycle time samplers )15 and 16 of the transmitter in Figure 1, or 42-46 in the receiver of Figure 2) serve to sample each cycle of the input waveform thereto, and to recreate or generate an almost pure sine wave corresponding thereto in time duration, while discriminating against amplitude variations in the input waveform. In the case of the transmitter, the original information is converted from its complex waveform to frequency through the cycle time modulation of the oscillator 14, and each output cycle of the oscillator 14 is sampled and a clean sine wave corresponding in duration thereto is ultimately generated by the final sampler 16. The final wave, comprising a series or train of individual sine waves, the duration of each corresponding to amplitude samplers of the original information, is transmitted and received by the receiver.Inasmuch as the received wave may be degraded by static and other interference thereby imposing amplitude variations thereon, each cycle of this received wave then is similarly sampled to ultimately recreate sine waves from final sampler stage 46, with each sine wave corresponding in time duration to the respective waves received by the receiver (and consequently the duration of each sine wave corresponds to the amplitude of the incremental samples of the original information). These approximated sine waves then are applied to a decoder to recover the original information, which decoder will be described in more detail below.

As noted earlier, a detector can be used in a sampler stage (such as at winding 102) to sense a given signal level and to turn on or off the output of the receiver when this signal level occurs.

Decoder The output of the final sampler stage 46 is connected by terminals 103-104 across an input potentiometer 110 of the decoder as shown in Figure 7a. A variable tap 111 of potentiometer 110 is connected to an input terminal 112 of a piezo-electric or crystal filter 113 of high Q. An output terminal 114 of the filter 113 is connected to the base of an emitter follower driver transistor 115, and to an upper terminal of a resonant capacitor 116 for the filter 113. The capacitor 116 is adjustable to compensate for the lead length of the filter 113.

This capacitor is used to cancel the inductance of the filter 113, as well as the inductance of winding 102 (Figure 5) reflected through to the decoder circuit. The capacitor 116 thus takes any knees out of the filter response, so as to provide a respose curve having a clean slope as seen in Figure 7b. Additionally, the leads of the filter 113 should be as short as possible (longer leads require a larger value for capacitor 116, and a larger value thereof causes high frequency degeneration). For example, terminal 112 can be soldered directly to the top 111 and terminal 114 directly to the base of transistor 115.

The lower terminal of the potentiometer 110, as well as a third or common terminal 117 of the filter 113 and the lower terminal of the capacitor 116 are connected to a supply or ground line 118. A potentiometer 119 and resistor 120 provide bias voltage for the transistor 122 in an emitter follower configuration which applies voltage across filter resistors 123-124.

The transistors 115 and 122 are connected in a modified Darlington configuration to function as an infinite impedance detector. Resistor 121 reduces the leakage current of the collector to base circuit of transistor 122, and the value of this resistor depends on the HFE of the transistor 122. Preferably, the transistors 115 and 122 have a high HFE (e.g., 300) and the resistors 120 and 121 are large (e.g. 470 K) so as to obtain good high frequency response. Capacitors 125-126 bypass the resulting average carrier frequency from the audio output across resistor 124 as prov d-d a terminals 127-128.

The filter 113 is similar to conventior.al crystal filters; however, both the crystals 113a and 113b thereof are a matched frequency pair (instead of slightly different), and the filter is of very high Q. As explained earlier, variable capacitor 116 is provided to remove any inductive component from the crystal filter 113 to enable the same to resonate at the proper frequency, which frequency is slightly off of the resultant average center frequency of the carrier waveform applied to the decoder. This enables the incoming sine waves to operate on the functional slope (from A to B in Figure 7b) of the decoder filter. Other variable components (i.e., adjustable resistances and/or adjustable inductance) may be incorporated into the filter circuit if desired to provide further variation of the resonant circuit. Also, a series combination of a diode and resistance may be coupled in parallel with the variable capacitor 116 to enable the Q of the filter circuit to be increased where there is a slight mismatch between the crystals 113a and 113b; however, such increase of Q is not necessary unless it is desired for the receiver to operate at a very narrow bandwidth (e.g., approximately below 50 cycles bandwidth). The higher the Q of the decoder as provided by the filter 113 and associated components, the less frequency deviation (and thus bandwidth) is needed for an efficient decoder output. If the output amplifier used (e.g., 4a of Figure 2) is relatively sensitive, good signal quality can be obtained with less frequency deviation; whereas, if the output amplifier is less sensitive, greater frequency deviation is needed to obtain good signal quality.

The potentiometer 119 may be adjusted to preset the bias of the transistor 115 to determine the operating point of the crystal 113. Resistor 120 supplies bias voltage to the transistor 115, and also provides an effective ground for the capacitive effect of the crystal 113.

Figure 7b illustrates the response characteristic of the filter 113 of the decoder. In this exemplary embodiment, the filter is a high pass, high Q filter. Assuming a center frequency at B of 454.7 Khz, a frequency at A of 455.5 Khz, and a frequency at C of 453.9 Khz, then the frequency change from A to C is 1600 Hz. The voltage on terminal 130 sets the frequency cutoff at A, and the setting of potentiometer 119 sets the cutoff at point C.

Assuming that this voltage range (from cutoff points A to C) is 6 volts, the voltage at point B is three volts (at the center frequency of 454.7 Khz). The frequency change from the center frequency at B to either A or C is 800 Hz. Dividing three volts (from B to A or B to C) by 800 Hz (from B to A or B to C) indicates that .0037 volts or 3.7 millivolts per cycle of carrier is provided in the decoding process. That is, as the frequency deviation between incoming sine waves varies 800 Hz, then a three volt variation likewise occurs, and this equates to 3.7 milivolts per cycle. The setting of the voltage on the collector of the transistor 115 (as provided by the voltage at terminal 130) and the setting of the potentionmeter 119 determine the upper and lower limits of detected variation of the incoming signal frequency for which an output information signal is provided.A higher voltage between these limits, provides voltage (e.g., millivolts) per cycle of devitation, and the variation of this voltage provides an adjustable Q for the filter circuit.

It should be noted that the HFE of th transistors 115 and 122, and the resistors 120-121 must be selected to be compatible with the time constant provided by the resistor 68 and capacitor 67 circuit of the cycle time modulator of Figure 4 so as to ensure appropriate cycle deviation relationships between the transmitter and receiver. The value of the time constant provided by resistor 68 and capacitor 67 will roll off the frequency of the transmitter. Thus, the design of a given receiver circuit is related to the design of the compatible transmitter.

For example, it may be desirable in compensating for the uneven frequency characteristics of the human ear to provide roll-off by changing the values of the resistor 68/capacitor 67 circuit, and/or the HFE of the transistors 115, 122 and the value of the resistors 120, 121.

Since the signal input to input terminals 103-104 is a sine wave of fixed amplitude, and whose individual cycles vary in time duration, and tuned crystal filter 113 modifies the time variations of the incoming sine waves and thereby produces an amplitude modulated carrier fed to the base of the transistor 115. This carrier supplies a minimum of two millivolts of audio where the variations of the cycle time of the sine waves vary 2 x 10-l6 seconds.

Therefore, the time cycle carrier cycles need only vary 1 x 10-15 seconds to produce 10 millivolts audio output. Thus a carrier frequency of any reasonable frequency of transmission need only change so the total frequency during one second would change only five hertz to produce this result. As an example, if a 25 Mhz carrier produces 0 volt, a 25.000005 Mhz carrier will produce 0.01 volts; or when the carrier is 100 Mhz, the output would be zero, but 100 Mhz + 5 Hz would produce 0101 volt. The time duration of each sine wave and the amount of time deviation from cycle to cycle (i.e., rate of change) of the incoming sine waves represents or carries the transmitted information. As another example, if eight cycles is varied plus or minus ten thousand times per second, a 10 Khz note can be received.A high frequency note will be higher in amplitude than a low frequency note as compared to the signal transmitted; therefore a contour setting which can be provided by variable capacitor 116 can be used for changing the Q or slope of the filter to provide compensation (similar to tone control compensation) and thus amplitude adjustment. Similarly, as noted previously, the values of resistor 68 and capacitor 67 (Figure 4) can be changed to thereby provide compression compensation for the transmitted signal. A low impedance substantially pure resistive, driver input to the filter 113 is useful, and if desired the crystal 113 can be proceeded by a standard emitter follower low impedance non-inductive driver.

In the present exemplary embodiment, the frequency of the crystals 113a and 113b preferably are less than 455 Khz, and are preferably approximately a 454.7 Khz matched pair. The filter 113 forms a series resonant circuit, and the output voltage will be Q times the input voltage across terminals 112 and 117. This output voltage falls at the 6 db point of the resonant circuit when this input voltage remains constant and the resonant circuit is tuned to the frequency of the input and, thus, the voltage out will be dependent on frequency. When the frequency of the input to the crystal is varied by the incoming signal, the output voltage varies in amplitude proportional to frequency.By adjusting this resonant circuit (through use of adjustable components as noted above) the bandwidth and the Q thereof can be changed to thereby supply the contour necessary to match the contour of the transmitter (although such adjustments have not been found necessary in the particular embodiment illustrated and described herein). For example, if the transmitter is transmitting with a maximum sine wave variation of 50 cycles, the capacitor 116 or other suitable variable component can be adjusted to the 6 db point to produce the voltage required to drive the following amplifier. The overall slope of the resonant circuit can be made to follow a square law; therefore, it is desirable to operate on the slope of the incoming sine wave near the peak thereof to thereby provide the greatest voltage variation as a function of frequency change.The bandwidth of the decoder thus can be adjusted as desired, with normal operation set at 50 to 500 cycles bandwidth to compensate for frequency drift of the transmitter and the local oscillator in the receiver. When adjusted to 50 cycles, and less, good response still is obtainable but drift can affect the output unless compensated for by variation or feedback to maintain the desired relationship between the resonant frequency of the filter and the center frequency of the transmitter and receiver local oscillator. In a test conducted with the system illustrated and described herein, stereo music from a local radio station in San Diego, California, was received, with right and left channels summed, and fed as an input to the transmitter.A maximum bandwidth of 500 Hz was all that was necessary to transmit the high fidelity music, and to receive and play back the same through the receiver of Figure 2. The high fidelity music was transmitted and received without loss of quality with the bandwidth of the decoder operating at approximately 50 cycles. There was believed to be an improvement because the signal resulting from the multiplex switching frequency in usual FM stereo was eliminated. Thus, it has been concluded that it is the rate of change of frequency within a relatively limited bandwidth as detected by the decoder that enables the original information or intelligence to be retrived, without the necessity of sidebands as is generally thought to be the case in, conventional systems.

Additional information can be conveyed by combining the present invention and conventional systems, such as by amplitude modulating (AM) the resultant carrier wave of the present system with separate information, and then retrieving the AM information from the transmitted wave in a conventional manner while retrieving information within the wave as explained with respect to the receiver of Figure 2.

Minimum Inductance Devices As noted earlier wit respect to diodes 98 and 99 of the cycle time sampler shown in Figure 5, it is necessary that the diode network have a minimum inductance in order to eliminate or minimize the overshoot or spike occasioned in the operation of the circuit. These diodes, because of their lead lengths, will cause spikes (note the left hand wave in Figure 8c) which will result in sidebands being transmitted by the transmitter. Therefore an arrangement is provided wherein a series of diodes (i.e., two diodes 98 and 99 in the receiver, or eight similar diodes in the transmitter) is used such that the inductance of their leads is cancelled.

Also, certain diodes will resonate at higher frequencies, such as a commercially available diode has been found to resonate at 240 Mhz, and this is a result of the inductance occasioned by lead lengths.

It has been found that this inductance can be substantially minimized or eliminated by arranging a plurality of diodes in a configuration as illustrated in Figures 8a-8b, and with an even number of diodes (and other components in some applications). For example, Figures 8a-8b illustrate eight diodes 140 through 147 wherein their respective leads (designated a and b) are tapered and connected to rings 150 and 151. Connecting leads 152-153 and 154-155 are respectively connected and soldered to the rings 150 and 151, and the diode leads are equally spaced and in line with its respective diode. In this construction, the number of diodes 140-147 is an even number and the diodes are evenly spaced. The space between junctions of the diodes (within the diode encapsulating member) preferably is equal to the diameter of the diode junction (pn) itself.Additionally, the diameter of the electrical conducting and connecting rings 150 and 151 preferably is one half of the diameter of the circle "X" circumscribed by the leads eminating from the diodes (note Figure 8b).

Additionally, the connecting leads 152-153 (and likewise 154 and 155) comprise two leads spaced 180 apart on the ring 150 (and the ring 151), and are formed in a "V" configuration with external connections to the diode network being made at points 156 and 157 (i.e., at the apex of the V's). The diameter of the rings 150 and 151 also is preferably approximately 50 times the diameter of a leads 140a-147a and 140b-147b.

This configuration of the diodes functions to cancel the inductance of the leads, and serves to eliminate spikes like S, and S2 in Figure 8c, and result in a waveform as shown at the right in Figure 8c. The inductance action of a lead cannot occur unless a magnetic field can be produced around it. When an even number of loads are placed in a parallel or circular configuration, the field of each pair of leads will be equal and opposite. That is, the inductance energy of the field of one lead will be of an opposite polarity or direction with respect to the inductance field of the second lead. Since the leads are of equal lengths and the currents therein are equal, a cancellation will occur, and the only inductive action will be a field that can exist around the two leads.The more pairs of leads that are used, the greater will be the cancellation since the current will be divided amond the leads and be less in each lead and the resultant field from each lead therefore is less, and also the resultant diameter of the total field becomes larger and weaker.

Circuit modifications of conventional transceiver For the sake of completeness, Figures 9a through 9c illustrate modified portions of the circuit of a conventional Citizen's Band transceiver to incorporate the concepts of the present invention therein. The modifications made are in accordance with the block diagrams of Figures 1 and 2, and the circuits of Figures 4, 5, 7 and 8. The commercial unit involved is a Johnson Messenger IIt, the circuit diagram drawing of which is identified by the numbers 242-0143-XXX and 025-0644-031, dated 4-10-67. The original components therein are identified with their original component nomenclature (e.g., C31, L7, D2, S1B, Q17, etc.).

In the transmitter portion of the circuit shown in Figure 9a, normally the audio output stage 160 is used to modulate the driver 161 and the power output stages 162, the latter two stages being fed by crystal oscillator 163. In the modified circuit, the secondaries 164-165 of transformer T2 are connected across a potentiometer 166 which provides an input to the cycle time modulator 13.This cycle time modulator is substantially identical to that shown in Figure 4, and like components are identified with like numerals, but with a letter suffix 'a". As will be readily apparent from the modified circuit shown in Figure 9a, the same operates in the manner described in connection with the discussion of Figure 4 to convert the audio output of transformer T2 into a corresponding frequency modulated waveform at the output of the oscillator 163 (at output transformer 79a). The driver 161 and power output 162 stages are used without change, except they are not directly driven by the audio output stage 160.

As to the receiver, modifications thereof are shown in Figure 9b wherein certain changes are made following the second IF of the receiver portion of the transceiver; namely, the addition of the cycle time samplers 42-46 and the decoder 48. The output of the decoder 48 is connected in the transceiver to the volume control (not shown) thereof which in turn feeds into audio output stages to feed speaker 170 (Figure 9a). Additional change is made in the local oscillator in the transceiver as shown in Figure 9c since this crystal oscillator must be variable. The 4.755 MC crystal 172 of the conventional unit (which is normally connected to line 173, has added thereto a variable capacitor 174 and parallel resistance 175. The variable capacitor 174 is used to vary the drive of the oscillator to enable the bandwidth of the receiver to be changed or minimized.The bandwidth thereof depends upon the amount of drive and modulation of the transmitter. The variable capacitor 174 thus can vary the frequency of the oscillator and can be used to reduce bandwidth. In an exemplary modification, a variable capacitor of 9-60 pf having a resistance of 1000 ohms was used which enabled variation of the frequency of the local oscillator from 4.755 Mhz to 4.855 Mhz. The resistor 175 in parallel with the capacitor 174 provides a path to ensure that both capacitors (the capacitance of the crystal 172 and the capacitor 174) can properly charge in accordance with standard practice. Greater improvements can be made by also replacing the 4300 Khz IF amplifier with a sampling IF stage of the type described herein instead of the conventional IF amplifier of the unit.

The following are tables of exemplary component values for the new circuits described herein.

Cycle time sampler Component Transmitter Receiver Resistor 84 820 Ohms 2200 Ohms Capacitor 85 220 pf .1 Microfarad Resistor 87 68 K 68K Resistor 88 33K 33K Ohms Diode 89 1N934 1N934 Resistor 91 5600 Ohms 12K Ohms Resistor 93 3300 Ohms 3300 Ohms Capacitor 94 .1 Microfarad Ceramic 2 Microfarads Resistor 95 1K Ohms 1K Ohms Capacitor 96 .1 Microfarad Ceramic 2 Microfarads Potentiometer 97 250 Variable 500 Ohms Variable Diodes 98-99 1N934 1N934 - Transistors 82-23 TI 6219-1 2 N5086 Coils 101-102 ---------Described in Specification------------------------- Resistor 105 6800 47K Ohms Capacitor 106 51 pf, 180 pf Cycle time modulator Component Transmitter Resistor 63 8 Ohms, or to mach output of preceeding modulator Capacitor 64 10 Microfarads Voltage Variable Capacitor 66 33 pf at 4 V, Motorola MV 1638 or equivalent Capacitor 67 220 pf Resistor 68 3300 Ohms Resistor 69 10 K adjustable for frequency Capacitor 70 220 pf Inductance 71 15 ph (for 27 Mhz use 68 turns on iron core Resistor 72 3300 Ohms Decoder Component Receiver Potentiometer 110 5K Ohms Crystal Filter 113 Described in specification Variable Capacitor 116 30 Microfarads Potentiometer 119 10K Ohms Resistor 120 470K Ohms Transistor 115 SK 7358 Resistor 121 470K Ohms Transistor 122 SK 7358 Capacitors 125,126 .02 microfarads (both) Resistor 123 820 Ohms Resistor 124 10K Ohms WHAT WE CLAIM IS: 1. A method of conveying information including the steps of dividing the information into increments, the time duration of each increment varying as a function of the amplitude of the information during the respective increment, generating substantially pure sine wave cycles, the cycling time duration of each sine wave cycle being proportional to the time duration of the corresponding increment to form a sine wave signal train, and

**WARNING** end of DESC field may overlap start of CLMS **.

Claims (18)

1. **WARNING** start of CLMS field may overlap end of DESC **.

   Cycle time modulator Component Transmitter Resistor 63 8 Ohms, or to mach output of preceeding modulator Capacitor 64 10 Microfarads Voltage Variable Capacitor 66 33 pf at 4 V, Motorola MV 1638 or equivalent Capacitor 67 220 pf Resistor 68 3300 Ohms Resistor 69 10 K adjustable for frequency Capacitor 70 220 pf Inductance 71 15 ph (for 27 Mhz use 68 turns on iron core Resistor 72 3300 Ohms Decoder Component Receiver Potentiometer 110 5K Ohms Crystal Filter 113 Described in specification Variable Capacitor 116 30 Microfarads Potentiometer 119 10K Ohms Resistor 120 470K Ohms Transistor 115 SK 7358 Resistor 121 470K Ohms Transistor 122 SK 7358 Capacitors 125,126 .02 microfarads (both) Resistor 123 820 Ohms Resistor 124 10K Ohms WHAT WE CLAIM IS: 1.A method of conveying information including the steps of dividing the information into increments, the time duration of each increment varying as a function of the amplitude of the information during the respective increment, generating substantially pure sine wave cycles, the cycling time duration of each sine wave cycle being proportional to the time duration of the corresponding increment to form a sine wave signal train, and

   conveying the signal train.
2. 2. A method as claimed in claim 1, wherein the step of dividing the information into increments comprises the steps of generating a narrow band frequency modulated signal responsive to the information and determining the zero-crossing points of the signal to determine the time duration of the increments.
3. 3. A method of conveying information contained in a frequency modulated waveform comprising the steps of determining the cycle time duration of each cycle of the waveform, generating substantially pure sine wave cycles as a function of said information, each cycle commencing near the 90.degree positive peak point, and each sine wave cycle having a duration of cycling time proportional to the waveform cycle time duration to form a sine wave signal train, and conveying the signal train.
4. 4. A method as claimed in any dne of the preceding claims, and further including the steps of receiving the sine wave signal train and generating from each cycle thereof respective new substantially pure sine wave cycles, the duration of cycling time of each new sine wave cycle corresponding to the cycling time duration of a corresponding cycle of the sine wave signal train, and decoding the new sine wave cycles to retrieve said information.
5. 5. An information conveying system for carrying out a method as claimed in any one of the preceding claims, the system including first means for receiving input information in the form of a complex waveform, modulator means coupled with said input means for converting said input information into a modulated signal, said modulator means comprising a cycle time modulator and oscillator circuit for converting said input information into a narrow band frequency modulated signal, cycle time sampler means comprising a plurality of cycle time samplers coupled with said modulator means and responsive thereto for producing as an output a train of substantially pure sine wave cycles, the duration of cycling time of each sine wave cycle being proportional to the cycling time duration of the respective cycle of the modulated signal, output means coupled with said cycle time sampler means and responsive thereto for providing an output for said system.
6. 6. An information receiving and decoding system for carrying out a method as claimed in claim 4, the system including input means for receiving the sine wave signal train, cycle time sampler means coupled with said input means for generating respective new substantially pure sine wave cycles, the duration of cycling time of each new sine wave cycle corresponding to the cycle time duration of each cycle of the received sine wave signal train, said cycle sampler means comprising a plurality of cycle time samplers with at least the last sampler thereof operating in a constant current mode, and decoding means coupled with said sampler means and responsive to the new sine wave cycles for retrieving said information by providing an output signal variation responsive to slope deviations of the sine wave cycles applied thereto.
7. 7. An information conveying system for carrying out a method as claimed in claim 4, the system comprising first means for receiving input information, modulator means coupled with said input means for converting said input information into a frequency modulated signal.

   first cycle time sampler means coupled with said modulator means and responsive thereto for producing as an output a train of substantially pure sine wave cycles, the duration of cycling time of each sine wave cycle varying as a function of the cycling time duration of each corresponding cycle of the frequency modulated signal, output means connected to the first cycle time sampler means for conveying the sine wave signal train, second means for receiving said sine wave signal train, second cycle time sampler means coupled with said second means for generating respective new substantially pure sine wave cycles, the duration of cycling time of each new sine wave cycle corresponding to the cycling time duration of each cycle of the received sine wave signal train, and decoding means coupled with said second sampler means for retrieving said information.
8. 8. An information conveying system as claimed in claim 7, wherein said modulator means is a cycle time modulator and oscillator circuit for converting said input information into a narrow band frequency modulated signal.
9. 9. An information conveying system as claimed in claim 7 or claim 8, wherein said first cycle time sampler means comprises a plurality of cycle time samplers, with at least the last sampler operating in a constant current mode.
10. 10. An information conveying system as claimed in any one of claims 7 to 9, wherein said second cycle time sampler means comprises a plurality of cycle time samplers, with at least the last sampler operating in a constant current mode.
11. 11. An information conveying system as claimed in any one of claims 7 to 10, wherein said decoding means comprises high Q crystal filter means for providing an output signal variation responsive to slope deviations of input sine wave cycles thereto, said filter means comprising a crystal filter having a pair of crystals substantially matched in frequency and at a frequency near the average frequency of said new substantially pure sine wave cycles.
12. 12. An information conveying system as claimed in claim 11, including output means including a capacitor connected with said filter means for varying the drive thereto.
13. 13. An information conveying system as claimed in claim 7, wherein each of said first cycle time sampler means and said second cycle time sampler means includes at least one cycle time sampler comprising input means for receiving signal cycles wherein the duration of cycling time of each cycle is proportional to information to be conveyed, first level detecting means responsive to each cycle for generating an output substantially square wave signal proportional thereto when said cycle exceeds a predetermined level, second level detecting means coupled with said first level detecting means operating in a constant current mode when said square wave signal is received thereby, and resonant circuit means coupled with said second level detecting means and responsive thereto for producing substantially pure sine wave cycles each having a duration of cycling time proportional to the time duration of each input signal cycle being sampled.
14. 14. A cycle time sampler for use in a method as claimed in any one of claims 1 to 4, comprising input means for receiving frequency modulated signal cycles wherein the time duration of each cycle is proportional to information to be conveyed, first level detecting means responsive to each cycle for generating an output substantially square wave signal proportional thereto when each cycle exceeds a predetermined level, second level detecting means coupled with said first level detecting means operating in a constant current mode when said square wave signal is received thereby, and resonant circuit means coupled with said second level detecting means and responsive thereto for producing a substantially pure sine wave cycle having a duration of cycling time proportional to the time duration of each input signal cycle being sampled, each sine wave cycle commencing near the ninety degree positive peak before the zero crossing point thereof.
15. 15. A method for conveying or retrieving information contained in a waveform wherein the cycle time duration of the waveform cycles is proportional to the amplitude of the information, including the steps of determining the cycle time duration of each cycle of the waveform generating substantially pure sine wave cycles of uniform amplitude as a function of said information, each cycle commencing near the 90 degree positive peak point, and each sine wave cycle having a duration of cycling time proportional to the waveform cycle time duration to form a sine wave signal train, and either: (i) conveying the signal train formed from the sine wave cycles or (ii) decoding the sine wave cycles to retrieve the information.
16. 16. A method as claimed in any one of claims 1 to 4, substantially as described herein with reference to the accompanying drawings.
17. 17. An information conveying system substantially as described herein with reference to, and as illustrated by, Figures 1, 4 and 5 of the accompanying drawings.
18. 18. An information receiving the decoding system substantially as described herein with reference to, and as illustrated by, Figures 2, 5, 7a and b of the accompanying drawings.