FR3103655A1 - Switching module and power converter comprising such a module - Google Patents

Abstract

Switching module and power converter comprising such a module The present description relates to a switching module comprising N switches (Qc1, Qc2; Qc3) connected in parallel, with N integer greater than or equal to 2, and a control circuit (500) configured to apply to the N switches respectively N periodic control signals phase-shifted two by two by 2π / N. Figure for the abstract: Fig. 5

Description

Switching module and power converter comprising such a module

This description relates generally to electronic circuits. It relates more particularly to a switching module, and a power converter comprising such a module.

Various power converter topologies have already been proposed, based on the use of switching-controlled switches at relatively high frequencies, typically from several tens of kHz to several hundreds of kHz. These converters are generally called switching converters.

For example, CFFB type converters (Current Fed Full Bridge) are direct-to-direct (DC/DC) converters particularly well suited to applications requiring a conversion ratio high voltage and sensitive to input current ripples, for example applications for converting the output power of photovoltaic panels or fuel cells to a higher voltage intended to be applied at the input of an inverter.

A CFFB DC/DC converter typically comprises a primary stage comprising an H-bridge controlled in series with an input inductor, modulating the input power at a relatively high frequency, followed by a transformer adapted to transmit the modulated power by the primary stage, followed by a secondary stage rectifying the power transmitted by the transformer and supplying the output power of the converter.

Examples of CFFB DC/DC converters are described in the article “Study and Implementation of a Current-Fed Full-Bridge Boost DC–DC Converter With Zero-Current Switching for High-Voltage Applications” by Ren-Yi Chen et al. . (IEEE Transactions on Industry Applications (Volume: 44 , Issue: 4 , July-aug. 2008)) and in the article entitled “Current-fed full-bridge DC-DC converter with nonlinear control scheme” by M. Phattanasak et al . (2014 14th International Conference on Environment and Electrical Engineering).

The article “Analysis and Design of Zero-Voltage-Switching Current-Fed Isolated Full-Bridge Dc/Dc Converter” by U R Prananna et al. (2011 IEEE Ninth International Conference on Power Electronics and Drive Systems) proposes a CFFB converter topology in which a branch comprising a capacitor in series with an active switching switch or blocking switch ("active clamp") is placed in parallel with the bridge in H of the primary stage. The switching switch is controlled synchronously with the H-bridge switches, so that the switching of the H-bridge switches from the open (off) state to the closed (on) state is carried out during zero crossings of the voltage at their terminals. This makes it possible to limit the switching losses of the converter. However, a limitation of this topology is that the blocking switch must switch at twice the frequency of the primary stage H-bridge switches.

It would be desirable to improve at least in part certain aspects of known switching converters, and in particular CFFB converters provided with a blocking switch in series with a capacitor in parallel with the H-bridge of the primary stage of the converter.

More generally, it would be desirable to be able to have a switching module that overcomes all or part of the drawbacks of known switching modules.

For this, one embodiment provides a switching module comprising N switches connected in parallel, with N integer greater than or equal to 2, and a control circuit configured to apply to the N switches respectively N periodic control signals phase-shifted two by two by 2π/N.

According to one embodiment, the N control signals have the same frequency and the same duty cycle.

According to one embodiment, the N control signals have a duty cycle less than 1/N.

According to one embodiment, the N switches are gallium nitride transistors.

Another embodiment provides a switching converter comprising an active switching branch comprising a switching module as defined above.

According to one embodiment, the number N of switches of the switching module is equal to two.

According to one embodiment, the converter further comprises, in parallel with the active switching branch, a controlled H-bridge, the control circuit being configured for, at each period of an operating cycle of the converter, successively:
a) controlling first and second diagonals of the H-bridge simultaneously in the closed state;
b) controlling the first diagonal in the closed state and the second diagonal in the open state;
c) controlling the first and second diagonals simultaneously in the closed state; And
d) controlling the first diagonal in the open state and the second diagonal in the closed state,
the control circuit being further configured for:
during step a), maintaining a first and a second of the two switches of the switching module simultaneously open;
during step b), keeping the first and second switches respectively closed and open;
during step c), keeping the first and second switches simultaneously open; And
during step d), keeping the first and second switches respectively open and closed.

According to one embodiment, the converter further comprises an input inductance in series with the H bridge.

According to one embodiment, the assembly comprising the active switching branch and the H-bridge forms a primary stage of the converter, the converter further comprising a transformer arranged to transmit a modulated power supplied by the primary stage and a secondary stage arranged to demodulate the power transmitted by the transformer.

According to one embodiment, the active switching branch further comprises a capacitor in series with the N switches of the switching module.

These characteristics and advantages, as well as others, will be set out in detail in the following description of particular embodiments made on a non-limiting basis in relation to the attached figures, among which:

Figure 1 is a simplified electrical diagram of an example of a CFFB converter provided with an active switching switch;

FIG. 2 schematically represents an example of control signals from the converter of FIG. 1;

FIG. 3 is a simplified electrical diagram of an example of a CFFB converter according to one embodiment;

FIG. 4 schematically represents an example of control signals from the converter of FIG. 3;

Figure 5 is an electrical diagram of an example of a switching module according to one embodiment; And

FIG. 6 schematically represents an example of control signals from the switching module of FIG. 5.

The same elements have been designated by the same references in the various figures. In particular, the structural and/or functional elements common to the various embodiments may have the same references and may have identical structural, dimensional and material properties.

For the sake of clarity, only the steps and elements useful for understanding the embodiments described have been represented and are detailed. In particular, the control circuits of the switches of the converters or switching modules described are not detailed, the realization of these control circuits being within the reach of those skilled in the art based on the functional indications of the present description.

Unless otherwise specified, when reference is made to two elements connected together, it means directly connected without intermediate elements other than conductors, and when reference is made to two elements connected or coupled together, it means that these two elements can be connected or be linked or coupled through one or more other elements.

Unless specified otherwise, the expressions “about”, “approximately”, “substantially”, and “of the order of” mean to within 10%, preferably within 5%.

Figure 1 is a simplified electrical diagram of an example of a CFFB type converter provided with an active switching switch.

The converter of figure 1 comprises a primary stage 110, followed by a transformer T, followed by a secondary stage 130.

In this example, the primary stage 110 comprises a controlled H-bridge H1, or primary bridge. The bridge H1 comprises four controlled switches Q1, Q2, Q3 and Q4, for example identical except for manufacturing dispersions, each comprising two main conduction nodes, and a control node. The switches Q1 and Q2 are connected in series, by their conduction nodes, between input nodes A and B of the bridge. Switches Q3 and Q4 are connected in series, via their conduction nodes, between nodes A and B, in parallel with the branch comprising switches Q1 and Q2. Midpoint C between switches Q1 and Q2 defines a first output node of the bridge, and midpoint D between switches Q3 and Q4 defines a second output node of the bridge. More particularly, in the example shown, switch Q1 has a first conduction node connected to node A and a second conduction node connected to node C, switch Q2 has a first conduction node connected to node C and a second conduction node connected to node B, switch Q3 has a first conduction node connected to node A and a second conduction node connected to node D, and switch Q4 has a first conduction node connected to node D and a second conduction node connected to node B.

In the example of figure 1, the bridge H1 also comprises four diodes D1, D2, D3 and D4 connected respectively in parallel with the switches Q1, Q2, Q3 and Q4. More specifically, in this example, diode D1 has its anode connected, for example connected, to node C and its cathode connected, for example connected, to node A, diode D2 has its anode connected, for example connected, to node B and its cathode connected, for example connected, to node C, diode D3 has its anode connected, for example connected, to node D and its cathode connected, for example connected, to node A, and diode D4 has its anode connected, for example connected, to node B and its cathode connected, for example connected, to node D.

The switches Q1, Q2, Q3 and Q4 are for example field effect transistors, for example gallium nitride transistors, silicon carbide transistors, or silicon transistors, the conduction nodes of the switches corresponding for example respectively to source and drain nodes of the transistors, and the control nodes of the switches corresponding to gate nodes of the transistors. The diodes D1, D2, D3 and D4 correspond for example to body diodes intrinsic to the transistors Q1, respectively Q2, respectively Q3, respectively Q4. As a variant, the diodes D1, D2, D3 and D4 can be specific components separate from the transistors Q1, Q2, Q3 and Q4. More generally, other types of controllable switches can be used to produce switches Q1, Q2, Q3 and Q4.

The primary stage 110 further comprises an input inductance L1 connected in series with the bridge H1, between terminals E1 and E2 for applying a DC input voltage VIN of the converter. More specifically, in this example, inductance L1 has a first end connected, for example connected, to node A, and a second end connected, for example connected, to terminal E1. Moreover, in this example, the node B is connected, for example connected, to the terminal E2.

The primary stage 110 further comprises an input capacitor C1 having a first electrode connected, for example connected, to the terminal E1, and a second electrode connected, for example connected, to the terminal E2.

The primary stage 110 further comprises, in parallel with the bridge H1, an active switching branch 112 comprising an active switching switch Qc in series with a capacitor Cc. More particularly, in this example, the switch Qc has a first conduction node connected, for example connected, to node A, and a second conduction node connected, for example connected, to an intermediate node E of branch 112, and the capacitor Cc has a first electrode connected, for example connected, to the node E and a second electrode connected, for example connected, to the node B. In the example of FIG. 1, the active switching branch 112 further comprises a diode Dc connected in parallel with switch Qc. More particularly, in this example, diode Dc has its anode connected, for example connected, to node A and its cathode connected, for example connected, to node E. Switch Qc is for example a transistor of the same type as transistors Q1, Q2, Q3 and Q4. The diode Dc corresponds for example to a body diode intrinsic to the transistor Qc.

The transformer T comprises a primary winding w1 and a secondary winding w2 magnetically coupled to each other.

The ends of primary winding w1 are respectively connected to output nodes C and D of bridge H1. More specifically, in the example shown, the primary winding w1 has a first end connected to node C and a second end connected to node D.

In the example of FIG. 1, the secondary stage 130 comprises a first branch or rectifying branch comprising two diodes D5 and D6 connected in series between output nodes F and G of the converter, forming nodes for supplying a DC output voltage VOUT and intended to be connected, for example connected, to the terminals of a load LD to be supplied. More particularly, in the example shown, diode D5 has its cathode connected, for example connected, to node F and its anode connected, for example connected, to an intermediate node H of the rectifying branch. Moreover, in this example, diode D6 has its anode connected, for example connected, to node G and its cathode connected, for example connected, to node H.

The secondary stage 130 of the converter of FIG. 1 further comprises, in parallel with the rectifying branch, a second branch or capacitive branch comprising two capacitors C2 and C3 connected in series between the nodes F and G. More particularly, in the In the example shown, capacitor C2 has a first electrode connected, for example connected, to node F and a second electrode connected, for example connected, to an intermediate node I of the capacitive branch. Moreover, in this example, the capacitor C3 has a first electrode connected, for example connected, to the node G, and a second electrode connected, for example connected, to the node I.

The ends of secondary winding w2 of transformer T are respectively connected to nodes H and I of secondary stage 130. More particularly, in the example shown, secondary winding w2 has a first end connected to node H and a second end connected to node I.

In the example of Figure 1, the converter comprises a control circuit 140 (CTRL) adapted to control the switches Q1, Q2, Q3, Q4 and Qc of the primary stage 110.

FIG. 2 schematically illustrates an example of an operating mode of the converter of FIG. 1. FIG. 2 more particularly represents the evolution, as a function of time, of a control signal S14 applied to the control nodes switches Q1 and Q4, a control signal S23 applied to the control nodes of transistors Q2 and Q3, and a signal Sc applied to the control node of transistor Qc. The signals S14, S23 and Sc are by example of the voltages applied respectively:
between the gate and the source of the transistor Q1 and between the gate and the source of the transistor Q2;
between the gate and the source of the transistor Q2 and between the gate and the source of the transistor Q3; And
between the gate and the source of the transistor Qc.

The signals S14, S23 and Sc are for example generated by the control circuit 140 and/or by close control circuits (not detailed) connected to the control nodes of the switches Q1, Q2, Q3, Q4 and Qc.

In this example, each of the switches Q1, Q2, Q3, Q4 and Qc is controlled in the closed state (on) when the signal S14, S23 or Sc applied to its control node is in the high state, and is controlled in the open state (off) when the signal S14, S23 or Sc applied to its control node is in the low state.

In this example, switches Q1 and Q4 are controlled alternately in the closed state then in the open state, according to a periodic switching cycle of frequency f, called switching frequency. Similarly, switches Q2 and Q3 are controlled alternately in the closed state then in the open state, according to a periodic switching cycle of the same frequency f, of the same duty cycle as the control cycle of switches Q1 and Q4 and in phase opposition with the control cycle of switches Q1 and Q4.

In this example, the cyclic switching ratio of switches Q1 and Q4, i.e. the ratio, in the same period T=1/f of the switching cycle, between the duration Ton of the control phase of the switches Q1 and Q4 in the closed state and the total duration of the period T, is greater than 1/2. Similarly, the switching ratio of switches Q2 and Q3 is greater than 1/2. For example, the duty cycle switching ratio of switches Q1 and Q4 is substantially equal to the duty cycle switching ratio of switches Q2 and Q3.

It follows from the above that at each period of duration T=1/f of the converter operating cycle:
- the switches Q1, Q2, Q3 and Q4 are simultaneously on during a phase ranging from a time t1 of closing of the switches Q1 and Q4 (transition of the signal S14 from the low state to the high state) to a time t2 d opening of switches Q2 and Q3 (transition of signal S23 from the high state to the low state);
- switches Q1 and Q4 are on and switches Q2 and Q3 are off during a phase going from time t2 to time t3 of closure of switches Q2 and Q3 (transition of signal S23 from the low state to the high state);
- switches Q1, Q2, Q3 and Q4 are simultaneously on during a phase going from time t3 to time t4 of opening of switches Q1 and Q4 (transition of signal S14 from the high state to the low state) ; And
- switches Q1 and Q4 are off and switches Q2 and Q3 are on during a phase ranging from time t4 to time t5 of closure of switches Q1 and Q4 (transition of signal S14 from the low state to the high state), instant t5 corresponding to instant t1 of the next period T of the cycle.

The switch Qc is for its part kept open throughout the duration of the phases of simultaneous closing of the switches Q1, Q2, Q3 and Q4, that is to say between the instants t1 and t2 on the one hand, and between the instants t3 and t4 on the other hand. On the other hand, switch Qc is kept closed during at least part of the phase of closing switches Q1 and Q4 and opening of switches Q2 and Q3, that is to say between times t2 and t3, and during at least part of the phase of closing switches Q2 and Q3 and opening of switches Q1 and Q4, that is to say between times t4 and t5.

Thus, the switch Qc is controlled in switching at a frequency Fc twice as high as the switching frequency f of the converter.

The control mode described in relation to FIG. 2 makes it possible to obtain that the switchings of the switches Q1, Q2, Q3 and Q4 of the bridge H1 from the open state to the closed state are carried out during zero crossings of the voltage at their terminals. This makes it possible to limit the switching losses of the converter compared to a CFFB converter having no active switching branch in parallel with the primary bridge.

However, a limitation of the converter described in relation to Figures 1 and 2 is linked to the high switching frequency of the switch Qc, which induces significant heating of the latter which can lead to a degradation of the converter and induce reliability problems. .

For example, in the case where switches Q1, Q2, Q3, Q4 and Qc are gallium nitride transistors, and for a switching frequency of 300 kHz and use of the converter at an ambient temperature of 70°C (corresponding to at a typical temperature of use in photovoltaic panel output power conversion applications), the internal temperature of switches Q1, Q2, Q3 and Q4 reaches approximately 90°C, which remains acceptable for this type of transistors, while that the internal temperature of the switch Qc (controlled in switching at 600 kHz) reaches approximately 120° C., which is close to the critical point of degradation of the transistor. It should also be noted that at high temperature, the switching losses are increased compared to operation at lower temperature.

Thus, the maximum switching frequency that can be applied to the converter of Figure 1 is limited by the fact that a frequency equal to twice the switching frequency is applied to the active switching switch Qc.

Figure 3 is a simplified electrical diagram of an example of a CFFB converter according to one embodiment.

The converter of FIG. 3 comprises elements common with the converter of FIG. 1. These elements will not be described again below. In the following, only the differences with respect to the converter of FIG. 1 will be detailed.

The converter of figure 3 differs from that of figure 1 mainly in that, in the embodiment of figure 3, the active switching switch Qc has been replaced by an active switching module 300 comprising two switching switches Qc1 and Qc2 connected in parallel. The switches Qc1 and Qc2 are for example identical except for manufacturing dispersions.

More specifically, in this example, switch Qc1 has a first conduction node connected to a first conduction node of switch Qc2 and a second conduction node connected to a second conduction node of switch Qc2. In the example shown, the first conduction nodes of switches Qc1 and Qc2 are connected, for example connected, to node A, and the second conduction nodes of switches Qc1 and Qc2 are connected, for example connected, to node E.

In the example of Figure 3, the active switching branch 112 further comprises, replacing the diode Dc of Figure 1, two diodes Dc1 and Dc2 respectively connected in parallel with the switch Qc1 and in parallel with the Qc2 switch. More particularly, in this example, diode Dc1 has its anode connected, for example connected, to node A and its cathode connected, for example connected, to node E, and diode Dc2 has its anode connected, for example connected, to node A and its cathode connected, for example connected, to the node E.

The switches Qc1 and Qc2 are for example transistors of the same type as the transistors Q1, Q2, Q3 and Q4. The diodes Dc1 and Dc2 correspond for example respectively to body diodes intrinsic to the transistors Qc1 and Qc2. As a variant, a single diode Dc, distinct from transistors Qc1 and Qc2, can be provided to replace diodes Dc1 and Dc2, diode Dc being connected in a manner similar to what has been described in relation to FIG. 1.

In the example of Figure 3, the control circuit 140 is adapted to control the switches Q1, Q2, Q3, Q4, Qc1 and Qc2 of the primary stage.

FIG. 4 schematically illustrates an example of an operating mode of the converter of FIG. 3. FIG. 3 more particularly represents the evolution, as a function of time, of a control signal S14 applied to the control nodes switches Q1 and Q4, a control signal S23 applied to the control nodes of transistors Q2 and Q3, a signal Sc1 applied to the control node of transistor Qc1, and a signal Sc2 applied to the node control of transistor Qc2. The signals S14, S23 are for example identical or similar to what was previously described in relation to FIG. 2. The signals Sc1 and Sc2 are for example voltages applied respectively between the gate and the source of the transistor Qc1 and between the gate and the source of transistor Qc2.

The signals S14, S23, Sc1 and Sc2 are for example generated by the control circuit 140 and/or by close control circuits (not detailed) connected to the control nodes of the switches Q1, Q2, Q3, Q4, Qc1 and Qc2 .

In this example, each of the switches Q1, Q2, Q3, Q4, Qc1 and Qc2 is controlled in the closed state (on) when the signal S14, S23, Sc1 or Sc2 applied to its control node is in the high state. , and is controlled in the open state (off) when the signal S14, S23, Sc1 or Sc2 applied to its control node is in the low state.

In the example of FIG. 3, the control of switches Q1 and Q4 on the one hand (signal S14), and Q2 and Q3 on the other hand (signal S23), is identical or similar to what has been described in relation with Figure 2.

Switches Qc1 and Qc2 are kept open throughout the duration of the simultaneous closing phases of switches Q1, Q2, Q3 and Q4, that is to say between times t1 and t2 on the one hand, and between times t3 and t4 on the other hand.

Switch Qc1 is kept closed for at least part of the closing phase of switches Q1 and Q4 and opening of switches Q2 and Q3, i.e. between times t2 and t3. Switch Qc2 is kept open for the entire duration of the closing phase of switches Q1 and Q4 and opening of switches Q2 and Q3, that is to say between times t2 and t3.

Switch Qc2 is kept closed for at least part of the closing phase of switches Q2 and Q3 and opening of switches Q1 and Q4, i.e. between times t4 and t5. Switch Qc1 is kept open for the entire duration of the closing phase of switches Q2 and Q3 and opening of switches Q1 and Q4, i.e. between times t4 and t5.

Thus, each of the switches Qc1 and Qc2 is controlled in switching at a frequency equal to the chopping frequency f of the converter. By way of example, signal Sc2 is complementary to signal S14, and signal Sc1 is complementary to signal S23.

Thus, in the embodiment of Figures 3 and 4, compared to the embodiment of Figures 1 and 2, the switch Qc controlled in switching periodically at a frequency Fc=2*f with a duty cycle of switching Rc, has been replaced by two switches Qc1 and Qc2 each controlled periodically at the same frequency Fc1c2=Fc/2 with the same switching duty cycle Rc1c2=Rc/2, the commands of the switches Qc1 and Qc2 being in opposition to phase.

This allows, as in the example of Figures 1 and 2, that the switching of the switches Q1, Q2, Q3 and Q4 of the bridge H1 from the open state to the closed state are carried out during zero crossings of the voltage at their terminals, thus limiting the switching losses in the H1 bridge.

In addition, an advantage of the embodiment of Figures 3 and 4 compared to the example of Figures 1 and 2 is that the switching speed of the switches Qc1 and Qc2 is equal to the chopping frequency f of the converter, that is that is to say half as high as the switching frequency of the switch Qc of the converter of FIG. 1. As a result, the heating of the switches Qc1 and Qc2 is limited compared to that of the switch Qc. This allows operation at a higher switching frequency, for example greater than 300 kHz, which in particular makes it possible to reduce the dimensions of the transformer T and of the input inductor L1.

By way of example, in the case where the switches Q1, Q2, Q3, Q4, Qc1 and Qc2 are gallium nitride transistors, and for a switching frequency of 400 kHz and use of the converter at an ambient temperature of 70°C, the internal temperature of switches Q1, Q2, Q3 and Q4 reaches approximately 96°C, and the internal temperature of transistors Qc1 and Qc2 reaches approximately 101°C, which remains acceptable for this type of transistor.

At identical switching frequencies, the reduction in the internal temperature of transistors Qc1 and Qc2 also makes it possible to reduce switching losses compared to the example of figures 1 and 2.

Another advantage of the example embodiment described in relation to FIGS. 3 and 4 is that the control signals Sc1 and Sc2 of the switches Qc1 and Qc2 are in practice simpler to generate than the control signal Sc of the switch Qc of the example of Figures 1 and 2. Indeed, in practice, in the example of Figures 3 and 4, the control signals Sc1 and Sc2 can be complementary to the control signals S23 and S14 respectively, and are then relatively simple to be generated, whereas, in the example of FIGS. 1 and 2, the generation of the signal Sc requires the implementation of an XOR function on the complements of the signals S23 and S14, a function which can be relatively complex to implement.

It will be noted that the proposed solution consisting in replacing a switch intended to be controlled in switching at a frequency Fc with a duty cycle Rc by two switches connected in parallel controlled in switching at a frequency Fc/2 with a duty cycle Rc/2, with a phase shift of π between the control signals of the two switches, can apply more generally to any switching device that can benefit from a reduction in the switching frequency of the switches, for example in a switching converter, or, more generally, in any electronic device comprising a periodic current and/or voltage switching device. For example, in the converter of FIG. 3, each of the switches Q1, Q2, Q3 and Q4 can be replaced by two switches connected in parallel and controlled in phase opposition at a frequency f/2 with a duty cycle R/2. This solution also makes it possible, for a given switching frequency of the switches, to increase the chopping frequency of the system, which makes it possible to reduce the size of the passive components of the system, for example the input inductance and the transformer.

Furthermore, the switching frequency of the switches can be reduced even further by increasing the number of switches connected in parallel, as explained in more detail below in relation to figures 5 and 6.

Figure 5 is an electrical diagram of another example of a switching module according to one embodiment.

The switching module of FIG. 5 comprises three switches Qc1, Qc2, Qc3 connected in parallel between conduction nodes NA and NB of the module. In this example, each of the switches Qc1, Qc2, Qc3 has a first conduction node connected to the node NA of the module and a second conduction node connected to the node NB of the module. The switches Qc1, Qc2, Qc3 each comprise a specific control node connected to a control circuit 500 (CTRL) of the module.

FIG. 6 schematically represents an example of control signals of the switching module of FIG. 5. FIG. 6 more particularly represents the evolution as a function of time of signals Sc1, Sc2, Sc3 applied respectively to the control nodes of switches Qc1, Qc2, Qc3. In this example, each switch is closed when the signal applied to its control node is high and is open when the signal applied to its control node is low.

In this example, the switches Qc1, Qc2, Qc3 are switched periodically at the same frequency Fc, with the same switching duty cycle Rc, preferably less than 1/3, and with a phase shift of 2π/3 between the commands of the various switches. Preferably, the cyclic switching ratio Rc of each of the switches Qc1, Qc2, Qc3 is less than 1/3.

Thus, the switching module of FIG. 5 makes it possible, by using only switches controlled at a switching frequency Fc, to obtain an operation equivalent to that of a single switch controlled in switching at a frequency equal to 3*Fc with a cyclical switching ratio equal to 3*Rc.

More generally, this operation can be adapted whatever the number N of switches connected in parallel, with N an integer greater than or equal to 2. The N switches Qc1, ... QcN are then controlled in switching periodically at the same frequency Fc, with the same cyclic switching ratio Rc, preferably less than 1/N, and with a phase shift of the order of 2π/N between the commands of the various switches (that is to say that the signals of respective control Sc1, ... ScN of switches Qc1, ... QcN are phase shifted two by two by approximately 2π/N, or, in other words, that each of the control signals Sc1, ... ScN is phase shifted by +2π /N with respect to one and only one other of the control signals Sc1, ... ScN).

This makes it possible, by using only switches controlled at a switching frequency Fc, to obtain an operation equivalent to that of a single switch controlled in switching at a frequency equal to N*Fc with a switching duty cycle equal to N* RC.

Various embodiments and variants have been described. Those skilled in the art will understand that certain features of these various embodiments and variations could be combined, and other variations will occur to those skilled in the art. In particular, the embodiments described are not limited to the examples described above in which the switches of the switching module are gallium nitride transistors. More generally, the embodiments described can apply to any type of switch, for example MOS transistors based on silicon or silicon carbide, bipolar transistors, but also to switches other than transistors, for example electromechanical relays.

Claims (10)

Switching module comprising N switches (Qc1, Qc2; Qc3) connected in parallel, with N integer greater than or equal to 2, and a control circuit (140; 500) configured to apply to the N switches respectively N control signals (Sc1, Sc2; Sc3) periodic phase-shifted two by two by 2π/N. Module according to Claim 1, in which the N control signals (Sc1, Sc2; Sc3) have the same frequency and the same duty cycle. Module according to Claim 2, in which the N control signals (Sc1, Sc2, Sc3) have a duty cycle of less than 1/N. Module according to any one of Claims 1 to 3, in which the N switches (Qc1, Qc2, Qc3) are gallium nitride transistors. Switching converter comprising an active switching branch (112) comprising a switching module according to any one of claims 1 to 4. Converter according to claim 5, in which the number N of switches of the switching module is equal to two. Converter according to claim 6, further comprising, in parallel with the active switching branch (112), a controlled H-bridge (H1), the control circuit (140) being configured for, at each period of a cycle of operation of the converter, successively:

1. controlling first (Q1, Q4) and second (Q2, Q3) diagonals of the H-bridge (H1) simultaneously in the closed state;
2. controlling the first diagonal (Q1, Q4) in the closed state and the second diagonal (Q2, Q3) in the open state;
3. driving the first (Q1, Q4) and second (Q2, Q3) diagonals simultaneously to the closed state; And
4. controlling the first diagonal (Q1, Q4) in the open state and the second diagonal (Q2, Q3) in the closed state,

the control circuit (140) being further configured to:

• during step a), maintaining a first (Qc1) and a second (Qc2) of the two switches of the switching module simultaneously open;
• during step b), keeping the first (Qc1) and second (Qc2) switches respectively closed and open;
• during step c), keeping the first (Qc1) and second (Qc2) switches simultaneously open; And
• during step d), keeping the first (Qc1) and second (Qc2) switches respectively open and closed.

Converter according to claim 7, further comprising an input inductor (L1) in series with the H-bridge (H1). Converter according to Claim 7 or 8, in which the assembly comprising the active switching branch (112) and the H-bridge (H1) forms a primary stage (110) of the converter, the converter further comprising a transformer (T) arranged to transmit modulated power supplied by the primary stage and a secondary stage (130) arranged to demodulate the power transmitted by the transformer. A converter according to any one of claims 5 to 9, wherein the active switching branch (112) further comprises a capacitor (Cc) in series with the N switches of the switching module.