FR2969847A1 - REACTIVE ENERGY COMPENSATION CIRCUIT AND METHOD IMPLEMENTED IN SUCH A CIRCUIT - Google Patents

Abstract

The invention relates to a circuit for compensating a reactive energy absorbed by a variable power load, this compensation circuit being arranged in parallel with the variable power load with respect to an alternating voltage source. This circuit comprises: - at least one compensation capacitor for supplying reactive energy to the load, - an absorption coil for absorbing a surplus of reactive energy, - a static converter controlled according to a switching operation, this converter statics consisting of power transistors for supplying the absorption coil generated by the static converter in the supply network; an anti-harmonic filter for eliminating harmonic components of the current supplying the load, and a regulation module for controlling the power transistors from a duty cycle which is an image of the phase shift between the voltage and the current of said voltage source, and so as to impose at any time a zero phase shift between the voltage and the current of said voltage source.

Description

1 "Reactive energy compensation circuit and method implemented in such a circuit."

The present invention relates to a compensation circuit of a reactive energy absorbed by a variable power load. It finds a particularly interesting application in domestic consumption or more generally "general public". The consumption of each home with appliances equipped with small motors (refrigerator, freezer, dishwasher, washing machine, etc.) or inductors (induction or wave oven) includes a large part of the reactive energy inherent in the operation of these devices. In general, any electrical appliance using an alternating current involves two forms of energy: active energy and reactive energy, the whole constituting the apparent energy supplied by the electrical network. Active energy provides useful work in mechanical or thermal form. Reactive energy makes it possible to create magnetic fields, it supplies electrical and magnetic circuits such as capacitors and coils included notably in asynchronous motors, transformers, converters, etc. In order to account for the consumption efficiency of the reactive power of an electrical apparatus, a power factor commonly called cosine phi (cos (p) is used which is the ratio between the active power consumed and the apparent power supplied, a power factor close to 1 indicates a low reactive power consumption and optimizes the operation of an installation In the range of small powers, the principle of power factor correction is widely used The general synoptic is composed of cascaded converters: a typical rectifier Single-phase Graetz bridge and a storage switching converter, the load is almost equivalent to pure resistance The depollution of the network is obtained by the insertion of special filters. The switching frequency is well above 20kHz. The power of the apparatus is less than 10kW. These converters generally use ultra fast Mosfet transistors. For example, document US5751138 discloses a conditioner for compensating the reactive energy and the harmonics of the mains current. The reactive energy is compensated by means of a "stepped" wave inverter consisting of GTO thyristors. Also known is US5187427 which discloses a static compensator of reactive energy. It notably describes an automatic regulation by capacitor bank. This compensator comprises an inverter which integrates IGBT transistors connected in pairs between the collector of one and the transmitter of the other. Document US Pat. No. 7,335,297 describes a system with two circuits. A first thyristor-based circuit for performing the reactive energy compensation function. A second circuit for compensating harmonics is realized by means of an active filter associated with a passive filter, the assembly being parallel to the first circuit. The systems of the prior art have many components ls which penalize the power consumption and make these systems bulky.

The present invention aims to overcome the aforementioned drawbacks by proposing a new solid conditioner, compact and inexpensive. Another object of the invention is a conditioner capable of compensating reactive energy for small electrical equipment and household appliances.

At least one of the aforementioned objects is achieved with a circuit for compensating a reactive energy absorbed by a variable power load, this compensation circuit being arranged in parallel with the variable power load with respect to a voltage source. alternative. According to the invention, this compensation circuit, also called a conditioner, comprises: at least one compensation capacitor for supplying reactive energy to the load; an absorption coil for absorbing a surplus of reactive energy; static converter controlled according to a switching operation, this static converter consisting of power transistors for supplying the absorption coil generated by the static converter in the supply network; An anti-harmonic filter for eliminating harmonic components of the current supplying the load; and a regulation module for controlling the power transistors from a duty cycle which is an image of the phase shift between the voltage and the current. of said voltage source, and so as to impose at any time a zero phase shift between the voltage and the current of said voltage source. The static converter according to the invention is of the chopper type for converting an AC signal to an AC signal. The compensation capacitor is a capacitor C sized to compensate for at least the maximum reactive energy required by the load. The absorption coil may be an inductance coil for example activated only to absorb any excess reactive energy within the system. The static converter may comprise two arms with power transistors for essentially supplying the absorption coil. The harmonic filter preferably attenuates any harmonic components generated by the static converter. This filter may be low-pass type and have the largest portion of the compensation capacitor.

With the compensation circuit according to the invention, the capacitor and the coil are designed to have a unit coscp at any time for a variable power load. This reduces energy consumption in residential homes and in industry. The conditioner thus developed having only one converter, is of simplified structure and topology. Compared to the systems of the prior art, the compensation circuit according to the invention is simple to design, economical, compact, complies with electromagnetic compatibility standards and can therefore be placed inside the variable power load which is advantageously a low power appliance. This compensation circuit according to the invention can also be placed outside the variable power load. In the second case, it can then compensate for a group of devices whose number is limited by the maximum compensation capacity.

According to a first advantageous variant of the invention, the filter is arranged at the input of the compensation circuit according to the invention, and this filter comprises a circuit RLC for Resistance-Inductance-Capacitor, the capacitor of this circuit RLC being said at least one capacitor used for compensation. This reduces the number of components used for the design of the compensation circuit: the same capacitor is used for compensation and filtering. With such a compensation circuit, it is not necessary to add sector filters to clean the grid current because the filter according to the invention is an integral part of the structure of the compensation circuit, resulting in increased compactness. . The realization of the control-command part is totally digital thanks to a DSP, so reliable and robust. According to a second variant of the invention, the absorption coil and the compensation capacitor are disposed at the output of the static converter. Preferably, the static converter supplies the capacitor and the coil directly. The absorption coil may be arranged in parallel or in series with the compensation capacitor. According to an advantageous characteristic of the invention, the static converter comprises two arms whose midpoint constitutes an output for supplying the absorption coil; each arm comprising two IGBT transistors connected in series by their emitter, and each IGBT transistor being directly in parallel with a diode disposed in opposite direction. Such a structure allows for efficient conversion. According to the invention, pulse width modulation (PWM) regulation can be carried out, the regulation module can then comprise: an angle measuring module which determines the phase difference between the voltage and the current of the source of food ; A module for transforming a DC voltage level differential from the previously measured angle; a generator that converts the differential DC voltage level previously obtained into a duty cycle; and a control circuit for the gates of the power transistors. And the angle measuring module can comprise: - a measuring circuit of a phase angle between the voltage and the power source current, - a circuit for transforming the phase angle into phase shift voltage, s - a comparison circuit between this phase shift voltage and a preset reference voltage, and - a proportional-integral type corrector fed by the comparison circuit and generating a continuous signal for the duty cycle generator. Advantageously, a modulated hysteresis regulation is carried out. In this case, the duty cycle generator may comprise: an adder for adding a triangular carrier with the DC signal output as a proportional-integral corrector output, a shaping circuit that generates the control triggers, and ls - a switching circuit which appropriately distributes the gate control signals of the power transistors. This makes it possible to improve the shape of the current wave delivered by the network. In this case, the harmonic distortion rate of the current is further improved and the power factor of the compensation circuit easily tends to unity. According to an advantageous characteristic of the invention, the regulation module is configured to work with a switching frequency fsw of less than five kHz, preferably between 1 and 2 kHz. In particular, it is the frequency of the triangular carrier that is set below 5 kHz, preferably between 1 and 2 kHz. With such a low switching frequency and hard commutation, the IGBT transistor is highly efficient. When using IGBT transistors, they are chosen such that gate resistors of these transistors have values in the range of 5 to 33 ohms so as to reduce common mode currents. The IGBT transistors may be preferably those of the trade already having internally an integrated diode and placed in reverse. Thus the diodes used in the converter are not independent chips reported to the converter.

The compensation capacitor can be subdivided into two elements: a large part of the anti-harmonic filter (at least 95%) and a smaller part (at most 5%) placed in parallel with the absorption coil. s The control module can be fully supported by a Dsp using three waves to fully perform its function: the voltage and current waves of the network, and the current wave in the absorber. This DSP can:. control to have a zero phase shift at any time between the voltage and the current of the power source; and. provide the control triggers for the drivers of the transistors; . ensure zero current flow without compromising the integrity of the power switches.

It is expected a close control strategy by generating the control triggers for the drivers of the transistors, principle of dead time at zero crossing of the current in the absorption coil is completely ensured by the Dsp. According to another aspect of the invention, there is provided a method implemented in a compensation circuit as described above. According to the invention, the power transistors are controlled so that the voltage across the absorption coil is kept substantially equal to the product between the duty cycle and the voltage of said voltage source. The converter then behaves like an autotransformer, whose transformation ratio is given directly by the continuously variable cyclic ratio. The voltage delivered to the absorption coil can thus be continuously regulated, so that an automated process is provided for continuous adjustment of the energy absorbed by the absorption coil.

An automated process is thus performed for the continuous adjustment of the reactive energy involved. An all-digital DSP can be used to fully digitalize the regulation module, a close control, a generation of the gate control triggers. power transistors, and control of the zero crossing of the current of the absorption coil. The DSP can be used to carry out control functions from three waves: the voltage and mains waves, and the current wave in the absorption coil. The static converter may comprise two arms whose midpoint constitutes an output for supplying the absorption coil; each arm comprising two IGBT transistors connected in series by their emitter, and each IGBT transistor being associated with an internal diode head to tail. The control module can be configured by means of a DSP-based module so that the currents of both arms are in perfect encroachment.

Advantageously, the regulation module is configured so that the currents of the two arms are in perfect encroachment.

Other advantages and features of the invention will appear on examining the detailed description of a non-limiting embodiment, and the appended drawings, in which: FIG. 1 is a schematic view of a system compensation device according to the invention; FIG. 2 is a graphical representation of the physical quantities, voltage and current, during a reactive energy compensation; Fig. 3 is a schematic diagram of the static converter used in the compensation system according to the present invention; Figures 4 and 5 show configurations and operating waves obtained by the qualitative analysis. The quantities indicated in these figures are those defined in FIG. 3. FIG. 6 is a view of the dynamic switching tracking of the IGBT transistors of the converter of FIG. 3; Fig. 7 is a view illustrating the waves of the control voltages of the IGBT transistors and currents in the converter of Fig. 3; Fig. 8 is a view illustrating the voltage and current waves at the terminals of an IGBT switch of the converter of Fig. 3; FIG. 9 gives the general control diagram, analog or digital, of the compensation; Fig. 10 is a view illustrating curves of mains voltage and mains and load currents for a cos) of 0.8; FIG. 11 is a view illustrating curves of the network voltage and the network and load currents for a cos of 0.3. FIG. 12 is a diagram showing the equivalence between (P '' P) and (R ') Fig. 13 shows an electronic circuit diagram corresponding to the whole of the anti-harmonic filter (LF, C), to the switches one of which is controlled during the raw time interval and the other during time interval rsw and to the absorption coil (Ra, La), and another diagram of this sinusoidal model set in which the converter behaves as a transformation ratio autotransformer equal to the cyclic ratio ls a, FIG. 14 is a diagram of an electrical circuit for the harmonic current of switching frequency fsw, taking into account all the real elements of the assembly, ie: the compensation capacitor C, the elements of the coil of the harmonic filter (resistance internal RF, inductor LF), l a load (R, L), the internal impedance of the network (p, to), Figure 15 is the equivalent diagram of the electrical circuit of the system referred to the frequency of the supply network fo, Figure 16 is a simplified diagram Figure 17 is a Bode curve for Ih, and the absorber Figure 18 is a Bode curve of E for the low load. Although the invention is not limited thereto, FIG. 1 shows a compensation circuit 3 according to the invention applying to a domestic appliance appliance. This device constitutes a load 1 supplied from the electrical network 2 in voltage v and current i reciprocating. Load 1 absorbs sinusoidal current i with more or less reactive energy. This load 1 has a variable equivalent load which can be characterized by an absorbed active power P and a cosine of the load argument (p) The compensation circuit 3 according to the invention is arranged in parallel with the load 1 relative to the 2. The input of the compensation circuit is represented by the connections E1 and E2, which means that the voltage v across the load and the same as the voltage between the connections E1 and E2. i of the electrical network 2 is divided into a current Icharge supplying the load 1 and a current (not shown) to the compensation circuit via E1 The compensation circuit 3 comprises at the input a filter 4 illustrated schematically and comprising a coil 5 receiving the input current via E1 and at least one capacitor C disposed between the output of the coil 5 and the input E2.The coil 5 is characterized by an inductance Lf and a resistor Rf. channel 4 supplies a static converter 6 which generates a voltage equal to av across an absorption coil 7; a being a duty cycle obtained from the voltage v and the current i of the electrical network 2. The absorption coil 7 is characterized by an inductance La and a resistor Ra. According to the invention, the capacitor C represents both a compensation capacitor for transmitting reactive energy to the load 1, and at the same time a filtering capacitor. Advantageously, these two functions are grouped together in the same capacitor, but they can be separated with: a filtering capacitor with a capacitance adapted to produce the filter 4 and arranged as shown in FIG. 1, and at least one compensation capacitor C1 or C2 of a capacity adapted to compensate for the reactive energy of the load and arranged in a first variant Cl in series with the absorption coil 7, and in a second variant C2 in parallel with the coil of absorption 7, as shown in dotted lines in FIG. 1. The converter 6 is controlled by a control circuit 8 which receives the duty cycle a from a duty cycle generator 9. A comparator system 10 measures in real time the voltage v and the current i of the electrical network 2 so as to transmit a voltage setpoint to the duty cycle generator 9. A household appliance operates in general between two load modes: at full load identified by (IMax, (PMin), and at idle or at rest with (IMin, (PMax) - In these two intervals of load variations, compensation circuit 3 reacts from such that the mains voltage v and the current flow i always remain in phase. In this case, the current delivered by the network may be lower than that before the compensation of the reactive energy: del link, it would be * COSq '. For the full load operating point (IMax, (PMin), the compensation circuit 3 is practically decommissioned, so the duty cycle io is close to zero In this situation, the capacitor C of the filter compensates the reactive energy absorbed by the load by supplying the reactive energy Qc, such that the total reactive energy of the installation is zero, According to the compensation vector diagram given in FIG. 2, the current delivered by the network i is less than charge current Icharge- At full load, the maximum active power absorbed is maximum and the reactive power absorbed is also maximum and compensated by the reactive energy Qc of the capacitor C. For the point of idle operation (IMin, (Max), the excess of reactive energy Qa is dissipated in the absorption coil 7. The excess reactive energy Qa is the difference between the reactive energy at maximum QMax and the reactive energy at least QMin- This is done for a given value of the duty cycle a. The current delivered by the network in this case remains close to that absorbed by the load. The static converter then behaves like a variable transformation ratio transformer, represented by the duty cycle a. The structure of the converter according to the invention is illustrated in FIG. 3. It consists of two arms, series 11 and parallel 12, connected at a common point A representing the output of the converter 6. On the series arm 11, we see a switch T1 constituted by an IGBT transistor in parallel with a diode. The emitter of the IGBT transistor is connected to the anode of the diode, the collector of the IGBT transistor being connected to the cathode of the diode. Subsequently, such an assembly consisting of an IGBT transistor and a diode arranged in parallel in the manner indicated above, will be called "IGBT switch". Thus, each arm has two IGBT switches arranged in series and connected to each other by their emitter. The arms 11 and 12 comprise two 11 IGBT transistors T1 and T2, T3 and T4 with their internal diodes back-to-back, these transistors are placed in series and connected by their source. The output signal at point A is a modulated current signal as a function of the conduction and interruption of the various switches of the inverter. These IGBT switches are controlled by control signals C1, C2, C3 and C4 coming from the control circuit 8. In FIGS. 4, 5, 6 and 7, operating configurations and waves obtained by the analysis are shown. qualitative. The quantities indicated in these figures are those defined in FIG. 3. FIG. 4 shows the configuration sequences for illustrating the high viability switching of the configuration represented by the transistor T1 and the diode d2 towards the configuration represented by the transistor T4. and the diode d3, and vice versa with E> 0 and Io> 0. Thus, Figure 4 (a) indicates the sequence with T1 passing: the current then passes through T1 and d2. In FIG. 4 (b), the control of transistor T4 is set to "1", the previous sequence continues. In FIG. 4 (c), after a time interval Δt, the T1 blocking control causes an encroaching phenomenon between the transistors T1 and T4. Figure 4 (d) shows the two switches T4 and d3 conducting the current Io. The control at 1 of the switch T1 resets the phenomenon of encroachment between T4 and T1 (Fig.4 (e)). When T1 is conducting completely, the diode d3 is blocked because its cathode anode voltage is practically equal to -E. Setting the T4 command to 0 does not change the previous configuration which is identical to that of Figure 4 (a). The waveforms associated with this analysis are given in FIG. 5. FIG. 6 shows the configuration sequences to illustrate the high viability switching of the configuration represented by transistor T2 and di di to the configuration represented by FIG. transistor T3 and diode d4, and vice versa with E> 0 and Io <0. Thus, Fig. 6 (a) indicates the sequence with passing T2: the current then passes through T2 and di. While keeping the control c2 = 1 in FIG. 6 (b) for a time interval, as soon as the control of the transistor T3 goes to "1", the phenomenon of encroachment occurs between the transistors T2 and T3. . When the current in T2 and di vanishes, the anode-cathode voltage of di is approximately equal to -E, so di is blocked. In Fig. 4 (c), the two switches T3 and d4 leading the current I0 and setting "0" to the command of T2 do not change the previous configuration. In FIG. 6 (d), setting the control of the switch T2 does not change the configuration of FIG. 6 (c) either. In Fig. 6 (e), the blocking control of T3 resets the encroaching phenomenon between T3 and T2. When T2 drives completely, the diode d3 is blocked because its cathode anode voltage is practically equal to -E. The configuration then returns to that of Figure 6 (a). The waveforms associated with this analysis are given in FIG. 7. For E <0 and Io> 0 and E <0 and Io <0, the same analyzes are carried out. FIG. 8 illustrates voltage and current waves at the terminals of a switch during the sign change of the charging current. On the left panel, no dead time is applied, ie At - 0, between the commands Cl and C3, and one causes a sudden cut of the current of the load. A peak voltage of about 800V then appears for a period of 33.82ps across T1. On the right board, a dead time At - 1 / (2fsw) has been set: the voltage peak has completely disappeared and the change of sign of the current is carried out without problem. This avoids any sudden opening of the converter during the change of sign 20 of the charging current. The converter according to the invention is of direct type. It can operate in pulse width modulation (PWM), according to the principle of comparison of a sinusoid and a triangular frequency signal. Due to the switching of the converter transistors, harmonic components of current are generated in the network. The main function of filter 4 is to eliminate these harmonic components. In addition, in order to reduce the converter losses, essentially switching, the switching frequency is chosen relatively low, between one and two kHz. In addition, such frequency values make it possible to select gate resistors which reduce common mode currents. The converter input filter 4, the low switching frequency and the gate resistance values give the compensation circuit according to the invention a very good behavior in comparison with the Electromagnetic Compatibility (EMC) standards for domestic appliances. to differential mode currents and common mode currents. Laminar wiring with grounding is also provided to reduce common mode currents in the converter. By way of non-limiting example, copper sheets (thickness 0.5 mm) and insulating sheets (thickness 0.035 mm) can be used for producing planar conductors. These drivers, sometimes called "bus bar", are used on the converter arms.

FIG. 9 shows a synoptic of the control of the compensation making it possible to impose at any instant a zero phase shift between the waves of the mains voltage and the current output. We find the comparator system 10 and the duty cycle generator 9. The set works completely analog. A measurement module 13 is distinguished for measuring the phase shift between the voltage v and the current i of the electrical network. The phase shift cp generated is then transformed into a voltage Vcp by a transformation module 14. This voltage Vcp is compared with a reference voltage for a zero phase shift by means of a comparator 15. The output of the comparator 15 supplies a proportional regulator. integral 16 which in turn powers a module 17 to generate the duty cycle a. Results obtained by simulation are given on: FIG. 10 for a point of load operation, with P = 2000W and coscp = 0.8, the duty ratio being 0.05, and FIG. 11 for a point of no-load operation, with P = 200W and coscp = 0.3, the duty ratio being 0.37. We see that the final phase shifts are practically zero, and the currents are sinusoidal. For the point of operation under load, the grid current is lower than that of the load, ie 2030 VA instead of 2500 VA. The compensation circuit according to the invention is substantially disconnected because the duty cycle is 0.05. Therefore, the network current has no harmonic components. For the point of no-load operation, the network current has low harmonic components: their value remains below the limits allowed by the EMC standards. The value of the grid current is slightly higher than that of the load because of the resistive components of the filter and absorber coil elements (5 and 7). In this case, the power supplied is 778 VA instead of 667 VA. Figure 12 shows a diagram illustrating the equivalence between s the torque (P, cos (p) and a circuit of passive elements (RL) of the load.The variable equivalent load is characterized by the power absorbed P and the cosine The equations linking these two pairs are: R = P * (l + tan2 (p) V 2 * tan (p L. _ We will now describe non-limiting calculation methods of values characteristics of the circuit according to the invention.

ls 1- Calculation of the compensation capacitor C. The value of the reactive energy to be compensated must be the largest value between (V.Imax.9dn (Pmin) and (v.1min.slnq'max), to take account of other inductive elements of the assembly, as in the case of active filters, a setting coefficient kQ of the global reactive energy is given, in which case, in a purely sinusoidal mode, the value of the compensation capacitor C can be calculated from full load, according to: 25 2- Current harmonic filter and absorption coil 2-1: The filter elements The switching operation of the converter generates harmonics of currents with the same frequency as the switching frequency fW of the converter A Butterworth type 2 simple structure lowpass filter is placed between the grating and the converter which is buffered by the capacitor. In Figure 13, it is noted that the capacitor C has an impedance. previously defined is part of this filter. For the second order Butterworth filter, we have the following relations: 0.707 2 * n * f * ~ r (Z a ~ az (3) C = 10 l 1.414q2 a 2 * 'n * f ~ r The value of the absorption coil will be defined for a cyclic ratio aMax, verifying the characteristic impedance of the filter, such that: IdiLF The maximum Ca Let: L = a 2 * LF a Max r The cutoff frequency of the filter will be chosen rather close or 15 slightly above 5OHz in view of a strong attenuation of the current harmonics for the switching frequency fw, the filtering of the low frequency components especially 150 Hz and 300 HZ 2-2: The higher harmonics of the network current.

Figure 14 shows the system by reporting all the passive components of the elements. In this Figure 12, by posing: zc _- - Jcc '': the impedance of the compensation capacitor; - zres = P + lie) the internal impedance of the network; - zeh: the impedance of the load; ZLF = rF + iIF60: the impedance of the filter coil; he comes: LF = (4) 1 - 16-ihr __ Zc * Zch

ih Zr * Zch Zr + Zch + ZLF + Z, Zr + Zch For the switching frequencyfsw, the calculation of the different elements Ihr (i2lIffsw) ~ 0s must lead us to the relation: Ih. Note that because of the non-linear behavior of the converter, the effects of low frequency switching are not necessarily eliminated in the network current. 2-3: Calculation of the reactive energy. As a first approximation, it is considered that the converter has no losses. By neglecting the effects of the splitting, the equivalent diagram of the absorption coil brought back to the source side is given in figure 15. The choice of the absorption coil must cancel the reactive energy for given cyclic ratios for the two loads . The impedance ls equivalent of the system vis-à-vis the network is written: * (Za ZC + ZLF) * zZa + ZC ch Ztotak - + Zr * (Za ZC + ZLF) + z Za + ZC ch (6 ) The resolution of the equation Im (Ztotak) - ° makes it possible to find: - a cyclic ratio ainf for the high load; 20 - A cyclic ratio as ° p for the low load. 2-4. Calculation of the output voltage of the converter. Due to the choice of the critical frequency of the filter rather close to 50 Hz, for the choice of the power transistor sizes, the phenomenon of antiresonance must be perfectly mastered. Thus, according to FIG. 16, there is: (5) - Z * Z Zch * (ZLF + Za a + ZC C Za ZC Zch + (ZLF + Za ZC Za + ZC + ZC + ZC + ZC Zch * (zLF + ZaZC) Za + ZC a + C Equations 1, 2, 3 and 4 make it possible to calculate the elements of the circuit according to the invention, for the load points considered.

The harmonic distortion rate (abbreviated THD) According to an advantageous embodiment of the invention, the preferred dimensioning of the circuit according to the invention is obtained with harmonic distortion rates (THD) currents of less than 10%, and a peak voltage across the absorber of less than 500V. In this case, we choose ideally:

fc = 55 Hz;

- kQ = 1.4; am. = 0.8 As a non-limiting example, the actual components of the circuit may be the following:

- IGBT calibrations: 600V; 30A

- For the filter: C = 140 pF; LF = 58mH; LA = 45mH.

Which give :

a cut-off frequency fc = 55.85 Hz; - A duty cycle max: OEmax = 0.8335 Thus, Figures 17 and 18 illustrate performance response curves of the circuit according to the invention. In Figure 17, we see the curve

Bode of the ratio lhY and it is found that the component at 150Hz is at -15dB, that at 250Hz at -25dB, that of 350Hz at -32dB, that of 1000Hz at -50d B. + Zres

Zch + (ZLF + Za + ZC a c (7) - 18- In Figure 18, we see the Bode curve of the absorber ratio [dB] and there is an overvoltage at the output of the converter.

Of course, the invention is not limited to the examples which have just been described and many adjustments can be made to these examples without departing from the scope of the invention. E

Claims (15)

REVENDICATIONS1. A compensation circuit for a reactive energy absorbed by a variable power load, said compensation circuit being arranged in parallel with the variable power load with respect to an alternating voltage source; characterized in that it comprises: - at least one compensation capacitor for supplying reactive energy to the load, - an absorption coil for absorbing excess reactive energy, - a static converter controlled according to an operation switching mode, this static converter consisting of power transistors for supplying the absorption coil generated by the static converter in the supply network; ls - an anti-harmonic filter for eliminating harmonic components of the current supplying the load, and - a regulation module for controlling the power transistors from a duty cycle which is an image of the phase shift between the voltage and the current of the load. said voltage source, and so as to impose at any time a zero phase shift between the voltage and the current of said voltage source. 2. Circuit according to claim 1, characterized in that the filter is arranged at the input of this compensation circuit, and in that the filter comprises a circuit RLC for Resistance-Inductance-Capacitor, the capacitor of this circuit RLC being said at least one compensation capacitor. 3. Circuit according to claim 1, characterized in that the absorption coil 30 and the compensation capacitor are disposed at the output of the static converter. 4. Circuit according to claim 3, characterized in that the absorption coil is arranged in parallel with the compensation capacitor or in series with this compensation capacitor. 5. Circuit according to any one of the preceding claims, characterized in that the static converter comprises two arms whose midpoint constitutes an output for the supply of the absorption coil; each arm comprising two IGBT transistors serially mounted by their emitter, and each IGBT transistor being directly in parallel with a diode disposed in opposite directions. Circuit arrangement according to one of the preceding claims, characterized in that the regulation module comprises: - an angle measuring module which determines the phase difference between the voltage and the current of the power source; a module for transforming a DC voltage level differential from the previously measured angle; a generator which converts the previously obtained differential voltage level ls into a duty cycle; and a control circuit for the gates of the power transistors. 7. Circuit according to claim 6, characterized in that the angle measuring module comprises: a circuit for measuring a phase shift angle between the voltage and the power source current; a circuit for transforming; the angle of phase difference in phase-shift voltage, - a comparison circuit between this phase-shift voltage and a preset reference voltage, and - a proportional-integral type corrector fed by the comparison circuit and generating a continuous signal for the cyclic report generator. 30 8. Circuit according to claim 6 or 7, characterized in that the duty cycle generator comprises: an adder for adding a triangular carrier with the DC signal output as a proportional-integral corrector output, a circuit forming circuit which generates the control triggers, and a switching circuit which appropriately distributes the gate control signals of the power transistors. 9. Circuit according to any one of the preceding claims, characterized in that the regulation module is configured to work with a switching frequency fsW less than five kHz. 10. Circuit according to claim 9, characterized in that the regulation module is configured to work with a switching frequency fsW between 1 and 2 kHz. io Circuit according to Claim 9 or 10, characterized in that when the power transistors are IGBT transistors, gate resistances of these transistors have values in the range of 5 to 33 ohms so as to reduce currents. common mode. 15 12. A method implemented in a compensation circuit of a reactive energy absorbed by a variable power load, the compensation circuit being arranged in parallel between an AC voltage source and the variable power load and comprising: - at least a compensation capacitor for supplying reactive energy to the load; - an absorption coil for absorbing a surplus of reactive energy; - a static converter controlled according to a switching operation, this static converter being constituted by transistors of power for supplying the absorption coil generated by the static converter in the supply network; an anti-harmonic filter for eliminating harmonic components of the current supplying the load, and a regulation module for controlling the power transistors from a duty cycle which is an image of the phase shift between the voltage and the current of the load. said voltage source, and so as to impose at any time a zero phase shift between the voltage and the current of said voltage source; wherein the power transistors are controlled so that the voltage across the absorption coil is maintained substantially equal to the product between the duty cycle and the voltage of said voltage source. 2969847 - 22 - 13. Method according to claim 12, characterized in that an all-digital DSP is used to fully emulate the regulation module in digital mode, a close control, a generation of the triggers for control of the gates of the power transistors, and a control the zero crossing of the current of the absorption coil. 14. The method of claim 13, characterized in that the DSP is used to perform control functions from three io waves: the voltage and current of the network, and the current wave in the coil. absorption. 15. Method according to one of claims 12 to 14, characterized in that the static converter comprises two arms whose midpoint ls constitutes an output for the supply of the absorption coil; each arm comprising two IGBT transistors connected in series by their emitter, and each IGBT transistor being associated with an internal diode head to tail; and in that the control module is configured by means of a DSP-based module so that the currents of the two arms are perfectly encroached.