FR2710802A1 - Device for driving a load in a vehicle direction indicator device - Google Patents

Abstract

The device comprises a charge/discharge capacitor (CO), a circuit for charging/discharging the capacitor with a given time constant so that its voltage varies between upper and lower limits; a drive circuit (31) for turning on a power transistor (16) in series with the path of the current supplied to a load (10L; 10R) from a DC source (14), which results in the charge being driven when the charging/discharging circuit charges the capacitor, or the voltage at the terminals of the capacitor is higher than or equal to a given voltage defined between the upper and lower limits; a means of detecting overcurrents in order to detect whether the voltage of the transistor is exceeding a certain value; a means of changing the upper voltage limit so as to change this voltage at which the charging/discharging circuit stops charging the capacitor and starts discharging it, in order to cause it to take a value normally higher than a detection voltage stopping the driving when the means of detecting overcurrents detects that the said voltage at the terminals exceeds the said overcurrent detection voltage, and a first means of stopping the driving (26) in order to turn off the transistor and also to keep the charging/discharging circuit in the charging state, by detecting whether the voltage of the capacitor is higher than or equal to the voltage for stopping the driving when this voltage becomes greater than or equal to the drive stop voltage.

Description

 The present invention generally relates to a load driving device comprising a power transistor, and more particularly to a protection circuit for a power transistor which is periodically turned on and off in response to an oscillating signal produced by the power transistor. charge / discharge of a capacitor.

 In the prior art, in any control device for turning on / off flashing lamps, or a variable width pulse width modulation (MILV) control apparatus, a power transistor in series with the current path is provided. used to supply a load with a DC current from a DC power source. This transistor is periodically conductive and non-conductive in response to an oscillating signal by charging / discharging a capacitor.

 In other words, in this type of device, the charge / discharge capacitor is charged / discharged via a resistor in such a way that the voltage across the capacitor varies between two preselected values. This periodic operation of the power transistor to make it conductive / non-conductive is performed in such a way that the transistor is turned on during the charging operation of the capacitor, and not passing during its discharge. Moreover, such a drive signal (MILV signal) having the desired duty cycle is generated by comparing the voltages present across the capacitor, which are modified by the charging / discharging operations, to a reference voltage, and the transistor power is then turned on / off in response to this MILV signal.

 On the other hand, a protection device, such as that described in Japanese Patent Application Laid-Open No. 59-32224 (1984), is employed in the power transistor used in this driver of a charge. More specifically, in order to protect the power transistor against overcurrent or dissipation of overheating, which are due to short-circuiting or disconnection of the current path in which this power transistor is inserted, this protection device operates as follows: at the end of a predetermined time, since the power transistor is turned on, it is appreciated whether overcurrents can pass through the power transistor by checking whether the voltage terminals of this transistor connected to the charge current path is greater than or not to a predetermined voltage. It will be noted that this voltage corresponds to the collector-emitter voltage VCE when a bipolar transistor is used as a power transistor, or this voltage corresponds to the drain-source voltage VDS when a field effect transistor (FET) is used. as a power transistor. When the evaluation indicates that overcurrents can pass through the power transistor, the latter is made non-conductive necessarily and continuously via an interlocking circuit consisting of a thyristor and transistors.

 However, the conventional protection device above raises the problem that the heat dissipation of the power transistor gradually increases and that the power transistor is then likely to be electrically destroyed even if this protective device is used. This is because when either the drive signal of the power transistor is turned off or the power switch is turned off, it is supplied in the OFF state. being separated from the current path between the power supply and the load, the latching operation performed by the latching circuit is initiated, and the power transistor can be turned on again if the next driving operation of the charge has begun, whereby when the power transistor is periodically turned on / off by the driver above, or when the power switch is repeatedly closed, the power transistor power will be subjected to overheating.

 In other words, according to the conventional protection device, after the lapse of a given time, since the power transistor is turned on, an overcurrent caused by the short-circuiting of the transistor is evaluated. charging, or increasing the internal resistance of the power transistor based on the voltage across this transistor, and therefore, the power transistor is made non-conductive. As a result, a current can necessarily flow in the power transistor for a specified time after it has been turned on. Accordingly, when the power transistor is periodically turned on / off by the driver above, or when the power switch is repeatedly closed / open, the current must flow through this switch for a long period of time. preselected time even if the protective device is used. As a result, this current can increase the heat dissipation of the power transistor, and ultimately can electrically destroy the transistor.

 In addition, the conventional protection device is likely to be subjected to the harmful effects of noise, and therefore raises the problem that the power transistor is necessarily made non-conductive even if it is not crossed by an overload; that is, by way of example, that overloads are erroneously detected as a result of the occurrence of noise in the current path in the case of the use of the power transistor, so the inrush circuit will operate, or this circuit will be directly powered by the noise.

 It should be noted that, in order to avoid such an erroneous assessment of the overloads caused by the noise, after ascertaining the overloads by the voltage at the terminals of the power transistor, this judgment signal can be delayed by the integral circuit, etc., to provide the judgment signal delayed at the latching circuit. In this case, a delay circuit such as the integral circuit must be provided separately, but it could not avoid the erroneous operation of the interlocking circuit caused by the noise.

 The object of the present invention is to solve the problems set out above and, consequently, its main object is to provide a protection device for a power transistor, capable of making a correct evaluation, without the harmful effect of noise, because whether or not an overload crosses a power transistor which is periodically turned on / off in response to an oscillating signal produced by charging / discharging a capacitor, as described above, and which is further capable of safely protect the power transistor after the evaluation of the overload.

 To achieve the main object, a charge driving device according to a first aspect of the present invention comprises: a charge / recharge capacitor; a charge / recharge circuit for charging / recharging the capacitor at a given time constant in such a way that the voltage across the capacitor varies between a predetermined upper limit and a pre-established lower limit; a driving circuit for conducting a power transistor connected in series to a current path for supplying a current between a DC power source and a load, thereby driving the load when the circuit charging / discharging causes charging of the capacitor, or the voltage across the capacitor is greater than or equal to a predetermined voltage defined between the upper limit and the lower limit; overload judging means for judging whether or not the voltage at the terminals of the power transistor connected to the current path exceeds the rating voltage of an overload; an upper voltage change means for changing the upper limit of the voltage at which the charging / discharging circuit stops the charging operation of the capacitor and begins discharging it to a value normally greater than a corresponding voltage at the judgment of the stopping of the attack when the means appreciating the overloads appreciates the fact that the voltage at the terminals exceeds the tension of appreciation of the overloads; and a first drive stop means for rendering the power transistor necessarily non-conductive and further for maintaining the charge / discharge circuit in the capacitor charge state by appreciating that the voltage across the capacitor is or not greater than or equal to the judgment voltage of the stopping of the attack when the voltage present across the capacitor becomes greater than or equal to the judgment voltage of the stopping of the attack.

 As described above, the charge driving device according to the first aspect of the present invention includes the driver for making the power transistor conductive / non-conductive in response to the oscillating signal obtained by charging. / discharging the capacitor. In this load driving device, the overcurrent appreciating means will appreciate whether or not the overload crosses the power transistor by checking whether the voltage at the terminals of the power transistor exceeds the value of appreciation or not. overloads. Then, when this means appreciating the overloads appreciates the fact that the voltage across the power transistor exceeds the corresponding voltage of appreciation of the overloads, the means changing the upper limit voltage modifies this voltage to make it pass to the appreciation voltage. stopping the attack, the upper limit voltage at which the charging / discharging circuit employed in the driver stops charging the capacitor to begin discharging. The first means stopping the attack appreciates whether or not the voltage across the capacitor is greater than or equal to the judgment voltage of the stopping of the attack. When the terminal voltage becomes greater than the normally high voltage corresponding to the evaluation of the stopping of the drive, the power transistor is made non-conductive force so as to maintain the charging / discharging circuit to the state of charge of this circuit.

 In other words, according to the present invention, when the voltage present at the terminals of the transistor exceeds the evaluation voltage of the overloads, and the means appreciating the overloads appreciate that the overload passes through the power transistor, the charging circuit / discharge continues to charge the capacitor until the voltage at its terminals becomes greater than or equal to the voltage normally greater than the judgment voltage of the stopping of the attack. When the voltage across the capacitor becomes greater than the stopping voltage of the drive, the first stopping means of the drive renders the power transistor non-conductive. As a consequence, according to the present invention, the duration defined after the means appreciating the overloads appreciates the appearance of an overload, and until the power transistor is made non-conductive, is delayed by the charging time of the capacitor. If the voltage across the power transistor is reduced to the normal voltage during this time, since the means changing the upper limit voltage stops operation, the charging / discharging circuit stops the charging operation of the capacitor and starts the process. discharge operation.

 As a consequence, in accordance with the load driving device of the present invention, even when the overload means temporarily and erroneously assess the occurrence of overload due to the adverse effect of noise, power is not necessarily made non-conductive. Further, since the charging / discharging circuit employed in the driver is used because of its delay, it can be realized without employing a specific delay circuit.

 Then, the means for stopping the attack not only renders the power transistor necessarily non-conductive when the voltage at the terminals of the capacitor is greater than the judgment voltage of the judgment of the attack, but also maintains the circuit charging / discharging the state of charge of the capacitor.

As a consequence, after the power transistor has been made non-conductive because the presence of an overload has been appreciated, the capacitor continues to be charged even after this terminal voltage reaches the judgment voltage of the shutdown. 'attack. Finally, this terminal voltage is maintained as the maximum rechargeable voltage in the charging / discharging circuit.

 Accordingly, according to the present invention, after the power transistor has been made non-conductive, this non-conductive state is maintained by charging the capacitor. Even when the first stopping means of the attack temporarily and erroneously assesses the fact that the voltage present at the terminals of the capacitor becomes lower than the judgment voltage of the stopping of the attack because of the noise after the first stopping means of the attack has rendered the power transistor nonconductive, there is no risk that the non-conductive state of the power transistor is initiated. When the attack stopping means erroneously appreciates the fact that the voltage across the capacitor becomes greater than the judgment voltage of the stopping of the attack because of the noise, the power transistor is rendered temporarily non-conducting in response to this erroneous assessment. In this case, since the capacitor is not charged in such a way that the voltage at its terminals becomes greater than or equal to the judgment voltage of the judgment of the attack, the non-conducting state of the transistor of power is not maintained.

 In addition, as described above, in the present invention, when the non-conductive state of the power transistor is triggered by the charge of the capacitor, and while the voltage across the capacitor becomes lower than the voltage judgment of the attack, this state of engagement is initiated. Therefore, even if the power switch used separately in the current path between the power supply and the load is repeatedly closed, the non-conductive state of the power transistor can be maintained, but this transistor will not be set to the non-conducting state whenever the power switch is closed.

 Applicant has studied the overvoltage protections of the above-mentioned power transistor, in the case where the above-mentioned device for driving a load is applied to a two-terminal type load driving device, where the transistor is used as what is called a switch on the high side.

 More specifically, in the conventional control device employing a semiconductor element, there is known a surge protection circuit such as that described in Japanese Patent Application Laid-open No. 50-36942 in which the element Output semiconductor connected to the power supply can not be destroyed when there is a power surge in the power supply.

Then, the output semiconductor element is forced through by the breaking current of the diode
Zener, and this overvoltage is absorbed in the load of the output semiconductor element.

 However, when such a conventional surge protection circuit is applied to the two-terminal driver (DC power source connection terminal and load connection terminal) to make the driver / non-driver N-channel power transistor used as "high-side switch" in the current path between the DC power source and the load, the power transistor is turned on / off at a preselected frequency during the occurrence of overvoltages and then this power transistor is overheated. In the worst case, there is the risk that the power transistor will be destroyed by heat. We will now explain in more detail this classic problem.

 First, as shown in FIG. 8, for example, the two-terminal type driver for driving the N-channel power transistor 86, connected in series with the current path between the battery 80 and the load 82. together with the main switch 84, includes an auxiliary power source 92. This source 92 provides power to the internal control circuit 88 and the driver 90, etc., by charging the capacitor C10 from the source supply via the diode D10 by the voltage appearing at the two terminals (drain source) of the power transistor 86 when the main switch 84 is in the closed state and the power transistor 86 is in the non-conductive state, and also by the electrical charges stored in this capacitor C10, and the voltage across the terminals of the power transistor 86.

Therefore, when the above-described conventional overvoltage protection circuit is applied to the two-terminal type driver, this circuit can be arranged, as indicated by reference 94 in FIG. 8, with diodes
Zener ZD11 to ZD13 become conductive when the voltage of the power source Vc output from the auxiliary power source 92 reaches a predetermined upper limit. In addition, this overvoltage protection circuit comprises the protective transistor TR15 to obligatorily conduct the drive transistor TR16 provided in the driver circuit 90, which has the effect of making the power transistor 86 conductive, and also resistors R51 and R52 for making the protective transistor TR15 (that is, the power transistor turned on) conductive by the breaking currents flowing through Zener diodes ZD11 to ZD13.

 It will be noted in the drive device of FIG. 8 that the driver circuit 90 for making the power transistor 86 conductive / non-conductive is constructed so as to turn on the power transistor 86 by applying the voltage Vc of the source supplying the resistors R14 and R15 to the gate of the power transistor 86 when the driving transistor TR16 turned on / off in response to the control signal from the control circuit 88 is turned off. In addition, the driving device comprises the overvoltage protection Zener diode ZD14 which limits the voltage of the gate of the power transistor 86 to a value lower than a preselected value when the voltage Vc of the power supply becomes abnormal (overvoltage ).

 In the overvoltage protection circuit 94 having the arrangement explained above, when the main switch 84 is in the closed state, and also when the power transistor 86 is in the non-conductive state, the voltage VB of the battery becomes a surge. If the terminal voltage (supply voltage Vc) of the capacitor C10 of the power supply supplied by the battery 80 exceeds the breaking voltages of the Zener diodes ZD11 to ZD13, the protective transistor TR15 is then made conductive. where it follows that the power transistor 86 is necessarily made conductive.

 When the power transistor 86 is turned on during the occurrence of an overvoltage, the current flows from the battery 80 via the power transistor 86 and the main switch 84 to the load 82, so that terminals of the load 82, that is to say the potential of the mass KG of the driving device, is substantially equal to the voltage obtained by subtracting the voltage in the conducting state of the power transistor 86 from the voltage VB of the battery 80. At this time, the supply voltage Vc in the driver corresponds to the voltage across the capacitor C10 of the auxiliary power supply 92 on the basis of the potential of the mass KG. Then, in this situation, since the potential at the positive polarity of the capacitor C10 of the power supply becomes greater than that of the battery 80, no charging operation of the capacitor C10 from the battery 80 is executed, and the supply voltage Vc is immediately lowered by the discharge operation between the capacitor C10 and the control circuit 88 and the circuit 94 for protection against overvoltages. In response to the lowering of this voltage Vc, the protective transistor TR15 employed in the overvoltage protection circuit 94 is made non-conductive and the power transistor 86 is also turned off. Then, when the power transistor 86 is made non-conductive, since the voltage VB of the battery equal to the overvoltage is again applied to the capacitor C10 of the power supply, the voltage Vc increases. As a result, the overvoltage protection circuit becomes active and the power transistor 86 is rendered non-conductive.

 In other words, as described above, when the conventional overvoltage protection circuit is applied to the two-terminal type driver, this driver acts as an oscillating circuit, and the transistor power is periodically made conductive / non-conductive.

 Then, at this time, the oscillation frequency is determined by the junction capacitances of Zener diodes ZD11 to ZD13 and the capacitance of capacitor C10 of the power supply normally becomes several hundred KHz. For example, when overvoltage occurs for a long time as a result of the decrease in charge, this driver will oscillate repeatedly.

 On the other hand, the operating frequency of the driving circuit 90 shown in FIG. 8 is determined on the basis of the resistances R14, R15 and the capacitance of the gate of the power transistor 86, so that, when the protective transistor TR15 is turned on / off at the above oscillation frequency, the gate voltage becomes stable at an intermediate potential which is defined between the voltage present when the protection transistor TR15 is turned on and the voltage at the moment when this transistor is made non-conductive. As a result, the power transistor 86 can not be driven into a perfect saturation zone where the internal resistance of the transistor becomes minimum, and therefore it will be fixed in a saturation region where it is in the half-state. driver.

 As a result, when the overvoltage is generated for a long time, the loss in the power transistor 86 is increased relative to the load current, and the heat dissipation of the transistor 86 increases. In the worst case, the power transistor 86 will be thermally destroyed.

 The present invention has been realized for the purpose of solving the problems that have just been described and, consequently, its secondary object is to provide an overvoltage protection circuit such that it can prevent the destruction of a transistor. power in the case of driving the power transistor used as a high side switch in the current path between a DC power source and a load by a two-terminal type driver; when overvoltages occur in the DC power source, the power transistor can be turned on securely without operating the driver as an oscillating circuit.

 To achieve the second object just described, an overvoltage protection device according to a second aspect of the invention is characterized in the charge driver circuit described above in that the current path has a connection terminal of a DC power source, and a load connection switch for connecting the load. This overvoltage protection device is employed in a load driving device having an auxiliary power source connected in parallel with the power transistor, for storing electric charges in a supply capacitor by a power supply. voltage that appears across the power transistor, and to produce a supply voltage by the electrical charges stored in the capacitor of the power supply and the voltage present at both terminals of the connection of the DC power source and the connection of the load; and a charging circuit operating by receiving the power supply from the auxiliary power supply, to turn the power transistor on / off in response to an externally applied control signal; which has the effect of attacking the charge. The overvoltage protection device includes: an overvoltage judgment means for judging whether an overvoltage occurs in the DC power source when the voltage applied from the power source auxiliary exceeds a predetermined judgment voltage; a judgment voltage setting means for setting the judgment voltage at a first judgment voltage unless the overvoltage judging means appreciates the occurrence of the overvoltage, and for establishing the a judgment voltage at a second evaluation voltage lower than the first judgment voltage according to a preselected value when the overvoltage judging means evaluates the appearance of the overvoltage, and then until overvoltage appreciation appreciates the appearance of the surge; and a protection means for making the power transistor conductive when the overvoltage judging means appreciates the occurrence of the overvoltage.

 In the overvoltage protection device having the arrangement just described, according to the second aspect of the present invention, the overvoltage judgment means appreciates the fact that an overvoltage occurs in the power supply. direct current when the voltage of the auxiliary power source exceeds a predetermined judgment voltage. When this overvoltage evaluation means appreciates the appearance of the overvoltage, the protection means turns on the power transistor. The appreciating voltage in the overvoltage judging means is set at the first judging voltage by the judgment voltage setting means unless the overvoltage judging means appreciates the appearance of the overvoltage. surge. When the appearance of the overvoltage is appreciated by the overvoltage judging means, the appraisal voltage setting means sets the judgment voltage at a second judgment voltage lower than the first following a preselected voltage. after the overvoltage judgment means does not appreciate the occurrence of the overvoltage.

 In other words, according to the second aspect of the present invention, a so-called "hysteresis characteristic" is provided in the judgment voltage of the overvoltage judgment means. When the overvoltage of the DC power source is appreciated by the overvoltage judgment means, the voltage used to assess the voltage is subsequently changed to the second rating voltage below the normal rating voltage. (first tension of appreciation). As a result, the time set after the power transistor has been turned on and until the voltage of the auxiliary power source is lowered to make the power transistor non-conductive can be extended, so that the transistor The power is turned on / off at high speed when the surge is produced in the DC power source.

 As a consequence, according to the second aspect of the present invention, the power transistor can be safely protected, not suffering from the conventional problem that, when an overvoltage occurs in the DC power source, the transistor Power is made conductive / non-conductive at high speeds and therefore is overheated, as in the conventional charge driving device.

 In addition, the Applicant has considered the thermal destruction of the power transistor in relation to the charge driver circuit described above.

More particularly, in the two-terminal type load driver described above, when the main switch
(Direction indicator switch) is closed, there is virtually no potential difference between the terminals of the power transistor. Accordingly, even in this situation, the auxiliary power source is needed, which employs the capacitor of the power supply used to operate the driver circuit to drive the power transistor.

 However, the voltage of the source of the auxiliary power supply decreases when the voltage produced across the power transistor is lowered in the case where the transistor corresponding to the switching element is made non-conductive, that is to say when the output voltage of the DC power supply is lowered in order to produce the supply voltage using the output voltage of the DC power source. As a result, when the output voltage of the DC power source is lowered, the power transistor is placed in the low voltage drive state. There is no problem if the driving voltage of the power transistor is the voltage in the saturation zone where the internal resistance of the transistor becomes minimum. When the drive voltage is thus lowered, the internal resistance increases, so that the dissipation of heat increases, which results in thermal destruction.

 A third object of the present invention is therefore to provide a load driving device which, when the output voltage is from the DC power source, or from the auxiliary power supply, is lowered. The driving operation of the power transistor is stopped to suppress the heat dissipation of this transistor, and then the thermal destruction of the transistor can be avoided.

 In order to achieve the third object above, a third aspect of the present invention is characterized in that, in the load driving device described above, means for detecting the output voltage for the DC power source or the auxiliary power source is employed, and this output voltage detecting means includes a second means for stopping the attack in order to stop the driving operation of the power transistor when the output voltage is from the DC power source, or from the auxiliary power source drops.

 According to the charge driving device having the third aspect above of the present invention, when the output voltage of the DC power source is lowered, the decrease of this output voltage is detected by the output voltage detection means and then the driving operation of the power transistor is stopped.

 As a consequence, it can be avoided that the power transistor is attacked under a low voltage such that its internal resistance becomes high.

Accordingly, the thermal destruction that is caused by the heat dissipation of the power transistor being driven under a low voltage can be avoided.

The present invention will be better understood from the following description made in conjunction with the accompanying drawings in which
Fig. 1 is a circuit diagram of a load driving device according to a first embodiment of the present invention;
Figure 2 schematically illustrates the circuitry of a load driver, wherein all embodiments of the present invention have been mounted;
Fig. 3 is a timing diagram for indicating the operations of the load driving device shown in Fig. 1;
Fig. 4 is a circuit diagram of a load driver according to a second embodiment of the present invention, corresponding to an exemplary variant of the first embodiment;
Fig. 5 is a diagram of an overvoltage protection circuit employed in the load driver of the present invention;
Fig. 6 is a timing diagram for showing the operations of the overload protection circuit of Fig. 5;
Fig. 7 is a diagram of an exemplary variant of the overload protection circuit employed in the load driver of Fig. 5;
Fig. 8 is a diagram of the conventional overload protection circuit for the load driver shown in Fig. 5 and Fig. 7;
Fig. 9 is a circuit diagram of another load driving device according to a fifth embodiment and a sixth embodiment of the present invention;
Fig. 10 is a timing diagram for indicating the operations of the load driving device shown in Fig. 9; and
Fig. 11 is a circuit diagram for an alternative embodiment of the load driving device shown in Fig. 9.

 In connection with the drawings, there will now be described a charge driving device according to an embodiment of the present invention.

 Figure 2 schematically illustrates an overall circuit arrangement of a vehicle direction indicator, to which the present invention is applicable. Figure 1 is a circuit diagram of a power MOSFET overload protection device employed in this direction indicator, wherein the power MOSFET corresponds to one of the power transistors.

 As shown in Fig. 2, the direction indicator of a vehicle according to a first embodiment of the present invention consists of direction indicators lOL, lOR which are mounted on the left and right sides of the vehicle, respectively; a direction indication switch 12 actuated by the driver of the vehicle so as to turn on one of the lamps lOL, lOR; and a N-channel power MOSFET 16. The MOSFET 16 is mounted in a current path consisting of the switch 12, a fuse 15 and a battery 14, whereby the energy of the battery 14 is supplied to the lamps. lOL and lOR. One of the lamps lOL and lOR, which is connected by the switch 12, flashes by periodically connecting and disconnecting this current path when the switch 12 is closed. To connect / disconnect this path from the current, the drain of the power MOSFET 16 is connected via a terminal TB to the battery 14, and its source via a terminal TL to the switch 12.

 An auxiliary power source consisting of a resistor R16, a diode D1, and a supply capacitor C1 is connected between the terminals TB and TL, to which the MOSFET 16 is connected, and this source is mounted in parallel with the MOSFET 16.

 This auxiliary power source charges the capacitor C1 with a voltage substantially equal to the output voltage of the battery 14 produced between the terminals TB and TL when the switch 12 is closed to connect the terminal TL to one of the indicator lamps lOL and lOR, and further, the MOSFET 16 is made non-conductive. This voltage generated between the terminals is applied to the driver of a load as a supply voltage. When the MOSFET 16 is brought to the conductive state, hence the presence of the voltage between the terminals TB and TL, the voltage across the capacitor C1 is applied to the device for driving a load as a supply voltage. .

 At capacitor C1 is connected a constant voltage circuit consisting of a constant current integrated circuit IC, a Zener diode D2 and a transistor TR1. Based on the constant voltage output from this circuit, various reference voltages can be produced which are used to control the power MOSFET 16 in the conductive / non-conductive state.

 On the other hand, there is provided in this vehicle direction indicator device, according to the present embodiment, a driver circuit 31 for driving the MOSFET 16; an integration circuit consisting of a charge / discharge capacitor CO and a resistor RO, and a flash circuit 22 to produce a flash signal S1 as a control signal as described above. The indicator lamps lOL and lOR are flashed by charging / discharging the capacitor CO by applying the charge / discharge control voltage via an inverter IN1 to the integration circuit when the switch 12 is closed. The direction indicator device further comprises a fast discharge circuit 24 for rapidly discharging the charge / discharge capacitor CO by detecting that the switch 12 is open and that the transistor TR2 is turned on; over-current appreciating means as defined in the first aspect of the present invention, corresponding to a circuit for preventing an overcurrent from flowing in the MOSFET 16 by detecting this flow, whereby the blinking of the lamps LOL and 1OR under the effect of the circuit 22 is prevented. The direction indicator device further comprises a driver stop circuit 26, having a higher voltage change means, a drive stop means; a circuit 28 producing a rating voltage to produce the rating voltage of the means appreciating the overcurrent in the circuit 26; an overvoltage appreciating means as defined in the second aspect of the present invention, which detects an extraordinary situation (overvoltage) of the battery voltage VB from the supply voltage Vc which is applied from the source auxiliary power supply, which has the effect of making the power MOSFET 16 necessarily conductive; and an overvoltage protection circuit 30 having a rating determining means and a protection means.

 It will be noted that the flashing circuit 22 is known in the art; it repeatedly charges / discharges the charge / discharge capacitor CO in accordance with the sequence mentioned below, and produces a flash signal S having a low level to turn on the power MOSFET 16 during the charging operation. Specifically, the constant voltage VT obtained from the constant voltage circuit is subdivided to produce an upper limit voltage and a lower limit voltage of the capacitor CO, and the output of the inverter IN1 is set high up to the voltage at the terminals of the charging / charging capacitor CO reaches the upper limit voltage, so that the capacitor CO is charged at a given time constant. When the voltage across the capacitor CO reaches the upper limit voltage, the output of the inverter IN1 is set low, whereas this terminal voltage then reaches the lower limit voltage, so that the capacitor CO is discharged at a lower voltage. predetermined time constant.

 The constituents of the driver stop circuit 26, the judgment voltage production circuit 28, the overvoltage protection circuit 30, the driver circuit 31, and the driver circuit 24 will be described in greater detail. fast discharge.

[FIRST EMBODIMENT]
First of all, FIG. 1 shows the present invention defined by its first aspect, namely a device for driving a load equipped with an overcurrent protection circuit according to a first embodiment of the present invention. invention.

 FIG. 1 shows in more detail the arrangement of the internal circuits of the blinking circuit 22, the driver stop circuit 26 and the circuit 28 for producing the evaluation voltage among the circuits represented in FIG. that the same references as in Figure 2 will be used to designate the same circuit elements in Figure 2, and also in the following drawings.

 As a flashing circuit 22 shown in Figure 2, the vehicle direction indicator device is equipped according to this embodiment of a charging / discharging circuit. This charging / discharging circuit consists of an integration circuit comprising the charge / discharge capacitor CO and the resistor RO; dividing resistors R1, R2, R3 and R4 for subdividing the output voltage VT obtained at the constant voltage circuit to produce the various reference voltages; and a comparator CO1 for comparing the voltage VOS across the capacitor CO to the reference voltages produced from these resistors R1 to R4 to produce a high signal when the voltage VOS becomes greater than the reference voltage. This charging / discharging circuit further comprises an inverter IN1 for inverting the output signal obtained to the comparator CO1 to supply the inverted signal to the integration circuit; and a transistor TR2 for receiving the output of the comparator CO1 via a resistor R5 at its base, and for grounding the junction point between the resistors R2 and R3 when the output of the comparator CO1 goes high.

 Normally, the junction point between the resistors R3 and R4 is grounded via a transistor TR3 operating as a means of changing the reference voltage. In this charging / discharging circuit, the upper limit voltage VTA and the lower limit voltage VTC are generally produced by the resistors R1 to R3 when the capacitor is charged and discharged, so that the capacitor CO is charged and discharged.

 In other words, the output of the comparator CO1 goes low until the voltage VOS across the charge / discharge capacitor CO reaches the upper limit voltage VTA = VT (R2 + R3) / (R1 + R2 + R3) obtained by subdividing the output voltage VT present at the constant voltage circuit by the resistor R1, and the resistors R2 and R3, during which time the capacitor CO is charged via the resistor RO by the signal at the high level coming from the inverter IN1.

 Then, the voltage VOS across the capacitor CO reaches the upper limit voltage VTA, the level of the output of the comparator CO1 is inverted to go high, and thus the electrical charges stored in the capacitor CO are discharged via the resistor RO into the inverter IN1.

 At this time, since the output of the comparator CO1 is at the high level, the transistor TR2 is turned on, and the reference voltages applied to the comparator CO1 go to the lower limit voltage VTC = VTxR2 / (R1 + R2) obtained by subdividing the output voltage VT from the constant voltage circuit by the resistors R1 and R2.

 As a result, during the period when, after the voltage VOS across the capacitor CO has reached the upper limit voltage VTA, and this voltage VOS becomes the lower limit voltage VTC, the signal level of the output of the comparator CO1 becomes a high level and then the discharge operation of the capacitor CO continues.

 Then, when the voltage VOS across the charging / discharging capacitor CO reaches the lower limit voltage VTC, the signal level of the comparator output CO1 goes low, and the reference voltage applied to the comparator CO1 switches to voltage. upper limit VTA.

 As a result, when the voltage VOS across the capacitor CO reaches the lower limit voltage VTC as a result of the discharge operation, the capacitor CO is recharged again until the voltage VOS reaches the upper limit voltage VTA. .

 Then, the output of the comparator CO1 is applied as a driving signal making the power MOSFET 16 conductive / non-conductive to the transistor TR6 used to constitute the driving circuit of the MOSFET 16. In other words, this signal entered into the Transistor TR6 corresponds to the flashing signal S1 shown in Figures 2, 5 or 7. This signal is applied in the same manner in the other embodiments. The transistor TR6 receives at its base the output signal from the comparator CO1 via a resistor R13, and is brought to the conductive state when this output signal is at the high level, whereby the voltage coming from the source Auxiliary power supply is applied to the gate of the MOSFET 16 from the point of junction between a resistor R14 and a resistor R15, which is grounded.

 That is, transistor TR6 causes MOSFET 16 to become conductive during the charging operation of capacitor CO in which the output level of comparator CO1 goes low, resulting in that the transistor TR6 causes the transition to the non-conductive state of the MOSFET 16 during the discharge operation of the capacitor CO, in which the output level of the comparator CO1 goes high, so that the MOSFET 16 is periodically made conductive / non-conductive in response to the outputs of the comparator CO1.

 In addition, there is provided in this direction indicating device according to the present embodiment another C02 comparator operating as a means of evaluating overcurrents. This comparator C02 looks for a voltage (hereinafter referred to as conductive voltage ")" VD "corresponding to the voltage between source and drain VDS of the MOSFET 16, and makes a decision as to whether or not overcurrent crosses the MOSFET 16 verifying whether or not this voltage VD exceeds a voltage Vref appreciation of overcurrent.

 It will be noted that the voltage Vref can be adjusted to the appropriate value determined by a circuit for producing an overcurrent evaluation voltage constituted by a circuit 20 for detecting the current of the load, a resistor R7 and a resistance R8 based on the conditions of use of the MOSFET 16, that is to say the supply voltage and the temperature.

 In other words, the circuit 20 synthesizes the output voltage VT obtained at the constant voltage circuit, the supply voltage being proportional to the voltage of the battery from the auxiliary power source, hence the production of a voltage corresponding to the current in the load passing through the direction indicators lOL or lOR when the power MOSFET 16 is turned on. In addition, this charge current detection circuit 20 produces the overcurrent evaluation voltage Vref which corresponds to the overcurrent flowing under the respective supply voltages and the respective temperatures when the MOSFET 16 is turned on, by subdividing this voltage produced by the resistor R7 and the resistor R8, because this voltage has been corrected to be adapted to a voltage which corresponds to the temperature characteristic of the MOSFET 16 by means of a diode.

 Then, the overcurrent judgment comparator C02 will output a VC judgment signal at the high level so as to turn on the transistor TR3 when the power MOSFET 16 is operating normally and the voltage in the conductive state VD is lower than the voltage Vref. The comparator C02 will output another evaluation signal VC having a low level so as to render the transistor TR3 non-conductive when an assessment is made that the voltage VD in the conductive state exceeds the voltage Vref, and the overcurrent crosses the Power MOSFET 16.

 As a result, as previously indicated, the upper limit voltage VTA is applied normally to the comparator CO1 as the reference voltage for the charging operation of the capacitor CO. When the comparator C02 appreciates that the overcurrent passes through the MOSFET 16, a second upper limit voltage VTB = VT (R2 + R3 + R4) / (R1 + R2 + R3 + R4) which is greater than the upper limit voltage VTA has entered the CO1 comparator. As a consequence, when the overcurrent passes through the MOSFET 16, the charge / discharge capacitor CO is charged to the second upper limit voltage VTB which exceeds the normal upper limit voltage VTA by means of the comparison operation performed by the comparator CO1. .

 Then, the direction indicating device further comprises according to this embodiment a comparator C03 operating as a drive stop means, for comparing the voltage VOS present across the capacitor CO which is loaded / discharged in the manner described. above with an attack judgment VTS voltage which is produced by the resistors R9 to R10 used to subdivide the output voltage VT obtained to the circuit with constant voltage, and which is more or less less than the second upper limit voltage VTB. The comparator C03 outputs a high level signal when the voltage VOS across the capacitor CO is greater than this voltage VTS. The direction indicating device further comprises a resistor R11 and a transistor TR7 to make the MOSFET 16 non-conductive by grounding the gate of this MOSFET via the resistor R15 when the output signal from the comparator C03 is at high level; and a resistor R12 and a transistor TR5 to regulate the voltage of the comparator CO1 at the low level by grounding the path of the driving signal of the MOSFET 16 which is defined between the comparator CO1 via the resistor R13 and the base of the transistor TR6 when the comparator output signal C03 is similarly high.

 As a consequence, when it is appreciated that the overcurrent passes through the MOSFET 16 from the comparator C02, the capacitor CO is charged to a voltage exceeding the normal upper limit voltage VTA, and then the voltage across this capacitor CO reaches the voltage VTS judgment of the cessation of the attack, the MOSFET 16 is made non-conductive and the output of the comparator COl is kept low. After the MOSFET 16 has been made nonconductive, the charging / discharging CO capacitor is continuously charged.

 The direction indicating device according to this embodiment further comprises a fast discharge transistor TR4 which is connected in parallel with the capacitor CO; and a fast discharge circuit 24 for rapidly discharging the capacitor CO by rendering the transistor TR4 nonconductive when the voltage VT leaving the constant voltage circuit is lowered to a predetermined level in connection with the reduction of the supply voltage.

 This fast discharge circuit 24 has such a function. Specifically, when the switch 12 is open, the power supply voltage supplied from the auxiliary power supply is lowered, so that this device can not operate under normal conditions, the charge / discharge capacitor CO is quickly discharged. As a result, the electrical charges stored in this capacitor CO are discharged to zero, whereby the charge operation of the capacitor CO starts from the initial condition when the switch 12 is closed, and therefore the Indicator lamps lOL or lOR flash at a constant period immediately after the closing of the switch 12. It will be appreciated that the operation of the fast discharge circuit 24 will be described in more detail in a fifth embodiment and in the subsequent embodiments.

 It will also be noted that the circuit portions surrounded by a dashed line in the circuit arrangement to constitute the direction indicating device of the present embodiment, with the exception of the power MOSFET 16, the R16 resistor. , of the integration circuit comprising the resistor RO and the charge / discharge capacitor CO, and the capacitor C1 of the power supply, are stored in a single integrated circuit assembly. In Fig. 1, the symbol "VR" indicates a voltage across the lamp lOR, and the symbol "VL" indicates a voltage across the lamp lOL, respectively.

 The operation of the vehicle direction indicating device having the circuit arrangement described above, in accordance with the first embodiment of the invention, will now be described in conjunction with the timing diagram shown in FIG. 3, the references VOS, VD, VC, VR and VL represent the voltages appearing at the various parts of the circuits of FIG.

 As shown in FIG. 3, when the switch 12 is closed by the driver of the vehicle to turn on the indicator lamp 1OR, and that this therefore turned on (Closed), the charging / discharging operations capacitor CO are initiated by the charging / discharging circuit consisting of the comparator CO1, the transistor TR2, the inverter IN1, etc. The power MOSFET 16 is turned on / off via the transistor TR6 in response to the output of the comparator CO1 to flash the lamp 1OR.

 Here, although the conductive voltage of the MOSFET 16 exceeds the overcurrent evaluation voltage Vref due to the current draw, the evaluation signal VC coming out of the comparator C02 goes low, indicating that the overcurrent passes through the MOSFET 16 when the indicator lamp lOR begins to be energized, since the voltage VOS present across the capacitor CO does not reach the voltage VTS judgment of the stop of the attack, the MOSFET 16 is not rendered non-conductive in response to the output from comparator C03. In addition, since the current draw is attenuated after a few tens of milliseconds, the output of the comparator C02 immediately returns to the high level, indicating that the MOSFET 16 is operating normally, so that no detrimental effect is caused by the flashing operation of the lOR indicator lamp.

 Furthermore, when the MOSFET 16 is turned off during the flashing of the indicator lamp lOR, since the voltage between the source and the drain VDS of the MOSFET 16 becomes equal to the battery voltage, the judgment signal VC coming out of the comparator C02 has a low level which indicates that the overcurrent passes through the MOSFET 16. At this time, since the capacitor CO is in discharge, the voltage VOS at its terminals does not pass to a value higher than the voltage VTS of appreciation of stopping the attack, but also the MOSFET is not made non-conductive by the comparator C03.

 As indicated by the arrow "a" in FIG. 3, even when the voltage VD exceeds the overcurrent evaluation voltage Vref due to the noise while the indicator lamp lOR is energized, since the voltage VOS at the terminals of the capacitor CO does not become greater than the voltage VTS, the MOSFET 16 is not made non-conductive by the comparator C03.

 As represented by the arrow "b" in FIG. 3, there are cases where the output of the comparator C03 goes high under the effect of the noise, and then the MOSFET 16 is made non-conductive. In this case, as the noise disappears and the output of the comparator C03 is brought down, the MOSFET 16 is in the on state and is not engaged.

 On the other hand, as represented by the arrow "c" in FIG. 3, when the overcurrent flows continuously during operation in the conducting state of the MOSFET 16, the capacitor CO is charged to a voltage exceeding the upper limit voltage VTA by virtue of operation of the comparator C02, the transistor TR3, the comparator CO1 and the inverter IN1. At the moment when the voltage at the terminals of the charging / discharging capacitor CO becomes greater than the voltage VTS of judgment of the stopping of the attack as a result of this charging operation, the comparator C03 switches the MOSFET 16 to the non-conductive state. In other words, according to this embodiment, when the overcurrent flows during the conducting state of the MOSFET 16, this conducting state is maintained during a charging period as a delay time such that the voltage VOS across the capacitor CO reaches the voltage VTS from the upper limit voltage VTA. After this time, the MOSFET 16 goes from the conductive state to the non-conductive state.

 At this time, since the comparator C03 keeps the output of the comparator CO1 low, the capacitor CO is discharged until the voltage VOS at its terminals reaches the level of the output of the inverter IN1. As a consequence, when the overcurrent passes through the MOSFET 16, after the comparator C03 has made the MOSFET 16 non-conductive, this state is maintained.

 As described above in this embodiment of the invention, in the case where the overcurrent passes through the MOSFET 16 while it is turned on, the charge / discharge capacitor CO is charged up to the voltage greater than the VTS tension of appreciation of the stop of the attack.

As such a delay time defined after the evaluation of the overcurrent until the transition to the non-conductive state of the MOSFET 16 is set, even when the MOSFET 16 is continuously made conductive for the time defined by the addition of the time of when the lOL and lOR lamps are switched on at this delay time, the MOSFET 16 can sufficiently withstand the thermal conditions. In the event that such an overcurrent occurs, and the MOSFET 16 can be instantly destroyed, since the fuse 15 is melting, the MOSFET 16 is not destroyed.

 Then, as represented by the arrow "d" in FIG. 3, if the judgment voltage VTS of the judgment of the attack which has entered the comparator C03 becomes greater than the voltage VOS at the terminals of the capacitor CO as a result of the noise when the comparator C03 causes the transition to the non-conductive state of the MOSFET 16, the output of the comparator C03 then temporarily becomes the low level, and thus the transistors TR5 and TR7 go to the non-conductive state.

However, since the voltage VOS at the terminals of the capacitor CO is greater than the second upper limit voltage VTB at this time, the comparator CO1 outputs a signal at the high level through which the capacitor CO is discharged to the lower limit voltage VTC.

 As a result, even if the output of the comparator C03 temporarily goes low due to noise while the comparator C03 renders the MOSFET 16 non-conductive, this MOSFET is maintained in the non-conductive state by the output of the comparator. CO1. At this time, the capacitor CO is discharged by the output of the comparator CO1. However, the noise application time is very short, and when the noise is eliminated, the output of the comparator C03 again becomes the high level, so that the voltage VOS of the capacitor CO is not lower than the voltage VTS , and the non-conductive state of the MOSFET 16 is securely maintained.

 In addition, the overcurrent flows through the MOSFET 16, and the voltage VOS of the capacitor CO becomes lower than the voltage VTS after the comparator C03 has rendered the MOSFET 16 non-conductive.

Such a situation occurs when the driver of the vehicle cuts switch 12 and the power supply voltage is lowered, and then the charge / discharge capacitor CO is quickly discharged by the operations of fast discharge circuit 24. As a result, even when the conductor closes the switch 12 after the comparator C3 has made the MOSFET 16 non-conductive and before the fast discharge circuit 24 operates, the MOSFET is not turned on. Then, there is no risk that the switch 12 is repeatedly closed, so that the MOSFET 16 is overheated and is electronically destroyed.

 As previously explained, in the direction indicating device according to the present embodiment, when it is appreciated that the overcurrent passes through the power MOSFET 16, the delay time is established using the load circuit. / discharge to determine the flashing period of the indicator lamp. When it is appreciated that overcurrent flows through MOSFET 16 during this delay time, the MOSFET is rendered non-conductive. Accordingly, when the occurrence of the overcurrent is mistakenly evaluated for noise, there is no risk that the MOSFET 16 will become non-conductive. In addition, since no delay specific circuit needs to be employed, the overall arrangement of the circuit can be simple.

 After the power MOSFET 16 has been made non-conductive, the charging / discharging circuit is employed. The non-conductive state of the MOSFET is maintained while the voltage VOS across the capacitor CO becomes lower than the judgment voltage VTS judgment of the attack, so that the MOSFET 16 can be made nonconductive with safety without receiving any influence from the noise after MOSFET 16 was made non-conductive.

 Since the non-conductive state of the MOSFET is initiated when the fast discharge circuit 24 is actuated in connection with the lowering of the power supply voltage, even when the switch 12 is repeatedly closed, which constitutes the power switch. power supply of the present device, after the MOSFET 16 has been made non-conducting, it is not made non-conducting every time this switching operation is performed. Then, it is possible to prevent the MOSFET 16 from being destroyed by repeating the operation of closing switch 12. This point will be described in more detail after the fifth embodiment.

[SECOND EMBODIMENT]
In the embodiment just described, it has been explained that the MOSFET 16 is made conductive / non-conductive in response to the output of the comparator CO1 to charge / discharge the capacitor CO, which has the effect of causing the indicator lamps. The present invention can be applied to another driving device such as the MOSFET 16 is turned on / off in response to a MILV signal having a predetermined duty cycle, thereby controlling the charging current of an engine and of a lamp. As to this arrangement of the circuit, its description will be made in conjunction with FIG. 4.

 As shown in FIG. 4, in the case where a load 40 is controlled by MILV using the power MOSFET 16, in addition to the circuit arrangement of the first embodiment described above, a C04 comparator. This comparator C04 compares the voltage across the capacitor CO to the reference voltage obtained by dividing the constant voltage from the constant voltage circuit by a variable resistor VR1 and a resistor R20, and produces a signal having a low level when the voltage capacitor terminals is greater than or equal to this reference voltage. The output of the comparator C04 is input via a resistor R13 into a transistor TR6 to drive the power MOSFET. The other arrangements of the circuits are similar to those of the first embodiment.

 In a similar manner to the embodiment explained above, in the MILV driver device having the circuit arrangement described above, the power MOSFET 16 may preferably be protected without adverse effect. Due to the noise, and moreover, when, for example, a motor is used to constitute the load 40, the advantages mentioned below can be obtained.

It is assumed that when the engine is started in response to the drive signal having a low duty cycle with a short conduction time, the motor is mechanically locked due to problems in the mechanical system. At this time, as the VDS voltage between the source and the drain of the MOSFET 16 increases, the output of the overcurrent evaluation comparator C02 goes low, so that the charge / discharge capacitor CO continues to be charged, the ratio cyclic drive signal for the
MOSFET 16 becomes 100%, and thus the drive torque of the motor becomes maximum. As a result, in the case of such an engine blocking phenomenon, the problem raised by the mechanical system can be automatically solved by the maximum driving torque, and therefore this blocking situation can be released.

 Note that when either the motor blocking phenomenon can be released by the drive signal having the duty cycle of 100%, or the overcurrent is caused as a result of a malfunction of the electrical system, the power MOSFET 16 is made conductive at the instant when the voltage VOS across the charging / discharging capacitor CO becomes greater than the voltage VTS appreciation of stopping the attack. Therefore, the MOSFET 16 can not be electrically destroyed, and the wiring patterns can not be damaged.

[THIRD EMBODIMENT]
In connection now with Figure 5, will be described in more detail the arrangement of the driver circuit 31 and the circuit 30 protection against overvoltages.

As shown in FIG. 5, the drive circuit 31 is arranged according to this embodiment so as to use the driver transistor TR6 which is turned on / off in response to the flash signal S1 coming out of the similar flashing circuit 22. to the conventional device shown in Figure 8; when the transistor TR6 is brought to the non-conductive state (that is to say, when the signal S1 is at the low level), the MOSFET 16 is turned on by applying the voltage Vc of the supply via the resistors R14 and R15 at the gate of the MOSFET 16. In addition, this driver 31 is equipped with diodes
Zener ZD4 and ZD5 overvoltage protection that can limit the voltage of the gate of the MOSFET 16 to a value less than a predetermined voltage when the voltage Vc of the power supply becomes abnormal (occurrence of an overvoltage).

Then the overvoltage protection circuit 30 of this embodiment will appreciate the abnormal (overvoltage) condition of the battery voltage VB on the basis of the voltage Vc of the power supply that is applied from the power source. auxiliary, and therefore necessarily makes the power MOSFET 16. conductive. In a manner similar to the conventional device shown in FIG. 8, this circuit 30 for protecting against overvoltages consists of Zener diodes ZD1 to ZD3 which are conductive when the voltage Vc of the supply reaches a predetermined upper limit value VHH; a protective transistor TR12 to necessarily make non-conducting the drive transistor TR6 used in the driver circuits 31 so as to make the MOSFET 16 conductive; and resistors R31 and R32 for turning on the protective transistor TR12 (i.e., the MOSFET is turned on) by applying a bias voltage to the protective transistor TR12 due to the breaking currents flowing through them. diodes
Zener ZD1 and ZD3 while these diodes are conductive.

In addition, the surge protection circuit 30 of the present embodiment comprises an NPN type transistor TR11 and a PNP type transistor TR13. The base of this transistor TR11 is connected via a resistor R35 to the junction point between a resistor R31 and a resistor R32, with its emitter connected to the earth line (potential KG of the mass) of this device, and its collector connected via a resistor R33 and a resistor R34 to the supply line (voltage VC of the power supply) of this device. The base of the transistor TR13 is connected to the junction point between the resistor R33 and the resistor R34, and the emitter and the collector of this transistor are connected to the two terminals of the diode
Zener ZD1.

 Therefore, in the surge protection circuit 30 of this embodiment, when the voltage Vc of the power supply reaches the upper limit VHH, the Zener diodes ZD1 to ZD3 are conductive and the protective transistor TR12 is turned on. At the same time, the transistor TR13 is turned on, and the two ends of the Zener diode ZD1 are short-circuited by the transistor TR13. Then, while the voltage Vc of the power supply goes below the lower voltage VHL which is lower than the upper limit voltage VHH determined by the breaking voltages of the Zener diodes ZD2 and ZD3 according to the breaking voltage of the Zener diode ZD1, the conductive state of the protective transistor TR12, namely the conducting state of the power MOSFET, is maintained.

 Now in conjunction with the timing diagram of FIG. 6, the operations and effects of the vehicle direction indicating device employing the arrangement of the circuits just described will be described.

 Since the arrangement of the flashing circuit 22 is similar to that of the first embodiment, the explanation will be omitted.

 First, an area "A" of Figure 6 represents the normal operation of the flashing circuit.

 Then, as shown in area "B" of Fig. 6, when an overvoltage occurs in a situation such that the direction indicator switch 12 is closed, since the device is open for battery 14, the MOSFET 16 and the internal circuits of this device are not destroyed electrically by this overvoltage.

 On the other hand, as shown in the "C" area of FIG. 6, when the switch 12 is in the closed state and the flashing signal S1 is high, ie the MOSFET 16 is in the non-conductive state, an overvoltage is produced. When the supply voltage Vc reaches the upper limit voltage VHH, there is a break in the Zener diodes ZD1 to ZD3 employed in the overvoltage protection circuit 30, and the protective transistor TR12 is turned on, so that the Power MOSFET 16 is forced into the conductive state. At the moment the overvoltage occurs, it is applied between the terminals TB and TL (that is, between the source and the drain of the MOSFET 16). However, after increasing this voltage to substantially the upper limit voltage VHH, the MOSFET 16 is made conductive, so that this overvoltage is immediately lowered.

 Then, as described above, when the MOSFET 16 is turned on, the voltage Vc of the power supply is progressively decreased from the upper limit voltage VHH according to the capacitance of the capacitor C1 of the power supply and also to the consumption currents of the internal circuit.

 On the other hand, when the MOSFET 16 is turned on, and at the same time, the two transistors TR11 and TR13 employed in the surge protection circuit 30 are made conductive, both ends of the Zener diode ZD1 are short. -circuitées. As a result, the conducting state of the MOSFET 16 is maintained until the voltage Vc of the power supply becomes the lower limit voltage VHL, and the normal conductive / non-conductive control of the MOSFET 16 is resumed. at the instant when the voltage Vc of the power supply is reduced to the lower limit voltage VHL.

 During this period, since no such overvoltage, exceeding the upper limit voltage VHH, is applied to the power MOSFET 16 and the internal circuit, there is no risk that the MOSFET 16 and the semi-conductors conductors used in the circuit are destroyed electrically because of this surge.

 Since the gate of the MOSFET 16 is biased by a voltage defined on the basis of the Zener diode breakdown voltages ZD4 and ZD5, the MOSFET 16 is brought to a very good conductive state. In addition, since the MOSFET 16 is not made conductive / non-conductive by the high frequency signal, there is no risk that the MOSFET 16 will be thermally destroyed, which is completely different from the prior art device.

 On the other hand, in the case where an overvoltage occurs, as shown in zone "D" of FIG. 6, at the moment when the MOSFET 16 is brought to the conductive state in response to the blinking signal S1 coming out of the circuit of FIG. blinking 22, this overvoltage is discharged from the MOSFET 16 either in the lamp lOL or in the lamp lOR, which constitute a load. Therefore, no overvoltage is applied to the MOSFET 16, as well as the internal circuit, and therefore, no problem is raised.

 As previously described in detail, in the direction indicating device according to this embodiment, Zener diodes ZD1 to ZD3 are employed as a surge arrester to appreciate the occurrence of such overvoltages. surge.

When the voltage Vc of the power supply reaches the upper limit voltage VHH determined on the basis of the breaking voltages of Zener diodes ZD1 to ZD3, a decision is made that an overvoltage occurs, and therefore the MOSFET 16 is driven in the conductive state. Furthermore, after the MOSFET 16 comes into the conductive state, the two terminals of the diode
Zener ZD1 are short-circuited. The overvoltage evaluation voltage is changed to the lower limit voltage VHL, which is determined by the breaking voltages of Zener diodes ZD2 and ZD3. There is an extension of the period during which, after the MOSFET 16 is brought into conductive state, the voltage Vc of the power supply is lowered, and then the MOSFET 16 is made non-conductive. Consequently, according to this embodiment, it is possible to solve the conventional problem in which the MOSFET 16 is made conductive / non-conductive at high speeds during the occurrence of the overvoltage, which results in the overheating of the MOSFET 16.

Thus, the power MOSFET can be safely protected.

[FOURTH EMBODIMENT]
It should be noted that in the embodiment described above, the overvoltage protection circuit 30 is arranged by using Zener diodes ZD1 to ZD3 to assess the presence of the overvoltage. Alternatively, this over-voltage protection circuit 30 may be constructed as shown in FIG. 7. Specifically, the voltage Vc of the power supply is subdivided by employing a dividing resistor R36 and a dividing resistor R37, and the constant voltage VT from the constant voltage circuit is also subdivided using three resistors R38 to R40 to produce a reference voltage Vref (= VHH) to assess the presence of an overvoltage. A comparator C07 is used to compare the reference voltage Vref with the subdivided voltage VCC of the voltage Vc of the supply obtained by the resistors R36 and R37.

When the subdivided voltage VCC becomes greater than or equal to the reference voltage Vref, an overvoltage judgment signal (high level) is produced in this comparator C07 so as to turn on the protective transistor TR12 via the resistor R31. In addition, during the overvoltage evaluation operation, the transistor TR13 is made conductive via a resistor R41 by receiving the judgment signal from the comparator C07, so that the dividing resistor R40 makes it possible to produce the reference voltage Vref is short-circuited, and therefore, the reference voltage Vref is replaced by another reference voltage Vref (= VHL) lower than the normal value. Consequently, in this variant embodiment, an effect similar to that of the embodiment described above can be obtained.

 Since the overvoltage protection circuit 30 is so arranged, the overvoltage detection voltage (VHH), the overvoltage-releasing voltage (VHL), and the overvoltage evaluation time can be established freely with respect to the circuit. overvoltage protection of the embodiment explained above. In addition, since this over-voltage protection circuit does not have any temperature characteristics such as a Zener diode, overvoltage protection can be exerted with high accuracy.

 Although the overvoltage protection device of the present invention has been applied to the direction indication device of a vehicle in the embodiment described above, the present invention is applicable to a device in which a N-channel power MOSFET used as a high-side switch is driven by a two-terminal type driver, whereby a similar advantage to that of the embodiment described above can be obtained.

[FIFTH EMBODIMENT]
A description will now be made of an embodiment of the third aspect of the present invention in connection with FIG. 9.

 It will be noted in the drawing that, since the basic operation of making the power MOSFET 16 conductive / non-conductive by the use of the charge / discharge capacitor CO has been explained in connection with the first embodiment, omit the description. In FIG. 9, the resistor R4 is suppressed.

 A characteristic part of the circuit of FIG. 9 is a circuit preventing a low voltage attack which has the following construction the circuit comprises a resistor R21 and a resistor R22 which subdivide the potential at the TR9 junction polarized by the voltage subdivided by the resistors R21 and R22, operating as an output voltage detecting means defined in the present invention. This circuit preventing low-voltage driving further comprises a transistor TR8 operating as a second circuit for stopping the attack in order to force the value of the output of the comparator CO1 to a low level corresponding to the potential of the mass KG in this device, wherein, when the voltage divided by the resistors R21 and R22 is lowered to make the transistor TR9 non-conductive, a bias voltage is applied thereto via the resistor R20, and then the output of the comparator COl is applied to the terminal TL, and a circuit preventing the application of a voltage <ie, the circuit ND1 to door NAND and in this embodiment) in order to prevent the application of this bias voltage to MOSFET 16 in response to the fact that the output value of the comparator CO1 is forcibly brought to the low level.

 By employing the low voltage driver circuit explained above, it is possible to prevent the MOSFET 16 from being driven at low voltage because the voltage across capacitor C1 of the power supply is lowered, and then the MOSFET 16 is thermally destroyed.

 It will be noted that the driving circuit of the MOSFET 16 consists of this NAND gate circuit ND1, a resistor R13, one end of which is connected to the positive terminal of the capacitor C1 of the power supply and whose other end is connected to the gate of the MOSFET 16, and a transistor TR7. This transistor TR7 is brought to the conductive state when the output of the circuit ND1 is high, and then the MOSFET 16 is made non-conductive by connecting the gate of the MOSFET to the terminal TL corresponding to the ground terminal of this device . The transistor TR6 is set to the non-conductive state when the output of the circuit ND1 is at the low level, and then the MOSFET 16 is made conductive by the application of the terminal voltage of the capacitor C1 between the gate and the source of the MOSFET 16.

 In fact, in the circuit shown in FIG. 9, another transistor TR10 is provided outside the transistor TR9, whereby the transistor TR8 corresponding to a second stop circuit of the drive is controlled. In addition, a fast discharge transistor TR4 is used which is connected in parallel with the transistor TR8, which operates as a switch for discharging the electric charges stored in the charge / discharge capacitor CO.

 This fast discharge transistor TR4 corresponds to a circuit such that, when the direction indication switch 12 is closed, the charge / discharge capacitor CO is discharged, whereby the circuit can be attacked from the state initialized when the switch 12 is subsequently closed. In other words, the fast discharge circuit 24 detects from the voltage VB of the battery which is applied to this device that the switch 12 is open when the power MOSFET 16 is made non-conductive, from which it results that the transistor TR4 is turned on to quickly discharge the capacitor CO. As a consequence, when the switch 12 is then closed, the charge operation of the capacitor CO starts from an initial condition such that there is no electric charge, so that either the lamp lOL or the lamp lOR flashes for a constant time just after switch 12 is closed.

 Note that this fast discharge circuit 24 can be used to quickly discharge the charge / discharge capacitor CO when the constant voltage VT leaving the constant voltage circuit becomes less than a preselected voltage, whereby this device can not operate under normal conditions.

 A description will now be given of how to detect with certainty that the switch 12 is open in such a "two-terminal" type of vehicle direction indicating device, which is represented in FIG. 9.

 The voltage that can drive the transistor is produced between the terminals TB and TL when the MOSFET 16 is turned off while the switch 12 is closed. When the switch 12 is open, no voltage is produced between the terminals TB and TL. In addition, the voltage produced between the terminals TB and TL when the switch 12 is closed while the MOSFET 16 is set to the non-conductive state is substantially equal to the voltage of the battery.

 Thus, according to this embodiment of the invention, it is possible to assess whether or not the voltage between the terminals TB and TL is lower than a predetermined voltage by checking whether or not the transistor TR9 is made non-conductive. in such a way that a bias voltage corresponding to the voltage between the terminals TB and TL is produced by means of the resistors R21 and R22, and this bias voltage is then applied to the base of the transistor TR9. The transistor TR10 is made conductive / non-conductive at the same time as the MOSFET 16, so that it is possible to assess the conductive / non-conductive states of the MOSFETs 16 from this state. When the respective transistors TR9 and TR10 are set to the non-conductive state, an assessment is made either of the open state of the switch 12 or of the state of lowering of the battery voltage, so that the transistor TR4 fast discharge is made conductive (at the same time, the transistor TR8 preventing direction of the present embodiment, with the exception of the MOSFET 16, the resistor R16, the integration circuit consisting of the resistor RO and the driver of charge / discharge CO, and capacitor C1 of the power supply, are stored in the same integrated circuit assembly.This is similar to the case of Figure 2. In Figure 9, the symbol "VR" indicates the voltage present across the terminals. the indicator lamp lOR, and the symbol "VL" the voltage present at the terminals of the lamp lOL, and the symbol "VC" the voltage present across the capacitor C1 of the power supply, respectively.

 The operations of the vehicle direction indicating device having the circuit arrangement described above according to the fifth embodiment will now be described in conjunction with the timing diagram of FIG. 10. In FIG. 10, the symbols VOS, VQ, VC, VR and VL represent the voltages appearing at the various parts of the circuit of FIG.

 As shown in FIG. 10, when the switch 12 is closed by the driver of the vehicle to turn on the lamp 1OR, and therefore is in the closed state, the two transistors TR4 and TR8 are turned on. if the voltage VB of the battery does not reach a predetermined voltage VTD, so that both ends of the charge / discharge capacitor CO are short-circuited, and the output of the comparator CO1 is also controlled to be low. Consequently, in the case where the battery voltage is low, the power MOSFET 16 is not made conductive, and the low voltage is applied to the gate of the MOSFET 16, so that it is impossible to avoid that this MOSFET is not thermally destroyed.

 Then, if the battery voltage increases and reaches a predetermined voltage VTD, the two transistors TR4 and TR8 are in the non-conductive state.

As the voltage VOS across the charge / discharge capacitor CO becomes OV at this time, the output of the comparator CO1 becomes a high level. Then, the capacitor CO is charged by the output of this comparator CO1 until the voltage VOS present at its terminals reaches the upper limit voltage VTA.

When this charging operation begins, since the voltage VOS across the capacitor CO is equal to OV, the output of the priming comparator C05 becomes the low level. This situation continues for a predetermined period TD defined after the start of the charging operation, and until the voltage VOS across the capacitor CO reaches the starting voltage VTB. As a result,
MOSFET 16 is not made conductive after the start of the charging operation of the capacitor CO and until the lapse of the predetermined period TD.

After the expiration of the predetermined period, the MOSFET 16 is made conductive.

 Then, after the beginning of the charging operation of the capacitor CO and also after the lapse of the predetermined period TD, and the output of the comparator C05 then becomes the high level, the MOSFET 16 is periodically made conductive / non-conductive in response to the output of the comparator CO1, which varies in accordance with the oscillating circuit described above. The lamp lOR flashes in response to the conductive / non-conductive state.

 Then, when the switch 12 is open during this flashing operation, no voltage is produced between the terminals TB and TL even if the MOSFET 16 is in the non-conducting state. This situation is appreciated by transistors TR9 and TR10, and therefore transistors TR4 and TR8 are controlled to become conductive.

 As a result, when the switch 12 goes from the closed state to the open state, the charge / discharge capacitor CO is rapidly discharged by the transistor TR4, so that the voltage across this capacitor becomes OV. In addition, since the output of the comparator CO1 is controlled at low level by the transistor TR8, the MOSFET 16 is also turned off.

As described above, when after switching of the switch 12 to the open state, the driver of the vehicle acts on the switch 12 to turn on the lamp 1OR, or the lamp 1OL, and then the switch 12 is put in the closed state closed or
Closed), the oscillating circuit starts the charging operation of the capacitor CO if the battery voltage reaches the preselected voltage VDT.

 At this time, as the output of the comparator COS becomes the low level until the voltage VOS across the capacitor CO reaches the voltage VTB, the MOSFET 16 is brought to the conductive state after the closing of the switch 11 and after that a given period of time TD has elapsed. Then, this MOSFET 16 is set to the non-conducting state at the moment when the constant period of time has elapsed during which the voltage VOS across the capacitor CO becomes the upper limit voltage VTA.

 As a result, according to this fifth embodiment, when the voltage of the battery reaches the preselected voltage VTD, the timing of the ignition of the lamp 1OL, or the lamp 1OR, immediately after the transition to the closed state of the Switch 12, and the firing time can be made permanently stable.

 The reason why the attack operation of the MOSFET 16 is prevented by the operation of the comparator C05 during such a given period TD while the switch 12 is placed in the closed state, hence the beginning of the operation of charging the capacitor CO by the oscillating circuit, and then the voltage VOS across this capacitor reaches the ignition voltage VTB, is to safely store the electrical charges capable of attacking the device during this period of prevention. As TD period, a period of time is established which requires charging in the capacitor C1 of the power supply, at least electric charges that are consumed to provide the supply voltage when the MOSFET 16 is made conductive.

 As a result, according to this embodiment, even when the switch 12 is closed while the capacitor of the power supply is completely discharged, the electrical charges stored in the capacitor C1 are discharged while the MOSFET 16 is turned on. There is no risk that the conduction time of MOSFET 16 becomes unstable. Thus, the stable conductive / non-conductive control of the MOSFET can be performed continuously.

[SIXTH EMBODIMENT]
The vehicle steering indication device according to this sixth embodiment consists of dividing resistors R17 and R18 for subdividing the output voltage VT of the constant voltage circuit so as to produce a reference voltage VTB '(voltage d priming) which is slightly lower than the lower limit voltage VTC. In addition, this device uses a boot comparator C05 to compare the firing voltage VTB '= VTxR18 / (R17 + R18) subdivided by the resistors R17 and R18 with the voltage VOS across the capacitor charge / discharge CO. This comparator C05 outputs a signal at the low level when this voltage VOS is lower than the voltage VTB ', and another signal at the high level when the voltage VOS becomes greater than the voltage VTB'.

In this device, the output of the comparator C05 is applied to the ND1 circuit with a NAND gate, so that the level of the signal leaving the circuit ND1 necessarily has the high level until the voltage VOS of the capacitor CO reaches the voltage VTB. whatever the output of oscillating comparator CO1.

Therefore, the power MOSFET 16 is made non-conductive.

 In other words, according to this sixth embodiment, the blinking signal S1 (driving signal) applied to the driving transistor TR6 can be obtained by subjecting to a NAND operation the signal that is produced by inverting the signal. output of the comparator CO1 by the inverter IN2, as well as the output signal of the comparator C05.

[SEVENTH EMBODIMENT]
It will be noted that in the fifth embodiment explained above, an evaluation circuit consisting of transistor TR9 is used to assess the state of lowering of the voltage between terminals TB and TL, and of transistor TR10 for appreciate the non-conductive state of the power MOSFET 16 as a judgment circuit for both the open state of the direction indicating switch 12 and the state of lowering the battery voltage on the basis of the voltage between terminals TB and TL when MOSFET 16 is made non-conductive. Since the evaluation operation can not be performed in such an evaluation circuit when the MOSFET 16 is turned on, when the switch 12 is open during the conducting state of the MOSFET 16, this situation can not be detected by this circuit. of appreciation, so that the charge / discharge capacitor CO can not be discharged quickly.

 To quickly detect the switching operation of the switch 12 from the closed state to the open state regardless of the conductive / non-conductive states of the MOSFET 16, a judgment comparator C06 can be used as is shown in FIG. 11. More precisely, this comparator C06 compares a voltage appearing at the junction point between the diode D1 and the resistor R16, which is applied via the resistor R25, to the reference voltage obtained by subdividing the constant voltage VT coming from the constant voltage circuit using resistors R26 and R27. Then, when this voltage of the junction is lower than the reference voltage, that is to say that the voltage present between the terminals TB and TL is lower than the reference voltage, a high level signal leaves the comparator C06 .

 In other words, if the switch 12 is in the closed state while the MOSFET 16 is in the conductive state, the current to turn on the lamp lOL, or the lamp lOR, then flows between the terminals TB and TL. As a result, although a very low voltage is produced between the terminals TB and TL, when the switch 12 is open, no current flows between the terminals TB and TL, so that there is no production of a very low voltage. As a result, as shown in FIG. 11, if such a measurement is taken, the operation of the switch 12 at closing-opening can be judged as unrelated to the conductive / non-conductive states of the MOSFET 16, so that the capacitor CO can be discharged quickly; that is, a very low reference voltage is produced by means of the resistors R26 and R27, the evaluation capacitor C06 appreciates the open state of the switch 12 by checking whether or not the voltage of the junction between the resistor R16 and the diode D1 is lower than this very low reference voltage. Then, the transistors TR4 and TR8 are turned on in response to the signal at the high level obtained from this comparator C06.

 Note that the comparator C06 can not appreciate the state of lowering of the voltage of the battery when the open state of the switch 12 can not be appreciated by this comparator C06 even if the MOSFET 16 is conductive. Therefore, as in the first embodiment of FIG. 1, another judgment circuit must be employed so that the state of lowering of the battery voltage is appreciated to stop the attack operation of the battery. MOSFET 16. This circuit can appreciate the state of lowering the battery voltage based on the voltage between terminals TB and TL during the non-conducting state of MOSFET 16.

 FIG. 11 shows an arrangement of the circuits of the direction indicating device according to the second embodiment, in which the evaluation circuit consisting of transistors TR9 and TR10 employed in the device indicated in FIG. 9 has been changed into a circuit. of appreciation, explained above, which consists of the comparator C06. Since parts of the circuit other than those described above are entirely similar to those of the device of FIG. 1, the same reference numbers used in FIG. 1 are employed, and their description is omitted.

 The present invention is not limited to the embodiments which have just been described; it is, on the contrary, susceptible of modifications and variants which will appear to those skilled in the art.

Claims (7)

 1 - Device for attacking a load, characterized in that it comprises  a charge / discharge capacitor (CO);  a charge / discharge circuit for charging / discharging the capacitor at a given time constant in such a way that the voltage across the capacitor varies between a given upper limit voltage and a given lower limit voltage;  a driving circuit (31) for conducting a power transistor (16) connected in series to the current path used to supply a load (lOL; lOR) with a current from a current supply source; continuous (14), from where the charging of the charge when the charging / discharging circuit causes charging of the capacitor, or the voltage present across the capacitor is greater than or equal to a predetermined voltage which is defined between the voltage upper limit and lower limit voltage;  overcurrent evaluation means for judging whether or not the voltage at the terminals of the power transistor which is connected to said current path exceeds an overcurrent judgment voltage;  - upper limit voltage change means for changing said upper limit voltage at which the charging / discharging circuit stops the charging operation of the capacitor and begins its discharge operation, to make it take a value normally greater than a voltage judgment of attack judgment when said overcurrent judgment means appreciates that said terminal voltage exceeds said overcurrent judgment voltage; and  a first drive stop means (26) for rendering the power transistor (16) necessarily non-conductive and furthermore for maintaining the charge / discharge circuit in the state of charge of the capacitor (CO), judging whether whether or not the voltage present at the terminals of the capacitor is greater than or equal to said judgment voltage of judgment of attack when this voltage at the terminals of the capacitor becomes greater than or equal to the judgment voltage of judgment of attack.  2 - Device according to claim 1, wherein the overcurrent evaluation means produces a voltage corresponding to a current flowing through the load, and furthermore, corrects this voltage to a value adapted to the temperature characteristic of the power transistor ( 16).  3 - Device according to claim 1 or 2, characterized in that  the current path has a connection terminal (TB) of the DC power source (14) and a connection terminal (TL) of the load, and a load connection switch (12) for connect the load connection terminal, and load is provided at this load connection terminal;  - an overvoltage protection device (30) is employed in the load driving device equipped with an auxiliary power source (C1; R16) connected in parallel with the power transistor (16), in order to storing electrical charges in a supply capacitor (C1) by a voltage appearing across said transistor (16), and producing a supply voltage by the charges stored in the capacitor and the voltage present at both terminals of the connection the DC power source and the load connection (TB; TL); and a circuit of the load actuated by receiving the power supply from said auxiliary power supply, to turn the power transistor (16) on / off in response to a control signal (S1) input from the exterior, whence the attack of the charge;  an overvoltage judging means for judging that an overvoltage occurs in the DC power source when the voltage applied from the auxiliary power source exceeds a given rating voltage;  a judgment voltage setting means for setting said judgment voltage to a first voltage unless said overvoltage judging means evaluates the appearance of the overvoltage, and for establishing the judgment voltage at a first voltage; second value lower than the first value of appreciation according to a given voltage when the overvoltage judging means appreciates the appearance of the overvoltage, and then until the overvoltage judging means appreciates the appearance of the overvoltage; surge; and  a protection means for making the power transistor (16) conductive when the overvoltage judging means appreciates the appearance of this overvoltage.  4 - Device according to any one of claims 1 to 3 above, characterized in that:  output voltage detection means is provided for the DC power source, or the auxiliary power source; and  when said output voltage detecting means detects a lowering state of the voltage output from the DC power source, or the auxiliary power source, it stops the driving operation of the power.  5 - Device according to claim 4, characterized in that, when the output voltage detection means detects the lowering state of the voltage output from the DC power source, or the power source auxiliary, the second drive stop means stops the driving operation of the power transistor by changing the output of the charging / discharging circuit.  6 - Device according to claim 4 or claim 5, characterized in that  - state detection means is provided which detects the lowering state of the output voltage of the output voltage detecting means and also the conductive / non-conductive states of the power transistor (16); and  when the non-conducting state of the power transistor is detected by this state detection means, the charge / discharge capacitor is rapidly discharged.  7 - Device according to any one of claims 4 to 6 above, characterized in that it further comprises  a circuit preventing the output of a driving signal so as to prevent the output of the driving signal of the power transistor between the charging / discharging circuit and the driving device of the power transistor when the voltage at the terminals the charge / discharge capacitor becomes less than or equal to a given voltage lower than two different values which are changed during charge / discharge operations.