FR2501447A1 - Transmission matching unit for sonar transducer - uses parallel inductor switched into circuit by thyristors to present zero phase shift for transmitter - Google Patents

Abstract

The output of an oscillator (1) is connected through a coupling transformer (2) to a transducer (3) whose equivalent circuit comprises an inductor (LM) equivalent to the mass of material. A capacitance (CM) is equivalent to the material rigidity and a resistor (RM) corresponds to the mechanical losses and radiated energy. A second capacitor (CO) corresponds to the dielectric losses and is the significant component. An inductor (L) is placed in parallel with the transducer and is in parallel with a variable inductor (LV) in series with back to back thyristors (Th1,2). Toroidal detectors (5,7) sense voltages proportional to phase of the respective currents (IE,IL) w.r.t. the voltage (VL) from a divider circuit. A control circuit (4) containing a servo loop compares the phase differences between the voltage and generator current (IE). The error signal controls the thyristor to maintain the phase difference close to zero by modifying the current in the variable inductor.

Description

Method and apparatus for automatically matching the impedance of an electro-acoustic transducer

 The present invention relates to a method and a device for automatically matching the impedance of an electroacoustic transducer, which is excited by an electronic transmitter.

 The technical field of the invention is that of sonar construction.

 It is recalled that the active sonars comprise an emitting antenna composed of electro-acoustic transducers, which are generally transducers composed of piezoelectric ceramic plates. These transducers are excited by emitters that are electronic oscillators.

 Piezoelectric transducers have a reactive impedance that varies with frequency.

 In the case where it is desired to transmit in a narrow frequency band, it is easy to adapt the impedance of the transducer to amounts in parallel with the transducer, a choke whose inductance is calculated to compensate for the reactance of the transducer in the Frequency band used, ie an inductance such that the total impedance of the transducer and the choke is close to pure resistance.

 When it is desired to use a transducer with a modulated frequency transmitter, for example with a transmitter that emits variable frequencies in a band having an octave width, this solution is no longer suitable. It is then possible to use a plurality of inductances calculated to correct the impedance for different frequencies and a switch which makes it possible to put the inductances into service one after the other according to the frequency transmitted, but this solution is bulky, expensive and not keep on going.

 The object of the present invention is to provide means for adapting the impedance of a transducer continuously, especially when this transducer is excited by a continuously modulated frequency transmitter in a wide frequency range, for example a range having a width of about one octave.

 To adapt the impedance of an electro-acoustic transducer which is excited by an electronic transmitter, a first inductance is known in parallel with the transducer. In a method according to the invention, a second inductor is mounted in series with a switch and said second inductor and said switch are placed in parallel with the first inductor, the phase difference between the intensity IE of the current and the voltage is continuously measured. VE issued by the transmitter and it automatically controls the closures of said switch by a control loop during a portion of each half-wave of the current IL through said first self, such that it tends to cancel said phase shift.

 According to one embodiment of the method, the phase difference between the intensity IE and the voltage VE is converted into a duration t proportional to this phase shift, the zero crossings of the intensity of the current IL are detected in said first inductor and the closing said switch after a time t after each of said zero crossings of the current IL.

 A device according to the invention for adapting the impedance of an electro-acoustic transducer is of known type having a first inductor placed in parallel or in series with the transducer.

 The device according to the invention furthermore comprises a second inductor mounted in series with a switch and in parallel with said first inductor, means for measuring the phase difference between the intensity of the current and the voltage delivered by the emitter and a loop. servo, which automatically controls the closures of said switch, during a portion of each half-wave of the current through said first inductor, so that the inductance value of the two inductors connected in parallel varies in the direction which tends to cancel said phase shift. Preferably, the switch is composed of two thyristors mounted in opposition.

 A device according to the invention comprises a phase comparator which measures the phase difference between the intensity and the voltage and which delivers a DC voltage proportional to this phase shift, a generator of a voltage which increases linearly from a zero value from each zero crossing of the intensity of the current through the first inductor, a comparator which compares said constant voltage with said linearly variable voltage and which emits a signal when the increasing voltage is greater than said constant voltage, a monoset which emits a signal at each rising edge of the signal emitted by said comparator and circuits which direct the pulses of said monostable alternately to one or the other of said thyristors.

 According to a preferred embodiment, a device according to the invention comprises a first sensor which measures the intensity of the current emitted by the transmitter, a second sensor which measures the intensity of the current through the first inductor, two comparator circuits, which transforms the voltages emitted by said sensors into rectangular slots, an "exclusive" gate on the two inputs from which the outputs of said comparators are connected, a low pass filter and a third comparator which compares the DC voltage emitted by said filter to a reference voltage and which emits a constant voltage proportional to said phase shift.

 The device further comprises a third sensor which measures the voltage across said first inductor, a fourth comparator which compares the voltage emitted by said third sensor to a threshold and converts said voltage into a rectangular slot, a circuit. or "exclusive" whose two inputs are connected to the outputs of said second and fourth comparators, a transistor whose base is connected to the output of said circuit "or exlcusif", an integrator which integrates a constant voltage and which is controlled by said transistor and a fifth comparator which compares the integrated voltage supplied by said integrator and the constant voltage delivered by said third comparator and emits a voltage when the integrated voltage is greater than the constant voltage.

 Preferably, the first inductor has an impedance Lu which is calculated to compensate for the reactance of the transducer when the transmitter is transmitting on the lowest frequency of the frequency band used.

 The invention results in new means for automatically matching the impedance of a transducer to maintain it between certain permissible limits for the transmitter and to reduce reactive energy losses.

 The means according to the invention being based on a control loop which is controlled by a voltage proportional to the phase shift, an automatic correction is obtained regardless of the causes of variation of the impedance of the transducer. In particular, the impedance of the transducer can be adapted to variations in frequency over a wide band, for example on a frequency-modulated transmission band within an octave. It is also possible to correct the impedance variations due to variations in the property of the transducer, or to factors related to the medium in which it is used.

 The use of a transducer equipped with an automatic impedance matching device according to the inveation, with a constant voltage or constant current emitter, makes it possible to obtain an optimum level of noise emission inside. a wide band of frequencies.

 The automatic impedance matching devices according to the invention are entirely static devices, which can be made in integrated circuits, and thus robust and space-saving circuits, which can be associated with an active sonar transmitting antenna on board a boat. or on an underwater vehicle.

 The following description refers to the accompanying drawings which show, without any limiting character, an exemplary embodiment of a device according to the invention.

 Figure I shows the equivalent diagram of a piezoelectric transducer connected in parallel with an inductor.

 FIG. 2 is a polar coordinate diagram showing the variations of the impedance of the circuit according to FIG. 1 as a function of frequency.

FIG. 3 is a diagram of a device according to the invention
FIG. 4 is a diagram in the form of a block diagram of the control loop of a device according to the invention.

 FIG. 5 is a diagram showing the variations of the intensity of the current in the two inductors of a device according to the invention.

 FIG. 6 is a diagram of one embodiment of the circuits for controlling a device according to the invention.

 FIG. 7 is a diagram showing the signals at various points of the circuit according to FIG. 6.

 Figures 8, 9 and 10 are curves taken at one oscillograph.

Figure 1 shows schematically the output amplifier of a transmitter, which is an electronic oscillator, which is connected through a coupling transformer 2 to a piezoelectric transducer 3, represented by a dashed rectangle. The equivalent diagram of the transducer, which is composed in parallel of a capacitance Co, of a resistance Ro corresponding to the dielectric losses, and of a series-connected assembly of a resistor RM corresponding to each other, is represented inside the rectangle. mechanical losses and radiated energy, with a CM capacity equivalent to the rigidity of the material and an LM inductance equivalent to the mass of the material.
Co is preponderant, so that the transducer has a very reactive impedance, close to that of a pure capacitance.

 If this impedance is not corrected, a large part of the energy consumed by the transmitter is lost in reactive energy.

 When the emitter emits in a narrow band fo (wo), the impedance of the transducer 3 is adapted by mounting in parallel therewith a choke having an inductance L whose value is given so that at the frequency fo, Impedance equivalent to the self and transducer connected in parallel is equivalent to pure resistance.

 FIG. 2 represents, in polar coordinates, the variations of the impedance of the inductor L and of the transducer 3 in parallel in an example where the frequency of the emitter 1 varies within an octave, for example between 2 , 5 KHz and 5 KHz and where the inductor L has an inductance computed so that the impedance is e-equivalent to a pure resistance (argument O = o) for a frequency fo = 3 KHz. We see on this diagram that the impedance is inductive (0> o) for the frequencies between 2.5 lCHz and 3 KHz and capacitive (O <o) for the frequencies between 3 KHz and 5 KHz.

 Electronic transmitters are constructed in such a way that they must charge on an impedance located between certain limits to deliver their nominal power. The argument of this impedance must be within a range of + 40 ", the limits of which have been represented by phantom lines OMI and OM2 in Figure 2. It can be seen in Figure 2 that if an impedance adapter device composed of an inductor L is used, the range in which the transmitter could operate would be from F'1 = 2670 Hz to F'2 = 3370 Hz.

 The objective of the invention is to provide means for adapting the impedance of a transducer by adding thereto a circuit whose impedance varies automatically so that the total impedance of the transducer and the circuit of corrections is close to pure resistance and remains within limits where most of the power of the transmitter is converted into acoustic energy and within a frequency range whose extent may be equal or greater than one octave.

 FIG. 3 represents the power circuits of a device according to the invention.

 This figure shows the transmitter 1, the connection transformer 2, the transducer 3 and the fixed inductor L.

A variable inductance Lv is connected in series with a switch which is composed for example of two thyristors Th1 and
Th2 mounted head to tail. These two thyristors can be replaced by an alternistor type circuit or any other power switch that can operate in both directions of the current through the self-L
v
The device further comprises electronic circuits 4 for controlling the switches, sensors which are connected to these circuits and N and M conductors which automatically control the closures of thyristors Th1 and Th2.

The sensors are a torus-type sensor 5 placed around the output of the transformer 2, which delivers a voltage proportional and in phase with the intensity IE delivered by the transmitter; a voltage divider 6 which delivers a voltage VL proportional and in phase with the voltage VE delivered by the transmitter, another torus-type sensor 7 placed around the circuit which passes through the inductor L, which delivers a proportional voltage and in phase with the intensity IL through the self-L and a sensor 8, of the torus type, which is placed around the circuit which passes through the self-L and
v which delivers a proportional voltage and in phase with the intensity 1Lv which crosses the self L
v
The control circuits 4 comprise a control loop which compares the phase difference Qf between the voltage VL and the intensity IE delivered by the generator 1 and which tends to automatically maintain this phase difference equal to or close to zero by passing through the current through the inductance L during
v part of each alternation only. The passage time of the current through L or angle of passage 9 'of the current changes the
v rms value of the current through the inductor and makes it possible to vary the inductance equivalent to the two inductances L and L mounted
v in parallel and correct this equivalent inductance in the direction that tends to cancel the phase difference A.

FIG. 4 is a schematic block diagram showing the essential parts of a servo loop according to the invention. This figure shows the transmitter 1, 2, the transducer 3, the fixed inductor L and the variable inductor L connected in series with a switch I
vn
Block 9 represents a phase comparator which measures the phase difference + between voltage VL and intensity IE and delivers a voltage e which is proportional to this phase difference.

Block 10 represents a comparator which compares the voltage e with a reference voltage eO. In the case where it is desired to maintain the phase difference between VL and IE equal to zero, this reference voltage e is zero. The comparator 10 delivers a voltage
o which is proportional to the difference e - e.

 The block 11 represents a circuit which delivers periodic signals whose frequency is twice that of the intensity IE and which are produced at a time t after each zero crossing of the current IL through the inductor L, the time t being proportional to the voltage el delivered by the comparator 10.

The periodic signals delivered by the circuit 11 control the closures of the switch I and thus determine the conduction of the self-inductance L during each half-cycle.
v current.

 Figure 5 shows the abscissa time, and ordinate, the voltages or currents. This dashed figure shows the voltage VL proportional to VE and, in solid lines, the intensity IL through the fixed inductor L. These two curves are sinusoidal and the voltage across the inductor is in advance of phase 90 on the intensity through the self.

The phantom curves ILv1 and 1Lv2 represent the variations in the intensity of the current through the inductor Lv. t1 and t2 are the times delivered by the circuit 11 of FIG. 4. Before the time t1, the two thyristors Th1 and Th2 are open, no current passes through the self-L. At the time tl the signal emitted
v by the circuit 11 controls the closing of Th1. A current ILv1 starts to flow in the self Lv and grows until the voltage VL at the terminals of the self becomes zero. During this period, the self stores reactive energy which it then restores until the current ILv1 becomes zero. At this time, Th1 opens automatically. At the beginning of the second half-wave of IL, the two thyristors are open. After a time t2, after the zero crossing of IL, the circuit 11 commands the closing of
Th2. A current ILv2, of opposite direction of the preceding one, starts to circulate in the self L and the absolute value of this current increases
v until the voltage VL crosses zero, then decrolt.

It can be seen in FIG. 5 that the evolution of the current ILv in the inductor L is substantially parallel to the evolution of the neck
IL in the self L. The angle in FIG.
O 'of current flow in t. This angle varies between 0
v the self Lv is not used until 180 when the self L
vv always stays in service.

 The value of the current passing angle # 'through the second inductor Lv, during each half-cycle, determines the value of the rms current through the inductor and hence the rms value of the inductance Lv.

A choke whose inductance is k takes the value L
qv for an angle 0 '= 1800 and higher and higher values when 0' decreases to reach an infinite value for 0 '= 0.

As a result, the total inductance of the two inductors L and Lv, connected in parallel, decreases by a value equal to L for 0 '= 0 to L.Lv a value equal to L + Lv for #' = 180.

 FIG. 6 represents an embodiment of the control circuits 4 of FIG. 1 in the form of a control loop according to FIG. 4.

On the left side of the figure are the sensors 5, 6 and 7, which respectively deliver voltages proportional to IE, VL and
These sinusoidal voltages are shaped by three limiter circuits 12, 13, 14 which deliver signals VLE> ILe and
IEe having the form of rectangular slots. Circuits 12, 13 and 14 are for example threshold comparators.

 The phase comparator 9 comprises a logic or exclusive circuit "15 on the two inputs of which are connected the outputs of the secretors 13 and 14. This circuit therefore receives the signals in crenellals ILe and IEe.It will be noted that ILe corresponds to the intensity at through the fixed inductor L which is in phase quadrature with the voltage VL.

 The output of the circuit 15 is connected to a low-pass filter 16.

 If IL and IE are in quadrature phase, that is to say if IE and VL are in phase, the slots obtained at the exit of the "exclusive" gate 15 are of the same width and are obtained at the output of the filters a null DC component.

 If, on the other hand, a phase difference occurs between the voltage VL and the current IE, the slots delivered by the circuit 15 no longer have the same width and the filter 16 delivers a direct current whose value is proportional to the phase shift.

 The output of the filter 16 is connected to a differential amplifier input 17 whose second input is connected to a reference voltage which is equal to zero in this embodiment. The differential amplifier 17 is equivalent to the block 10 of FIG. 4. The output of the amplifier 17 is connected to an input terminal of a comparator amplifier 18.

 The outputs of the clippers 12 and 13 are connected to the two inputs of a second "exclusive" circuit 19.

 It will be recalled that VL and IL are in quadrature and rectangular slots are obtained at the output of the circuit 19, the falling edges of which correspond to the zero crossings of the current IL in the self-L. The output of the circuit 19 is connected to the base of a transistor 20 which controls an integrator circuit 21 which integrates a DC voltage delivered by a source 22.

 The falling edges of the slots delivered by the circuit 19, which correspond to the zero crossings of the current 1L1 control the upright of an integration from which the circuit 21 delivers a voltage that grows linearly. The rising edges of the slots delivered by the circuit 19 control the start of the discharge of the capacitor 21a from which the voltage delivered by the integrator 21 decreases linearly.

 The output of the integrator circuit 21 is connected to the second input of the comparator 18 which compares the variable voltage delivered by the integrator 21 to the DC voltage delivered by the comparator 17.

 From the moment when the integrated voltage begins to increase, an instant which coincides with a zero crossing of the current IL, the time t necessary for the integrated voltage to be equal to the DC voltage delivered by the comparator 17 is proportional to this DC voltage since the integrated voltage increases proportionally to the time. The comparator 18 transmits a logic signal that changes sign each time the integrated voltage is equal to the DC voltage delivered by the comparator 17. The output of the comparator 18 is connected to an input of an AND gate 23. The output of the gate 23 is connected to a monos table 24.

 The second input of the AND gate 23 is connected through an inverter 25 to the output of the circuit 19.

The output of the monos table 24 is connected in parallel to inputs of two AND gates 27 and 27, whose outputs are connected through two inverter circuits 28 and 29 on the bases of two M and N transistors mounted in switches, which control, through coupling transformers 30 and 31, the closing gates of thyristors Th1 and Th2,
FIG. 7 is a diagram which represents on the abscissa the time and on the ordinates, on different lines, the signals at various points of the circuit of FIG. 6 identified by letters.

 Line 1 represents the rectangular slots IL delivered by the nibbler circuit 13.

 Line 2 represents the slots VLe delivered by the circuit 12.

 Line 3 represents the slots at point A located at the exit of the "exclusive" circuit 19 which receives on its two in tren slots ILe and VLe, which are in quadrature phase.

 Line -4 represents the signals at point B at the output of the inverter 25.

Line 5 represents in solid lines the integrated voltage delivered at point D by the circuit 21,
We see that the decreasing slope is stronger than the increasing slope. FIG. 5 also shows, by two horizontal dashed lines, the DC voltages Vb 1 and Vb 2 delivered by the comparator 17 when the phase shift between VE and IE is close to -100 and +100, respectively. mites correspond to the range in which the servo-control acts and the phase-shift must therefore remain between these two limits.

Line 6 corresponds to the signals delivered at point F by comparator 18 whenever the integrated voltage is equal to voltage VBI.
Line 7 corresponds to the signals at point H, at the output of the AND gate 23, which combines the signals F and B. The gate 23 is an inverting gate. The rising edges of the signal F are transformed by this gate into falling fronts which control the monostable 24. The line 8 represents in full lines the signals emitted by the monostable during the falling edges of the signal 7,
At the start of the installation, when the servocontrol has not yet reduced the phase shift in the range between Vbl and Vb2, If the phase shift is less than -100, the integrated voltage is always greater than Vb1 and the signal F has a constant positive value. The AND gate 23 then follows the variations of the signal B and therefore controls the monostable during each zero crossing of the current IL. The inductor L is in operation permanently until
that the phase shift becomes greater than -100 and that the servo becomes operational.

If, on the other hand, the initial phase shift is greater than +100, the integrated voltage 5 is always smaller than Vu2, the signal 6 is always zero and the AND gate remains constantly closed. In this case, the variable choke L is permanently switched off.
v until the phase shift becomes less than + 1Q0.

 Lines 9 and 10 represent the signals sent on the bases of transistors M and N.

Line 11 represents, in solid lines, a signal IEe in quadrature of phase with IL and in dotted lines the same signal 1E in advance or in delay with respect to a signal in quadrature phase,
Line 12 represents in full lines the signal at the point
C at the output of the circuit "or exclusive" 15, when 1E and VE are in phase and dotted the signal at point C when IE and VE have a phase difference.

Line 13 represents the DC voltage delivered by the filter circuit 16. The dc voltages Vb 1 and V obtained for respective phase shifts of FIG.
b2 - 100 and + 100 The gain of the servo loop is adjusted for example to maintain the phase shift between these two limits,
A value of the fixed inductance L is chosen such that it compensates the reactance of the transducer for a frequency F within the frequency range of the transmitter. Preferably, the value L is chosen to compensate for the reactance corresponding to the lowest frequency F1. Thus, for low frequencies, the variable inductance L does not intervene. On the contrary, for the frequencies
v the strongest, close to F2, the variable impedance L is
v- constantly in use, that is to say that the current passage angle 0 'is close to 1800. As a result, there is no distortion at both ends of the frequency range and the Acoustic signals are sinusoldals. On the other hand, for intermediate frequencies, the inductance Lv will be switched on during part of each half-cycle (o <0 '<1800).

 However, by construction, it is generally transmitted in a frequency band which frames the natural frequency or resonant frequency of the transducer. As a result, at the center of the frequency band, the transducer forms a mechanical filter that corrects the distortion.

Figure 8 illustrates this correction well. This figure corresponds to real curves recorded on the oscillograph. The upper sinusoidal curve represents the variations of the voltage VE. The second curve represents the variations of the current 1E The third curve represents the current ILv through the inductance
L in an average case where the angle O 'of current flow is
v the order of 900 and where the frequency Fo is of the order of 3 KHz for a transmission band from 2.5 KHz to 5 KHz. The fourth curve P is that of the sound pressure emitted by the transducer.

It is clearly seen in this figure that VE and IE are in phase, that i is strongly distorted, but that the acoustic signal P is perfectly sinusoidal.

 FIGS. 9 and 10 are diagrams of the same type recorded on the oscillograph in the case where the emitter transmits on a frequency slightly lower than F2 which is the highest frequency of the emission band.

FIG. 9 represents the phenomenon in the case where the servocontrol loop (that is to say
ILv = o). It can be seen that the voltage VE and the current IE are strongly out of phase and almost in quadrature.

Figure 10 shows the phenomenon when putting the servo loop into service. In this case, the angle and current flow through the variable inductance L is close
v from 1800, as shown in curve IL. The voltage VE and the current IE are in phase. There is no distortion of the sound signal P.

 Figure 10 corresponds to a lower power transmitter operation than in the case of Figure 9 where the reactive energy is consumed.

 Divided in FIG. 6 are optional circuit portions which comprise a low-pass filter 32 to which is connected the output of the sensor 8, a differential amplifier 33 and a comparator and limiter amplifier 34, the two inputs of which are connected to one another. one on the output of the differential amplifier 33 and the other on the output of the sensor 7.

These circuits make full use of the inductor
L avoiding that the current IL which circulates in it does not compor
v Lv te a continuous component if the closing signals of the thyristors during the successive half-cycles are not symmetrical in time. In this case, the DC component is eliminated by moving the reference of the comparator 34 which delivers a slot-shaped signal ILv1 and it is this signal ILyl which is connected to an input of the "exclusive or" gate 19 instead of the signal ILe, as shown in dashed lines in Figure 6.

 The maxima of the integrated voltage delivered by the integrator 21 (signal D) vary inversely with the frequency, but this is not a problem since the high parts are only used at low frequencies. However, to remedy any risk of malfunction, is advantageously used a voltage D whose slope increases when the frequency increases.

 This slope variation is easily obtained by using as a constant voltage source 22 a frequency converter in a constant voltage proportional to the frequency and by connecting the output of the shaping circuit 12 to this converter in order to obtain the frequency signal according to FIG. link shown in dashed lines in FIG.

 The device for adapting the impedance of a transducer which has just been described proceeds by measuring the phase difference between the voltage and the intensity of the current delivered by the emitter and by a control loop comprising a variable inductance of which the value is automatically corrected to reduce this phase shift to zero and maintain it around zero.

 Due to the fact that these devices operate by phase shift control, they have the advantage of automatically correcting the impedance of the transducer not only as a function of the frequency, but whatever the reasons that require such a correction, such as for example the variations of the transducer's own impedance due to aging or to the physical parameters of the ambient medium: pressure, salinity, temperature.

 In the case where the emitter is at constant voltage, the devices according to the invention make it possible to best use the power of the emitter. In the case where the transmitter is at constant current, they make it possible to obtain a sound emission level as a function of the substantially constant frequency.

Figures 3, 4 and 6 show an embodiment in which the inductors L and L are connected in parallel with the
v transducer.

 It would also be possible to mount a constant self and a variable inductor in parallel with each other and in series with the transducer.

 An embodiment of a device according to the invention composed of analog circuits has been described. Of course, one could also use digital circuits that fulfill the same functions.

Claims (8)

1 - Method for automatically matching the impedance of a transducer  electroacoustic transmitter which is excited by an electronic transmitter  that, according to which a first self is placed in parallel or in  series with said transducer, characterized in that one ascends  a second self in serine with a switch and place said  second self and said switch in parallel with the first  self, we measure continuously the phase difference between the intensity (IE) of the  current and voltage (VE) delivered by said transmitter and  automatically controls the closures of said switch by a  servo loop during part of each half-alternation  this of the current (IL) through said first self as one  tends to cancel said phase shift. 2 - Process according to claim 1, characterized in that one  converts the phase difference between the intensity (IE) and the voltage (VE)  in a duration (t) proportional to this phase shift, it detects  the zero crossings of the current intensity (ZL) in said  first self and one controls the closures of said switch  after a time (t) after each of said zero crossings. 3 - Device for automatically matching the impedance of a transducer  electro-acoustic transmitter (3) which is excited by an electro transmitter  nique (1) of the type comprising a first self (t) place er, paral  or in series with said transducer, characterized in that  has a second self (Lv) connected in series with an interrup  (Th1, Th2) and in parallelwith said first self,  means (9) for measuring the phase difference between the intensity (IE) of the  current and voltage (VE) delivered by the transmitter and a  servo loop (9, 10, 11) which automatically controls  the closures of said switch (Th1, Th2) during a game  of each half-wave of the current through said first  self, so that the value of the inductance of the two selfs  mounted in parallel varies in the direction that tends to cancel said  phase shift. 4 - Device according to claim 3, characterized in that said  switch is composed of two thyristors (Th1, Th2) mounted in  opposition. 5 - Device according to claim 4, characterized in that  comprises a phase comparator (9) which measures the phase difference between  the intensity (IE) and the voltage (VE) and which delivers a voltage  continuous proportional to this phase shift, a generator (11) of a  voltage that grows linearly from a zero value from  of each zero crossing of the intensity (IL) of the current through  the first self (L), a comparator (8) which compares said ten  constant voltage at said linearly variable voltage and which emits  a signal when the increasing voltage is greater than said  constant voltage, a monostable (24) which gives a signal when  each rising edge of the signal seeded by said comparator (18) and  circuits (26, 27, 28 and 29) which direct the pulses delivered  by said monostable alternately to one or the other of said  thyristors. 6 - Device according to claim 5, characterized in that  has a first sensor (5) which measures the intensity of the current  (IE) seeded by the transmitter, a second sensor (7) that measures  the intensity of the current (IL) through the first self, two cir  cookers comparators (13 and 14), which transform the vol  its by said sensors in rectangular slots, a door  "or exclusive" (15) on the two inputs from which the outputs  said comparators (13 and 14) are connected, a filter passes  base (16) and a comparator (17) which compares the DC voltage  transmitted by said filter to a reference voltage and which gives  a constant voltage proportional to said phase shift. 7 - Device according to claim 6, characterized in that  comprises a third sensor (6) which measures the voltage (VL) at  terminals of said first self, a comparator (12) which compares  the voltage delivered by said third sensor to a threshold and which  transforms said voltage into a rectangular slot, a circuit  "exclusive" (19) whose two entries are connected to the  outputs of said comparators (12, 13), a transistor (20) whose  the base is connected to the output of said "or exclusive" circuit  (19), an integrator (21) which integrates a constant voltage and  which is controlled by said transistor and a comparator (18) which  compares the integrated voltage supplied by said integrator (21) and  the constant voltage delivered by said comparator (17) and which  resets when the integrated voltage is greater than the  constant voltage. 8 - Device according to any one of claims 3 to 7  for adapting the impedance of an electro-acoustic transducer  who is excited by an electronic transmitter that delivers a frequency  this modulated in a broad frequency band between a  lower limit (F1) and an upper limit (F2) of  on both sides of a frequency (Fo) corresponding to the frequency  of said transducer, characterized in that the impedance (L)  of said first self is calculated to compensate the reactance  said transducer corresponding to the lowest frequency (F1).