FI92892C - Method for limiting the output current of a flyback type switch mode power supply in overload situations and a flyback type switch mode power supply - Google Patents

Description

92892

Method for limiting the output current of a flyback-type switch mode power supply in overload situations and a flyback-type switch mode power supply 5

The invention relates to a method according to the preamble of appended claim 1 for limiting the output current of a flyback type switch mode power supply in overload situations. The invention also relates to a flyback-type switch power supply according to the preamble of claim 5.

The share of switched-mode power supply in power supply design is constantly growing. This is due to their many advantages, such as good efficiency, a wide input voltage range and the ability to implement compact and lightweight power supplies. Switchback power supplies now increasingly use a flyback topology (topology refers to the circuit configuration that determines how power is transferred in a power supply). The biggest advantage of the Flyback-type 20-point power supply is its simple and inexpensive design, which is also suitable for use in multi-source power supplies. However, a flyback-type power supply has one drawback that causes problems, especially if the power supply has multiple outputs. This drawback 25 is an unreasonably large output current in the event of a short circuit or similar overload. As the load of the flyback power supply increases so that the control based on primary current detection commonly used in the power supply begins to narrow the width of the pulse controlling the switch, the power-30 source switches to a nearly constant power mode. As the load is increased and the output voltage decreases, the output current increases. Short-circuit currents are often unreasonably large, especially when the power reserved for other outputs is transferred to an overloaded output. Separate 35 current measuring circuits for solving the problem also become unreasonably expensive 2,92892 and require special arrangements, because the control circuit is now usually placed in the primary for cost reasons.

U.S. Patent 4,908,755 discloses a way to limit the output current of a flyback power supply. The adjustment takes place by adjusting the peak value of the primary current as a function of the input and output voltages. In this case, too, the control is performed with a rather complex circuit, in which the parameters have to be dimensioned so that the formal dependence between the input and output voltages and the peak value of the primary current 10 in the flyback power supply can be simulated.

The object of the present invention is to overcome the drawbacks described above with a solution which guarantees the most economical practical implementation. This is achieved by a method and a power supply according to the invention, which method is characterized by what is described in the characterizing part of the appended claim 1 and the power supply by what is described in the characterizing part of the appended claim 5.

The idea according to the invention is to utilize the secondary voltage reflected back to the primary side through the transformer 20 by forming a control signal from this voltage, which controls the control circuit controlling the switch.

Thanks to the solution according to the invention, the cooling and foil of the rectifiers do not have to be oversized, nor does it have to be prepared in the design of the secondary windings of the transformer that unreasonably large currents can flow in each winding. In addition, damage on the load side is less in the event of a short circuit.

In the following, the invention and its preferred embodiments will be described in more detail with reference to the examples of the accompanying drawings, in which Fig. 1 shows a flyback type power supply according to the invention, Fig. 2 is a block diagram of the current limiting circuit shown in Fig. 1, F 3 92892 of the power supply output voltage, Fig. 4 shows a more detailed implementation of the power supply 5 shown in Fig. 1, and Fig. 5 shows the behavior of the output current and voltage in both the prior art and the power supply shown in Fig. 4.

Fig. 1 shows a flyback-10 type power supply according to the invention, which converts the rectified voltage Uin applied to the terminals of the input capacitor Cin to a second direct voltage Uout present at the terminals of the output capacitor Cout. As is known per se, the power supply firstly comprises a transformer 10, through which power is transferred from the primary 15 to the secondary, a switch SW in the primary circuit to cut off the primary current through the primary winding 10a and a switch control circuit 13 which controls the output voltage by adjusting the duty cycle of the switch. The adjustment takes place by means of pulse width modulation (PWM), i.e. by adjusting the relationship between the lengths of the ON and OFF periods of the switch. In the secondary circuit, a rectifier diode D1 and an output capacitor Cout are connected in series in parallel with the secondary winding 10b.

The flyback power supply works as follows. When switch SW 25 is closed (ON state), a positive voltage is generated at the • end of the transformer. The output rectifier diode D1 then has a blocking voltage, so it does not conduct. As a result, the secondary current is zero during the ON state of the switch. On the primary side, the current 30 passing through the switch, on the other hand, increases linearly during the ON state. The transformer stores energy in its magnetic flux (air gap) during this phase, so in fact the transformer is a secondary inductance. When the switch is controlled by a non-conductive state (open, or OFF state), translates the transformer magnetic-35 stored in the tivuohon energy winding voltage 4 92 892 reversed (flyback phenomenon), whereby the secondary side O-rectifying diode D becomes conductive and the transformer toisiokää-in current begins to flow. In contrast to the primary current, the secondary current decreases linearly during the OFF state. The Sa-5 racket secondary current maintains the required output voltage across the output capacitor Cout.

If the load at the output increases, only the length of the ON state of the switch needs to be adjusted longer, as a result of which the primary current has time to increase, so that during the 10 OFF states the secondary current is correspondingly higher. The flyback power supply can operate either in a continuous state (secondary energy does not have time to be completely discharged at the end of the flyback phase) or in a discontinuous state in which the energy is completely discharged during each cycle. There are also 15 flyback power supplies that operate in continuous and uninterrupted image mode, depending on the load. The power supply according to the present invention can be of any type described above.

The speed of the primary winding of the transformer is indicated in Fig. 20 by the reference numeral Np and the speed of the secondary winding by the reference numeral Ns, respectively. The switch SW is shown in the figure only as an ideal element describing its operation, in practice the switch is typically implemented with a MOSFE or bipolar transistor. The control circuit 13 controlling the width of the switch pulse 25 can operate either in the voltage mode based on the output voltage (so-called voltage mode) or in the current mode based on the primary current and the output voltage (so-called current mode). Most (approx. 80%) of the current flyback chippers use current mode circuits (because the current mode 30 control provides better control phase margin than the voltage mode control). Therefore, the control circuit shown in the example of Fig. 1 is a control circuit 13 operating in the current mode, which performs the control on the basis of the voltage information received from the differential amplifier 15 and the current information received from the switch. The voltage information is generated by comparing the output voltage in the differential amplifier with the reference voltage and supplying the difference signal, e.g. via the optoisolator 14, to the differential voltage input EV of the control circuit. The current information is obtained from the switch SW via the resistor Rcs at the current measurement input CS of the control circuit. The information is obtained as the effective voltage across the current measuring resistor R7 (which has a small value compared to the value of the resistor Rcs). The control circuit 13 may be, for example, type UC3843 (or another circuit of the same family) manufactured by Unitrode Corporation, USA. Similar circuits are also available from muil-10 la manufacturers.

The solution according to the invention utilizes the secondary voltage reflected back to the primary side. After all, the fly-back power supply has a voltage Vs across the switch during the OFF state of the magnitude:

Vs = Uin + ““ “(Uni + Uout) (1)

The so-called D1 15 UDI is the voltage across the secondary rectifier diode D1. Since this voltage is small compared to the output voltage Uout, it need not necessarily be taken into account. According to the invention, a separate current-limiting circuit 12 is added to the primary side, to the input A of which this voltage acting over the switch 20 is applied. The output signal (output current) Ice of the current limiting circuit 12 is in turn connected to the current measuring input CS of the control circuit, where the resistor Rcs | a control voltage for the control circuit 13 is formed from the current Ice.

The control circuit has a high input impedance, so no current flows inside it.

The output signal Ice of the current limiting circuit is active only in cases of overload, in which case it limits the output current lout of the power supply, as shown below.

Figure 2 shows the two main blocks of the current limiting circuit 12 according to the invention, which are a peak value rectification circuit 22 and a controllable current generator 21, 196 92892 which is controlled by a peak value rectification circuit. The rectifier circuit 22 receives at its input the above-mentioned voltage Vs. The current generator, in turn, is connected to the positive terminal of the input voltage Uin, and it generates at its output a control current Ice, 5 which is inversely proportional to the output voltage Uout of the power supply. (Since the current generator is connected to the positive terminal of the input voltage Uin, it is affected by a voltage Ug corresponding to the latter part of formula (1) independent of the input voltage Uin.) Figure 3 shows the control current Ice output as a function of the output voltage Uout.

When the output voltage has dropped from its nominal value Uoutl to a predetermined value kl1Uoutl, the current generator 21 starts operating. If the nominal value of the output voltage 15 Uoutl of the power supply is e.g. 5 V, this starting point could correspond to e.g.

80% of the rated voltage (kl = 0.8). Thus, the operation of the current generator starts when the output voltage drops to 4 V. In addition, the current generator is dimensioned so that its maximum current (corresponding to a complete short circuit 20, i.e. Uout = 0 V) cannot completely close the control circuit 13. Namely, the control circuit 13 has a threshold value which completely closes the control, whereby no power is output from the power supply. In this presentation, a typical control circuit threshold value of 1 V is used as an example, which corresponds to a current of Ith = lmA when the value of the resistor Rcs is 1 kn (the value of the resistor R7

* I

* the value is very small, eg 1Ω, so it has no effect). The maximum value of the control current Ice is thus some predetermined proportion, e.g. about 75% (k2 = 0.75) of said current limit value Ith, which completely closes the control circuit 13.

Figure 4 shows a more detailed embodiment of the power supply shown in Figures 1 and 2. For the sake of simplicity, only the implementation of the primary circuit is shown in Figure 4, since the secondary circuit corresponds in this case to the structure shown in Figure 1. In addition, the 2 2 7 92892 feedback loop formed by the differential amplifier and the optoisolator is not shown. A zener diode Z1 and a resistor R3 are connected in series between the input terminals (the input capacitor Cin is not shown in Fig. 4). From one of these 5 terminals, a resistor R2 is connected to the base of the pnp transistor Tri at a point P1. The emitter of the transistor is connected via a resistor Rg to the positive terminal of the input voltage Uin. The collector of the transistor is connected via a resistor R4 to a common point of the current measuring input CS of the control circuit 13 and the resistor Rcs 10. The base of the transistor is further connected via a resistor R1 to a point P of the rectifier circuit 22, which is connected via a capacitor C1 to the negative terminal of the input voltage. In addition, a common terminal of the primary winding and the switch SW is connected to the point P via a series connection of the resistor R5 and the directional diode D2. Resistor R5, diode D5 and capacitor C1 form a peak rectifier circuit 22, and point P thus forms a supply point from which the voltage of formula (1) above is supplied to a current generator 21 formed by ze-20 nerdiode Z1, resistors R1-R4 and Rg, and transistors Tri. As can be seen from formula (1), this voltage applied to point P decreases as the output voltage Uout decreases (short-circuit case).

In other respects, the primary circuit corresponds to the structure already shown in Fig. 1, i.e. the current measuring input CS of the control circuit 13 is connected via a resistor Rcs to the second terminal of the switch, which is connected via a resistor R7 to the negative terminal Uin.

In order for the current generator 21 to operate as described above 30, the resistance values of the circuit must be dimensioned correctly. The following exemplary values are used in the following as in Fig. 3, and it is further assumed that the number of turns of the primary winding of the transformer is 13, the number of turns of the secondary winding is 3, and that the resistor R3 biases the zener diode Z13 so that the voltage across the zener diode is 6.2 V.

92892 δ

In the equilibrium situation (i.e., in a situation where the output voltage Uout has dropped to the limit (4 V) at which the current generator begins to operate), the transistor Tri is just conducting, resulting in a base-emitter voltage of approximately 0 V 5 (0-0.2 V). The current through resistor Rg is still zero, so the voltage across resistor R2 must match the voltage across the zener diode. Assuming that the value of resistor R2 is e.g. 56 kΩ, the current 12 through resistor R2 is about 110 μΑ. In the equilibrium situation, the voltage at the base of the transistor 10 Tri (at the point P1) «Uin, and the base current of the transistor is zero, so that the current 12 can only be obtained through the resistor R1. Since the voltage across the resistor R1 is <17.3 V (13/3 * 4 V), the value of the resistor R1 is thus obtained

Rl «150 kn.

When the output voltage Uout is greater than 4 V, the current flowing through the resistor R1 is correspondingly higher, and the base emitter voltage keeps the transistor in the closed state. When the output voltage drops to 4 V, the control current Ice starts to flow, and the lower the output voltage drops, the smaller the cancellation effect obtained through the resistor R1, respectively, whereby the control current Ice is correspondingly higher.

In the second extreme situation, there is a complete short circuit at the output of the power supply (Uout = 0 V), whereby the voltage VP «Uin at point P (assuming that the diode D1 is ideal, i.e. the volume of the voltage 25 above it is zero). In this case (assuming that the gain of the transistor is e.g. 30) the value of the resistor Rg is obtained

Rg «5.7 kn, when a maximum value of about 750 μΑ is required for the current Ice (the voltage across the resistor Rg is 6.2 V minus the base emitter voltage of the transistor of approximately 30 0.5 V and the voltage across the resistor R2 which is approx. 1.4 V).

The value of resistor R3 must be dimensioned in such a way that the current flowing through resistor R2 cannot interfere with the bias of zener diode Z1.

35 By changing the ratio of resistors R1 and R2, the 9 92892 current generator limit threshold (Uout = 4V) can be changed differently. On the other hand, by changing the value of the resistor Rg, various limitation curves can be obtained, which are described below.

Fig. 5 shows the output current lout of the power supply shown in Fig. 4 as a function of the output voltage Uout. The Ba-spot point is denoted by the reference symbol B. The normal operating range is that in which the output voltage Uout remains at its nominal value Uoutl (e.g. 5 V). The corner point L corresponds to the threshold of the control circuit 10 13 at which the primary current detection of the control circuit starts to narrow the pulse width, and the power supply switches to a nearly constant power mode, where the dependence of the output voltage and output current is represented by curve D. However, according to the invention a balance point 15B is provided. using the control current Ice provided by the generator 21. When the output voltage Uout decreases to the equilibrium point, the output current is thus limited even more effectively, whereby the dependence of the output voltage and the output current is described, for example, by one of the 20 lines F1-F5. The direction in which this constraint curve travels depends on the value of the resistor Rg. The usable area is indicated in the figure by the arrow H. If the value of the resistor Rg is increased to be greater than the value corresponding to the line F5, the area H is moved towards the curve D, whereby the solution according to the invention is no longer very useful. On the other hand, if the value of the resistor Rg is reduced to less than the value corresponding to the line F1, the power supply remains in the lock. As stated above, by changing the ratio of resistors R1 and R2, the position of the equilibrium point B on curve D can be changed.

Although the invention has been described above with reference to the examples according to the accompanying drawings, it is clear that the invention is not limited thereto, but can be modified within the scope of the inventive idea set forth above and in the appended claims. For example, practical power sources may have multiple outputs, although the above example structures have only one output. The more detailed implementation of the power generator can also vary in many ways. However, the structure 5 described above makes it possible to add the additional features according to the invention to known flyback-type switch power supplies as economically as possible. In principle, it is also possible to use the solution according to the invention in connection with a control circuit operating in voltage mode, although the invention has been described above only in connection with a circuit operating in current mode. However, if a voltage-mode circuit is to be used, a control signal suitable for a voltage-mode circuit (e.g., UC 3524, manufactured by Unitrode Corporation, USA) must be formed from the control current Ice.

In this case, the solution becomes more complex than described above, and at the same time the advantages which the control circuit operating in the current mode has over the control circuit operating in the voltage mode are lost.

Claims (7)

A method for limiting the outflow (lout) from a flyback-type chopper current source in overload situations, according to which the outflow (lout) method is limited by pulse-width modulation (PWM) by known control circuit (13) regulate the relationship between the lengths of ON and OFF periods in a primary circuit breaker (SW), characterized in that a primary side of a transformer (10) visible, dependent on the output voltage (Uout) dependent voltage ( Vs) is used to control a current generator (21), from whose output current (Ice), a control signal is generated for the control circuit (13). 15 2. A method according to claim 1, characterized in that from the current generator (21) is given a zero-zero deviation and against the output source voltage (Uout) inversely proportional current (Ice), where the output voltage is lowered to a predetermined part (kl). of its nominal value (Uoutl). 3. A method according to claim 1, characterized in that the maximum current from the current generator (21) is poured less than the threshold current (Ith) which completely closes the control circuit (13). 25 A method according to claim 3, wherein the control circuit (13) operates in a current state, characterized in that the output current (21) of the current generator (21) is initiated in the current measurement input (13) of the control circuit, in a manner known per se. a signal proportional to the primary current of the transformer (10) is also obtained. A flyback type chopper power source comprising a transformer (10a, 10b) provided with primary and secondary windings (10) through which the power is transmitted from the primary to the secondary, a switch (SW) located in the primary circuit, through which primary winding (10a) the primary primary current is interrupted and a control circuit (13) which controls the switch and regulates the output voltage (Uout) of the power source by means of pulse-width modulation by controlling the relationship between the lengths of 0N-5 and OFF periods in the switch (SW), characterized in that it comprises means (21, 23) for generating a single control signal (Icc) in response to a visible side of the transformer, which is dependent on the output voltage (Uout) voltage (Vs) ), which means 10 connect said control signal to the control circuit (13). Power source according to claim 5, whose control circuit (13) is a current supply circuit, characterized in that said means comprises a peak value rectifier circuit (22), to which said voltage (Vs) is coupled, and a a current generator (21), to which the output of the rectifier circuit is coupled, which output of the generator is connected to the current measurement input (CS) of the control circuit (13). 7. Power source according to claim 6, characterized in that the current generator (21) is coupled to the positive terminal of the input voltage (Uin). t «t