FI70104C - ADJUSTMENT OF THE LINE - Google Patents

Description

[755 ^ 1 m miKUU, LUTUSJULKA, su 70104 (11) UTLÄGGNINGSSKRIFT 'C (45) F · titi titi (51) Kv.lk. * / Lnt.CI.4 H 04 N 3/16 FINLAND - FINLAND (21 ) Patent application - Patentansökning 800673 (22) Filing date - Ansökningsdag 05.03 .8 0 (Fl) (23) Starting date - Giltighetsdag 05 · 03 .80 (41) Published public - Blivit offentlig 1 3-09.80

National Board of Patents and Registration ^ Date of filing and monthly publication— 31 01 .86

Patent and registration authorities '' Ansökan utlagd och utl.skriften publicerad (32) (33) (31) Privilege claimed - Begärd priority 12.03.79 USA (US) 019581 (71) RCA Corporation, 30 Rockefeller Plaza, New York, New York 10022, USA (72) Wolfgang Friedrich Wilhelm Dietz, New Hope, Pennsylvania, USA (74) Oy Koister Ab (54) Television Line Deflection Device - Reglerad 1injeavlänkningsapparat This invention relates to a guided line deflection device adapted to be fed from an uncontrolled DC voltage source and comprising: a) a primary switching means comprising a half-conductor switch having a controlled current path and a control electrode which can be made non-conductive by commutation; b) a transformer comprising a primary winding connected to the controlled current path and the uncontrolled DC voltage source , and a secondary winding electrically isolated from the primary winding, (c) a deflection winding and a charge capacitor both are connected to a secondary winding, d) a deflection circuit isolated from the source and comprising a line switch connected to the deflection winding and used at a line frequency to enhance the deflection current in the deflection windings during repeated line and return cycles, the voltage across the deflection winding e) a control circuit to which a detection signal from the detection circuit representing the energy circulating in the deflection device is applied and which provides a detection signal to the semiconductor switch control electrode as a semiconductor to determine its time, available during the start-up period immediately after switching on the line deflection device.

Horizontal offset circuits are used in conjunction with television picture tubes in television display devices. Typically, the horizontal deflection circuits include a magnetic winding associated with the picture tube and a switching circuit that connects energy from the DC voltage source to this winding and associated reactances. The switching circuit is synchronized by means of synchronization marks which are related to the data portion of the video to be displayed on the picture tube. In order to avoid distorted images in the raster shown, the size of the horizontal sweep line and the peak deviation, i.e., the sweep current, must be kept constant over the relevant time periods.

Many conditions can cause horizontal sweep line size variation. If the DC current voltage for the horizontal deflection circuit varies, the energy of the sweep and thus the line width of the horizontal sweep must also vary. In the past, it has been conventional to control the DC voltage supplied to the horizontal deflection circuit using a power-dissipating regulator. The requirements for low power consumption in television receivers reduce the use of such power consuming controls in favor of non-power consuming, i.e., switching types.

Among the switching controllers used in television deflection circuits are reverse current controllers, thus called because the energy in excess of the maximum amount required in the deflection circuit flows back from the controller to the power supply during a portion of the deflection interval. Such a regulator is described in U.S. Patent No. 4,013,923, issued March 22, 1977 to den Hollander. When this is used with an SCR deflection system, as described in this den Hollander patent, the adjustment takes place during the time period in which it is open, i.e. due to 3 ^ 0104 unattended. This time interval is approximately 38 μs in duration, which represents more than half of the 63 ^ μs horizontal deflection time interval. The principle of reverse current control is not conveniently applicable to use in a transistor horizontal deflection system because only a relatively short reset time is available for control operation and also because the Port Control of the reset current switch changes the impedance of the input source and adversely affects reset time.

Direct current controllers, as illustrated in U.S. Patent No. 4,002,965, issued January 11, 1977, by Den Hollander, may provide conductivity isolation when used in conjunction with a transistor deflection circuit. However, such arrangements require either high voltage control switches or diodes to prevent excessive voltages in the blocking direction from being applied to this control switch during the reset interval. Likewise, the variation in gating time required to maintain a constant output voltage under conditions of mains current and load variations is relatively large, thus limiting the correction range.

A directional current regulator having a bidirectional control switch, as described in U.S. Patent No. 4,071,810, issued January 31, 1978, to Dobbert, and when used in conjunction with an SCR deflection circuit, provides galvanic or current conduction isolation to a power supply source and deflection circuit. and has a limited blocking voltage across the control switch, but requires a separate regulated resonant circuit to allow the control switch to become non-conductive.

It is desirable to have a controller that can be applied to a transistor deflection circuit that provides a conductive isolation situation from the power grid that does not have any significant voltage across the control switch during the reset interval, has a wide control range and is switched off and has relatively low losses and consequent losses. which does not require any independent tripping circuit or where the transient phenomena of the tripping are small if a tripping circuit is used.

The line deflection device according to the invention is characterized by a starting circuit comprising a rectifier connected between the auxiliary energy source and the charging capacitor and biased during the start-up period to supply energy to the deflection circuit and then biased to separate the auxiliary energy source from the deflection circuit.

The accompanying drawings show:

Figures 1 and 3 illustrate in block diagram and schematic form the proportion of controlled deflection in a television receiver incorporating the present invention.

Figures 2 and 4 are diagrams between amplitude and time, i.e. waveforms of electrical quantities, related to the arrangements of Figures 1 and 3, respectively.

In the upper left of Fig. 1, the DC power supply, generally indicated by reference numeral 10, has switch terminals 12 and 14, which are suitable for connection to a power supply or other AC voltage supply source, not shown. The full wave rectifier 16 is connected to the switch bases 12 and 14. The filter capacitor 18 is connected to the bridge rectifier 16 in a known manner to filter the pulsating DC current to provide an unregulated DC voltage. This DC voltage is applied to a control circuit, generally indicated by reference numeral 20. The control circuit 20 includes, in series, the inductance of the winding 22, the primary winding 24a of the transformer 24, and a bidirectional control switch 26. This series is connected across the capacitor 18. The switch 26 includes a controlled silicon rectifier (SCR) or thyristor 28 connected in parallel with the diode 30.

A high AC voltage is provided to the secondary winding 24b of the transformer 24. This high voltage is rectified by a rectifier as represented by diode 32 and this high voltage is connected from the cathode of diode 32 to the last electrode of the picture tube, which is not shown. Yet another secondary winding 24c from the transformer 24 is connected to the reference voltage generating circuit 33. Current flows from the winding 24c through the diode 34 and the resistor 36 to a resistor-capacitor filter, generally indicated at 38, to provide a direct voltage proportional to the energy rotate. This DC voltage is used as a reference in the controller.

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The horizontal deflection circuit, generally indicated at 40, is connected to one end of the secondary winding 24d in the transformer 24. The other end of the winding 24d is connected to ground through a filter capacitor 41. The horizontal deflection circuit 40 includes a drawing switch 42 formed by an NPN horizontal output transistor 44 having an emitter connected to ground and an attenuator diode 46 from which its cathode is connected to the collector of transistor 42 and from which an anode is connected to ground. The return capacitor 48 and the series connection of the deflection winding 50 and the S-shaped capacitor 52 are connected in parallel with the switch 42. The deflection circuit 40 is powered by power, which is supplied through a transformer 24 and stored in a capacitor 41.

An input source for horizontal frequency synchronization signals, such as an oscillator 60 from the bottom of Figure 1, is connected to the base of the NPN control transistor 62, from which the emitter is connected to ground. The collector 64 of the transistor 62 is connected to the base 43 of the transistor 44 by a transformer 66. The collector 64 of the control transistor 62 also controls the controller control circuit, generally indicated at 68. The control circuit 68 includes a bypass controller 69 consisting of a series connection of a resistor 70 and a zener diode 72 connected between the transistor 64 and the ground. The charging resistor 74 connects the charging capacitor 80 to one end of the output of the bypass controller 69, from where the other end is connected to ground. From the point of view of the transistor 76, the connection to the emitter is connected in series with the stabilizing resistor 78 over the capacitor 80.

The collector 77 of the transistor 76 is connected to one end of the primary winding 82a in the isolation transformer 82 and the other end of the winding 82a is connected to the collector 64. The attenuator resistor 84 is connected across the winding 82a. The secondary winding 82b of the transformer 82 is connected across the gate and cathode terminals of the SCR rectifier 28. The control circuit is made to operate on the basis of the energy circulating in the controlled deflection system by means of a series connection of the zener diode 86 and the resistor 88, which is connected between the transistor 76 and the base and the filter 38. The current for the control transistor 62 and the control circuit 68 is provided by a power supply including a transformer 90, the primary winding of which is connected to terminals 12 and 14 and the secondary side of which is connected to a rectifier frequency diode 92 and a filter including capacitors 93 and 94 and a resistor 95. A starting circuit 96 including a resistor 97 and a diode 98 are connected between the capacitor 93 and the capacitor 41 to provide a horizontal deflection of the operating voltage for the circuit 40 when it is initially turned on.

In general, the arrangement of Figure 1 brings energy from the power supply 10 to the deflection circuit 40 through the transformer 24, thus providing a level of conductivity isolation between the power grid and its equipment. The horizontal deflection circuit 40 is controlled at the horizontal frequency and the SCR rectifier 88 at the gate is controlled by pulse time modulated signals at the same frequency to control the amount of deflection and / or the voltage of the last electrode. The SCR rectifier 28 is commutated off by a horizontal reset pulse, which is reconnected from the deflection circuit 40 via a transformer 24. After the initial power-up period, the starting circuit 96 is disconnected by the diode 98 from the deflection circuit 40 as the DC voltage across the capacitor 41 increases as a result of the rectification of the alternating current provided by the winding 24d in the drawing switch 42.

During operation immediately before time T2, as shown in Figure 2, transistor 62 is conductive, horizontal output transistor 44 is non-conductive, and attenuator diode 46 conducts a decreasing current deflection through winding 50 and capacitor 52 as winding 50 supplies energy to capacitor 52. Control switch 30 also conducts d according to Fig. 2j, when the energy stored in the current form in the inductance 22 is transferred to the capacitor 18 and via the transformer 24 to the capacitor 41. When the diodes 30 and 46 are conductive, the voltages across the control switch 26 and the drawing switch 42 are substantially zero, such as is shown at V26 in Figure 2k and at V42 in Figure 2d, respectively. The SCR rectifier 28 and the transistor 44 receive a slightly blocking forward voltage and are therefore non-conductive. Transistor 76 lacks collector bias voltage and is non-conductive and capacitor 80 has a voltage across it less than the emitter blocking voltage VB76 for transistor 76 and discharges as illustrated in V80 in Figure 2f through resistor 88 and zener diode 86.

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At time T2, the control voltage V60 provided by the oscillator 60 shifts to a negative transition point, as illustrated in Fig. 2a, and the transistor 62 is consequently made non-conductive. The energy stored in the leakage inductance of the transformer 66 provides a positive pulse voltage to the collector 64, as illustrated at V64 in Figure 2b, and also to the base 43 of the transistor 44, resulting in a directional base current, as illustrated by 143 in Figure 2c. Transistor 44 is thus made conductive before time T6, at which time the current 150 in the deflection winding 50 reaches zero, as illustrated in Figure 2e. Between T2 and T10, a positive voltage V64 at the collector of transistor 62 provides a positive voltage across zener diode 62, as illustrated in V72 in Figure 2b. A positive voltage across the zener diode 72 allows current to flow through resistor 74, which directs the discharge current through resistor 88 and provides a positive voltage rise across capacitor 80, as illustrated in V80 in Figure 2f. However, the transistor 76 remains non-conductive after the time T2 until a certain later time T8, when the increasing rise voltage V80 reaches the value of the blocking voltage VB76 for the emitter for the transistor 76.

Switching on the transistor 62 at time T2 does not affect the energy transfer from the inductance 22 to the supply 10 and the capacitor 41. The energy is still continuously transferred in this way until a certain later time T4. At time T4, the energy associated with the magnetic field of the inductance 22 is depleted and the current ceases to flow along the circular path through the inductance 22, the diode 30 and the winding 24a. Diode 30 becomes non-conductive and control switch 26 opens. When no current is flowing in the inductance 22 or in the winding 24a, the voltage across the control switch 26 rises equal to the unregulated voltage, which then occurs over the capacitor 18, as illustrated in V26 in the interval after time T4 in Fig. 2k. The SCR rectifier 28 remains non-conductive even though it receives a forward bias voltage until a certain later time T8, at which time the gating pulse once again conducts it. No energy is transferred between the inductance 22 and the deflection winding 40 during the time interval T4-T8, when the control switch 26 is open.

70104

Between T4 and T6, the current that travels in a circular path including deflection winding 50, capacitor 52, and attenuator diode 46 decreases toward zero, and at time T6, the current reaches zero and diode 46 becomes non-conductive. Transistor 44 becomes conductive before time T6, the current reaches zero, and diode 46 becomes non-conductive. Transistor 44 becomes conductive before time T6, as mentioned, and current begins to flow at time T6 through winding 50 and transistor 44 under voltage across capacitor 52, as illustrated at 150 in Figure 2e. However, current does not flow in the time interval T4-T8 from the capacitor 41 through the winding 24d to ground through the transistor 43, because the winding 24d is connected to the winding 24a, which is open in its circuit under the action of the open switch 26.

Between T6 and T8, the rise voltage V80 continues to increase at a rate determined by the zener diode 72, resistor 74, resistor 88, zener diode 86, and the voltage stored in filter 38. The SCR rectifier 28 remains non-conductive even though it receives a forward bias voltage. At time T8, the rise voltage V80 becomes equal to the side-shift voltage at the emitter in transistor 76 and transistor 76 becomes conductive. When transistor 76 is conductive, the voltage V77 on the collector of transistor 76 shifts to negative as a transition point, as illustrated in Figure 2g, and a control pulse is applied to SCR rectifier 28, as illustrated in V82b in Figure 2h. At time T8, the SCR 28 is turned on and the control switch 26 is closed. The voltage across the control switch 26 decreases toward zero, as illustrated in V26 in Figure 2k. When the switch 26 is closed in the time interval T8-T12, a complete series-connected circuit is formed between the supply capacitor 18, the inductor 22 and the transformer winding 24a. The effective impedance over the secondary winding 24d decreases and the conventional current flows upward through the winding 24d due to the voltage across the capacitor 42. Due to the polarity of the transformer 24, this results in a voltage across the primary winding 24a, which increases the voltage applied across the inductance 22, i.e., the voltage across the winding 24a assists an unregulated DC voltage, enhancing current flow in the inductance 22. Thus T8-T12 winding 22.

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Between T8 and T10, the output voltage V60 of the oscillator 60 remains low and the transistor 62 remains non-conductive. The collector voltage V64 of the transistor 62 remains high, as does the voltage V72 across the zener diode 72. Rising voltage V80 and rising current 150 continue to increase. At time T10, oscillator 60 provides a positive transition point and transistor 62 becomes conductive. This provides a negative transition point to the voltage applied to the base 43 of the output transistor 44 and the removal of charge conveyors from the base area of the transistor 44 begins, as illustrated at 143 in Figure 2c.

The negative transition point of the voltage V64 at the collector of the transistor 62 at time T10 also removes control from the zener diode 72 and clears the supply voltage available at the collector 77 of the transistor 76. The voltage V77 is therefore maintained at zero by removing the collector control regardless of the base voltage. Capacitor 80 continues to discharge through resistor 88 and zener diode 86 after time T10, but is discharged through resistors 70 and 74 to a low voltage at collector 64 instead of being charged therethrough. Thus, the voltage V80 becomes a rising component with a negative slope. At time T16, which in the illustration case occurs during the reset interval, but which may be during the drawing interval, the decreasing rise voltage V80 becomes equal to and later less than the emitter side off-voltage VB76 for transistor 76. However, no shift occurs in the collector V77 of transistor 76. because the supply voltage V64 for transistor 77 is currently zero. Thus, V77 cannot implement the transition until V64 becomes positive again.

In the time interval T10-T12, just before the start of the reset interval, the charge conveyors continue to exit from point 43, as illustrated 143, and the transistor 44 remains conductive, as indicated by the low voltage V44 across it in Figure 2d. The collector current of transistor 44 continues to increase as the deflection current 150 increases. The current continues to increase in inductance 22 as energy is transferred there from the unregulated supply source and capacitor 41 and the SCR 28 of the control switch 26 remains closed. At time T12, the charge conveyors in the base area of transistor 44 are depleted and the base current is reduced to zero, as illustrated 143 although the forward bias voltage applied to the base still exists. Transistor 44 becomes non-conductive and the currents that then flow in the deflection winding 50 and winding 24d begin to flow to the return capacitor 48, starting the return interval. The voltage across capacitor 48 increases until it is equal to the voltage across capacitor 41 and continues to rise. When the voltage across the capacitor 48 exceeds the voltage across the capacitor 41, the voltage across the secondary winding 24d changes in polarity compared to the time interval of the drawing. This alternating voltage appears over the pri-winding 24a and begins to resist the flow of current in the inductance 22 instead of helping it. The energy associated with the magnetic field of the inductance 22 forcing the current to continue to flow up to the opposite voltage provided to the primary winding 24a. Thus, shortly after the time T12, when the recovery time interval begins, the energy stored in the winding 22 begins to transfer to the recovery capacitor 48 via the secondary winding 24d.

During the first half T12-T14 from the recovery interval, the deflection winding 50 and the inductance 22 transfer substantially all of their energy to the capacitor 48 as the currents 150 and 126 decrease toward zero. At time T14, capacitor 48 reaches its maximum return voltage, as illustrated at V48 in Figure 2d, and the current in winding 50 and inductance 22 reaches zero. When the current in switch 26 reaches zero, SCR 28 becomes non-conductive. Immediately after time T14, the current in the deflection winding 50 and also in the inductance 22 and the switch 26 starts to increase in the opposite direction. The current in the switch 26 passes through the diode 30, thus generating a slightly forward bias voltage to the SCR rectifier 28. As the current increases, the return voltage across the transistor 44 and the return capacitor 48 decreases.

During the second half time interval, which extends from the time T14 to the time T18, energy is transferred from the capacitor 48 to the capacitor 42 by the current flowing through the winding 24d.

During this same time, energy is supplied from the winding 24d to the winding 24a and transferred to the inductance 22 as well as the unregulated

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11 7 010 4 just before the time T18, the voltage across the reset capacitor 48 becomes equal and decreases below the voltage then present in the capacitor 41, after which the voltage across the winding 24d and consequently across the winding 24a becomes opposite to that. existed during the principal portion of the recovery period. Thus, near the time T18 at the end of the recovery period, the energy transfer from the deflection circuit 40 to the inductance 22 ceases and the energy again begins to shift from the inductance 22 back to the deflection circuit 40, as well as to the unregulated supply source 10.

At time T18, the voltage across capacitor 48 decreases slightly to a negative value, diode 46 becomes conductive, and the reset interval ends. The decreasing current forming the current gain continues to flow through the deflection winding 50 and the attenuator diode 46 and the inductance 22 continues to transfer energy to the unregulated supply 10 and the transformer 24 to the capacitor 42 and the cycle starts again.

As described, the arrangement of Figure 1 transfers energy to a horizontal deflection circuit via a current path that includes a transformer 24 that is bidirectional for both current and instantaneous power. The instantaneous direction of power flow during the respective portions of each cycle is illustrated in Figure 2 using arrows. The bidirectional switch 26 is commutated to non-conductive by reducing the voltage to more than zero, and since the switch is bidirectional, the maximum voltage acting over it during the reset interval does not exceed the value of the unregulated B + voltage. Since the switch 26 is commutated, there are no switching losses that occur in the controllers during the switch-on intervals, and the efficiency is therefore high.

It has now been found that a rapid rise in the unregulated B + voltage in the arrangement of Figure 1 may result in a situation where the control switch 26 does not turn off during the reset interval. It is believed that this is due to an increase in the amount of energy stored in the inductance 22 without a corresponding increase in the energy of the return pulse. Figure 3 illustrates an arrangement using a transistor for a control switch. In the arrangement of Fig. 3, the control switch is switched off in normal operation, thus providing all the advantages described in connection with the arrangement 12 701 04 of Fig. 1. In the event that the control switch does not turn off by the reset pulse during the reset interval, the transistor is turned off by the control circuit, thus providing a backup control.

In Fig. 3, those parts corresponding to those in Fig. 1 are provided with the same reference numerals using a three-hundred-digit set. In Figure 3, alternating current applied to the input switch terminals 312 and 314 and rectified by a bridge rectifier 316 provides an unregulated DC voltage across the filter capacitor 318. The control switch 326 is connected in series with the inductance 322 and the primary winding 324a of the transformer 324. The control switch 326 includes an NPN transistor 328, from which its collector is connected to the positive switch terminal of an unregulated supply source. From diode 330, its cathode is connected to the collector of transistor 328 and its anode is connected to the emitter. The equalizing circuit 327 is connected in parallel with the diode 330.

The last electrode winding 324b and rectifier 322 provide the last electrode voltage for the picture tube, which is not shown. The deflection circuit 340 closely corresponds to the circuit 40 of Figure 1, as does the reference voltage supply circuit 333, which provides a direct voltage representing the energy circulating in the system. However, the control circuit 368 is arranged to account for the different control needs of the transistor 328 compared to the SCR rectifier 28. The synchronized signal input as illustrated block 408 is connected to the control transistor 366 which is generally indicated by reference numeral 413. Circuit 413 includes an NPN transistor 416 connected as a common emitter amplifier, from which its base is connected to a diode 412. The collector of transistor 416 is connected to B + voltage by resistor 418, ground resistor 420 and NPN transistor 426 base by capacitor 422. B + voltage adjustable resistor 424. Transistor 426 is connected to a common emitter or inverting structure and its collector 427 is connected to B + voltage by a load resistor 428 and back to the base of transistor 416 by a feedback resistor 414. A timing circuit 413 against , which is delayed during adjustment of resistor 424 to provide a turn-off time for control transistor 328.

The second timing circuit 429 includes a capacitor 430 in series with a resistor 432 adjustable from point 431 between the B + point and the output point 427 of the timing circuit 413. For transistor 434, the connection to the emitter is connected between circuit point 431 and ground. The collector of transistor 434 is connected to B + voltage through resistor 436. The Darlington transistor circuit 438 including the NPN transistors 440 and 442 is emitter-equivalent connected to ground and base-equivalent connected to the collector of transistor 434. The other end of the primary winding 382a in the isolation transformer 382 is connected to the collector of the Darlington circuit 438. The other end of the winding 382a is connected to the B + voltage by an insulating resistor 448 and is also connected to ground by a capacitor 450. A damping circuit including a resistor 444 and a diode 446 are connected across the winding 382a. The secondary winding of the transformer 382 is connected to the emitter across the junction of the control transistor 328. The control circuit 368 is made to operate based on the amount of total energy circulating in the deflection system using a zener diode 386 and a resistor 388 connected in series between the circuit point 431 and the reference voltage generating circuit 333.

During operation, the delay circuit 413 is adjusted so as to initiate the offset portion of the control pulse at the base of transistor 328 during the second half reset time interval. Thus, if the reset pulse does not commutate switch 326 as non-conductive, as may occur due to a rapid increase in unregulated supply voltage, the control circuit turns off transistor 328, ensuring operation without excessive energy storage. The timing circuit 429 is adjusted by a resistor 432 to provide the correct amount of circulating energy by adjusting the relative turn-on time in the control transistor 328.

Immediately before time T1, as illustrated in Figure 4, the output of the synchronization signal input source 408 is in the low state and the output transistor 344 receives a forward bias voltage and conducts a decreasing current along a circular path 70104 through a capacitor 352 and a winding 350. Likewise, immediately before time T1, the control switch 326 is conductive and the incremental current flows through the inductance 322 and through the transformer winding 324a under the control of an unregulated DC voltage, as illustrated at 1322 in Figure 4d. Transistors 426 and 434 are conductive and transistors 416 and 434 are non-conductive. Circuit point 423 is slightly positive with respect to ground for transistor 426 to the emitter due to the side-setting voltage, and capacitor 422 is charged to the polarity shown.

At time T1, a positive transition point is formed at the output of the input source 408, as illustrated at V408 in Figure 4a. The transition initiates the removal of charge conveyors from the base area of the horizontal output transistor 344, as illustrated in 1343 in Figure 4b. The transition point also turns on transistor 416, providing a negative transition voltage to its collector. This transition is connected to circuit point 423, as illustrated at V423 in Figure 4f. Transistor 427 becomes non-conductive and its collector voltage rises, as illustrated in V427 in Figure 4e. The positive transition point is reconnected to the base of transistor 416 via feedback resistor 414 to keep transistor 416 conductive.

Between T1 and T3, the charge conveyors are further removed from the base area of transistor 344, as illustrated in 1343 in Figure 2b. At the same time, the voltage T1-T3 at the circuit point 423 rises, forming a rising portion as the capacitor 422 discharges. At time T3, the charge conveyors from the base area of transistor 344 are exhausted and the base current drops to zero. Transistor 344 becomes non-conductive, initiating a recovery period. The current flowing in the deflection winding 350 and the winding 324d resonantly flows to the return capacitor 348, providing a high return voltage for the time interval T3-T11, as illustrated in V348 in Figure 2c.

During the first half T3-T5, the recovery interval T3-T11 causes the high recovery voltage in the current deflection winding to decrease towards zero and also reduces the current 1322

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15 701 04 per zero. In normal operation, the current 1322 at inductance 322 reaches a point near zero at time T5 in the center of the reset.

This also reduces the current in the transistor 328 of the control switch to zero, regardless of whether the connection to the emitter receives a forward bias voltage.

During the latter half T5-T11 from the reset interval, the reset voltage causes an increasing current to flow through the deflection winding 350 when current is stored there and also causes an increasing current to flow through the inductance 322 and diode 330 when energy is stored from the deflection circuit 340 to the inductance 322 and supply 310.

At time T7 in the illustrating case, during the last half T5-T11 of the reset interval, but which may be adjustable as needed at any suitable time, the rise voltage V423 becomes equal to the line-to-emitter voltage in transistor 426. Transistor 426 becomes conductive again and its collector voltage V427 is illustrated in Figure 4e and keeps transistor 416 inoperative. The negative transition point V427 is also introduced through capacitor 430 to circuit point 431 at the base of transistor 434, rendering it non-conductive. The collector voltage of transistor 434 rises, as illustrated in VC434 in Figure 4h. Darlington transistor 438 becomes conductive and its collector voltage drops rapidly, as illustrated in VC438 in Figure 2i. A change in this voltage is applied to the base of transistor 328 so as to provide a forward bias voltage.

In the subsequent time interval, the T7-T9 charge conveyors are removed from the base area of the transistor 328, but the transistor 328 is still conductive. A rise voltage V431 is provided to circuit point 431 as the capacitor 430 is discharged through the resistor 432 and under the influence of the reference voltage generator 333.

At time T9, all charge conveyors have been removed from the base area of transistor 328 and will no longer be able to conduct until it again receives the line forward voltage. Since the conductivity of the switch 326 in normal operation occurs through the diode 330 after the time T5 in the center of the reset time interval, disabling the transistor 328 from the time T9 16 701 04 does not cause any difference in the operation of this circuit. Between the time T9, when the transistor 328 is rendered non-conductive, and between the time T11 at the end of the reset interval, the reset voltage continues to decrease. At a certain point in time before T11, the return voltage becomes equal to the voltage across the stealth capacitor 341 and the voltage applied across the winding 324d changes to zero and then reverses, as described in connection with Figure 1, resulting in a change in the direction of energy flow.

At time T11, the reset time slot ends and the drawing time slot begins when the attenuator diode 346 begins to conduct a decreasing rise current through the deflection winding 350. The diode 330 of the control switch 326 also conducts a decreasing current through the winding 322 in the time interval T11-T13 as the inductance 322 transfers its energy to the unregulated supply source 310 and the capacitor 341. At time T13 the current of the inductance 322 reaches zero, open until the next time interval when the base control is applied to the transistor 328. When the switch 326 is open, the voltage V322 and 324a over the series connection of the inductance 322 and the winding 324a decreases to zero, as illustrated in Fig. 4k. As illustrated, the waveform in Fig. 4k shows transient portions in the interval immediately after the switch-off time T13.

At time T15 before the center of the drawing time slot, the synchronization signal generator 408 provides a negative transition point, which is coupled through the transformer 366 to the base of the transistor 344 so that the transistor receives a forward bias voltage and is made conductive. Near time T15, switch 326 remains non-conductive and the rise voltage V431 continues to develop at circuit point 431 as capacitor 430 is discharged. At time T17, the increasing rise voltage V431 at the circuit point 431 becomes equal to the voltage to the emitter for the transistor 434. Transistor 434 becomes conductive again, its collector voltage VC434 decreases and makes transistor 438 non-conductive. The collector voltage VC428 of transistor 438 performs a positive oscillation, which provides a forward bias to transistor 326.

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Current begins to flow through transistor 328 and inductance 322 after time T17. Thus, the period in which no energy is transferred between the regulated supply source 310, the inductance 322 and the deflection circuit 340 ends at time T17. The moment at which time T17 occurs during each renewable control period may vary from time to time depending on the amount of energy drawn by the picture tube from the circuit and also on the magnitude of the unregulated DC voltage.

Energy is still transferred to inductance 322 and through transformer 324 to capacitor 341 until slightly after the next time T3, when the energy transfer sequence begins, resulting in a net shift to deflection circuit 340 to compensate for circuit losses.

Other embodiments of this invention will be apparent to those skilled in the art. For example, switch 326 can be connected to any series-connected path that includes inductance 322 and winding 324a. The power of the last electrode can be taken from a separate transformer, which is connected in parallel with the deflection winding 350. The amount of energy circulating in the system can be detected by means other than using the reference voltage generators 33 and 333, such as using a voltage across the capacitors 52 or 352. Likewise, the shape of the control circuit required to provide the relevant pulses to control the SCR rectifier 28 or transistor 328 is not critical. The inductance provided by inductances 22 and 322 can instead be provided by leakage inductance associated with the primary windings of transformers 24 and 324, respectively.

Claims (1)

70104 Claim: A controlled television line deflection device adapted to be supplied from an uncontrolled DC voltage source (10) and comprising a) a primary switching means (26) comprising a semiconductor switch (28) having a controlled current path and a control electrode and which can be made non-conductive by commutation , b) a transformer (24) comprising a primary winding (24a) connected to a controlled current path and uncontrolled DC voltage source (10) for supplying energy from said source to the primary winding (24a) and a secondary winding (24d) electrically isolated from the primary winding, c) a deflection winding (50) and a charge capacitor (41) both connected to the secondary winding (24d); d) a deflection circuit (40) isolated from the source (10) and including a line switch (42) connected to the deflection winding ( 50) and which is used at a line frequency to enhance the deflection current in the deflection winding (50) during repeated line and return cycles, whereby the return during the cycle, a voltage across the deflection winding is connected to the primary winding (24a) to commutate half of the wire switch (28) to a non-conductive state; e) a control circuit (68, 82) a detection signal for the control electrode of the semiconductor switch (28) for determining the time during which the semiconductor switch turns on, f) an additional energy source (90, 92, 93) available during the start-up period immediately after switching on the line deflector, characterized by g) a rectifier (98) connected between the auxiliary energy source (90, 92, 93) and the charge capacitor (41) and biased during the start-up period to supply energy to the deflection circuit (40) and then biased to separate the auxiliary energy source from the deflection circuit. tl