FI103612B - Hydroacoustic distance measurement system - Google Patents

Description

103612

Hydroacoustic distance measurement system

The present invention relates to a device for transmitting and receiving hydroacoustic pulses used to determine the spatial difference between the pairs of devices. In a preferred embodiment, the invention relates in particular to high frequency hydroacoustic digital signal processing transceivers used at known locations along hydrophone tow lines to determine their shape and geodetic locations.

10 In search of geologic formations that are likely to contain oil or gas, the offshore seismic exploration industry is investigating the outer layers of the submarine ocean crust by towing a hydrophone group, periodically triggering a source of acoustic energy, and storing hydrophone responses. . The hydrophone group is arranged linearly in a line whose depth is controlled. A fishing line, which may be a few kilometers long, may also include a front buoy tied to a flame at the front end of the line and a rear buoy tied at the back end as surface marks.

20 Historically, only one line containing a hydrophone group was used in the research boat during the study. The accuracy of the study depends, among other things, on the accuracy of the hydrophone line shape estimation and the position of the known point on the line.

One way to evaluate the shape is to mechanically model the line and calculate its dynamic performance at different towing speeds and in different environmental conditions. Of course, the accuracy of the estimation is only as good as what the model is. Installing magnetic compasses and depth sensors along the line improves the estimation of the line shape. Depth and magnetic heading data for line components are transmitted from distributed compasses and depth sensors to a guide on the tow boat to immediately calculate line shape and store the original data for further detailed processing. An accurate shape estimate is achieved in this way.

Just as important as evaluating the shape of a fishing line is attaching its location to a geodetic reference coordinate system. In-boat radio-35 position receivers are typically used to attach the 103612 2-point on the boat to the geodetic reference coordinate. Precise optical positioning systems, such as lasers, are then used to attach the front buoy to the geodetic reference coordinate system. It is also common to use a radio position receiver in the boat to secure the position of the rear buoy. The locations of the distributed compasses and 5 depth sensors with respect to the buoys are then estimated based on the model of the line and buoy flares. Inaccuracies in the model result in absolute errors in the transfer of the geodetic reference coordinate system from buoys to the fishing line. Further, the performance of optical positioning systems is deteriorating in severe weather.

An important benefit of oil and gas research is the development of three-dimensional seismic research, which often uses one or more hydrophone lines. When towing one or more boats with multiple lines, more seismic hydrophone data can be collected in a much shorter time than one line, resulting in a significant reduction in research costs. 15 When using multiple lines, accurate estimates of the exact positions of the hydrophone lines with respect to each other and to the acoustic source are essential. Fortunately, several lines towed more or less parallel form geometry that is advantageous in determining the position of the lines relative to each other, the boat, the acoustic source, or the 20 guns, and acoustic distance measurement to the buoys. With private hydroacoustic transceivers positioned along fishing lines, acoustic source, boat or boats, and buoys, acoustic passage times of transceiver pulses from transceivers and surrounding transceivers can be transmitted to a controller on the boat where the location solution can be performed and for further processing. Using the speed of sound in the water, the controller converts the travel times into spatial differences between the transceiver pairs while developing a location solution. Combined with information from the radio positioning system and grouped depth sensors and compasses, the positioning solution is perfect.

In a typical three dimensional survey using one or more lines, the towing boat or tows follow a more or less constant direction at a more or less constant speed through field 35. Waves, wind, current and inevitable fluctuations in boat speed and direction constantly affect the shape of the lines. The acoustic source or weapon 103612 3 is triggered periodically, for example, every 10 seconds. The compressed air pulse is forced into the water to form a bubble. The bubble size randomizer causes an acoustic pulse to radiate through water to the ground. Pulse reflections from geological structures are received by hydrophones and data representing these reflections are transmitted to a controller on board the boat. Each shot and the associated period of time during which the acoustic echoes are detected is known as the firing point. It is important that sufficient data is available for each trigger point to complete the perfect location solution. In a group of long lines, each of which has acoustic transceivers distributed, many acoustic distances need to be measured. In theory, it would be best if all the distances to be measured could be determined at the same time before the fishing line was forced to change shape and position. Unfortunately, this is not possible in practice. The idea is then to measure all acoustic distances in as short a time as possible, which requires a high throughput for each transceiver.

The difference between the transceiver pairs is usually measured either by one-way or two-way distance measurement. In unidirectional distance measurement, the first transceiver transmits a hydroacoustic pulse at time ζ. The pulse propagates through the water where it is received by the second transceiver 20 at time t1. The time difference is proportional to the spatial difference between the two transceivers. In accurate unidirectional distance measurement, the timers of both transceivers must be closely synchronized with each other, since the value t is determined by the receiving transceiver. In bidirectional distance measurement, each transmitter-25 receiver transmits a pulse, the first at time t1 and the second at time t2s. The first transceiver receives the second pulse at time t1r, and the second transceiver receives the first pulse at time t2r. Even if the timers of both transceivers are not synchronized, the spatial difference is proportional to [(t1r-t1s) + (t2r-t2s)] / 2, because the offset between the 30 timers is removed in subtraction. Thus, the exact synchronization required for unidirectional distance measurement is not required in bidirectional distance measurement systems.

While bidirectional distance measurement avoids the synchronization problem of unidirectional distance, each transceiver has to perform more processing in the bidirectional distance measurement method, i.e., each transceiver must receive a pulse for each 103612 of the distance it is involved in measuring. The arrival times of the received pulses and the time of transmission of the transmitted pulse, or their differences, must be sent to the controller on the boat for each launch point. The transceiver, which is involved in multi-range measurement, has to process a lot of data. Thus, only a transceiver with high throughput can be effectively used in a bidirectional range measurement system.

Thus, it is an object of the present invention to provide a hydroacoustic transceiver capable of the high execution speeds required for bidirectional distance measurement without the need for accurate time synchronization.

More particularly, the invention relates to a hydroacoustic distance measuring system according to claim 1, and to a method according to claim 9.

If all the transceivers in the range measuring system transmit only 15 on a single frequency, the only way to measure multiple distances is by time division multiplexing, i.e. staggering the transmissions so that no two pulses transmitted by different transceivers can arrive at any one receiver at a time. This requirement, in addition to making the timing of the transmission nightmare, results in the long distance measurement of many distances 20, which causes errors in the location solution.

Another object of the present invention is to provide a transceiver capable of transmitting and receiving hydroacoustic pulses having Selected Properties.

A: The problems with acoustic distance measurement are. still errors caused by multipath interference 25. The straight path from the transmitting transceiver to the receiving transceiver is the direct path that defines the true spatial difference. Other paths are due to reflections of the transmitted pulse from the ocean surface or bottom. Depending on differences in the length of the reflected roads directly to the road, the reflected pulses may interfere with 30 direct pulses. Such interference can be destructive, preventing or distorting pulse detection, leading to an error in determining the direct pulse arrival time. In addition, the shorter the transmitted pulse, the more susceptible it is to multipath interference and the higher the spatial resolution. It is well known that the narrower the pulse, the wider the bandwidths of the transmitter and receiver 35 should be. In other words, there is a connection between resolution (pulse width) and bandwidth.

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The wider bandwidths for each pulse of a given carrier frequency require that in a frequency division multiplexed system each channel is further discriminated. In typical hydroacoustic transceivers, it is difficult to accommodate a wide range of carrier frequencies.

One way to compress more channels to a particular transceiver bandwidth is to synthesize narrow transmit pulses and detect them using a filter matched receiver. With a filter matched receiver, it is possible to achieve a lower pulse width and bandwidth input than conventional receivers. However, a real filter matched receiver cannot be implemented in the typically used linear analogue transceivers. Thus, analog transceivers must sacrifice resolution to enjoy the flexibility offered by multiple channels, or must sacrifice frequency-specific flexibility to improve resolution.

One technique used in analogue transceivers to avoid the multipath problem is to transmit pulses alternately on different channels and to analyze the travel times measured for each channel. The idea is that for the same reflected path, the interference between direct and reflected pulses is different at different frequencies, and that at one of these frequencies the interference is not destructive and distance measurement can be made. This use of multiple frequencies to solve the multipath problem requires more time, since each transceiver must transmit more than one pulse in order to obtain a valid distance measurement.

Thus, it is a further object of the present invention to provide a hydro-acoustic transceiver operating on a plurality of efficiently compressed channels and transmitting sufficiently narrow hydroacoustic pulses to minimize multipath interference.

These and other objects and advantages of the invention will become apparent and partially apparent from the following, and are accomplished by the present invention which provides a device for transmitting narrow hydroacoustic pulses and determining arrival times for received hydroacoustic pulses for measuring spatial differences between the pairs of devices. An example of such a device is a hydroacoustic transceiver used as part of an acoustic distance measurement system that evaluates the locations and shapes of hydrophone lines to improve the accuracy of a staging study. In this distance measurement system, the individual transceivers can be attached to various 103612 6 points along hydrophone lines, to the gun, to the front buoy, to the rear buoy, to the towed embedded towing bracket, or to the tow boat hull. The controller on the boat, which is a processing device of some kind, controls the operation of the system and collects data from the transceivers along the communication links 5.

The present invention relates to a transceiver having a converter for converting hydroacoustic energy into electrical energy and vice versa. In a preferred embodiment, the transducer is a ceramic sphere having a bandwidth ranging from 50kHz to 100kHz. The transducer is alternately connected by either an electronic transmission path of the transceiver 10 or a transmission / reception switch on its receiving path.

With the switch in the receiving position, the transceiver listens to pulses from other transceivers. The receiving path leads to a hydroacoustic energy-converting electrical energy conversion device 15, such as a sampling analog-to-digital converter, which converts the input electric energy into a resulting series of digital words, or into samples. In a preferred embodiment, the receiving path includes a high pass filter for attenuating low frequency noise, which may be significant in the marine environment. In a queue of digital samples, the detector detects the presence of pulses transmitted from the other 20 transceivers, the pulses having known characteristics. In a preferred embodiment, known features include pulse shape and carrier frequency, and the detector is a multi-channel digital filter implemented with an integrated: digital signal processing circuit (DSP). Digital filter coefficients stored in a memory device such as EPROM or 25 RAM are designed to detect pulses of known shape at one of five carrier frequencies, or channels, in the range of about 50kHz to 100kHz. The digital filter detects pulses on each channel by correlating the queue filter coefficients of the digital samples. The relative maximum correlation values obtained from the filter represent the detected pulses, the magnitudes of the correlation values indicating their signal strengths. The time count value of the timer device that occurs when pulse detection occurs, which represents the pulse arrival time, is stored in memory. The detection device likewise saves the signal strength of each received pulse. Up to eight pulse arrival times and signal strengths can be saved.

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When the transducer is connected to the electrical transmission path of the transceiver, the transceiver outputs a hydroacoustic pulse of known shape and carrier frequency. In a preferred embodiment, one pulse is transmitted for each trigger point at one of five carrier frequencies. Pulse 5 is digitally synthesized in the synthesizer at the frequency of the trigger point. The counting comparator, in cooperation with the timer, determines the transmission time period by comparing the counting value of the timer with the value stored in the register. In a preferred embodiment, the count value of the timer is set to zero when the count value matches the count value in the register. The arrival times of the received pulses are compared with the transmission time. The digitally synthesized pulse is converted to an analog signal by a 12-bit digital-to-analog converter and fed to the converter via a switch along an electrical transmission path including a bandpass filter to attenuate digital switching noise and power amplifier pulse power to a level sufficient The pulse is connected to the water by means of a converter. In a preferred embodiment, the transmit / receive switch is in the transmit position for about 500 microseconds for each trigger point period. To save power, the power amplifier is only turned on for a short transmission time.

In a preferred embodiment, the timer, comparator, detector, and synthesizer other than the digital-to-analog converter is implemented on a DSP chip, its support circuitry, and its operating machine code. The DSP chip is used because it can rapidly perform many arithmetic and logic functions, such as those required to implement a multichannel digital matched filter. The DSP chip provides high throughput, nearly simultaneous bidirectional distance measurement, and good multi-path interference tolerance on two or more lines.

In a typical embodiment, the transceivers are attached to a plurality of positions, the front and rear buoys, the gun, the tug booster and the 30 boat hull. Prior to deployment while the lines are still on the boat deck, the controller will position each transceiver through a communication link. The adjustable parameters are: a) the transmission frequency; (b) the time period of the inquiry, ie the transmission speed; c) Transmission time, that is, transmission time to zero with respect to the setting, which occurs at the beginning of the polling period or when 35 basic synchronization settings are performed; d) reception window opening hours and closing times, i.e., an acceptance time interval for each incoming pulse relative to the beginning of 836 of query 103612; (e) the receiver channel number (or carrier frequency); (f) the acceptance threshold; and g) a calibration value for the reception time. The adjustable parameters are stored in registers. Eight registers are reserved for each position d-g, allowing up to eight pulses to be received for each of the 5 trigger points that meet the criteria defined by the corresponding settings in d-g. In addition, each transceiver may be configured as a response device which transmits a pulse to a selected channel only when it receives a pulse on a particular channel. Responders are used in system locations, such as rear buoys, that do not have a communications link to the boat controller. 10 The communication link between the controller and each transceiver is accomplished through a common line. The microcontroller in each transceiver processes the communication and stores the layout parameters in the registers that the DSP chip has access to. In a preferred embodiment, the multipath interference is eliminated by terminating the acceptance period for a particular pulse after it is detected.

During operation and distance measurement, the controller on the boat freely synchronizes each transceiver from time to time via a communication link. When using bidirectional distance measurement, a few milliseconds synchronization is sufficient and can be performed via a communication link. For each launch point, the transceiver always transmits eight pulse arrival time and signal strength data back to the boat controller, where it is processed and stored for further resolution of the hydrophone line location and shape.

The foregoing and other features of the present invention will be more fully understood from the following description with reference to the accompanying drawings, in which:

Figure 1 is a block diagram illustrating one of a plurality of transceivers used in a positioning device in accordance with the present invention.

Figures 2 to 5 show flowcharts representing a program of a processor located in transceivers 30.

Fig. 6 shows the desired frequency response of a transceiver channel.

Figure 7a shows an idealized frequency spectrum of transmitted pulses for five channels.

Figure 7b shows an idealized frequency spectrum of received pulses for 35 channels sub-sampled at 100 kHz.

9 1Π3619 I ^ <4_

Figure 8 represents the actual transmitted pulse and its envelope on a 95 kHz carrier.

Figure 9 represents 50 in-phase and quadrature receiver correlation coefficients for a channel.

Figure 10 is a block diagram of a pulse detection technique used in a preferred embodiment.

Figure 11 shows the autocorrelation function of the envelope of the received pulse.

Figure 12 shows a top view of the use of transceivers used in the present invention.

Referring now to Figure 1, a block diagram of a device according to the present invention is shown. The converter 20 converts the acoustic energy received through the water into the received electrical energy and inversely converts the electrical pulses into acoustic pulses and connects them to the water. An example of such a transducer is a ceramic ball having a relatively uniform response in the range 50kHz to 100kHz 15 and a more or less omnidirectional radiation beam pattern. The converter 20 is alternately connected by either an electronic transmission path 22 or an electronic reception path 24 via a T / R switch 26 (transmit / receive). In a preferred embodiment, the T / R switch is an electromechanical single-pole and two-position relay, although it should be noted that other types of switching may also be used, for example semiconductor switches.

With the T / R switch 26 in the receiving position shown in Fig. 1, the converter is coupled to the electrical receiving path 24. The receiving path comprises an active high-pass filter 28 which provides a substantial voltage gain. In a preferred embodiment, the high pass filter 28 is a fourth 25 degree Butterworth filter having a cutoff frequency of about 40 kHz and a voltage gain of about 300. The filter attenuates noise in the range of 0-50 kHz. Although a preferred embodiment uses an active filter with amplification, it is also possible to use a separate preamplifier together with a passive filter as the amplification stage. The filtered electric energy at the output of the high-pass filter 28 30 is sampled and converted into a queue of digital samples by a sampling AD converter (analog-to-digital converter) 30 which provides a 12-bit output. The sampling frequency is determined by the output frequency fs of the frequency divider 32. The frequency divider divides the input frequency supplied thereto by a source of clock signals from oscillator 34. In a preferred embodiment, the oscillator frequency f 1 is 20 MHz, the frequency divider 32 divides by a factor 200 when the sampling frequency f s is 100 kHz.

10 103612

Thus, the AD converter 30 provides new samples at a frequency of 100,000 samples per second.

In a preferred embodiment, the 12-bit digital sample obtained from the AD converter 30 is read by processor 36 via its bidirectional data bus 385. The conversion stop signal 40 from the AD converter 30 informs the processor 36 that the digital sample is ready to be read. The processor then commands the AD converter to output the digital sample to bus 38 together with the AD read control signal 42. The processor implements a multi-channel digital matched filter programmed to detect the presence of acoustic pulses with known characteristics in a queue of digital samples. The details of the disclosure will be explained below.

When the converter 20 is coupled to the electrical transmitter 22 selected by the incoming transmit / receive control line 44 of the processor 36, the electrical pulses synthesized in the processor control can be directed to the converter for connection to water 15. When synthesizing the pulse to be transmitted, processor 36 outputs a sequence of 12-bit data words over data bus 38. In a preferred embodiment, a given pulse string contains about 200 digital words. When printing consecutive words at 400,000 words per second, the entire queue takes about 0.5 ms, i.e. the duration of the transmission pulse. The sequential digital words from the queue are encrypted by a DA latch signal from 20 data buses from 38 processors to 50 inputs of a 12-bit DA converter (digital analog) at 400 kHz. The DA converter converts the 12-bit digital word in its input to its analogue voltage output. Since the analog output of the DA converter is undefined during its conversion time, i.e., the time period beginning at and ending at the end of the 25 conversions, the deglitcher circuit 52 is used. During the conversion time, the interrupt voltage off circuit 52 short-circuits its output. When the output of the DA converter 50 is stable, the interference voltage peak output circuit 52 overrides the output of the DA converter directly to the reconstruction low pass filter 56. The interference voltage output circuit control line. The control 54 switches, via the latch 48, the output of the disturbance voltage peak output circuit 52 to ground and the output of the DA converter 50, respectively, via the latch 48. The low pass filter 56 smooths out the output of the interference voltage peak circuit 52, eliminating most of the digital switching noise. In a preferred embodiment, the filter 56 is a passive third order Bessel filter 35 which provides the smallest phase distortion in the frequency range of the pulses. The waveform at the output of the filter 56 is the waveform of the desired synthesized pulse 4, 103612.

The synthesized pulse from the filter is applied to the converter 20 via an electrical transmission path 22 which includes a controller stage 58 and a power amplifier 60. The controller stage 58 acts to buffer the low voltage analog portion from the high voltage power amplifier 60 5. In a preferred embodiment, the controller stage 58 also provides voltage amplification by a factor of two for the synthesized pulse. A power amplifier 60 having an operating frequency range of about 50 to 100 kHz and gain of about 32 dB for a rated load of 100 ohms amplifies the pulse to appropriate levels for transmission.

The communication interface 37 enables the processor 36 to communicate with the external controller 59 via line 61A. The parameters defining the transmitter and receiver layouts can be selected in the controller and then sent to processor 36 and stored in memory. The controller may also synchronize processor 36 with processors 15 in a plurality of similar circuits via common lines 61A or separate lines 61B to 61N through an interface. Further, the arrival times of the received pulses and their signal strengths are transmitted to the external controller 59 via interface 37. The processor 36 also reads data collected by the data acquisition module 35, which may include sensors, such as pressure transducers for depth measurement and temperature transducers for seawater temperature measurement. The data obtained from the data acquisition module can be transmitted to the external controller via the communication interface 37.

The synthesis of the transmission pulses is controlled by the processor 36 of FIG. 1 and the action program is stored in its memory. The flowchart of Figure 2 illustrates 25 transmission subroutines used to synthesize a transmission pulse. When it is time to transmit the pulse, the transmission subroutine is called and program execution jumps to step 62 of FIG. 2. First, the sequence of digital words or transmission coefficients cN (i) corresponding to the carrier frequency or channel N through which the pulse is to be transmitted. Shipment. The layout register 30 is a memory location in the processor that can be programmed via the communication interface 37 of FIG. In the preferred embodiment, the transmission layout register contains one of six possible values. Values from one to five are defined by channels N = 1 to 5, or corresponding carrier frequencies of 55 kHz, 65 kHz, 85 kHz and 95 kHz. A sixth possible value of 0, 35 prevents pulse transmission. For example, if the transmission layout register contains a value of 3, a pulse with a carrier frequency of 75 kHz is synthesized.

12 103612

When the correct set of transmission coefficients is selected, in step 64 the data pointer i indicating the first member of the selected sequence cN (0) is reset. Next, in step 66, the disturbance voltage peak output circuit 52 of Figure 1 is short-circuited through latch 48 and control line 54 in anticipation of a DA conversion. The conversion begins when the transmission coefficient cN (0) of i th 5 is applied to bus 38 and locked to the input of the DA converter 50 via the latch 48 and the DA control line 46. The processor then delays, at step 70, a time sufficient to complete the conversion. After the delay, the short circuit in the output of the interference voltage peak output circuit is interrupted and the output is coupled to the output of the DA converter as shown in step 72. According to step 74 10, the data pointer i is then compared to the total number of coefficients XMTCNT. In a preferred embodiment, XMTCNT = 203. If the data pointer is less than an integer, execution continues to step 76. Otherwise, indicating that the entire string has been converted and pulse synthesis is completed, execution continues in steps 80 and 82 which short circuit 15 interrupt voltage peak output circuits execution of the main call program. The data pointer i is incremented in step 76 to indicate the next successive transmission factor. The second processor delay is achieved in step 78 before the operation returns to step 66. The delays of steps 70 and 78 are selected such that during pulse synthesis, the interference voltage peak output circuit is resolved half the time. Delays are further selected such that, in a preferred embodiment, the passage of the loop defined from step 66 through step 78 and back to step 66 takes 2.5 microseconds at an output sample rate of 400 kHz. Five queues cN (i) are stored in the memory of the processor 36. A change to another set of synthesized signals only requires that the values in memory 25 be changed. This change can even be made through the telecommunication interface 37. In an analog system, the hardware should be replaced.

A device for detecting the presence of acoustic pulses from the queue of digital samples obtained from the AD converter 30 in Figure 1 is implemented by a correlation routine performed by processor 36. The AD converter 30 produces new digital: 30 samples at 100 kHz set by oscillator 34 and frequency divider 32.

At the end of each conversion, the AD converter communicates to the processor 36 via the conversion termination line 40, which causes the processor to immediately perform an interrupt routine. The routine, referred to as A / D READ in Figure 4, forces the AD converter to export a 35 sample of the digital bus 38 to the data bus through a signal passing through the AD reader line 42. The processor then reads the sample and saves it to the next available location 103612 13 in the rotation buffer 83 in its memory. In a preferred embodiment, the rotation buffer 83 stores 50 samples in memory. When the buffer is full, the next sample replaces the oldest 50 samples, so that the buffer always holds up to 50 recent samples. Step 124 in Figure 4 performs this function before returning to pre-program execution as shown in step 126. At this point, it should be remembered that the transmission pulses are synthesized at about 200 times the frequency of 400 kHz, resulting in a 0.5 ms pulse. Because the AD converter in the receiver only samples at 100 kHz, 50 consecutive samples cover a 0.5 ms time period, i.e. the pulse width. Multiple transmission factors are used to simplify the hardware required to implement the reconstruction low pass filter 56.

The correlation detection is implemented in the processor 36 by the detection routine whose flowchart is shown in Figure 3. To simplify the drawing, the flowchart of Figure 3 is drawn to detect a pulse on a single channel. In reality, a detection routine can be performed simultaneously for up to five channels simultaneously. The first step 84 of the routine resets the reception data pointer j, which is used as a count value index for the start of the detection routine. In step 86, the processor waits for a timer interrupt, which occurs every 0.1 s or at 10 kHz. A flowchart of the timer interrupt routine is shown in Figure 4. 20 When the timer interrupt occurs and the AD converter is read, the program execution continues to steps 88 and 90 which reset the correlation pointer k and the correlation accumulators yi (j) and yq (j). The iteratively performed steps 92, 94, 96 and 98 perform the 50 most recent sampling samples x (j), j = -49 ... 0, in-phase and quadrature correlation with 50 in-phase correlation coefficients hi (k) and 50 quadrature 25 coefficients hq (k) per channel. The odds are stored in the processor memory. A total of 500 multipliers have been deposited for the five channels. Just as the transmission coefficients cN could be modified without significant hardware changes, so can the correlation coefficients. As described in more detail below, the coefficients represent a known form of transmitted pulses for each of the 30 channels. The iteration through steps 92, 94, 96 and 98 continues until the correlation indicator k, incremented in each iteration, reaches one-fourth of the number of values in the queue of transmission pulses. In a preferred embodiment, XMTCNT = 203 and XMTCNT / 4 = 50. Thus, each correlation battery represents a sum of 50 terms. In standard math-35 terms, in-phase correlation is given by yi (j) = S [x (jk) hi (k)] and quadrature correlation is obtained by yq (j) = S [x (jk) hq (k) j, 103612 14 where the sum of k is between O ... 49.

When the correlation calculations are completed, the execution of the program continues in step 100 which calculates the sum of the net correlation values y (j), the vector sum of the in-phase and quadrature correlation steps, or the square root of the squares 5 of yi (j) and yq (j). In step 102, the net correlation value is compared to the reception threshold value THLD, which is a selectable value in the reception layout register in the processor memory. If the net correlation value is less than the threshold value, the net correlation value y (j) is set to zero in step 104. In the second case, the execution of the program continues in step 106, 10 which determines the maximum correlation value. In step 106, the last net correlation value (night) is compared to the previous maximum peak value p (j-1) · If the last value is higher, it replaces the previous value in p (j). Otherwise, the previous value p (j-1) is placed in p (j). Thus, p (j) contains the maximum correlation calculated so far. Step 108 causes program 15 to override the peak detector implemented in steps 110, 112, 114, and 116 if the peak net correlation value is already detected. If the peak net correlation value has not yet been expressed, the execution of the program proceeds to step 110, which compares the most recent net correlation value night) to the maximum correlation value pi found so far. If the night) is less than 20 d), the correlation values will start to decrease and the present night) will be the first value after the peak. If the peak is detected and the device acts as a response device, the timer time count value is reset to zero and the program execution returns to the paging routine as shown in steps 111, 113, 122. If the peak is detected and the device is standard, program execution proceeds to step 112 , which corresponds to the peak ö'-1), in the processor memory to the location marked PLSTIM. Because the data pointer is incremented each time a new correlation value is calculated, i.e., every 0.1 s, the data pointer acts as a timer that produces periodic time count values. Step 112 causes the device time count value to be positioned as a maximum correlation value, indicating pulse gain time relative to the beginning of the detection routine. The signal strength of the received pulse, which is its peak correlation value p (j), is stored in step 114 at a location in the processor memory designated PLSLVL. Step 116 sets the SKIP flag indicating that the peak has been configured. Notwithstanding the results of the decisions of steps 108 and 110, program execution continues at step 35,118, which determines whether pulse detection on a particular channel has been completed. The receiver layout register in processor memory contains a value of 103612 15 RCVCNT, which defines an acceptance time period relative to the beginning of the detection routine during which the correlation detection routine operates on a particular channel. When the data pointer j reaches the value RCVCNT, program execution returns to the routine run above through step 122. Otherwise, the data pointer j is incremented in step-120 in step 120, and the program execution is looped back to step 86 to compute the next correlation value. It should be noted that the square root function performed by step 100 can be eliminated without affecting arrival times. If the phase is eliminated, the net correlation values are merely squares of the correlation values, and the thresholds must be squared accordingly. The purpose of the square-jute-10 row step 100 in the preferred embodiment is to scale the net correlation value to the same units as the sampling samples.

The software timer is implemented with a timer interrupt routine, the flowchart of which is shown in Figure 4. The routine is run every 0.1 second, or at 10 kHz. The oscillator 34 of Figure 1 is a source of clock signals to which the execution frequency of the timer routine 15 is compared. Each time the timer routine is run, the time count value stored in the processor memory is incremented to the location designated by TIMER. If the time count value reaches the maximum time count value TCNT, the time count value is set to zero as shown in steps 128 and 136. Further, if the base synchronization command is received, the time count value is set to zero and the base synchronization is reset so that the timer routine does not hold 132 and 134. The basic synchronization command transmitted through the communication interface 37 of FIG. 1 is used to synchronize devices relative to each other, as usual once for each trigger point. If devices are not synchronized for each of the 25 trigger points, TCNT determines the polling time period that sets the transmission rate.

The timing of the transmission and detection routines is addressed by the main program shown in Figure 5. The run of the main program starts from the setting and the run continues in the loop until the next setting. First, according to step 140, zero. A representative digital word 30 is applied to the data bus 46 of Figure 1 and blocked to the input of the DA converter 50 by the latch control line 46 passing through the latch 48. Next, step 142 sets the T / R switch 26 to its receiving position via the control line 44. The program execution then proceeds to the beginning of the loop, step 144. When the timer is interrupted (every 0.1 ms), the loop is executed once. First, the routine determines in step 146 whether it is time to place the T / R switch 26 in the transmit station, and if so, in step 148.

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In step 150, the time count value TIMER is compared to the time to begin synthesizing and transmitting the XMTTIM pulse. If they match, the transmission subroutine shown in Figure 2 is called for execution in step 152. If there is no time to transmit, or if the transmission routine has stopped executing, step 154 determines 5 whether it is time to place the T / R switch 26 at the receiving station. If so, step 156 sets the switch. In a preferred embodiment, the T / R switch 26 is held at the transmit station for a sufficient period of time to ensure passage through the synthesized pulse, or just above 0.5 ms. The T / R switch is in the receiving station for the rest of the time. In step 158, the time slip TIMER is compared to the value RCVTIM, which represents the start of the 10 acceptance period on a particular receive channel. If they match, step 160 calls the expression shown in Figure 3 to execute the subroutine. Otherwise, or after the expression subroutine is complete, the loop is restarted in step 144.

For the sake of simplicity, the flow charts described so far have been drawn to illustrate a single channel expression. In a truly preferred embodiment, simultaneous detection of five channels is always possible. In addition, eight pulse detection events can be recorded for each trigger point. The eight pulse detection events are defined by the values in the eight receiver setting registers 20 representing the preset states. The eight registers are stored in the processor memory. In a preferred embodiment, each of the eight receiver alignment registers includes: a) a receiving channel number that selects in-phase and quadrature correlation coefficients for use in the detection routine; 25 b) a reception threshold (THLD) which defines the minimum correlation value used in peak detection; c) Reception Window Opening Time (RCVTIM), which defines the start of the acceptance window during which the detection routine is run; (d) the Reception Window Closing Time (RCVCNT), which defines point (c). 30 acceptance window expiration relative to time RCVTIM; e) a bit that determines that the device acts as an independent response device by transmitting only in response to receiving a pulse as defined in a) to d) above.

Other preset registers used in the preferred embodiment include; 35 a) a transmission layout register which selects the transmission channel N and determines the coefficients cN (i); 103612 17 b) a query time period register containing a value (TCNT) which is a modulus of the timer count value, so whenever TIMER is equal to TCNT, TIMER is set to zero, restarting the transmission time period; c) a transition register including a value (XMTTIM) that defines 5 transmission times relative to the start of the transmission time period (TIMER = 0); and d) a calibration register containing a calibration value (CALTIM) that measures the pulse arrival time for otherwise undesired delays.

When each of the eight pulses is detected according to the conditions set by each of the eight receiver location registers, 10 of each arrival time (PLSTIM) and signal strength (PLSLVL) are stored in memory. Arrival time is deposited in one of the eight reception time registers; the signal strength is stored in one of eight signal strength registers. The eight reception time and signal strength registers correspond to the eight reception setting registers. The arrival time 15 is calculated with respect to the transmission time XMTTIM as follows: (RCVTIM + PLSTIM -XMTTIM + CALTIM).

Because the direct acoustic path is shorter than the reflected paths, the pulse arriving at the direct path arrives at the detector earlier than the pulses arriving along the reflected paths. By selecting the first net correlation peak in each of the 20 acceptance periods, the detection routine solves the multipath problem.

The preset registers are set by the external controller through the communication interface 37 of FIG. The values in the reception time and signal strength registers are transmitted to the controller via communication interface 37 for use in acoustic distance solutions.

As described above, the transmission signal sequence comprises a 0.5 ms pulse at five carrier frequencies: 55 kHz, 65 kHz, 75 kHz, 85 kHz and 95 kHz. The actual pulse shape is derived by an iterative technique known as the Parks-McClellan algorithm and described in Digital Signal Processing, Oppenheimer and Schafer, Prentice-Hall, 1975. Parks-McClellan -: 30 algorithm generates an optimal estimate of the finite pulse shape at the desired frequency spectrum . The actual output of the algorithm is a series of coefficients representing the magnitude of the pulses at even intervals. Several criteria have been used in the design of the signal series. First, a finite-length, narrow pulse is desirable to maximize spatial resolution and minimize multipath interference. Second, the channel separation required by the receiver to avoid channel crosstalk interference must be small to allow the transceiver to handle many 103612 18 channels for more flexible operation of the range measurement system. It is known in the art that the best receiver for a given set of transmitted signals is a filter matched receiver. A matched filter is defined as a filter whose impulse response has the same shape as the signal to which it is applied, but inversely with time. With a symmetric transmission pulse which is identical in time inverse, the impulse response is identical to the shape of the signal to which it is fitted. Like the symmetric transmission pulse, the filter matched receiver has a frequency response identical to that of the pulse frequency. Thus, selecting the desired frequency response of the receiver also sets the pulse frequency spectrum. Another noteworthy aspect of matched filters is that they cannot be implemented in linear analog systems because the impulse response inverse with time cannot be realized.

Fig. 6 shows the desired frequency response 170 and the actual response 172 of a particular receiving channel, which approximates the desired response with a pulse shape generated by the Parks-McClellan algorithm. The desired frequency response 170 can be observed to be in the form of a symmetrical triangle that decreases from the 0 dB peak at the carrier frequency to 80 dB downward at the carrier frequencies of the adjacent channels and remains constant at 80 dB 20 below them. Actual response 172 represents an approximation of the desired response resulting in a 60 dB suppression of channel crosstalk. Good channel crosstalk attenuation allows only 10 kHz difference between receiver channels, thus allowing equal frequency converter use. The bandwidth of the actual frequency response 172 of the receiver channel, which is the same as the pulse band-25, to which it is fitted, is about 10 kHz. FIG. 7a shows the overlap of the desired frequency responses of the five channels and the efficient use of the available spectrum in the range 50 kHz to 100 kHz. In an analog implementation, five separate bandpass filters would be required, each comprising a plurality of components.

Fig. 8 is a graph showing a transmission pulse envelope 180 and a transmission pulse at a carrier frequency of 95 kHz 182. Each of the drawings has about 200 values. In a preferred embodiment, 203 values are used. If each of the 203 values plotted on the envelope is denoted by cE (i), the transmission coefficients for each channel are given by 35 cN (i) = cE (i) cos [2p (yft) i], 103612 19 where f, = 400 kHz, transmission sample rate, fN = carrier frequency on channel N, and i runs between -101 ... +101. Processor 36 synthesizes a transmission pulse, for example, for channel 5 by transmitting successive coefficients to the c5 (i) DA converter 50 at a frequency of 400,000 times per second. The result is a transmitted pulse 5 of the form 182, but further smoothed by a reconstruction filter 56. By synthesizing pulses from digital words converted at a frequency four times the highest significant transmitted frequency (100 kHz), digital switching noise can be effectively removes the third order Bessel low pass filter 56. The Bes Selel filter is used to minimize phase distortion in the filtering process. Slower conversion speeds require more complex and expensive higher order filters.

Although in a preferred embodiment, the total duration of the pulse envelope curve is about 0.5 ms, most of its energy is in a window of 0.1 ms that encloses the center of the pulse envelope. Thus, it is possible to achieve a time resolution of about 0.1 ms, which in contrast to a spatial resolution of 0.15 m in seawater. The narrow pulses also make it possible to distinguish the desired pulses from the reflected pulses having path differences of about 0.1 ms, which allows the detector to distinguish the direct pulse from the reflected pulses, thereby effectively eliminating the multipath problem.

As a general rule, when designing a receiver for sampled data, the sampling frequency must be at least twice the highest frequency in the received signals to avoid false reproduction. * False repetition is a phenomenon in which the frequency component fa, sampled at another frequency fb, where fb <2fa, appears as a component shifted to the frequency f3 = fb - fa. If the other frequency component is actually at frequency f3, the displaced component will distort it. It is impossible to distinguish between these two separate components. If this general rule were followed in a preferred embodiment, the sampling frequency fs: 30 of the AD converter 30 should be about 200 kHz. Such a high sampling rate would make it difficult to implement a five-channel receiver. In the present preferred embodiment, the known frequency transfer characteristics are utilized to allow for a slower sampling rate. Sub-sampling the signals in the 50 kHz to 100 kHz band at 100 kHz transmits these 35 signals to the 0 Hz to 50 kHz band. In reality, the spectrum above 50 kHz folds around 50 kHz (half the sampling frequency). To operate this 103612 20 sub-sampling technique, a high pass filter 28 is used to eliminate the pre-sampled energy in the 0 Hz to 50 kHz band to avoid false repetition. A cut-off frequency of about 40 kHz minimizes phase distortion in the pass band while sufficiently attenuating noise in the block. The spectrum of Fig. 7a 5 is converted from 50 kHz to 100 kHz into 0 Hz to 50 kHz of Fig. 7b. The carrier frequency of the transmitted channel 1, 55 kHz, sampled at 100 kHz, appears as a frequency of 45 kHz in the transmitted spectrum; the carrier frequencies of the other channels appear as follows: channel 2 - 35 kHz, channel 3 - 25 kHz, channel 4 - 15 kHz and channel 5 - 5 kHz. The natural infiltration of the higher frequencies by seawater and the attenuation of the transducer 12 dB / octave above the cut-off frequency minimize the effects of false repetition at even higher frequencies in the 0 Hz to 50 kHz band.

A transmission pulse synthesized from a coefficient of 203,400 kHz is sampled at the receiver at 100 kHz and correlated with 15 in-phase and quadrature sequences, each having 50 coefficients. The in-phase and quadrature receiver coefficients hi (k) and hq (k) are plotted in Figure 9 for channel 5. The in-phase sequence of coefficients hi (k) is generated by multiplying the shape of envelope 180 by a 5 kHz sine waveform whose peak coincides with selecting 190 samples from the resulting waveform which are always 0.01 ms apart. The quadrature coefficients are similarly developed except that the envelope 180 is multiplied by a 5 kHz sinusoidal waveform shifted 90 °. A diagram of the resulting queue hq (k) is shown in Figure 9.

In the quadrature detection diagram shown in Fig. 10, 50 successive samples of AD converter 30 are correlated x (k) covering 0.5 ms with two 50-factor sequences hi (k) and hq (k) in correlators 200 and 202. (AD converter) With the exception of 30, the other blocks of FIG. 10 are implemented with program code). The resulting correlation values yi (j) and yq (j) are squared. 30, and summed in adder 206. Finally, the net correlation value y (j) is derived by the square root block 208. It should be noted that the square root block 208 is used merely to scale the net correlation value to the same units as the sampled unit. The use of a matched filter in the detection graph has the great advantage that the sequence y (j) is the autocorrelation function of the transmission pulse envelope. 35 An interesting feature of the autocorrelation function of the symmetric pulse is that it is also symmetric, but twice as wide. Because the autocorrelation function defined by y (j) 21 103612 is twice as wide over time, its frequency spectrum is about half the pulse width. It should be remembered that each transmission pulse has a bandwidth of about 10 kHz, which makes the autocorrelation function have a bandwidth of 5 kHz. Thus, the correlation calculations only need to be performed at 5 kHz at 10 kHz in order to avoid falsifying the correlation function. Thus, the use of a filter matched receiver has the advantage that the output of the correlator can be sampled at the same frequency as the bandwidth of the transmitted pulses instead of twice the bandwidth, as in unmatched filter receivers. This advantage greatly reduces the number of required correlation calculations.

A typical sequence of the net correlation values y (j) covering the pulse arrival time is plotted in Figure 11. It should be noted that the envelope of the values plotted in Figure 11 illustrates the autocorrelation function of the transmission pulse envelope and is therefore twice as wide as the pulse itself. The peak detector 209 of FIG. 10 selects from a queue a maximum of y (j) representing the pulse arrival time.

Performing many calculations on a multi-channel system requires a high performance processor. In a preferred embodiment of the invention, an integrated digital signal processing circuit (DSP) is used to synthesize the transmitted pulses and detect the received pulses. An example of these 20 DSP chips is the "DSP 56000" manufactured by Motorola, Inc., Scaumburg, IL. To further unleash the DSP chip from other functions so that it can operate more efficiently, a low-cost microcontroller is used to handle other functions such as timing and communication via communication interface 37 and to read data acquisition module 35. An example of such microcontroller is a 25 "Motorola MC68HC11AT ' Also manufactured by Motorola, Inc., Schaumburg, IL. Thus, the processor 36 according to the preferred embodiment comprises a DSP chip for pulse synthesis and detection, a general controller I / O function, timing, communication and control, and associated memories.

Referring to Figure 12, a top view of a seismic tugboat 220 towing two submersible hydrophone lines 222A and 222B is shown. The front buoy 226A and the rear buoy 230A are tied by a tie line 222A at its front and rear ends respectively. Floating buoys 226A and 230A are used as surface markings that mark the ends of the recessed line 222A. Line 222B is marked 35 with its front buoy 226B and its rear buoy 230B, respectively. Hydroacoustic transceiver capsules 236A-J containing the device of the invention are shown attached to embedded lines 222A and 222B at position # 2-4, front buoys 226A and 226B at position # 1, and rear buoys 230A and 230B at position # 5. To simplify the representation of Figure 12, the transceivers are shown attached to only two lines 222A and 222B and 5 to their buoys 226A-B and 230A-B. In a typical seismic study, additional transceivers may also be attached to the tug boat hull 220 and the air gun 234. These additional transceivers would typically be attached via a telecommunication link to an external controller, but could, if appropriate, act as counterparts. In addition, if the vertical thermal profile of the sea 10 obstructs the acoustic path between the transceivers in the buoys and the transceivers in the line, the transceivers installed in the tow reinforcements can be towed below the buoys to provide a new acoustic path. Further, more than two fishing lines can be used, each of which may have much more transceiver than shown in Figure 12, distributed along the fishing line. A typical 3km line can have about six transceivers along its length.

The transceivers 236B, C, D and 236G, H, I, which are directly attached to the fishing line, may communicate with the controller 20 on the towing boat 220 through the communication lines in the fishing lines 222A and 222B. In most applications, the buoys are tied to the line by line without electrical connection along the flame. Thus, transceivers mounted on a buoy cannot normally communicate with the controller. By way of illustration, it is assumed that the transceivers 236A, E, F, and J in FIG. 12 may not be in communication with the controller. Transceivers mounted on buoys are intended as transponders that transmit only when a pulse is detected on a designated channel. The dashed lines in Figure 12 represent the distances between the transceiver pairs 236. In typical use, all transceivers on line 222A are intended to transmit on the same channel 30 and all transceivers on line 222B on another channel. The response devices in the buoys are intended to transmit in response to pulse reception on the same channel as the corresponding response device itself transmits. Each transceiver, with the exception of the transceivers, is intended to receive pulses from surrounding transceivers. For example, transceiver 236B 35 is designed to receive five pulses: from transceivers 236A, C, F, G, and H. Pulse transmission times are staggered to prevent pulses 103612 23 on the same channel from arriving at the same transceiver. In typical use, the transceivers that are directly opposite each other send more or less together, the one closest to the boat first relative to those farthest from the boat.

In a preferred embodiment, the distance between the transceiver pairs is measured by bidirectional distance measurement to avoid the need for an accurate synchronization command. In bidirectional distance measurement, each transceiver transmits a pulse, the first at time t1x compared to the transceiver # 1 timer and the second at time t1x compared to the transceiver # 2 timer. The first receives the second pulse at time t1r, and the second receives the first pulse at time t2r. Even if the timers of the two transceivers are not synchronized with each other, the distance is proportional to [(ίι-ίιχ) + α2Γ-ω] / 2, 15 because the offset between the two timers is eliminated by subtraction. The time differences (t1r-t1x) in transceiver # 1 and (t2r · ^) in transceiver # 2 are values that are transmitted via the communication interface 37 to the external controller. The external controller then computes the distance by C [(tir-tix) + (t2r-t2x)] / 2, 20 where c is the speed of sound in water. The described calculation function is also applicable to the distance 240 in Fig. 12 between transceivers 236D and 236I, which may be in communication with an external controller in towing boat 220.

When calculating the distances between the transceiver and the response device, the calculation function 25 is different. The distance 242 between the transceiver # 3 and the transponder # 4 on the same line, where the transponder is intended to transmit in response to the pulse detection from the transceiver # 3, is proportional to [(VU-tj ^ · Difference (t3r-t3x)). transceiver # 3 informs the external controller that there is a time interval between transmitting 30 pulses of transceiver # 3 and receiving pulse from response device # 4. The delay t *, which is equal to XMTTIM + CALTIM and occurring on all response devices, is the pulse detection performed by the response device and This transmission function is suitable for the distance 242 between the transceiver 236D and the response device 35 236E in FIG. 12.

103612 The distance 244 between the transceiver 236D and the response device 236J, wherein the response device transmits in response to a pulse detection from a transceiver other than 236D, namely the transceiver 236I, is more complex. In this case, to simplify the shape 5 of the distance equations used, the subscript I is used to refer to the transceiver 236I, the subscript J to refer to the response device 236J, and the subscript D to refer to the response device 236D. Distance 244 is proportional to: (t | rJ "t | x)" (t | rD "tfc)" td 10 ~ [(^ DrJ '^ Dx) "td] / 2 + [(t | rO" t | x ) + (tori - ^ 0χ)] / 2, where (t, rJ - t, x) is the pulse arrival time of the pulse 236J relative to the pulse transmission time of 236l, (tlr0 - tlx) is the pulse from the transceiver 236D arrival time to 236l relative to pulse transmission time from 15 236l, (tQfj - t ^) is arrival time to pulse 236D from pulse 236J relative to pulse transmission time from 236D, (tDr1 - tDx) to 236I pulse to 236I relative to the pulse transmission time of 236D, and is the delay within the response device that occurs between pulse reception and transmission.

The terms in arc brackets are the arrival times that are sent to the external controller via the communication interface and t <is a known value. From these terms, the external controller may calculate distance 244.

An essential element in a perfect location solution is the depth of each transceiver at the distance measurement. Therefore, in the preferred embodiment 25, the data acquisition module of Figure 1 can transfer the depth information from the pressure transducer to the processor 36, which transmits it to an external controller via communication interface 35. In combination with the depth data, the data acquisition module may provide temperature information from the temperature sensor, which may be respectively transmitted to an external controller for estimating the local speed of the sound.

Referring again to Figure 1, the invention disclosed in this system may also be used in a unidirectional distance measurement system using an external synchronization signal in processor 36. An external synchronization signal is used to set the timer count value to zero. The synchronization signal 35 shuts down the processor and immediately sets the timer count to zero. In addition, each transceiver may be specifically designed as a receiver or transmitter only.

Thus far, devices and methods for a hydroacoustic positioning system have been described, and although the present invention has been described with respect to such specific methods and devices, it is not intended that such specific references be considered as limitations of the invention.

Claims (13)

103612 26 A hydroacoustic range measurement system used to determine the acoustic distances between private hydroacoustic transceiver locations at a plurality of submerged objects, comprising a plurality of hydroacoustic transceivers attached to different locations at said submersible transmitters (236A to 236J), each an energy converter (20) converter (20) adapted for underwater transmission and reception of acoustic pulses, characterized in that it comprises: an electronic system (22-58) coupled to the converter (20), comprising means for transmitting hydroacoustic pulse energy (20, 22) and 15 for receiving (20, 24, 30) which pulse energy is a predetermined property in the region, including processor means (36) for processing pulses that are in the excited feature area, the pulse detecting means (24, 28, 30, 40, 36) in the electronic system for detecting pulses received by the converter within the predetermined feature area, local time clock means (34, 36) in the electronic system for generating a local reference time 20) for received pulses having properties within a predetermined range, pulse synthesizing means (36, 48, 50, 52, 54) in the electronic system for generating a waveform having predetermined properties for transmission from said transducer (20) during pulses in the region, means (59, 61, 37) for synchronizing the local time clock means (34, 36) of the plurality of hydroacoustic transceivers 30, eva storage means (36, 83) for storing a timing diagram of local pulse transmitting and receiving events, 27 103612 timing means (36, 38, 42, 44, 46) at each transceiver in cooperation with local time clock means (34, 36) and storage means (36,83); for scheduling, scheduling, and initiating a sequence of local pulse transmitting and receiving events according to an event scheduling scheme stored in the storage means and in response to a synchronization pulse from the synchronization means (59). The hydroacoustic range measurement system according to claim 1, characterized in that the pulse detection means further comprises an mode of detecting simultaneously received pulses having different independent properties within a predetermined characteristic range. Distance measuring system according to claim 1 or 2, characterized in that the waveforms in the predetermined characteristic area are characterized by a pulse shape (172) and that the pulse detection means comprise pulse shape determination means (30, 32, 34, 36, 40) operating in a digital in a sampling mode, for sampling a pulse to determine the shape of the received pulses. A distance measuring system according to claim 1 or 2, characterized in that it comprises receiving hydroacoustic pulses having a predetermined carrier frequency above about 40 kHz and an electronic high-pass filter (28) coupled to said electronic system at about 40 kHz. to attenuate the received hydroacoustic noise at frequencies below. Distance measuring system according to claim 1 or 2, characterized in that it comprises a system which connects said electronic system (22-58) to an electronic system (220, 234) having a different underwater location and means for separating the distance between the two devices. to determine by measuring the time with the formula [(t1r-t1s) + (t2r-t2s)] / 2, wherein the transceiver in the first device transmits at time t1s and the transceiver in the second device transmits at time t2s and wherein the subscript r identifies the pulse reception time. 28 103612 Distance measuring system according to claim 1 or 2, characterized in that it comprises a bidirectional distance measuring means (59) for measuring the discriminating distance between the two transceivers (20) connected respectively to each of the 5 underwater objects by measuring and averaging pulse transmission time from each. from one transceiver to another. Distance measuring system according to claim 1 or 2, characterized in that it comprises means for binding the location 10 of the device in question to a geodetic reference coordinate system (220). Distance measuring system according to claim 1 or 2, characterized in that it comprises a difference calculation system (59) for measuring the spatial difference of the first transceiver (20) and the associated electronic system, and the second transceiver (20) and the associated electronic system. sometimes using an intermediate transceiver (20) and associated electronic system. A method for determining acoustic distances between a plurality of private underwater hydroacoustic transceiver locations 20 in a plurality of underwater objects (236A-236J) having hydroacoustic transducers (20) adapted for underwater transmitting and receiving acoustic pulses, characterized in that it comprises steps : processing hydroacoustic pulse energy, which is a predetermined 25 characteristic area, to detect the hydroacoustic pulses received by the converters (20), which is a reference time synchronization clock to provide pulse arrival times for the transducer (20) received pulses having a predetermined range, synthesizing and transmitting pulses in the characteristic domain from said converter (20), synchronizing the operation of said transducers (20) with local pulses to schedule transmission and reception events, 29 103612 storing a private pulse transmission event scheduler for each converter (20) and triggering the local stored event transmission and reception events according to the scheduling scheme. Method according to claim 9, characterized in that it further comprises the steps of measuring the difference in the position of two transducers (20) in the water by timing the transmission times of the hydroacoustic pulses from each transducer relative to each other and averaging the two transmission times. The method of claim 9, further comprising digitally sampling the hydroacoustic pulses received in steps to determine the waveform shape and timing the arrival of the predetermined shape pulses with respect to the digital sampling time. A method according to claim 9, characterized in that it comprises the step of comparing the locations of the transducers with the geodetic reference coordinate system (220, 224). The method of claim 9, further comprising the step of transmitting hydroacoustic pulses at a predetermined carrier frequency of greater than about 40 kHz and the step of filtering out the frequency components of the received pulses above 40 kHz to the received hydroacoustic noise to suppress. 25 103612 30