ES2565822T3 - Transmission device, transmission method, reception device and reception method - Google Patents

Abstract

A transmission device for Single-Input, Single-Output, SISO, Multiple-Input, Single-Output, MISO, and Multiple-Input, Multiple-Output, MIMO, comprising the transmission device: a signal generator (7608) that generates: a control signal that includes a first signal that indicates at least one transmission scheme to be used to transmit flow data through physical layer pipes, PLP, and a second signal indicating a transmission scheme to be used to transmit the signal of control; a signal processor (7612) that generates data symbols by modulating the flow data and that generates a control symbol by modulating the control signal; a frame configurator (7610) that configures at least one SISO frame and at least one MISO / MIMO frame; a pilot symbol inserter that inserts pilot symbols in each SISO frame and in each MISO / MIMO frame; a preamble inserter (7622) that generates a preamble that carries the second signal and that inserts the preamble in each SISO frame and in each MIMO / MISO frame; and a transmitter (7262) that performs transmissions; wherein the frame configurator (7610) is adapted to configure the at least one SISO frame from the control symbol and the data symbols resulting from the modulation of data for SISO in the flow data and the at least one frame of MISO / MIMO from the control symbol and the data symbols resulting from the modulation of any or both data for MISO and data for MIMO in the flow data; the pilot symbol inserter is adapted to insert pilot symbols in each MISO / MIMO frame in which common pattern pilots are inserted into data for MISO and data for MIMO; The transmitter (7262) can be operated to switch between a first transmission process to transmit the at least one SISO frame and a second transmission process to transmit the at least one MISO / MIMO frame; the second signal identifies the transmission scheme of the control signal from among a plurality of schemes that include at least SISO, MISO and MIMO; and the first signal identifies, for the PLPs in each SISO frame and in each MISO / MIMO frame, if a transmission scheme for flow data transmitted by each PLP is SISO, MISO or MIMO.

Description

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DESCRIPTION

Transmission device, transmission method, reception device and reception method Field of the invention

The present invention relates to a transmission device and a reception device for communication using multiple antennas.

Conventional technology allows a transmission device to enable communications in systems (for example, Non-patent Bibliography 14) SISO (Single-Input, Single-Output) and MISO (Multiple-Input, Single-Output).

[List of quotes]

[Patent bibliography]

[Patent bibliography 1]

International Patent Application Publication No. WO2005 / 050885 [Patent Bibliography 2]

International Patent Application Publication No. WO 2011/043618 discloses a transmission device that creates a frame according to at least one of the MIMO, MISO and SISO schemes.

[Non-patent bibliography]

[Non-patent bibliography 1]

“Achieving near-capacity on a multiple-antenna channel” IEEE Transaction on communications, vol. 51, n ° 3, pag. 389-399, March 2003

[Non-patent bibliography 2]

“Performance analysis and design optimization of LDPC-coded MIMO OFDM systems” IEEE Trans. Signal Processing, vol. 52, n ° 2, pag. 348-361, February 2004

[Non-patent bibliography 3]

“BER performance evaluation in 2x2 MIMO spatial multiplexing systems under Rician fading channels” IEICE Trans. Fundamentals, vol. E91-A, No. 10, p. 2798-2807, October 2008

[Non-patent bibliography 4]

“Turbo space-time codes with time varying linear transformations” IEEE Trans. Wireless communications, vol. 6, n ° 2, pag. 486-493, February 2007

[Non-patent bibliography 5]

“Likelihood function for QR-MLD suitable for soft-decision turbo decoding and its performance” IEICE Trans. Commun., Vol. E88-B, No. 1, p. 47-57, January 2004

[Non-patent bibliography 6]

“A tutorial on 'Parallel concatenated (Turbo) coding', 'Turbo (iterative) decoding' and related topics” IEICE, Technical Report IT98-51

[Non-patent bibliography 7]

"Advanced signal processing for PLCs: Wavelet-OFDM" Proc. of IEEE International symposium on ISPLC 2008, pag. 187-192, 2008

[Non-patent bibliography 8]

D. J. Love and R. W. Heath Jr., “Limited feedback unitary precoding for spatial multiplexing systems” IEEE Trans. Inf. Theory, vol. 51, No. 8, p. 2967-2976, August 2005

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[Non-patent bibliography 9]

Document DVB A122, Framing structure, channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2), June 2008

[Non-patent bibliography 10]

L. Vangelista, N. Benvenuto, and S. Tomasin “Key technologies for next-generation digital terrestrial television standard DVB-T2,” IEEE Commun. Magazine, vol. 47, n ° 10, pag. 146-153, October 2009

[Non-patent bibliography 11]

T. Ohgane, T. Nishimura, and Y. Ogawa, “Applications of space division multiplexing and those performance in a MIMO channel” IEICE Trans. Commun., Vol. E88-B, No. 5, p. 1843-1851, May 2005

[Non-patent bibliography 12]

R. G. Gallager “Low-density parity-check codes,” IRE Trans. Inform. Theory, IT-8, pag. 21-28, 1962 [Non-patent bibliography 13]

D. J. C. Mackay, “Good error-correcting codes based on very sparse matrices,” IEEE Trans. Inform. Theory, vol. 45, No. 2, p. 399-431, March 1999.

[Non-patent bibliography 14]

ETSI EN 302 307, “Second generation framing structure, channel coding and modulation systems for broadcasting, interactive services, news gathering and other broadband satellite applications” v.1.1.2, June 2006

[Non-patent bibliography 15]

Y.-L. Ueng, and C.-C. Cheng “A fast-convergence decoding method and memory-efficient VLSI decoder architecture for irregular LDPC codes in the IEEE 802.16e standards” IEEE VTC-2007 Fall, pag. 1255-1259

[Non-patent bibliography 16]

S. M. Alamouti “A simple transmit diversity technique for wireless communications” IEEE J. Select. Areas Commun., Vol.16, n ° 8, pag. 1451-1458, October 1998

[Non-patent bibliography 17]

V. Tarokh, H. Jafrkhani, and A. R. Calderbank “Space-time block coding for wireless communications: Performance results” IEEE J. Select. Areas Commun., Vol.17, n ° 3, n ° 3, pag. 451-460, March 1999

[Non-patent bibliography 16]

“Digital Video Braodcasting (DVB) M Frame structure channel coding and modulation for a second generation digital terrestrial television broadcasting system (DVB-T2)”, European Standard (Telecommunications Series), European Telecommunications Standards Institute (ETSI), 650, Route Des Lucioles , F-06921 Sophia-Antipolis; France n ° V1.1.1, July 1, 2009

The present invention is intended to provide a frame configuration that allows, when used in transmission signals that switch between SISO and MISO / MIMO, easy detection of the signals on the receiver side.

This is achieved by the characteristics of the independent claims.

As before, the present invention provides a transmission method, reception method, transmission device and reception device that allow each side of the receiver to easily detect signals when the signals are transmitted using SISO, MISO and MIMO.

Figure 1 illustrates an example of a transmission and reception device in a spatial multiplexing MIMO system.

Figure 2 illustrates a sample frame configuration.

Figure 3 illustrates an example of a transmission device that applies a phase change method.

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Figure 4 illustrates another example of a transmission device that applies a phase change method.

Figure 5 illustrates another sample frame configuration.

Figure 6 illustrates another method of changing the sample phase.

Figure 7 illustrates a sample configuration of a receiving device.

Figure 8 illustrates a sample configuration of a signal processor in the receiving device.

Figure 9 illustrates another sample configuration of a signal processor in the receiving device.

Figure 10 illustrates an iterative decoding method.

Figure 11 illustrates sample reception conditions.

Figure 12 illustrates a further example of a transmission device that applies a phase change method.

Figure 13 illustrates a further example of a transmission device that applies a phase change method.

Figure 14 illustrates another sample frame configuration.

Figures 15A and 15B illustrate another sample frame configuration.

Figures 16A and 16B illustrate another sample frame configuration.

Figures 17A and 17B illustrate another sample frame configuration.

Figures 18A and 18B illustrate another sample frame configuration.

Figures 19A and 19B illustrate examples of a mapping method.

Figures 20A and 20B illustrate additional examples of a mapping method.

Figure 21 illustrates a sample configuration of a weighting unit.

Figure 22 illustrates a method of reorganizing the sample symbol.

Figure 23 illustrates another example of a transmission and reception device in a spatial multiplexing MIMO system.

Figures 24A and 24B illustrate sample BER characteristics.

Figure 25 illustrates another method of changing the sample phase.

Figure 26 illustrates another method of changing the sample phase.

Figure 27 illustrates another method of changing the sample phase.

Figure 28 illustrates another method of changing the sample phase.

Figure 29 illustrates another method of changing the sample phase.

Figure 30 illustrates a sample symbol arrangement for a modulated signal that provides high quality of received signal.

Figure 31 illustrates a sample frame configuration for a modulated signal that provides high quality of received signal.

Figure 32 illustrates a sample symbol arrangement for a modulated signal that provides high quality of received signal.

Figure 33 illustrates a sample symbol arrangement for a modulated signal that provides high quality of received signal.

Figure 34 illustrates a variation in numbers of symbols and intervals required per pair of coded blocks when block codes are used.

Figure 35 illustrates another variation in numbers of symbols and intervals required per pair of coded blocks when block codes are used.

Figure 36 illustrates a global configuration of a digital broadcast system.

Figure 37 is a block diagram illustrating a sample receiver.

Figure 38 illustrates multiplexed data configuration.

Figure 39 is a schematic diagram illustrating multiplexing of data encoded in flows.

Figure 40 is a detailed diagram illustrating a video stream according to this content in a PES packet sequence.

Figure 41 is a structural diagram of TS packages and source packages in multiplexed data.

Figure 42 illustrates configuration of PMT data.

Figure 43 illustrates information as configured in multiplexed data.

Figure 44 illustrates the configuration of flow attribute information.

Figure 45 illustrates the configuration of a video presentation and audio output device.

Figure 46 illustrates a sample configuration of a communications system.

Figures 47A and 47B illustrate sample symbol arrangements for a modulated signal that provides high quality of received signal.

Figures 48A and 48B illustrate sample symbol arrangements for a modulated signal that provides high quality of received signal.

Figures 49A and 49B illustrate sample symbol arrangements for a modulated signal that provides high quality of received signal.

Figures 50A and 50B illustrate sample symbol arrangements for a modulated signal that provides high quality of received signal.

Figure 51 illustrates a sample configuration of a transmission device.

Figure 52 illustrates another sample configuration of a transmission device.

Figure 53 illustrates an additional sample configuration of a transmission device.

Figure 54 illustrates an additional sample configuration of more than one transmission device.

Figure 55 illustrates a baseband signal changer.

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Figure 56 illustrates an additional sample configuration of even more than one transmission device.

Figure 57 illustrates sample operations of a distributor.

Figure 58 illustrates additional sample operations of a distributor.

Figure 59 illustrates a sample communications system that indicates the relationship between base stations and terminals.

Figure 60 illustrates an example of the transmission signal frequency assignment.

Figure 61 illustrates another example of assigning the transmission signal frequency.

Figure 62 illustrates a sample communications system that indicates the relationship between a base station, repeaters and terminals.

Figure 63 illustrates an example of assigning the transmission signal frequency with respect to the base station.

Figure 64 illustrates an example of assigning the transmission signal frequency with respect to the repeaters.

Figure 65 illustrates a sample configuration of a receiver and a transmitter in the repeater.

Figure 66 illustrates a signal data format used for transmission through the base station.

Figure 67 illustrates another sample configuration even more of a transmission device.

Figure 68 illustrates another baseband signal changer.

Figure 69 illustrates a sample weighting, baseband signal change and phase change method.

Figure 70 illustrates a sample configuration of a transmission device using an OFDM method. Figures 71A and 71B illustrate another sample frame configuration.

Figure 72 further illustrates the numbers of intervals and phase change values that correspond to a modulation method.

Figure 73 further illustrates the numbers of intervals and phase change values that correspond to a modulation method.

Figure 74 illustrates the global frame configuration of a signal transmitted by a diffuser using DVB-12.

Figure 75 illustrates two or more types of signals in the same time indication.

Figure 76 illustrates an additional sample configuration of more than one transmission device.

Figure 77 illustrates an alternative sample frame configuration.

Figure 78 illustrates another alternative sample frame configuration.

Figure 79 illustrates an additional alternative sample frame configuration.

Figure 80 illustrates a further additional alternative sample frame configuration.

Figure 81 illustrates yet another alternative sample frame configuration.

Figure 82 illustrates yet another alternative sample frame configuration.

Figure 83 illustrates a further additional alternative sample frame configuration.

Figure 84 further illustrates two or more types of signals in the same time indication.

Figure 85 illustrates an alternative sample configuration of a transmission device.

Figure 86 illustrates an alternative sample configuration of a receiving device.

Figure 87 illustrates another alternative sample configuration of a receiving device.

Figure 88 illustrates another alternative sample configuration more than one receiving device.

Figures 89A and 89B illustrate additional alternative sample frame configurations.

Figures 90A and 90B illustrate further additional alternative sample frame configurations.

Figures 91A and 91B illustrate more alternative sample frame configurations.

Figures 92A and 92B illustrate even more alternative sample frame configurations.

Figures 93A and 93B illustrate additional alternative sample frame configurations.

Figure 94 illustrates a sample frame configuration used when using space-time block codes.

Figure 95 illustrates an example of signal point distribution for 16-QAM in the I-Q plane.

Figure 96 indicates a sample configuration for a signal generator when cyclic Q delay is applied.

Figure 97 illustrates a first example of a generation method for s1 (t) and s2 (t) when Q delay is used

clical

Figure 98 indicates a sample configuration for a signal generator when cyclic Q delay is applied.

Figure 99 indicates a sample configuration for a signal generator when cyclic Q delay is applied.

Figure 100 illustrates a second example of a generation method for s1 (t) and s2 (t) when using Q-delay.

Figure 101 indicates a sample configuration for a signal generator when cyclic Q delay is applied.

Figure 102 indicates a sample configuration for a signal generator when cyclic Q delay is applied.

Figure 103A indicates restrictions pertaining to the transmission of a single antenna and multiple antennas in the DVB-T2 standard, while Figure 103B indicates a desirable future standard.

Figure 104 indicates a sample sub-frame configuration based on the transmission antenna configuration.

Figure 105 indicates a sample sub-frame configuration based on the transmission antenna configuration.

Figure 106 indicates the transmission frame configuration.

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Figure 107 illustrates an example of an SP pilot for a sub-frame start symbol and a sub-frame close symbol.

Figure 108A illustrates a real DVB-T2 service network (SISO).

Figure 108B illustrates a distributed MISO system that employs an existing transmission antenna.

Figure 108C illustrates a co-located MIMO configuration.

Figure 108D illustrates a configuration in which MISO distributed and MIMO co-located are combined.

Figure 109 indicates an example of sub-frame configuration based on the transmission antenna configuration (taking into account polarization).

Figure 110 indicates the transmission frame configuration.

Figure 111 indicates an example of sub-frame configuration based on the transmission antenna configuration (taking into account the transmission power).

Figure 112 indicates the transmission frame configuration.

Figure 113 indicates an example of sub-frame configuration based on the transmission antenna configuration (taking into account polarization and transmission power).

Figure 114 indicates the transmission frame configuration.

Figure 115 indicates a sample sub-frame configuration based on the transmission antenna configuration.

Figure 116 indicates a sample sub-frame configuration (an appropriate sub-frame order) based on the transmission antenna configuration.

Figure 117 indicates a sample sub-frame configuration (an appropriate sub-frame order) based on the transmission antenna configuration.

Figure 118 indicates the transmission frame configuration.

Figure 119 indicates an example of sub-frame configuration based on the transmission antenna configuration (taking into account polarization).

Figure 120 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account polarization) based on the transmission antenna configuration.

Figure 121 indicates the transmission frame configuration.

Figure 122 illustrates an example of a transmission power change pattern for SISO and MISO / MIMO.

Figure 123 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the transmission power change pattern) based on the transmission antenna configuration. Figure 124 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the transmission power change pattern) based on the transmission antenna configuration. Figure 125 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the transmission power change pattern) based on the transmission antenna configuration. Figure 126 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the transmission power change pattern) based on the transmission antenna configuration. Figure 127 indicates the transmission frame configuration.

Figure 128 illustrates a sample transmission power change pattern (taking into account polarization) for SISO and MiSO / MIMO.

Figure 129 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the pattern of transmission power change and polarization) based on the transmission antenna configuration.

Figure 130 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the transmission power change pattern and polarization) based on the transmission antenna configuration.

Figure 131 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the transmission power change pattern and polarization) based on the transmission antenna configuration.

Figure 132 indicates a sample sub-frame configuration (an appropriate sub-frame order, taking into account the transmission power change pattern and polarization) based on the transmission antenna configuration.

Figure 133 indicates the transmission frame configuration.

Figure 134 indicates the transmission frame configuration.

Figure 135 indicates the transmission frame configuration.

Figure 136 indicates the transmission frame configuration.

Figure 137 indicates the transmission frame configuration.

Figure 138 indicates the transmission frame configuration.

Figure 139 indicates the transmission frame configuration.

Figure 140 indicates the transmission frame configuration.

Figure 141 indicates the transmission frame configuration.

Figure 142A indicates control information S1, and Figure 142B indicates control information belonging to the sub-frame.

Figure 143 indicates control information pertaining to the sub-frame.

Figure 144 indicates the transmission frame configuration.

Figure 145A indicates signaling data L1, and Figure 145B indicates control information S1.

Figure 146 indicates the transmission frame configuration.

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Figure 147A indicates signaling data L1, and Figure 147B indicates control information S1.

Figure 148A indicates the transmission frame configuration.

Figure 148B indicates the transmission frame configuration.

Figure 149A indicates L1 signaling data in portion (a) and sub-frame control information in the

portion (b).

Figure 149B indicates control information S1.

Figure 150A indicates the transmission frame configuration.

Figure 150B indicates the transmission frame configuration.

Figure 151A indicates signaling data L1, and Figure 151B indicates control information S1.

Figure 152 indicates control information pertaining to an AGC synchronization preamble.

(Discoveries of the inventor)

MIMO (Multiple-Input, Multiple-Output) is an example of a conventional communication system that uses multiple antennas. In communication of multiple antennas, of which MIMO is typical, each of the multiple transmission signals is modulated, and each modulated signal is transmitted simultaneously from a different antenna to increase the speed of data transmission.

Figure 23 illustrates a sample configuration of a transmission and reception device that has two transmission antennas and two reception antennas, and which uses two modulated transmission signals (transmission streams). In the transmission device, encoded data is interleaved, the interleaved data is modulated, and frequency conversion and the like are performed to generate transmission signals, which are then transmitted from the antennas. In this case, the scheme to simultaneously transmit different modulated signals from different transmission antennas at the same time indication and on a common frequency is MIMO of spatial multiplexing.

In this context, Patent Bibliography 1 suggests using a transmission device provided with a different intercalation pattern for each transmission antenna. That is, the transmission device from Figure 23 should use two different interleaving patterns made by two interleavers (pa and pb). As for the receiving device, Non-patent Bibliography 1 and Non-Patent Bibliography 2 describe improving quality reception by iteratively using flexible values for the detection method (by means of the MIMO detector of Figure 23).

As usual, real propagation environment models in wireless communications include those of NLOS (Non-Line-of-Vision), characterized by a Rayleigh fading environment, and those of lOs (Line-of-Vision), characterized by a fading environment of Rician. When the transmission device transmits a single modulated signal, and the reception device performs maximum ratio combination on the signals received by a plurality of antennas and then demodulates and decodes the resulting signals, excellent reception quality can be achieved in a LOS environment. , particularly in an environment where Rician's factor is large. The Rician factor represents the power received from direct waves relative to the power received from scattered waves. However, depending on the transmission system (for example, a MIMO system of spatial multiplexing), there is a problem where reception quality deteriorates as Rician's factor increases (see Non-patent Bibliography 3).

Figures 24A and 24B illustrate an example of simulation results of the characteristics (vertical axis: BER, horizontal axis: SNR (signal to noise ratio) for data encoded with LDPC codes (low density parity check) of BER ( Bit Error Rate) and transmitted through a 2x2 spatial multiplexing MIMO system (two transmitting antennas, two receiving antennas) in a Rayleigh fade environment and a Rician fade environment with factors of Rician of K = 3, 10 and 16 dB Figure 24A provides the BER characteristics of logarithmic probability relationship based on the Max-Log approximation (i.e., Max-log APP, where APP is the posterior probability) without detection iterative phase (see Bibliography no of patent 1 and Bibliography non-patent 2), while Figure 24B provides the BER feature of Max-log APP with iterative phase detection (see Bibliografla no of patent 1 and Bibliogr patent number 2) (number of iterations: five). Figures 24A and 24B clearly indicate that, regardless of whether or not iterative phase detection is performed, reception quality degrades in the MIMO system of spatial multiplexing as the Rician factor increases. Therefore, the problem of reception quality degradation after stabilization of the propagation environment in the MIMO system of spatial multiplexing, which does not take place in a conventional single modulation signal system, is unique to the MIMO system of spatial multiplexing

Broadcast or multicast communication is a service that must be applied to various propagation environments. The radio wave propagation environment between the diffuser and the receivers that belong to the users is often a LOS environment. When a spatial multiplexing MIMO system is used that has the above problem for broadcast or multicast communication, a situation may occur in which the intensity of the received electric field is high in the receiving device, but in which the degradation in reception quality makes reception of service impossible. In other words, to use a MIMO system of

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Spatial multiplexing in broadcast or multicast communication in both an NLOS and LOS environment, a MIMO system that offers a certain degree of reception quality is desirable.

Non-patent Bibliography 8 describes a method for selecting a code book used in precoding (ie, a precoding matrix, also referred to as a precoding weighting matrix) based on feedback information from a communication part. However, the non-patent Bibliography 8 does not disclose at all a method for precoding in an environment in which no feedback information can be obtained from the other party, such as in the previous broadcast or multicast communication.

On the other hand, the non-patent Bibliography 4 discloses a method for changing the precoding matrix over time. This method is applicable when feedback information is not available. Non-patent Bibliography 4 discloses a unitary matrix as the precoding matrix, and changing the unitary matrix randomly, but does not disclose at all a method applicable to the degradation of reception quality in the environment of LOS described above. Non-patent Bibliography 4 simply relates jumping between precoding matrices randomly. Obviously, the non-patent Bibliography 4 makes no mention of a precoding method, or a structure of a precoding matrix, to remedy the degradation of reception quality in an LOS environment.

An objective of the present invention is to provide a MIMO system that improves reception quality in a LOS environment.

The embodiments of the present invention are described below with reference to the accompanying drawings. [Embodiment 1]

The following describes, in detail, a transmission method, a transmission device, a reception method, and a reception device belonging to the present embodiment.

Before beginning the appropriate description, a summary of transmission schemes and decoding schemes in a conventional spatial multiplexing MIMO system is provided.

Figure 1 illustrates the structure of a NtxNr spatial multiplexing MIMO system. An information vector z is encoded and interleaved. The vector of coded bits u = (u1, ..., uNt) is obtained as the interleaving output. At this point, ui = (un, .... uiM) (where M is the number of bits transmitted per symbol). For a transmission vector s = (s1, ..., SNt), a received serial is found if = map (ui) for the transmission antenna # i. Normalizing the transmission energy, this is expressible as E {| si | 2} = Es / Nt (where Es is the total energy per channel). The reception vector y = (y1, ... yNr) T is expressed in the Calculation. 1 (formula 1), below.

[Calculation 1]

'- (v-O'aJ

= s + n (formula 1)

At this point, HNtNr is the channel matrix, n = (m, ..., nNr) is the noise vector, and the average value of ni is zero for independent and identically distributed complex Gaussian noise (iid) of variance o2 . Based on the relationships between transmitted symbols introduced in a receiver and received symbols, the probability distribution of the received vectors can be expressed as Calculation 2 (formula 2), below, for a multi-dimensional Gaussian distribution.

[Calculation 2]

image 1

At this point, it is considered a receiver that performs iterative decoding. A receiver of this type is illustrated in Figure 1 as being composed of an external flexible input / flexible output decoder and a MIMO detector. The log likelihood ratio vector (L value) for Figure 1 is provided by Calculation 3 (formula 3) through Calculation 5 (formula 5), as follows.

[Calculation 3]

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£ (u) = (£ («,),

(formula 3)

image2

[Calculation 5]

image3

(Iterative Detection Method)

The following describes the iterative MIMO signal detection performed using the NtxNr spatial multiplexing MIMO system. The logarithmic probability ratio of umn is defined by Calculation 6 (formula 6).

[Calculation 6]

image4

Through the application of Bayes' theorem, Calculus 6 (formula 6) can be expressed as Calculus 7 (formula 7).

[Calculation 7]

image5

Note that Umn, ± i = {u | umn = ± 1}. Through the lnXaj ~ max ln aj approximation, Calculation 7 (formula 7) can approximate as Calculation 8 (formula 8). The symbol ~ is used herein to mean approximation.

[Calculation 8]

image6

In Calculus 8 (formula 8), P (u | umn) and In P (u | umn) can be expressed as follows. [Calculation 9]

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image7

[Calculation 10]

[Calculation 11]

image8

image9

Note that the logarithmic probability of the equation provided in the Calculation. 2 (formula 2) can be expressed as Calculation 12 (formula 12).

[Calculation 12]

image10

Therefore, given Calculation 7 (formula 7) and Calculation 13 (formula 13), the subsequent L value for MAP or APP (a posteriori probability) can be expressed as follows.

[Calculation 13]

, «P '- ^ || y-Hs (u) | f + i /« ^ y

! y) = in ------: --------------------. -----------'------ --J

Zu

This is hereinafter referred to as iterative APP decoding. Also, given Calculus 8 (formula 8) and Calculus 12 (formula 12), the subsequent L value for the Max-log APP can be expressed as follows.

[Calculation 14]

(formula 13)

1 y) = KM “.y.M«>)} - sMu, y, K »))}

[Calculation 15]

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image11

This is hereafter referred to as iterative Max-log APP decoding. As such, the external information required by the iterative decoding system is obtainable by subtracting the previous entry from Calculation 13 (formula 13) or from Calculation 14 (formula 14).

(System model)

Figure 23 illustrates the basic configuration of a system related to the following explanations. The illustrated system is a 2x2 spatial multiplexing MIMO system that has an outer decoder for each of the two streams A and B. The two outer decoders perform identical LDPC coding. (Although the present example considers a configuration in which the external encoders use LDPC codes, the external encoders are not restricted to the use of LDPC as the error correction codes. The example can also be performed using other error correction codes, such as turbo codes, convolutional codes or convolutional codes of LDPC In addition, while the external encoders are currently described as individually configured for each transmission antenna, no limitation is intended in this regard.A unique exterior encoder can be used for a plurality of transmission antennas, or the number of external encoders may be greater than the number of transmission antennas). The system also has interleavers (Pa, Pb) for each of the A and B streams. At this point, the modulation method is 2h-QAM (i.e. h bits transmitted by symbol).

The receiver performs iterative detection (decoding APP (or Max-log APP) iterative) of MIMO signals, as described above. LDPC codes are decoded using, for example, sum-product decoding.

Figure 2 illustrates the frame configuration and describes the symbol order after the intercalation. At this point, (iaja) and (ibjb) can be expressed as follows.

[Calculation 16]

[Calculation 17]

(ib’j) = xb (Q? lbJl)

(formula 16)

(formula 17)

At this point, ia and ib represent the order of the symbol after the interleaving, ja and jb represent the bit position in the modulation method (where jajb = 1, ... h), Pa and Pb represent the interleavers of the flows A and B, and Wa iaja and Wb ibjb represent the order of data of flows A and B before interleaving. Note that Figure 2 illustrates a situation where ia = ib.

(Iterative decoding)

The following describes, in detail, the sum-product decoding used when decoding the LDPC codes and the MIMO signal iterative detection algorithm, both used by the receiver.

Sum-product decoding

A two-dimensional MxN matrix H = {Hmn} is used as the test matrix for LDPC codes undergoing decoding. For the set [1, N] = {1, 2 ... N}, the partial sets A (m) and B (n) are defined as follows. [Calculation 18]

image12

[Calculation 19]

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image13

At this point, A (m) means the set of Column Indices equal to 1 for row m of the check matrix H, while B (n) means the set of Row Indices equal to 1 for row n of the test matrix H. The sum-product decoding algorithm is as follows.

Stage A-1 (Initialization): for all pairs (m, n) that satisfy Hmn = 1, establish the previous logarithmic relationship bmn = 0. Set the loop variable (number of iterations) lsum = 1, and set the number maximum loops l sum, max.

Stage A-2 (Processing): for all pairs (m, n) that satisfy Hmn = 1 in the order m = 1, 2, ... M, update the log relation of extrinsic value amn using the following update formula .

[Calculation 20]

image14

[Calculation 21]

sign (x) =

image15

x> 0 x <0

(formula 21)

[Calculation 22]

f (x) = lnEm ± l

exp (x) ~ l

(formula 22)

where f is the Gallager function. It can then be calculated as follows.

Stage A-3 (Column operations): for all pairs (m, n) that satisfy Hmn = 1 in the order n = 1, 2, ... N, update the logarithmic relationship of extrinsic value bmn using the following formula of update.

[Calculation 23]

image16

Stage A-4 (Calculation of log probability ratio): For ne [1, N], the log probability ratio Ln is calculated as follows.

[Calculation 24]

image17

Stage A-5 (Iteration count): If lsum <lsum, max, then lsum is incremented and the process returns to stage A-2. The sum-product decoding ends when l sum = lsum, max.

The above describes an iteration of the sum-product decoding operations. Subsequently, iterative MIMO signal detection is performed. The variables m, n, amn, bmn, In and Ln used in the previous explanation of the sum-product decoding operations are expressed as ma, na, aamana, bamana, Ina and Lna for flow A and as mb, nb , abmbnb, Pbmbnb, Inb and Lnb for flow B.

(Iterative MIMO signal detection)

The following describes the calculation of ln for iterative MIMO signal detection. The following formula can be obtained from Calculation 1 (formula 1).

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image18

Given the plot configuration illustrated in Figure 2, the following functions can be obtained from Calculation 16 (formula 16) and Calculation 17 (formula 17).

[Calculation 26]

image19

image20

(formula 26)

[Calculation 27]

image21

where na, nb e [1, N]. For the iteration k of MIMO signal iterative detection, the variables Ina, Lna, Inb and Lnb are

express as 1k, na, Lk, na, lK, nb and Lk, nb.

Stage B-1 (Initial detection; k = 0) For the initial wave detection, 1o, na and 10, nb are calculated as follows. For iterative APP decoding:

[Calculation 28]

image22

For Max-log APP decoding: [Calculation 29]

image23

max Mu (^), and (/ *))} - max Mu (^), and (^))}

(formula 29)

[Calculation 30]

'*' (• '(iAy <A)) = - r ^ || and (iA') - H220x) s (u (1>) |

(formula 30)

where X = a, b. Next, the iteration count for the iterative MIMO signal detection is set to a minimum = 0, the maximum iteration count being a minimum, max.

Stage B-2 (Iterative detection; Iteration k): when the iteration count is k, Calculus 11 (formula 11), Calculus 13 (formula 13) through Calculus 15 (formula 15), Calculus 16 (formula 16), and Calculation 17 (formula 17) can be expressed as Calculation 31 (formula 31) through Calculation 34 (formula 34), below. Note that (X, Y) = (a, b) (b, a).

For iterative APP decoding:

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image24

[Calculation 32]

image25

(formula 32)

For iterative Max-log APP decoding:

[Calculation 33]

(formula 33)

[Calculation 34]

'•' («<;, * uQ'j) - -2? || and (iA -) - H22 (iA-) s (u (iA-» go +

(formula 34)

Stage B-3 (Iteration count and code word estimation) If minimum <maximum, then maximum is increased

and the process returns to stage B-2. When I

follow.

[Calculation 35]

image26

where X = a, b.

Figure 3 shows a sample configuration of a transmission device 300 belonging to the present embodiment. An encoder 302A takes information (data) 301A and a frame configuration signal 313 as input (which includes the error correction method, coding rate, block length and other information used by the encoder 302A in error correction coding of the data, so that the designated method is used by the frame configuration signal 313. The error correction method can be changed). According to the frame configuration signal 313, the encoder 302A performs error correction coding, such as convolutional coding, LDPC coding, turbo coding or the like and issues the encoded data 303A.

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An interleaver 304A takes the encoded data 303A and the frame configuration signal 313 as input, performs intercalation, that is, reorganizes the order thereof, and then issues the interleaved data 305A.

(Depending on the frame configuration signal 313, the interleaving method may be changed.)

A mapper 306A takes the interleaved data 305A and the frame configuration signal 313 as input and performs modulation, such as (Quadrature Phase Shift Modulation), 16-QAM (Quadrature Amplitude Modulation of 16) or 64-QAM (Quadrature Amplitude Modulation of 64) in them, then emits a baseband signal 307A. (Depending on the frame configuration signal 313, the modulation method can be changed).

Figures 19A and 19B illustrate an example of a QPSK modulation mapping method for a baseband signal composed of a phase I component and a quadrature component Q in the IQ plane. For example, as shown in Figure 19A, when the input data is 00, then the output is I = 1.0, Q = 1.0. Similarly, when the input data is 01, the output is I = -1.0, Q = 1.0, and as! successively. Figure 19B illustrates an example of a QPSK modulation mapping method in the IQ plane that differs from Figure 19A in that the signal points of Figure 19A have been rotated around the origin to obtain the signal points of the Figure 19B Non-patent Bibliography 9 and Non-Patent Bibliography 10 describe a method of constellation rotation of this type. As an alternative, the Clical Q Delay described in the non-patent Bibliography 9 and in the non-patent Bibliography 10 can also be adopted. An alternative example, other than Figures 19A and 19B, is shown in Figures 20A and 20B, which illustrate a signal point distribution for 16-QAM in the IQ plane. The example in Figure 20A corresponds to Figure 19A, while that in Figure 20B corresponds to Figure 19B.

An encoder 302B takes information (data) 301B and the frame configuration signal 313 as input (which includes the error correction method, coding rate, block length and other information used by the encoder 302B in error correction coding of the data, so that the designated method is used by the frame configuration signal 313. The error correction method can be changed). According to frame configuration signal 313, encoder 302B performs error correction coding, such as convolutional coding, LDPC coding, turbo coding or the like, and issues the encoded data 303B.

An interleaver 304B takes the encoded data 303B and the frame configuration signal 313 as input, performs interleaving, that is, reorganizes the order thereof, and issues the interleaved data 305B.

(Depending on the frame configuration signal 313, the interleaving method may be changed.)

A mapper 306B takes the interleaved data 305B and the frame configuration signal 313 as input and performs modulation, such as QPSK, 16-QAM or 64-QAM therein, then issues a baseband signal 307B. (Depending on the frame configuration signal 313, the modulation method can be changed).

A signal processing method information generator 314 takes the frame configuration signal 313 as input and consequently issues the signal processing method information 315. The signal processing method information 315 designates the fixed precoding matrix to be used, and includes information about the phase change pattern used to change the phase.

A weighting unit 308A takes a baseband signal 307A, the baseband signal 307B and the signal processing method information 315 as input and, in accordance with the information of signal processing method 315, weights on the baseband signals 307A and 307B, then emits a weighted signal 309A. The weighting method is described in detail, later.

A wireless unit 310A takes the weighted signal 309A as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification and as! successively, next it emits the transmission signal 311A. The transmission signal 311A is then broadcast as radio waves by means of an antenna 312A.

A weighting unit 308B takes the baseband signal 307A, the baseband signal 307B and the signal processing method information 315 as input and, in accordance with the information of signal processing method 315, weights on the baseband signals 307A and 307B, then output the weighted signal 316B.

Figure 21 illustrates the configuration of the weighting units 308A and 308B. The area of Figure 21 enclosed in the dashed line represents one of the weighting units. The baseband signal 307A is multiplied by w11 to obtain w11s1 (t), and multiplied by w21 to obtain w21s1 (t). Similarly, baseband signal 307B is multiplied by w12 to obtain w12s2 (t), and multiplied by w22 to obtain w22s2 (t). TO

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Then you get z1 (t) = w11s1 (t) + w12s2 (t) and z2 (t) = w21s1 (t) + w22s22 (t). At this point, as explained in embodiment 1, s1 (t) and s2 (t) are the baseband signals modulated according to a modulation method such as BPSK (Binary Phase Shift Modulation), QPSK, 8 -PSK (Phase Shift Modulation of 8), 16-QAM, 32-QAM (Quadrature Amplitude Modulation of 32), 64-QAM, 256- QAM 16-APSK (16-Phase Shift and Amplitude Modulation) and so! successively.

Both weighting units perform weighting using a fixed precoding matrix. The precoding matrix uses, for example, the method of Calculation 36 (formula 36), and satisfies the conditions of Calculation 37 (formula 37) or Calculation 38 (formula 38), all found below. However, this is only an example. The value of a is not restricted to Calculation 37 (formula 37) and Calculation 38 (formula 38), and may take other values, for example, a = 1.

At this point, the precoding matrix is [Calculation 36]

'wt'l wl2'

^ vt> 21 w22j

image27

In Calculation 36 (formula 36) above, a is provided by: [Calculation 37]

(formula 36)

to-

V2 + 4

V2 + 2

(formula 37)

As an alternative, in the calculation. 36 (formula 36) above, can be provided by: [Calculation 38]

a =

V2 + 3 + V5 V2 + 3-V5

(formula 38)

The precoding matrix is not restricted to that of Calculation 36 (formula 36), but may also be as indicated by Calculation 39 (formula 39).

[Calculation 39]

image28

In Calculation 39 (formula 39), let a = Aejd11, b = Bejd12, c = Cejd21 and d = Dejd22. In addition, one of a, b, c and d can be zero. For example, the following configurations are possible: (1) a can be zero while b, c and d are non-zero, (2) b can be zero while a, c and d are non-zero, (3) c can be zero while a , b and d are nonzero, or (4) d can be zero while a, b and c are nonzero.

When any of the modulation method, error correction codes and the codification rate thereof is changed, the precoding matrix can also be established, changed and set for use.

A phase changer 317B takes the weighted signal 316B and the signal processing method information 315 as input, then regularly changes the phase of the signal 316B to emit. This regular change is a phase change made according to a predetermined phase change pattern that has a predetermined period (cycle) (for example, each n symbols (where n is an integer, n> 1) or at a predetermined interval).

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The details of the phase change pattern are explained below, in realization 4.

The wireless unit 310B takes the post-phase change signal 309B as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification and as! successively, next it emits the transmission signal 311B. The transmission signal 311B is then broadcast as radio waves by means of a 312B antenna.

Figure 4 illustrates a sample configuration of a transmission device 400 that differs from that of Figure 3. The points of difference of Figure 4 of Figure 3 are described below.

An encoder 402 takes information (data) 401 and the frame configuration signal 313 as input, and, in accordance with the frame configuration signal 313, performs error correction coding and issues encoded data 402.

A distributor 404 takes the encoded data 403 as input, distributes it, and issues the data 405A and the data 405B. Although Figure 4 illustrates only one encoder, the number of encoders is not limited as such. The present invention can also be carried out using m encoders (where m is an integer, m> 1) so that the distributor divides the encoded data created by each encoder into two groups for distribution.

Figure 5 illustrates an example of a time domain frame configuration for a transmission device according to the present embodiment. Symbol 500_1 is a symbol for notifying the reception device of the transmission scheme. For example, symbol 500_1 carries information such as the error correction method used to transmit data symbols, the coding rate thereof, and the modulation method used to transmit data symbols.

The symbol 501_1 is for estimating channel fluctuations for the modulated signal z1 (t) (where t is time) transmitted by the transmission device. Symbol 502_1 is a data symbol transmitted by the modulated signal z1 (t) as the symbol number u (in the time domain). Symbol 503_1 is a data symbol transmitted by the modulated signal z1 (t) as the symbol number u + 1.

The symbol 501_2 is for estimating channel fluctuations for the modulated signal z2 (t) (where t is time) transmitted by the transmission device. Symbol 502_2 is a data symbol transmitted by the modulated signal z2 (t) as the symbol number u. Symbol 503_2 is a data symbol transmitted by the modulated signal z1 (t) as the symbol number u + 1.

At this point, the z1 (t) and z2 (t) symbols that have the same time indication (identical timing) are transmitted from the transmitting antenna using the same (shared / common) frequency.

The following describes the relationships between the modulated signals z1 (t) and z2 (t) transmitted by the transmission device and the signals received r1 (t) and r2 (t) received by the receiving device.

In Figure 5, 504 No. 1 and 504 No. 2 indicate the transmitting antennas of the transmission device, while 505 No. 1 and 505 No. 2 indicate the receiving antennas of the receiving device. The transmission device transmits the modulated signal z1 (t) from the transmission antenna 504 # 1 and transmits the modulated signal z2 (t) from the transmission antenna 504 # 2. At this point, the modulated signals z1 (t) and z2 (t) are assumed to occupy the same (shared / common) frequency (bandwidth). The channel fluctuations in the transmitting antennas of the transmitting device and the antennas of the receiving device are hn (t), h12 (t), h21 (t) and h22 (t), respectively. Assuming that the receiving antenna 505 # 1 of the receiving device receives the received signal r1 (t) and that the receiving antenna 505 # 2 of the receiving device receives the received signal r2 (t), the following relation.

[Calculation 40]

image29

Figure 6 pertains to the weighting method (precoding method) and the phase change method of the present embodiment. A weighting unit 600 is a combined version of the weighting units 308A and 308B from Figure 3. As shown, the flow s1 (t) and the flow s2 (t) correspond to the baseband signals 307A and 307B of Figure 3. That is, the flows s1 (t) and s2 (t) are the baseband signals composed of a phase I component and a quadrature component Q according to the mapping by a modulation method such as QPSK , 16-QAM, and 64-QAM. As indicated by the frame configuration of Figure 6, the flow s1 (t) is represented as s1 (u) in the number of symbol u, as s1 (u + 1) in the number of symbol u + 1, and ace! successively. Similarly, the flow s2 (t) is represented as s2 (u) in the number of symbol u, as s2 (u + 1)

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in the number of symbol u + 1, and so! successively. The weighting unit 600 takes the baseband signals 307A (s1 (t)) and 307B (s2 (t)) as! As the signal processing method information 315 from Figure 3 as input, it performs weighting in accordance with the signal processing method information 315, and issues the weighted signals 309A (z1 (t)) and 316B (z2 '(t)) from Figure 3. Phase changer 317B changes the phase of the weighted signal 316B (z2' (t)) and issues the post-phase change signal 309B (z2 (t)).

At this point, given the vector W1 = (w11, w12) from the first row of the fixed precoding matrix F, z1 (t) can be expressed as Calculation 41 (formula 41), below.

[Calculation 41]

Zl (0 - Wlx (sl (f), (formula 41)

Similarly, given the vector W2 = (w21, w22) from the second row of the fixed precoding matrix F, and the phase change formula being applied by the phase changer by y (t), then z2 ( t) can be expressed as Calculation 42 (formula 42), below.

[Calculation 42]

z2 (t) - ^ (£) X-W2x (sl (f} ts2 (/)) (formula 42)

At this point, and (t) is a phase change formula that meets a predetermined method. For example, given a period (cycle) of four and time indication u, the phase change formula can be expressed as Calculation 43 (formula 43), below.

[Calculation 43]

and (li) - eJ (formula 43)

Similarly, the phase change formula for the time indication u + 1 may be, for example, as provided by Calculation 44 (formula 44).

[Calculation 44]

JC

and (u +1)

& (formula 44)

That is, the phase change formula for the time indication u + k is generalized in Calculation 45 (formula 45). [Calculation 45]

'.kx

and (u + k) £ (formula 45)

Note that Calculation 43 (formula 43) through Calculation 45 (formula 45) is provided only as an example of a regular phase change.

The regular phase change is not restricted to a period (cycle) of four. Improved reception capabilities (error correction capabilities, to be exact) can potentially be promoted in the receiving device by increasing the number (this does not mean that a longer period (cycle) is better, although avoiding small numbers such as two is equally ideal) of period (cycle).

Additionally, although Calculation 43 (formula 43) through Calculation 45 (formula 45), above, represents a configuration in which a change in phase is carried out through rotation by consecutive predetermined phases (in the above formula, every p / 2), the change in phase is not necessary to be the rotation by a constant amount, but can be random. For example, according to the predetermined period (cycle) of y (t), the phase can be changed through sequential multiplication as shown in Calculation 46 (formula 46) and in Calculation 47 (formula 47). The key point of the regular phase change is that the phase of the modulated signal is changed regularly. The rate of variance of the degree of phase change is preferably as even as possible, such as from -n radians to n radians. However, since this refers to a distribution, random variance is also possible.

[Calculation 46]

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(formula 46)

(formula 47)

As such, the weighting unit 600 of Figure 6 performs precoding using fixed predetermined precoded weights, and the phase changer 317B changes the phase of the signal introduced thereto while regularly varying the degree of phase change.

When a specialized precoding matrix is used in the LOS environment, reception quality is likely to improve greatly. However, depending on the direct wave conditions, the phase and amplitude components of the direct wave may differ considerably from the specialized precoding matrix, upon receipt. The LOS environment has certain rules. Therefore, the quality of data reception is greatly improved through a regular change of the transmission signal phase that complies with these rules. The present invention offers a method of signal processing to improve the LOS environment.

Figure 7 illustrates a sample configuration of a receiving device 700 belonging to the present embodiment. The wireless unit 703_X receives, as input, the received signal 702_X received by the antenna 701_X, performs processing such as frequency conversion, quadrature demodulation, and the like, and emits the baseband signal 704_X.

The channel fluctuation estimator 705_1 for the modulated signal z1 transmitted by the transmission device takes the baseband signal 704_X as input, extracts the reference symbol 501_1 for the channel estimation from Figure 5, estimates the value of hn from Calculation 40 (formula 40), and issues channel estimate signal 706_1.

The channel fluctuation estimator 705_2 for the modulated signal z2 transmitted by the transmission device takes the baseband signal 704_X as input, extracts the reference symbol 502_2 for the channel estimation from Figure 5, estimates the value of h12 from Calculation 40 (formula 40), and emits channel estimate signal 706_1.

The wireless unit 703_Y receives, as input, the received signal 702_Y received by the antenna 701_Y, performs processing such as frequency conversion, quadrature demodulation and the like, and emits the baseband signal 704_Y.

The channel fluctuation estimator 707_1 for the modulated signal z1 transmitted by the device

Transmission takes a baseband signal 704_Y as input, extracts the reference symbol 501_1 for the

channel estimate from Figure 5, estimates the value of hn from Calculation 40 (formula 40), and issues channel estimate signal 708_1.

The channel fluctuation estimator 707_2 for the modulated signal z2 transmitted by the device

transmission takes a baseband signal 704_Y as input, extracts the reference symbol 502_2 for the

channel estimate from Figure 5, estimates the value of hn from Calculation 40 (formula 40), and issues the channel estimate signal 708_2.

A control information decoder 709 receives the baseband signal 704_X and the baseband signal 704_Y as input, detects the symbol 500_1 indicating the transmission scheme from Figure 5, and issues a method information signal of 710 transmission for the transmission device.

A signal processor 711 takes the baseband signals 704_X and 704_Y, the channel estimation signals 706_1, 706_2, 708_1 and 708_2, and the transmission method information signal 710 as input, performs detection and decoding, and then issues the data received 712_1 and 712_2.

ej0 - »

.2x .3x Ax

j ~ a J-r J ~ r

e 5 - »e 5 -» e 5

6x

.7 n

, 8jt

Ax

J-r J ~ r J ~ r 7-

e -> e 5 -> e 5 -> e 5 ~> e 5

[Calculation 47]

, 7t_

e1 ej * - ¥ e 2 en ”-> eJj

, 3, 5Jt .Ik

j — 7t J ---] ---

4V '4V * 4

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Next, the operations of signal processor 711 are described in detail from Figure 7. Figure 8 illustrates a sample configuration of signal processor 711 belonging to the present embodiment. As shown, the signal processor 711 is mainly composed of an internal MIMO detector, a flexible input / flexible output decoder, and a coefficient generator. Non-patent Bibliography 2 and Non-Patent Bibliography 3 describe the method of iterative decoding with this structure. The MIMO system described in the non-patent Bibliography 2 and in the non-patent Bibliography 3 is a spatial multiplexing MIMO system, although the present embodiment differs from the non-patent Bibliography 2 and the non-patent Bibliography 3 when describing a MIMO system that regularly changes the phase over time, while using the precoding matrix. Taking the (channel) matrix H (t) of Calculation 36 (formula 36), then F is the precoding weighting matrix from Figure 6 (at this point, a fixed precoding matrix remains unchanged for a signal given given) and with the phase change formula used by the phase changer from Figure 6 Y (t) (at this point, Y (t) changes with time t), then the reception vector R ( t) = (r1 (t), r2 (t)) T and the flow vector S (t) = (s1 (t), s2 (t)) T the following function is deduced:

[Calculation 48]

where

image30

image31

fl

<9

0 ^

and (t) j

(formula 48)

At this point, the receiving device can use the decoding methods of Non-patent Bibliography 2 and 3 in R (t) by calculating H (t) xY (t) xF.

Accordingly, the coefficient generator 819 from Figure 8 takes a transmission method information signal 818 (corresponding to 710 from Figure 7) indicated by the transmission device (information to specify the precoding matrix fixed in use and the phase change pattern used when the phase is changed) and emits an 820 signal processing method information signal.

The internal MIMO detector 803 takes the signal processing method information signal 820 as input and performs iterative detection and decoding using the signal and the relation thereof to Calculation 48 (formula 48). Its operations are described below.

The processing unit illustrated in Figure 8 should use a processing method, as illustrated in Figure 10, to perform iterative decoding (iterative detection). First, the detection of a code word (or a frame) of the modulated signal (flow) s1 and a code word (or a frame) of the modulated signal (flow) s2 is performed. As a result, the flexible input / flexible output decoder obtains the log probability ratio of each bit of the code word (or frame) of the modulated signal (flow) s1 and the code word (or frame) of the signal modulated (flow) s2. Next, the logarithmic probability relationship is used to perform a second round of detection and decoding. These operations (referred to as iterative decoding (iterative detection)) are performed multiple times. The following explanations focus on the method of creating the logarithmic probability relationship of a symbol at a specific time in a frame.

In Figure 8, a memory 815 takes a baseband signal 801X (corresponding to the baseband signal 704_X from Figure 7), the group of 802X channel estimation signal (which corresponds to estimation signals of channel 706_1 and 706_2 from Figure 7), the baseband signal 801Y (corresponding to the baseband signal 704_Y from Figure 7), and the channel estimation group of channel 802Y (corresponding to channel estimation signals 708_1 and 708_2 from Figure 7) as input, execute (calculate) H (t) xY (t) xF from Calculation 48 (formula 48) to perform iterative decoding (iterative detection), and stores the resulting matrix as a group of transformed channel signal. Memory 815 then issues the signals described above as necessary, specifically as the baseband signal 816X, the transformed channel estimation signal group 817X, the baseband signal signal 816Y and the transformed channel estimation signal group 817Y.

Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection).

(Initial detection)

The internal MIMO detector 803 takes the baseband signal 801X, the 802X channel estimation signal group,

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the 801Y baseband signal and the 802Y channel estimation signal group as input. At this point, the modulation method for the modulated signal (flow) s1 and the modulated signal (flow) s2 is described as 16-QAM.

The internal MIMO detector 803 first calculates H (t) xY (t) xF from the 802X and 802Y channel estimation signal groups, thereby calculating a candidate signal point corresponding to the band signal 801X base. Figure 11 represents such a calculation. In Figure 11, each black point is a candidate signal point in the IQ plane. Since the modulation method is 16-QAM, there are 256 candidate signal points: (However, Figure 11 is only one representation and does not indicate all 256 candidate signal points). Being the four bits in the modulated signal s1 b0, b1, b2 and b3 and being four bits transmitted in the modulated signal s2 b4, b5, b6 and b7, there are the candidate signal points corresponding to (b0, b1, b2 , b3, b4, b5, b6, b7) in Figure 11. The square distance between each candidate signal point and each received signal point 1101 (corresponding to the baseband signal 801X) is calculated below. The euclldea square distance between each point is divided by the noise variance o2. Therefore, Ex (b0, b1, b2, b3, b4, b5, b6, b7) is calculated. That is, the square distance euclldea between a candidate signal point that corresponds to (b0, b1, b2, b3, b4, b5, b6, b7) and a received signal point is divided by the noise variance. At this point, each of the baseband signals and the modulated signals s1 and s2 is a complex signal.

Similarly, the internal MIMO detector 803 calculates H (t) xY (t) xF from the 802X and 802Y channel estimation signal groups, calculates the candidate signal points that correspond to the baseband signal 801Y , calculate the euclldea square distance between each of the candidate signal points and the received signal points (corresponding to the baseband signal 801Y), and divide the euclldea square distance by the noise variance o2. Therefore, EY is calculated (b0, b1, b2, b3, b4, b5, b6, b7). That is, Eg is the euclldea square distance between a candidate signal point that corresponds to (b0, b1, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance.

Then EX (b0, b1, b2, b3, b4, b5, b6, b7) + EY (b0, b1, b2, b3, b4, b5, b6, b7) = E (b0, b1, b2, b3, b4, b5, b6, b7).

The internal MIMO detector 803 emits E (b0, b1, b2, b3, b4, b5, b6, b7) as signal 804.

Logarithmic probability calculator 805A takes the signal 804 as input, calculates the logarithmic probability of bits b0, b1, b2 and b3, and issues the logarithmic probability signal 806A. Note that this calculation of log probability produces the log probability of a bit being 1 and the log probability of a bit being 0. The calculation method is as shown in Calculation 28 (formula 28), Calculation 29 (formula 29) and Calculation 30 (formula 30), and the details thereof are provided by the non-patent Bibliography 2 and 3.

Similarly, the logarithmic probability calculator 805B takes the signal 804 as input, calculates the logarithmic probability of bits b4, b5, b6 and b7, and issues the logarithmic probability signal 806B.

A deinterleaver (807A) takes the logarltmic probability signal 806A as input, performs deintercalation corresponding to that of the interleaver (the interleaver (304A) from Figure 3), and issues the logarithmic probability signal uninterleaved 808A.

Similarly, a deinterleaver (807B) takes the logarithmic probability signal 806B as input, performs deintercalation corresponding to that of the interleaver (the interleaver (304B) from Figure 3), and issues the uninterleaved logarithmic probability signal 808B .

Logarithmic probability ratio calculator 809A takes the logarithmic probability signal uninterleaved 808A as input, calculates the logarithmic probability ratio of the bits encoded by encoder 302A from Figure 3, and issues the logarithmic probability relationship signal 810A .

Similarly, the logarithmic probability ratio calculator 809B takes the logarithmic probability signal uninterleaved 808B as input, calculates the logarithmic probability ratio of the bits encoded by the encoder 302B from Figure 3, and issues the relation signal Logarithmic probability 810B.

The flexible input / flexible output decoder 811A takes the logarithmic probability ratio signal 810A as input, performs decoding, and issues a decoded logarithmic probability ratio 812A.

Similarly, the flexible input / flexible output decoder 811B takes the logarithmic probability ratio signal 810B as input, performs decoding, and issues the decoded logarithmic probability ratio 812B.

(Iterative decoding (iterative detection), k iterations)

The interleaver (813A) takes the k-th decoded logarithmic probability ratio 812A decoded by the flexible input decoder / flexible output as input, performs interleaving, and issues a relationship of

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Logarithmic probability interspersed 814A. At this point, the interleaving pattern used by the interleaver (813A) is identical to that of the interleaver (304A) from Figure 3.

Another interleaver (813B) takes the k-th decoded logarithmic probability ratio 812B decoded by the flexible input decoder / flexible output as input, performs interleaving, and issues the interleaved logarithmic probability ratio 814B. At this point, the interleaving pattern used by the interleaver (813B) is identical to that of the other interleaver (304B) from Figure 3.

The internal MIMO detector 803 takes the baseband signal 816X, the transformed channel estimation signal group 817X, the baseband signal signal 816Y, the transformed channel estimation signal group 817Y, the interleaved logarithmic probability ratio 814A and the interrelated logarithmic probability ratio 814B as input. At this point, the baseband signal 816X, the transformed channel estimate signal group 817X, the baseband signal signal 816Y, and the transformed channel estimate signal group 817Y are used instead of the baseband signal 801X, the 802X channel estimation signal group, the 801Y baseband signal group and the 802Y channel estimation signal group since the latter produces delays due to iterative decoding.

The iterative decoding operations of the internal MIMO detector 803 differ from the initial detection operations thereof in that the logarithmic probability interleaved ratios 814A and 814B are used in signal processing for the latter. The internal MIMO detector 803 first calculates E (b0, b1, b2, b3, b4, b5, b6, b7) in the same way as for the initial detection. In addition, the coefficients corresponding to Calculus 11 (formula 11) and Calculus 32 (formula 32) are calculated from the interleaved logarithmic probability ratios 814A and 814B. The value of E (b0, b1, b2, b3, b4, b5, b6, b7) is corrected using the coefficients as! calculated to obtain E '(b0, b1, b2, b3, b4, b5, b6, b7), which is emitted as signal 804.

Logarithmic probability calculator 805A takes the signal 804 as input, calculates the logarithmic probability of bits b0, b1, b2 and b3, and issues the logarithmic probability signal 806A. Note that this calculation of log probability produces the log probability of a bit being 1 and the log probability of a bit being 0. The calculation method is as shown in Calculation 31 (formula 31) through Calculation 35 ( formula 35), and the details are provided by the non-patent Bibliography 2 and 3.

Similarly, the logarithmic probability calculator 805B takes the signal 804 as input, calculates the logarithmic probability of bits b4, b5, b6 and b7, and issues the logarithmic probability signal 806B. The operations performed by the forward deinterleaver are similar to those performed for the initial detection.

Although Figure 8 illustrates the configuration of the signal processor when performing iterative detection, this structure is not absolutely necessary since good reception improvements are obtainable by iterative detection alone. Whenever the components necessary for iterative detection are present, the configuration does not need to include interleavers 813A and 813B. In such a case, the internal MIMO detector 803 does not perform iterative detection.

The key point for the present embodiment is the calculation of H (t) xY (t) xF. As shown in Non-patent Bibliography 5 and the like, QR decomposition can also be used to perform initial detection and iterative detection.

Also, as indicated by the non-patent Bibliography 11, the linear operations of MMSE (Minimum Medium Square Error) and ZF (Forced to Zero) can be performed based on H (t) xY (t) xF when the initial detection is performed .

Figure 9 illustrates the configuration of a signal processor, unlike that of Figure 8, which serves as the signal processor for the modulated signals transmitted by the transmission device from Figure 4. The point of difference from Figure 8 is the number of flexible input / flexible output decoders. A flexible input / flexible output decoder 901 takes the logarithmic probability ratio signals 810A and 810B as input, performs decoding, and issues a decoded logarithmic probability ratio 902. A distributor 903 takes the decoded logarithmic probability ratio 902 as input for distribution. Otherwise, the operations are identical to those explained for Figure 8.

As described above, when a transmission device according to the present embodiment using a MIMO system transmits a plurality of the modulated signals from a plurality of antennas, change the phase over time while multiplying by the matrix of Precoding to regularly change the phase results in improvements to the quality of data reception for a receiving device in a LOS environment, where direct waves are dominant, compared to a conventional MIMO system of spatial multiplexing.

In the present embodiment, and particularly in the configuration of the receiving device, the number of antennas

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It is limited and explanations are provided accordingly. However, the embodiment can also be applied to a larger number of antennas. In other words, the number of antennas in the receiving device does not affect the operations or advantageous effects of the present embodiment.

Also, although LDPC codes are described as a particular example, the present embodiment is not limited in this way. Additionally, the decoding method is not limited to the sum-product decoding example provided for the flexible input / output decoder. flexible. Other flexible input / flexible output decoding methods, such as the BCJR algorithm, SOVA and the Max-Log-Map algorithm can also be used. Details are provided in the non-patent Bibliography 6.

Furthermore, although the present embodiment is described using a single carrier method, no limitation is intended in this regard. The present embodiment is also applicable to multi-carrier transmission. Accordingly, the present embodiment can also be performed using, for example, spread spectrum communications, OFDM, SC-FDMA (Multiple Access by Single Carrier Frequency Division), SC-OFDM, wavelet OFDM as described in Bibliography no. of patent 7, and so on! successively. Additionally, in the present embodiment, the symbols other than the data symbols, such as pilot symbols (preamble, unique word and so on) or the symbols that transmit control information, can be arranged in the frame in any way.

The following describes an example in which OFDM is used as a multi-carrier method.

Figure 12 illustrates the configuration of a transmission device using OFDM. In Figure 12, components that operate in the same manner described for Figure 3 use identical reference numbers.

A processor related to OFDM 1201A takes a weighted signal 309A as input, performs processing related to OFDM therein, and issues transmission signal 1202A. Similarly, the processor related to OFDM 1201B takes the post-phase change signal 309B as input, performs processing related to OFDM therein, and issues the transmission signal 1202B.

Figure 13 illustrates a sample configuration of the OFDM-related processors 1201A and 1201B and forward from Figure 12. Components 1301A to 1310A are between 1201A and 312A from Figure 12, while components 1301B to 1310B They are between 1201B and 312B.

Serial to parallel converter 1302A performs serial to parallel conversion on weighted signal 1301A (corresponding to weighted signal 309A from Figure 12) and outputs parallel signal 1303A.

The reorganizer 1304A takes the parallel signal 1303A as input, performs rearrangement thereof, and issues the rearranged signal 1305A. The reordering is described in detail later.

The IFFT (Inverse Fast Fourier Transform) unit 1306A takes the reordered signal 1305A as input, applies an IFFT to it, and issues the post-IFFT signal 1307A.

The wireless unit 1308A takes the post-IFFT signal 1307A as input, performs processing such as frequency conversion and amplification, and emits the modulated signal 1309A. The modulated signal 1309A is then emitted as radio waves by the antenna 1310A.

The serial to parallel converter 1302B performs serial to parallel conversion on the weighted signal 1301B (corresponding to the post-phase change 309B from Figure 12) and emits the parallel signal 1303B.

The reorganizer 1304B takes the parallel signal 1303B as input, performs rearrangement thereof, and issues the rearranged signal 1305B. The reordering is described in detail later.

The IFFT unit 1306B takes the reordered signal 1305B as input, applies an IFFT to it, and issues the post-IFFT signal 1307B.

The wireless unit 1308B takes the post-IFFT signal 1307B as input, performs processing such as frequency conversion and amplification therein, and outputs modulated signal 1309B. The modulated signal 1309B is then emitted as radio waves by the antenna 1310A.

The transmission device from Figure 3 does not use a multi-carrier transmission method. Therefore, as shown in Figure 6, a phase change is made to achieve a period (cycle) of four and the post-phase change symbols are arranged in the time domain. As shown in Figure 12, when multi-carrier transmission is used, such as OFDM, then, of course, the precoded post-phase change symbols can be arranged with respect to the time domain as in Figure 3, and this It applies to each (sub-) carrier. However, for multi-carrier transmission, the arrangement can also be in the frequency domain, or both in the frequency domain and in the time domain. The following describes these

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dispositions

Figures 14A and 14B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13. The frequency axes they are composed of the (sub-) carriers 0 to 9. The modulated signals z1 and z2 share common time indications (timing) and use a common frequency band. Figure 14A illustrates a rearrangement method for the modulated signal z1 symbols, while Figure 14B illustrates a rearrangement method for the z2 modulated signal symbols. With respect to the symbols of the weighted signal 1301A introduced to the serial to parallel converter 1302A, the assigned order is # 0, # 1, # 2, # 3, and so! successively. At this point, given that the example refers to a period (cycle) of four, No. 0, No. 1, No. 2 and No. 3 are equivalent to a period (cycle). Similarly, # 4n, # 4n + 1, # 4n + 2, and # 4n + 3 (where n is a positive integer other than zero) are also equivalent to a period (cycle).

As shown in Figure 14A, the symbols # 0, # 1, # 2, # 3, and so! successively they are arranged in order, starting on carrier 0. The symbols # 0 to # 9 are given the time indication $ 1, followed by the symbols # 10 to # 19 provided with the time indication # 2, and so on! successively in a regular disposition. At this point, the modulated signals z1 and z2 are complex signals.

Similarly, with respect to the symbols of the weighted signal 1301B introduced to the serial to parallel converter 1302B, the assigned order is # 0, # 1, # 2, # 3, and so on! successively. At this point, since the example refers to a period (cycle) of four, a different phase change is applied to each of no. 0, no. 1, no. 2 and no. 3, which are equivalent to a period (cycle). Similarly, a different phase change is applied to each of # 4n, # 4n + 1, # 4n + 2, and # 4n + 3 (where n is a positive integer other than zero ), which are also equivalent to a period (cycle).

As shown in Figure 14B, the symbols # 0, # 1, # 2, # 3, and so! successively they are arranged in order, starting on carrier 0. The symbols # 0 to # 9 are given the time indication $ 1, followed by the symbols # 10 to # 19 provided with the $ 2 time indication, and so on! successively in a regular disposition.

The group of symbols 1402 shown in Figure 14B corresponds to a period (cycle) of symbols when the phase change method of Figure 6 is used. Symbol # 0 is the symbol obtained using the phase in the indication of time u in Figure 6, the symbol # 1 is the symbol obtained using the phase in the time indication u + 1 in Figure 6, the symbol # 2 is the symbol obtained using the phase in the indication of time u + 2 in Figure 6, and the symbol # 3 is the symbol obtained using the phase in the time indication u + 3 in Figure 6. Therefore, for any symbol # x, the symbol n . ° x is the symbol obtained using the phase in the time indication u in Figure 6 when x mod 4 equals 0 (that is, when the remainder of x divided by 4 is 0, the mod operator being mod), the Symbol No. x is the symbol obtained using the phase in the time indication u + 1 in Figure 6 when x mod 4 equals 1, the symbol No. x is the symbol obtained using the phase in the time indication u + 2 in Figure 6 when x mod 4 equals 2, and the symbol # x is the symbol obtained using the phase in the time indication u + 3 in Figure 6 when x mod 4 equals 3.

In the present embodiment, the modulated signal z1 shown in Figure 14A has not undergone a phase change.

As such, when a multi-carrier transmission method such as OFDM is used, and unlike single carrier transmission, the symbols can be arranged in the frequency domain. Of course, the method of arrangement of symbols is not limited to those illustrated by Figures 14A and 14B. Additional examples are shown in Figures 15A, 15B, 16A and 16B.

Figures 15A and 15B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from of Figures 14A and 14B. Figure 15A illustrates a rearrangement method for the modulated signal z1 symbols, while Figure 15B illustrates a rearrangement method for the z2 modulated signal symbols. Figures 15A and 15B differ from Figures 14A and 14B in the rearrangement method applied to the symbols of the modulated signal z1 and the symbols of the modulated signal z2. In Figure 15B, the symbols No. 0 to No. 5 are arranged in carriers 4 to 9, the symbols No. 6 to No. 9 are arranged in carriers 0 to 3, and this arrangement is repeated for symbols # 10 to # 19. At this point, as in Figure 14B, the group of symbols 1502 shown in Figure 15B corresponds to a period (cycle) of symbols when using the method of changing the phase of Figure 6.

Figures 16A and 16B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from of Figures 14A and 14B. Figure 16A illustrates a rearrangement method for the modulated signal z1 symbols, while Figure 16B illustrates a rearrangement method for the z2 modulated signal symbols. Figures 16A and 16B differ from Figures 14A and 14B in that,

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While Figures 14A and 14B show the slmboios arranged in sequential carriers, Figures 16A and 16B do not arrange the symbols in sequential carriers. Obviously, for Figures 16A and 16B, different rearrangement methods can be applied to the symbols of the modulated signal z1 and to the symbols of the modulated signal z2 as in Figures 15A and 15B.

Figures 17A and 17B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from those of Figures 14A to 16B. Figure 17A illustrates a reordering method for the z1 modulated signal symbols while Figure 17B illustrates a reordering method for the z2 modulated signal symbols. While Figures 14A to 16B show the symbols arranged with respect to the frequency axis, Figures 17A and 17B use the frequency and time axes together in a single arrangement.

Although Figure 6 describes an example where the phase change takes place over a period (cycle) of four intervals, the following example describes a period (cycle) of eight intervals. In Figures 17A and 17B, the group of symbols 1702 is equivalent to a period (cycle) of symbols when the phase change scheme (ie, for eight symbols) is used so that the symbol # 0 is the symbol obtained using the phase in the time indication u, the symbol # 1 is the symbol obtained using the phase in the time indication u + 1, the symbol # 2 is the symbol obtained using the phase in the indication of time u + 2, the symbol # 3 is the symbol obtained using the phase in the time indication u + 3, the symbol # 4 is the symbol obtained using the phase in the time indication u + 4, Symbol No. 5 is the symbol obtained using the phase in the time indication u + 5, the symbol No. 6 is the symbol obtained using the phase in the time indication u + 6, and the symbol No. 7 is the symbol obtained using the phase in the time indication u + 7. Therefore, for any symbol # x, the symbol # x is the symbol obtained using the phase in the time indication u when x mod 8 equals 0, the symbol # x is the symbol obtained using the phase in the time indication u + 1 when x mod 8 equals 1, the symbol # x is the

Symbol obtained using the phase in the time indication u + 2 when x mod 8 equals 2, the symbol No. x is the

Symbol obtained using the phase in the time indication u + 3 when x mod 8 equals 3, the symbol No. x is the

Symbol obtained using the phase in the time indication u + 4 when x mod 8 equals 4, the symbol No. x is the

Symbol obtained using the phase in the time indication u + 5 when x mod 8 equals 5, the symbol No. x is the

symbol obtained using the phase in the time indication u + 6 when x mod 8 equals 6, and the symbol # x is the symbol obtained using the phase in the time indication u + 7 when x mod 8 equals 7 In Figures 17A and 17B four intervals are used along the time axis and two intervals along the frequency axis for a total of 4x2 = 8 intervals, in which a period (cycle) of symbols is arranged. At this point, given mxn symbols per period (cycle) (ie, different phases are available for multiplication), then n intervals (carriers) in the frequency domain and m intervals in the time domain should be used to arrange the symbols of each period (cycle), so that m> n. This is because the phase of the direct waves fluctuates little by little in the time domain relative to the frequency domain. Accordingly, the present embodiment performs a regular phase change that reduces the effect of sustained direct waves. Therefore, the period (cycle) of phase change should preferably reduce direct wave fluctuations. Therefore, m should be greater than n. Given the above, using the time and frequency domains together for rearrangement, as shown in Figures 17A and 17B, it is preferable to use either the frequency domain or the time domain alone due to the strong probability of that the direct waves become regular. As a result, the effects of the present invention are more easily obtained. However, reordering in the frequency domain can lead to diversity gain due to the fact that the frequency domain fluctuations are abrupt. As such, using the frequency and time domains together for reordering is not always ideal.

Figures 18A and 18B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from of Figures 17A and 17B. Figure 18A illustrates a rearrangement method for the modulated signal z1 symbols, while Figure 18B illustrates a rearrangement method for the z2 modulated signal symbols. Like Figures 17A and 17B, Figures 18A and 18B illustrate the use of time and frequency axes together. However, in contrast to Figures 17A and 17B, where the frequency axis is prioritized and the time axis is used for the secondary symbol arrangement, Figures 18A and 18B prioritize the time axis and use the frequency axis to the provision of secondary symbol. In Figure 18B, the group of symbols 1802 corresponds to a period (cycle) of symbols when the phase change method is used.

In Figures 17A, 17B, 18A and 18B, the reordering method applied to the symbols of the modulated signal z1 and the symbols of the modulated signal z2 can be identical or can be differentiated as in Figures 15A and 15B. Any approach allows good reception quality to be obtained. Also, in Figures 17A, 17B, 18A and 18B, the symbols may be arranged non-sequentially as in Figures 16A and 16B. Any approach allows good reception quality to be obtained.

Figure 22 indicates frequency on the horizontal axis and time on the vertical axis thereof, and illustrates an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from the previous one. Figure 22 illustrates a regular phase change method using four intervals,

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similar to the indications of time ua or + 3 from Figure 6. The characteristic feature of Figure 22 is that, although the symbols are rearranged with respect to the frequency domain, when read along the time axis, a periodic displacement of n (n = 1 in the example of Figure 22) is evident. The group of frequency domain symbols 2210 in Figure 22 indicates four symbols to which the phase change is applied in the time indications u to u + 3 from Figure 6.

At this point, the symbol No. 0 is obtained through a phase change in the time indication u, the symbol No. 1 is obtained through a phase change in the time indication u + 1, the symbol # 2 is obtained through a phase change in the time indication u + 2 and the symbol # 3 is obtained through a phase change in the time indication u + 3.

Similarly, for the frequency domain symbol group 2220, the symbol # 4 is obtained through a phase change in the time indication u, the symbol # 5 is obtained through a change phase in the time indication u + 1, the symbol # 6 is obtained through a phase change in the time indication u + 2 and the symbol # 7 is obtained through a phase change in the time indication u + 3.

The phase change described above applies to the symbol in the time indication $ 1. However, to apply periodic displacement with respect to the time domain, the following phase change is applied to the groups of symbols 2201, 2202, 2203 and 2204.

For the time domain symbol group 2201, the # 0 symbol is obtained through a phase change in the time indication u, the # 9 symbol is obtained through a phase change in the indication of time u + 1, the symbol # 18 is obtained through a phase change in the time indication u + 2 and the symbol # 27 is obtained through a phase change in the indication of time u + 3.

For the time domain symbol group 2202, the symbol # 28 is obtained through a phase change in the time indication u, the symbol # 1 is obtained through a phase change in the time indication u + 1, the symbol # 10 is obtained through a phase change in the time indication u + 2, and the symbol # 19 is obtained through a phase change in the indication of time u + 3.

For the time domain symbol group 2203, the # 20 symbol is obtained through a phase change in the time indication u, the # 29 symbol is obtained through a phase change in the time indication u + 1, the symbol # 2 is obtained through a phase change in the time indication u + 2, and the symbol # 11 is obtained through a phase change in the indication of time u + 3.

For the time domain symbol group 2204, the # 12 symbol is obtained through a phase change in the time indication u, the # 21 symbol is obtained through a phase change in the time indication u + 1, the symbol # 30 is obtained through a phase change in the time indication u + 2, and the symbol # 3 is obtained through a phase change in the indication of time u + 3.

The characteristic feature of Figure 22 can be seen in that, taking the symbol No. 11 as an example, the two neighboring symbols thereof having the same time indication in the frequency domain (No. 10 and No. 12) are both symbols changed using a different phase than that of symbol # 11, and the two neighboring symbols of the same having the same carrier in the time domain (# 2 and # 20) are both symbols changed using a different phase than that of symbol No. 11. This is maintained not only for symbol No. 11, but also for any symbol that has two symbols that are neighbors in the frequency domain and in the time domain . Therefore, the phase change is carried out effectively. This is highly likely to improve the quality of data reception since the influence of regularizing direct waves is less prone to reception.

Although Figure 22 illustrates an example in which n = 1, the invention is not limited in this way. The same can be applied to a case in which n = 3. Additionally, although Figure 22 illustrates the realization of the previously described effects by arranging the symbols in the frequency domain and advancing in the time domain to achieve the characteristic effect of imparting a periodic displacement to the order of disposition of the symbol, the symbols can also be arranged randomly (or regularly) for the same effect.

[Embodiment 2]

In the embodiment 1, described above, phase change is applied to a weighted z (t) signal (precoded with a fixed precoding matrix). The following embodiments describe various phase change methods by which the effects of embodiment 1 can be obtained.

In the previously described embodiment, as shown in Figures 3 and 6, the phase changer 317B is configured to perform a phase change in only one of the signals emitted by the weighting unit 600.

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However, the phase change can also be applied before the precoding is carried out by the weighting unit 600. In addition to the components illustrated in Figure 6, the transmission device may also have the weighting unit 600 before the weight changer. phase 317B, as shown in Figure 25.

In such circumstances, the following configuration is possible. The phase changer 317B performs a regular phase change with respect to a baseband signal s2 (t), in which mapping has been performed according to a selected modulation method, and emits s2 '(t) = s2 (t ) and (t) (where y (t) varies with time t). The weighting unit 600 executes precoding in s2't, emits z2 (t) = W2s2 '(t) (see Calculation 42 (formula 42)) and the result is transmitted below.

Alternatively, the phase change can be performed on both modulated signals s1 (t) and s2 (t). As such, the transmission device is configured to include a phase changer that takes both signals emitted by the weighting unit 600, as shown in Figure 26.

Like the phase changer 317B, the phase changer 317A performs a regular phase change in the signal introduced thereto, and as such changes the phase of the signal z1 '(t) precoded by the weighting unit. The post-phase change signal z1 (t) is then transmitted to a transmitter.

However, the phase change rate applied by phase changers 317A and 317B varies simultaneously to perform the phase change shown in Figure 26. (The following describes a non-limiting example of the phase change method). For time indication u, the phase changer 317A from Figure 26 performs the phase change so that z1 (t) = y1 (t) z1 '(t), while the phase changer 317B makes the change phase so that z2 (t) = y2 (t) z2 '(t). For example, as shown in Figure 26, for the time indication u, y1 (u) = e0 and y2 (u) = e-jp / 2, for the time indication u + 1, y1 (u + 1 ) = ep / 4 and y2 (u + 1) = e-j3p / 4, and for the time indication u + k, y1 (u + k) = ekp / 4 and y2 (u + k) = e (k3p / 4-P / 2). At this point, the period (cycle) of regular phase change may be the same for both phase changers 317A and 317B, or it may vary for each.

Also, as described above, a phase change can be made before precoding is performed by the weighting unit. In such a case, the transmission device should be configured as illustrated in Figure 27 instead of as illustrated in Figure 26.

When a phase change takes place in both modulated signals, each of the transmission signals is, for example, control information that includes information about the phase change pattern. By obtaining the control information, the reception device knows the phase change method by which the transmission device regularly varies the change, that is, the phase change pattern, and can therefore demodulate (decode) the signals. correctly.

Next, variants of the sample configurations shown in Figures 6 and 25 are described with reference to Figures 28 and 29. Figure 28 differs from Figure 6 in the inclusion of ON / OFF phase change information 2800 and in which the phase change takes place only in one of z1 '(t) and z2' (t) (that is, performed in one of z1 '(t) and z2' (t), which have identical time indications or a common frequency). Therefore, to perform the phase change at one of z1 '(t) and z2' (t), the phase changers 317A and 317B shown in Figure 28 may each be ON, and perform the phase change, or OFF, and do not make the phase change. The information of phase change ON / OFF 2800 is control information thereof. The ON / OFF phase change information 2800 is output by the signal processing method information generator 314 shown in Figure 3.

The phase changer 317A of Figure 28 changes the phase to produce z1 (t) = y1 (t) z1 '(t), while the phase changer 317B changes the phase to produce z2 (t) = y2 (t) z2 '(t).

At this point, a phase change that has a period (cycle) of four applies, for example, to z1 '(t). (Meanwhile, the phase of z2 '(t) is not changed). Therefore, for the time indication u, y1 (u) = eP and y2 (u) = 1, for the time indication u + 1, y1 (u + 1) = ejp / 2 and y2 (u + 1) = 1, for the time indication u + 2, y1 (u + 2) = e y2 (u + 2) = 1, and for the time indication u + 3, y1 (u + 3) = e3p / 2 e y2 (u + 3) = 1.

Next, a phase change having a period (cycle) of four applies, for example, to z2 '(t). (Meanwhile, the phase of z1 '(t) is not changed). Therefore, for the time indication u + 4, y1 (u + 4) = 1 e y2 (u + 4) = e0, for the time indication u + 5, y1 (u + 5) = 1 e y2 ( u + 5) = eT / 2, for the time indication u + 6, y1 (u + 6) = 1 e y2 (u + 6) = e ^, and for the time indication u + 7, y1 (u +7) = 1 and y2 (u + 7) = e3p / 2.

Therefore, given the previous examples.

for any time indication 8k, y1 (8k) = ej0 and y2 (8k) = 1,

for any time indication 8k + 1, y1 (8k + 1) = ejp / 2 and y2 (8k + 1) = 1,

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for any time indication 8k + 2, yi (8k + 2) = ejp and y2 (8k + 2) = 1,

for any time indication 8k + 3, y1 (8k + 3) = ej3p / 2 and y2 (8k + 3) = 1,

for any time indication 8k + 4, y1 (8k + 4) = 1 and y2 (8k + 4) = ej0,

for any time indication 8k + 5, y1 (8k + 3) = 1 and y2 (8k + 5) = ejp / 2,

for any time indication 8k + 6, y1 (8k + 6) = 1 and y2 (8k + 6) = ejp, e

for any time indication 8k + 7, y1 (8k + 7) = 1 and y2 (8k + 7) = ej3p / 2.

As described above, there are two intervals, one where the phase change takes place in z1 '(t) only, and one where the phase change takes place in z2' (t) only. Additionally, the two intervals form a period (cycle) of phase change. Although the above explanation describes the interval where the phase change is performed in z1 '(t) only and the interval where the phase change is performed in z2' (t) only as being equal, no limitation is intended in this way. . The two intervals can also be differentiated. In addition, although the above explanation describes performing a phase change that has a period (cycle) of four in z1 '(t) only and then a phase change that has a period (cycle) of four in z2' (t) only , no limitation is intended in this way. Phase changes can be made in z1 '(t) and in z2' (t) in any order (for example, the phase change can alternate between being made in z1 '(t) and in z2' (t), or it can be be done in random order).

The phase changer 317A of Figure 29 changes the phase to produce s1 '(t) = y1 (t) s1 (t), while the phase changer 317B changes the phase to produce s2' (t) = y2 (t) s2 (t).

At this point, a phase change that has a period (cycle) of four applies, for example, to s1 (t). (Meanwhile, s2 (t) remains unchanged). Therefore, for the time indication u, y1 (u) = e0 and y2 (u) = 1, for the time indication u + 1, y1 (u + 1) = ejp / 2 and y2 (u + 1) = 1, for the time indication u + 2, y1 (u + 2) = e y2 (u + 2) = 1, and for the time indication u + 3, y1 (u + 3) = ej3p / 2 e y2 (u + 3) = 1.

Next, a phase change that has a period (cycle) of four applies, for example, to s2 (t). (Meanwhile, s1 (t) remains unchanged). Therefore, for the time indication u + 4, y1 (u + 4) = 1 e y2 (u + 4) = e0, for the time indication u + 5, y1 (u + 5) = 1 e y2 ( u + 5) = eA / 2, for the time indication u + 6, y1 (u + 6) = 1 and y2 (u + 6) = eA, and for the time indication u + 7, y1 (u + 7) = 1 and y2 (u + 7) = e3p / 2.

Therefore, given the previous examples,

for any time indication 8k, y1 (8k) = ej0 ey2 (8k) = 1,

for any time indication 8k + 1, y1 (8k + 1) = ejp / 2 and y2 (8k + 1) = 1,

for any time indication 8k + 2, y1 (8k + 2) = ejp and y2 (8k + 2) = 1,

for any time indication 8k + 3, y1 (8k + 3) = ej3p / 2 and y2 (8k + 3) = 1,

for any time indication 8k + 4, y1 (8k + 4) = 1 and y2 (8k + 4) = ej0,

for any time indication 8k + 5, y1 (8k + 5) = 1 and y2 (8k + 5) = ejp / 2,

for any time indication 8k + 6, y1 (8k + 6) = 1 and y2 (8k + 6) = ejp, e

for any time indication 8k + 7, y1 (8k + 7) = 1 and y2 (8k + 7) = ej3p / 2.

As described above, there are two intervals, one where the phase change is performed in s1 (t) only, and one where the phase change is performed in s2 (t) only. Additionally, the two intervals form a period (cycle) of phase change. Although the above explanation describes the interval where the phase change is performed in s1 (t) only and the interval where the phase change is performed in s2 (t) only as being equal, no limitation is intended in this way. The two intervals can also be differentiated. In addition, although the above explanation describes performing the phase change that has a period (cycle) of four in s1 (t) only and then performing the phase change that has a period (cycle) of four in s2 (t) only, No limitation is intended in this way. Phase changes can be made in s1 (t) and in s2 (t) in any order (for example, you can switch between being done in s1 (t) and in s2 (t), or it can be done in random order).

Therefore, the reception conditions under which the reception device receives each transmission signal z1 (t) and z2 (t) are equalized. By periodically changing the phase of the symbols in the received signals z1 (t) and z2 (t), the ability of the corrected error codes to correct errors can be improved, thereby improving the quality of the signal received in the LOS environment.

Accordingly, realization 2 as described above may produce the same results as realization 1 described above.

Although the present embodiment uses a unique carrier method, that is, time domain phase change, as an example, no limitation is intended in this regard. The same effects are also achievable using multi-carrier transmission. Accordingly, the present embodiment can also be performed using, for example, spread spectrum communications, OFDM, SC-FDMA (Multiple Access by Single Carrier Frequency Division), SC-OFDM, particle OFDM as described in Bibliography no. of patent 7 and so!

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successively. As described above, although the present embodiment explains the phase change as changing the phase with respect to the time domain t, the phase can be changed as an alternative with respect to the frequency domain as described in embodiment 1. That is, considering phase change method in the time domain t described in the present embodiment and substituting t for f (where f is the frequency of ((sub-) carrier)) leads to a phase change applicable to the frequency domain. Also, as explained above for embodiment 1, the phase change method of the present embodiment is also applicable to a phase change with respect to both the time domain and the frequency domain.

Therefore, although Figures 6, 25, 26 and 27 illustrate phase changes in the time domain, replacing time t with carrier f in each of Figures 6, 25, 26 and 27 corresponds to a phase change in the frequency domain. In other words, substitute (t) for (t, f) where t is time and f is frequency corresponds to the phase change in time-frequency blocks.

Additionally, in the present embodiment, the symbols other than the data symbols, such as pilot symbols (preamble, unique word, etc.) or the symbols that transmit control information, can be arranged in the frame in any way.

[Embodiment 3]

Embodiments 1 and 2, described above, analyze regular phase changes. Embodiment 3 describes a method for allowing the receiving device to obtain good quality of the received signal for data, regardless of the arrangement of the receiving device, considering the location of the receiving device with respect to the transmission device.

Embodiment 3 refers to the arrangement of the symbol in signals obtained through a phase change.

Figure 31 illustrates an example of frame configuration for a portion of the symbols in a signal in the time-frequency domains, given a transmission method where a regular phase change is performed for a multi-carrier method such as OFDM.

First, an example is explained in which the phase change is performed in one of two baseband signals, precoded as explained in embodiment 1 (see Figure 6).

(Although Figure 6 illustrates a phase change in the time domain, changing the time t by the carrier f in Figure 6 corresponds to a phase change in the frequency domain. In other words, replace (t) with ( t, f) where t is time and f is frequency corresponds to making phase changes in time-frequency blocks).

Figure 31 illustrates the frame configuration of the modulated signal z2 ', which is introduced to the phase changer 317B from Figure 12. Each square represents a symbol (although both signals s1 and s2 are included for precoding purposes, depending of the precoding matrix, only one of the signals s1 and s2 can be used).

Consider symbol 3100 on carrier 2 and the time indication $ 2 of Figure 31. The carrier described at this point may be referred to as a subcarrier as an alternative.

In carrier 2, there is a very strong correlation between the channel conditions for the symbol 3100A in the carrier 2, the time indication $ 2 and the channel conditions for the nearest neighboring symbols in the time domain $ 2, that is, the symbol 3013 in the time indication $ 1 and the symbol 3101 in the time indication $ 3 in the carrier 2.

Similarly, for the $ 2 time indication, there is a very strong correlation between the channel conditions for the 3100 symbol in the carrier 2, the $ 2 time indication and the channel conditions for the nearest neighboring symbols in the domain of frequency to carrier 2, that is, symbol 3104 in carrier 1, time indication $ 2 and symbol 3104 in time indication $ 2, carrier 3.

As described above, there is a very strong correlation between the channel conditions for the symbol 3100 and the channel conditions for each symbol 3101, 3102, 3103 and 3104.

The present description considers N different phases (N being an integer, N> 2) for multiplication in a transmission method where the phase is changed regularly. The symbols illustrated in Figure 31 are indicated as ej0, for example. This means that this symbol is the signal z2 'from Figure 6 that has undergone a change in phase through multiplication by ej0. That is, the values indicated in Figure 31 for each of the symbols are the values of y (t) from Calculation 42 (formula 42), which are also the values of z2 (t) = y2 (t) z2 '(t) described in embodiment 2.

The present embodiment takes advantage of the high correlation in the existing channel conditions between symbols

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which are neighbors in the frequency domain and / or slmboios that are neighbors in the time domain in a arrangement of slmboios that allows high quality data reception to be obtained by the receiving device that receives the changed phase symbols.

To achieve this high quality of data reception, Conditions No. 1 and No. 2 are necessary.

(Condition # 1)

As shown in Figure 6, for a transmission method that involves a regular phase change performed on the precoded baseband signal z2 'using multi-carrier transmission such as OFDM, time X, the carrier

And it must be a symbol to transmit data (hereinafter, data symbol), symbols that are neighbors in the time domain, that is, in time X-1, the carrier Y and in time X + 1 the carrier And they must be data symbols, and a different phase change must be made in the precoded baseband signal z2 'corresponding to each of these three data symbols, that is, in the precoded baseband signal z2' in the time X, carrier Y, at time X-1, carrier Y and at time X + 1, carrier Y.

(Condition # 2)

As shown in Figure 6, for a transmission method that involves a regular phase change performed on the precoded baseband signal z2 'using multi-carrier transmission such as OFDM, time X, the carrier

And it must be a data symbol, symbols that are neighbors in the frequency domain, that is, at time X, carrier Y-1 and at time X, carrier Y + 1 must also be data symbols, and a different phase change must be made in the precoded baseband signal z2 'corresponding to each of these three data symbols, that is, in the precoded baseband signal z2' at time X, the carrier Y, in time X, carrier Y-1 and time X, carrier Y + 1.

Ideally, data symbols that satisfy Condition # 1 should be present. Similarly, data symbols that satisfy Condition # 2 should be present.

The reasons for supporting Conditions No. 1 and No. 2 are as follows.

There is a very strong correlation between the channel conditions of the given symbol of a transmission signal (hereinafter, the symbol A) and the channel conditions of the symbols that are neighboring the symbol A in the time domain, as has been previously described.

Therefore, when three symbols that are neighbors in the time domain each have different phases, then in spite of the degradation in reception quality in the LOS environment (poor signal quality produced by degradation in conditions due to phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols that are neighbors to symbol A are highly likely to provide good reception quality. As a result, good signal quality received after error correction and decoding is attainable.

Similarly, there is a very strong correlation between the channel conditions of the given symbol of a transmission signal (hereinafter, the symbol A) and the channel conditions of the symbols that are neighboring the symbol A in the frequency domain , as described above.

Therefore, when three symbols that are neighbors in the frequency domain each have different phases, then despite the degradation in reception quality in the LOS environment (poor signal quality caused by degradation in conditions due to Direct wave phase ratios despite high signal quality in terms of SNR) for symbol A, the remaining two symbols that are neighbors to symbol A are highly likely to provide good reception quality. As a result, good signal quality received after error correction and decoding is attainable.

By combining Conditions # 1 and # 2, it is likely that even higher data reception quality for the receiving device is achievable. Therefore, the following Condition # 3 can be obtained.

(Condition # 3)

As shown in Figure 6, for a transmission method that involves a regular phase change performed on the precoded baseband signal z2 'using multi-carrier transmission such as OFDM, time X, the carrier

And it must be a data symbol, symbols that are neighbors in the time domain, that is, at time X-1, carrier Y and at time X + 1, carrier Y must also be data symbols, and symbols that are neighbors in the frequency domain, that is, at time X, carrier Y-1 and at time X, carrier Y + 1 must also be data symbols, and a different phase change must also be made in the precoded baseband signal z2 'corresponding to each of these five data symbols, that is, in the precoded baseband signal z2' at time X, carrier Y, at time X, carrier Y- 1 at time X, the carrier Y + 1,

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in a time X-1, the carrier Y, and in time X + 1, the carrier Y.

At this point, the different changes in phase are as follows. Phase changes are defined from 0 radians to 2p radians. For example, for time X, carrier Y, a phase change of eBXY is applied to the precoded baseband signal z2 'from Figure 6, for time X-1, carrier Y, a change of phase of ej0X-1Y to the precoded baseband signal z2 'from Figure 6, for time X + 1, carrier Y, a phase shift of ejBX + 1Y is applied to the precoded baseband signal z2' from Figure 6, so that 0 <0x, and <2p, 0 <0x-1, and <2p, and 0 <0x + 1, and <2p, all units being in radians. Therefore, for Condition No. 1, it follows that 0x, and t 0x-1, y, 0x, yt0x + 1, y, and that 0x-1, yt0x + 1, y. Similarly, for Condition 2, it follows that 0x, and t 0x, y-1, 0x, and t 0x, y + 1, and that 0x, yt0x, y + 1. And, for Condition No. 3, it follows that 0x, yt 0x-1, y, 0x, yt 0X + 1, Y, 0X, Y t 0X, Y-1, 0X, Y t 0X, Y-1 , 0X-1, Y t 0X + 1, Y, 0X-1, Y t 0X, Y-1, 0X-1, Y t 0X + 1, Y, 0X + 1, Y t 0X-1, Y, 0X + 1, Y t 0X, Y + 1, and that 0X, Y-1t 0x, and + 1.

Ideally, the data symbols that satisfy Condition # 3 should be present.

Figure 31 illustrates an example of Condition # 3 where the symbol A corresponds to the symbol 3100. The symbols are arranged so that the phase by which the precoded baseband signal z2 'is multiplied from Figure 6 it differs for the symbol 3100, for both neighboring symbols thereof in the time domain 3101 and 3102, and for both neighboring symbols thereof in the frequency domain 3102 and 3104. Consequently, despite the signal quality degradation received from the 3100 symbol for the receiver, good signal quality is highly likely for neighboring signals, thus guaranteeing good signal quality after error correction.

Figure 32 illustrates a distribution of symbols obtained through phase changes under these conditions.

As is evident from Figure 32, with respect to any data symbol, a different phase change is applied to each symbol that is neighboring in the time domain and in the frequency domain. As such, the ability of the receiving device to correct errors can be improved.

In other words, in Figure 32, when all symbols that are neighbors in the time domain are data symbols, Condition # 1 is satisfied for all X and all Y.

Similarly, in Figure 32, when all symbols that are neighbors in the frequency domain are data symbols, Condition # 2 is satisfied for all X and all Y.

Similarly, in Figure 32, when all symbols that are neighbors in the frequency domain are data symbols and all symbols that are neighbors in the time domain are data symbols, Condition # 3 is satisfied to all X and all Y.

The following describes an example in which a phase change is performed on two precoded baseband signals, as explained in embodiment 2 (see Figure 26).

When a phase change is made in the precoded baseband signal z1 'and in the precoded baseband signal z2' as shown in Figure 26, various phase change methods are possible. The details of them are explained below.

Scheme 1 implies a phase change of the precoded baseband signal z2 'as described above, to achieve the phase change illustrated by Figure 32. In Figure 32, a phase change having a period (cycle ) of ten is applied to the precoded baseband signal z2 '. However, as described above, to satisfy Conditions No. 1, No. 2 and No. 3, the phase change applied to the precoded baseband signal z2 'on each (sub-) carrier rod over time. (Although such changes are applied in Figure 32 with a period (cycle) of ten, other methods of phase change are also possible). Then, as shown in Figure 33, the phase change made in the precoded baseband signal z1 'produces a constant value that is one tenth that of the phase change made in the precoded baseband signal z2'. In Figure 33, for a period (cycle) (of phase change made in the precoded baseband signal z2 ') including the time indication $ 1, the value of the phase change made in the precoded baseband signal z1' It's ej0. Then, for the following period (cycle) (of phase change made in the precoded baseband signal z2 ') including the time indication $ 2, the value of the phase change made in the precoded baseband signal z1' is ep / 9, and so on! successively.

The symbols illustrated in Figure 33 are indicated as ej0, for example. This means that this symbol is the signal z1 'from Figure 26 to which a change in phase has been applied through multiplication by e0. That is, the values indicated in Figure 33 for each of the symbols are the values of z1 (t) = y1 (t) z1 '(t) described in embodiment 2 for y1 (t).

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As shown in Figure 33, the phase change made in the precoded baseband signal z1 'produces a constant value that is one tenth that of the phase change made in the precoded baseband signal z2' so that the Post-phase change value varies with the number of each period (cycle). (As described above, in Figure 33, the value is ej0 for the first period (cycle), e ^ / 9 for the second period (cycle), and so on).

As described above, the change in phase made in the precoded baseband signal z2 'has a period (cycle) of ten, but the period (cycle) can be made effectively greater than ten taking into account the change in phase applied to the precoded baseband signal z1 'and the precoded baseband signal z2'. Accordingly, the quality of data reception for the receiving device can be improved.

Scheme 2 involves a phase change of the precoded baseband signal z2 'as described above, to achieve the phase change illustrated by Figure 32. In Figure 32, a phase change having a period (cycle ) of ten is applied to the precoded baseband signal z2 '. However, as described above, to satisfy Conditions No. 1, No. 2 and No. 3, the phase change applied to the precoded baseband signal z2 'on each (sub-) carrier rod over time. (Although such changes are applied in Figure 32 with a period (cycle) of ten, other methods of phase change are also possible). Then, as shown in Figure 30, the phase change made in the precoded baseband signal z1 'differs from that made in the precoded baseband signal z2' in that it has a period (cycle) of three instead of ten.

The symbols illustrated in Figure 30 are indicated as ej0, for example. This means that this symbol is the signal z1 'from Figure 26 to which a change in phase has been applied through multiplication by e0. That is, the values indicated in Figure 30 for each of the symbols are the values of z1 (t) = y1 (t) z1 '(t) described in realization 2 for y1 (t).

As described above, the phase change made in the precoded baseband signal z2 'has a period (cycle) of ten, but taking into account the phase changes applied to the precoded baseband signal z1' and the signal of precoded baseband z2 ', the period (cycle) can be effectively made equivalent to 30 for both precoded baseband signals z1' and z2 '. Accordingly, the quality of data reception for the receiving device can be improved. An effective way to apply method 2 is to make a phase change in the precoded baseband signal z1 'with a period (cycle) of N and make a phase shift in the precoded baseband signal z2' with a period ( cycle) of M so that N and M are coprime. As such, taking into account both precoded baseband signals z1 'and z2', a period (cycle) of NxM is easily achievable, effectively making the period (cycle) longer when N and M are coprime.

The above describes an example of the phase change method belonging to embodiment 3. The present invention is not limited in this way. As explained for embodiments 1 and 2, a phase change can be made with respect to the frequency domain or the time domain, or in time-frequency blocks. Improvement similar to the quality of data reception for the receiving device can be obtained in all cases.

The same also applies to frames that have a different configuration from the one described above, where the pilot symbols (SP symbols) and the symbols that transmit control information are inserted between the data symbols. The details of the phase change in such circumstances are as follows.

Figures 47A and 47B illustrate the frame configuration of the modulated signals (precoded baseband signals) z1 or z1 'and z2' in the time-frequency domain. Figure 47A illustrates the frame configuration of the modulated signal (precoded baseband signal) z1 or z1 'while Figure 47B illustrates the frame configuration of the modulated signal (precoded baseband signal) z2'. In Figures 47A and 47B, 4701 marks pilot symbols while 4702 marks data symbols. The 4702 data symbols are the symbols in which precoding or precoding and a phase change have been performed.

Figures 47A and 47B, like Figure 6, indicate the arrangement of the symbols when a phase change is applied to the precoded baseband signal z2 '(although no phase change is made in the precoded baseband signal z1) . (Although Figure 6 illustrates a change in phase with respect to the time domain, changing the time t by the carrier f in Figure 6 corresponds to a change in phase with respect to the frequency domain. In other words, replace (t) for (t, f) where t is time and f is frequency corresponds to a phase change in time-frequency blocks). Therefore, the numerical values indicated in Figures 47A and 47B for each of the symbols are the values of the precoded baseband signal z2 'after a phase change is made. Values for the precoded baseband signal z1 '(z1) symbols are not provided since no phase change is made therein.

The key point of Figures 47A and 47B is that a phase change is made in the data symbols of the precoded baseband signal z2 ', that is, in the precoded symbols. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no change is made in phase

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in the pilot slmboios inserted in z2 '.

Figures 48A and 48B illustrate the frame configuration of the modulated signals (precoded baseband signals) z1 or z1 'and z2' in the time-frequency domain. Figure 48A illustrates the frame configuration of the modulated signal (precoded baseband signal) z1 or z1 'while Figure 48B illustrates the frame configuration of the modulated signal (precoded baseband signal) z2'. In Figures 48A and 48B, 4701 marks pilot symbols while 4702 marks data symbols. The 4702 data symbols are the symbols in which precoding or precoding and a phase change have been performed.

Figures 48A and 48B, like Figure 26, indicate the arrangement of the symbols when a phase change is applied to the precoded baseband signal z1 'and to the precoded baseband signal z2'. (Although Figure 26 illustrates a change in phase with respect to the time domain, changing the time t by the carrier f in Figure 26 corresponds to a change in phase with respect to the frequency domain. In other words, replace (t) for (t, f) where t is time and f is frequency corresponds to a phase change in time-frequency blocks). Therefore, the numerical values indicated in Figures 48A and 48B for each of the symbols are the values of the precoded baseband signal z1 'and z2' after a phase change.

The key point of Figures 48A and 48B is that a phase change is made in the data symbols of the precoded baseband signal z1 ', that is, in the precoded symbols thereof, and in the data symbols of the precoded baseband signal z2 ', that is, in the precoded symbols thereof. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no phase change is made in the pilot symbols inserted in z1 ', nor in the pilot symbols inserted in z2'.

Figures 49A and 49B illustrate the frame configuration of the modulated signals (precoded baseband signals) z1 or z1 'and z2' in the time-frequency domain. Figure 49A illustrates the frame configuration of the modulated signal (precoded baseband signal) z1 or z1 'while Figure 49B illustrates the frame configuration of the modulated signal (precoded baseband signal) z2'. In Figures 49A and 49B, 4701 marks pilot symbols, 4702 marks data symbols, and 4901 marks null symbols for which the phase component of the baseband signal I = 0 and the quadrature component Q = 0. As such , the 4702 data symbols are the symbols in which precoding or precoding and a change in phase have been performed. Figures 49A and 49B differ from Figures 47A and 47B in the configuration method for symbols other than data symbols. The times and carriers in which the pilot symbols are inserted in the modulated signal z1 'are null symbols in the modulated signal z2'. Conversely, the times and carriers in which the pilot symbols are inserted in the modulated signal z2 'are null symbols in the modulated signal z1'.

Figures 49A and 49B, like Figure 6, indicate the arrangement of the symbols when a phase change is applied to the precoded baseband signal z2 '(although no phase change is made in the precoded baseband signal z1) . (Although Figure 6 illustrates a change in phase with respect to the time domain, changing the time t by the carrier f in Figure 6 corresponds to a change in phase with respect to the frequency domain. In other words, replace (t) for (t, f) where t is time and f is frequency corresponds to a phase change in time-frequency blocks). Therefore, the numerical values indicated in Figures 49A and 49B for each of the symbols are the values of the precoded baseband signal z2 'after a phase change is made. Values for the precoded baseband signal z1 '(z1) symbols are not provided since no phase change is made therein.

The key point of Figures 49A and 49B is that a phase change is made in the data symbols of the precoded baseband signal z2 ', that is, in the precoded symbols. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no change in phase is made in the pilot symbols inserted in z2 '.

Figures 50A and 50B illustrate the frame configuration of the modulated signals (precoded baseband signals) z1 or z1 'and z2' in the time-frequency domain. Figure 50A illustrates the frame configuration of the modulated signal (precoded baseband signal) z1 or z1 'while Figure 50B illustrates the frame configuration of the modulated signal (precoded baseband signal) z2'. In Figures 50A and 50B, 4701 marks pilot symbols, 4702 marks data symbols, and 4901 marks null symbols for which the phase component of the baseband signal I = 0 and the quadrature component Q = 0. As such , the 4702 data symbols are the symbols in which precoding or precoding and a change in phase have been performed. Figures 50A and 50B differ from Figures 48A and 48b in the configuration method for symbols other than data symbols. The times and carriers in which the pilot symbols are inserted in the modulated signal z1 'are null symbols in the modulated signal z2'. Conversely, the times and carriers in which the pilot symbols are inserted in the modulated signal z2 'are null symbols in the modulated signal z1'.

Figures 50A and 50B, like Figure 26, indicate the arrangement of the symbols when a phase change is applied to the precoded baseband signal z1 'and to the precoded baseband signal z2'. (Although Figure 26 illustrates a change in phase with respect to the time domain, changing the time t by the carrier f in Figure 26 corresponds to a change in phase with respect to the frequency domain. In other words, replace (t) by (t, f)

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where t is time and f is frequency corresponds to a phase change in time-frequency blocks). Therefore, the numerical values indicated in Figures 50A and 50B for each of the symbols are the values of the precoded baseband signal z1 'and z2' after the change in phase.

The key point of Figures 50A and 50B is that a phase change is made in the data symbols of the precoded baseband signal z1 ', that is, in the precoded symbols thereof, and in the data symbols of the precoded baseband signal z2 ', that is, in the precoded symbols thereof. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no phase change is made in the pilot symbols inserted in z1 ', nor in the pilot symbols inserted in z2'.

Figure 51 illustrates a sample configuration of a transmission device that generates and transmits the modulated signal having the frame configuration of Figures 47A, 47B, 49A and 49B. The components thereof that perform the same operations as those in Figure 4 use the same reference symbols between them.

In Figure 51, the weighting units 308A and 308B and the phase changer 317B only operate at the times indicated by the frame configuration signal 313 as corresponds to data symbols.

In Figure 51, a pilot symbol generator 5101 (which also generates null symbols) issues the baseband signals 5102A and 5102B for a pilot symbol each time the frame configuration signal 313 indicates a pilot symbol (and a null symbol ).

Although not indicated in frame configurations from Figures 47A to 50B, when no precoding (or phase rotation) is performed, such as when a modulated signal is transmitted using only one antenna (so that the other antenna does not transmits signal) or when a space-time coding transmission method (particularly, space-time block coding) is used to transmit control information symbols, then frame configuration signal 313 takes information information symbols control 5104 and control information 5103 as input. When frame configuration signal 313 indicates a control information symbol, baseband signals 5102A and 5102B thereof are output.

The wireless units 310A and 310B of Figure 51 take a plurality of baseband signals as input and select a desired baseband signal according to the frame configuration signal 313. The wireless units 310A and 310B then apply processing of OFDM signal and emit modulated signals 311A and 311B according to the frame configuration.

Figure 52 illustrates a sample configuration of a transmission device that generates and transmits the modulated signal having the frame configuration of Figures 48A, 48B, 50A and 50B. The components thereof that perform the same operations as those in Figures 4 and 51 use the same reference symbols between them. Figure 51 shows an additional phase changer 317A that only operates when frame configuration signal 313 indicates a data symbol. In all other cases, the operations are identical to those explained for Figure 51.

Figure 53 illustrates a sample configuration of a transmission device that differs from that of Figure

51. The following describes the points of difference. As shown in Figure 53, phase changer 317B takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 317B makes the phase change in the precoded baseband signal 316B. When the frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, the phase changer 317B pauses the phase change operations so that the baseband signal symbols are emitted as is, just as it is. (This can be interpreted as performing forced rotation corresponding to ej0).

A selector 5301 takes the plurality of the baseband signals as input and selects a baseband signal having a symbol indicated by the frame configuration signal 313 to be broadcast.

Figure 54 illustrates a sample configuration of a transmission device that differs from that of Figure

52. The following describes the points of difference. As shown in Figure 54, phase changer 317B takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 317B makes the phase change in the precoded baseband signal 316B. When the frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, the phase changer 317B pauses the phase change operations so that the baseband signal symbols are emitted as is, just as it is. (This can be interpreted as performing forced rotation corresponding to ej0).

Similarly, as shown in Figure 54, phase changer 5201 takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 5201 makes the phase change in the precoded baseband signal 309A. When

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the frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, the phase changer 5201 pauses the phase change operations so that the baseband signal symbols are emitted such which. (This can be interpreted as performing forced rotation corresponding to ej0).

The above explanations are provided using pilot symbols, control symbols and data symbols as examples. However, the present invention is not limited in this way. When symbols are transmitted using methods other than precoding, such as single antenna transmission or transmission using space-time block coding, not making a phase change is important. Conversely, making a phase change in the symbols that have been precoded is the key point of the present invention.

Therefore, a characteristic feature of the present invention is that the phase change is not carried out in all the symbols in the frame configuration in the time-frequency domain, but only in the signals that have been precoded.

[Embodiment 4]

Embodiments 1 and 2, described above, analyze a regular phase change. Embodiment 3, however, discloses a different phase change in symbols that are neighbors.

The present embodiment describes a phase change method that varies according to the modulation method and the coding rate of the error correction codes used by the transmission device.

Table 1, below, is a list of phase change method settings that correspond to the settings and parameters of the transmission device.

[Table 1]

 No. of modulated transmission signals

 Modulation scheme Coding rate Phase change pattern

 2

 # 1: QPSK, # 2: QPSK # 1: 1/2, # 2 2/3 # 1: -, # 2: A

 2

 # 1: QPSK, # 2: QPSK # 1: 1/2, # 2: 3/4 # 1: A, # 2: B

 2

 # 1: QPSK, # 2: QPSK # 1: 2/3, # 2: 3/5 # 1: A, # 2: C

 2

 # 1: QPSK, # 2: QPSK # 1: 2/3, # 2: 2/3 # 1: C, # 2: -

 2

 # 1: QPSK, # 2: QPSK # 1: 3/3, # 2: 5/6 # 1: D, # 2: E

 2

 # 1: QPSK, # 2: 16-QAM # 1: 1/2, # 2: 2/3 # 1: B, # 2: A

 2

 # 1: QPSK, # 2: 16-QAM # 1: 1/2, # 2: 3/4 # 1: A, # 2: C

 2

 # 1: QPSK, # 2: 16-QAM # 1: 1/2, # 2: 3/5 # 1: -, # 2: E

 2

 # 1: QPSK, # 2: 16-QAM # 1: 2/3, # 2: 3/4 # 1: D, # 2: -

 2

 # 1: QPSK, # 2: 16-QAM # 1: 2/3, # 2: 5/6 # 1: D, # 2: B

 2

 # 1: 16-QAM, # 2: 16-QAM # 1: 1/2, # 2: 2/3 # 1: -, # 2: E

 :

 :

 :

 :

In Table 1, No. 1 indicates the modulated signal s1 from the above-described embodiment 1 (the baseband signal s1 modulated with the modulation method established by the transmission device) and # 2 indicates the signal modulated s2 (the baseband signal s2 modulated with the modulation method established by the transmission device). The coding rate column of Table 1 indicates the coding rate of the error correction codes for modulation methods # 1 and # 2. The phase change pattern column of Table 1 indicates the phase change method applied to the precoded baseband signals z1 (z1 ') and z2 (z2'), as explained in embodiments 1 to 3. Although the phase change patterns are labeled A, B, C , D, E, and so on! successively, this refers to the degree of phase change applied, for example, in a phase change pattern provided by Calculation 46 (formula 46) and Calculation 47 (formula 47), above. In the phase change pattern column of Table 1, the dash means that no phase change is applied.

The combinations of modulation method and coding rate listed in Table 1 are examples. Other modulation methods (such as 128-QAM and 256-QAM) and coding rates (such as 7/8) not listed in Table 1 may also be included. Also, as described in embodiment 1, the error correction codes used for s1 and s2 may differ (Table 1 is provided for cases where a single type of error correction codes is used, as in Figure 4) . Additionally, the same modulation method and coding rate with different phase change patterns can be used. The transmission device transmits information indicating the phase change patterns to the receiving device. The receiving device specifies the phase change pattern by cross-reference to the information and to Table 1, then performs demodulation and decoding. When the modulation method and the error correction method determine a single phase change pattern, then whenever the transmission device transmits the modulation method and information regarding the error correction method, the receiving device

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Know the phase change pattern by obtaining that information. As such, the information pertaining to the phase change pattern is not strictly necessary.

In embodiments 1 to 3, the phase change is applied to the precoded baseband signals. However, the amplitude can also be modified along with the phase to apply regular periodic changes. Therefore, an amplification modification pattern that regularly modifies the amplitude of the modulated signals according to Table 1 can also be made. In such circumstances, the transmission device should include an amplification modifier that modifies the amplification after the unit of Weighting 308A or the weighting unit 308B from Figure 3 or 4. In addition, the amplification modification can be performed in only one or both of the precoded baseband signals z1 (t) and z2 (t) (in in the latter case, the amplification modifier is only necessary after one of the weighing unit 308A and 308B).

Additionally, although not indicated in Table 1 above, the mapping scheme can also be modified regularly by means of the mapper, without a regular phase change.

That is, when the mapping method for the modulated signal s1 (t) is 16-QAM and the mapping method for the modulated signal s2 (t) is also 16-QAm, the mapping method applied to the modulated signal s2 ( t) can be changed regularly as follows: from 16-QAM to 16-APSK, to 16-QAm in the IQ plane, to a first mapping method that produces a signal point distribution unlike 16-APSK, to 16- QAM in the IQ plane, to a second mapping method that produces a signal point distribution unlike 16-APSK, and so on! successively. As such, the quality of data reception can be improved for the receiving device, as well as the results obtained by a regular phase change described above.

In addition, the present invention can use any combination of methods for a regular phase change, mapping method and amplitude, and the transmission signal can be transmitted taking all these into account.

The present embodiment can be performed using single carrier methods as well! as multi-carrier methods. Accordingly, the present embodiment can also be performed using, for example, spread spectrum communications, OFDM, SC-FDM, SC-OFDM, particle OFDM as described in Non-patent Bibliography 7, and so on! successively. As described above, the present embodiment describes changing the phase, amplitude and mapping methods by making modifications of phase, amplitude and mapping method with respect to the time domain t. However, like embodiment 1, the same changes can be made with respect to the frequency domain. That is, considering the phase modification, amplitude and method of mapping in the time domain t described in the present embodiment and substituting t for f (where f is the frequency of ((sub-) carrier)) leads to phase modification, amplitude and mapping method applicable to the frequency domain. Also, the modification of phase, amplitude and method of mapping of the present embodiment is also applicable to modification of phase, amplitude and method of mapping in both the time domain and the frequency domain.

Additionally, in the present embodiment, the symbols other than the data symbols, such as pilot symbols (preamble, unique word, etc.) or the symbols that transmit control information, can be arranged in the frame in any way.

[Embodiment A1]

The present embodiment describes a method for regularly changing the phase when coding is performed using block codes as described in Non-patent Bibliography 12 to 15, such as LDPC Codes (not only QC-LDPC but also LDPC codes can be used ) QC (Quasi-Cyclic), concatenated LDPC and BCH codes (Bose-Chaudhuri-Hocquenghem), Turbo codes or Turbo Duo-binary codes that use tail bits, and so on! successively. The following example considers a case where two flows s1 and s2 are transmitted. When coding has been carried out using block codes and control information and they are not necessary, the number of bits that make up each block encoded coincides with the number of bits that make up each block code (even control information can be included and thus ! successively described below). When coding has been performed using block codes or the like and control information or the like is required (eg, CRC transmission parameters), then the number of bits that make up each encoded block is the sum of the number of bits that make up the block codes and the number of bits that make up the information.

Figure 34 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used. Figure 34 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams s1 and s2 are transmitted as indicated by the transmission device from Figure 4, and The transmission device has only one encoder. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 34, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

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Next, since the transmission device from Figure 4 transmits two streams simultaneously, 1500 of the 3,000 above-mentioned symbols needed when the modulation method is QPSK are assigned to each other and the other 1500 symbols are assigned to s2. As such, 1500 intervals are required to transmit the 1500 symbols (hereinafter intervals) for each of s1 and s2.

By the same reasoning, when the modulation method is 16-QAM, 750 intervals are necessary to transmit all the bits that make up each encoded block, and when the modulation method is 64-QAM, 500 intervals are necessary to transmit all the bits that make up each coded block.

The following describes the relationship between the previously defined intervals and the multiplication phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change. That is, five different phase change values (or phase change assemblies) have been prepared for the phase changer of the transmission device from Figure 4 (equivalent to the period (cycle) from embodiments 1 to 4) (As in Figure 6, five phase change values are necessary to perform a phase change with a period (cycle) of five in the precoded baseband signal z2 'only. Also, as in Figure 26, two phase change values are required for each interval to perform the phase change in both precoded baseband signals z1 'and z2'. These two phase change values are called a phase change set. ideally prepare five sets of phase change to perform a phase change that has a period (cycle) of five in such circumstances). These five phase change values (or phase change sets) are expressed as PHASE [0], PHASE [1], PHASE [2], PHASE [3] and PHASE [4].

For the 1500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is QPSk, PHASE [0] is used in 300 intervals, PHASE [1] is used in 300 intervals, used PHASE [2] in 300 intervals, PHASE [3] is used in 300 intervals and PHASE [4] is used in 300 intervals. This is due to the fact that any deviation in phase use causes enormous influence to be exerted by the most frequently used phase, and that the reception device is dependent on such influence for the quality of data reception.

Moreover, for the 500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is 64-QAM, PHASE [0] is used in 150 intervals, PHASE [1] is used in 150 intervals, PHASE [2] is used in 150 intervals, PHASE [3] is used in 150 intervals and FASe [4] is used in 150 intervals.

Moreover, for the 500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is 64-QAM, PHASE [0] is used in 100 intervals, PHASE [1] is used in 100 intervals, PHASE [2] is used in 100 intervals, PHASE [3] is used in 100 intervals and FASe [4] is used in 100 intervals.

As described above, a method for a regular phase change requires the preparation of N phase change values (or phase change sets) (where the N different phases are expressed as PHASE [0], PHASE [1] , PHASE [2] ... PHASE [N-2], PHASE [N-1]). As such, to transmit all the bits that make up a single encoded block, PHASE [0] is used in K0 intervals, PHASE [1] is used in K1 intervals, PHASE [i] is used in K1 intervals (where i = 0, 1, 2 ... N-1), and PHASE [N-1] is used in Kn-1 intervals, so that Condition No. A01 is met.

(Condition # A01)

K0 = K1 ... = Ki = ... Kn-1. That is, Ka = Kb (Va and Vb where a, b, = 0, 1.2 ... N-1; (being an integer not less than zero and not greater than N-1) a t b).

Then, when a communication system that supports multiple modulation methods selects a supported modulation method of this type for use, Condition No. A01 must be met for the supported modulation method.

However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Condition No. A01 may not be satisfied for some modulation schemes. In such a case, the following condition applies in place of Condition No. A01.

(Condition # A02)

The difference between Ka and Kb must be 0 or 1. That is, | Ka - Kb | must be 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1, a t b)

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Figure 35 illustrates the variable numbers of slots and intervals required in each coded block when block codes are used. Figure 35 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams are transmitted if and s2 as indicated by the transmission device from Figure 3 and the Figure 12, and the transmission device has two encoders. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 35, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

The transmission device from Figure 3 and the transmission device from Figure 12 each transmit two streams at once, and have two encoders. As such, the two streams each transmit different blocks of code. Therefore, when the modulation method is QPSK, two blocks extracted from s1 and s2 are transmitted in the same interval, for example, a first coded block extracted from s1 is transmitted, then a second coded block extracted from s2 is transmitted. As such, 3000 intervals are necessary to transmit the first and second coded blocks.

By the same reasoning, when the modulation scheme is 16-QAM, 1500 intervals are necessary to transmit all the bits that make up the two encoded blocks, and when the modulation scheme is 64-QAM, 1000 intervals are necessary to transmit all the bits that make up the two coded blocks.

The following describes the relationship between the previously defined intervals and the multiplication phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change. That is, five different phase change values (or phase change sets) have been prepared for the phase changer of the transmission device from Figures 3 and 12 (equivalent to the period (cycle) from the embodiments 1 to 4) (As in Figure 6, five phase change values are necessary to perform a phase change with a period (cycle) of five in the precoded baseband signal z2 'only. Also, as in Figure 26, two phase change values are required for each interval to perform the phase change in both precoded baseband signals z1 'and z2'. These two phase change values are called a phase change set. , five sets of phase change should ideally be prepared to perform a phase change that has a period (cycle) of five in such circumstances). These five phase change values (or phase change sets) are expressed as PHASE [0], PHASE [1], PHASE [2], PHASE [3] and PHASE [4].

For the 3000 intervals described above necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation method is QPSK, PHASE [0] is used in 600 intervals, PHASE [1] is used in 600 intervals, used PHASE [2] in 600 intervals, PHASE [3] is used in 600 intervals and PHASE [4] is used in 600 intervals. This is due to the fact that any deviation in phase use causes enormous influence to be exerted by the most frequently used phase, and that the receiving device is dependent on such influence for the quality of data reception.

Additionally, to transmit the first encoded block, PHASE [0] is used at 600 times intervals, the

PHASE [1] at intervals 600 times, PHASE [2] is used at intervals 600 times, PHASE [3] is used at

600 times intervals and PHASE [4] is used at 600 times intervals. Additionally, to transmit the second encoded block, PHASE [0] is used at 600 times intervals, PHASE [1] is used at 600 times intervals, PHASE [2] is used at 600 times intervals, used PHASE [3] at intervals 600 times and PHASE [4] is used at intervals 600 times.

Similarly, for the 1500 intervals described above necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation method is 16-QAM, PHASE [0] is used in 300

intervals, PHASE [1] is used in 300 intervals, PHASE [2] is used in 300 intervals, PHASE [3] is used in 300

intervals and PHASE [4] is used in 300 intervals.

Additionally, to transmit the first encoded block, PHASE [0] is used at intervals 300 times, the

PHASE [1] at intervals 300 times, PHASE [2] is used at intervals 300 times, PHASE [3] is used at

intervals 300 times and PHASE [4] is used at intervals 300 times. Additionally, to transmit the second encoded block, PHASE [0] is used at 300 times intervals, PHASE [1] is used at 300 times intervals, PHASE [2] is used at 300 times intervals, used PHASE [3] at intervals 300 times and PHASE [4] is used at intervals 300 times.

Similarly, for the 1000 intervals described above necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation method is 64-QAM, PHASE [0] is used in 200

intervals, the fAsE [1] is used in 200 intervals, the PHASE [2] is used in 200 intervals, the PHASE [3] is used in 200

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intervals and PHASE [4] is used in 200 intervals.

Additionally, to transmit the first encoded block, PHASE [0] is used at 200 times intervals, PHASE [1] is used at 200 times intervals, PHASE [2] is used at 200 times intervals, used PHASE [3] at intervals 200 times and PHASE [4] at intervals 200 times is used. Additionally, to transmit the second coded block, PHASE [0] is used at 200 times intervals, PHASE [1] is used at 200 times intervals, PHASE [2] is used at 200 times intervals, used PHASE [3] at intervals 200 times and PHASE [4] at intervals 200 times is used.

As described above, a method for regularly changing the phase requires the preparation of phase change values (or phase change sets) expressed as PHASE [0], PHASE [1], PHASE [2] ... PHASE [N-2], PHASE [N-1]. As such, to transmit all the bits that make up two coded blocks, PHASE [0] is used in K0 intervals, PHASE [1] is used in K1 intervals, PHASE [i] is used in Ki intervals (where i = 0, 1, 2 ... N-1), and PHASE [N-1] is used in Kn-1, intervals, so that Condition No. A03 is met.

(Condition # A03)

K0 = K1 ... = Ki = ... Kn-1. That is, Ka = Kb (Va and Vb where a, b, = 0, 1, 2 ... N-1, (being an integer not less than zero and not greater than N-1) a t b).

In addition, to transmit all the bits that make up the first encoded block, PHASE [0] Kc is used, 1 times, PHASE [1] K11 times is used, PHASE [i] Ki is used, 1 times (where i = 0, 1, 2 ... N-1), and PHASE [N-1] Kn-1.1 times is used, so that Condition No. A04 is met.

(Condition # A04)

Kc, 1 = K11 = ... Ki, i = ... Kn-1,1. That is, Ka, 1 = Kb, 1 (Va and Vb where a, b, = 0, 1, 2 ... N-1, a t b).

Additionally, to transmit all the bits that make up the second encoded block, PHASE [0] K0.2 times is used, PHASE [1] K12 times is used, PHASE [i] Ki is used, 2 times (where i = 0, 1, 2 ... N-1), and PHASE [N-1] Kn-1.2 times is used, so that Condition No. A05 is met.

(Condition # A05)

Kq, 2 = K1,2 = ... Ki, 2 = ... Kn-1,2. That is, Ka, 2 = Kb, 2 (Va and Vb where a, b, = 0, 1, 2 ... N-1, a t b).

Then, when a communication system that supports multiple modulation methods selects a supported modulation method of this type for use, Condition # A03, # A04, and # A05 must be met for the supported modulation method .

However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Conditions No. A03, n A04 and A05 may not be satisfied by some method of modulation. In such a case, the following conditions apply in place of Condition No. A03, No. A04 and No. A05.

(Condition # A06)

The difference between Ka and Kb satisfies 0 or 1. That is, | Ka - Kb | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1,2 ... N-1, a t b)

(Condition # A07)

The difference between Ka, 1 and Kb, 1 satisfies 0 or 1. That is, | Ka, 1 - Kb, 1 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1, a t b)

(Condition # A08)

The difference between Ka, 2 and Kb, 2 satisfies 0 or 1. That is, | Ka, 2 - Kb, 2 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1, a t b)

As described above, the deviation between the phases that are used to transmit the encoded blocks is eliminated by creating a relationship between the encoded block and the multiplication phase. As such, the quality of data reception for the reception device can be improved.

In the present embodiment, N phase change values (or phase change sets) are necessary to perform a phase change that has a period (cycle) of N with the method for a regular phase change. How

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such, N phase change values (or phase change sets) are prepared PHASE [0], PHASE [1], PHASE [2] ... PHASE [N-2] and PHASE [N-1]. However, there are schemes to reorder the phases in the order established with respect to the frequency domain. No limitation is intended in this regard. The N phase change values (or phase change sets) can also change the phases of the blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement as described in embodiment 1. Although the above examples analyze a phase change method with a period (cycle) of N, the same effects are obtainable using N phase change values (or phase change sets) randomly. That is, the N phase change values (or phase change sets) do not always have to have regular periodicity. Provided that the conditions described above are satisfied, huge improvements in receiving quality data for the receiving device are achievable.

Additionally, given the existence of modes for spatial multiplexing MIMO schemes, MIMO schemes that use a fixed precoding matrix, space-time block coding schemes, single stream transmission and schemes that use a regular phase change ( the transmission schemes described in embodiments 1 to 4), the transmission device (diffuser, base station) can select any one of these transmission schemes.

As described in Non-patent Bibliography 3, MIMO methods of spatial multiplexing involve transmitting signals s1 and s2, which are mapped using a selected modulation method, on each of two different antennas. As described in embodiments 1 to 4, MIMO methods using a fixed precoding matrix involve only precoding (without phase change). In addition, the space-time block coding methods are described in the non-patent Bibliography 9, 16 and 17. The single stream transmission methods involve transmitting the signal s1, mapped with a selected modulation method, from an antenna after performing default processing.

Schemes using multi-carrier transmission such as OFDM involve a first group of carriers composed of a plurality of carriers and a second group of carriers composed of a plurality of carriers different from the first group of carriers, and so on! successively, so that the multicarrier transmission is performed with a plurality of carrier groups. For each group of carriers, any of the spatial multiplexing MIMO schemes, MIMO schemes that use a fixed precoding matrix, space-time block coding schemes, single stream transmission, and schemes that use a change in time may be used. regular phase In particular, schemes that use a regular phase change in a selected (sub-) carrier group are preferably used to perform the present embodiment.

When a phase change is made, then for example, a phase change value is made for the PHASE [i] of X radians in only one precoded baseband signal, the phase changers of Figures 3, 4, 5, 12, 25, 29, 51 and 53 multiply the precoded baseband signal z2 'by ex. Then, when a phase change is made by, for example, a phase change set for the PHASE [i] of X radians and Y radians in both precoded baseband signals, the phase changers from the Figures 26, 27, 28, 52 and 54 multiply the precoded baseband signal z2 'by ejX and multiply the precoded baseband signal z1' by ejY.

[Realization B1]

The following describes an example configuration of an application of the transmission methods and reception methods discussed in the previous embodiments and a system using the application.

Figure 36 illustrates the configuration of a system that includes devices that execute transmission methods and reception methods described in the previous embodiments. As shown in Figure 36, the devices that execute transmission methods and reception methods described in the previous embodiments include various receivers such as a diffuser, a television 3611, a DVD recorder 3612, an STB (room decoder) 3613 , a 3620 computer, a 3641 vehicle mounted television, a 3630 mobile phone and so on! successively in a digital broadcast system 3600. Specifically, the diffuser 3601 uses a transmission method analyzed in the previously described embodiments to transmit multiplexed data, in which video, audio and other data is multiplexed, through a predetermined transmission band .

The signals transmitted by the diffuser 3601 are received by an antenna (such as antenna 3660 or 3640) embedded in or connected externally to each of the receivers. Each receiver obtains the multiplexed data using reception methods analyzed in the previously described embodiments to demodulate the signals received by the antenna. Accordingly, the digital broadcast system 3600 can realize the effects of the present invention, as discussed in the previously described embodiments.

The video data included in the multiplexed data is encoded with a video coding method that complies with a standard such as MPEG-2 (Group of Experts in Motion Images), MPEG4-AVC (Advanced Video Coding), VC- 1 or similar. The audio data included in the multiplexed data is encoded with an audio coding method that complies with a standard such as Dolby AC-3.

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Audio), Dolby Digital Plus, MLP (Meridian Lossless Packaging), DTS (Digital Cinema Systems), DTS-HD, Linear PCM (Coded Pulse Modulation) or the like.

Figure 37 illustrates the configuration of a 7900 receiver that executes a reception method described in the previously described embodiments. The receiver 3700 corresponds to a receiver included in one of the television 3611, the DVD recorder 3612, the STB 3613, the computer 3620, the television mounted in vehicle 3641, the mobile phone 3630 and so! successively from Figure 36. The receiver 3700 includes a tuner 3701 that converts a high frequency signal received by an antenna 3760 into a baseband signal, and a demodulator 3702 that demodulates the baseband signal as! converted to obtain multiplexed data. The demodulator 3702 executes a reception method analyzed in the previously described embodiments, and thus achieves the effects of the present invention as explained above.

Receiver 3700 additionally includes a stream interface 3720 that demultiplexes audio and video data into multiplexed data obtained by demodulator 3702, a signal processor 3704 that decodes video data obtained from video data demultiplexed into a video signal applying a video decoding method that corresponds to them and decodes the audio data obtained from the demultiplexed audio data in an audio signal applying an audio decoding method corresponding to them, a Audio output unit 3706 that emits the decoded audio signal through a speaker or the like, and a video presentation unit 3707 that outputs the decoded video signal on a screen or the like.

When, for example, a user uses a 3750 remote control, the information for a selected channel (selected program (television) or audio broadcast) is transmitted to an operation input unit 3710. Next, the receiver 3700 performs processing in the received signal received by the 3760 antenna that includes demodulating the signal corresponding to the selected channel, performing error correction decoding, and so on! successively, to obtain the data received. At this point, receiver 3700 obtains control symbol information that includes information on the transmission method (the transmission method, modulation method, error correction method, and so on from the previously described embodiments) (as described above). described using Figures 5 and 41) from control symbols included in the signal corresponding to the selected channel. As such, the receiver 3700 can correctly establish the reception operations, the demodulation method, the error correction method and so on! successively, thus enabling the data included in the data symbols transmitted by the diffuser (base station) to be obtained. Although the above description is provided for an example of the user using the 3750 remote control, the same operations apply when the user presses a selection key embedded in the 3700 receiver to select a channel.

According to this configuration, the user can view programs received by the 3700 receiver.

The receiver 3700 belonging to the present embodiment additionally includes a unit 3708 which can be a magnetic disk, an optical disk, a non-volatile semiconductor memory, or a similar recording medium. Receiver 3700 stores data included in demultiplexed data obtained through demodulation by means of demodulator 3702 and error correction decoding (in some circumstances, data obtained through demodulation by means of demodulator 3702 may not be subject to correction of errors Also, the 3700 receiver can perform additional processing after the correction of errors.The same applies hereinafter to similar statements related to other components), data corresponding data of this type (for example, data obtained through compression of such data), data obtained through audio and video processing, and so on! successively, in unit 3708. At this point, an optical disc is a recording medium, such as DVD (Digital Versatile Disc) or BD (Blu-ray Disc), which is readable and writable with the use of a laser beam. A magnetic disk is a flexible disk, a hard disk or similar recording medium in which the information is storable through the use of magnetic flux to magnetize a magnetic body. A non-volatile semiconductor memory is a recording medium, such as flash memory or ferroelectric random access memory, composed of semiconductor element or elements. Specific examples of non-volatile semiconductor memory include an SD card that uses flash memory and a SSD (Solid State Drive) Flash. Obviously, the specific types of recording media mentioned herein are merely examples. Other types of recording media can also be used.

According to this structure, the user can record and store programs received by the 3700 receiver, and can thus display programs at any given time after broadcasting by reading the data recorded from it.

Although the above explanations describe the receiver 3700 that stores multiplexed data obtained through demodulation by means of demodulator 3702 and decoding error correction in unit 3708, a portion of the data included in the multiplexed data can be extracted and recorded in its place. For example, when data dissemination services or similar content are included along with the audio and video data in the multiplexed data obtained through demodulation by means of demodulator 3702 and error correction decoding, audio data and video can be extracted from the data

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multiplexed demodulated by demodulator 3702 and stored as new multiplexed data. Additionally, unit 3708 can store any of the audio data or video data included in the multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding as new multiplexed data. The above-mentioned data broadcast service content included in the multiplexed data can also be stored in unit 3708.

Additionally, when a television, recording device (for example, a DVD recorder, BD recorder, HDD recorder, SD card or similar), or mobile phone incorporating the 3700 receiver of the present invention receives multiplexed data multiplexed data obtained through demodulation using demodulator 3702 and error correction decoding that includes data to correct programming errors in software used to operate the television or recording device, to correct programming errors in software to prevent information loss personal and recorded data, and so on, such software programming errors can be corrected by installing the data on the television or recording device. As such, programming errors in receiver 3700 are corrected through the inclusion of data to correct programming errors in receiver software 3700. Accordingly, television, recording device or mobile phone incorporating receiver 3700 can be manufactured to operate more reliably.

At this point, the process for extracting a portion of the data included in the multiplexed data obtained through demodulation by means of demodulator 3702 and decoding error correction is performed by, for example, flow interface 3703. Specifically, Flow interface 3703 demultiplexes the various data included in the multiplexed data demodulated by demodulator 3702, such as audio data, video data, data broadcast service content, and so on, as ordered by a non-controller. diagrammed such as a CPU. The flow interface 3703 then extracts and multiplexes only the indicated demultiplexed data, thereby generating new multiplexed data. The data to be extracted from the demultiplexed data can be determined by the user or can be determined in advance according to the type of recording medium.

According to such a structure, the receiver 3700 can extract and record only the data necessary to view the recorded program. As such, the amount of data to be recorded can be reduced.

Although the above explanation describes the unit 3708 as storing multiplexed data obtained through demodulation by means of demodulator 3702 and decoding error correction, the video data included in the multiplexed data thus obtained can be converted using a coding method of video different from the original video coding method applied to them, to reduce the amount of data or the bit rate thereof. Unit 3708 can then store the converted video data as new multiplexed data. At this point, the video coding method used to generate the new video data can conform to a different standard than the one used to generate the original video data. Alternatively, the same video coding method can be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation through demodulator 3702 and error correction decoding can be converted using an audio coding method different from the original audio coding method applied to the same, to reduce the amount of data or the bit rate thereof. Unit 3708 can then store the converted audio data as new multiplexed data.

At this point, the process by which the audio or video data included in the multiplexed data obtained through demodulation through demodulator 3702 and error correction decoding are converted to reduce the amount of data or the rate of bits thereof are realized by, for example, the flow interface 3703 or the serial processor 3704. Specifically, the flow interface 3703 demultiplexes the various data included in the multiplexed data demodulated by demodulator 3702, such as audio data , video data, data broadcast service content, and so on, as ordered by an ungrammed controller such as a CPU. The serial 3704 processor then performs processing to convert the video data thus demultiplexed using a video encoding method different from the original video encoding method applied thereto, and performs processing to convert the audio data thus demultiplexed using a method video encoding different from the original audio coding method applied to them. As ordered by the controller, the flow interface 3703 then multiplexes the converted audio and video data, thereby generating new multiplexed data. The serial 3704 processor can, according to instructions from the controller, perform conversion processing on any of the video data or audio data, alone, or it can perform conversion processing on both types of data. In addition, the amounts of video data and audio data or the bit rate thereof to be obtained by conversion can be specified by the user or determined in advance according to the type of recording medium.

According to such a structure, the receiver 3700 can modify the amount of data or the bit rate of the audio and video data to be stored according to the data storage capacity of the recording medium or according to the data read or write speed of unit 3708. Therefore, programs can be stored in the unit despite the storage capacity of the media

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recording is less than the amount of multiplexed data obtained through demodulation by means of demodulator 3702 and decoding error correction, or the speed of reading or writing data from the unit is less than the bit rate of the data demultiplexes obtained through demodulation through demodulator 3702. As such, the user can watch programs at any given time after broadcasting by reading the recorded data.

The receiver 3700 additionally includes a flow output interface 3709 that transmits the multiplexed data demultiplexed by demodulator 3702 to external devices through a communication means 3730. The flow output interface 3709 can be, for example, a device for Wireless communication that transmits multiplexed modulated data to an external device using a wireless transmission method conforming to a wireless communication standard such as Wi-Fi ™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE 802.11n, and so! successively), WiGiG, WirelessHD, Bluetooth ™, Zig-Bee ™, and so on! successively through a wireless medium (corresponding to communications medium 3730). The flow output interface 3709 may also be a wired communication device that transmits modulated multiplexed data to an external device using a communication method conforming to a wired communication standard such as Ethernet ™, USB (Universal Serial Bus), PLC ( Communication via Electric Line), HDMI (High Definition Multimedia Interface) and so! successively through a wired transmission path (corresponding to communications medium 3730) connected to the flow output interface 3709.

According to this configuration, the user can use an external device with multiplexed data received by the 3700 receiver using the reception method described in the previously described embodiments. The use of multiplexed data by the user at this point includes the use of multiplexed data to visualize in real time on an external device, recording of multiplexed data by means of a recording unit included in an external device, and transmission of multiplexed data from an external device to another external device.

Although the above explanations describe the receiver 3700 that emits multiplexed data obtained through demodulation by means of demodulator 3702 and decoding error correction through flow output interface 3709, a portion of the data included in the data may be extracted. data multiplexed instead and issued. For example, when data dissemination services or similar content are included along with the audio and video data in the multiplexed data obtained through demodulation by means of demodulator 3702 and error correction decoding, audio data and Video can be extracted from the multiplexed data obtained through demodulation by means of demodulator 3702 and decoding error correction, multiplexed and output via flow output interface 3709 as new multiplexed data. In addition, the stream output interface 3709 can store any of the audio data or video data included in the multiplexed data obtained through demodulation by means of demodulator 3702 and error correction decoding as new multiplexed data.

At this point, the process for extracting a portion of the data included in the multiplexed data obtained through demodulation by means of demodulator 3702 and decoding error correction is performed by, for example, flow interface 3703. Specifically, The flow interface 3703 demultiplexes the various data included in the multiplexed data demodulated by demodulator 3702, such as audio data, video data, data broadcast service content, and so on, as ordered by an ungrammed controller. such as a CPU. The flow interface 3703 then extracts and multiplexes only the indicated demultiplexed data, thereby generating new multiplexed data. The data to be extracted from the demultiplexed data can be determined by the user or can be determined in advance according to the type of flow output interface 3709.

According to this structure, the receiver 3700 can only extract and emit the required data to an external device. As such, less multiplexed data is emitted using less communication bandwidth.

Although the above explanation describes flow output interface 3709 as it emits multiplexed data obtained through demodulation by demodulator 3702 and error correction decoding, video data included in the multiplexed data thus obtained can be converted using a method of video coding different from the original video coding method applied to them, to reduce the amount of data or the bit rate thereof. The stream output interface 3709 can then output the converted video data as new multiplexed data. At this point, the video coding method used to generate the new video data can conform to a different standard than the one used to generate the original video data. Alternatively, the same video coding method can be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation through demodulator 3702 and error correction decoding can be converted using an audio coding method different from the original audio coding method applied to the same, to reduce the amount of data or the bit rate thereof. The stream output interface 3709 can then output the converted audio data as new multiplexed data.

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At this point, the process by which the audio or video data included in the multiplexed data obtained through demodulation through demodulator 3702 and error correction decoding are converted to reduce the amount of data or the rate of bits thereof are realized by, for example, the flow interface 3703 or the signal processor 3704. Specifically, the flow interface 3703 demultiplexes the various data included in the multiplexed data demodulated by demodulator 3702, such as audio data , video data, data broadcast service content, and so on, as ordered by an ungrammed controller. The signal processor 3704 below performs processing to convert the video data thus demultiplexed using a video encoding method different from the original video coding method applied thereto, and performs processing to convert the audio data thus demultiplexed using a video coding method different from the original audio coding method applied to them. As ordered by the controller, the flow interface 3703 then multiplexes the converted audio and video data, thereby generating new multiplexed data. The signal processor 3704 can, according to instructions from the controller, perform conversion processing on any of the video data or audio data, alone, or it can perform conversion processing on both types of data. In addition, the amounts of video data and audio data or the bit rate thereof to be obtained by conversion can be specified by the user or determined in advance according to the type of flow output interface 3709.

According to this structure, the receiver 3700 can modify the bit rate of the video and audio data to be broadcast according to the communication speed with the external device. Therefore, despite the fact that the communication speed with an external device is slower than the bit rate of multiplexed data obtained through demodulation through demodulator 3702 and error correction decoding, issuing new multiplexed data From the flow output interface to the external device, the user can use the new data multiplexed with other communication devices.

The receiver 3700 additionally includes an audiovisual output interface 3711 that emits audio and video signals decoded by the signal processor 3704 to the external device through an external communications medium. The audiovisual output interface 3711 can be, for example, a wireless communication device that transmits modulated audiovisual data to an external device using a wireless transmission method according to a wireless communication standard such as Wi-Fi ™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE 802.11n, and so on), WiGig, WirelessHD, Bluetooth ™, ZigBee ™, and so on through a wireless medium. The flow output interface 3709 may also be a wired communication device that transmits modulated audiovisual data to an external device using a communication method in accordance with a wired communication standard such as Ethernet ™, USB, PLC, HDMI, and so on through a wired transmission path connected to the flow output interface 3709. Additionally, the flow output interface 3709 can be a terminal for connecting a cable that emits audio signals and analog video signals as is.

According to such a structure, the user can use the audio signals and the video signals decoded by the signal processor 3704 with an external device.

In addition, receiver 3700 includes an operation input unit 3710 that receives user operations as input. The receiver 3700 behaves in accordance with control signals introduced by the operation input unit 3710 according to user operations, such as CONNECT or DISCONNECT the power supply, change the channel being received, CONNECT or DISCONNECT the display subtitle, switch between languages, change the volume output using the 3706 audio output unit, and various other operations, including changing the settings for channels that can be received and the like.

The receiver 3700 may additionally include functionality to present an antenna level that represents the quality of the signal received while the receiver 3700 is receiving a signal. The antenna level can be, for example, an index that presents the received signal quality calculated according to the RSSI (Received Signal Intensity Indicator), the received magnetic signal strength, the C / N ratio (carrier to noise), the BER, the packet error rate, the frame error rate, the channel status information, and so on, received by the 3700 receiver and indicating the level and quality of a received signal. In such circumstances, demodulator 3702 includes a signal quality calibrator that measures the RSSI, the received magnetic field strength, the C / N ratio, the BER, the packet error rate, the frame error rate. , the channel status information, and so on. In response to user operations, the receiver 3700 presents the antenna level (signal level, signal quality) in a format recognizable by the user in the video display unit 3707. The display format for the antenna level ( signal level, signal quality) can be a numerical value presented in accordance with the RSSI, the received magnetic field strength, the C / N ratio, the BER, the packet error rate, the error rate of plot, the channel status information, and so on, or it can be an image display that varies according to the RSSI, the received magnetic field strength of the signal, the C / N ratio, the BER, the error rate of packets, frame error rate, channel status information, and so on. The receiver 3700 may have the level of multiple antennas (signal level, signal quality) calculated for each flow s1, s2, and so on demultiplexed using the reception method analyzed in the previously described embodiments, or it may

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present a single antenna level (signal level, signal quality) calculated for all such flows. When the video data and audio data that make up a program are transmitted hierarchically, the signal level (signal quality) can also be presented for each hierarchical level.

According to the above structure, the user is provided with an understanding of the antenna level (signal level, signal quality) numerically or visually during reception using the reception methods analyzed in the previously described embodiments.

Although the above example describes the receiver 3700 as including the audio output unit 3706, the video display unit 3707, the unit 3708, the flow output interface 3709, and the audio-visual output interface 3711, all these components They are not strictly necessary. Provided that the receiver 3700 includes at least one of the components described above, the user may use the multiplexed data obtained through demodulation by means of demodulator 3702 and error correction decoding. Any receiver can be freely combined with the components described above according to the method of use.

(Multiplexed data)

The following is a detailed description of a multiplexed data sample configuration. The configuration of data normally used in broadcasting is a transport stream (TS) of MPEG-2. Therefore the following description describes an example related to MPEG2-TS. However, the data configuration of multiplexed data transmitted by the transmission and reception methods discussed in the above described embodiments is not limited to MPEG2-TS. The advantageous effects of the embodiments described above are also achievable using any other data structure.

Figure 38 illustrates a sample configuration for multiplexed data. As shown, multiplexed data are elements that make up programs (or events, which are a portion of them) currently provided by various services. For example, one or more video streams, audio streams, presentation chart streams (PG), interactive chart streams (IG) and other such item streams are multiplexed to obtain multiplexed data. When a broadcast program provided by multiplexed data is a movie, the video streams represent the main video and the sub video of the movie, the audio streams represent the main audio of the movie and the sub-audio to be mixed with the main audio, and the presentation graphics streams represent subtitles for the movie. The main video refers to video images normally presented on a screen, while sub-video refers to video images (for example, text images that explain the summary of the movie) to be presented in a small window inserted in the video images The interactive graphic streams represent an interactive presentation composed of the GUI (Graphical User Interface) components presented on a screen.

Each stream included in the multiplexed data is identified by an identifier, called a PID, uniquely assigned to the stream. For example, the PID 0x1011 is assigned to the video stream used for the main video of the movie, the PID 0x1100 to 0x1nF is assigned to the audio streams, the PIDs 0x1200 to 0x121F are assigned to the presentation graphics, the PID 0x1400 at 0x141F they are assigned to the interactive graphics, the PIDs 0x1B00 to 0x1B1F are assigned to the video streams used for the sub-video of the movie, and the PIDs 0x1A00 to 0x1A1 F are assigned to the audio streams used as sub-audio to mix with the main audio of the movie.

Figure 39 is a schematic diagram illustrating an example of multiplexed data that is multiplexed. First, a video stream 3901, composed of a plurality of frames, and an audio stream 3904, composed of a plurality of audio frames, respectively become the sequence of PES packets 3902 and 3905, then additionally they become TS 3903 and 3906 packages. Similarly, a flow of presentation graphics 3911 and a flow of interactive graphics 3914 respectively become the sequence of PES packages 3912 and 3915, then they are further converted into packages of TS 3913 and 3916. The multiplexed data 3917 are composed of the packages of TS 3903, 3906, 3913 and 3916 multiplexed in a single stream.

Figure 40 illustrates additional details of a sequence of PES packets as contained in the video stream. The first level of Figure 40 shows a sequence of video frames in the video stream. The second level shows a sequence of PES packages. The arrows yy1, yy2, yy3 and yy4 indicate the plurality of Video Presentation Units, which are instantaneous I, instantaneous B and instantaneous P, in the video stream as it is divided and stored individually as the useful space of a PES package . Each PES package has a PES heading. A PES header contains a PTS (Display Time Indication) in which the instant is to be presented, a DTS (Decode Time Indication) in which the instant is to be decoded, and so on.

Figure 41 illustrates the structure of a TS packet as it is finally written in multiplexed data. A TS packet is a 188-byte fixed-length packet composed of a 4-byte PID that identifies the flow and an 184-byte TS payload that contains the data. The PES packages described above are divided and stored individually as the useful capacity of TS. For a BD-ROM, each TS package has a

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Extra_TP_header fixed to 4 bytes set to it to create a 192-byte source packet, which should be written as multiplexed data. TheTP_Extra_Header contains information such as an Arrived_Time_Indication (ATS). The ATS indicates a time to start the transfer of the TS packet to the PID filter of a decoder. The multiplexed data is composed of source packets arranged as indicated in the lower level of Figure 41. An SPN (source packet number) is incremented for each packet beginning at the header of the multiplexed data.

In addition to video streams, audio streams, presentation graphics streams, and the like, the TS packages included in the multiplexed data also include a PAT (Program Association Table), a PMT (Program Map Table) , a PCR (Program Clock Reference) and as! successively. The PAT indicates the PID of a PMT used in the multiplexed data, and the PID of the PAT itself is recorded as 0. The PMT includes the PIDs that identify the respective flows, such as video, audio and subtitles, contained in the multiplexed data and attribute information (frame rate, aspect ratio and the like) of the flows identified by the respective PIDs. In addition, the PMT includes various types of descriptors related to multiplexed data. A descriptor of this type may be copy control information indicating whether or not it is allowed to copy multiplexed data. The PCR includes information to synchronize the aTc (Arrival Time Clock) that serves as the chronological axis of the ATS to the STC (System Time Clock) that serves as the chronological axis of the PTS and DTS. Each PCR packet includes an STC time that corresponds to the ATS in which the packet is to be transferred to the decoder.

Figure 42 illustrates the detailed data configuration of a PMT. The PMT starts with a PMT header that indicates the length of the data contained in the PMT. After the PMT heading, the descriptors that belong to the multiplexed data are arranged. An example of a descriptor included in the PMT is the copy control information described above. After the descriptors, the flow information that belongs to the respective flows included in the multiplexed data is arranged. Each piece of flow information is composed of flow descriptors that indicate a type of flow that identifies a compression codec used for a corresponding flow, a PID for the flow and attribute information (frame rate, aspect ratio and the like) of the flow. The PMT includes the same number of flow descriptors as the number of flows included in the multiplexed data.

When recording on a recording medium or the like, multiplexed data is recorded together with a multiplexed data information file.

Figure 43 illustrates a sample configuration for the multiplexed data information file. As shown, the multiplexed data information file is management information for the multiplexed data, is provided in one-to-one correspondence with the multiplexed data, and is composed of multiplexed data information, flow attribute information and a map. input

The multiplexed data information is composed of a system speed, a playback start time and an end play time. The system speed indicates the maximum transfer rate of the multiplexed data to the PID filter of an objective decoder of the system described below. Multiplexed data includes the ATS in a set interval to not exceed the system speed. The playback start time is set to the time specified by the PTS of the first video frame in the multiplexed data, while the playback end time is set to the time calculated by adding the playback duration of a frame to the PTS of the Last video frame in multiplexed data.

Figure 44 illustrates a sample configuration for the flow attribute information included in the multiplexed data information file. As shown, the flow attribute information is attribute information for each flow included in the multiplexed data, recorded for each PID. That is, different pieces of attribute information are provided for different streams, in particular for video streams, audio streams, presentation chart streams and interactive chart streams. The video stream attribute information indicates the compression codec used to compress the video stream, the resolution of the individual instants that constitute the video stream, the aspect ratio, the frame rate, and so on. The audio stream attribute information indicates the compression codec used to compress the audio stream, the number of channels included in the audio stream, the language of the audio stream, the sampling frequency, and so on. This information is used to initialize the decoder before playing through a player.

In the present embodiment, the type of flow included in the PMT is used among the information included in the multiplexed data. When multiplexed data is recorded on a recording medium, the video stream attribute information included in the multiplexed data information file is used. Specifically, the video coding method and the device described in any of the above embodiments can be modified to further include a stage or unit to adjust a specific piece of information in the type of flow included in the PMT or in the attribute attribute information. video stream The specific piece of information is to indicate that video data is generated using the video coding method and the device described in

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the realization. According to such a structure, the video data generated by the video coding method and the device described in any of the above embodiments are distinguishable from the video data that complies with other standards.

Figure 45 illustrates a sample configuration of an audiovisual output device 4500 that includes a receiving device 4504 that receives a modulated signal that includes video and audio data transmitted by a diffuser (base station) or data intended for diffusion. The configuration of the reception device 4504 corresponds to the reception device 3700 from Figure 37. The audiovisual output device 4500 incorporates, for example, an OS (Operating System), or incorporates a communication device 4506 to connect to the internet ( for example, a communication device intended for a wireless LAN (Local Area Network) or for Ethernet ™). As such, a video presentation unit 4501 can simultaneously present video and audio data, or video in video data for broadcast 4502, and hypertext 4503 (from the Global Computer Network) provided through the internet. By operating a 4507 remote control (as an alternative, a mobile phone or keypad), any of the video in the video data to broadcast 4502 and the hypertext 4503 provided through the internet can be selected to change operations. For example, when the hypertext 4503 provided through the internet is selected, the presented website can be changed by remote control operations. When the video and audio data, or the video in the video data is selected to broadcast 4502, the information from a (selected program (television) or audio broadcast) can be transmitted by the remote control 4507. As such, a interface 4505 obtains the information transmitted through the remote control. The receiving device 4504 performs processing such as demodulation and error correction corresponding to the selected channel, thus obtaining the received data. At this point, the receiving device 4504 obtains the control symbol information that includes information on the transmission method (as described using Figure 5) from control symbols included in the signal corresponding to the selected channel. As such, the reception device 4504 can correctly establish the reception operations, demodulation method, the error correction method and so on, thereby enabling the data included in the data symbols transmitted by the diffuser to be obtained ( Base station). Although the description above is provided for an example of the user using the 4507 remote control, the same operations apply when the user presses a selection key embedded in the audiovisual output device 4500 to select a channel.

In addition, the audiovisual output device 4500 can be operated using the internet. For example, the audiovisual output device 4500 can be manufactured to record (store) a program through another terminal connected to the internet. (Consequently, the audiovisual output device 4500 should include unit 3708 from Figure 37). The channel is selected before recording begins. As such, the receiving device 4504 performs processing such as demodulation and error correction corresponding to the selected channel, thereby obtaining the received data. At this point, the receiving device 4504 obtains control symbol information that includes information on the transmission method (the transmission method, modulation method, the error correction method, and so on from the previously described embodiments) ( as described using Figure 5) from control symbols included in the signal corresponding to the selected channel. As such, the reception device 4504 can correctly establish the reception operations, demodulation method, the error correction method and so on, thereby enabling the data included in the data symbols transmitted by the diffuser to be obtained ( Base station).

(Supplement)

The present description considers a communications / broadcast device such as a broadcaster, a base station, an access point, a terminal, a mobile phone or the like provided with the transmission device, and a communications device such as a television, radio , terminal, personal computer, mobile phone, access point, base station or similar provided with the receiving device. The transmission device and the reception device belonging to the present invention are communication devices in a way that can execute applications, such as a television, radio, personal computer, mobile phone, or the like, through the connection of any type of interface (for example, USB).

Additionally, in the present embodiment, symbols other than data symbols, such as pilot symbols (specifically preamble, unique word, epilogue, reference symbols, scattered pilot symbols and so on), the intended symbols for control information, and so on they can be freely arranged in the plot. Although the pilot symbols and the symbols intended for control information are currently named, such symbols may be freely named in another way since their function remains the important consideration.

With the proviso that a pilot symbol is, for example, a known symbol modulated with PSK modulation at the transmitter and receiver (alternatively, the receiver can be synchronized so that the receiver knows the symbols transmitted by the transmitter), the receiver You can use this symbol for frequency synchronization, time synchronization, channel estimation (CSI estimation (Channel Status Information) for each modulated signal), signal detection and the like

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The intended slots for control information are slmboios that transmit information (such as the modulation method, error correction coding method, error correction code coding rate, and adjustment information for the upper layer used in communications) which must be transmitted to the receiving party to execute transmission of no data (i.e. applications).

The present invention is not limited to embodiments, but can also be done in various other ways. For example, although the above embodiments describe communication devices, the present invention is not limited to such devices and can be implemented as software for the corresponding communications method.

Although the embodiments described above describe phase change methods for methods for transmitting two modulated signals from two antennas, no limitation is intended in this regard. Precoding and a phase change can be performed on four signals that have been mapped to generate four modulated signals transmitted using four antennas. That is, the present invention is applicable to perform a phase change in N signals that are mapped and precoded to generate N modulated signals transmitted using N antennas.

Although the embodiments described above describe examples of systems where two modulated signals are transmitted from two antennas and received by two respective antennas in a MIMO communications system, the present invention is not limited in this regard and is also applicable to communications systems. MISO (Multiple Input Single Output). In a MISO system, the receiving device does not include the antenna 701_Y, the wireless unit 703_Y, the channel fluctuation estimator 707_1 for the modulated signal z1, and the channel fluctuation estimator 707_2 for the modulated signal z2 from Figure 7. However, the processing described in embodiment 1 can be executed even to estimate r1 and r2. The technology for receiving and decoding a plurality of signals transmitted simultaneously on a common frequency that are received by a single antenna is widely known. The present invention is additional processing that complements the conventional technology for a signal processor that reverses a phase changed by the transmitter.

Although the present invention describes examples of systems where two modulated signals are transmitted from two antennas and received by two respective antennas in a MIMO communications system, the present invention is not limited in this regard and is also applicable to MISO systems. In a MISO system, the transmission device performs precoding and phase change so that the points described so far are applicable. However, the receiving device does not include the antenna 701_Y, the wireless unit 703_Y, the channel fluctuation estimator 707_1 for the modulated signal z1, and the channel fluctuation estimator 707_2 for the modulated signal z2 from Figure 7 However, the processing described in the present description can be executed even to estimate the data transmitted by the transmission device. The technology for receiving and decoding a plurality of signals transmitted simultaneously on a common frequency that are received by a single antenna is widely known (a single antenna receiver can apply ML operations (Max-log APP or similar)). The present invention may have the signal processor 711 from Figure 7 which performs demodulation (detection) taking into account the precoding and phase change applied by the transmitter.

The present description uses terms such as precoding, precoding weights, precoding matrix and so on! successively. The terminology itself may be otherwise (for example, a code book may be referred to as an alternative) since the key point of the present invention is the signal processing itself.

Additionally, although the present description analyzes examples that primarily use OFDM as the method of transmission, the invention is not limited in this way. Multi-carrier methods other than OFDM and single carrier methods can all be used to achieve similar embodiments. At this point, spread spectrum communications can also be used. When single carrier methods are used, the phase change takes place with respect to the time domain.

In addition, although this description analyzes the use of ML, APP, Max-log APP, ZF, MMSE and so! successively by means of the reception device, all these operations can be generalized as wave detection, demodulation, detection, estimation and demultiplexing such as flexible results (logarithmic probability and logarithmic probability relation) and the definitive results (zeros and ones) obtained in this way they are the individual bits of data transmitted by the transmission device.

Different data can be transmitted through each stream s1 (t) and s2 (t) (sl (i), s2 (i)), or identical data can be transmitted in this way.

The two flux baseband signals s1 (i) and s2 (i) (where i indicates sequence (with respect to time or frequency (carrier))) undergo precoding and a regular phase change (the order of operations can be reversed freely) to generate two post-processing baseband signals z1 (i) and z2 (i). To post-process the baseband signal z1 (i), the phase I component is h (i) while the quadrature component is Q1 (i), and

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To post-process the baseband signal z2 (i), the phase component is Ii (i) while the quadrature component is Q2 (i). Baseband components can be changed, provided the following is true.

Be the phase component and the quadrature component of the changed baseband signal r1 (i) Ii (i) and Q2 (i), and be the phase component and the quadrature component of the changed baseband signal r2 (i) I2 (i) and Qi (i).

The modulated signal corresponding to the changed baseband signal r1 (i) is transmitted by the transmission antenna 1 and the modulated signal corresponding to the changed baseband signal r2 (i) is transmitted from the transmission antenna 2, simultaneously on a common frequency. As such, the modulated signal corresponding to the changed baseband signal r1 (i) and the modulated signal corresponding to the changed baseband signal r2 (i) are transmitted from different antennas, simultaneously on a common frequency. As an alternative,

For the changed baseband signal r1 (i), the phase component may be h (i) while the quadrature component may be I2 (i), and for the changed baseband signal r2 (i), the component in phase it can be Q1 (i) while the quadrature component can be Q2 (i).

For the changed baseband signal r1 (i), the phase component may be h (i) while the quadrature component may be h (i), and for the changed baseband signal r2 (i), the component in phase it can be Q1 (i) while the quadrature component can be Q2 (i).

For the changed baseband signal r1 (i), the phase component may be h (i) while the quadrature component may be h (i), and for the changed baseband signal r2 (i), the component in phase it can be Q2 (i) while the quadrature component can be Q1 (i).

For the changed baseband signal r1 (i), the phase component may be h (i) while the quadrature component may be h (i), and for the changed baseband signal r2 (i), the component in phase it can be Q2 (i) while the quadrature component can be Q1 (i).

For the changed baseband signal r1 (i), the phase component may be h (i) while the quadrature component may be Q2 (i), and for the changed baseband signal r2 (i), the component in phase it can be Q1 (i) while the quadrature component can be I2 (i).

For the changed baseband signal r1 (i), the phase component may be Q2 (i) while the quadrature component may be h (i), and for the changed baseband signal r2 (i), the component in phase it can be 12 (i) while the quadrature component can be Q1 (i).

For the changed baseband signal r1 (i), the phase component may be Q2 (i) while the quadrature component may be h (i), and for the changed baseband signal r2 (i), the component in phase it can be Q1 (i) while the quadrature component can be h (i).

For the changed baseband signal r2 (i), the phase component may be h (i) while the quadrature component may be h (i), and for the changed baseband signal r1 (i), the component in phase it can be Q1 (i) while the quadrature component can be Q2 (i).

For the changed baseband signal r2 (i), the phase component may be h (i) while the quadrature component may be h (i), and for the changed baseband signal r1 (i), the component in phase it can be Q1 (i) while the quadrature component can be Q2 (i).

For the changed baseband signal r2 (i), the phase component may be h (i) while the quadrature component may be h (i), and for the changed baseband signal r1 (i), the component in phase it can be Q2 (i) while the quadrature component can be Q1 (i).

For the changed baseband signal r2 (i), the phase component may be h (i) while the quadrature component may be h (i), and for the changed baseband signal r1 (i), the component in phase it can be Q2 (i) while the quadrature component can be Q1 (i).

For the changed baseband signal r2 (i), the phase component may be h (i) while the quadrature component may be Q2 (i), and for the changed baseband signal r1 (i), the component in phase it can be 12 (i) while the quadrature component can be Q1 (i).

For the changed baseband signal r2 (i), the phase component may be h (i) while the quadrature component may be Q2 (i), and for the changed baseband signal r1 (i), the component in phase it can be Q1 (i) while the quadrature component can be h (i).

For the changed baseband signal r2 (i), the phase component may be Q2 (i) while the quadrature component may be h (i), and for the changed baseband signal r1 (i), the component in phase it can be 12 (i) while the quadrature component can be Q1 (i)

For the changed baseband signal r2 (i), the phase component may be Q2 (i) while the quadrature component may be h (i), and for the changed baseband signal r1 (i), the component in phase it can be Q1 (i) while the quadrature component can be h (i).

Alternatively, although the above description analyzes performing two types of signal processing in both flow signals to change the phase component and the quadrature component of the two signals, the invention is not limited in this way. The two types of signal processing can be performed in more than two flows, to change the component in phase and the quadrature component thereof.

Alternatively, although the above examples describe the change made in baseband signals that have a common time indication (frequency of (sub-) common carrier), the baseband signals that are changed do not necessarily need to have an indication of common time (frequency of (sub-) carrier) common). For example, any of the following is possible.

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For the changed baseband signal r1 (i), the phase component may be Ii (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r2 (i ), the phase component can be h (i + w) while the quadrature component can be Qi (i + v).

For the changed baseband signal r1 (i), the phase component may be Ii (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r2 (i ), the phase component can be Q1 (i + v) while the quadrature component can be Q2 (i + w).

For the changed baseband signal r1 (i), the phase component may be I2 (i + v) while the quadrature component may be Q1 (i + w), and for the changed baseband signal r2 (i ), the phase component can be Q1 (i + v) while the quadrature component can be Q2 (i + w).

For the changed baseband signal r1 (i), the phase component may be h (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r2 (i ), the phase component can be Q2 (i + w) while the quadrature component can be Q1 (i + v).

For the changed baseband signal r1 (i), the phase component may be I2 (i + v) while the quadrature component may be Q1 (i + w), and for the changed baseband signal r2 (i ), the phase component can be Q2 (i + w) while the quadrature component can be Q1 (i + v).

For the changed baseband signal r1 (i), the phase component may be h (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r2 (i ), the phase component can be Q1 (i + v) while the quadrature component can be h (i + w).

For the changed baseband signal r1 (i), the phase component may be Q2 (i + w) while the quadrature component may be h (i + v), and for the changed baseband signal r2 (i ), the phase component can be h (i + w) while the quadrature component can be Q1 (i + v).

For the changed baseband signal r1 (i), the phase component may be Q2 (i + w) while the quadrature component may be h (i + v), and for the changed baseband signal r2 (i ), the phase component can be Q1 (i + v) while the quadrature component can be h (i + w).

For the changed baseband signal r2 (i), the phase component may be h (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r1 (i ), the phase component can be Q1 (i + v) while the quadrature component can be Q2 (i + w).

For the changed baseband signal r2 (i), the phase component may be h (i + v) while the quadrature component may be Q1 (i + w), and for the changed baseband signal r1 (i ), the phase component can be Q1 (i + v) while the quadrature component can be Q2 (i + w).

For the changed baseband signal r2 (i), the phase component may be h (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r1 (i ), the phase component can be Q2 (i + w) while the quadrature component can be Q1 (i + v).

For the changed baseband signal r2 (i), the phase component may be I2 (i + v) while the quadrature component may be Q1 (i + w), and for the changed baseband signal r1 (i ), the phase component can be Q2 (i + w) while the quadrature component can be Q1 (i + v).

For the changed baseband signal r2 (i), the phase component may be h (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r1 (i ), the phase component can be h (i + w) while the quadrature component can be Q1 (i + v).

For the changed baseband signal r2 (i), the phase component may be h (i + v) while the quadrature component may be Q2 (i + w), and for the changed baseband signal r1 (i ), the phase component can be Q1 (i + v) while the quadrature component can be h (i + w).

For the changed baseband signal r2 (i), the phase component may be Q2 (i + w) while the quadrature component may be h (i + v), and for the changed baseband signal r1 (i ), the phase component can be h (i + w) while the quadrature component can be Q1 (i + v).

For the changed baseband signal r2 (i), the phase component may be Q2 (i + w) while the quadrature component may be h (i + v), and for the changed baseband signal r1 (i ), the phase component can be Q1 (i + v) while the quadrature component can be h (i + w).

Figure 55 illustrates a baseband signal changer 5502 that explains the above. As shown, of the two processed baseband signals z1 (i) 5501_1 and z2 (i) 5501_2, the processed baseband signal z1 (i) 5501_1 has the phase component h (i) and the quadrature component Q1 (i), while the processed baseband signal z2 (i) 5501_2 has the phase component h (i) and the quadrature component Q2 (i). Then, after changing, the changed baseband signal r1 (i) 5503_1 has the phase component Ir1 (i) and the quadrature component Qr1 (i), while the changed baseband signal r2 (i) 5503_2 has the phase component Ir2 (i) and the quadrature component Qr2 (i). The phase component Ir1 (i) and the quadrature component Qr1 (i) of the changed baseband signal r1 (i) 5503_1 and the phase component Ir2 (i) and the quadrature component Qr2 (i) of the signal of changed baseband r2 (i) 5503_2 can be expressed as any of the above. Although this example describes the change made in the baseband signals that have an indication of common time (frequency of (sub-) common carrier) and that have undergone two types of signal processing, the same can be applied to the signals of baseband that have undergone two types of signal processing but have different time indications (different (sub-) carrier frequencies).

Each of the transmitting antennas of the transmitting device and each of the receiving antennas of the receiving device shown in the figures can be formed by a plurality of antennas.

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The present description uses symbol V, which is the universal quantifier, and symbol 3, which is the existential quantifier.

Additionally, the present description uses radian as the unit of phase in the complex plane, for example, for its argument.

When referring to the complex plane, the coordinates of complex numbers are expressible by means of polar coordinates. For a complex number z = a + jb (where a and b are real numbers and j is the imaginary unit), the corresponding point (a, b) in the complex plane is expressed with the polar coordinates [r, 0], converted as follows:

a = r x cqsO

b = r xsenfi

[Calculation 49]

image32

(formula 49)

where r is the absolute value of z (r = | z |), and 0 is its argument. As such, z = a + jb is expressible as rej0.

In the present invention, the baseband signals si, s2, z1, and z2 are described as being complex signals. A complex signal composed of the phase I signal and the quadrature signal Q is also expressible as the complex signal I + jQ. At this point, any of I and Q can be zero.

Figure 46 illustrates a sample diffusion system using the phase change method described in the present description. As shown, a 4601 video encoder takes video as input, performs video encoding and outputs encoded video data 4602. An audio encoder 4603 takes audio as input, performs audio encoding and outputs encoded audio data 4604. An encoder of data 4605 takes data as input, performs data coding (for example, data compression), and issues data encoded 4606. Taken as a whole, these components form a source information encoder 4600.

A transmitter 4607 takes the encoded video data 4602, the encoded audio data 4604 and the encoded data 4606 as input, performs error correction coding, modulation, precoding and phase change (for example, signal processing by the device of transmission from Figure 3) in a subset or in all of these, and issues the transmission signals 4608_1 to 4608_N. Transmission signals 4608_1 to 4608_N are then transmitted through antennas 4609_1 to 4609_N as radio waves.

A receiver 4612 takes the signals received 4611_1 to 4611_M received through the antennas 4610_1 to 4610_M as input, performs processing such as frequency conversion, phase change, decoding of the precoding, calculation of logarithmic probability ratio and error correction decoding (for example, the processing by the reception device from Figure 7), and emits the received data 4613, 4615 and 4617. A source information decoder 4619 takes the received data 4613, 4615 and 4617 as input. A video decoder 4614 takes the received data 4613 as input, performs video decoding and emits a video signal. The video is presented below on a television screen. An audio decoder 4616 takes the received data 4615 as input. Audio decoder 4616 performs audio decoding and emits an audio signal, the audio is then played through speakers. A data decoder 4618 takes the received data 4617 as input, performs data decoding and issues information.

In the previously described embodiments pertaining to the present invention, the number of encoders in the transmission device that use a multi-carrier transmission method such as OFDM can be any number, as described above. Therefore, as in Figure 4, for example, the transmission device may have only one encoder and apply a method for distributing the output to the multi-carrier transmission method such as OFDM. In such circumstances, the wireless units 310A and 310B from Figure 4 should replace the processors related to OFDM 1301A and 1301B from Figure 12. The description of the processors related to OFDM is as provided for realization 1.

Although embodiment 1 provides Calculation 36 (formula 36) as an example of a precoding matrix,

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Another precoding matrix can also be used, when the following method is applied. [Calculation 50]

'wll k> 12 ^

vw2l w22 j

image33

image34

ax

jo

ax

JO

e)

(formula 50)

In the precoding matrices of Calculation 36 (formula 36) and Calculation 50 (formula 50), the value of a is set as provided by Calculation 37 (formula 37) and Calculation 38 (formula 38). However, no limitation is intended in this way. A simple precoding matrix is obtainable by setting a = 1, which is also a valid value.

In embodiment A1, the phase changers from Figures 3, 4, 6, 12, 25, 29, 51 and 53 are indicated as having a phase change value of PHASE [i] (where i = 0 , 1, 2, ..., N-2, N-1) to achieve a period (cycle) of N (value reached given that Figures 3, 4, 6, 12, 25, 29, 51 and 53 perform a phase change in only one baseband signal). The present description analyzes a phase change in a precoded baseband signal (that is, in Figures 3, 4, 6, 12, 25, 29, 51 and 53) specifically in the precoded baseband signal z2 ' . At this point, PHASE [k] is calculated as follows.

[Calculation 51]

radians (formula 51)

where k = 0, 1, 2, ..., N-2, N-1. When N = 5, 7, 9, 11 or 15, the receiving device can obtain good data reception quality.

Although the present description analyzes the details of the phase change methods that involve two modulated signals transmitted by a plurality of antennas, no limitation is intended in this regard. The precoding and a phase change can be performed in three or more baseband signals in which mapping has been performed according to a modulation method, followed by predetermined processing in the post-phase change of the baseband signals and transmission using a plurality of antennas, to perform the same results.

image35

The programs for executing the above transmission method can be stored in advance, for example, in the ROM (Read Only Memory) and read for operation by means of a CPU.

Additionally, the programs for executing the above transmission method can be stored in a computer readable recording medium, the programs stored in the recording medium can be loaded into the RAM (Random Access Memory) of the computer, and the computer can be operated in I agree with the programs.

The components of the above described embodiments can normally be assembled as an LSI (Large Scale Integration), a type of integrated circuit. The individual components can be manufactured respectively in discrete chips, or a subset or all of the components can be manufactured in a single chip. Although an LSI has been previously mentioned, the expressions IC (Integrated Circuit), LSI system, super LSI or ultra LSI can also be applied, depending on the degree of integration. Additionally, the integrated circuit assembly method is not limited to LSI. A specialized circuit or a general purpose processor can be used. After LSI assembly, an FPGA (Programmable Door Matrix Field) can be used or a reconfigurable processor can be used.

Additionally, if progress in the field of semiconductors or emerging technologies led to the replacement of LSI by other integrated circuit methods, then such technology could of course be used to integrate the functional blocks. Biotechnology applications are also feasible.

[Realization C1]

Embodiment 1 explains that the precoding matrix in use can be changed when the transmission parameters change. The present embodiment describes a detailed example of such a case, where, as described above (in the supplement), the transmission parameters change so that the flows s1 (t) and s2 (t) change between transmitting different data and transmit identical data, and the precoding matrix and the phase change method that are used are changed accordingly.

The example of the present embodiment describes a situation where two modulated signals transmitted from two different transmission antennas alternate between having modulated signals that include identical data and having the

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modulated signals that each includes different data.

Figure 56 illustrates a sample configuration of a transmission device that switches between transmission methods, as described above. In Figure 56, components that operate in the same manner described for Figure 54 use identical reference numbers. As shown, Figure 56 differs from Figure 54 in that a distributor 404 takes frame configuration signal 313 as input. The operations of distributor 404 are described using Figure 57.

Figure 57 illustrates the operations of distributor 404 when identical data is transmitted and when different data is transmitted. As shown, given the encoded data x1, x2, x3, x4, x5, x6 and as! successively, when identical data is transmitted, distributed data 405 is provided as x1, x2, x3, x4, x5, x6 and so! successively, while the distributed data 405B is similarly provided as x1, x2, x3, x4, x5, x6 and so! successively.

On the other hand, when different data is transmitted, the distributed data 405A is provided as x1, x3, x5, x7, x9 and so! successively, while the distributed data 405B is provided as x2, x4, x6, x8, x10 and as! successively.

The distributor 404 determines, according to the frame configuration signal 313 taken as input, if the transmission mode is identical data transmission or different data transmission.

An alternative method to the above is shown in Figure 58. As shown, when identical data is transmitted, distributor 404 issues distributed data 405A as x1, x2, x3, x4, x5, x6 and so! successively, while the 405B data is not issued as distributed. Therefore, when frame configuration signal 313 indicates transmission of identical data, distributor 404 operates as described above, while interleaver 304B and mapper 306B from Figure 56 do not operate. Therefore, only the baseband signal 307A issued by the mapper 306A from Figure 56 is valid, and is taken as input by both weighting units 308A and 308B.

A characteristic feature of the present embodiment is that, when the transmission mode changes from transmission of identical data to transmission of different data, the precoding matrix can also be changed. As indicated by Calculation 36 (formula 36) and Calculation 39 (formula 39) in embodiment 1, given a matrix composed of w11, w12, w21 and w22, the precoding matrix used to transmit identical data can be as follows .

[Calculation 52]

rwll

^ w21

image36

(formula 52)

where a is a real number (a may also be a complex number, but since the baseband signal introduced as a result of the precoding undergoes a phase change, a real number is preferred for circuit size considerations and reduction of complexity). Also, when a is equal to one, the weighting units 308A and 308B do not perform weighting and emit the input signal as is.

Therefore, when identical data is transmitted, the weighted baseband signals 309A and 316B are identical signals emitted by the weighting units 308A and 308B.

When the frame configuration signal 313 indicates identical transmission mode, a phase changer 5201 makes a phase change in the weighted baseband signal 309A and outputs the post-phase change baseband signal 5202. Similarly , when the frame configuration signal indicates identical transmission mode, the phase changer 317B makes a phase change in the weighted baseband signal 316B and outputs the post-phase change baseband signal 309B. The phase change performed by the 5201 phase changer is eA (t) (alternatively, ejA (f) or ejA (tf)) (where t is time and f is frequency) (Therefore, ejA (t) (as alternative, eA (f) or ejA (tf)) is the value by which the input baseband signal is multiplied), and the phase change made by the phase changer 317B is of ejB (t) (as an alternative , ejB (f) or ejB (tf)) (where t is time and f is frequency) (therefore, ejB (t) (alternatively, ejB (f) or ejB (tf)) is the value by which the signal Baseband input is multiplied). As such, the following condition is satisfied.

[Calculation 53]

Some time t please

eMO) ^)

(Or, some frequency (carrier) f satisfies ejA (f) t ejB (f))

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(Or, some frequency (carrier) f and time t satisfy eiA (tf t eB (t, f))

As such, the transmission signal can reduce the multi-route influence and thus improve the quality of data reception for the receiving device. (However, the phase change can also be performed by only one of the weighted baseband signals 309A and 316B).

In Figure 56, when OFDM is used, processing such as IFFT and frequency conversion is performed on the post-phase change baseband signal 5202, and the result is transmitted by a transmission antenna. (See Figure 13) (Accordingly, the post-phase change baseband signal 5202 can be considered the same as the signal 1301A from Figure 13). Similarly, when OFDM is used, processing such as IFFT and frequency conversion is performed on the post-phase change baseband signal 309B, and the result is transmitted by a transmitting antenna. (See Figure 13) (Accordingly, the post-phase change baseband signal 309B can be considered the same as signal 1301B from Figure 13).

When the selected transmission mode indicates transmission of different data, then any of Calculation 36 (formula 36), Calculation 39 (formula 39) and Calculation 50 (formula 50) given in embodiment 1 can be applied. Significantly, the changers of Phase 5201 and 317B from Figure 56 use a different phase change method than when identical data is transmitted. Specifically, as described in embodiment 1, for example, the phase changer 5201 performs the phase change while the phase changer 317B does not, or the phase changer 317B performs the phase change while the phase changer 5201 does not. Only one of the two phase changers performs the phase change. As such, the receiving device obtains good quality of data reception in the environment of LOS as! as the environment of NLOS.

When the selected transmission mode indicates different data transmission, the precoding matrix can be provided as in the Calculation. 52 (formula 52), or as any of Calculation 36 (formula 36), Calculation 50 (formula 50) and Calculation 39 (formula 39), or it may be a precoding matrix unlike that provided in Calculation 52 ( formula 52). Therefore, the receiving device is especially likely to experience improvements in the quality of data reception in the LOS environment.

Additionally, although the present embodiment analyzes examples using OFDM as the method of transmission, the invention is not limited in this way. All multi-carrier methods other than OFDM and single carrier methods can be used to achieve similar embodiments. At this point, spread spectrum communications can also be used. When single carrier methods are used, the phase change takes place with respect to the time domain.

As explained in embodiment 3, when the transmission method involves transmission of different data, the phase change is carried out in the data symbols only. However, as described in the present embodiment, when the transmission method involves transmission of identical data, then the phase change does not need to be limited to the data symbols but can also be performed in pilot symbols, control symbols and others. Symbols of this type inserted in the transmission frame of the transmission signal. (The phase change does not always need to be done on symbols such as pilot symbols and control symbols, although this is preferred to achieve diversity gain).

[Realization C2]

The present embodiment describes a configuration method for a base station corresponding to embodiment C1.

Figure 59 illustrates the relationship of base stations (diffusers) to terminals. A terminal P (5907) receives the transmission signal 5903A transmitted by the antenna 5904A and the transmission signal 5905A transmitted by the antenna 5906A of the diffuser A (5902A), then performs predetermined processing therein to received data obtained.

A terminal Q (5908) receives the transmission signal 5903A transmitted by the antenna 5904A of the base station A (5902A) and the transmission signal 593B transmitted by the antenna 5904B of the base station B (5902B), then performs predetermined processing in the same to received data obtained.

Figures 60 and 61 illustrate the frequency assignment of the base station A (5902A) for the transmission signals 5903A and 5905A transmitted by the antennas 5904A and 5906A, and the frequency assignment of the base station B (5902B) for the signals of transmission 5903B and 5905B transmitted by antennas 5904B and 5906B. In Figures 60 and 61, the frequency is on the horizontal axis and the transmission power is on the vertical axis.

As shown, the transmission signals 5903A and 5905A transmitted by the base station A (5902A) and the transmission signals 5903B and 5905B transmitted by the base station B (5902B) use at least the frequency band X and the frequency band Y. The frequency band X is used to transmit data from a

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first channel, and the frequency band Y is used to transmit data from a second channel.

Accordingly, the terminal P (5907) receives the transmission signal 5903A transmitted by the antenna 5904A and the transmission signal 5905A transmitted by the antenna 5906A of the base station A (5902A), extracts the frequency band X from it, performs default processing, and therefore obtains the data of the first channel. Terminal Q (5908) receives the transmission signal 5903A transmitted by the antenna 5904A of the base station A (5902A) and the transmission signal 5903B transmitted by the antenna 5904B of the base station B (5902B), extracts the frequency band And from it, it performs default processing, and therefore obtains data from the second channel.

The following describes the configuration and operations of the base station A (5902A) and the base station B (5902B).

As described in embodiment C1, both the base station A (5902A) and the base station B (5902B) incorporate a transmission device configured as illustrated by Figures 56 and 13. When transmitted as illustrated by Figure 60 , the base station A (5902A) generates two different modulated signals (on which precoding and a phase change is performed) with respect to the frequency band X as described in embodiment C1. The two modulated signals are transmitted respectively through the antennas 5904A and 5906A. With respect to the frequency band Y, the base station A (5902A) operates the interleaver 304A, the mapper 306A, the weighting unit 308A and the phase changer from Figure 56 to generate the modulated signal 5202. Next , a transmission signal corresponding to modulated signal 5202 is transmitted by antenna 1310A from Figure 13, that is, by antenna 5904A from Figure 59. Similarly, base station B (5902B) The interleaver 304A, the mapper 306A, the weighting unit 308A and the phase changer 5201 are operated from Figure 56 to generate the modulated signal 5202. Next, the transmission signal corresponding to the modulated signal 5202 is transmitted by antenna 1310A from Figure 13, that is, by antenna 5904B from Figure 59.

The creation of encoded data in the frequency band Y may involve, as shown in Figure 56, generating encoded data at individual base stations, or it may involve having one of the base stations that generating such encoded data for transmission to other base stations . As an alternative method, one of the base stations can generate the modulated signals and be configured to pass the modulated signals as! generated to other base stations.

Also, in Figure 59, signal 5901 includes information pertaining to the transmission mode (transmission of identical data or transmission of different data). The base stations obtain this signal by changing in this way between generation methods for the modulated signals in each frequency band. At this point, signal 5901 is indicated in Figure 59 as being input from another device or from a network. However, configurations are also possible where, for example, base station A (5902) is a master station that passes a signal that corresponds to signal 5901 to base station B (5902B).

As explained above, when the base station transmits different data, the precoding matrix and the phase change method are established in accordance with the transmission method to generate the modulated signals.

On the other hand, to transmit identical data, two base stations generate and transmit respectively the modulated signals. In such circumstances, each of the base stations that generates modulated signals for transmission from a common antenna can be considered to be two base stations combined using the precoding matrix provided by Calculation 52 (formula 52). The phase change method is as explained in embodiment C1, for example, and satisfies the conditions of Calculation 53 (formula 53).

In addition, the method of transmission of the frequency band X and the frequency band Y may vary over time. Therefore, as illustrated in Figure 61, as time passes, the frequency assignment changes from that indicated in Figure 60 to that indicated in Figure 61.

According to the present embodiment, not only the reception device can obtain improved data reception quality for transmission of identical data as well! As different data transmission, but the transmission devices can also share a phase changer.

Additionally, although the present embodiment analyzes examples using OFDM as the transmission method, the invention is not limited in this way. All multi-carrier methods other than OFDM and single carrier methods can be used to achieve similar embodiments. At this point, spread spectrum communications can also be used. When single carrier methods are used, the phase change takes place with respect to the time domain.

As explained in embodiment 3, when the transmission method involves the transmission of different data, the phase change is carried out in the data symbols only. However, as described in the present embodiment, when the transmission method involves transmission of identical data, then the change of

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phase does not need to be limited to data slots but can also be carried out in pilot slots, control slots and other symbols of this type inserted in the transmission frame of the transmission signal. (The phase change does not always need to be done on symbols such as pilot symbols and control symbols, although this is preferred to achieve diversity gain).

[Realization C3]

The present embodiment describes a method of configuration for a repeater that corresponds to embodiment C1. The repeater can also be called a repetition station.

Figure 62 illustrates the relationship of base stations (diffusers) to repeaters and terminals. As shown in Figure 63, the base station 6201 at least transmits the modulated signals in the frequency band X and in the frequency band Y. The base station 6201 transmits the respective modulated signals in the antenna 6202A and in the antenna 6202B . The transmission method used at this point is described below, with reference to Figure 63.

The repeater A (6203A) performs processing such as demodulation in the received signal 6205A received by the receiving antenna 6204A and in the received signal 6207A received by the receiving antenna 6206A, thereby obtaining the received data. Then, to transmit the data received to a terminal, the repeater A (6203A) performs transmission processing to generate the modulated signals 6209A and 6211A for transmission on the respective antennas 6210A and 6212A.

Similarly, the repeater B (6203B) performs processing such as demodulation in the received signal 6205B received by the receiving antenna 6204B and in the received signal 6207B received by the receiving antenna 6206B, thereby obtaining the received data. Then, to transmit the data received to a terminal, the repeater B (6203B) performs transmission processing to generate the modulated signals 6209B and 6211B for transmission on the respective antennas 6210B and 6212B. At this point, the repeater B (6203B) is a master repeater that emits a control signal 6208. The repeater A (6203A) takes the control signal as input. A master repeater is not strictly necessary. The base station 6201 can also transmit individual control signals to repeater A (6203A) and to repeater B (6203B).

Terminal P (5907) receives the modulated signals transmitted by repeater A (6203A), thus obtaining data. Terminal Q (5908) receives signals transmitted by repeater A (6203A) and by repeater B (6203B), thus obtaining data. Terminal R (6213) receives the modulated signals transmitted by repeater B (6203B), thus obtaining data.

Figure 63 illustrates the frequency assignment for a modulated signal transmitted by the antenna 6202A between the transmission signals transmitted by the base station, and the frequency assignment of the modulated signals transmitted by the antenna 6202B. In Figure 63, the frequency is on the horizontal axis and the transmission power is on the vertical axis.

As shown, the modulated signals transmitted by the antenna 6202A and by the antenna 6202B use at least the frequency band X and the frequency band Y. The frequency band X is used to transmit data from a first channel, and the band of frequency Y is used to transmit data from a second channel.

As described in embodiment C1, the data of the first channel is transmitted using the frequency band X in different data transmission mode. Therefore, as shown in Figure 63, the modulated signals transmitted by the antenna 6202A and by the antenna 6202B include components of the frequency band X. These components of the frequency band X are received by the repeater A and by the repeater B. Therefore, as described in embodiment 1 and in embodiment C1, the signals modulated in the frequency band X are signals in which mapping has been performed, and to which precoding (weighting) and phase change.

As shown in Figure 62, the data of the second channel is transmitted by the antenna 6202A of Figure 2 and transmits data in components of the frequency band Y. These components of the frequency band Y are received by the repeater A and by repeater B.

Figure 64 illustrates the frequency assignment for the transmission signals transmitted by the repeater A and the repeater B, specifically for the modulated signal 6209A transmitted by the antenna 6210A and the modulated signal 6211A transmitted by the antenna 6212A of the repeater 6210A, and for the modulated signal 6209B transmitted by the antenna 6210B and the modulated signal 6211B transmitted by the antenna 6212B of the repeater B. In Figure 64, the frequency is on the horizontal axis and the transmission power is on the vertical axis.

As shown, the modulated signal 6209A transmitted by the antenna 6210A and the modulated signal 6211A transmitted by the antenna 6212A use at least the frequency band X and the frequency band Y. Also, the modulated signal 6209B transmitted by the antenna 6210B and the 6211B modulated signal transmitted by the

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Antenna 6212B similarly uses at least the frequency band X and the frequency band Y. The frequency band X is used to transmit data from a first channel, and the frequency band Y is used to transmit data from a second channel .

As described in embodiment C1, the data of the first channel is transmitted using the frequency band X in different data transmission mode. Therefore, as shown in Figure 64, the modulated signal 6209A transmitted by the antenna 6210A and the modulated signal 6211A transmitted by the antenna 6212B include components of the frequency band X. These components of the frequency band X are received by terminal P. Similarly, as shown in Figure 64, the modulated signal 6209B transmitted by the antenna 6210B and the modulated signal 6211B transmitted by the antenna 6212B include components of the frequency band X. These components of the frequency band X frequencies are received through terminal R. Therefore, as described in embodiment 1 and in embodiment C1, the signals modulated in the frequency band X are signals in which mapping has been performed, and to which it has been applied precoding (weighting) and phase change.

As shown in Figure 64, data from the second channel is carried by modulated signals transmitted by antenna 6210A of repeater A (6203A) and by antenna 6210B of repeater B (6203) from Figure 62 and transmit data in components of the frequency band Y. At this point, the components of the frequency band Y in the modulated signal 6209A transmitted by the antenna 6210A of the repeater A (6203A) and those in the modulated signal 6209B transmitted by the antenna 6210B of the Repeater B (6203B) is used in a transmission mode that involves transmission of identical data, as explained in embodiment C1. These components of the frequency band Y are received by terminal Q.

The following describes the configuration of repeater A (6203A) and repeater B (6203B) from Figure 62, with reference to Figure 65.

Figure 65 illustrates a sample configuration of a receiver and transmitter in a receiver. The components that operate identically to those in Figure 56 use the same reference numbers between them. The receiver 6203X takes the received signal 6502A received by the receiving antenna 6501A and the received signal 6502B received by the receiving antenna 6501B as input, performs signal processing (demultiplexing or signal composition, error correction decoding and so on) ) in the components of the frequency band X thereof to obtain 6204X data transmitted through the base station using the frequency band X, issues the data to the distributor 404 and obtains transmission method information included in control information (and information of transmission method when transmitted by means of a repeater), and emits frame configuration signal 313.

The 6203X and forward receiver constitutes a processor to generate a modulated signal to transmit the frequency band X. In addition, the receiver described at this point is not only the receiver for the frequency band X as shown in Figure 65, but It also incorporates receivers for other frequency bands. Each receiver forms a processor to generate the modulated signals to transmit a respective frequency band.

The global operations of distributor 404 are identical to those of the distributor in the base station described in embodiment C2.

When transmitted as indicated in Figure 64, repeater A (6203A) and repeater B (6203B) generate two different modulated signals (in which precoding and phase change is performed) in the frequency band X as described in realization C1. The two modulated signals are transmitted respectively by the antennas 6210A and 6212A of the repeater A (6203) from Figure 62 and by the antennas 6210B and 6212B of the repeater B (6203B) from Figure 62.

As for the frequency band Y, the repeater A (6203A) operates a processor 6500 that belongs to the frequency band Y and that corresponds to the signal processor 6500 that belongs to the frequency band X shown in Figure 65 (the processor of signal 6500 is the signal processor that belongs to the frequency band X, but since an identical signal processor is incorporated for the frequency band Y, this description uses the same reference numbers), interleaver 304A, the mapper 306A, the weighting unit 308A and the phase changer 5201 to generate the modulated signal 5202. A transmission signal corresponding to the modulated signal 5202 is then transmitted by the antenna 1301A from Figure 13, that is, by antenna 6210A from Figure 62. Similarly, repeater B (6203 B) operates interleaver 304A, mapper 306A, weighting unit 308A and phase changer 5201 from Figure 62 which allows it has the frequency band Y to generate the modulated signal 5202. Next, a transmission signal corresponding to the modulated signal 5202 is transmitted by the antenna 1310A from Figure 13, that is, by the antenna 6210B from of Figure 62.

As shown in Figure 66 (Figure 66 illustrates the frame configuration of the modulated signal transmitted by the base station, with time on the horizontal axis and frequency on the vertical axis), the base station

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transmits transmission method information 6601, phase change information applied to the repeater 6602 and data slots 6603. The repeater obtains and applies the transmission method information 6601, the phase change information applied to the repeater 6602 and the symbols of 6603 data to the transmission signal, thereby determining the phase change method. When the phase change information applied to the repeater 6602 from Figure 66 is not included in the signal transmitted by the base station, then as shown in Figure 62, the repeater B (6203B) is the master and indicates the phase change method to repeater A (6203A).

As explained above, when the repeater transmits different data, the precoding matrix and the phase change method are established according to the transmission method to generate the modulated signals.

On the other hand, to transmit identical data, two repeaters generate and transmit respectively the modulated signals. In such circumstances, the repeaters that each generate the modulated signals for transmission from a common antenna can be considered to be two combined repeaters using the precoding matrix provided by Calculation 52 (formula 52). The phase change method is as explained in realization C1, for example, and satisfies the conditions of Calculation 53 (formula 53).

Also, as explained in embodiment C1 for the frequency band X, the base station and the repeater can each have two antennas that transmit respective modulated signals and two antennas that receive identical data. The operations of a base or repeater station of this type are as described for embodiment C1.

According to the present embodiment, not only the reception device can obtain improved data reception quality for transmission of identical data as well! as a different data transmission, but the transmission devices can also share a phase changer

Additionally, although the present embodiment analyzes examples using OFDM as the transmission method, the invention is not limited in this way. All multi-carrier methods other than OFDM and single carrier methods can be used to achieve similar embodiments. At this point, spread spectrum communications can also be used. When single carrier methods are used, the phase change takes place with respect to the time domain.

As explained in embodiment 3, when the transmission method involves transmission of different data, the phase change is carried out in the data symbols only. However, as described in the present embodiment, when the transmission method involves transmission of identical data, then the phase change does not need to be limited to the data symbols but can also be performed in pilot symbols, control symbols and others. Symbols of this type inserted in the transmission frame of the transmission signal. (The phase change does not always need to be done on symbols such as pilot symbols and control symbols, although this is preferred to achieve diversity gain).

[Realization C4]

The present embodiment refers to a phase change method different from the phase change methods described in embodiment 1 and in the supplement.

In embodiment 1, Calculation 36 (formula 36) is provided as an example of a precoding matrix, and in the supplement, Calculation 50 (formula 50) is similarly provided as another example of this type. In embodiment A1, the phase changers from Figures 3, 4, 6, 12, 25, 29, 51 and 53 are indicated as having a phase change value of PHASE [i] (where i = 0 , 1, 2, ..., N-2, N- 1) to achieve a period (cycle) of N (value reached given that Figures 3, 4, 6, 12, 25, 29, 51 and 53 perform a phase change in only one baseband signal). The present description analyzes a phase change in a precoded baseband signal (that is, in Figures 3, 4, 6, 12, 25, 29, 51 and 53) specifically in the precoded baseband signal z2 ' . At this point, PHASE [k] is calculated as follows.

[Calculation 54]

PHASE [k \ = ~

N radians (formula 54)

where k = 0, 1, 2, ..., N-2, N-1.

Therefore, the receiving device can achieve improvements in the quality of data reception in the LOS environment, and especially in a radio wave propagation environment. In the LOS environment, when the phase change has not been made, a regular phase relationship takes place. However, when the phase change takes place, the phase relationship is modified, avoiding poor conditions in a propagation environment.

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similar to bursts. As an alternative to Calculation 54 (formula 54), PHASE [k] can be calculated as follows. [Calculation 55]

kn

N radians (formula 55)

PHASE [£]

where k = 0, 1, 2, ..., N-2, N-1.

As an additional alternative phase change method, PHASE [k] can be calculated as follows. [Calculation 56]

image37

(formula 56)

where k = 0, 1, 2, ..., N-2, N-1.

As an additional alternative phase change method, PHASE [k] can be calculated as follows. [Calculation 57]

image38

where k = 0, 1, 2, ..., N-2, N-1.

As such, by performing the phase change according to the present embodiment, the receiving device becomes more likely to obtain good reception quality.

The phase change of the present embodiment is applicable not only to single carrier methods but also to multi-carrier methods. Accordingly, the present embodiment can also be performed using, for example, spread spectrum communications, OFDM, SC-FDMA, SC-OFDM, particle OFDM as described in Non-patent Bibliography 7, and so on! successively. As described above, although the present embodiment explains the phase change as a phase change with respect to the time domain t, the phase can be changed as an alternative with respect to the frequency domain as described in embodiment 1. That is , considering the phase change with respect to the time domain t described in the present embodiment and substituting t for f (where f is the frequency of ((sub-) carrier)) leads to a phase change applicable to the frequency domain. Also, as explained above for embodiment 1, the phase change method of the present embodiment is also applicable to a phase change with respect to both the time domain and the frequency domain. Furthermore, when the phase change method described in the present embodiment satisfies the conditions indicated in embodiment A1, the receiving device is highly likely to obtain good data quality.

[Realization C5]

The present embodiment refers to a phase change method different from the phase change methods described in embodiment 1, in the supplement and in embodiment C4.

In embodiment 1, Calculation 36 (formula 36) is provided as an example of a precoding matrix, and in the supplement, Calculation 50 (formula 50) is similarly provided as another example of this type. In embodiment A1, the phase changers from Figures 3, 4, 6, 12, 25, 29, 51 and 53 are indicated as having a phase change value of PHASE [i] (where i = 0 , 1, 2, ..., N-2, N- 1) to achieve a period (cycle) of N (value reached given that Figures 3, 4, 6, 12, 25, 29, 51 and 53 perform a phase change in only one baseband signal). The present description analyzes a phase change in a precoded baseband signal (that is, in Figures 3, 4, 6, 12, 25, 29, 51 and 53) specifically in the precoded baseband signal z2 ' .

The characteristic feature of the phase change method that belongs to the present embodiment is the period (cycle) of N = 2n + 1. To achieve the period (cycle) of N = 2n + 1, n + 1 different values of phase change. Among these n + 1 different phase change values, n phase change values are used twice per period (cycle), and a phase change value is used only once per period (cycle), thus achieving the period (cycle) of N = 2n + 1. The following describes these phase change values in detail.

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The n + 1 different phase change values required to achieve a phase change method in which the phase change value is regularly changed in a period (cycle) of N = 2n +1 are expressed as PHASE [0] , PHASE [1], PHASE [i] ... PHASE [n-1], PHASE [n] (where i = 0, 1, 2 ... n-2, n-1, n). At this point, the n + 1 different phase change values of PHASE [0], PHASE [1], PHASE [i] ... PHASE [n-1], PHASE [n] are expressed as follows.

[Calculation 58]

image39

(formula 58)

where k = 0, 1, 2, ..., n-2, n-1, n. The n + 1 different phase change values PHASE [0], PHASE [1] ... PHASE [i] ... PHASE [n-1], PHASE [n] are provided by Calculation 58 (formula 58) . PHASE [0] is used once, while PHASE [1] to PHASE [n] is used every two times (ie, PHASE [1] is used twice, PHASE [2] is used twice and so on! successively, until PHASE [n-1] is used twice and PHASE [n] is used twice). As such, through this phase change method in which the phase change value is regularly changed in a period (cycle) of N = 2n +1, a phase change method is performed in which the value Phase change is changed regularly between less phase change values. Therefore, the receiving device can achieve better data reception quality. Since the phase change values are smaller, their effect on the transmission device and the reception device can be reduced. According to the above, the receiving device can achieve improvements in the quality of data reception in the LOS environment, and especially in a radio wave propagation environment. In the LOS environment, when the phase change has not been made, a regular phase relationship takes place. However, when the phase change is made, the phase relationship is modified, in turn avoiding poor conditions in a burst-like propagation environment. As an alternative to Calculation 58 (formula 58), PHASE [k] can be calculated as follows.

[Calculation 59]

image40

where k = 0, 1, 2, ..., n-2, n-1, n.

The n + 1 different phase change values PHASE [0], PHASE [1] ... PHASE [i] ... PHASE [n-1], PHASE [n] are provided by Calculation 59 (formula 59) . PHASE [0] is used once, while PHASE [1] to PHASE [n] is used every two times (ie, PHASE [1] is used twice, PHASE [2] is used twice, and so ! successively, until PHASE [n-1] is used twice and PHASE [n] is used twice). As such, through this phase change method in which the phase change value is regularly changed in a period (cycle) of N = 2n +1, a phase change method is performed in which the value Phase change is changed regularly between less phase change values. Therefore, the receiving device can achieve better data reception quality. Since the phase change values are smaller, their effect on the transmission device and the reception device can be reduced.

As an additional alternative, PHASE [k] can be calculated as follows.

[Calculation 60]

rr _ 2k? u _

PHASE [k] = ------------------ + Z

'"2n + 1 radians (formula 60)

where k = 0, 1, 2, ..., N-2, N-1.

The n + 1 different phase change values PHASE [0], PHASE [1] ... PHASE [i] ... PHASE [n-1], PHASE [n] are provided by Calculation 60 (formula 60) . PHASE [0] is used once, while PHASE [1] to PHASE [n] is used every two times (ie, PHASE [1] is used twice, PHASE [2] is used twice, and so ! successively, until PHASE [n-1] is used twice and PHASE [n] is used twice). As such, through this phase change method in which the phase change value is regularly changed in a period (cycle) of N = 2n +1, a phase change method is performed in which the value Phase change is changed regularly between less phase change values. Therefore, the receiving device can achieve better data reception quality. Since the phase change values are smaller, their effect on the transmission device and the reception device can be reduced.

As an additional alternative, PHASE [k] can be calculated as follows.

[Calculation 61]

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2 jtjT

PHASE [k] = - + Z

2n + 1 radians (formula 61)

where k = 0, 1, 2, ..., n-2, n-1, n.

The n + 1 different phase change values PHASE [0], PHASE [1] ... PHASE [i] ... PHASE [n-1], PHASE [n] are provided by Calculation 61 (formula 61) . PHASE [0] is used once, while PHASE [1] to PHASE [n] is used every two times (ie, PHASE [1] is used twice, PHASE [2] is used twice, and so ! successively, until PHASE [n-1] is used twice and PHASE [n] is used twice). As such, through this phase change method in which the phase change value is regularly changed in a period (cycle) of N = 2n +1, a phase change method is performed in which the value Phase change is changed regularly between less phase change values. Therefore, the receiving device can achieve better data reception quality. Since the phase change values are smaller, their effect on the transmission device and the reception device can be reduced.

As such, by performing the phase change according to the present embodiment, the receiving device becomes more likely to obtain good reception quality.

The phase change of the present embodiment is applicable not only to single carrier methods but also to transmission using multi-carrier methods. Accordingly, the present embodiment can also be performed using, for example, spread spectrum communications, OFDM, SC-FDMA, SC-OFDM, particle OFDM as described in Non-patent Bibliography 7, and so on! successively. As described above, although the present embodiment explains the phase change as a phase change with respect to the time domain t, the phase can be changed as an alternative with respect to the frequency domain as described in embodiment 1. That is , considering the phase change with respect to the time domain t described in the present embodiment and substituting t for f (where f is the frequency of ((sub-) carrier)) leads to a phase change applicable to the frequency domain. Also, as explained above for embodiment 1, the phase change method of the present embodiment is also applicable to a phase change with respect to both the time domain and the frequency domain.

[Realization C6]

The present embodiment describes a method for regularly changing the phase, specifically that of embodiment C5, when coding is performed using block codes as described in Non-patent Bibliography 12 to 15, such as LDPC Codes (not only QC- LDPC but also LDPC codes) QC, LDPC codes (blocks) and concatenated BCH codes, Turbo codes or Turbo Duo-binary codes using tail bits, and so! successively. The following example considers a case where two flows s1 and s2 are transmitted. When coding has been carried out using block codes and control information and they are not necessary, the number of bits that make up each block encoded coincides with the number of bits that make up each block code (even control information can be included and thus ! successively described below). When coding has been performed using block codes or the like and control information or the like is required (eg, CRC transmission parameters), then the number of bits that make up each encoded block is the sum of the number of bits that make up the block codes and the number of bits that make up the information.

Figure 34 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used. Figure 34 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams s1 and s2 are transmitted as indicated by the transmission device from Figure 4, and The transmission device has only one encoder. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 34, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

Next, since the transmission device from Figure 4 transmits two streams simultaneously, 1500 of the 3,000 above-mentioned symbols required when the modulation method is QPSK are assigned to s1 and the other 1500 symbols are assigned to s2. As such, 1500 intervals are required to transmit the 1500 symbols (hereinafter intervals) for each of s1 and s2.

By the same reasoning, when the modulation method is 16-QAM, 750 intervals are necessary to transmit all the bits that make up each encoded block, and when the modulation method is 64-QAM, 500 intervals are necessary to transmit all the bits that make up each coded block.

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The following describes the relationship between the previously defined intervals and the phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change, which has a period (cycle) of five. That is, the phase changer of the transmission device from Figure 4 uses five phase change values (or phase change sets) to achieve the period (cycle) of five. However, as described in embodiment C5, three different phase change values are present. Therefore, some of the five phase change values necessary for the period (cycle) of five are identical. (As in Figure 6, five phase change values are necessary to perform a phase change that has a period (cycle) of five in the precoded baseband signal z2 'only. Also, as in Figure 26, they are Two phase change values are required for each interval to perform the phase change in both pre-encoded baseband signals z1 'and z2'. These two phase change values are called a phase change set. ideally five sets of phase change to perform a phase change that has a period (cycle) of five in such circumstances). The five phase change values (or phase change sets) necessary for the period (cycle) of five are expressed as P [0], P [1], P [2], P [3] and P [4 ].

The following describes the relationship between the previously defined intervals and the phase, since it belongs to methods for a regular phase change.

For the 1500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is QPSK, the phase change value P [0] is used in 300 intervals, the phase change value P [ 1] is used in 300 intervals, the phase change value P [2] is used in 300

intervals, the phase change value P [3] is used in 300 intervals and the phase change value P [4] is used in 300

intervals. This is due to the fact that any deviation in the use of the phase change value causes enormous influence to be exerted by the most frequently used phase change value, and that the receiving device is dependent on such influence for the quality of data reception

Similarly, for the 1500 intervals described above necessary to transmit the 6000 bits that make up the pair of coded blocks when the modulation method is 16-QAM, the phase change value P [0] is used in 150 intervals, the phase change value P [1] is used in 150 intervals, the phase change value P [2] is used in 150 intervals, the phase change value P [3] is used in 150 intervals and the value of P phase change [4] is used in 150 intervals.

In addition, for the 500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is 64-QAM, the phase change value P [0] is used in 100 intervals, the change value of phase P [1] is used in 100 intervals, the value of phase change P [2] is used in 100

intervals, the phase change value P [3] is used in 100 intervals and the phase change value P [4] is used in 100

intervals.

As described above, a phase change method for regularly varying the phase change value as provided in embodiment C5 requires the preparation of N = 2n + 1 phase change values P [0], P [1 ] ... P [2n-1], P [2n] (where P [0], P [1] ... P [2n-1], P [2n] are expressed as PHASE [0], PHASE [ 1], PHASE [2] ... PHASE [n-1], PHASE [n] (see embodiment C5)). As such, to transmit all the bits that make up the encoded block, the phase change value P [0] is used in K0 intervals, the phase change value P [1] is used in K1 intervals, the change value of phase P [i] is used in Ki intervals (where i = 0, 1, 2, ..., 2n-1, 2n), and the phase change value P [2n] is used in K2n intervals, of so that Condition No. C01 is met.

(Condition # C01)

K0 = K1 ... = Ki = ... K2n. That is, Ka = Kb (Va and Vb where a, b, = 0, 1,2 ... 2n-1, 2n (being a, b integers between 0 and 2n, a

* b).

A phase change method for a regular change of phase change value as provided in embodiment C5 having a period (cycle) of N = 2n +1 requires the preparation of phase change values PHASE [0], PHASE [1], PHASE [2] ... PHASE [n-1], PHASE [n]. As such, to transmit all the bits that make up a single encoded block, the phase change value PHASE [0] is used in G0 intervals, the phase change value PHASE [1] is used in G1 intervals, the value of Phase change PHASE [i] is used in Gi intervals (where i = 0, 1, 2, ..., n-1, n), and the phase change value PHASE [n] is used in Gn intervals, so that Condition No. C01 is met. Condition No. C01 can be modified as follows.

(Condition # C02)

2xG0 = G1 ... = Gi = ... Gn. That is, 2xG0 = Ga (It goes where a = 1, 2 ... n-1, n (being an integer between 1 and n).

So, when a communication system that supports multiple modulation methods selects a method

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supported of this type for use, Condition No. C01 (or Condition No. C02) must be met for the supported modulation method.

However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Condition No. C01 (or Condition No. C02) may not be satisfied for some method of modulation. In such a case, the following condition applies in place of Condition No. C01.

(Condition # C03)

The difference between Ka and Kb satisfies 0 or 1. That is, | Ka - Kb | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... 2n-1, 2n (being a and b integers between 0 and 2n) a t b).

Alternatively, Condition No. C03 can be expressed as follows.

(Condition # C04)

The difference between Ga and Gb satisfies 0, 1, or 2. That is, | Ga - Gb | satisfies 0, 1, or 2 (Va, Vb, where a, b = 1, 2 ... n-1, n (where a and b are integers between 1 and n) a t b)

Y

The difference between 2xG0 and Ga satisfies 0, 1, or 2. That is, | 2xG0 - Ga | satisfies 0, 1, or 2 (Va, where a = 1, 2 ... n-1, n (being an integer between 1 and n)).

Figure 35 illustrates the variable numbers of symbols and intervals required in two coded blocks when using block codes. Figure 35 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams s1 and s2 are transmitted as indicated by the transmission device from Figure 3 and the Figure 12, and the transmission device has two encoders. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 35, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

The transmission device from Figure 3 and the transmission device from Figure 12 each transmit two streams at once, and have two encoders. As such, the two streams each transmit different blocks of code. Therefore, when the modulation method is QPSK, two blocks extracted from s1 and s2 are transmitted in the same interval, for example, a first encoded block extracted from s1 is transmitted, then a second encoded block extracted from s2 is transmitted. As such, 3000 intervals are necessary to transmit the first and second coded blocks.

By the same reasoning, when the modulation method is 16-QAM, 1500 intervals are necessary to transmit all the bits that make up two encoded blocks, and when the modulation method is 64-QAM, 1000 intervals are necessary to transmit all the bits which make up the two coded blocks.

The following describes the relationship between the previously defined intervals and the phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change, which has a period (cycle) of five. That is, the phase changer of the transmission device from Figure 4 uses five phase change values (or phase change sets) to achieve the period (cycle) of five. However, as described in realization C5, three different phase change values are present. Therefore, some of the five phase change values necessary for the period (cycle) of five are identical. (As in Figure 6, five phase change values are necessary to perform a phase change that has a period (cycle) of five in the precoded baseband signal z2 'only. Also, as in Figure 26, they are Two phase change values are required for each interval to perform the phase change in both pre-encoded baseband signals z1 'and z2'. These two phase change values are called a phase change set. ideally five sets of phase change to perform a phase change that has a period (cycle) of five in such circumstances). The five phase change values (or phase change sets) necessary for the period (cycle) of five are expressed as P [0], P [1], P [2], P [3] and P [4 ].

For the 3000 intervals described above necessary to transmit the 6000x2 bits that make up the pair of coded blocks when the modulation method is QPSK, the phase change value P [0] is used in 600 intervals, the phase change value P [1] is used in 600 intervals, the phase change value P [2] is used in 600 intervals, the phase change value P [3] is used in 6100 intervals and the phase change value P [4] ] is used in 600 intervals. This is due to the fact that any deviation in the use of the phase change value produces that

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enormous influence is exerted by the most frequently used phase change value, and that the receiving device is dependent on such influence for the quality of data reception.

In addition, to transmit the first encoded block, the phase change value P [0] is used at 600 times intervals, the phase change value P [1] is used at 600 times intervals, the change value of P phase [2] is used at 600 times intervals, the P phase change value [3] is used at 600 times intervals and the PHASE phase change value [4] is used at 600 times intervals. Additionally, to transmit the second encoded block, the phase change value P [0] is used at 600 times intervals, the phase change value P [1] is used at 600 times intervals, the change value of P phase [2] is used at 600 times intervals, the P phase change value [3] is used at 600 times intervals and the P phase change value [4] is used at 600 times intervals.

Similarly, for the 1500 intervals described above necessary to transmit the 6000x2 bits that make up the pair of coded blocks when the modulation method is 16-QAM, the phase change value P [0] is used in 300 intervals, the phase change value P [1] is used in 300 intervals, the phase change value P [2] is used in 300 intervals, the phase change value P [3] is used in 300 intervals and the value of P phase change [4] is used in 300 intervals.

Additionally, to transmit the first encoded block, the phase change value P [0] is used in the intervals 300 times, the phase change value P [1] is used in the intervals 300 times, the change value of Phase P [2] is used at intervals 300 times, the phase change value P [3] is used at intervals 300 times and the phase change value P [4] is used at intervals 300 times. Additionally, to transmit the second coded block, the phase change value P [0] is used in the intervals 300 times, the phase change value P [1] is used in the intervals 300 times, the change value of P phase [2] is used at intervals 300 times, the phase change value P [3] is used at intervals 300 times, and the phase change value P [4] is used at intervals 300 times.

Similarly, for the 1000 intervals described above necessary to transmit the 6000x2 bits that make up the pair of coded blocks when the modulation method is 64-QAM, the phase change value P [0] is used in 200 intervals, the phase change value P [1] is used in 200 intervals, the phase change value P [2] is used in 200 intervals, the phase change value P [3] is used in 200 intervals, and the value P phase change [4] is used in 200 intervals.

Additionally, to transmit the first encoded block, the phase change value P [0] is used in the intervals 200 times, the phase change value P [1] is used in the intervals 200 times, the change value of P phase [2] is used in the intervals 200 times, the phase change value P [3] is used in the intervals 200 times and the phase change value P [4] is used in the intervals 200 times. Additionally, to transmit the second coded block, the phase change value P [0] is used in the intervals 200 times, the phase change value P [1] is used in the intervals 200 times, the change value of P phase [2] is used in the intervals 200 times, the phase change value P [3] is used in the intervals 200 times, and the phase change value P [4] is used in the intervals 200 times.

As described above, a phase change method for regularly varying the phase change value as provided in embodiment C5 requires the preparation of N = 2n + 1 phase change values P [0], P [1 ] ... P [2n-1], P [2n] (where P [0], P [1] ... P [2n-1], P [2n] are expressed as PHASE [0], PHASE [ 1], PHASE [2] ... PHASE [n-1], PHASE [n] (see embodiment C5)). As such, to transmit all the bits that make up the two encoded blocks, the phase change value P [0] is used in K0 intervals, the phase change value P [1] is used in K1 intervals, the value of Phase change P [i] is used in Ki intervals (where i = 0, 1, 2 ... 2n-1, 2n), and the phase change value P [2n] is used in K2n intervals.

(Condition # C05)

K0 = K1 ... = Ki = ... K2n. That is, Ka = Kb (Va and Vb, where a, b, = 0, 1, 2 ... 2n-1, 2n (being a, b integers between 0 and 2n, atb). To transmit all the bits that make up the first coded block, the phase change value P [0] is used K0.1 times, the phase change value P [1] is used K11 times, the phase change value P [i] is used Ki, 1 times (where i = 0, 1, 2 ... 2n-1, 2n), and the phase change value P [2n] is used K2n, 1 times.

(Condition # C06)

K0,1 = K11 ... = Ki, 1 = ... K2n, 1. That is, Ka, 1 = Kb, 1 (Va and Vb where a, b, = 0, 1,2 ... 2n-1, 2n (being a, b integers between 0

and 2n, a t b).

To transmit all the bits that make up the second coded block, the phase change value P [0] is used K0.2 times, the phase change value P [1] is used K1.2 times, the change value of phase P [i] Ki, 2 is used (where i = 0, 1, 2 ... 2n-1, 2n), and the phase change value P [2n] is used K2n, 2 times.

(Condition # C07)

K0.2 = K1.2 ... = Ki, 2 = ... K2n, 2. That is, Ka, 2 = Kb, 2 (Va and Vb where a, b, = 0, 1,2 ... 2n-1, 2n (being a, b integers between 0 and 2n, a t b).

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A phase change method for regularly varying the phase change value as provided in embodiment C5 having a period (cycle) of N = 2n +1 requires the preparation of phase change values PHASE [0], PHASE [1], PHASE [2] ... PHASE [n-1], PHASE [n]. As such, to transmit all the bits that make up the two coded blocks, the phase change value PHASE [0] is used in G0 intervals, the phase change value PHASE [1] is used in G1 intervals, the value of Phase change PHASE [i] is used in Gi intervals (where i = 0, 1, 2 ... n-1, n) and the phase change value PHASE [n] is used in Gn intervals, so that Condition No. C05 is met.

(Condition # C08)

2xG0 = G1 ... = Gi = ... Gn. That is, 2xG0 = Ga (It goes where a = 1, 2 ... n-1, n (being an integer between 1 and n) .To transmit all the bits that make up the first coded block, the phase change value PHASE [0] is used G01 times, the phase change value PHASE [1] is used G1.1 times, the phase change value PHASE [i] is used Gi, 1 (where i = 0, 1.2 ... n-1, n), and the phase change value PHASE [n] is used Gn, 1 times.

(Condition # C09)

2xG0.1 = G1.1 ... = Gi, 1 = ... Gn, 1. That is, 2xG0,1 = Ga, 1 (It goes where a = 1, 2 ... n-1, n (being an integer between 1 and n). To transmit all the bits that make up the second coded block, the value phase change PHASE [0] is used G0.2 times, the phase change value PHASE [1] is used G1.2 times, the phase change value PHASE [i] is used Gi, 2 (where i = 0, 1, 2 ... n-1, n), and the phase change value PHASE [n] is used Gn, 1 times.

(Condition # C10)

2xG0.2 = G1.2 ... = Gi, 2 = ... Gn, 2 That is, 2xG0.2 = Ga, 2 (Go where a = 1, 2 ... n-1, n (where a an integer between 1 and n).

Then, when a communication system that supports multiple modulation methods selects such a supported method for use, Condition # C05, Condition # C06, and Condition # C07 (or Condition #. ° C08, Condition No. C09, and Condition No. C10) must be met for the supported modulation method.

However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Condition No. C05, the Condition # C06, and Condition # C07 (or Condition # C08, Condition # C09, and Condition # C10) may not be satisfied for some modulation method. In such a case, the following conditions apply in place of Condition No. C05, Condition No. C06, and Condition No. C07.

(Condition # C11)

The difference between Ka and Kb satisfies 0 or 1. That is, | Ka - Kb | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... 2n-1, 2n (being a and b integers between 0 and 2n) a t b).

(Condition # C12)

The difference between Ka, 1 and Kb, 1 satisfies 0 or 1. That is, | Ka, 1 - Kb, 1 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... 2n-

1, 2n (being a and b integers between 0 and 2n) a t b).

(Condition # C13)

The difference between Ka, 2 and Kb, 2 satisfies 0 or 1. That is, | Ka, 2 - Kb, 2 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... 2n- 1, 2n (being a and b integers between 0 and 2n) a t b).

Alternatively, Condition No. C11, Condition No. C12, and Condition No. C13 may be expressed as follows. (Condition # C14)

The difference between Ga and Gb satisfies 0, 1, or 2. That is, | Ga - Gb | satisfies 0, 1, or 2 (Va, Vb, where a, b = 1, 2 ... n-1, n (where a and b are integers between 1 and n) a t b)

Y

the difference between 2xG0 and Ga satisfies 0, 1, or 2. That is, | 2xG0 - Ga | satisfies 0, 1, or 2 (Va, where a = 1, 2 ... n-1, n (being an integer between 1 and n)).

(Condition # C15)

The difference between Ga, 1 and Gb, 1 satisfies 0, 1, or 2. That is, | Ga, 1 - Gb, 1 | satisfies 0, 1, or 2 (Va, Vb, where a, b = 1, 2 ... n-1, n (where a and b are integers between 1 and n) a t b)

Y

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the difference between 2xG0, i and Ga, i satisfies 0, 1, or 2. That is, | 2xG0, i - Ga, i | satisfies 0, 1, or 2 (Va, where a = 1, 2 ... n- 1, n (being an integer between 1 and n))

(Condition # C16)

The difference between Ga, 2 and Gb, 2 satisfies 0, 1, or 2. That is, | Ga, 2 - Gb, 2 | satisfies 0, 1, or 2 (Va, Vb, where a, b = 1, 2 ... n-1, n (where a and b are integers between 1 and n) a t b)

Y

the difference between 2xG0,2 and Ga, 2 satisfies 0, 1, or 2. That is, | 2xG0,2 - Ga, 2 | satisfies 0, 1, or 2 (Va, where a = 1, 2 ... n- 1, n (being an integer between 1 and n))

As described above, the deviation between the phase change values that are used to transmit the encoded blocks is eliminated by creating a relationship between the encoded block and the phase change values. As such, the quality of data reception can be improved for the receiving device.

In the present embodiment, N phase change values (or phase change sets) are necessary to perform a phase change that has a period (cycle) of N with the method for a regular phase change. As such, N phase change values (or phase change sets) P [0], P [1], P [2] ... P [N-2], and P [N-1] are prepared . However, there are schemes to order the phases in the order established with respect to the frequency domain. No limitation is intended in this regard. The N phase change values (or phase change sets) P [0], P [1], P [2] ... P [N-2], and P [N-1] can also change the phases of the blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement as described in embodiment 1. Although the previous examples analyze a phase change scheme with a period (cycle) of N , the same effects are obtainable using N phase change values (or phase change sets) randomly. That is, the N phase change values (or phase change sets) do not always have to have regular periodicity. Provided that the conditions described above are satisfied, improvements in receiving quality data for the receiving device are achievable.

Additionally, given the existence of modes for spatial multiplexing MIMO methods, MIMO methods using a fixed precoding matrix, space-time block coding methods, single stream transmission, and methods that use a phase change Regular, the transmission device (diffuser, base station) can select any one of these transmission methods.

As described in Non-patent Bibliography 3, MIMO methods of spatial multiplexing involve transmitting signals s1 and s2, which are mapped using a selected modulation method, on each of two different antennas. MIMO methods that use a fixed precoding matrix involve precoding only (no change in phase). In addition, the space-time block coding methods are described in the non-patent Bibliography 9, 16 and 17. The single stream transmission methods involve transmitting the signal s1, mapped with a selected modulation method, from an antenna after performing default processing.

Schemes using multi-carrier transmission such as OFDM involve a first group of carriers composed of a plurality of carriers and a second group of carriers composed of a plurality of carriers different from the first group of carriers, and so on! successively, so that the multicarrier transmission is performed with a plurality of carrier groups. For each group of carriers, any of the spatial multiplexing MIMO schemes, MIMO schemes that use a fixed precoding matrix, space-time block coding schemes, single stream transmission, and schemes that use a change in time may be used. regular phase In particular, schemes using a regular phase change are preferably used in a group of (sub-) carrier selected to perform the present embodiment.

When a phase change is made by, for example, a phase change value for the P [i] of X radians in only one precoded baseband signal, the phase changers of Figures 3, 4, 6, 12 , 25, 29, 51 and 53 multiply the precoded baseband signal z2 'by ex. Then, when a phase change is made by, for example, a phase change set for P [i] of X radians and Y radians in both precoded baseband signals, the phase changers from Figures 26 , 27, 28, 52 and 54 multiply the precoded baseband signal z2 'by ejX and multiply the precoded baseband signal z1' by ejY.

[Realization C7]

The present embodiment describes a method for regularly changing the phase, specifically as done in embodiment A1 and in embodiment C6, when coding is performed using block codes as described in Non-patent Bibliography 12 to 15, such as Codes of LDPC (not only QC-LDPc but also codes (block) of LDPC) QC, concatenated LDPC and BCH codes, Turbo codes or Turbo Duo-binary codes and so! successively. The following example considers a case where two flows s1 and s2 are transmitted. When coding has been done using block codes and control information and they are not necessary, the number of bits that make up each block encoded matches the number of bits that it composes

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each block code (even control information and so on described below can be included). When codification has been performed using block codes or the like and control information or the like is required (eg, CRC transmission parameters), then the number of bits that make up each encoded block is the sum of the number of bits that make up the block codes and the number of bits that make up the information.

Figure 34 illustrates the variable numbers of symbols and intervals required in an encoded block when block codes are used. Figure 34 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams are transmitted if and s2 as indicated by the transmission device from Figure 4, and The transmission device has only one encoder. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 34, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

Next, since the transmission device from Figure 4 transmits two streams simultaneously, 1500 of the 3,000 above-mentioned symbols required when the modulation method is QPSK are assigned to s1 and the other 1500 symbols are assigned to s2. As such, 1500 intervals are required to transmit the 1500 symbols (hereinafter intervals) for each of s1 and s2.

By the same reasoning, when the modulation method is 16-QAM, 750 intervals are necessary to transmit all the bits that make up two encoded blocks, and when the modulation method is 64-QAM, 500 intervals are necessary to transmit all the bits which make up the two coded blocks.

The following describes the relationship between the previously defined intervals and the phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change, which has a period (cycle) of five. The phase change values (or phase change sets) prepared to regularly change the phase with a period (cycle) of five are P [0], P [1], P [2], P [3] and P [4]. However, P [0], P [1], P [2], P [3] and P [4] should include at least two different phase change values (ie P [0], P [1 ], P [2], P [3] and P [4] may include identical phase change values). (As in Figure 6, five phase change values are necessary to perform a phase change that has a period (cycle) of five in the precoded baseband signal z2 'only. Also, as in Figure 26, they are Two phase change values are required for each interval to perform the phase change in both pre-encoded baseband signals z1 'and z2'. These two phase change values are called a phase change set. ideally five sets of phase change to perform a phase change that has a period (cycle) of five in such circumstances).

For the 1500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is QPSk, the phase change value P [0] is used in 300 intervals, the phase change value P [ 1] is used in 300 intervals, the phase change value P [2] is used in 300

intervals, the phase change value P [3] is used in 300 intervals and the phase change value P [4] is used in 300

intervals. This is due to the fact that any deviation in the use of the phase change value causes enormous influence to be exerted by the most frequently used phase change value, and that the reception device is dependent on such influence for the quality of data reception

In addition, for the 750 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is 16-QAM, the phase change value P [0] is used in 150 intervals, the change value of phase P [1] is used in 150 intervals, the value of phase change P [2] is used in 150

intervals, the phase change value P [3] is used in 150 intervals and the phase change value P [4] is used in 150

intervals.

In addition, for the 500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is 64-QAM, the phase change value P [0] is used in 100 intervals, the change value of phase P [1] is used in 100 intervals, the value of phase change P [2] is used in 100

intervals, the phase change value P [3] is used in 100 intervals and the phase change value P [4] is used in 100

intervals.

As described above, the phase change values used in the phase change method that regularly change between phase change values with a period (cycle) of N are expressed as P [0], P [1]. .. P [N-2], P [N-1]. However, P [0], P [1] ... P [N-2], P [N-1] should include at least two different phase change values (ie, P [0], P [ 1] ... P [N-2], P [N-1] may include identical phase change values). To transmit all the bits that make up a single coded block, the phase change value P [0] is used in K0 intervals, the

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phase change value P [1] is used in Ki intervals, the phase change value P [i] is used in Ki intervals (where i = 0, 1, 2 ... N-1), and the value Phase change P [N-1] is used in Kn-1 intervals, so that Condition # C17 is met.

(Condition # C17)

K0 = K1 ... = Ki = ... Kn-1. That is, Ka = Kb (Va and Vb where a, b, = 0, 1, 2 ... N-1 (being a and b integers between zero and N-1) a t b).

Then, when a communication system that supports multiple modulation methods selects such a supported method for use, Condition No. C17 must be met for the supported modulation method.

However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Condition # C17 may not be satisfied for some method of modulation. In such a case, the following condition applies in place of Condition No. C17.

(Condition # C18)

The difference between Ka and Kb satisfies 0 or 1. That is, | Ka - Kb | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1 (being a and b integers between 0 and 2n) a t b).

Figure 35 illustrates the variable numbers of symbols and intervals required in two coded blocks when using block codes. Figure 35 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams s1 and s2 are transmitted as indicated by the transmission device from Figure 3 and the Figure 12, and the transmission device has two encoders. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 35, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

The transmission device from Figure 3 and the transmission device from Figure 12 each transmit two streams at once, and have two encoders. As such, the two streams each transmit different blocks of code. Therefore, when the modulation method is QPSK, two blocks extracted from s1 and s2 are transmitted in the same interval, for example, a first coded block extracted from s1 is transmitted, then a second coded block extracted from s2 is transmitted. As such, 3000 intervals are necessary to transmit the first and second coded blocks.

By the same reasoning, when the modulation method is 16-QAM, 1500 intervals are necessary to transmit all the bits that make up two encoded blocks, and when the modulation method is 64-QAM, 1000 intervals are necessary to transmit all the bits which make up the two coded blocks.

The following describes the relationship between the previously defined intervals and the phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change, which has a period (cycle) of five. That is, the phase changer of the transmission device from Figure 4 uses five phase change values (or phase change sets) P [0], P [1], P [2], P [3 ] and P [4] to achieve the period (cycle) of five. However, P [0], P [1], P [2], P [3] and P [4] should include at least two different phase change values (ie P [0], P [1 ], P [2], P [3] and P [4] may include identical phase change values). (As in Figure 6, five phase change values are necessary to perform a phase change that has a period (cycle) of five in the precoded baseband signal z2 'only. Also, as in Figure 26, they are Two phase change values are required for each interval to perform the phase change in both pre-encoded baseband signals z1 'and z2'. These two phase change values are called a phase change set. ideally five sets of phase change to perform a phase change that has a period (cycle) of five in such circumstances). The five phase change values (or phase change sets) necessary for the period (cycle) of five are expressed as P [0], P [1], P [2], P [3] and P [4 ].

For the 3000 intervals described above necessary to transmit the 6000x2 bits that make up the pair of coded blocks when the modulation method is QPSK, the phase change value P [0] is used in 600 intervals, the phase change value P [1] is used in 600 intervals, the phase change value P [2] is used in 600 intervals, the phase change value P [3] is used in 600 intervals and the phase change value P [4] ] is used in 600 intervals. This is due to the fact that any deviation in the use of the phase change value produces that

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enormous influence is exerted by the most frequently used phase change value, and that the receiving device is dependent on such influence for the quality of data reception.

In addition, to transmit the first encoded block, the phase change value P [0] is used at 600 times intervals, the phase change value P [1] is used at 600 times intervals, the change value of P phase [2] is used at 600 times intervals, the P phase change value [3] is used at 600 times intervals, and the PHASE phase change value [4] is used at 600 times intervals. Additionally, to transmit the second encoded block, the phase change value P [0] is used at 600 times intervals, the phase change value P [1] is used at 600 times intervals, the change value of P phase [2] is used at 600 times intervals, the P phase change value [3] is used at 600 times intervals and the P phase change value [4] is used at 600 times intervals.

Similarly, for the 1500 intervals described above necessary to transmit the 6000x2 bits that make up the pair of coded blocks when the modulation method is 16-QAM, the phase change value P [0] is used in 300 intervals, the phase change value P [1] is used in 300 intervals, the phase change value P [2] is used in 300 intervals, the phase change value P [3] is used in 300 intervals and the value of P phase change [4] is used in 300 intervals.

Additionally, to transmit the first encoded block, the phase change value P [0] is used in the intervals 300 times, the phase change value P [1] is used in the intervals 300 times, the change value of Phase P [2] is used at intervals 300 times, the phase change value P [3] is used at intervals 300 times and the phase change value P [4] is used at intervals 300 times. Additionally, to transmit the second coded block, the phase change value P [0] is used in the intervals 300 times, the phase change value P [1] is used in the intervals 300 times, the change value of Phase P [2] is used at intervals 300 times, the phase change value P [3] is used at intervals 300 times and the phase change value P [4] is used at intervals 300 times.

Additionally, for the 1000 intervals described above necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation method is 64-QAM, the phase change value P [0] is used in 200 intervals, the change value phase P [1] is used in 200 intervals, the phase change value P [2] is used in 200 intervals, the phase change value P [3] is used in 200 intervals and the phase change value P [4] is used in 200 intervals.

Additionally, to transmit the first encoded block, the phase change value P [0] is used in the intervals 200 times, the phase change value P [1] is used in the intervals 200 times, the change value of P phase [2] is used in the intervals 200 times, the phase change value P [3] is used in the intervals 200 times and the phase change value P [4] is used in the intervals 200 times. Additionally, to transmit the second coded block, the phase change value P [0] is used in the intervals 200 times, the phase change value P [1] is used in the intervals 200 times, the change value of P phase [2] is used in the intervals 200 times, the phase change value P [3] is used in the intervals 200 times and the phase change value P [4] is used in the intervals 200 times.

As described above, the phase change values used in the phase change method that regularly change between phase change values with a period (cycle) of N are expressed as P [0], P [1]. .. P [N-2], P [N-1]. However, P [0], P [1] ... P [N-2], P [N-1] should include at least two different phase change values (ie, P [0], P [ 1] ... P [N-2], P [N-1] may include identical phase change values). To transmit all the bits that make up a single coded block, the phase change value P [0] is used in K0 intervals, the phase change value P [1] is used in K1 intervals, the phase change value P [i] is used in Ki intervals (where i = 0, 1, 2 ... N-1), and the phase change value P [N-1] is used in Kn-1 intervals, so that Condition No. C19 is met.

(Condition # C19)

N0 = K1 ... = Ki = ... Kn-1. That is, Ka = Kb (Va and Vb where a, b, = 0, 1, 2 ... N-1 (being a and b integers between zero and N-1) a t b).

To transmit all the bits that make up the first encoded block, the phase change value P [0] is used K0.1 times, the phase change value P [1] is used K11 times, the phase change value P [i] Ki is used, 1 times (where i = 0, 1, 2 ... N-1), and the phase change value P [N-1] is used Kn-1.1 times.

(Condition # C20)

K0,1 = K1,1 = ... Ki, 1 = ... Kn-1,1. That is, Ka, 1 = Kb, 1 (Va and Vb where a, b, = 0, 1, 2 ... N-1, a t b).

To transmit all the bits that make up the second coded block, the phase change value P [0] is used K0.2 times, the phase change value P [1] is used K1.2 times, the change value of phase P [i] Ki, 2 (where i = 0, 1, 2 ... N-1) is used, and the phase change value P [N-1] is used Kn-1.2 times.

(Condition # C21)

Nq, 2 = K1,2 = ... Ki, 2 = ... Kn-1,2. That is, Ka, 2 = Kb, 2 (Va and Vb where a, b, = 0, 1,2 ... N-1, a t b).

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Then, when a communication system that supports multiple modulation methods selects such a supported method for use, Condition # C19, Condition # C20 and Condition # C21 are preferably met for the method of modulation supported.

However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Condition No. C19, the Condition # C20 and Condition # C21 may not be satisfied for some method of modulation. In such a case, the following conditions apply in place of Condition No. C19, Condition No. C20, and Condition No. C21.

(Condition # C22)

The difference between Ka and Kb satisfies 0 or 1. That is, | Ka - Kb | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1 (being a and b integers between 0 and N-1) a t b).

(Condition # C23)

The difference between Ka, 1 and Kb, 1 satisfies 0 or 1. That is, | Ka, 1 - Kb, 1 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1 (being a and b integers between 0 and N-1) a t b).

(Condition # C24)

The difference between Ka, 2 and Kb, 2 satisfies 0 or 1. That is, | Ka, 2 - Kb, 2 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1 (being a and b integers between 0 and N-1) a t b).

As described above, the deviation between the phase change values that are used to transmit the encoded blocks is eliminated by creating a relationship between the encoded block and the phase change values. As such, the quality of data reception can be improved for the reception device.

In the present embodiment, N phase change values (or phase change sets) are necessary to perform a phase change that has a period (cycle) of N with the method for a regular phase change. As such, N phase change values (or phase change sets) P [0], P [1], P [2] ... P [N-2] and P [N-1] are prepared. However, there are methods for ordering the phases in the order established with respect to the frequency domain. No limitation is intended in this regard. The N phase change values (or phase change sets) P [0], P [1], P [2] ... P [N-2] and P [N-1] can also change the phases of the blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement as described in embodiment 1. Although the above examples analyze a phase change method with a period (cycle) of N, the same effects are obtainable using N phase change values (or phase change sets) randomly. That is, the N phase change values (or phase change sets) do not always have to have regular periodicity. Provided that the conditions described above are satisfied, huge improvements in receiving quality data for the receiving device are achievable.

Additionally, given the existence of modes for spatial multiplexing MIMO methods, MIMO methods that use a fixed precoding matrix, space-time block coding methods, single stream transmission, and methods that use a phase change Regular, the transmission device (diffuser, base station) can select any one of these transmission methods.

As described in Non-patent Bibliography 3; MIMO methods of spatial multiplexing involve transmitting signals s1 and s2, which are mapped using a selected modulation method, on each of two different antennas. MIMO methods that use a fixed precoding matrix involve precoding only (no change in phase). In addition, the space-time block coding methods are described in the non-patent Bibliography 9, 16 and 17. The single stream transmission methods involve transmitting the signal s1, mapped with a selected modulation method, from an antenna after performing default processing.

Schemes using multi-carrier transmission such as OFDM involve a first group of carriers composed of a plurality of carriers and a second group of carriers composed of a plurality of carriers different from the first group of carriers, and so on! successively, so that the multicarrier transmission is performed with a plurality of carrier groups. For each group of carriers, any of the spatial multiplexing MIMO schemes, MIMO schemes that use a fixed precoding matrix, space-time block coding schemes, single stream transmission, and schemes that use a change in time may be used. regular phase In particular, schemes using a regular phase change are preferably used in a group of (sub-) carrier selected to perform the present embodiment.

When a phase change is made by, for example, a phase change value for P [i] of X radians in only one precoded baseband signal, the phase changers of Figures 3, 4, 6, 12, 25, 29, 51 and

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53 multiply the precoded baseband signal z2 'by ejX. Then, when a phase change is made by, for example, a set of phase change by P [i] of X radians and Y radians in both precoded baseband signals, the phase changers from Figures 26 , 27, 28, 52 and 54 multiply the precoded baseband signal z2 'by ejX and multiply the precoded baseband signal z1' by ejY.

[Realization D1]

The present embodiment is first described as a variation of embodiment 1. Figure 67 illustrates a sample transmission device belonging to the present embodiment. The components thereof that operate identically to those of Figure 3 use the same reference numbers between them, and their description is omitted for simplicity below. Figure 67 differs from Figure 3 in the insertion of a 6702 baseband signal changer that follows directly the weighting units. Accordingly, the following explanations focus primarily on the baseband signal changer 6702.

Figure 21 illustrates the configuration of the weighting units 308A and 308B. The area of Figure 21 enclosed in the dashed line represents one of the weighting units. The baseband signal 307A is multiplied by w11 to obtain w11s1 (t), and multiplied by w21 to obtain w21s1 (t). Similarly, baseband signal 307B is multiplied by w12 to obtain w12s2 (t), and multiplied by w22 to obtain w22s2 (t). Then you get z1 (t) = w11s1 (t) + w12s2 (t) and z2 (t) = w21s1 (t) + w22s22 (t). At this point, as explained in embodiment 1, s1 (t) and s2 (t) are the baseband signals modulated according to a modulation method such as BPSK, QPSK, 8-PsK, 16-QAM, 32 -QAM, 64-QAM, 256-QAM, 16-APSK and so! successively. Both weighting units perform weighting using a fixed precoding matrix. The precoding matrix uses, for example, the method of Calculation 62 (formula 62), and satisfies the conditions of Calculation 63 (formula 63) or Calculation 64 (formula 64), all found below. However, this is only an example. The value of a is not limited to Calculation 63 (formula 63) and Calculation 64 (formula 64), and can be, for example, 1, it can be 0 (a is preferably a real number greater than or equal to 0, but it can also be an imaginary number).

At this point, the precoding matrix is [Calculation 62]

fwll *> 12 ^ ^ w> 21 w22y

image41

f JO

and

jo

laxe

image42

(formula 62)

In Calculation 62 (formula 62), above, a is provided by: [Calculation 63]

to

Vz + 4

■ Jl + 2

(formula 63)

As an alternative, in the calculation. 62 (formula 62), above, can be provided by:

V2 + 3 + V5

(X - —p = ------------- p = - (formula 64)

V2 + 3-V5

As an alternative, the precoding matrix is not restricted to that of Calculation 62 (formula 62), but may also be:

[Calculation 65]

image43

where a = Aejd11, b = Bejd12, c = Cejd21, and d = Dejd22. In addition, one of a, b, c, and d can be zero. For example:

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(1) a can be zero while b, c, and d are nonzero, (2) b can be zero while a, c, and d are nonzero, (3) c can be zero while a, b, and d are nonzero, or (4) d can be zero while a, b, and c are nonzero.

Alternatively, any two of a, b, c, and d can be zero. For example, (1) a and d can be zero while b and c are nonzero, or (2) b and c can be zero while a and d are nonzero.

When any of the modulation method, error correction codes, and the coding rate thereof are changed, the precoding matrix in use can also be established and changed, or the same precoding matrix can be used as is.

Next, the baseband signal changer 6702 is described from Figure 67. The baseband signal changer 6702 takes the weighted signal 309A and the weighted signal 316B as input, performs baseband signal change, and issues the changed baseband signal 6701A and the changed baseband signal 6701B. The details of the change of baseband signal are as described with reference to Figure 55. The change of baseband signal made in the present embodiment differs from that of Figure 55 in terms of the signal used to change. The following describes the change of baseband signal of the present embodiment with reference to Figure 68.

In Figure 68, the weighted signal 309A (p1 (i)) has a phase I component of Ip1 (i) and a quadrature component Q of Qp1 (i), while the weighted signal 316B (p2 (i)) has a phase I component of Ip2 (i) and a quadrature component Q of Qp2 (i). In contrast, the changed baseband signal 6701A (q1 (i)) has a phase I component of Iq1 (i) and a quadrature component Q of Qq1 (i), while the changed baseband signal 6701B (q2 ( i) has a phase I component of Iq2 (i) and a quadrature component Q of Qq2 (i). (At this point, i represents (time or frequency order (carrier). In the example of Figure 67, i represents time, although i can also represent frequency (carrier) when Figure 67 is applied to an OFDM scheme, as in Figure 12. These points are detailed below).

At this point, the baseband components are changed using the 6702 baseband signal changer, so that:

• For the changed baseband signal q1 (i), the phase I component may be Ip1 (i) while the quadrature component Q may be Qp2 (i), and for the changed baseband signal q2 (i) , the phase I component can be Ip2 (i) while the quadrature component Q can be Qp1 (i). The modulated signal corresponding to the changed baseband signal q1 (i) is transmitted by the transmission antenna 1 and the modulated signal corresponding to the changed baseband signal q2 (i) is transmitted from the transmission antenna 2, simultaneously on a common frequency. As such, the modulated signal corresponding to the changed baseband signal q1 (i) and the modulated signal corresponding to the changed baseband signal q2 (i) are transmitted from different antennas, simultaneously on a common frequency. As an alternative,

For the changed baseband signal q1 (i), the phase component may be Ip1 (i) while the quadrature component may be Ip2 (i), and for the changed baseband signal q2 (i), the component in phase it can be Qp1 (i) while the quadrature component can be Qp2 (i).

For the changed baseband signal q1 (i), the phase component may be Ip2 (i) while the quadrature component may be Ip1 (i), and for the changed baseband signal q2 (i), the component in phase it can be Qp1 (i) while the quadrature component can be Qp2 (i).

For the changed baseband signal q1 (i), the phase component may be Ip1 (i) while the quadrature component may be Ip2 (i), and for the changed baseband signal q2 (i), the component in phase it can be Qp2 (i) while the quadrature component can be Qp1 (i).

For the changed baseband signal q1 (i), the phase component may be Ip2 (i) while the quadrature component may be Ip1 (i), and for the changed baseband signal q2 (i), the component in phase it can be Qp2 (i) while the quadrature component can be Qp1 (i).

For the changed baseband signal q1 (i), the phase component may be Ip1 (i) while the quadrature component may be Qp2 (i), and for the changed baseband signal q2 (i), the component in phase it can be Qp1 (i) while the quadrature component can be Ip2 (i).

For the changed baseband signal q1 (i), the phase component may be Qp2 (i) while the quadrature component may be Ip1 (i), and for the changed baseband signal q2 (i), the component in phase it can be Ip2 (i) while the quadrature component can be Qp1 (i).

For the changed baseband signal q1 (i), the phase component may be Qp2 (i) while the quadrature component may be Ip1 (i), and for the changed baseband signal q2 (i), the component in phase it can be Qp1 (i) while the quadrature component can be Ip2 (i).

For the changed baseband signal q2 (i), the phase component may be Ip1 (i) while the quadrature component may be Ip2 (i), and for the changed baseband signal q1 (i), the component in phase it can be Qp1 (i) while the quadrature component can be Qp2 (i).

For the changed baseband signal q2 (i), the phase component may be Ip2 (i) while the quadrature component may be Ip1 (i), and for the changed baseband signal q1 (i), the component in phase it can be Qp1 (i) while the quadrature component can be Qp2 (i).

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• For the changed baseband signal q2 (i), the phase component may be Ipi (i) while the quadrature component may be Ip2 (i), and for the changed baseband signal q1 (i), the phase component can be Qp2 (i) while the quadrature component can be Qpi (i).

• For the changed baseband signal q2 (i), the phase component may be Ip2 (i) while the quadrature component may be Ipi (i), and for the changed baseband signal q1 (i), the phase component can be Qp2 (i) while the quadrature component can be Qp1 (i).

• For the changed baseband signal q2 (i), the phase component may be Ip1 (i) while the quadrature component may be Qp2 (i), and for the changed baseband signal q1 (i), the Phase component can be Ip2 (i) while the quadrature component can be Qp1 (i).

• For the changed baseband signal q2 (i), the phase component may be Ip1 (i) while the quadrature component may be Qp2 (i), and for the changed baseband signal q1 (i), the Phase component can be Qp1 (i) while the quadrature component can be Ip2 (i).

• For the changed baseband signal q2 (i), the phase component may be Qp2 (i) while the quadrature component may be Ip1 (i), and for the changed baseband signal q1 (i), the Phase component can be Ip2 (i) while the quadrature component can be Qp1 (i).

• For the changed baseband signal q2 (i), the phase component may be Qp2 (i) while the quadrature component may be Ip1 (i), and for the changed baseband signal q1 (i), the Phase component can be Qp1 (i) while the quadrature component can be Ip2 (i).

Alternatively, the weighted signals 309A and 316B are not limited to the previously described change of the phase component and the quadrature component. The change can be made in phase components and quadrature components larger than those of the two signals.

Also, although the above examples describe change made in baseband signals that have a common time indication (common (sub-) carrier frequency), the baseband signals that are changed do not necessarily need to have a time indication common (frequency of (sub-) carrier) common). For example, any of the following is possible.

For the changed baseband signal q1 (i), the phase component may be Ip1 (i + v) while the quadrature component may be Qp2 (i + w), and for the changed baseband signal q2 (i ), the phase component can be Ip2 (i + w) while the quadrature component can be Qp1 (i + v).

For the changed baseband signal q1 (i), the phase component may be Ip1 (i + v) while the quadrature component may be Ip2 (i + w), and for the changed baseband signal q2 (i ), the phase component can be Qp1 (i + v) while the quadrature component can be Qp2 (i + w).

For the changed baseband signal q1 (i), the phase component may be Ip2 (i + w) while the quadrature component may be Ip1 (i + v), and for the changed baseband signal q2 (i ), the phase component can be Qp1 (i + v) while the quadrature component can be Qp2 (i + w).

For the changed baseband signal q1 (i), the phase component may be Ip1 (i + v) while the quadrature component may be Ip2 (i + w), and for the changed baseband signal q2 (i ), the phase component can be Qp2 (i + w) while the quadrature component can be Qp1 (i + v).

For the changed baseband signal q1 (i), the phase component may be Ip2 (i + w) while the quadrature component may be Ip1 (i + v), and for the changed baseband signal q2 (i ), the phase component can be Qp2 (i + w) while the quadrature component can be Qp1 (i + v).

For the changed baseband signal q1 (i), the phase component may be Ip1 (i + v) while the quadrature component may be Qp2 (i + w), and for the changed baseband signal q2 (i ), the phase component can be Qp1 (i + v) while the quadrature component can be Ip2 (i + w).

For the changed baseband signal q1 (i), the phase component may be Qp2 (i + w) while the quadrature component may be Ip1 (i + v), and for the changed baseband signal q2 (i ), the phase component can be Ip2 (i + w) while the quadrature component can be Qp1 (i + v).

For the changed baseband signal q1 (i), the phase component may be Qp2 (i + w) while the quadrature component may be Ip1 (i + v), and for the changed baseband signal q2 (i ), the phase component can be Qp1 (i + v) while the quadrature component can be Ip2 (i + w).

For the changed baseband signal q2 (i), the phase component may be Ip1 (i + v) while the quadrature component may be Ip2 (i + w), and for the changed baseband signal q1 (i ), the phase component can be Qp1 (i + v) while the quadrature component can be Qp2 (i + w).

For the changed baseband signal q2 (i), the phase component may be Ip2 (i + w) while the quadrature component may be Ip1 (i + v), and for the changed baseband signal q1 (i ), the phase component can be Qp1 (i + v) while the quadrature component can be Qp2 (i + w).

For the changed baseband signal q2 (i), the phase component may be Ip1 (i + v) while the quadrature component may be Ip2 (i + w), and for the changed baseband signal q1 (i ), the phase component can be Qp2 (i + w) while the quadrature component can be Qp1 (i + v).

For the changed baseband signal q2 (i), the phase component may be Ip2 (i + w) while the quadrature component may be Ip1 (i + v), and for the changed baseband signal q1 (i ), the phase component can be Qp2 (i + w) while the quadrature component can be Qp1 (i + v).

For the changed baseband signal q2 (i), the phase component may be Ip1 (i + v) while the quadrature component may be Qp2 (i + w), and for the changed baseband signal q1 (i ), the component in

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phase can be Ip2 (i + w) while the quadrature component can be Qpi (i + v).

• For the changed baseband signal q2 (i), the phase component may be Ipi (i + v) while the quadrature component may be Qp2 (i + w), and for the changed baseband signal q1 ( i), the phase component can be Qpi (i + v) while the quadrature component can be Ip2 (i + w).

• For the changed baseband signal q2 (i), the phase component may be Qp2 (i + w) while the quadrature component may be Ipi (i + v), and for the changed baseband signal q1 ( i), the phase component can be Ip2 (i + w) while the quadrature component can be Qp1 (i + v).

• For the changed baseband signal q2 (i), the phase component may be Qp2 (i + w) while the quadrature component may be Ip1 (i + v), and for the changed baseband signal q1 ( i), the phase component can be Qp1 (i + v) while the quadrature component can be Ip2 (i + w).

At this point, the weighted signal 309A (p1 (i)) has a phase I component of Ip1 (i) and a quadrature component Q of Qp1 (i), while the weighted signal 316B (p2 (i)) has a phase I component of Ip2 (i) and a quadrature component Q of Qp2 (i). In contrast, the changed baseband signal 6701A (q1 (i)) has a phase I component of Iq1 (i) and a quadrature component Q of Qq1 (i), while the changed baseband signal 6701B (q2 ( i)) has a phase component Iq2 (i) and a quadrature component Q of Qq2 (i).

In Figure 68, as described above, the weighted signal 309A (p1 (i)) has a phase I component of Ip1 (i) and a quadrature component Q of Qp1 (i), while the weighted signal 316B ( p2 (i)) has a phase I component of Ip2 (i) and a quadrature component Q of Qp2 (i). In contrast, the changed baseband signal 6701A (q1 (i)) has a phase I component of Iq1 (i) and a quadrature component Q of Qq1 (i), while the changed baseband signal 6701B (q2 ( i)) has a phase component Iq2 (i) and a quadrature component Q of Qq2 (i).

As such, the phase I component of Iq1 (i) and the quadrature component Q of Qq1 (i) of the changed baseband signal 6701A (q1 (i)) and the phase component Iq2 (i) and the component Quadrature Q of Qq2 (i) of the baseband signal 6701B (q2 (i)) are expressible as any of the above.

As such, the modulated signal corresponding to the changed baseband signal 6701A (q1 (i)) is transmitted from the transmission antenna 312A, while the modulated signal corresponding to the changed baseband signal 6701B (q2 (i )) is transmitted from the 312B transmission antenna, both being transmitted simultaneously on a common frequency. Therefore, the modulated signals corresponding to the changed baseband signal 6701A (q1 (i)) and the changed baseband signal 6701B (q2 (i)) are transmitted from different antennas, simultaneously on a common frequency.

The phase changer 317B takes the changed baseband signal 6701B and the signal processing method information 315 as input and regularly changes the phase of the changed baseband signal 6701B to emit. This regular change is a phase change made according to a predetermined phase change pattern that has a predetermined period (cycle) (for example, each n symbols (where n is an integer, n> 1) or at a predetermined interval). The phase change pattern is described in detail in embodiment 4.

The wireless unit 310B takes the post-phase change signal 309B as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification, and so on! successively, next it emits the transmission signal 311B. The transmission signal 311B is then broadcast as radio waves by means of a 312B antenna.

Figure 67, like Figure 3, is described as having a plurality of encoders. However, Figure 67 may also have an encoder and a distributor like Figure 4. In such a case, the signals emitted by the distributor are the respective input signals for the interleaver, while the subsequent processing remains as it has been. described above for Figure 67, despite the changes required in this way.

Figure 5 illustrates an example of a frame configuration in the time domain for a transmission device according to the present embodiment. Symbol 500_1 is a symbol for notifying the reception device of the transmission method. For example, symbol 500_1 carries information such as the error correction method used to transmit data symbols, the codification rate thereof, and the modulation method used to transmit data symbols.

The symbol 501_1 is for estimating channel fluctuations for the modulated signal z1 (t) (where t is time) transmitted by the transmission device. Symbol 502_1 is a data symbol transmitted by the modulated signal z1 (t) as the symbol number u (in the time domain). Symbol 503_1 is a data symbol transmitted by the modulated signal z1 (t) as the symbol number u + 1.

The symbol 501_2 is for estimating channel fluctuations for the modulated signal z2 (t) (where t is time) transmitted by the transmission device. Symbol 502_2 is a data symbol transmitted by the modulated signal z2 (t) as the symbol number u. Symbol 503_2 is a data symbol transmitted by the modulated signal z1 (t) as the symbol number u + 1.

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At this point, the slots of z1 (t) and z2 (t) that have the same time indication (identical timing) are transmitted from the transmitting antenna using the same (shared / common) frequency.

The following describes the relationships between the modulated signals z1 (t) and z2 (t) transmitted by the transmission device and the signals received r1 (t) and r2 (t) received by the reception device.

In Figure 5, 504 No. 1 and 504 No. 2 indicate transmitting antennas of the transmitting device, while 505 # 1 and 505 # 2 indicate receiving antennas of the receiving device. The transmission device transmits the modulated signal z1 (t) from the transmission antenna 504 # 1 and transmits the modulated signal z2 (t) from the transmission antenna 504 # 2. At this point, the modulated signals z1 (t) and z2 (t) are assumed to occupy the same (shared / common) frequency (bandwidth). The channel fluctuations in the transmitting antennas of the transmitting device and the antennas of the receiving device are hn (t), h12 (t), h21 (t) and h22 (t), respectively. Assuming that the receiving antenna 505 # 1 of the receiving device receives the received signal r1 (t) and that the receiving antenna 505 # 2 of the receiving device receives the received signal r2 (t), the following relation.

[Calculation 66]

image44

Figure 69 belongs to the weighting method (precoding method), the baseband change method and the phase change method of the present embodiment. The weighting unit 600 is a combined version of the weighting units 308A and 308B from Figure 67. As shown, the flow s1 (t) and the flow s2 (t) correspond to the baseband signals 307A and 307B of Figure 3. That is, the flows s1 (t) and s2 (t) are the baseband signals composed of a phase I component and a quadrature component Q according to mapping by a modulation method such as QPSK , 16-QAM and 64-QAM. As indicated by the frame configuration of Figure 69, the flow s1 (t) is represented as s1 (u) in the number of symbol u, as s1 (u + 1) in the number of symbol u + 1, and ace! successively. Similarly, the flow s2 (t) is represented as s2 (u) in the number of symbol u, as s2 (u + 1) in the number of symbol u + 1, and so! successively. The weighting unit 600 takes the baseband signals 307A (s1 (t)) and 307B (s2 (t)) as! as the signal processing method information 315 from Figure 67 as input, it performs weighting in accordance with the signal processing method information 315, and issues the weighted signals 309A (p1 (t)) and 316B (p2 (t)) from Figure 67.

At this point, given the vector W1 = (w11, w12) from the first row of the fixed precoding matrix F, p1 (t) can be expressed as Calculation 67 (formula 67), below.

[Calculation 67]

Jpl (0 = W lsl (£) (formula 67)

At this point, given the vector W2 = (w21, w22) from the first row of the fixed precoding matrix F, p2 (t) can be expressed as Calculation 68 (formula 68), below.

[Calculation 68]

p2 (t) = W2s2 (t) (formula 68)

Accordingly, the precoding matrix F can be expressed as follows.

[Calculation 69]

F =

rw \\

vw21

wl2 ^

w22 ^

(formula 69)

After the baseband signals have been changed, the changed baseband signal 6701A (q1 (i)) has a phase I component of Iq1 (i) and a quadrature component Q of Qp1 (i), and the Changed baseband signal 6701B (q2 (i)) has a phase I component of Iq2 (i) and a quadrature component Q of Qq2 (i). The relationships between all these as established above. When the phase changer uses the phase change formula and (t), the post-phase change baseband signal 309B (q'2 (i)) is provided by Calculation 70 (formula 70), below. .

[Calculation 70]

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^ 2 (f) and (/) <^ 2 (f) (formula 70)

At this point, and (t) is a phase change formula that meets a predetermined method. For example, given a period (cycle) of four and time indication u, the phase change formula can be expressed as Calculation 71 (formula 71), below.

[Calculation 71]

and (^ ^ (formula 71)

Similarly, the phase change formula for the time indication u + 1 can be, for example, as provided by Calculation 72 (formula 72).

[Calculation 72]

and (u + l)

image45

(formula 72)

That is, the phase change formula for the time indication u + k is generalized in Calculation 73 (formula 73). [Calculation 73]

. k n

and (u + k) = eJl (formu, a73)

Note that Calculation 71 (formula 71) to Calculation 73 (formula 73) are provided only as an example of a regular phase change.

The regular phase change is not restricted to a period (cycle) of four. Improved reception capabilities (error correction capabilities, to be exact) can potentially be promoted in the receiving device by increasing the number (this does not mean that a longer period (cycle) is better, although avoiding small numbers such as two is equally ideal) of period (cycle).

Additionally, although Calculation 71 (formula 71) to Calculation 73 (formula 73), above, represents a configuration in which a phase change is carried out through rotation by consecutive predetermined phases (in the previous formula, each p / 2), the phase change does not need to be rotated by a constant amount but can also be random. For example, according to the predetermined period (cycle) of y (t), the phase can be changed through sequential multiplication as shown in Calculation 74 (formula 74) and in Calculation 75 (formula 75). The key point of the regular phase change is that the phase of the modulated signal is changed regularly. The rate of variance of degree of phase change is preferably as even as possible, such as from -p radians to p radians. However, since this refers to a distribution, random variance is also possible.

[Calculation 74]

.n .2 7t Hit An

eJ0 ->e'7-> eT ->e'T-> 7

.6 n .7x_ .8n_ .9 n

- »ein -> e 5 -» eJ 5 -> eJ 5 - »e 5

[Calculation 75]

.7> n .x

in -a e-> eJT ^ er2 * -A e7

.3, 5n .lx

j — nj— j—

-> e 4 -ye 4 4

As such, the weighting unit 600 of Figure 6 performs precoding using fixed predetermined precoded weights, the baseband signal changer performs baseband signal change as described above, and the phase changer changes the phase of the signal introduced to it while regularly varying the degree of change.

(formula 74)

(formula 75)

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When a specialized precoding matrix is used in the LOS environment, reception quality is likely to improve greatly. However, depending on the direct wave conditions, the phase and amplitude components of the direct wave may differ greatly from the specialized precoding matrix, upon receipt. The LOS environment has certain rules. Therefore, the quality of data reception is greatly improved through a regular change of the transmission signal phase that complies with these rules. The present invention offers a method of signal processing to improve the LOS environment.

Figure 7 illustrates a sample configuration of a receiving device 700 belonging to the present embodiment. The wireless unit 703_X receives, as input, the received signal 702_X received by the antenna 701_X, performs processing such as frequency conversion, quadrature demodulation, and the like, and emits the baseband signal 704_X.

The channel fluctuation estimator 705_1 for the modulated signal z1 transmitted by the transmission device takes a baseband signal 704_X as input, extracts the reference symbol 501_1 for the channel estimation from Figure 5, estimates the value of hn from Calculation 66 (formula 66), and issues channel estimate signal 706_1.

The channel fluctuation estimator 705_2 for the modulated signal z2 transmitted by the transmission device takes a baseband signal 704_X as input, extracts the reference symbol 501_2 for the channel estimation from Figure 5, estimates the value of h12 from Calculation 66 (formula 66), and emits channel estimate signal 706_2.

The wireless unit 703_Y receives, as input, the received signal 702_Y received by the antenna 701_X, performs processing such as frequency conversion, quadrature demodulation, and the like, and emits the baseband signal 704_Y.

The channel fluctuation estimator 707_1 for the modulated signal z1 transmitted by the transmission device takes a baseband signal 704_Y as input, extracts the reference symbol 501_1 for the channel estimation from Figure 5, estimates the value of h21 from Calculation 66 (formula 66), and issues channel estimate signal 708_1.

The channel fluctuation estimator 707_2 for the modulated signal z2 transmitted by the transmission device takes a baseband signal 704_Y as input, extracts the reference symbol 501_2 for the channel estimation from Figure 5, estimates the value of h22 from Calculation 66 (formula 66), and emits channel estimate signal 708_2.

A control information decoder 709 receives the baseband signal 704_X and the baseband signal 704_Y as input, detects the symbol 500_1 indicating the transmission method from Figure 5, and issues a method information signal of 710 transmission device transmission.

A signal processor 711 takes the baseband signals 704_X and 704_Y, the channel estimation signals 706_1, 706_2, 708_1, and 708_2, and the transmission method information signal 710 as input, performs detection and decoding, and then issues the received data 712_1 and 712_2.

Next, the operations of signal processor 711 are described in detail from Figure 7. Figure 8 illustrates a sample configuration of signal processor 711 belonging to the present embodiment. As shown, the signal processor 711 is mainly composed of an internal MIMO detector, a flexible input / flexible output decoder, and a coefficient generator. Non-patent Bibliography 2 and Non-Patent Bibliography 3 describe the method of iterative decoding with this structure. The MIMO system described in the non-patent Bibliography 2 and in the non-patent Bibliography 3 is a spatial multiplexing MIMO system, although the present embodiment differs from the non-patent Bibliography 2 and the non-patent Bibliography 3 when describing a MIMO system that regularly changes the phase over time, while using the precoding matrix and changing the baseband signal. Taking the (channel) matrix H (t) of Calculation 66 (formula 66), where F is the precoding weighting matrix from Figure 69 (at this point, a fixed precoding matrix remains unchanged for a given received signal ) and being the phase change formula used by the phase changer from Figure 69 Y (t) (at this point, Y (t) changes with time t), then since the change of band signal base, the reception vector R (t) = (r1 (t), r2 (t)) T and the flow vector S (t) = (s1 (t), s2 (t)) T lead to the decoding method of Non-patent Bibliography 2 and Non-Patent Bibliography 3, thus enabling MIMO detection.

Accordingly, the coefficient generator 819 from Figure 8 takes a transmission method information signal 818 (corresponding to 710 from Figure 7) indicated by the transmission device (information to specify the precoding matrix fixed in use and the phase change pattern used when the phase is changed) and emits an 820 signal processing method information signal.

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Internal MIMO detector 803 takes signal processing method information signal 820 as input and performs iterative detection and decoding using the signal. The operations are described below.

The processing unit illustrated in Figure 8 should use a processing method, as illustrated in Figure 10, to perform iterative decoding (iterative detection). First, the detection of a code word (or a frame) of the modulated signal (flow) is performed and of a code word (or a frame) of the modulated signal (flow) s2. As a result, the flexible input / flexible output decoder obtains the logarithmic probability ratio of each bit of the code word (or frame) of the modulated signal (flow) if and of the code word (or frame) of the signal modulated (flow) s2. Next, the logarithmic probability relationship is used to perform a second round of detection and decoding. These operations (referred to as iterative decoding (iterative detection)) are performed multiple times. The following explanations focus on the method of creating the logarithmic probability relation of a symbol at a specific time in a frame.

In Figure 8, a memory 815 takes a baseband signal 801X (corresponding to the baseband signal 704_X from Figure 7), the group of 802X channel estimation signal (which corresponds to estimation signals of channel 706_1 and 706_2 from Figure 7), the baseband signal 801Y (corresponding to the baseband signal 704_Y from Figure 7), and the channel estimation group of channel 802Y (corresponding to channel estimation signals 708_1 and 708_2 from Figure 7) as input, performs iterative decoding (iterative detection), and stores the resulting matrix as a group of transformed channel signal. Memory 815 then issues the signals described above as necessary, specifically as the baseband signal 816X, the transformed channel estimation signal group 817X, the baseband signal signal 816Y and the transformed channel estimation signal group 817Y.

Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection).

(Initial Detection)

The internal MIMO detector 803 takes an 801X baseband signal, the 802X channel estimation signal group, the 801Y baseband signal group and the 802Y channel estimation signal group as input. At this point, the modulation method for the modulated signal (flow) s1 and the modulated signal (flow) s2 is described as 16-QAM.

The internal MIMO detector 803 first calculates a candidate signal point corresponding to the baseband signal 801X from the 802X and 802Y channel estimation signal groups. Figure 11 represents such a calculation. In Figure 11, each black point is a candidate signal point in the IQ plane. Since the modulation method is 16-QAM, there are 256 candidate signal points. (However, Figure 11 is only a representation and does not indicate all 256 candidate signal points). Being the four bits transmitted in the modulated signal s1 b0, b1, b2 and b3 and the four bits being transmitted in the modulated signal s2 b4, b5, b6 and b7, there are candidate signal points corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) in Figure 11. The square distance between each candidate signal point and each received signal point 1101 (corresponding to the baseband signal 801X) is calculated below. The euclldea square distance between each point is divided by the noise variance o2. Therefore, EX is calculated (b0, b1, b2, b3, b4, b5, b6, b7). That is, the square distance euclldea between a candidate signal point that corresponds to (b0, b1, b2, b3, b4, b5, b6, b7) and a received signal point is divided by the noise variance. At this point, each of the baseband signals and the modulated signals s 1 and s2 is a complex signal.

Similarly, the internal MIMO detector 803 calculates candidate signal points corresponding to the baseband signal 801Y from the 802X channel estimation signal group and the 802Y channel estimation signal group, calculates the euclldea square distance between each of the candidate signal points and the received signal points (corresponding to the baseband signal 801Y), and divide the square distance euclldea by the noise variance o2. Therefore, Ey (b0, b1, b2, b3, b4, b5, b6, b7) is calculated. That is, Ey is the square distance euclldea between a candidate signal point that corresponds to (b0, b1, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance.

Then EX (b0, b1, b2, b3, b4, b5, b6, b7) + EY (b0, b1, b2, b3, b4, b5, b6, b7) = E (b0, b1, b2, b3, b4, b5, b6, b7).

The internal MIMO detector 803 emits E (b0, b1, b2, b3, b4, b5, b6, b7) as signal 804.

Logarithmic probability calculator 805A takes the signal 804 as input, calculates the logarithmic probability of bits b0, b1, b2 and b3, and emits a logarithmic probability signal 806A. Note that this calculation of log probability produces the log probability of a bit being 1 and the log probability of a bit being 0. The calculation method is as shown in Calculation 28 (formula 28), in Calculation 29 ( formula 29), and in Calculus 30 (formula 30), and the details are provided by the non-patent Bibliography 2 and 3.

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Similarly, the logarithmic probability calculator 805B takes the signal 804 as input, calculates the logarithmic probability of bits b4, b5, b6 and b7, and issues the logarithmic probability signal 806B.

A deinterleaver (807A) takes the logarithmic probability signal 806A as input, performs deintercalation corresponding to that of the interleaver (the interleaver (304A) from Figure 67), and issues the logarithmic probability signal uninterleaved 808A.

Similarly, a deinterleaver (807B) takes the logarithmic probability signal 806B as input, performs deintercalation corresponding to that of the interleaver (the interleaver (6704B) from Figure 67), and issues the logarithmic probability signal uninterleaved 808B .

Logarithmic probability ratio calculator 809A takes the logarithmic probability signal uninterleaved 808A as input, calculates the logarithmic probability ratio of the bits encoded by the encoder 6702A from Figure 67, and issues the logarithmic probability relationship signal 810A .

Similarly, the logarithmic probability ratio calculator 809B takes the logarithmic probability signal uninterleaved 808B as input, calculates the logarithmic probability ratio of the bits encoded by the encoder 302B from Figure 67, and issues the relation signal Logarithmic probability 810B.

The flexible input / flexible output decoder 811A takes the logarithmic probability relation signal 810A as input, performs decoding and issues a decoded logarithmic probability relation 812A.

Similarly, the flexible input / flexible output decoder 811B takes the logarithmic probability ratio signal 810B as input, performs decoding, and issues the decoded logarithmic probability ratio 812B.

(Iterative Decoding (iterative detection), k iterations)

The interleaver (813A) takes the k-th decoded logarithmic probability ratio 812A decoded by the flexible input decoder / flexible output as input, performs interleaving, and issues the interleaved logarithmic probability ratio 814A. At this point, the interleaving pattern used by the interleaver (813A) is identical to that of the interleaver (304A) from Figure 67.

Another interleaver (813B) takes the k-th decoded logarithmic probability ratio 812B decoded by the flexible input decoder / flexible output as input, performs interleaving, and issues the interleaved logarithmic probability ratio 814B. At this point, the interleaving pattern used by the interleaver (813B) is identical to that of the other interleaver (304B) from Figure 67.

The internal MIMO detector 803 takes a baseband signal 816X, the transformed channel estimation signal group 817X, the baseband signal signal 816Y, the transformed channel estimation signal group 817Y, the interleaved logarithmic probability ratio 814A and the interrelated logarithmic probability ratio 814B as input. At this point, the baseband signal 816X, the transformed channel estimate signal group 817X, the baseband signal signal 816Y, and the transformed channel estimate signal group 817Y are used instead of the baseband signal 801X, the 802X channel estimation signal group, the 801Y baseband signal group and the 802Y channel estimation signal group since the latter produces delays due to iterative decoding.

The iterative decoding operations of the internal MIMO detector 803 differ from the initial detection operations thereof in that the logarithmic probability interleaved ratios 814A and 814B are used in signal processing for the latter. The internal MIMO detector 803 first calculates E (b0, b1, b2, b3, b4, b5, b6, b7) in the same way as for the initial detection. In addition, the coefficients corresponding to Calculus 11 (formula 11) and Calculus 32 (formula 32) are calculated from the interleaved logarithmic probability ratios 814A and 914B. The value of E (b0, b1, b2, b3, b4, b5, b6, b7) is corrected using the coefficients as! calculated to obtain E '(b0, b1, b2, b3, b4, b5, b6, b7), which is emitted as signal 804.

Logarithmic probability calculator 805A takes the signal 804 as input, calculates the logarithmic probability of bits b0, b1, b2 and b3, and issues the logarithmic probability signal 806A. Note that this logarithmic probability calculation produces the log probability of a bit being 1 and the logarithmic probability that a bit is 0. The calculation method is as shown in Calculation 31 (formula 31) to Calculation 35 (formula 35 ), and the details are provided by the non-patent Bibliography 2 and 3.

Similarly, the logarithmic probability calculator 805B takes the signal 804 as input, calculates the logarithmic probability of bits b4, b5, b6 and b7, and issues the logarithmic probability signal 806B. The operations performed by the forward deinterleaver are similar to those performed for the initial detection.

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Although Figure 8 illustrates the configuration of the signal processor when performing iterative detection, this structure is not absolutely necessary since good reception improvements are obtainable by iterative detection alone. Whenever the components necessary for iterative detection are present, the configuration does not need to include interleavers 813A and 813B. In such a case, the internal MIMO detector 803 does not perform iterative detection.

As shown in Non-patent Bibliography 5 and the like, QR decomposition can also be used to perform initial detection and iterative detection. Also, as indicated by the non-patent Bibliography 11, the linear operations of MmSE and ZF can be performed when the initial detection is performed.

Figure 9 illustrates the configuration of a signal processor unlike that of Figure 8, which serves as the signal processor for the modulated signals transmitted by the transmission device from Figure 4 as used in Figure 67. The point of difference from Figure 8 is the number of flexible input / flexible output decoders. A flexible input / flexible output decoder 901 takes the logarithmic probability ratio signals 810A and 810B as input, performs decoding, and issues a decoded logarithmic probability ratio 902. A distributor 903 takes the decoded logarithmic probability ratio 902 as input for distribution. Otherwise, the operations are identical to those explained for Figure 8.

As described above, when a transmission device according to the present embodiment using a MIMO system transmits a plurality of the modulated signals from a plurality of antennas, change the phase over time while multiplying by the matrix of Precoding to regularly change the phase results in improvements to the quality of data reception for a receiving device in a LOS environment, where direct waves are dominant, compared to a conventional MIMO system of spatial multiplexing.

In the present embodiment, and particularly in the configuration of the receiving device, the number of antennas is limited and explanations are provided accordingly. However, the embodiment can also be applied to a larger number of antennas. In other words, the number of antennas in the receiving device does not affect the operations or advantageous effects of the present embodiment.

In addition, in the present embodiments, the coding is not particularly limited to LDPC codes. Similarly, the decoding method is not limited to implementation through a flexible input / flexible output decoder that uses sum-product decoding. The decoding method used by the flexible input decoder / flexible output can also be, for example, the BCJR, SOVA algorithm and the Max-Log-Map algorithm. Details are provided in the non-patent Bibliography 6.

Furthermore, although the present embodiment is described using a single carrier method, no limitation is intended in this regard. The present embodiment is also applicable to multi-carrier transmission. Accordingly, the present embodiment can also be performed using, for example, spread spectrum communications, OFDM, SC-FDMA, SC-OFDM, particle OFDM as described in Non-patent Bibliography 7, and so on! successively. Additionally, in the present embodiment, the symbols other than the data symbols, such as pilot symbols (preamble, unique word, and so on) or the symbols that transmit control information, can be arranged in the frame in any way.

The following describes an example in which OFDM is used as a multi-carrier method.

Figure 70 illustrates the configuration of a transmission device using OFDM. In Figure 70, the components operating in the same manner described for Figures 3, 12 and 67 use identical reference numbers.

A processor related to OFDM 1201A takes a weighted signal 309A as input, performs processing related to OFDM therein, and issues transmission signal 1202A. Similarly, the processor related to OFDM 1201B takes the post-phase change signal 309B as input, performs processing related to OFDM therein, and issues the transmission signal 1202B.

Figure 13 illustrates a sample configuration of the OFDM-related processors 1201A and 1201B and forward from Figure 70. Components 1301A to 1310A are between 1201A and 312A from Figure 70, while components 1301B to 1310B They are between 1201B and 312B.

The serial to parallel converter 1302A performs serial to parallel conversion in the changed baseband signal 1301A (corresponding to the changed baseband signal 6701A from Figure 70) and issues the parallel signal 1303A.

The reorganizer 1304A takes the parallel signal 1303A as input, performs rearrangement thereof, and issues the rearranged signal 1305A. The reordering is described in detail later.

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The IFFT unit 1306A takes the reordered signal 1305A as input, applies an IFFT to it, and issues the post-IFFT signal 1307A.

The wireless unit 1308A takes the post-IFFT signal 1307A as input, performs processing such as frequency conversion and amplification, and emits the modulated signal 1309A. The modulated signal 1309A is then emitted as radio waves by the antenna 1310A.

The serial to parallel converter 1302B performs serial to parallel conversion in the post-phase change 1301B (corresponding to the post-phase change 309B from Figure 12) and emits the parallel signal 1303B.

The reorganizer 1304B takes the parallel signal 1303B as input, performs rearrangement thereof, and issues the rearranged signal 1305B. The reordering is described in detail later.

The IFFT unit 1306B takes the reordered signal 1305B as input, applies an IFFT to it, and issues the post-IFFT signal 1307B.

The wireless unit 1308B takes the post-IFFT signal 1307B as input, performs processing such as frequency conversion and amplification therein, and outputs modulated signal 1309B. The modulated signal 1309B is then emitted as radio waves by the antenna 1310A.

The transmission device from Figure 67 does not use a multi-carrier transmission method. Therefore, as shown in Figure 69, a phase change is made to achieve a period (cycle) of four and the post-phase change symbols are arranged in the time domain. As shown in Figure 70, when multi-carrier transmission is used, such as OFDM, then obviously the symbols on the precoded baseband signals that have undergone change and phase change can be arranged in the time domain as in Figure 67, and this can be applied to each (sub-) carrier. However, for multi-carrier transmission, the arrangement can also be in the frequency domain, or both in the frequency domain and in the time domain. The following describes these provisions.

Figures 14A and 14B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13. The frequency axes they are composed of (sub-) carriers 0 to 9. The modulated signals z1 and z2 share common time indications (timing) and use a common frequency band. Figure 14A illustrates a rearrangement method for the modulated signal z1 symbols, while Figure 14B illustrates a rearrangement method for the z2 modulated signal symbols. With respect to the symbols of the changed baseband signal 1301A introduced to the serial to parallel converter 1302A, the arrangement is # 0, # 1, # 2, # 3, and so! successively. At this point, given that the example refers to a period (cycle) of four, No. 0, No. 1, No. 2 and No. 3 are equivalent to a period (cycle). Similarly, # 4n, # 4n + 1, # 4n + 2, and # 4n + 3 (where n is a positive integer other than zero) are also equivalent to a period (cycle).

As shown in Figure 14A, the symbols # 0, # 1, # 2, # 3, and so! successively they are arranged in order, starting at carrier 0. The symbols # 0 to # 9 are given the time indication $ 1, followed by the symbols # 10 to # 19 which are given the time indication # 2, and so on! successively in a regular disposition. At this point, the modulated signals z1 and z2 are complex signals.

Similarly, with respect to the symbols of the weighted signal 1301B introduced to the serial to parallel converter 1302B, the assigned order is # 0, # 1, # 2, # 3, and so on! successively. At this point, since the example refers to a period (cycle) of four, a different phase change is applied to each of no. 0, no. 1, no. 2 and no. 3, which are equivalent to a period (cycle). Similarly, a different phase change is applied to each of # 4n, # 4n + 1, # 4n + 2, and # 4n + 3 (where n is a positive integer other than zero ), which are also equivalent to a period (cycle).

As shown in Figure 14B, the symbols # 0, # 1, # 2, # 3, and so! successively they are arranged in order, starting on carrier 0. The symbols # 0 to # 9 are given the time indication $ 1, followed by the symbols # 10 to # 19 provided with the $ 2 time indication, and so on! successively in a regular disposition.

The group of symbols 1402 shown in Figure 14B corresponds to a period (cycle) of symbols when the phase change method of Figure 69 is used. Symbol # 0 is the symbol obtained using the phase in the indication of time u in Figure 69, the symbol # 1 is the symbol obtained using the phase in the time indication u + 1 in Figure 69, the symbol # 2 is the symbol obtained using the phase in the indication of time u + 2 in Figure 69, and the symbol # 3 is the symbol obtained using the phase in the time indication u + 3 in Figure 69. Therefore, for any symbol # x, the symbol n . ° x is the symbol obtained using the phase in the time indication u in Figure 69 when x mod 4 equals 0 (that is, when the remainder of x divided by 4 is 0, the mod operator being mod), the Symbol No. x is the symbol obtained using the phase in the time indication x + 1 in Figure 69 when x mod 4 equals 1, the symbol No. x is the symbol or obtained using the phase in the indication of

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time x + 2 in Figure 69 when x mod 4 equals 2, and the symbol # x is the symbol obtained using the phase in the time indication x + 3 in Figure 69 when x mod 4 equals 3.

In the present embodiment, the modulated signal z1 shown in Figure 14A has not undergone a phase change.

As such, when a multi-carrier transmission method such as OFDM is used, and unlike single carrier transmission, the symbols can be arranged in the frequency domain. Of course, the method of arrangement of symbols is not limited to those illustrated by Figures 14A and 14B. Additional examples are shown in Figures 15A, 15B, 16A and 16B.

Figures 15A and 15B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a symbol rearrangement scheme used by reorganizers 1301A and 1301B from Figure 13 that differs from of Figures 14A and 14B. Figure 15A illustrates a rearrangement scheme for the symbols of the modulated signal z1, while Figure 15B illustrates a rearrangement scheme for the symbols of the modulated signal z2. Figures 15A and 15B differ from Figures 14A and 14B in that different rearrangement methods are applied to the symbols of the modulated signal z1 and to the symbols of the modulated signal z2. In Figure 15B, the symbols No. 0 to No. 5 are arranged in carriers 4 to 9, the symbols No. 6 to No. 9 are arranged in carriers 0 to 3, and this arrangement is repeated for symbols # 10 to # 19. At this point, as in Figure 14B, the group of symbols 1502 shown in Figure 15B corresponds to a period (cycle) of symbols when using the method of changing the phase of Figure 6.

Figures 16A and 16B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from that of Figures 14A and 14B. Figure 16A illustrates a rearrangement method for the modulated signal z1 symbols, while Figure 16B illustrates a rearrangement method for the z2 modulated signal symbols. Figures 16A and 16B differ from Figures 14A and 14B in that, while Figures 14A and 14B show the symbols arranged in sequential carriers, Figures 16A and 16B do not arrange the symbols in sequential carriers. Obviously, for Figures 16A and 16B, different rearrangement methods can be applied to the symbols of the modulated signal z1 and to the symbols of the modulated signal z2 as in Figures 15A and 15B.

Figures 17A and 17B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from those of Figures 14A to 16B. Figure 17A illustrates a reordering method for the modulated signal z1 symbols and Figure 17B illustrates a reordering method for the z2 modulated signal symbols. Although Figures 14A to 16B show the symbols arranged with respect to the frequency axis, Figures 17A and 17B use the frequency and time axes together in a single arrangement.

Although Figure 69 describes an example where the phase change takes place over a period (cycle) of four intervals, the following example describes a period (cycle) of eight intervals. In Figures 17A and 17B, the group of symbols 1702 is equivalent to a period (cycle) of symbols when the phase change scheme (ie, for eight symbols) is used so that the symbol # 0 is the symbol obtained using the phase in the time indication u, the symbol # 1 is the symbol obtained using the phase in the time indication u + 1, the symbol # 2 is the symbol obtained using the phase in the indication of time u + 2, the symbol # 3 is the symbol obtained using the phase in the time indication u + 3, the symbol # 4 is the symbol obtained using the phase in the time indication u + 4, Symbol No. 5 is the symbol obtained using the phase in the time indication u + 5, the symbol No. 6 is the symbol obtained using the phase in the time indication u + 6, and the symbol No. 7 is the symbol obtained using the phase in the time indication u + 7. Therefore, for any symbol # x, the symbol # x is the symbol obtained using the phase in the time indication u when x mod 8 equals 0, the symbol # x is the symbol obtained using the phase in the time indication u + 1 when x mod 8 equals 1, the symbol # x is the

Symbol obtained using the phase in the time indication u + 2 when x mod 8 equals 2, the symbol No. x is the

Symbol obtained using the phase in the time indication u + 3 when x mod 8 equals 3, the symbol No. x is the

Symbol obtained using the phase in the time indication u + 4 when x mod 8 equals 4, the symbol No. x is the

Symbol obtained using the phase in the time indication u + 5 when x mod 8 equals 5, the symbol No. x is the

symbol obtained using the phase in the time indication u + 6 when x mod 8 equals 6, and the symbol # x is the symbol obtained using the phase in the time indication u + 7 when x mod 8 equals 7 In Figures 17A and 17B four intervals are used along the time axis and two intervals along the frequency axis for a total of 4x2 = 8 intervals, in which a period (cycle) of symbols is arranged. At this point, given mxn symbols per period (cycle) (ie, different phases are available for multiplication), then n intervals (carriers) in the frequency domain and m intervals in the time domain should be used to arrange the symbols of each period (cycle), so that m> n. This is because the phase of the direct waves fluctuates little by little in the time domain relative to the frequency domain. Accordingly, the present embodiment performs a regular phase change that reduces the effect of sustained direct waves. Therefore, the period (cycle) of phase change should preferably reduce direct wave fluctuations. Therefore, m should be greater than n. Given the above, use the time and frequency domains together to

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rearrangement, as shown in Figures 17A and 17B, is preferable to using any of the frequency domain

0 of the time domain alone due to the strong probability that direct waves become regular. As a result, the effects of the present invention are more easily obtained. However, rearranging in the frequency domain can lead to diversity gain due to the fact that the frequency domain fluctuations are abrupt. As such, using the frequency and time domains together for reordering is not always ideal.

Figures 18A and 18B indicate frequency in the horizontal axes and time in the vertical axes thereof, and illustrate an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from that of Figures 17A and 17B. Figure 18A illustrates a rearrangement method for the modulated signal z1 symbols, while Figure 18B illustrates a rearrangement method for the z2 modulated signal symbols. Like Figures 17A and 17B, Figures 18A and 18B illustrate the use of time and frequency domains together. However, in contrast to Figures 17A and 17B, where the frequency domain is prioritized and the time domain is used for the secondary symbol arrangement, Figures 18A and 18B prioritize the time domain and use the frequency domain to the provision of secondary symbol. In Figure 18B, the group of symbols 1802 corresponds to a period (cycle) of symbols when the phase change method is used.

In Figures 17A, 17B, 18A and 18B, the rearrangement method applied to the symbols of the modulated signal z1 and the symbols of the modulated signal z2 can be identical or can be differentiated as in Figures 15A and 15B. Any approach allows good reception quality to be obtained. Also, in Figures 17A, 17B, 18A and 18B, the symbols may be arranged non-sequentially as in Figures 16A and 16B. Any approach allows good reception quality to be obtained.

Figure 22 indicates frequency on the horizontal axis and time on the vertical axis thereof, and illustrates an example of a method of reordering symbols used by reorganizers 1301A and 1301B from Figure 13 which differs from the previous one. Figure 22 illustrates a method of regular phase change using four intervals, similar to the indications of time ua or + 3 from Figure 69. The characteristic feature of Figure 22 is that, although the symbols are rearranged with respect to the frequency domain, when read along the time axis, a periodic displacement of n (n = 1 in the example in Figure 22) is evident. The group of frequency domain symbols 2210 in Figure 22 indicates four symbols to which the phase changes in the time indication u to u + 3 from Figure 69 apply.

At this point, the symbol No. 0 is obtained through a phase change in the time indication u, the symbol No.

1 is obtained through a phase change in the time indication u + 1, the symbol # 2 is obtained through a phase change in the time indication u + 2, and the symbol # 3 it is obtained through a phase change in the time indication u + 3.

Similarly, for the frequency domain symbol group 2220, the symbol # 4 is obtained through a phase change in the time indication u, the symbol # 5 is obtained through a change phase in the time indication u + 1, the symbol # 6 is obtained through a phase change in the time indication u + 2, and the symbol # 7 is obtained through a change of phase in the time indication u + 3.

The phase change described above applies to the symbol in the time indication $ 1. However, to apply periodic displacement with respect to the time domain, the following phase change is applied to the groups of symbols 2201, 2202, 2203 and 2204.

For the time domain symbol group 2201, the # 0 symbol is obtained through a phase change in the time indication u, the # 9 symbol is obtained through a phase change in the time indication u + 1, the symbol # 18 is obtained through a phase change in the time indication u + 2, and the symbol # 27 is obtained through a phase change in the indication of time u + 3.

For the time domain symbol group 2202, the symbol # 28 is obtained through a phase change in the time indication u, the symbol # 1 is obtained through a phase change in the time indication u + 1, the symbol # 10 is obtained through a phase change in the time indication u + 2, and the symbol # 19 is obtained through a phase change in the indication of time u + 3.

For the time domain symbol group 2203, the # 20 symbol is obtained through a phase change in the time indication u, the # 29 symbol is obtained through a phase change in the time indication u + 1, the symbol # 2 is obtained through a phase change in the time indication u + 2, and the symbol # 11 is obtained through a phase change in the indication of time u + 3.

For the time domain symbol group 2204, the # 12 symbol is obtained through a phase change in the time indication u, the # 21 symbol is obtained through a phase change in the time indication u + 1, the symbol # 30 is obtained through a phase change in the time indication u + 2, and the symbol # 3 is obtained through a phase change in the indication of time u + 3.

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The characteristic feature of Figure 22 can be seen in that, taking the symbol No. 11 as an example, the two neighboring symbols thereof having the same time indication in the frequency domain (No. 10 and No. 12) are both symbols changed using a different phase than that of symbol # 11, and the two neighboring symbols of the same having the same carrier in the time domain (# 2 and # 20) are both symbols changed using a different phase than that of symbol No. 11. This is maintained not only for symbol No. 11, but also for any symbol that has two symbols that are neighbors in the frequency domain and in the time domain . Therefore, the phase change is carried out effectively. This is highly likely to improve the quality of data reception since the influence of regularizing direct waves is less prone to reception.

Although Figure 22 illustrates an example in which n = 1, the invention is not limited in this way. The same can be applied to a case in which n = 3. Additionally, although Figure 22 illustrates the realization of the previously described effects by arranging the symbols in the frequency domain and advancing in the time domain to achieve the characteristic effect of imparting a periodic displacement to the order of disposition of the symbol, the symbols can also be arranged randomly (or regularly) for the same effect.

Although the present embodiment describes a variation of embodiment 1 in which a baseband signal changer is inserted before the phase change, the present embodiment can also be performed as a combination with embodiment 2, so that the signal changer Baseband is inserted before the phase change in Figures 26 and 28. Accordingly, in Figure 26, the phase changer 317A takes the changed baseband signal 6701A (q1 (i)) as input, and the changer Phase 317B takes the changed baseband signal 6701B (q2 (i)) as input. The same applies to phase changers 317A and 317B from Figure 28.

The following describes a method for allowing the receiving device to obtain good quality of the received signal for data, regardless of the arrangement of the receiving device, considering the location of the receiving device with respect to the transmission device.

Figure 31 illustrates an example of frame configuration for a portion of the symbols in a signal in the time-frequency domains, given a transmission method where a regular phase change is performed for a multi-carrier method such as OFDM.

Figure 31 illustrates the frame configuration of the modulated signal z2 'corresponding to the changed baseband signal introduced to the phase changer 317B from Figure 67. Each square represents a symbol (although both signals s1 and s2 are included for precoding purposes, depending on the precoding matrix, only one of the signals s1 and s2 can be used).

Consider symbol 3100 on carrier 2 and the time indication $ 2 of Figure 31. The carrier described at this point may be referred to as an alternative subcarrier.

In carrier 2, there is a very strong correlation between the channel conditions for the symbol 3100A in the carrier 2, time indication $ 2 and the channel conditions for the nearest neighboring symbols in the time domain to the time indication $ 2 , that is, symbol 3013 in time indication $ 1 and symbol 3101 in time indication $ 3 on carrier 2.

Similarly, for the time indication $ 2, there is a very strong correlation between the channel conditions for the symbol 3100 on the carrier 2, time indication $ 2 and the channel conditions for the nearest neighboring symbols in the frequency domain to carrier 2, that is, symbol 3104 in carrier 1, time indication $ 2 and symbol 3104 in time indication $ 2, carrier 3.

As described above, there is a very strong correlation between the channel conditions for the symbol 3100 and the channel conditions for each of the symbols 3101, 3102, 3103 and 3104.

The present description considers N different phases (where n is an integer, N> 2) for multiplication in a transmission method where the phase is changed regularly. The symbols illustrated in Figure 31 are indicated as ej0, for example. This means that this symbol is the signal z2 'from Figure 6 that has undergone a change in phase through multiplication by ej0. That is, the values given for the symbols in Figure 31 are the value of y (t) as provided by Calculation 70 (formula 70).

The present embodiment takes advantage of the high correlation in the existing channel conditions between symbols that are neighbors in the frequency domain and / or symbols that are neighbors in the time domain in a symbol arrangement that allows high quality reception of data through the receiving device that receives the post-phase change symbols.

To achieve this high quality of data reception, conditions No. D1-1 and No. D1-2 must be met.

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(Condition # D1-1)

As shown in Figure 69, for a transmission method that involves a regular phase change performed on the changed baseband signal q2 using a multi-carrier method such as OFDM, time X, the carrier

And it must be a symbol to transmit data (hereinafter, data symbol), symbols that are neighbors in the time domain, that is, in time X-1, the carrier Y and in time X + 1, the Carrier Y must also be data symbols, and a different phase change must be made in the changed baseband signal q2 corresponding to each of these three data symbols, that is, in the changed baseband signal q2 in the time X, carrier Y, at time X-1, carrier Y and at time X + 1, carrier Y.

(Condition # D1-2)

As shown in Figure 69, for a transmission method that involves a regular phase change performed on the changed baseband signal q2 using a multi-carrier method such as OFDM, time X, the carrier

And it must be a symbol to transmit data (hereinafter, data symbol), symbols that are neighbors in the time domain, that is, at time X, carrier Y + 1 and at time X, carrier Y -1 must also be data symbols, and a different phase change must be made in the changed baseband signal q2 corresponding to each of these three data symbols, that is, in the changed baseband signal q2 in the time X, carrier Y, at time X, carrier Y-1 and at time X, carrier Y + 1.

Ideally, a data symbol should satisfy Condition # D1-1. Similarly, data symbols should satisfy Condition # D1-2.

The reasons for supporting Conditions No. D1-1 and No. D1-2 are as follows.

There is a very strong correlation between the channel conditions of the given symbol of a transmission signal (hereinafter, the symbol A) and the channel conditions of the symbols that are neighboring the symbol A in the time domain, as has been previously described.

Therefore, when three symbols that are neighbors in the time domain each have different phases, then despite the degradation in reception quality in the LOS environment (poor signal quality produced by degradation in conditions due to phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols that are neighbors to symbol A are highly likely to provide good reception quality. As a result, good signal quality received after error correction and decoding is attainable.

Similarly, there is a very strong correlation between the channel conditions of the given symbol of a transmission signal (symbol A) and the channel conditions of the symbols that are neighboring the symbol A in the frequency domain, as described previously.

Therefore, when three symbols that are neighbors in the frequency domain each have different phases, then despite the degradation in reception quality in the LOS environment (poor signal quality caused by degradation in conditions due to Direct wave phase ratios despite high signal quality in terms of SNR) for symbol A, the remaining two symbols that are neighbors to symbol A are highly likely to provide good reception quality. As a result, good signal quality received after error correction and decoding is attainable.

By combining Conditions No. D1-1 and No. D1-2, it is likely that even higher data reception quality for the receiving device is achievable. Therefore, the following Condition No. D1-3 can be deduced.

(Condition # D1-3)

As shown in Figure 69, for a transmission method that involves a regular phase change performed on the changed baseband signal q2 that uses a multi-carrier method such as OFDM, time X, the carrier Y must be a symbol to transmit data (data symbol), symbols that are neighbors in the time domain, that is, at time X-1, carrier Y and at time X + 1, carrier Y must also be data symbols, and symbols that are neighbors in the frequency domain, that is, at time X, carrier Y-1 and at time X, carrier Y + 1 must also be data symbols, so that a change of different phase in the changed baseband signal q2 corresponding to each of these five data symbols, that is, in the changed baseband signal q2 at time X, carrier Y, at time X, carrier Y- 1, at time X, carrier Y + 1, at time X-1, carrier Y and at time X + 1, carrier Y.

At this point, the different changes in phase are as follows. Phase changes are defined from 0 radians to 2p radians. For example, for time X, carrier Y, a phase change of e ^ XY is applied to the precoded baseband signal q2 from Figure 69, for time X-1, carrier Y, a change is applied e / BX-1Y phase

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to the precoded baseband signal q2 from Figure 69, for time X + 1, carrier Y, a phase change of eyBX + 1Y is applied to the precoded baseband signal q2 from Figure 69, so that 0 <0x, and <2p, 0 <0x-1, and <2p, and 0 <0x + 1, and <2p, .. all units being in radians. Therefore, for Condition No. D1-1, it follows that 0x, and t 0x, y-1, 0x, and t 0x, y + 1, and that 0x, y-1 t 0x, y + 1. Similarly, for Condition No. D1-2, it follows that 0x, and t 0x, y-1, 0x, and t 0x, y + 1, and that 0x, y-1 t 0x, y + 1. And, for Condition No. D1-3, it follows that 0x, yt 0x- 1, y, 0x, yt 0x + 1, y, 0x, yt 0x, y-1, 0x, yt 0x, y + 1 , 0x-1, yt 0x + 1, y, 0x-1, yt 0x, y-1, 0x-1, yt 0x, y + 1, 0x + 1, yt 0x, y-1, 0x + 1, yt 0x, y + 1, and that 0x, y-1 t 0x, y + 1.

Ideally, a data symbol should satisfy Condition # D1-3.

Figure 31 illustrates an example of Condition No. D1-3, where symbol A corresponds to symbol 3100. The symbols are arranged such that the phase by which the changed baseband signal q2 is multiplied from the Figure 69 differs for the symbol 3100, for both neighboring symbols thereof in the time domain 3101 and 3102, and for both neighboring symbols thereof in the frequency domain 3102 and 3104. Consequently, despite the quality degradation Signal received from the 3100 symbol for the receiver, good signal quality is highly likely for neighboring signals, thus guaranteeing good signal quality after error correction.

Figure 32 illustrates a distribution of symbols obtained through phase changes under these conditions.

As is evident from Figure 32, with respect to any data symbol, a different phase change is applied to each symbol that is neighboring in the time domain and in the frequency domain. As such, the ability of the reception device to correct errors can be improved.

In other words, in Figure 32, when all symbols that are neighbors in the time domain are data symbols, Condition No. D1-1 is satisfied for all X and all Y.

Similarly, in Figure 32, when all symbols that are neighbors in the frequency domain are data symbols, Condition No. D1-2 is satisfied for all X and all Y.

Similarly, in Figure 32, when all symbols that are neighbors in the frequency domain are data symbols and all symbols that are neighbors in the time domain are data symbols, Condition # D1-3 is satisfies for all X and all Y.

The following analyzes the example described above for a case where the phase change takes place in two changed baseband signals q1 and q2 (see Figure 68).

Several methods of phase change are applicable to perform a phase change in two changed baseband signals q1 and q2. Its details are explained below.

Method 1 involves a phase change of the changed baseband signal q2 as described above, to achieve the phase change illustrated by Figure 32. In Figure 32, a phase change having a period (cycle) out of ten applies to the changed baseband signal q2. However, as described above, to satisfy Conditions No. D1-1, No. D1-2, and No. D1-3, the phase change applied to the changed baseband signal q2 in each (sub-) carrier changes over time. (Although such changes apply in the

Figure 32 with a period (cycle) of ten, other methods of phase change are also applicable). Then,

as shown in Figure 33, the degree of phase change made in the changed baseband signal q2 produces a constant value that is one tenth that of the phase change made in the baseband signal

changed q2. In Figure 33, for a period (cycle) (of phase change performed on the baseband signal

changed q2) which includes the time indication $ 1, the value of the phase change made in the changed baseband signal q1 is ej0. Then, for the next period (cycle) (of phase change performed on the changed baseband signal q2) that includes the time indication $ 2, the value of the degree of phase change performed on the precoded baseband signal q1 It's e ^ 9, and so! successively.

The symbols illustrated in Figure 33 are indicated as ej0, for example. This means that this symbol is the signal q1 from Figure 26 that has undergone a phase change through multiplication by ej0.

As shown in Figure 33, the phase change applied to the changed baseband signal q1 produces a constant value that is one tenth of that of the phase change made in the precoded changed baseband signal q2 so that the value Post-phase change varies with the number of each period (cycle). (As described above, in Figure 33, the value is ej0 for the first period (cycle), ep / 9 for the second period (cycle), and so on).

As described above, the change in phase made in the changed baseband signal q2 has a period (cycle) of ten, but the period (cycle) can be made effectively greater than ten taking into account the

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degree of phase change applied to the changed baseband signal q1 and to the changed baseband signal q2. Accordingly, the quality of data reception for the receiving device can be improved.

Scheme 2 implies a phase change of the changed baseband signal q2 as described above, to achieve the phase change illustrated by Figure 32. In Figure 32, a phase change having a period ( cycle) from ten to the changed baseband signal q2. However, as described above, to satisfy Conditions No. D1-1, No. D1-2 and No. D1-3, the phase change applied to the changed baseband signal q2 at each ( sub-) carrier changes over time. (Although such changes are applied in Figure 32 with a period (cycle) of ten, other phase change methods are also applicable). Then, as shown in Figure 33, the phase change made in the changed baseband signal q2 produces a constant value that is one tenth of that made in the changed baseband signal q2.

The symbols illustrated in Figure 30 are indicated as ej0, for example. This means that this symbol is the changed baseband signal q 1 that has undergone a phase change through multiplication by ej0.

As described above, the change in phase made in the changed baseband signal q2 has a period (cycle) of ten, but the period (cycle) can be made effectively greater than ten taking into account the phase changes applied to the changed baseband signal q1 and the changed baseband signal q2. Accordingly, the quality of data reception for the receiving device can be improved. An effective way to apply method 2 is to make a phase change in the changed baseband signal q1 with a period (cycle) of N and make a phase change in the precoded baseband signal q2 with a period (cycle) of M so that N and M are coprimos. As such, taking into account both changed baseband signals q1 and q2, a period (cycle) of NxM is easily achievable, effectively making the period (cycle) longer when N and M are coprime.

Although the above analyzes an example of the phase change method described above, the present invention is not limited in this way. The phase change can be made with respect to the frequency domain, the time domain or in time-frequency blocks. Improvement similar to the quality of data reception for the receiving device can be obtained in all cases.

The same also applies to frames that have a different configuration from the one described above, where pilot symbols (SP symbols) and the symbols that transmit control information are inserted between the data symbols. The details of the phase change in such circumstances are as follows.

Figures 47A and 47B illustrate the frame configuration of the modulated signals (changed baseband signals q1 and q2) z1 or z1 'and z2' in the time-frequency domain. Figure 47A illustrates the frame configuration of the modulated signal (changed baseband signal q1) z1 or z1 'while Figure 47B illustrates the frame configuration of the modulated signal (changed baseband signal q2) z2'. In Figures 47A and 47B, 4701 marks pilot symbols while 4702 marks data symbols. Data symbols 4702 are symbols in which change or change and change in phase have been made.

Figures 47A and 47B, like Figure 69, indicate the arrangement of the symbols when a phase change is applied to the changed baseband signal q2 (although no phase change is made in the changed baseband signal q1). (Although Figure 69 illustrates a change in phase with respect to the time domain, changing the time t by the carrier f in Figure 69 corresponds to a change in phase with respect to the frequency domain. In other words, replace (t) for (t, f) where t is time and f is frequency corresponds to a phase change in time-frequency blocks). Therefore, the numerical values indicated in Figures 47A and 47B for each of the symbols are the values of the changed baseband signal q2 after the change in phase. Values for the symbols of the changed baseband signal q1 (z1) are not provided from Figures 47A and 47B since no phase change is made therein.

The important point of Figures 47A and 47B is the phase change made in the data symbols of the changed baseband signal q2, that is, in the symbols that have undergone precoding or precoding and change. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no change in phase is made in the pilot symbols inserted in z2 '.

Figures 48A and 48B illustrate the frame configuration of the modulated signals (changed baseband signals q1 and q2) z1 or z1 'and z2' in the time-frequency domain. Figure 48A illustrates the frame configuration of the modulated signal (changed baseband signal q1) z1 or z1 'while Figure 48B illustrates the frame configuration of the modulated signal (changed baseband signal q2) z2'. In Figures 48A and 48B, 4701 marks pilot symbols while 4702 marks data symbols. The 4702 data symbols are the symbols in which precoding or precoding and a phase change have been performed.

Figures 48A and 48B indicate the arrangement of the symbols when a phase change is applied to the changed baseband signal q1 and the changed baseband signal q2. Therefore, the numerical values indicated in Figures 48a and 48B for each of the symbols are the baseband signal values

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changed q1 and q2 after a change in phase.

The important point of Figures 48A and 48B is that the phase change is made in the data symbols of the changed baseband signal q1, that is, in the precoded or precoded and changed symbols thereof, and in the symbols data of the changed baseband signal q2, that is, in the precoded or precoded and changed symbols thereof. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no phase change is made in the pilot symbols inserted in z1 ', nor in the pilot symbols inserted in z2'.

Figures 49A and 49B illustrate the frame configuration of the modulated signals (changed baseband signals q1 and q2) z1 or z1 'and z2' in the time-frequency domain. Figure 49A illustrates the frame configuration of the modulated signal (changed baseband signal q1) z1 or z1 'while Figure 49B illustrates the frame configuration of the modulated signal (changed baseband signal q2) z2'. In Figures 49A and 49B, 4701 marks pilot symbols, 4702 marks data symbols, and 4901 marks null symbols for which the phase component of the baseband signal I = 0 and the quadrature component Q = 0. As such , the 4702 data symbols are the symbols in which precoding or precoding and a phase change have been performed. Figures 49A and 49B differ from Figures 47A and 47B in the configuration scheme for symbols other than data symbols. The times and carriers in which the pilot symbols are inserted in the modulated signal z1 'are null symbols in the modulated signal z2'. Conversely, the times and carriers in which the pilot symbols are inserted in the modulated signal z2 'are null symbols in the modulated signal z1'.

Figures 49A and 49B, like Figure 69, indicate the arrangement of the symbols when a phase change is applied to the changed baseband signal q2 (although no phase change is made in the changed baseband signal q1). (Although Figure 69 illustrates a change in phase with respect to the time domain, changing the time t by the carrier f in Figure 6 corresponds to a change in phase with respect to the frequency domain. In other words, replace (t) for (t, f) where t is time and f is frequency corresponds to a phase change in time-frequency blocks). Therefore, the numerical values indicated in Figures 49A and 49B for each of the symbols are the values of the changed baseband signal q2 after the change in phase. Values for the symbols of the changed baseband signal q1 are not provided from Figures 49A and 49B since no phase change is made therein.

The important point of Figures 49A and 49B is the phase change made in the data symbols of the changed baseband signal q2, that is, in the symbols that have undergone precoding or precoding and change. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no change in phase is made in the pilot symbols inserted in z2 '.

Figures 50A and 50B illustrate the frame configuration of the modulated signals (changed baseband signals q1 and q2) z1 or z1 'and z2' in the time-frequency domain. Figure 50A illustrates the frame configuration of the modulated signal (changed baseband signal q1) z1 or z1 'while Figure 50B illustrates the frame configuration of the modulated signal (changed baseband signal q2) z2'. In Figures 50A and 50B, 4701 marks pilot symbols, 4702 marks data symbols, and 4901 marks null symbols for which the phase component of the baseband signal I = 0 and the quadrature component Q = 0. As such , the 4702 data symbols are the symbols in which precoding or precoding and a phase change have been performed. Figures 50A and 50B differ from Figures 48A and 48B in the configuration scheme for symbols other than data symbols. The times and carriers in which the pilot symbols are inserted in the modulated signal z1 'are null symbols in the modulated signal z2'. Conversely, the times and carriers in which the pilot symbols are inserted in the modulated signal z2 'are null symbols in the modulated signal z1'.

Figures 50A and 50B indicate the arrangement of the symbols when a phase change is applied to the changed baseband signal q1 and the changed baseband signal q2. Therefore, the numerical values indicated in Figures 50a and 50B for each of the symbols are the values of changed baseband signals q1 and q2 after a change in phase.

The important point of Figures 50A and 50B is that a phase change is made in the data symbols of the changed baseband signal q1, that is, in the precoded or precoded and changed symbols thereof, and in the symbols data of the changed baseband signal q2, that is, in the precoded or precoded and changed symbols thereof. (The symbols under analysis, which are precoded, actually include both the s1 and s2 symbols). Therefore, no phase change is made in the pilot symbols inserted in z1 ', nor in the pilot symbols inserted in z2'.

Figure 51 illustrates a sample configuration of a transmission device that generates and transmits the modulated signal having the frame configuration of Figures 47A, 47B, 49A and 49B. Its components that perform the same operations as those in Figure 4 use the same reference symbols between them. Figure 51 does not include a baseband signal changer as illustrated in Figures 67 and 70. However, Figure 51 may also include a baseband signal changer between the weighting unit and the phase changer, when same as Figures 67 and 70.

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In Figure 51, the weighting units 308A and 308B, the phase changer 317B, and baseband signal changer only operate at the times indicated by the frame configuration signal 313 as corresponds to data symbols.

In Figure 51, a pilot symbol generator 5101 (which also generates null symbols) issues the baseband signals 5102A and 5102B for a pilot symbol each time the frame configuration signal 313 indicates a pilot symbol (and a null symbol ).

Although not indicated in the frame configurations from Figures 47A to 50B, when no precoding (and phase rotation) is performed, such as when a modulated signal is transmitted using only one antenna (so that the other antenna does not transmits signal) or when a space-time coding transmission method (particularly, space-time block coding) is used to transmit control information symbols, then frame configuration signal 313 takes information information symbols control 5104 and control information 5103 as input. When frame configuration signal 313 indicates a control information symbol, baseband signals 5102A and 5102B thereof are output.

The wireless units 310A and 310B of Figure 51 take a plurality of baseband signals as input and select a desired baseband signal according to the frame configuration signal 313. The wireless units 310A and 310B then apply processing of OFDM signal and emit modulated signals 311A and 311B according to the frame configuration.

Figure 52 illustrates a sample configuration of a transmission device that generates and transmits the modulated signal having the frame configuration of Figures 48A, 48B, 50A and 50B. The components thereof that perform the same operations as those in Figures 4 and 51 use the same reference symbols between them. Figure 52 shows an additional phase changer 317A that only operates when frame configuration signal 313 indicates a data symbol. In all other cases, the operations are identical to those explained for Figure 51. Figure 52 does not include a baseband signal changer as illustrated in Figures 67 and 70. However, Figure 52 may also include a baseband signal changer between the weighting unit and the phase changer, as well as Figures 67 and 70.

Figure 53 illustrates a sample configuration of a transmission device that differs from that of Figure

51. Figure 53 does not include a baseband signal changer as illustrated in Figures 67 and 70. However, Figure 53 may also include a baseband signal changer between the weighting unit and the phase changer , like Figures 67 and 70. The following describes the points of difference. As shown in Figure 53, phase changer 317B takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 317B makes the phase change in the precoded baseband signal 316B. When the frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, the phase changer 317B pauses the phase change operations so that the baseband signal symbols are emitted as is, just as it is. (This can be interpreted as performing forced rotation corresponding to ej0).

A selector 5301 takes the plurality of the baseband signals as input and selects a baseband signal having a symbol indicated by the frame configuration signal 313 to be broadcast.

Figure 54 illustrates a sample configuration of a transmission device that differs from that of Figure

52. Figure 54 does not include a baseband signal changer as illustrated in Figures 67 and 70. However, Figure 54 may also include a baseband signal changer between the weighting unit and the phase changer , like Figures 67 and 70. The following describes the points of difference. As shown in Figure 54, phase changer 317B takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 317B makes the phase change in the precoded baseband signal 316B. When the frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, the phase changer 317B pauses the phase change operations so that the baseband signal symbols are emitted as is, just as it is. (This can be interpreted as performing forced rotation corresponding to ej0).

Similarly, as shown in Figure 54, phase changer 5201 takes a plurality of baseband signals as input. Then, when the frame configuration signal 313 indicates a data symbol, the phase changer 5201 makes the phase change in the precoded baseband signal 309A. When the frame configuration signal 313 indicates a pilot symbol (or null symbol) or a control information symbol, the phase changer 5201 pauses the phase change operations so that the baseband signal symbols are emitted as is, just as it is. (This can be interpreted as performing forced rotation corresponding to ej0).

The above explanations are provided using pilot symbols, control symbols and data symbols as examples. However, the present invention is not limited in this way. When symbols are transmitted using methods other than precoding, such as single antenna transmission or transmission using

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space-time block coding, the absence of phase change is important. Conversely, making the phase change in the symbols that have been precoded is the key point of the present invention.

Therefore, a characteristic feature of the present invention is that the change in phase is not made in all the symbols in the frame configuration in the time-frequency domain, but is only made in the baseband signals that have been precoded and have undergone change.

The following describes a scheme to regularly change the phase when coding is performed using block codes as described in Non-patent Bibliography 12 to 15, such as LDPC Codes (not only QC-LDPC but also LDPC codes can be used) QC, concatenated LDPC and BCH codes, Turbo codes or Turbo Duo-binary codes that use tail bits, and so on! successively. The following example considers a case where two flows s1 and s2 are transmitted. When coding has been carried out using block codes and control information and they are not necessary, the number of bits that make up each block encoded coincides with the number of bits that make up each block code (even control information can be included and thus ! successively described below). When coding has been performed using block codes or the like and control information or the like is required (eg, CRC transmission parameters), then the number of bits that make up each encoded block is the sum of the number of bits that make up the block codes and the number of bits that make up the information.

Figure 34 illustrates the variable numbers of symbols and intervals needed in two coded blocks when using block codes. Unlike Figures 69 and 70, for example, Figure 34 illustrates the variable numbers of symbols and intervals required in each coded block when block codes are used when, for example, two streams s1 and s2 are transmitted as indicated in Figure 4, with an encoder and distributor. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 34, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

Next, since the transmission device described above transmits two streams simultaneously, 1500 of the 3000 above-mentioned symbols needed when the modulation method is QPSK are assigned to s1 and the other 1500 symbols are assigned to s2. As such, 1500 intervals are required to transmit the 1500 symbols (hereinafter intervals) for each of s1 and s2.

By the same reasoning, when the modulation method is 16-QAM, 750 intervals are necessary to transmit all the bits that make up two encoded blocks, and when the modulation method is 64-QAM, 500 intervals are necessary to transmit all the bits which make up the two coded blocks.

The following describes the relationship between the previously defined intervals and the multiplication phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change. That is, the phase changer of the transmission device described above uses five phase change values (or phase change sets) to achieve the period (cycle) of five. (As in Figure 69, five phase change values are necessary to perform a phase change that has a period (cycle) of five in the baseline signal changed only q2. Similarly, to make the change in phase in both changed baseband signals q1 and q2, two phase change values are necessary for each interval, these two phase change values are called a phase change set, therefore, at this point, to make a change phase that has a period (cycle) of five, five such sets of phase change should be prepared). The five phase change values (or phase change sets) are expressed as PHASE [0], PHASE [1], PHASE [2], PHASE [3] and PHASE [4].

For the 1500 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation method is QPSK, PHASE [0] is used in 300 intervals, PHASE [1] is used in 300 intervals, used PHASE [2] in 300 intervals, PHASE [3] is used in 300 intervals and PHASE [4] is used in 300 intervals. This is due to the fact that any deviation in phase use causes enormous influence to be exerted by the most frequently used phase, and that the reception device is dependent on such influence for the quality of data reception.

Additionally, for the 750 intervals described above necessary to transmit the 6000 bits that make up a single encoded block when the modulation scheme is 16-QAM, PHASE [0] is used in 150 intervals, PHASE [1] is used in 150 intervals, PHASE [2] is used in 150 intervals, PHASE [3] is used in 150 intervals and PHASE [4] is used in 150 intervals.

Even more, for the 500 intervals described above necessary to transmit the 6000 bits that make up a

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The only coding block when the modulation method is 64-QAM, PHASE [0] is used in 100 intervals, PHASE [1] is used in 100 intervals, PHASE [2] is used in 100 intervals, PHASE is used [3] in 100 intervals and PHASE [4] in 100 intervals is used.

As described above, a scheme for a regular phase change requires the preparation of N phase change values (or phase change sets) (where the N different phases are expressed as PHASE [0], PHASE [1] , PHASE [2] ... PHASE [N-2], PHASE [N-1]). As such, to transmit all the bits that make up a single encoded block, PHASE [0] is used in K0 intervals, PHASE [1] is used in K1 intervals, PHASE [i] is used in Ki intervals (where i = 0, 1, 2 ... N-1), and PHASE [N-1] is used in Kn-1 intervals, so that Condition No. D1-4 is met.

(Condition # D1-4)

K0 = K1 ... = Ki = ... Kn-1. That is, Ka = Kb (for Va and Vb where a, b, = 0, 1,2 ... N-1, a t b).

Then, when a communication system that supports multiple modulation methods selects such a supported method for use, Condition No. D1-4 must be met for the supported modulation method.

However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Condition No. D1-4 may not be satisfied for some method of modulation. In such a case, the following condition applies in place of Condition No. D1-4.

(Condition # D1-5)

The difference between Ka and Kb satisfies 0 or 1. That is, | Ka - Kb | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1, a t b)

Figure 35 illustrates the variable numbers of symbols and intervals required in two coded blocks when using block codes. Figure 35 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams s1 and s2 are transmitted as indicated by the transmission device from Figure 67 and the Figure 70, and the transmission device has two encoders. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 35, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

The transmission device from Figure 67 and the transmission device from Figure 70 each transmit two streams at once and have two encoders. As such, the two streams each transmit different blocks of code. Therefore, when the modulation method is QPSK, two blocks extracted from s1 and s2 are transmitted in the same interval, for example, a first coded block extracted from s1 is transmitted, then a second coded block extracted from s2 is transmitted. As such, 3000 intervals are necessary to transmit the first and second coded blocks.

By the same reasoning, when the modulation scheme is 16-QAM, 1500 intervals are necessary to transmit all the bits that make up the two encoded blocks, and when the modulation scheme is 64-QAM, 1000 intervals are necessary to transmit all the bits that make up the two coded blocks.

The following describes the relationship between the previously defined intervals and the multiplication phase, since it belongs to methods for a regular phase change.

At this point, it is assumed that five different phase change values (or phase change sets) have been prepared for use in the method for a regular phase change. That is, the phase changer of the transmission device from Figure 67 and Figure 70 uses five phase change values (or phase change sets) to achieve the period (cycle) of five. (As in Figure 69, five phase change values are necessary to perform a phase change that has a period (cycle) of five in the baseline signal changed only q2. Similarly, to make the change in phase in both changed baseband signals q1 and q2, two phase change values are necessary for each interval, these two phase change values are called a phase change set, therefore, at this point, to make a change phase that has a period (cycle) of five, five such sets of phase change should be prepared). The five phase change values (or phase change sets) are expressed as PHASE [0], PHASE [1], PHASE [2], PHASE [3] and PHASE [4].

For the 3000 intervals described above necessary to transmit the 6000x2 bits that make up the two

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coded blocks when the modulation method is QPSK, PHASE [0] is used in 600 intervals, PHASE [1] is used in 600 intervals, PHASE [2] is used in 600 intervals, PHASE [3] is used in 600 intervals and PHASE [4] is used in 600 intervals. This is due to the fact that any deviation in phase use causes enormous influence to be exerted by the most frequently used phase, and that the receiving device is dependent on such influence for the quality of data reception.

Additionally, to transmit the first encoded block, PHASE [0] is used at 600 times intervals, PHASE [1] is used at 600 times intervals, PHASE [2] is used at 600 times intervals, used PHASE [3] at intervals 600 times and PHASE [4] is used at intervals 600 times. Additionally, to transmit the second encoded block, PHASE [0] is used at 600 times intervals, PHASE [1] is used at 600 times intervals, PHASE [2] is used at 600 times intervals, used PHASE [3] at intervals 600 times and PHASE [4] is used at intervals 600 times.

Similarly, for the 1500 intervals described above necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation method is 16-QAM, PHASE [0] is used in 300

intervals, the fAsE [1] is used in 300 intervals, the PHASE [2] is used in 300 intervals, the PHASE [3] is used in 300

intervals and PHASE [4] is used in 300 intervals.

Additionally, to transmit the first encoded block, PHASE [0] is used at intervals 300 times, PHASE [1] is used at intervals 300 times, PHASE [2] is used at intervals 300 times, used PHASE [3] at intervals 300 times and PHASE [4] is used at intervals 300 times. Additionally, to transmit the second encoded block, PHASE [0] is used at 300 times intervals, PHASE [1] is used at 300 times intervals, PHASE [2] is used at 300 times intervals, used PHASE [3] at intervals 300 times and PHASE [4] is used at intervals 300 times.

Similarly, for the 1000 intervals described above necessary to transmit the 6000x2 bits that make up the two coded blocks when the modulation scheme is 64-QAM, PHASE [0] is used in 200

intervals, the fAsE [1] is used in 200 intervals, the PHASE [2] is used in 200 intervals, the PHASE [3] is used in 200

intervals and PHASE [4] is used in 200 intervals.

Additionally, to transmit the first encoded block, PHASE [0] is used at 200 times intervals, PHASE [1] is used at 200 times intervals, PHASE [2] is used at 200 times intervals, used PHASE [3] at intervals 200 times and PHASE [4] at intervals 200 times is used. Additionally, to transmit the second coded block, PHASE [0] is used at 200 times intervals, PHASE [1] is used at 200 times intervals, PHASE [2] is used at 200 times intervals, used PHASE [3] at intervals 200 times and PHASE [4] at intervals 200 times is used.

As described above, a method for a regular phase change requires the preparation of N phase change values (or phase change sets) (where the N different phases are expressed as PHASE [0], PHASE [1] , PHASE [2] ... PHASE [N-2], PHASE [N-2]). As such, to transmit all the bits that make up a single encoded block, PHASE [0] is used in K0 intervals, PHASE [1] is used in K1 intervals, PHASE [i] is used in Ki intervals (where i = 0, 1, 2 ... N-1), and PHASE [N-1] is used in Kn-1 intervals, so that Condition No. D1-6 is met.

(Condition # D1-6)

K0 = K1 ... = Ki = ... Kn-1. That is, Ka = Kb (for Va and Vb where a, b, = 0, 1, 2 ... N-1, a t b).

In addition, to transmit all the bits that make up the first encoded block, PHASE [0] K0.1 times is used, PHASE [1] K11 times is used, PHASE [i] Ki is used, 1 times (where i = 0, 1, 2 ... N-1), and PHASE [N-1] Kn-1.1 times is used, so that Condition No. D1-7 is met.

(Condition # D1-7)

Kq, 1 = K11 = ... Ki, 1 = Kn-1,1. That is, Ka, 1 = Kb, 1 (Va and Vb where a, b, = 0, 1,2 ... N-1, a t b).

Additionally, to transmit all the bits that make up the second encoded block, PHASE [0] K0.2 times is used, PHASE [1] K1.2 times is used, PHASE [i] Ki is used, 2 times ( where i = 0, 1, 2 ... N-1), and PHASE [N-1] Kn-1.2 times is used, so that Condition No. D1-8 is met.

(Condition # D1-8)

Kq, 2 = K1,2 = ... Ki, 2 = ... Kn-1, 2. That is, Ka, 2 = Kb, 2 (Va and Vb where a, b, = 0, 1, 2 ... N-1, a t b).

Then, when a communication system that supports multiple modulation methods selects a supported method of this type for use, Condition # D1-6, Condition # D1-7 and Condition # D1-8 must Comply with the supported modulation method.

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However, when multiple modulation methods are supported, each modulation method of this type normally uses symbols that transmit a different number of bits per symbol (although something may occur to use the same number), Condition No. D1-6 , Condition No. D1-7 and Condition No. D1-8 may not be satisfied for some modulation method. In such a case, the following conditions apply in place of Condition No. D1-6, Condition No. D1-7 and Condition No. D1-8.

(Condition # D1-9)

The difference between Ka and Kb satisfies 0 or 1. That is, | Ka - Kb | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1, a t b)

(Condition # D1-10)

The difference between Ka1 and Kb1 satisfies 0 or 1. That is, | Ka1 - Kb 1 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1,2 ... N-1, a t b)

(Condition # D1-11)

The difference between Ka, 2 and Kb, 2 satisfies 0 or 1. That is, | Ka, 2 - Kb, 2 | satisfies 0 or 1 (Va, Vb, where a, b = 0, 1, 2 ... N-1, a t b)

As described above, the deviation between the phases that are used to transmit the encoded blocks is eliminated by creating a relationship between the encoded block and the multiplication phase. As such, the quality of data reception for the reception device can be improved.

As described above, N phase change values (or phase change sets) are necessary to perform a phase change that has a period (cycle) of N with the method for regular phase change. As such, N phase change values (or phase change sets) are prepared PHASE [0], PHASE [1], PHASE [2] ... PHASE [N-2], and PHASE [N-1] . However, there are schemes to order the phases in the order established with respect to the frequency domain. No limitation is intended in this regard. The N phase change values (or phase change sets) PHASE [0], fAsE [1], PHASE [2] ... PHASE [N-2], and PHASE [N-1] can also change the phases of the blocks in the time domain or in the time-frequency domain to obtain a symbol arrangement. Although the above examples analyze a phase change method with a period (cycle) of N, the same effects are obtainable using N phase change values (or phase change sets) randomly. That is, the N phase change values (or phase change sets) do not always have to have regular periodicity. Provided that the conditions described above are satisfied, huge improvements in receiving quality data for the receiving device are achievable.

Additionally, given the existence of modes for spatial multiplexing MIMO methods, MIMO methods that use a fixed precoding matrix, space-time block coding methods, single stream transmission and methods that use a regular phase change , the transmission device (diffuser, base station) can select any one of these transmission methods.

As described in Non-patent Bibliography 3, MIMO methods of spatial multiplexing involve transmitting signals s1 and s2, which are mapped using a selected modulation method, on each of two different antennas. MIMO methods that use a fixed precoding matrix involve precoding only (no change in phase). In addition, the space-time block coding methods are described in the non-patent Bibliography 9, 16 and 17. The single stream transmission methods involve transmitting the signal s1, mapped with a selected modulation method, from an antenna after performing default processing.

Schemes using multi-carrier transmission such as OFDM involve a first group of carriers composed of a plurality of carriers and a second group of carriers composed of a plurality of carriers different from the first group of carriers, and so on, so that the Multi-carrier transmission is performed with a plurality of carrier groups. For each group of carriers, any of the spatial multiplexing MIMO methods, MIMO methods that use a fixed precoding matrix, space-time block coding methods, single stream transmission, and methods that use a change of time can be used. regular phase In particular, methods using a regular phase change in a group of (sub-) carrier selected to perform the above are preferably used.

Although the present description describes the present embodiment as a transmission device that applies precoding, baseband change and phase change, all of these can be combined in various ways. In particular, the phase changer analyzed for the present embodiment can be freely combined with the phase change analyzed in all other embodiments.

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[Realization D2]

The present embodiment describes a phase change initialization method for the regular phase change described throughout the present description. This initialization method is applicable to the transmission device from Figure 4 when a multi-carrier method such as OFDM is used, and to the transmission devices of Figures 67 and 70 when a single encoder and distributor is used, similar to Figure 4.

The following is also applicable to a method for regularly changing the phase when coding is performed using block codes as described in Non-Patent Bibliography 12 to 15, such as LDPC Codes (not only QC-LDPC but also codes can be used of LDPC) QC, concatenated LDPC and BCH codes, Turbo codes or Turbo Duo-binary codes that use tail bits, and so on! successively.

The following example considers a case where two flows s1 and s2 are transmitted. When coding has been carried out using block codes and control information and they are not necessary, the number of bits that make up each block encoded coincides with the number of bits that make up each block code (even control information can be included and thus ! successively described below). When coding has been performed using block codes or the like and control information or the like is required (eg, CRC transmission parameters), then the number of bits that make up each encoded block is the sum of the number of bits that make up the block codes and the number of bits that make up the information.

Figure 34 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used. Figure 34 illustrates the variable numbers of symbols and intervals required in each encoded block when block codes are used when, for example, two streams s1 and s2 are transmitted as indicated by the transmission device described above, and the transmission device It has only one encoder. (At this point, the transmission method can be any single carrier method or multi-carrier method such as OFDM).

As shown in Figure 34, when block codes are used, there are 6000 bits that make up a single encoded block. To transmit these 6000 bits, the number of symbols required depends on the modulation method, with 3000 for QPSK, 1500 for 16-QAM and 1000 for 64-QAM.

Next, since the transmission device described above transmits two streams simultaneously, 1500 of the 3000 above-mentioned symbols needed when the modulation method is QPSK are assigned to s1 and the other 1500 symbols are assigned to s2. As such, 1500 intervals are required to transmit the 1500 symbols (hereinafter intervals) for each of s1 and s2.

By the same reasoning, when the modulation scheme is 16-QAM, 750 intervals are necessary to transmit all the bits that make up each encoded block, and when the modulation scheme is 64-QAM, 500 intervals are necessary to transmit all the bits that make up each coded block.

The following describes a transmission device that transmits modulated signals that have a frame configuration illustrated by Figures 71A and 71B. Figure 71A illustrates a frame configuration for the modulated signal z1 'or z1 (transmitted by the antenna 312A) in the time and frequency domains. Similarly, Figure 71B illustrates a frame configuration for the modulated signal z2 (transmitted by the antenna 312B) in the time and frequency domains. At this point, the frequency (band) used by the modulated signal z1 'or z1 and the frequency (band) used for the modulated signal z2 are identical, which carry the modulated signals z1,' or z1 and z2 at the same time.

As shown in Figure 71A, the transmission device transmits a preamble (control symbol) during the interval A. The preamble is a symbol that transmits control information to another party. In particular, this preamble includes information on the modulation method used to transmit a first and a second coded block. The transmission device transmits the first encoded block during the interval B. The transmission device then transmits the second encoded block during the interval C.

In addition, the transmission device transmits a preamble (control symbol) during the interval D. The preamble is a symbol that transmits control information to another party. In particular, this preamble includes information on the modulation method used to transmit a third or fourth encoded block and so on! successively. The transmission device transmits the third encoded block during the interval E. The transmission device then transmits the fourth encoded block during the interval D.

Also, as shown in Figure 71B, the transmission device transmits a preamble (control symbol) during the interval A. The preamble is a symbol that transmits control information to another party. In particular, this preamble includes information on the modulation method used to transmit a first and a second coded block. The transmission device transmits the first encoded block during the interval B. The transmission device then transmits the second encoded block during the interval C.

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In addition, the transmission device transmits a preamble (control symbol) during the interval D. The preamble is a symbol that transmits control information to another party. In particular, this preamble includes information on the modulation method used to transmit a third or fourth encoded block and so! successively. The transmission device transmits the third encoded block during the interval E. The transmission device then transmits the fourth encoded block during the interval D.

Figure 72 indicates the number of intervals used when transmitting the encoded blocks from Figure 34, specifically using 16-QAM as the modulation method for the first encoded block. At this point, 750 intervals are necessary to transmit the first encoded block.

Similarly, Figure 72 also indicates the number of intervals used to transmit the second encoded block, using QPSK as the modulation method for them. At this point, 1500 intervals are necessary to transmit the second coded block.

Figure 73 indicates the intervals used when transmitting the encoded blocks from Figure 34, specifically using QPSK as the modulation method for the third encoded block. At this point, 1500 intervals are necessary to transmit the encoded block.

As explained throughout this description, the modulated signal z1, that is, the modulated signal transmitted by the antenna 312A, does not undergo a change in phase, while the modulated signal z2, that is, the modulated signal transmitted by The 312B antenna experiences a change in phase. The following phase change method is used for Figures 72 and 73.

Before the phase change can take place, seven different phase change values must be prepared. The seven phase change values are labeled # 0, # 1, # 2, # 3, # 4, # 5 and # 6. The phase change is regular and newspaper. In other words, the phase change values are applied regularly and periodically, so that the order is No. 0, No. 1, No. 2, No. 3, No. 4, No. 5, No. 6, No. 0, No. 1, No. 2, No. 3, No. 4, No. 5, No. 6, No. 0, No. 1, # 2, # 3, # 4, # 5, # 6 and so on! successively.

As shown in Figure 72, since 750 intervals are necessary for the first coded block, the phase change value # 0 is initially used, so that # 0, # 1, # 2, No. 3, No. 4, No. 5, No. 6, No. 0, No. 1, No. 2 ... No. 3, No. 4, No. # 5, # 6 are used in series, with the 750th interval using # 0 in the final position. The phase change is then applied to each interval for the second coded block. The present description involves multi-diffusion transmission and broadcast applications. As such, a reception terminal may not have to need the first encoded block and only extract the second encoded block. In such circumstances, since the final interval used for the first coded block uses the phase change value # 0, the initial phase change value used for the second coded block is # 1. As such, they are conceivable the following methods:

(a): the aforementioned terminal monitors the transmission of the first encoded block, that is, monitors the pattern of the phase change values through the final interval used to transmit the first encoded block, and then estimates the change value of phase used for the initial interval of the second coded block;

(b): (a) does not take place, and the transmission device transmits information on the phase change values in use in the initial interval of the second coded block.

Scheme (a) leads to greater energy consumption through the terminal due to the need to monitor the transmission of the first coded block. However, scheme (b) leads to reduced data transmission efficiency.

Therefore, there is a need to improve the phase change value assignment described above. Considering a method in which the phase change value used to transmit the initial interval of each encoded block is set. Therefore, as indicated in Figure 72, the phase change value used to transmit the initial interval of the second encoded block and the phase change value used to transmit the initial interval of the first encoded block are identical, where n . ° 0.

Similarly, as indicated in Figure 73, the phase change value used to transmit the initial interval of the third coded block is not # 3, but is instead identical to the phase change value used for transmit the initial interval of the first and second coded blocks, being # 0.

As such, the problems that accompany both methods (a) and (b) described above can be restricted while maintaining the effects thereof.

In the present embodiment, the method used to initialize the phase change value for each encoded block, that is, the phase change value used for the initial interval of each encoded block, is set to be # 0. However, other methods for single frame units may also be used. For example, the phase change value used for the initial interval of a symbol that transmits information after it has been

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Once the preamble or control symbol has been transmitted, it can be set to No. 0.

[Realization D3]

The above-described embodiments analyze a weighting unit that uses a precoding matrix expressed in complex numbers for precoding. However, the precoding matrix can also be expressed in real numbers.

That is, assuming that the two baseband signals s1 (i) and s2 (i) (where i is time or frequency) have been mapped (using a modulation scheme), and precoded to obtain the precoded baseband signals z1 (i) and z2 (i). As such, the mapped baseband signal s1 (i) has a phase component of Isi (i) and a quadrature component of Qs1 (i), and the mapped baseband signal s2 (i) has a phase component of Is2 (i) and a quadrature component of Qs2 (i), while the precoded baseband signal z1 (i) has a phase component of Iz1 (i) and a quadrature component of Qz1 (i), and the Precoded baseband signal z2 (i) has a phase component of Iz2 (i) and a quadrature component of Qz2 (i), which provides the following precoding matrix Hr when all values are real numbers.

[Calculation 76]

image46

The precoding matrix Hr can also be expressed as follows, where all values are real numbers. [Calculation 77]

image47

where an, a12, a13, a14, a21, a22, a23, a24, a31, a32, a33, a34, a41, a42, a43 and a44 are real numbers. However, none of the following can be maintained: {an = 0, a12 = 0, a13 = 0 and a14 = 0}, {a21 = 0, a22 = 0, a23 = 0 and a24 = 0}, {a31 = 0 , a32 = 0, a33 = 0 and a34 = 0}, and {a41 = 0, a42 = 0, a43 = 0 and a44 = 0}. Also, none of the following can be maintained: {an = 0, a21 = 0, a31 = 0 and a41 = 0}, {a12 = 0, a22 = 0, a32 = 0 and a42 = 0}, {a13 = 0, a23 = 0, a33 = 0 and a43 = 0}, and {a14 = 0, a24 = 0, a34 = 0 and a44 = 0}.

[Embodiment E1]

The present embodiment describes a transmission scheme as an application of the phase change to precoded signals (or precoded signals having changed base bands) for a broadcast system using the DVB-T2 (Digital Video Broadcast for a broadcast system) standard. of second generation digital terrestrial television). First, the configuration of a frame in a broadcast system using the DVB-T2 standard is described.

Figure 74 illustrates the global frame configuration of a signal transmitted by a diffuser using the DVB-T2 standard. Since DVB-T2 uses an OFDM method, the frame is configured in the time-frequency domain. Therefore, Figure 74 illustrates the frame configuration in the time-frequency domain. The frame includes signaling data P1 (7401), pre-signaling data L1 (7402), post-signaling data L1 (7403), a common PLP (Physical Layer Pipe) (7404), and PLP no. 1 to No. N (7405_1 to 7405_N). (At this point, the pre-signaling data L1 (7402) and the post-signaling data L1 (7403) are called symbols P2). As such, signaling data P1 (7401), pre-signaling data L1 (7402), post-signaling data L1 (7403), a common PLP (Physical Layer Pipe) (7404), and PLP n. 1 to No. N (7405_1 to 7405_N) form a frame, which is called a T2 frame, thus constituting a frame configuration unit.

The signaling data P1 (7401) is a symbol used by the receiving device for signal detection and frequency synchronization (including frequency offset estimation), which serves

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simultaneously to transmit information such as the FFT size and if the modulated signal is transmitted by a SISO or MISO method. (With SISO methods, only a modulated signal is transmitted, while with MISO methods, a plurality of modulated signals are transmitted. In addition, the space-time blocks described in Non-Patent Bibliography 9, 16 and 17 can be used ).

The pre-signaling data L1 (7402) is used to transmit information with respect to the methods used to transmit the frame, with respect to the guard interval, the signal processing method information used to reduce the PAPR (Peak Ratio to Medium Power), the modulation method used to transmit the post-signaling data L1, the FEC method, the coding rate thereof, the length and size of the post-signaling data L1, the useful fit pattern, the specific cell numbers (frequency region), and if normal or extended mode is in use (where normal mode and extended mode differ in terms of subcarrier numbers used to transmit data).

The L1 post-signaling data (7403) is used to transmit information such as the number of PLPs, the region of frequency in use, the specific numbers of PLPs, the modulation method used to transmit the PLPs, the FEC method , the coding rate thereof, the number of blocks transmitted by each PLP, and so on! successively.

Common PLP (7404) and PLP # 1 to No. N (7405_1 to 7405_N) are areas used for data transmission.

The frame configuration from Figure 74 illustrates the signaling data P1 (7401), pre-signaling data L1 (7402), post-signaling data L1 (7403), the common PLP (Physical Layer Pipe) ( 7404), and PLPs No. 1 to No. N (7405_1 to 7405_N) divided with respect to the time domain for transmission. However, two or more of these signals can take place simultaneously. Figure 75 illustrates such a case. As shown, the pre-signaling data L1, post-signaling data L1, and the common PLP take place in the same time indication, while the PLP # 1 and PLP # 2 take place simultaneously in another time indication That is, each signal can coexist at the same point with respect to the time or frequency domain in the frame configuration.

Figure 76 illustrates a sample configuration of a transmission device (for example, a diffuser) that applies a transmission method in which a phase change is made in precoded (or precoded and changed) signals according to DVB- T2

A PLP 7602 signal generator takes transmission data from PLP 7601 (data for PLP) and a control signal 7609 as input, performs error correction coding according to the error correction code information for the included PLPs in the control signal 7609 and performs mapping according to the modulation method similarly included in the control signal 7609, and then emits a baseband signal of PLP (quadrature) 7603.

A P2 7605 signal signal generator takes P2 7604 signal transmission data and the 7609 control signal as input, performs error correction coding according to the error correction code information for the P2 symbol included in the signal of control 7609 and performs mapping according to the modulation method similarly included in the control signal 7609, and then emits a baseband signal of symbol P2 (quadrature) 7606.

A control signal generator 7608 takes signal transmission data P1 7607 and transmission data symbol P2 7604 as input and issues control signal 7609 for the group of symbols from Figure 74 (signaling data P1 ( 7401), L1 pre-signaling data (7402), L1 post-signaling data (7403), common PLP (7404), and PLP # 1 to No. N (7405_1 to 7405_N)) . The control signal 7609 is composed of transmission method information (such as error correction codes and the coding rate thereof, the modulation method, the block length, the frame configuration, the transmission method selected in which the precoding matrix is changed regularly, the pilot symbol insertion method, IFFT / FFT information, the PAPR reduction method and the guard interval insertion method) for the symbol group.

A frame configurator 7610 takes a baseband signal of PLP 7603, the baseband signal of symbol P2 7606, and the control signal 7609 as input, performs rearrangement with respect to the time and frequency domains according to the information of frame configuration included in the control signal, and accordingly emits (quadrature) the baseband signal 7611_1 for flow 1 (a mapped signal, that is, a baseband signal in which the method of use has been used modulation) and (quadrature) the baseband signal 7611_2 for flow 2 (also a mapped signal, that is, a baseband signal in which the modulation method has been used).

A signal processor 7612 takes the baseband signal for flow 1 7611_1, the baseband signal for flow 2 7611_2, and control signal 7609 as input, and then outputs the modulated signals 1 (7613_1) and 2 ( 7613_2), processed in accordance with the transmission method included in control signal 7609.

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At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor makes the phase change in the precoded (or precoded and changed signals) ) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2).

A pilot inserter 7614_1 takes the processed modulated signal 1 (7613_1) and the control signal 7609 as input, inserts pilot symbols into the processed modulated signal 1 (7613_1) in accordance with the information of the pilot symbol insertion method included in the control signal 7609, and emits a modulated signal of insertion of post-pilot symbol 7615_1.

Another pilot inserter 7614_2 takes the processed modulated signal 2 (7613_2) and the control signal 7609 as input, inserts pilot symbols into the processed modulated signal 2 (7613_2) in accordance with the information of the pilot symbol insertion method included in the control signal 7609, and emits a modulated signal of insertion of post-pilot symbol 7615_2.

An IFFT unit 7616_1 takes the modulated post-pilot symbol insert signal 7615_1 and the control signal 7609 as input, applies an IFFT according to the IFFT method information included in the control signal 7609, and issues the signal post-IFFT 7617_1.

Another IFFT unit 7616_2 takes the modulated post-pilot symbol insert signal 7615_2 and the control signal 7609 as input, applies an IFFT according to the IFFT method information included in the control signal 7609, and issues the signal post-IFFT 7617_2.

The PAPR reducer 7618_1 takes the post-IFFT signal 7617_1 and the control signal 7609 as input, applies PAPR reduction processing to the post-IFFT signal 7617_1 according to the PAPR reduction information included in the control signal 7609 , and issues the post-PAPR reduction signal 7619_1.

The PAPR reducer 7618_2 takes the post-IFFT signal 7617_2 and the control signal 7609 as input, applies PAPR reduction processing to the post-IFFT signal 7617_2 in accordance with the PAPR reduction information included in the control signal 7609 , and issues the post-PAPR reduction signal 7619_2.

The guard interval inserter 7620_1 takes the post-PAPR reduction signal 7619_1 and the control signal 7609 as input, inserts guard intervals in the post-PAPR reduction 7619_1 according to the included guard interval insertion method information in control signal 7609, and issues the post-interval guard insertion signal 7621_1.

The guard interval inserter 7620_2 takes the post-PAPR reduction signal 7619_2 and the control signal 7609 as input, inserts guard intervals in the post-PAPR reduction 7619_2 in accordance with the included guard interval insertion method information in control signal 7609, and issues the post-interval guard insertion signal 7621_2.

A P1 7622 symbol inserter takes the P1 7607 symbol transmission data and the 7621_1 and 7621_2 guard post-interval insertion signals as input, generates P1 signal signals from the P1 7607 symbol transmission data, add the symbols P1 to the respective guard post-interval insertion signals 7621_1 and 7621_2, and issues the post-P1 symbol addition signals 7623_1 and 7623_2. The P1 symbol signals can be added to one or both of the guard post-interval insert signals 7621_1 and 7621_2. In the latter case, the signal to which nothing is added has zero signals as the baseband signal in the interval to which the symbols are added to the other signal.

The 7624_1 wireless processor takes the post-P1 7623_1 symbol input signal as input, performs processing such as frequency conversion and amplification therein, and issues the 7625_1 transmission signal. The transmission signal 7625_1 is then broadcast as radio waves through the antenna 7626_1.

The 7624_2 wireless processor takes the post-P1 7623_2 symbol addition signal as input, performs processing such as frequency conversion and amplification therein, and issues the 7625_2 transmission signal. The transmission signal 7625_2 is then broadcast as radio waves by the antenna 7626_2.

Figure 77 illustrates a sample frame configuration in the time-frequency domain where a plurality of PLP is transmitted after the symbol P1, symbol P2 and Common PLP have been transmitted. As shown, with respect to the frequency domain, flow 1 (a mapped signal, that is, a baseband signal in which the modulation method has been used) uses subcarriers # 1 to # M, as does flow 2 (also a mapped signal, that is, a baseband signal in which the modulation method has been used). Therefore, when both s1 and s2 have a symbol on the same subcarrier at the same time indication, a symbol from each of the two flows is present at a single frequency. As explained in other embodiments, when a transmission method is used that involves performing a phase change in signals

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precoded (or precoded and changed), the phase change can be made in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

As shown in Figure 77, the interval 1 is used to transmit the group of symbols 7701 of the PLP # 1 using the flow s1 and the flow s2. The data is transmitted using a spatial multiplexing MIMO system as illustrated by Figure 23, or using a MIMO system with a fixed precoding matrix (where no phase change is made).

Interval 2 is used to transmit the group of symbols 7702 of PLP # 2 using the flow s1. Data is transmitted using a modulated signal.

The interval 3 is used to transmit the group of symbols 7703 of the PLP # 3 using the flow s1 and the flow s2. The data is transmitted using a transmission method in which a phase change is made in precoded (or precoded and changed) signals.

The interval 4 is used to transmit the group of symbols 7704 using the flow s1 and the flow s2. The data is transmitted using the time-space block codes described in the non-patent Bibliography 9, 16 and 17.

When a diffuser transmits the PLPs as illustrated by Figure 77, the receiving device from Figure 77 that receives the transmission signals must know the transmission method of each PLP. Therefore, as described above, the post-signaling data L1 (7403 from Figure 74), which is the symbol P2, should transmit the transmission scheme for each PLP. The following describes an example of a configuration method for symbols P1 and P2 in such circumstances.

Table 2 lists specific examples of control information carried by the symbol P1.

[Table 21

 S1 (3 bits)

 Control information

 000

 T2_SISO (transmission of a signal modulated in DVB-T2)

 001

 T2_MISO (transmission using space-time block codes in DVB-T2 standard)

 010

 NO_T2 (using a standard other than DVB-T2)

In the DVB-T2 standard, control information S1 (three data bits) is used by the receiving device to determine whether DVB-T2 is being used or not, and if so, to determine the transmission method.

As indicated in Table 2, above, the 3 data bits S1 are each set to 000 to indicate that the modulated signals are being transmitted according to the transmission of a signal modulated in the DVB-T2 standard.

Alternatively, the 3 data bits S1 are set to 001 to indicate that the modulated signals are being transmitted according to the use of time-space block codes in the DVB-T2 standard.

In DVB-T2, 010 to 111 are reserved for future use. To apply the present invention while maintaining compatibility with DVB-T2, the 3 bits of data S1 should be set to 010, for example (any other than 000 and 001 may be used), and should indicate that a standard other than DVB-T2 for modulated signals. Therefore, the receiving device or terminal can determine that the diffuser is transmitting using the modulated signals according to a standard other than DVB-T2 detecting that the data read is 010.

The following describes an example of a configuration method for a P2 symbol used when the modulated signals transmitted by the diffuser conform to a standard other than DVB-T2. In the first example, a scheme to use the P2 symbol in the DVB-T2 standard.

Table 3 lists a first example of control information transmitted by the post-signaling data L1 in the symbol P2.

[Table 31

 PLP MODE (2 bits)

 Control information

 00

 SISO / SIMO

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 01

 MISO / MIMO (time space block codes)

 10

 MIMO (make a phase change in precoded signals (or precoded signals that have changed base bands))

 eleven

 MIMO (use a fixed precoding matrix, or use spatial multiplexing)

The tables provided above use the following abbreviations.

SISO: Input-Unica Output-Unica (a modulated signal transmitted and received by an antenna)

SIMO: Input-Single Output-Multiple (a modulated signal transmitted and received by multiple antennas)

MISO: Input-Multiple Output-Single (multiple modulated signals transmitted by multiple antennas and received by a single antenna)

MIMO: Input-Multiple Output-Multiple (multiple modulated signals transmitted and received by multiple antennas)

The two data bits listed in Table 3 are the information of MODO_PLP. As shown in Figure 77, this information is control information to inform the terminal of the transmission method (the group of symbols of PLP No. 1 to No. 4 in Figure 77; hereinafter, the group of symbols). MODO_PLP information is present in each PLP. That is, in Figure 77, the information of MODO_PLP for PLP No. 1, for PLP No. 2, for PLP No. 3, for PLP No. 4, and so on! successively, it is transmitted by the diffuser. Obviously, the terminal acknowledges the transmission method used by the diffuser for the PLPs by demodulating (or performing the error correction decoding in) this information.

When the MODE_PLP is set to 00, the data is transmitted by that PLP using a method in which a single modulated signal is transmitted. When the MODE_PLP is set to 01, the data is transmitted by that PLP using a method in which multiple modulated signals are transmitted using space-time block codes. When the MODE_PLP is set to 10, the data is transmitted by that PLP using a method in which a phase change is made to precoded (or precoded and changed) signals. When the MODE_PLP is set to 11, the data is transmitted by that PLP using a method in which a fixed precoding matrix is used, or in which a spatial multiplexing MIMO system is used.

When the MODE_PLP is set to any one from 01 to 11, the broadcaster must transmit the specific processing (for example, the specific transmission method by which the phase change is applied to precoded (or precoded and changed) signals, the method of coding of space-time block codes, or the configuration of the precoding matrix) to the terminal. The following describes an alternative to Table 3, such as a configuration method for control information that includes the necessary control information under such circumstances.

Table 4 lists a second example of control information transmitted by the post-signaling data L1 in the symbol P2, different from that of Table 3.

[Table 41

 Name

 No. of bits Control information

 PLP MODE (1 bit)

 0 SISO / SIMO

 1 MISO / MIMO, using one of (i) space-time block codes; (ii) change in phase made in precoded signals (or precoded signals that have changed base bands); (iii) a fixed precoding matrix; and (iv) spatial multiplexing

 MINIMUM MODE (1 bit)

 0 phase change in precoded signals (or precoded signals that have changed base bands) is OFF

 one

 phase change in precoded signals (or precoded signals that have changed base bands) is ON

 MIMO PATTERN # 1 (2 bits)

 00 space-time block codes

 01

 fixed precoding matrix # 1

 10

 fixed precoding matrix # 2

 eleven

 spatial multiplexing

 PATRON_MIMO # 2 (2 bits)

 00 phase change in precoded signals (or precoded signals that have changed base bands), version # 1

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 01 phase change in precoded signals (or precoded signals that have changed base bands), version # 2

 10

 phase change in precoded signals (or precoded signals that have changed base bands), version # 3

 eleven

 phase change in precoded signals (or precoded signals that have changed base bands), version # 4

As indicated in Table 4, four types of control information are possible: 1 bit MODE_PLP information, 1 bit MODE_MIMO information, 2 bit PATRON_MIMO information 1 and 2 PATRON_MIMO information No. 2 of 2 bits As shown in Figure 77, the terminal of the transmission method is notified for each PLP (specifically PLPs No. 1 to No. 4) by this information. The four types of control information are present in each PLP. That is, in Figure 77, the information of MODO_PLP, the information of MODO_MIMO, information of PATRON_MIMO No. 1, and information of PATRON_MIMO No. 2 for PLP No. 1, for PLP No. 2, for PLP # 3, for PLP # 4, and so on! successively, it is transmitted by the diffuser. Obviously, the terminal acknowledges the transmission method used by the diffuser for the PLPs by demodulating (or performing the error correction decoding in) this information.

When the MODE_PLP is set to 0, the data is transmitted by that PLP using a method in which a single modulated signal is transmitted. When the MODE_PLP is set to 1, the data is transmitted by that PLP using a method in which any one of the following applies: (i) space-time block codes are used; (ii) a MIMO system is used where a phase change is made in precoded (or precoded and changed) signals; (iii) a MIMO system is used where a fixed precoding matrix is used; and (iv) spatial multiplexing is used.

When the MODO_PLP is set to 1, the information of MODO_MIMO is valid. When the MODE_MIMO information is set to 0, data is transmitted without using a phase change made to precoded (or precoded and changed) signals. When the MODE_MIMO information is set to 1, data is transmitted using a phase change made to the precoded (or precoded signals that have changed base bands).

When the MODO_PLP is set to 1 and the information in MODO_MIMO is set to 0, the information in PATRON_MIMO No. 1 is valid. When PATRON_MIMO # 1 information is set to 00, data is transmitted using space-time block codes. When the PATRON_MIMO # 1 information is set to 01, data is transmitted using the fixed precoding matrix # 1 for weighting. When the PATRON_MIMO # 1 information is set to 10, data is transmitted using the fixed precoding matrix # 2 for weighting. (The precoding matrix # 1 and the precoding matrix # 2 are different matrices). When PATRON_MIMO # 1 information is set to 11, data is transmitted using MIMO spatial multiplexing.

When the MODO_PLP is set to 1 and the information in MODO_MIMO is set to 1, the information in PATRON_MIMO No. 2 is valid. When the PATRON_MIMO No. 2 information is set to 00, data is transmitted using version # 1 of a phase change in the precoded (or precoded signals that have changed base bands). When the PATRON_MIMO # 2 information is set to 01, data is transmitted using version # 2 of a phase change in the precoded (or precoded signals that have changed base bands). When the PATRON_MIMO # 2 information is set to 10, data is transmitted using version # 3 of a phase change in the precoded (or precoded signals that have changed base bands). When PATR6n_MIMO No. 2 information is set to 11, data is transmitted using version # 4 of a phase change in the precoded (or precoded signals that have changed base bands). Although the change in phase is carried out in four different versions No. 1 to 4, the following three approaches are possible, given two different methods No. A and No. B:

Phase changes made using method # A and made using method # B include changes

identical and different.

A phase change value included in method # A is not included in method # B; Y

Multiple phase changes used in method # A are not included in method # B.

The control information listed in Table 3 and Table 4, above, is transmitted by the post-signaling data L1 in the symbol P2. However, in DVB-T2, the amount of information transmitted as a P2 symbol is limited. Therefore, the information listed in Tables 3 and 4 must be added to the information that must be transmitted by means of the P2 symbol in the DVB-T2 standard. When this leads to exceeding the limit in transmissible information such as the symbol P2, then as shown in Figure 78, a signaling PLP (7801) can be prepared to transmit necessary control information (at least partially, that is, transmit the data of post-signaling L1 and signaling PLP) not included in the DVB-T2 specification. Although Figure 78 illustrates an identical frame configuration to that of Figure 74, no limitation is intended in this regard. A specific time and specific carrier region can also be assigned in the domain of

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time-frequency for signaling PLP, as in Figure 75. That is, signaling PLP can be freely assigned in the time-frequency domain.

As described above, select a transmission method that uses a multi-carrier method such as OFDM and retains compatibility with the DVB-T2 standard, and in which the phase change is performed on precoded (or precoded and changed) signals. It has the advantages of driving at a better reception quality in the LOS environment and at higher transmission speeds. Although the present invention describes possible transmission methods for carriers as they are MIMO of spatial multiplexing, MIMO using a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals , space-time block codes, and transmission methods that only transmit the flow s1, no limitation is intended in this way.

Also, although the description indicates that the diffuser selects one of the above-mentioned transmission methods, these are not the only transmission methods available for selection. Other options include:

MIMO that uses a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals, space-time block codes, and transmission methods that transmit only the flow s1;

MIMO that uses a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals, and space-time block codes;

MIMO using a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals, and transmission methods that only transmit the s1 flow;

A transmission method that performs a phase change in precoded (or precoded and changed) signals, space-time block codes and transmission methods that transmit only the flow s1;

MIMO that uses a fixed precoding matrix and a transmission method that performs a phase change in precoded signals (or precoded and changed);

A transmission method that performs a phase change in precoded signals (or precoded and changed) and space-time block codes;

A transmission method that performs a phase change in precoded (or precoded and changed) signals and transmission methods that transmit only the s1 stream.

As such, including a transmission method that performs a phase change in precoded signals (or in precoded and changed signals), the advantages of driving at higher data rates in the LOS environment and better reception quality are achieved. receiving device

At this point, since, as described above; S1 must be set for the P1 symbol, another configuration method is possible for the control information (with respect to the transmission method for each PLP), different from that of Table 3. For example, Table 5, below.

[Table 51

 PLP MODE (2 bits)

 Control information

 00

 SISO / SIMO

 01

 MISO / MIMO (space-time block codes)

 10

 MIMO (phase change in precoded signals (or precoded signals that have changed base bands))

 eleven

 Reserved

Table 5 differs from Table 3 in that setting the information of MODO_PLP to 11 is reserved. As such, when the transmission method for PLPs is as described in one of the previous examples, the number of bits that form the information of MODO_PLP as in the examples of Tables 3 and 5 can be made greater or lesser according to the transmission methods available for selection.

Similarly, for Table 4, when, for example, a MIMO method is used with a transmission method that does not support changing the phase of precoded signals (or precoded signals that have changed base bands), the control information of MODO_MIMO is not necessary. Also, when, for example, MIMO schemes that use a fixed precoding matrix are not supported, then PATTERN_MIMO # 1 is not necessary. Also, when multiple precoding matrices are not necessary, 1-bit information can be used instead of 2-bit information. Additionally, two or more bits may be used when a plurality of precoding matrices are available.

The same principles apply to PATRON_MIMO # 2 information. When the transmission method does not require a plurality of methods to perform a phase change in precoded (or precoded and changed) signals, 1 bit information may be used instead of information of 2 bits. Additionally, two or more bits may be used when a plurality of phase change schemes are available.

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Additionally, although the present embodiment describes a transmission device having two antennas, no limitation is intended in this regard. Control information can also be transmitted using more than two antennas. In such circumstances, the number of bits in each type of control information can be increased as required to perform transmission using four antennas. The transmission of control information of the above description in the symbol P1 and P2 also applies to such cases.

Although Figure 77 illustrates the frame configuration for the PLP symbol groups transmitted by the diffuser as being divided with respect to the time domain, the following variation is also possible.

Unlike Figure 77, Figure 79 illustrates an example of a method for arranging the flow if and the flow 2 of the symbols in the time-frequency domain, after the Pi symbol, the P2 symbol has been transmitted and the Common PLP. In Figure 79, the symbols labeled No. 1 are the symbols of the symbol group of PLP No. 1 from Figure 77. Similarly, the symbols labeled No. 2 are the symbols of the group of PLP symbols # 2, the symbols labeled # 3 are the symbols of the symbol group of the PLP # 3, and the symbols labeled # 4 are the symbols of the symbol group of the PLP No. # 4, all from Figure 77. As in Figure 77, PLP # 1 is used to transmit data using a spatial multiplexing MIMO system as illustrated by Figure 23, or using a system of MIMO with a fixed precoding matrix. PLP # 2 is used to transmit data using only a modulated signal. PLP # 3 is used to transmit data using a transmission method in which a phase change is made in precoded (or precoded and changed) signals. PLP # 4 is used to transmit data using space-time block codes.

In Figure 79, when both s1 and s2 have a symbol on the same subcarrier (given as the carrier in Figure 79) at the same time indication, a symbol from each of the two flows is present in the frequency common. As explained in other embodiments, when a transmission method is used that involves performing a phase change in precoded (or precoded and changed) signals, the phase change can be performed in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

As described above, Figure 79 differs from Figure 77 in that the PLPs are divided with respect to the time domain. In addition, Figure 79 has a plurality of PLPs arranged with respect to time and frequency domains. That is, for example, the symbols of PLP No. 1 and PLP No. 2 are in the time indication 1, while the symbols of PLP No. 3 and PLP No. 4 are in the time indication 3 As such, PLP symbols that have a different Index (# X, where X = 1, 2, and so on) can be assigned to each symbol (consisting of a time indication and a subcarrier).

Although, for the sake of simplicity, Figure 79 lists only No. 1 and No. 2 in the time indication 1, no limitation is intended in this regard. The indices of PLP symbols other than No. 1 and No. 2 may be in the time indication No. 1. Additionally, the relationship between PLP indices and sub-carriers in the time indication 1 is not limited. to that illustrated by Figure 79. The indices of any PLP symbol can be assigned to any subcarrier. The same applies to other time indications, in which the indices of any PLP symbol can be assigned to them.

Unlike Figure 77, Figure 80 illustrates an example of a method for arranging the flow s1 and the flow 2 of the symbols in the time-frequency domain, after the symbol P1, the symbol P2 and the Common PLP. The characteristic feature of Figure 80 is that, supposing that a plurality of antennas is used for transmission is the basis of the PLP transmission method, then transmission using only flow 1 is not an option for frame T2.

Accordingly, in Figure 80, the PLP 8001 symbol group transmits data using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix. Also, the group of symbols 8002 of PLP # 2 transmits data using a transmission method that performs a phase change in precoded (or precoded and changed) signals. In addition, the group of symbols 8003 of PLP # 3 transmits data using space-time block code. The group of PLP symbols following the group of symbols 8003 of PLP # 3 transmits data using one of these methods, in particular using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix , using a transmission method that performs a phase change in precoded (or precoded and changed) signals, or using space-time block codes.

Unlike Figure 79, Figure 81 illustrates an example of a method for arranging the flow s1 and the flow 2 of the symbols in the time-frequency domain, after the symbol P1, the symbol P2 and the Common PLP. In Figure 81, the symbols labeled # 1 are the symbols of the group of symbols of the PLP # 1 from Figure 80. Similarly, the symbols labeled # 2 are the symbols of the group of PLP symbols # 2, the symbols labeled # 3 are the symbols of the symbol group of the PLP # 3, and the symbols labeled # 4 are the symbols of the symbol group of the PLP No. # 4, all from Figure 80. As in Figure 80, PLP # 1 is used to transmit data using a spatial multiplexing MIMO system as described.

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illustrated by Figure 23, or using a MIMO system with a fixed precoding matrix. PLP # 2 is used to transmit data using a transmission method in which a phase change is made in precoded (or precoded and changed) signals. PLP # 3 is used to transmit data using space-time block codes.

In Figure 81, when both if and s2 have a symbol on the same subcarrier (given as the carrier in Figure 81) at the same time indication, a symbol from each of the two flows is present in the frequency common. As explained in other embodiments, when a transmission method is used that involves performing a phase change in precoded (or precoded and changed) signals, the phase change can be performed in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

Figure 81 differs from Figure 80 in that the PLPs are divided with respect to the time and frequency domains. That is, for example, the PLP # 1 and PLP # 2 symbols are both in the time indication 1. As such, the PLP symbols having a different Index (# X, where X = 1, 2, and so on) can be assigned to each symbol (consisting of a time indication and a subcarrier).

Although, for the sake of simplicity, Figure 81 lists only No. 1 and No. 2 in the time indication 1, no limitation is intended in this regard. Indices of PLP symbols other than No. 1 and No. 2 may be in the time indication No. 1. Additionally, the relationship between PLP Indices and sub-carriers in the time indication 1 is not limited. to that illustrated by Figure 81. The indices of any PLP symbol can be assigned to any subcarrier. The same applies to other time indications, in which the indices of any PLP symbol can be assigned to them. On the other hand, a time indication can also have the symbols of only one PLP assigned to it, as is the case for the time indication 3. In other words, any assignment of PLP symbols in the time-frequency domain It is permissible.

Therefore, since the T2 frame does not include PLP using transmission methods that only transmit the s1 stream, the dynamic range of the signals received through the terminal can be restricted, which is likely to lead to improved received signal quality.

Although Figure 81 is described using examples to select one of transmitting data using a spatial multiplexing MIMO system, or an MlMO system using a fixed precoding matrix, transmitting data using a transmission method that performs a phase change in signals precoded (or precoded and changed), and transmitting data using space-time block codes, the selection of the transmission method is not limited as such. Other possibilities include:

select one to transmit data using a transmission method that performs a phase change in precoded (or precoded and changed) signals, transmit data using space-time block codes, and transmit data using a MIMO system using a matrix of fixed precoding;

select one to transmit data using a transmission method that performs a phase change in precoded (or precoded and changed) signals, and transmit data using space-time block codes; Y

select one to transmit data using a transmission method that performs a phase change in precoded (or precoded and changed) signals and transmit data using a MIMO system using a fixed precoding matrix.

Although the above explanation is provided for a T2 frame that has multiple PLPs, the following describes a T2 frame that has only one PLP.

Figure 82 illustrates a sample frame configuration for flow s1 and flow s2 in the time-frequency domain where frame T2 has only one PLP. Although Figure 82 indicates control symbols, these are equivalent to the previously described symbols, such as symbols P1 and P2. In Figure 82, interval 1 is used to transmit a first frame T2, interval 2 is used to transmit a second frame T2, interval 3 is used to transmit a third frame T2, and interval 4 is used to transmit a fourth frame T2.

Additionally, the first frame T2 in Figure 82 transmits the group of symbols 8101 of PLP No. 1-1. The selected transmission method is MIMO of spatial multiplexing or MIMO that uses a fixed precoding matrix.

The second frame T2 transmits the group of symbols 8102 of the PLP No. 2-1. The transmission method is a transmission that uses a unique modulated signal.

The third frame T2 transmits the group of symbols 8103 of the PLP No. 3-1. The transmission method is a transmission that makes a phase change in precoded signals (or in precoded and changed signals).

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The fourth frame T2 transmits the group of slots 8104 of the PLP No. 4-1. The transmission method is transmission using space-time block codes.

In Figure 82, when both s1 and s2 have a symbol on the same subcarrier at the same time indication, a symbol from each of the two flows is present in the common frequency. As explained in other embodiments, when a transmission method is used that involves performing a phase change in precoded signals (or precoded signals having changed base bands), the phase change can be made in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

As such, the transmission method can be set to take into account the data transmission rate and the terminal data reception rate for each PLP. This has the double advantage of allowing data transmission speed to be improved and high quality data reception to be ensured. The configuration method for the control information pertaining to the transmission method and so! successively for the symbols P1 and P2 (and the signaling PLP, where applicable) can be provided as per Tables 2 to 5, thereby obtaining the same effects. Figure 82 differs from Figure 77 in that, while the frame configuration from Figure 77 and the like includes multiple PLPs in a single frame T2, therefore requiring the control information pertaining to the transmission method and thus ! successively of each PLP, the frame configuration of Figure 82 includes only one PLP per frame T2. As such, the only control information needed is for the transmission information and so on! successively it belongs to the PLP.

Although the above description analyzes methods for transmitting information pertaining to the transmission method of the PLPs using symbols P1 and P2 (and the signaling PLP, where applicable), the following describes a method for transmitting information pertaining to the transmission method of PLPs without using the P2 symbol.

Figure 83 illustrates a frame configuration in the applicable time-frequency domain when a terminal receiving data transmitted through a diffuser is not compatible with DVB-T2. In Figure 83, components that operate in the same manner described for Figure 74 use identical reference numbers. The plot of Figure 83 includes signaling data P1 (7401), first signaling data (8301), second signaling data (8302), a common PLP (7404), and the PLP # 1 to No. N (7405_1 to 7405_N). As such, the signaling data P1 (7401), the first signaling data (8301), the second signaling data (8302), the common PLP (7404), and the pLp No. 1 to No. N ( 7405_1 to 7405_N) form a frame, thus constituting a frame unit.

The signaling data P1 (7401) is a symbol used for signal reception by the reception device and for frequency synchronization (including frequency offset estimation). In addition, these data convey identification with respect to whether or not the frame conforms to DVB-T2, for example, using the S1 data as indicated in Table 2 for this purpose.

The first signaling data (8301) is used to transmit information regarding the methods used to transmit the frame, with respect to the guard interval, the signal processing method information used to reduce the PAPR, the modulation method used to transmit the post-signaling data L1, the FEC method, the codification rate thereof, the length and size of the post-signaling data L1, the useful fit pattern thereof, the specific cell numbers (frequency region), and if normal or extended mode is in use, and other information of this type. At this point, the first signaling data (8301) need not necessarily be data in accordance with the DVB-T2 standard.

The second signaling data (8302) is used to transmit information such as the PLP number, the region of frequency in use, the specific numbers of PLP, the modulation method used to transmit the PLPs, the FEC method, the rate encoding them, the number of blocks transmitted by each PLP, and so on! successively.

The frame configuration from Figure 83 illustrates the first signaling data (8301), the second signaling data (8302), the post-signaling data L1 (7403), the common PLP (7404), and the PLP No. 1 to No. N (7405_1 to 7405_N) divided with respect to the time domain for transmission. However, two or more of these signals can take place simultaneously. Figure 84 illustrates such a case. As shown in Figure 84, the first signaling data, the second signaling data, and the common PLP share a common time indication, while the PLP # 1 and PLP # 2 share a common time indication different. That is, each signal can coexist at the same point with respect to the time or frequency domain in the frame configuration.

Figure 85 illustrates a sample configuration of a transmission device (eg, a diffuser) that applies a transmission method in which a phase change is made in precoded (or precoded and changed) signals as explained so far. , but according to a standard other than the DVB-T2 standard. In Figure 85, components that operate in the same manner described for Figure 76 use identical numbers of

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reference and invoke the above descriptions.

A control signal generator 7608 takes first and second signaling data 8501 and signal transmission data P1 7607 as input, and issues control signal 7609 (composed of information such as error correction codes and coding rate of the same, the modulation method, the block length, the frame configuration, the selected transmission method in which the precoding matrix is changed regularly, the pilot symbol insertion method, IFFT / FFT information, the method PAPR reduction, and the guard interval insertion method) for the transmission method of each group of symbols in Figure 83.

A control symbol signal generator 8502 takes the first and second data of signaling data transmission 8501 and the control signal 7609 as input, performs error correction coding according to the error correction code information for the first and second signaling data included in control signal 7609 and mapping according to the modulation method similarly included in control signal 7609, and then emits a first baseband signal of first and second signaling data (square) 8503.

In Figure 85, the frame configurator 7610 takes the baseband signal 8503 generated by the control symbol signal generator 8502 as input, instead of the baseband signal 7606 generated by the symbol signal generator P2 7605 from Figure 76.

The following describes, with reference to Figure 77, a transmission method for control information (information transmitted by means of the P1 symbol and by the first and second signaling data) and for the frame configuration of the transmission signal for a diffuser (the base station) by applying a transmission method in which a phase change is made in precoded (or precoded and changed) signals in a system not conforming to DVB-T2.

Figure 77 illustrates a sample frame configuration in the time-frequency domain where a plurality of PLP is transmitted after the first and second signaling data and the Common PLP have been transmitted. In Figure 77, flow s1 uses subcarrier # 1 to subcarrier # M in the frequency domain. Similarly, flow s2 also uses subcarrier # 1 to subcarrier # M in the frequency domain. Therefore, when both s1 and s2 have a symbol on the same subcarrier at the same time indication, a symbol from each of the two flows is present on a single frequency. As explained in other embodiments, when a transmission method is used that involves performing a phase change in precoded (or precoded and changed) signals, the phase change can be performed in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

As shown in Figure 77, the interval 1 is used to transmit the group of symbols 7701 of the PLP # 1 using the flow s1 and the flow s2. The data is transmitted using a spatial multiplexing MIMO system as illustrated by Figure 23, or using a MIMO system with a fixed precoding matrix.

Interval 2 is used to transmit the group of symbols 7702 of PLP # 2 using the flow s1. Data is transmitted using a modulated signal.

The interval 3 is used to transmit the group of symbols 7703 of the PLP # 3 using the flow s1 and the flow s2. The data is transmitted using a transmission method in which a phase change is made in precoded (or precoded and changed) signals.

The interval 4 is used to transmit the group of symbols 7704 of the PLP # 4 using the flow s1 and the flow s2. The data is transmitted using the time-space block codes.

When a diffuser transmits the PLPs as illustrated by Figure 77, the receiving device from Figure 64 receiving the transmission signals must know the transmission method of each PLP. Therefore, as described above, the first and second signaling data should be used to transmit the transmission method for each PLP. The following describes an example of a configuration method for the P1 symbol and for the first and second signaling data in such circumstances. A specific example of control information carried by the symbol P1 is provided in Table 2.

In the DVB-T2 standard, the control information S1 (three data bits) is used by the receiving device to determine whether or not DVB-T2 is being used, and if so, to determine the transmission method. The 3 data bits S1 are set 000 to indicate that the modulated signals are being transmitted according to the transmission of a signal modulated in the DVB-T2 standard.

Alternatively, the 3 bits of data S1 are set to 001 to indicate that the modulated signals are being

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transmitting according to the use of time-space block codes in the DVB-T2 standard.

In DVB-T2, 010 to 111 are reserved for future use. To apply the present invention while maintaining compatibility with DVB-T2, the 3 bits of data S1 should be set to 010, for example (any other than 000 and 001 may be used), and should indicate that a standard other than DVB-T2 for modulated signals. Therefore, the reception device or the terminal can determine that the diffuser is transmitting using the modulated signals according to a standard other than DVB-T2 detecting that the data read is 010.

The following describes a configuration method for the first and second signaling data used when the modulated signals transmitted by the diffuser are not conforming to the DVB-T2 standard. A second example of control information for the first and second signaling data is provided by Table 3.

The two bits of data listed in Table 3 are the information of MODO_PLP. As shown in Figure 77, this information is control information to inform the terminal of the transmission method for each PLP (PLP No. 1 to No. 4 in Figure 77). MODO_PLP information is present in each PLP. That is, in Figure 77, the information of MODO_PLP for PLP No. 1, for PLP No. 2, for PLP No. 3, for PLP No. 4, and so on! successively, it is transmitted by the diffuser. Obviously, the terminal acknowledges the transmission method used by the diffuser for the PLPs by demodulating (or performing the error correction decoding in) this information.

When the MODE_PLP is set to 00, the data is transmitted by that PLP using a method in which a single modulated signal is transmitted. When the MODE_PLP is set to 01, the data is transmitted by that PLP using a method in which multiple modulated signals are transmitted using space-time block codes. When the MODE_PLP is set to 10, the data is transmitted by that PLP using a method in which a phase change is made to precoded (or precoded and changed) signals. When the MODE_PLP is set to 11, the data is transmitted by that PLP using a method in which a fixed precoding matrix is used, or in which a spatial multiplexing MIMO system is used.

When the MODE_PLP is set to any one from 01 to 11, the broadcaster must transmit the specific processing (for example, the specific transmission method by which a phase change is applied to precoded (or precoded and changed) signals, the method of codification of space-time block codes, or the configuration of the precoding matrix) to the terminal. The following describes an alternative to Table 3, such as a configuration method for control information that includes the necessary control information under such circumstances.

A second example of control information for the first and second signaling data is provided by Table 4.

As indicated in Table 4, four types of control information are possible: 1-bit MODE_PLP information, 1-bit MODE_MlMO information, 2-bit PATRON_MIMO # 1 information, and PATRON_MIMO # 2 information 2 bit As shown in Figure 77, the terminal of the transmission method is notified for each PLP (specifically PLP No. 1 to No. 4) by this information. The four types of control information are present in each PLP. That is, in Figure 77, the information of MODO_PLP, the information of MODO_MIMO, information of PATRON_MIMO No. 1, and information of PATRON_MIMO No. 2 for PLP No. 1, for PLP No. 2, for PLP # 3, for PLP # 4, and so on! successively, it is transmitted by the diffuser. Obviously, the terminal acknowledges the transmission method used by the diffuser for the PLPs by demodulating (or performing the error correction decoding in) this information.

When the MODE_PLP is set to 0, the data is transmitted by that PLP using a method in which a single modulated signal is transmitted. When the MODE_PLP is set to 1, the data is transmitted by that PLP using a method in which any one of the following applies: (i) space-time block codes are used; (ii) a MIMO system is used where a phase change is made in precoded (or precoded and changed) signals; (iii) a MIMO system is used where a fixed precoding matrix is used; and (iv) spatial multiplexing is used.

When the MODO_PLP is set to 1, the information of MODO_MIMO is valid. When the information of MODE_MIMO is set to 0, data is transmitted without using a phase change made to recoded signals (or precoded signals that have changed base bands). When the MODE_MIMO information is set to 1, data is transmitted using a phase change made to recoded signals (or precoded signals that have changed base bands).

When the information of MODO_PLP is set to 1 and the information of MODO_MIMO is set to 0, the information of PATRON_MIMO No. 1 is valid. As such, when PATRON_MIMO No. 1 information is set to 00, data is transmitted using space-time block codes. When the information of

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PATRON_MIMO No. 1 is set to 01, data is transmitted using the fixed precoding matrix No. 1 for weighting. When the information in PATRON_MIMO No. 1 is set to 10, data is transmitted using the fixed precoding matrix No. 2 for weighting. (The precoding matrix # 1 and the precoding matrix # 2 are different matrices). When PATRON_MIMO # 1 information is set to 11, data is transmitted using MIMO spatial multiplexing.

When the information of MODO_PLP is set to 1 and the information of MODO_MIMO is set to 1, the information in PATRON_MIMO No. 2 is valid. When the PATRON_MIMO # 2 information is set to 00, data is transmitted using version # 1 of a phase change in precoded (or precoded and changed) signals. When the PATRON_MIMO # 2 information is set to 01, data is transmitted using version # 2 of a phase change in the precoded (or precoded signals that have changed base bands). When the information in PATR6n_MIMO No. 2 is set to 10, data is transmitted using version # 3 of a phase change in the precoded (or precoded signals that have changed base bands). When PATR6n_MIMO No. 2 information is set to 11, data is transmitted using version # 4 of a phase change in the precoded (or precoded signals that have changed base bands). Although the change in phase is carried out in four different versions No. 1 to 4, the following three approaches are possible, given two different methods No. A and No. B:

Phase changes made using method # A and made using method # B include identical and different changes.

Some phase change values are included in method # A but are not included in method # B;

Y

Multiple phase changes used in method # A are not included in method # B.

The control information listed in Table 3 and Table 4, above, is transmitted by the first and second signaling data. In such circumstances, there is no particular need to use PLPs to transmit control information.

As described above, select a transmission method that uses a multi-carrier method such as OFDM while it is identifiable as being different from the DVB-T2 standard, and in which a phase change is made in precoded signals (or precoded and changed) has the advantages of leading to better reception quality in the LOS environment and higher transmission speeds. Although the present invention describes possible transmission methods for carriers as they are MIMO of spatial multiplexing, MIMO using a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals , space-time block codes, and transmission methods that only transmit the flow s1, limitation is not intended in this way.

Also, although the description indicates that the diffuser selects one of the above-mentioned transmission methods, these are not the only transmission methods available for selection. Other options include:

MIMO that uses a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals, space-time block codes, and transmission methods that transmit only the flow s1;

MIMO that uses a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals, and space-time block codes;

MIMO using a fixed precoding matrix, a transmission method that performs a phase change in precoded (or precoded and changed) signals, and transmission methods that only transmit the s1 flow;

A transmission method that performs a phase change in precoded (or precoded and changed) signals, space-time block codes, and transmission methods that transmit only the flow s1;

MIMO that uses a fixed precoding matrix and a transmission method that performs a phase change in precoded signals (or precoded and changed);

A transmission method that performs a phase change in precoded signals (or precoded and changed) and space-time block codes; Y

A transmission method that performs a phase change in precoded (or precoded and changed) signals and transmission methods that transmit only the s1 stream.

As such, including a transmission method that performs a phase change in precoded signals (or in precoded and changed signals), the advantages of driving at higher data rates in the LOS environment and better reception quality are achieved. receiving device

At this point, since, as described above, the data S1 must be established for the P1 symbol, another configuration method is possible for the control information (with respect to the transmission method for each PLP) transmitted as the first and second signaling data, different from that of Table 3. For example, see Table 5, above.

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Table 5 differs from Table 3 in that setting the information of MODO_PLP to 11 is reserved. As such, when the transmission method for PLPs is as described in one of the previous examples, the number of bits that form the information of MODO_PLP as in the examples of Tables 3 and 5 can be made greater or lesser according to the transmission methods available for selection.

Similarly, for Table 4, when, for example, a MIMO method is used with a transmission method that does not support changing the phase of precoded (or precoded and changed) signals, the control information of MODO_MIMO is not necessary . Also, when, for example, MIMO schemes that use a fixed precoding matrix are not supported, then PATTERN_MIMO # 1 is not necessary. Also, when multiple precoding matrices are not necessary, 1-bit information can be used instead of 2-bit information. Additionally, two or more bits may be used when a plurality of precoding matrices are available.

The same principles apply to PATRON_MIMO No. 2 information. When transmission schemes do not require a plurality of methods to perform a phase change in precoded (or precoded and changed) signals, 1 bit information may be used instead of information of 2 bits. Additionally, two or more bits may be used when a plurality of phase change schemes are available.

Additionally, although the present embodiment describes a transmission device having two antennas, no limitation is intended in this regard. Control information can also be transmitted using more than two antennas. In such circumstances, the number of bits in each type of control information can be increased as required to perform transmission using four antennas. The transmission of control information of the above description in the P1 symbol and in the first and second serialization data also applies to such cases.

Although Figure 77 illustrates the frame configuration for the PLP symbol groups transmitted by the diffuser as being divided with respect to the time domain, the following variation is also possible.

Unlike Figure 77, Figure 79 illustrates an example of a method for arranging flow s1 and flow 2 of the symbols in the time-frequency domain, after the symbol p1 has been transmitted, the first and second signaling data, and the Common PLP.

In Figure 79, the symbols labeled No. 1 are the symbols of the symbol group of PLP No. 1 from Figure 77. Similarly, the symbols labeled No. 2 are the symbols of the group of PLP symbols # 2, the symbols labeled # 3 are the symbols of the symbol group of the PLP # 3, and the symbols labeled # 4 are the symbols of the symbol group of the PLP No. # 4, all from Figure 77. As in Figure 77, PLP # 1 is used to transmit data using a spatial multiplexing MIMO system as illustrated by Figure 23, or using a system of MIMO with a fixed precoding matrix. PLP # 2 is used to transmit data using only a modulated signal. PLP # 3 is used to transmit data using a transmission method in which a phase change is made in precoded (or precoded and changed) signals. PLP # 4 is used to transmit data using space-time block codes.

In Figure 79, when both s1 and s2 have a symbol on the same subcarrier at the same time indication, a symbol from each of the two flows is present in the common frequency. As explained in other embodiments, when a transmission method is used that involves performing a phase change in precoded (or precoded and changed) signals, the phase change can be performed in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

As described above, Figure 79 differs from Figure 77 in that the PLPs are divided with respect to the time domain. In addition, Figure 79 has a plurality of PLPs arranged with respect to time and frequency domains. That is, for example, the symbols of PLP No. 1 and PLP No. 2 are in the time indication 1, while the symbols of PLP No. 3 and PLP No. 4 are in the time indication 3 As such, PLP symbols that have a different Index (# X, where X = 1, 2, and so on) can be assigned to each symbol (consisting of a time indication and a subcarrier).

Although, for the sake of simplicity, Figure 79 lists only No. 1 and No. 2 in the time indication 1, no limitation is intended in this regard. The indices of PLP symbols other than No. 1 and No. 2 may be in the time indication No. 1. Additionally, the relationship between PLP indices and sub-carriers in the time indication 1 is not limited. to that illustrated by Figure 79. The indices of any PLP symbol can be assigned to any subcarrier. The same applies to other time indications, in which the indices of any PLP symbol can be assigned to them.

Unlike Figure 77, Figure 80 illustrates an example of a method for arranging the symbols the flow s1 and the flow s2 in the time-frequency domain, after the symbol P1 has been transmitted, the first and

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second signaling data, and the Common PLP. The characteristic feature of Figure 80 is that, supposing that a plurality of antennas is used for transmission is the basis of the PLP transmission method, then transmission using only the flow 1 is not an option for the T2 frame.

Accordingly, in Figure 80, the PLP 8001 symbol group transmits data using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix. Also, the group of symbols 8002 of PLP # 2 transmits data using a transmission method that performs a phase change in precoded (or precoded and changed) signals. In addition, the group of symbols 8003 of PLP # 3 transmits data using space-time block code. The group of symbols of the PLPs following the group of symbols 8003 of the PLP # 3 transmits data using one of these methods, in particular using a spatial multiplexing MIMO system, or a MIMO system using a precoding matrix fixed, using a transmission method that performs a phase change in precoded (or precoded and changed) signals, or using space-time block codes.

Unlike Figure 79, Figure 81 illustrates an example of a method for arranging the symbols the flow s1 and the flow s2 in the time-frequency domain, after the symbol P1 has been transmitted, the first and second data of signaling, and the Common PLP.

In Figure 81, the symbols labeled # 1 are the symbols of the group of symbols of the PLP # 1 from Figure 80. Similarly, the symbols labeled # 2 are the symbols of the group of PLP symbols # 2, the symbols labeled # 3 are the symbols of the symbol group of the PLP # 3, and the symbols labeled # 4 are the symbols of the symbol group of the PLP No. .4, all from Figure 80. As in Figure 80, PLP # 1 is used to transmit data using a spatial multiplexing MIMO system as illustrated by Figure 23, or using a system of MIMO with a fixed precoding matrix. PLP # 2 is used to transmit data using a transmission method in which a phase change is made in precoded (or precoded and changed) signals. PLP # 3 is used to transmit data using space-time block codes.

In Figure 81, when both s1 and s2 have a symbol on the same subcarrier at the same time indication, a symbol from each of the two flows is present in the common frequency. As explained in other embodiments, when a transmission method is used that involves performing a phase change in precoded (or precoded and changed) signals, the phase change can be performed in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

As described above, Figure 81 differs from Figure 80 in that the PLPs are divided with respect to the time domain. In addition, Figure 81 has a plurality of PLPs arranged with respect to time and frequency domains. That is, for example, the PLP # 1 and PLP # 2 symbols are both in the time indication 1. As such, the PLP symbols having a different Index (# X, where X = 1, 2, and so on) can be assigned to each symbol (consisting of a time indication and a subcarrier).

Although, for the sake of simplicity, Figure 81 lists only No. 1 and No. 2 in the time indication 1, no limitation is intended in this regard. The indices of PLP symbols other than No. 1 and No. 2 may be in the time indication No. 1. Additionally, the relationship between PLP indices and sub-carriers in the time indication 1 is not limited. to that illustrated by Figure 81. The indices of any PLP symbol can be assigned to any subcarrier. The same applies to other time indications, in which the indices of any PLP symbol can be assigned to them. On the other hand, a time indication can also have the symbols of only one PLP assigned to it, as is the case for the time indication 3. In other words, any assignment of PLP symbols in the time-frequency domain It is permissible.

Therefore, since the frame unit does not include PLP using transmission methods that only transmit the flow s1, the dynamic range of the signals received through the terminal can be restricted, which is likely to lead to improved received signal quality.

Although Figure 81 is described using examples to select one of transmitting data using a spatial multiplexing MIMO system, or a MIMO system using a fixed precoding matrix, transmitting data using a transmission method that performs a phase change in signals precoded (or precoded and changed), and transmitting data using space-time block codes, the selection of the transmission method is not limited as such. Other possibilities include:

select one to transmit data using a transmission method that performs a phase change in precoded (or precoded and changed) signals, transmit data using space-time block codes, and transmit data using a MIMO system using a matrix of fixed precoding;

select one to transmit data using a transmission method that performs a phase change in precoded (or precoded and changed) signals, and transmit data using space-time block codes; Y

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select one to transmit data using a transmission method that performs a phase change in precoded (or precoded and changed) signals and transmit data using a MIMO system using a fixed precoding matrix.

Although the above explanation is provided for a frame unit that has multiple PLPs, the following describes a frame unit that has only one PLP.

Figure 82 illustrates a sample frame configuration for the si stream and the s2 stream in the time-frequency domain where the frame unit has only one PLP.

Although Figure 82 indicates control symbols, these are equivalent to the Pi symbol described above and to the first and second signaling data. In Figure 82, the interval 1 is used to transmit a first frame unit, the interval 2 is used to transmit a second frame unit, the interval 3 is used to transmit a third frame unit and the interval 4 is used to transmit a fourth frame unit.

Additionally, the first frame unit in Figure 82 transmits the group of symbols 8101 of PLP No. 1-1. The transmission method is MIMO of spatial multiplexing or MIMO that uses a fixed precoding matrix.

The second frame unit transmits the group of symbols 8102 of the PLP No. 2-1. The transmission method is a transmission that uses a unique modulated signal.

The third frame unit transmits the group of symbols 8103 of the PLP No. 3-1. The transmission method is a transmission method that performs a phase change in precoded (or precoded and changed) signals.

The fourth frame unit transmits the group of symbols 8104 of the PLP No. 4-1. The transmission method is transmission using space-time block codes.

In Figure 82, when both s1 and s2 have a symbol on the same subcarrier at the same time indication, a symbol from each of the two flows is present in the common frequency. When a transmission method is used that involves making a phase change in precoded (or precoded and changed) signals, the phase change can be made in addition to the weighting using the precoding matrix (and, where applicable, after changing the baseband signal). Therefore, signals z1 and z2 are obtained. Signals z1 and z2 are each emitted by a different antenna.

As such, the transmission method can be set to take into account the data transmission rate and the terminal data reception rate for each PLP. This has the double advantage of allowing data transmission speed to be improved and high quality data reception to be ensured. The configuration method for the control information pertaining to the transmission method and so on! successively for the P1 symbol and for the first and second signaling data it can be provided as per Tables 2 to 5, thereby obtaining the same effects. The frame configuration of Figure 82 differs from that of Figures 77 and the like, where each frame unit has multiple PLPs and control information pertaining to the transmission method is required for each of the pLp. In Figure 82, each frame unit has only one PLP, and therefore, the only necessary control information is for the transmission information and so on! successively it belongs to that unique PLP.

The present embodiment describes a method applicable to a system that uses a DVB standard and in which the transmission method involves making a phase change in precoded (or precoded and changed) signals. The transmission method that involves making a phase change in precoded signals (or precoded signals that have changed base bands) is described herein. Although the present embodiment uses "control symbol" as an expression of the technique, this expression has no influence on the present invention.

The following describes the space-time block codes analyzed in the present description and included in the present embodiment.

Figure 94 illustrates the configuration of a modulated signal using space-time block codes. As shown, a space-time block encoder (9402) takes a baseband signal based on a modulated signal as input. For example, the space-time block encoder (9402) takes the symbol s1, the symbol s2, and so! successively as input. Next, as shown in Figure 94, space-time block codification is performed, resulting in z1 (9403A) that takes s1 as the symbol No. 0, -s2 * as the symbol No. 1, s3 as symbol # 2, -s4 * as symbol # 3, and so on! successively, and z2 (9403B) which takes s2 as the symbol No. 0, s1 * as the symbol No. 1, s4 as the symbol No. 2, s3 * as the symbol No. 3, and so! successively. At this point, the # X symbol of z1 and the # X symbol of z2 are simultaneous signals on a common frequency, each broadcast from a different antenna. The arrangement of symbols in the space-time block codes is not restricted to the time domain. A group of symbols can also be arranged in the frequency domain, or in the time-frequency domain, as required. Additionally, the method of

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space-time block coding of Figure 94 is provided as an example of space-time block codes. Other space-time block codes can also be applied to each realization analyzed in the present description.

[Realization E2]

The present embodiment describes a reception method and a reception device applicable to a communication system that uses the DVB-T2 standard when the transmission method described in embodiment E1 is used, which involves making a phase change in precoded signals ( or in precoded and changed).

Figure 86 illustrates a sample configuration for a receiving device in a terminal, for use when the diffuser transmission device from Figure 76 applies a transmission method that involves a phase change of precoded (or precoded) signals and changed). The components thereof that operate identically to those in Figure 7 use the same reference numbers between them.

In Figure 86, a detector and decoder of symbol P1 8601 receives the signal transmitted by the diffuser and takes the baseband signals 704_X and 704_Y as input, thus performing signal detection and frequency synchronization. The detector and decoder of symbol P1 8601 simultaneously obtains the control information included in the symbol P1 (performing demodulation and decoding of error correction therein) and issues the information of control of symbol P1 8602 as! obtained.

The OFDM 8600_X and 8600_Y related processors take the P1 8602 symbol control information as input and modify the OFDM signal processing method (such as the Fourier transform) accordingly. (This is possible since, as described in embodiment E1, the signals transmitted by the diffuser include information on the transmission method in the symbol P1). The processors related to OFDM 8600_X and 8600_Y a continuation emit the baseband signals 704_X and 704_Y after demodulation in the same according to the signal processing method.

A P2 8603 symbol demodulator (which can also be applied to the signaling PLP) takes the baseband signals 704_X and 704_Y and the P1 8602 symbol control information as input, performs signal processing and demodulation (which includes decoding of error correction) in accordance with the P1 symbol control information, and issues P2 8604 symbol control information.

A control information generator 8605 takes the symbol control information P1 8602 and the symbol control information P2 8604 as input, groups the control information (pertaining to reception operations), and issues a control signal 8606. Then, as shown in Figure 86, control signal 8606 is introduced into each component.

A 711 signal processor takes signals 706_1, 706_2, 708_1, 708_2, 704_X and 704_Y, as! as control signal 8606, as input, it performs demodulation and decoding in accordance with the information included in control signal 8606, and issues the data received 712. The information included in the control signal belongs to the transmission method, transmission method, modulation, error correction coding method and its coding rate, error correction code block size, and so on! successively used for each PLP.

When the transmission method used for PLPs is one of spatial multiplexing MIMO, MIMO that uses a fixed precoding matrix, and a transmission method that performs a phase change in precoded (or precoded and changed) signals, it is performed demodulation obtaining the received signals (base band) using the output of the channel estimators (705_1, 705_2, 707_1, and 707_2) and the ratio of the received signals (base band) to the transmission signals. When the transmission method involves making a phase change in precoded (or precoded and changed) signals, demodulation is performed using the output of the channel estimators (705_1, 705_2, 707_1, and 707_2), the received signals (base band ), and the relationship provided by Calculation 48 (formula 48).

Figure 87 illustrates a sample configuration for a receiving device in a terminal, for use when the diffuser transmission device from Figure 85 applies a transmission method that involves a phase change of precoded (or precoded) signals and changed). The components thereof that operate identically to those of Figures 7 and 86 use the same reference numbers between them.

The receiving device from Figure 87 differs from that in Figure 86 in that, although the latter receives data from signals according to DVB-T2 and other standards, the former receives data only from signals according to a different standard of DVB-T2.

In Figure 87, a symbol detector and decoder P1 8601 receives the signal transmitted by the diffuser and takes the baseband signals 704_X and 704_Y as input, thereby performing signal detection and frequency synchronization. The detector and decoder of symbol P1 8601 simultaneously obtains the

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control information included in the P1 symbol (performing demodulation and decoding error correction therein) and issues the P1 8602 as control symbol information! obtained.

The OFDM 8600_X and 8600_Y related processors take the P1 8602 symbol control information as input and modify the OFDM signal processing method accordingly. (This is possible since, as described in realization E1, the signals transmitted by the diffuser include transmission method information in the symbol P1). The processors related to OFDM 8600_X and 8600_Y a continuation emit the baseband signals 704_X and 704_Y after demodulation in the same according to the signal processing method.

A first and second serialization data demodulator 8701 (which can also be applied to the serialization PLP) takes the baseband signals 704_X and 704_Y and the symbol control information P1 8602 as input, performs signal processing and demodulation (including error correction decoding) in accordance with the symbol control information P1, and issues first and second signaling data control information 8702.

A control information generator 8605 takes the symbol control information P1 8602 and the first and second signaling data control information 8702 as input, groups the control information (pertaining to reception operations), and issues a signal control 8606. Next, as shown in Figure 86, control signal 8606 is introduced into each component.

A 711 signal processor takes signals 706_1, 706_2, 708_1, 708_2, 704_X and 704_Y, as! as control signal 8606, as input, it performs demodulation and decoding in accordance with the information included in control signal 8606, and issues the data received 712. The information included in the control signal belongs to the transmission method, transmission method, modulation, error correction codification method and its codification rate, error correction code block size, and so on! successively used for each PLP.

When the transmission method used for PLPs is one of spatial multiplexing MIMO, MIMO that uses a fixed precoding matrix, and a transmission method that performs a phase change in precoded (or precoded and changed) signals, it is performed demodulation obtaining the received signals (base band) using the output of the channel estimators (705_1, 705_2, 707_1, and 707_2) and the ratio of the received signals (base band) to the transmission signals. When the transmission method involves making a phase change in precoded (or precoded and changed) signals, demodulation is performed using the output of the channel estimators (705_1, 705_2, 707_1, and 707_2), the received signals (base band ), and the ratio provided by Calculation 48 (formula 48).

Figure 88 illustrates the configuration of a reception device for a terminal compatible with DVB-12 and with standards other than DVB-T2. The components thereof that operate identically to those of Figures 7 and 86 use the same reference numbers between them.

Figure 88 differs from Figures 86 and 87 in that the reception device of the first one is compatible with signals according to DVB-T2 as! as signals according to other standards. As such, the reception device includes a P2 symbol or demodulator of first and second signaling data 8801, to enable demodulation.

The P2 symbol or the first and second signaling data demodulator 8801 takes the baseband signals 704_X and 704_Y, as! as the symbol control information P1 8602, as input, uses the symbol control information P1 to determine if the received signals conform to DVB-T2 or another standard (for example, using the Table in a determination of this type), performs signal processing and demodulation (including error correction decoding), and issues control information 8802, which includes information indicating the standard to which the received signals conform. Otherwise, the operations are identical to those explained for Figures 86 and 87.

A reception device configured as described in the previous embodiment and receiving signals transmitted by a diffuser having the transmission device described in realization E1 provides superior data quality received by applying appropriate signal processing. In particular, when transmitted signals are received using a transmission method that implies a phase change applied to precoded (or precoded and changed) signals, the effectiveness of data transmission as! as signal quality both are improved in the LOS environment.

Although the present embodiment is described as a reception device compatible with the transmission method described in realization E1, and therefore having two antennas, no limitation is intended in this regard. The reception device may also have three or more antennas. In such cases, the quality of data reception can be further enhanced by enhancing the diversity gain. Also, the diffuser transmission device may have three or more transmission antennas and transmit three or more modulated signals.

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The same effects are attainable by increasing the number of antennas in the terminal receiving device accordingly. Alternatively, the receiving device may have only one antenna and apply maximum probability detection or approximate maximum probability detection. In such circumstances, the transmission method is preferably one that implies a phase change of precoded (or precoded and changed) signals.

The transmission method need not be limited to the specific methods explained in this description. Whenever precoding takes place and is preceded or followed by a change in phase, the same results are obtainable for the present realization.

[Realization E3]

The system of embodiment E1, which applies, to the DVB-T2 standard, a transmission method that involves a phase change made in precoded (or precoded and changed) signals, includes control information indicating the pilot insertion method in the pre-signaling information L1. The present embodiment describes a method for applying a transmission method that involves a phase change made in precoded signals (or precoded signals that have changed base bands) when the pilot insertion method in the pre-signaling information L1 is changed.

Figures 89A, 89B, 90A and 90B illustrate sample frame configurations according to the DVB-T2 standard in the time-frequency domain in which a common frequency region is used in a transmission method by which a plurality of Modulated signals are transmitted from a plurality of antennas. At this point, horizontal axes represent frequency, that is, carrier numbers, while vertical axes represent time. Figures 89A and 90A illustrate frame configurations for the modulated signal z1 while Figures 89B and 90B illustrate frame configurations for the modulated signal z2, both of which are as explained in the previous embodiments. Carrier numbers are labeled f0, f1, f2, and as! successively, while time is labeled t1, t2, t3 and ace! successively. Also, the symbols indicated on the same carrier and time are simultaneous to the symbols on a common frequency.

Figures 89A, 89B, 90A and 90B illustrate examples of pilot symbol insert positions according to DVB-T2. (In DVB-T2, eight pilot insertion modes are possible when a plurality of antennas are used to transmit a plurality of the modulated signals. Two of these are currently illustrated.) Two types of symbols are indicated, namely pilot symbols and data symbols. As described for other embodiments, when the transmission method involves making a phase change in precoded signals (or precoded signals that have changed base bands), or involves precoding using a fixed precoding matrix, then the data symbols of the modulated signal z1 are the symbols of the flow s1 and the flow s2 that have undergone weighting, since they are the data symbols of the modulated signal z2. (However, a phase change is also made when the transmission scheme involves doing this). When using space-time block codes or a spatial multiplexing MIMO system, the data symbols of the modulated signal z1 are the symbols of either the flow s1 or the flow s2, since they are the symbols of the modulated signal z2. In Figures 89A, 89B, 90A and 90B, the pilot symbols are labeled with an Index, which is either PP1 or PP2. These represent pilot symbols that use different configuration methods. As described above, eight pilot insertion methods are possible in DVB-T2 (which vary in terms of the frequency at which the pilot symbols are inserted in the frame), one of which is indicated by the diffuser. Figures 89A, 89B, 90A and 90B illustrate two methods of pilot insertion between these eight. As described in embodiment E1, the information pertaining to the pilot insertion method selected by the diffuser is transmitted to the receiving terminal as the pre-signaling data L1 in the symbol P2.

The following describes a method for applying a transmission method that involves a phase change made to precoded signals (or precoded signals that have changed base bands) that complements the pilot insertion method. In this example, the transmission method involves preparing ten different phase change values, namely F [0], F [1], F [2], F [3], F [4], F [5], F [6], F [7], F [8] and F [9]. Figures 91A and 91B illustrate the assignment of these phase change values in the time-frequency domain frame configuration of Figures 89A and 89B when a transmission method is applied that involves a phase change made in precoded signals ( or precoded and changed). Similarly, Figures 92A and 92B illustrate the assignment of these phase change values in the time-frequency domain frame configuration of Figures 90A and 90B when a transmission method is applied that involves a change in phase performed. in precoded signals (or precoded and changed). For example, Figure 91A illustrates the frame configuration of the modulated signal z1 while Figure 91B illustrates the frame configuration of the modulated signal z2. In both cases, the symbol # 1 in f1, t1 is a symbol in which frequency modification has been made using the phase change value F [1]. Therefore, in Figures 91A, 91B, 92A and 92B, a symbol on the carrier fx (where x = 0, 1, 2, and so on), time t (where y = 1, 2, 3, and as ! successively) No. Z is labeled to indicate that the frequency modification was made using the phase change value F [Z] in the fx, ty.

Obviously, the insertion method (insertion interval) for the frequency-time frame configuration of Figures 91A and 91B differs from that of Figures 92A and 92B. The transmission method in which it is performed

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A phase change in precoded signals (or precoded signals that have changed base bands) does not apply to pilot symbols. Therefore, although the same transmission method implies a phase change performed on the same synchronized precoded (or precoded and changed) signals (for which a different number of phase change values may have been prepared), the change value The phase assigned to a single symbol on a carrier and time given in Figures 91A and 91B may be different in Figures 92A and 92B. This becomes evident by reference to the drawings. For example, the symbol in f5, t2 in Figures 91A and 91B is labeled # 7, which indicates that a phase change has been made therein using the phase change value F [7]. On the other hand, the symbol et f5, t2 in Figures 92A and 92B is labeled # 8, which indicates that a phase change has been made therein using the phase change value F [8].

Therefore, although the diffuser transmits control information indicating the pilot pattern (pilot insertion method) in the pre-signaling information L1, when the transmission method selected by the diffuser method implies a phase change made in precoded signals (or precoded signals having changed base bands), the control information may additionally indicate the phase change value assignment method used in the selected method through the control information given by Table 3 or Table 4. Therefore, the terminal reception device that receives the modulated signals transmitted by the diffuser can determine the method of assigning phase change value by obtaining the control information indicated by the pilot pattern in the pre-data. L1 signaling. (This assumes that the transmission method selected by the diffuser for the PLP transmission of Table 3 or Table 4 is one that implies a phase change in precoded signals (or precoded signals that have changed base bands)). Although the above description uses the example of pre-signaling data L1, the control information described above may also be included in the first and second signaling data when, as described for Figure 83, no P2 symbols are used.

The following describes additional variant examples. Table 6 lists sample phase change patterns and corresponding modulation methods.

[Table 6]

 No. of modulated signals

 Modulation scheme Phase change pattern

 2

 # 1: QPSK, # 2: QPSK # 1: -, # 2: A

 2

 # 1: QPSK, # 2: 16-QAM # 1: -, # 2: B

 2

 # 1: 16-QAM, # 2: 16-QAM # 1: -, # 2: C

For example, as shown in Table 6, when the modulation method is indicated and the phase change values to be used in the transmission method have been determined which implies a phase change made in precoded signals (or precoded signals that they have changed base bands), the principles described above apply. That is, only transmit the control information pertaining to the pilot pattern, the PLP transmission method, and the modulation method is sufficient to enable the terminal receiving device to estimate the method of assigning phase change value (in the time-frequency domain) obtaining this control information. In Table 6, the phase change method column shows a hyphen to indicate that no phase change is made, and lists # A, # B, or # C to indicate phase change methods No. A, No. B, and No. C. Similarly, as shown in Table 1, when the modulation method and the error correction codification method are indicated and the values have been determined of phase change to be used in the transmission method that implies a phase change of precoded signals (or precoded signals that have changed base bands), then transmit only the control information pertaining to the pilot pattern, the PLP transmission method , the modulation method, and the error correction codes in the symbol P2 is sufficient to enable the terminal receiving device to estimate the method of assigning phase change value (in the time-frequency domain) obtaining this information of control.

However, unlike Table 1 and Table 6, two or more types of different transmission schemes can be selected that imply a change in phase made in precoded signals (or precoded signals that have changed base bands), despite the scheme of modulation that has been determined (for example, transmission schemes may have a different period (cycle), or use different phase change values). As an alternative, two or more different types of transmission schemes can be selected that imply a change in phase made in precoded signals (or precoded signals that have changed base bands), despite the modulation scheme and the error correction scheme that is has determined. Additionally, two or more different types of transmission schemes can be selected that imply a change in phase made in precoded signals (or precoded signals that have changed base bands), despite the error correction scheme that has been determined. In such cases, as shown in Table 4, the transmission scheme involves switching between phase change values. However, the information pertaining to the scheme of assigning the phase change values (in the time-frequency domain) can also be transmitted.

Table 7 lists examples of control information configuration for information pertaining to such allocation methods.

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[Table 7]

 PHASE FRAME PROVISION (2 bits)

 Control information

 00

 allocation scheme # 1

 01

 allocation scheme # 2

 10

 allocation scheme # 3

 eleven

 allocation scheme # 4

For example, assuming that the diffuser transmission device selects Figures 89A and 89B as the pilot pattern insertion method, and selects the transmission method A, which implies a phase change in precoded signals (or precoded signals having base bands changed). Therefore, the transmission device may select Figures 91A and 91B or Figures 93A and 93B as the method of assigning phase change value (in the time-frequency domain). For example, when the transmission device selects Figures 91A and 91B, the PHASE_THE PROVISION information in Table 7 is set to 00. When the transmission device selects Figures 93A and 93B, the PHASE_DRAFT_POSITION information is set to 01. As such, the receiving device can determine the method of assigning phase change value (in the time-frequency domain) by obtaining the control information of Table 7. The control information of Table 7 is also applicable to transmission by means of the symbol P2, and to the transmission by means of the first and second signaling data.

As described above, a phase change value assignment method for the transmission method that involves a phase change made in precoded (or precoded and changed) signals can be performed through the pilot insertion method. In addition, by reliably transmitting such allocation method information to the receiving party, the receiving device obtains the double benefit of improved data transmission efficiency and enhanced received signal quality.

Although the present embodiment describes a diffuser that uses two transmission signals, the same applies to diffusers that use a transmission device that has three or more transmission antennas that transmit three or more signals. The transmission method need not be limited to the specific methods explained in the present description. Whenever precoding takes place and is preceded or followed by a change in phase, the same results are obtainable for the present embodiment.

The pilot signal configuration method is not limited to the present embodiment. When the transmission method involves making a phase change in precoded (or precoded and changed) signals, the receiving device only needs to implement the ratio provided by Calculation 48 (formula 48) (for example, the receiving device can know the pilot pattern signals transmitted by the transmission device in advance). This applies to all the embodiments analyzed in the present description.

The transmission devices belonging to the present invention, as illustrated by Figures 3, 4, 12, 13, 51, 52, 67, 70, 76, 85 and so! successively they transmit two modulated signals, in particular the modulated signal No. 1 and the modulated signal No. 2, in two different transmission antennas. The average transmission power of modulated signals No. 1 and No. 2 can be freely set. For example, when the two modulated signals each have a different average transmission power, the transmission power control technology used in wireless transmission systems can be applied to them. Therefore, the average transmission power of modulated signals # 1 and # 2 may differ. In such circumstances, transmission power control can be applied to the baseband signals (for example, when mapping using the modulation method), or it can be performed by a power amplifier immediately before the antenna.

(Regarding Delay Q Clclico)

The following describes the application of the Clical Delay Q mentioned throughout this disclosure. Non-patent Bibliography 10 describes the overall concept of Q-Delay. The following describes a specific example of a generation method for signals s1 and s2 when using Clone Q Delay.

Figure 95 illustrates an example of a signal point arrangement in the I-Q plane when the modulation method is 16-QAM. As shown, when the input bits are b0, b1, b2 and b3, the bits take either a value of 0000 or a value of 1111. For example, when bits b0, b1, b2 and b3 are to be expressed as 0000, then signal point 9501 of Figure 95 is selected, a phase component value is taken based on signal point 9501 as the phase component of the baseband signal, and a component value is taken quadrature based on signal point 9501 as the quadrature component of the baseband signal. When bits b0, b1, b2 and b3 are to be expressed as a different value, the phase component and the quadrature component of the baseband signal are generated in a similar manner.

Figure 96 illustrates a sample configuration of a signal generator to generate the modulated signals

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s1 (t) (where t is time) (as an alternative, s1 (f), where f is frequency) and s2 (t) (as an alternative, s2 (f)) from data (binary) when the cyclic Q delay is applied .

A mapper 9602 takes data 9601 and a control signal 9606 as input, and performs mapping according to the modulation method of the control signal 9606. For example, when 16-QAM is selected as the modulation method, mapping is performed as illustrated in Figure 95. The continuation mapper emits a 9603_A phase component and a 9603_B quadrature component for the mapped baseband signal. No limitation is intended for the modulation method to be 16-QAM, and the operations are similar for other modulation methods.

At this point, the data at time 1 corresponding to bits b0, b1, b2 and b3 from Figure 95 are indicated respectively as b01, b11, b21 and b31. Mapper 9602 issues the phase component I1 and quadrature component Q1 for the baseband signal at time 1, according to data b0, b1, b2 and b3 at time 1. Similarly, another mapper 9602 emits the component in phase I2 and the quadrature component Q2 and as! successively for the baseband signal at time 2.

A memory and signal changer 9604 take the phase component 9603_A and the quadrature component 9603_B of the baseband signal as input and, in accordance with a control signal 9606, store the phase component 9603_A and the quadrature component 9603_B of the baseband signal, the signals change, and emit the modulated signal s1 (t) (9605_A) and the modulated signal s2 (t) (9605_B). The generation method for the modulated signals s1 (t) and s2 (t) is described in detail below.

As described elsewhere in the disclosure, precoding and phase change is performed on the modulated signal s1 (t) and s2 (t). At this point, as described elsewhere, signal processing that involves phase change, power change, signal change and so! successively it can be applied at any stage. Therefore, the modulated signals r1 (t) and r2 (t), obtained respectively by applying precoding and phase change to the modulated signals s1 (t) and s2 (t), are transmitted using the same (common) band of frequency in the same (common) time.

Although the above description is provided with respect to the time domain, s1 (t) and s2 (t) can be known as s1 (f) and s2 (f) (where f is the frequency of (sub-) carrier) when used a multicarrier transmission scheme such as OFDM. In contrast to the modulated signals s1 (f) and s2 (f), the modulated signals r1 (f) and r2 (f) obtained using a precoding scheme in which the precoding matrix is changed regularly are transmitted therein ( common) time (r1 (f) and r2 (f) being, of course) signals of the same frequency band). Also, as described above, s1 (t) and s2 (t) can be treated as s1 (t, f) and s2 (t, f).

The following describes the generation method for the modulated signals s1 (t) and s2 (t). Figures 97A, 97B, and 97C illustrate a first example of a generation method for s1 (t) and s2 (t) when a cyclic Q delay is used.

The portion (a) of Figure 97 indicates the phase component and the quadrature component of the baseband signal obtained by the mapper 9602 of Figure 96. As shown in Figure 87A and as described with reference to the mapper 9602 of Figure 96, the mapper 9602 issues the phase component and the quadrature component of the baseband signal so that the phase component I1 and the quadrature component Q1 take place at time 1, the phase component I2 and the quadrature component Q2 take place at time 2, the phase component I3 and the quadrature component Q3 take place at time 3, and so! successively.

Portion (b) of Figure 97 illustrates a sample set of phase components and quadrature components for the baseband signal when change is made by memory and signal changer 9604 of Figure 96. As shown, pairs of quadrature components are changed at each of time 1 and time 2, time 3 and time 4, and time 5 and time 6 (i.e. time 2i + 1 and time 2i + 2, i being a positive integer other than zero) so that, for example, the components at time 1 and t2 are changed.

Therefore, since the signal change is not performed in the phase component of the baseband signal, the order thereof is such that the phase component I1 takes place at time 1, the phase component I2 has place at time 2, the baseband signal I3 takes place at time 3, and so on! successively.

Next, a signal change is made in the pairs of the quadrature components for the baseband signal. Therefore, the quadrature component Q2 takes place at time 1, the quadrature component Q1 takes place at time 2, the quadrature component Q4 takes place at time 3, the quadrature component Q3 takes place at time 4, and so on! successively.

Portion (c) of Figure 97 indicates a sample configuration for the modulated signals s1 (t) and s2 (t) before precoding, when the scheme applied involves precoding and phase change. For example, as shown in portion (c), the baseband signal generated in portion (b) is assigned alternately to s1 (t) and s2 (t). Therefore, the first interval of s1 (t) takes (I1, Q2) and the first interval of s2 (t) takes (I2, Q1).

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Similarly, the second interval of s1 (t) takes (I3, Q4) and the second interval of s2 (t) takes (I4, Q3). This continues in a similar way.

Although Figure 97 describes an example with reference to the time domain, the same applies to the frequency domain (exactly as described above). In such cases, the descriptions belong to s1 (f) and 2 (f).

Next, the modulated signals r1 (t) and r2 (t) precoded from interval N and changed in phase are obtained after applying the precoding and phase change to the modulated signals s1 (t) and s2 (t) of interval N This point is described elsewhere in this disclosure.

Figure 98 illustrates a configuration that differs from that of Figure 96 and is used to obtain the interval N s1 (t) and s2 (t) from Figures 97A to 97C. The mapper 9802 takes data and a control signal 9804 as input and, according to the modulation method of the control signal 9804, for example, performs mapping taking into account the change from Figure 97, generates a mapped signal (i.e., the phase components and the quadrature components of the baseband signal) and generates the modulated signal s1 (t) (9803_A) and the modulated signal s2 (t) (9803_B) from the mapped signal. The modulated signal (s 1 (t) (9803_A) is identical to the modulated signal 9605_A from Figure 96, and the modulated signal s2 (t) (9803_B) is identical to the modulated signal 9605_B from Figure 6 This is as indicated in portion (c) of Figure 97. Accordingly, the first interval of the modulated signal s1 (t) (9803_A) takes (I1, Q2), the first interval of the modulated signal s2 ( t) (9803_B) takes (I2, Q1), the second interval of the modulated signal s1 (t) (9803_A) takes (I3, Q4), the second interval of the modulated signal s2 (t) (9803_B) takes (I4 , Q3), and so on.

The generation method for the first interval (I1, Q2) of the modulated signal s1 (t) (9803_A) and the first interval (I2, Q1) of the modulated signal s2 (t) (9803_B) using mapper 9802 from of Figure 98 is described below, as a supplement.

Data 9801 indicated in Figure 98 are composed of time data 1 b01, b11, b21, b31 and time data 2 b02, b12, b22, b32. Mapper 9802 of Figure 98 generates I1, Q1, I2 and Q2 as described above using data b01, b11, b21, b31 and b02, b12, b22 and b32. Therefore, the mapper 9802 of Figure 98 can generate the modulated signals s1 (t) and s2 (t) from I1, Q1, I2 and Q2.

Figure 99 illustrates a configuration that differs from that of Figures 96 and 98 and is used to obtain the interval N s1 (t) and s2 (t) from Figures 97A to 97C. The mapper 9901_A takes data 9801 and a control signal 9804 as input and, according to the method of modulation of the control signal 9804, for example, performs mapping taking into account the change from Figure 97, generates a signal mapped (i.e. the phase components and quadrature components of the baseband signal) and generates a modulated signal s1 (t) (9803_A) from the mapped signal. Similarly, the mapper 9901_B takes data 9801 and a control signal 9804 as input and, in accordance with the modulation method of the control signal 9804, for example, performs mapping taking into account the change from Figure 97 , generates a mapped signal (that is, the phase components and the quadrature components of the baseband signal) and generates a modulated signal s2 (t) (9803_B) from the mapped signal.

The data 9801 entered to the mapper 9901_A and the data 9801 introduced to the mapper 9901_B are, of course, identical data. The modulated signal s1 (t) (9803_A) is identical to the modulated signal 9605_A from Figure 96, and the modulated signal s2 (t) (9803_B) is identical to the modulated signal 9605_B from Figure 6. This It is as indicated in portion (c) of Figure 97.

Therefore, the first interval of the modulated signal s1 (t) (9803_A) takes (I1, Q2), the first interval of the modulated signal s2 (t) (9803_B) takes (I2, Q1), the second interval of the modulated signal s1 (t) (9803_A) takes (I3, Q4), the second interval of the modulated signal s2 (t) (9803_B) takes (I4, Q3), and so! successively.

The generation method for the first interval (I1, Q2) of the modulated signal s1 (t) (9803_A) by the mapper 9901_A from Figure 99 is described below, as a supplement. Data 9901 indicated in Figure 99 are composed of time data 1 b01, b11, b21, b31 and time data 2 b02, b12, b22, b32. Mapper 9901_A of Figure 99 generates I1 and Q2 as described above using data b01, b11, b21, b31 and b02, b12, b22 and b32. The mapper 9901_A of Figure 99 then generates the modulated signal s1 (t) from I1 and Q2.

The generation method for the first interval (I2, Q1) of the modulated signal s2 (t) (9803_B) by the mapper 9901_B from Figure 99 is described below. Data 9801 indicated in Figure 99 are composed of time data 1 b01, b11, b21, b31 and time data 2 b02, b12, b22, b32. Mapper 9901_B of Figure 99 generates I2 and Q1 as described above using data b01, b11, b21, b31 and b02, b12, b22 and b32. Therefore, mapper 9901_B of Figure 99 can generate the modulated signal s2 (t) from I2 and Q1.

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Next, Figures 100A to 100C illustrate a second example that differs from the generation method of s1 (t) and s2 (t) from Figures 97A to 97C, is provided for a case where the cyclic Q delay is used. In Figures 100A to 100C, the reference signs corresponding to elements found in Figures 97A to 97C are identical (ie, the phase component and the quadrature component of the baseband signal).

Portion (a) of Figure 100 indicates the phase component and the quadrature component of the baseband signal obtained by the mapper 9602 of Figure 96. Portion (a) of Figure 100 is identical to the portion ( a) of Figure 97. The explanations thereof are therefore omitted.

Portion (b) of Figure 100 illustrates the configuration of the phase component and the quadrature component of the baseband signals s1 (t) and s2 (t) before signal change. As shown, the baseband signal is assigned to s1 (t) at times 2i + 1, and is assigned to s2 (t) at times 2i + 2 (i being a positive integer other than zero).

Portion (c) of Figure 100 illustrates a sample set of phase components and quadrature components for the baseband signal when change is made by memory and the signal changer 9604 of Figure 96. The main point of The portion (c) of Figure 100 (and point of difference of the portion (c) of Figure 97) is that the signal change takes place in s1 (t) as well as in s2 (t).

Therefore, in contrast to the portion (b) of Figure 100, Q1 and Q3 of s1 (t) are changed in the portion (c) of Figure 100, such as Q5 and Q7. Also, in contrast to the portion (b) of Figure 100, Q2 and Q4 of s2 (t) are changed in the portion (c) of Figure 100, such as Q6 and Q8.

Therefore, the first interval of s1 (t) has a component in phase I1 and a quadrature component Q3, and the first interval of s2 (t) has a component in phase I2 and a quadrature component Q4. Also, the second interval of s1 (t) has a component in phase I3 and a quadrature component Q1, and the second interval of s2 (t) has a component in phase I4 and a quadrature component Q4. The third and fourth intervals are as indicated in portion (c) of Figure 100, and the subsequent intervals are similar.

Next, the modulated signals r1 (t) and r2 (t) precoded from interval N and changed in phase are obtained after applying the precoding and phase change to the modulated signals s1 (t) and s2 (t) of interval N This point is described elsewhere in this disclosure.

Figure 101 illustrates a configuration that differs from that of Figure 96 and is used to obtain the interval N s1 (t) and s2 (t) from Figures 100A to 100C. The mapper 9802 takes data 9801 and a control signal 9804 as input and, according to the modulation method of the control signal 9804, for example, performs mapping taking into account the change from Figure 100, generates a signal mapped (i.e. the phase components and quadrature components of the baseband signal) and generates the modulated signal s1 (t) (9803_A) and the modulated signal s2 (t) (9803_B) from the mapped signal. The modulated signal s1 (t) (9803_A) is identical to the modulated signal 9605_A from Figure 96, and the modulated signal s2 (t) (9803_B) is identical to the modulated signal 9605_B from Figure 6. This it is as indicated in portion (c) of Figure 100. Therefore, the first interval of the modulated signal s1 (t) (9803_A) takes (I1, Q3), the first interval of the modulated signal s2 (t) (9803_B) takes (I2, Q4), the second interval of the modulated signal s1 (t) (9803_A) takes (I3, Q1), the second interval of the modulated signal s2 (t) (9803_B) takes (I4, Q2 ), and so on.

The generation method for the first interval (I1, Q3) of the modulated signal s1 (t) (9803_A), the first interval (I2, Q4) of the modulated signal s2 (t) (9803_B), the second interval (I3 , Q1) of the modulated signal s1 (t) (9803_A), and the second interval (I4, Q2) of the modulated signal s2 (t) (9803_B) by the mapper 9802 from Figure 101 is described below, As a supplement.

The data 9801 indicated in Figure 101 are composed of time data 1 b01, b11, b21, b31, time data 2 b02, b12, b22, b32, time data 3 b03, b13, b23, b33, and data of time 4 b04, b14, b24, b34. Mapper 9802 of Figure 101 generates the aforementioned I1, Q1, I2, q2, I3, Q3, I4 and Q4 from data b01, b11, b21, b31, b02, b12, b22, b32, b03, b13, b23 , b33, b04, b14, b24, b34. Therefore, the mapper 9802 of Figure 101 can generate the modulated signals s1 (t) and s2 (t) from I1, Q1, I2, Q2, I3, Q3, I4 and Q4.

Figure 102 illustrates a configuration that differs from that of Figures 96 and 101 and is used to obtain the interval N of s1 (t) and s2 (t) from Figures 100A to 100C. A distributor 10201 takes data 9801 and control signal 9804 as input, distributes the data according to control signal 9804, and issues first data 10202_A and second data 10202_B. The mapper 9901_A takes the first data 10202_A and the control signal 9804 as input and, according to the modulation method of the control signal 9804, for example, performs mapping taking into account the change from Figure 100, generates a mapped signal (i.e., the phase components and the quadrature components of the baseband signal) and generates a modulated signal s1 (t) (9803_A) from the mapped signal. Similarly, the mapper 9901_B takes second data 10202_B and the control signal 9804 as input and, according to the modulation method of the control signal 9804, for example, performs mapping taking into account the change from the Figure 100, generates a mapped signal (i.e., the phase components and the quadrature components of the baseband signal) and generates a signal

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modulated s2 (t) (9803_B) from the mapped signal.

Therefore, the first interval of the modulated signal s1 (t) (9803_A) takes (I1, Q3), the first interval of the modulated signal s2 (t) (9803_B) takes (I2, Q4), the second interval of the modulated signal s1 (t) (9803_A) takes (I3, Q1), the second interval of the modulated signal s2 (t) (9803_B) takes (I4, Q2), and so! successively.

The generation method for the first interval (I1, Q3) of the modulated signal s1 (t) (9803_A) and the first interval (I3, Q1) of the modulated signal s2 (t) (9803_B) using the mapper 9901_A from of Figure 102 is described below, as a supplement. Data 9801 indicated in Figure 102 are composed of time data 1 b01, b11, b21, b31, time data 2 b02, b12, b22, b32, time data 3 b03, b13, b23, b33, and data of time 4 b04, b14, b24, b34. Distributor 10201 issues time data 1 b01, b11, b21, b31 and time data 3 b03, b13, b23, b33, such as the first data 10202_A, and issues time data 2 b02, b12, b22, b32 and time data 4 b04, b14, b24, b34 as the second data 10202_B. Mapper 9901_A of Figure 102 generates the first interval as (I1, Q3) and the second interval as (I3, Q1) from data b01, b11, b21, b31, b03, b13, b23, b33. The third interval and subsequent intervals are generated similarly.

The generation method for the first interval (I2, Q4) of the modulated signal s2 (t) (9803_B) and the second interval (I4, Q2) by the mapper 9901_B from Figure 102 is described below. Mapper 9901_B from Figure 102 generates the first interval as (I2, Q4) and the second interval as (I4, Q2) from time data 2 b02, b12, b22, b32 and time data 4 b04, b14, b24, b34. The third interval and subsequent intervals are generated similarly.

Although two methods using cyclic Q delay have been described above, when the signals are switched between pairs of intervals as for Figures 97A to 97C, the demodulator (detector) of the receiving device may restrict the number of candidate signal points. This has the advantage of reducing the calculation scale (circuit scale). Also, when the signals are changed in s1 (t) and s2 (t), as for Figures 100A to 100C, the demodulator (detector) of the receiving device finds a large number of candidate signal points. However, the time diversity gain (or frequency diversity gain when change is made with respect to the frequency domain) is available, which has the advantage of enabling further improvements to the quality of data reception.

Although the above description uses examples of a 16-QAM modulation method, no limitation is intended. The same applies to other modulation methods, such as QPSK, 8-QAM, 32-QAM, 64-QaM, 128-QAM, 256-QAM and as! successively.

Also, the cyclic Q delay method is not limited to the two schemes provided above. For example, either of the two schemes provided above may involve changing either the quadrature component or the phase component of the baseband signal. Also, although the above describes the change made in two stages (for example, changing the quadrature components of the baseband signal at times 1 and 2), the phase components and (or) the quadrature components of the signal Baseband can also be changed in a plurality of times. Therefore, when the phase components and quadrature components of the baseband signal are generated and cyclic Q delay is performed as in Figures 97a to 97C, then the phase component of the baseband signal after delay Q clical at time i is li, and the quadrature component of the baseband signal after the delay Q cyclic at time i is Qj (where itj). Alternatively, the phase-based component of the baseband signal after the time delay Q in time i is lj, and the quadrature component of the baseband signal after the time delay Q in time i is Qi (where itj ). As an alternative, the phase component of the baseband signal after the time delay Q cyclic is Ij, and the quadrature component of the baseband signal after the time delay Q cyclic is Qk (where itj , itk, jtk).

The precoding and phase change are then applied to the modulated signals s1 (t) (or s1 (f), or s1 (t, f)) and s2 (t) (or s2 (f) or s2 (t, f)) obtained by applying the cyclic Q delay described above. (At this point, as described elsewhere, signal processing that involves phase change, power change, signal change and so on can be applied at any stage). At this point, the method of applying precoding and phase change used in the modulated signal obtained by the cyclic Q delay can be any of the precoding and phase change methods described in the present disclosure.

[Realization F1]

In embodiment E1, the transmission method for performing a phase change in the precoded signals (or in precoded signals that have changed base bands) is applied to a broadcast system according to DVB-T2, and to a broadcast system according to another standard that is not DVB-T2. The present embodiment describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration to the E1 embodiment.

Figure 103A illustrates restrictions pertaining to single antenna transmission (SISO) and multi-stream transmission.

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antenna (MISO) in the DVB-T2 standard involving STBC. As described in the non-patent Bibliography 9, the DVB-T2 standard allows a selection between transmitting the entire frame through a single antenna and transmitting the entire frame through multiple antennas. When transmitted through multiple antennas, the P1 symbol is transmitted as an identical symbol through all the antennas. That is, the signaling data L1 carried by the symbol P2 and the entire PLP is transmitted through a selected one single antenna and multiple antennas.

Figure 103B indicates a future standard to be desired. In contrast to the DVB-T standard of the previous generation, a main feature of the DVB-T2 standard is that the transmission of parameters such as modulation method, codification rate, time intercalation depth, and so on! successively they are independently selected for each PLP. Therefore, it would be preferred to independently select whether each PLP is transmitted using a single antenna or multiple antennas. In addition, it would also be preferred to select whether the signaling data L1 is carried by the P2 symbol using a single antenna or multiple antennas.

As indicated in Figure 103B, a pilot symbol insertion position (pilot pattern) is a problem to consider to enable the presence of single antenna and multi-antenna transmission combined in a single frame. Non-patent Bibliography 9 explains that the pilot pattern for dispersed pilots (hereinafter, SP), which are a type of pilot symbol, difference between single antenna transmission (SISO) and multi-antenna transmission (MISO). Therefore, when a plurality of PLP # 1 and PLP # 2 are combined at the same time (as a common OFDM symbol) as shown in Figure 75, and when PLP # 1 is multi - antenna and PLP # 2 is of a single antenna as shown in Figure 77, the pilot pattern of SP is indefinable.

To solve this problem, Figure 104 illustrates a sub-frame based on the configuration of the transmission antenna. As shown, the frame includes a sub-frame for multi-antenna transmission (MISO, MIMO) and a sub-frame for single antenna transmission (SISO). Specifically, PLPs for MISO and / or MIMO (for example, Common PLP, PLP # 1) are assembled and a multi-antenna transmission subframe is provided, so that an SP pilot pattern is applicable multi-antenna transmission (when the number of transmission antennas is the same, a common SP pilot pattern for MISO and MIMO is usable). Meanwhile, PLPs for SISO (for example, PLP No. 2 to PLP No.) are assembled and a single antenna transmission subframe is provided so that a transmission SP pilot pattern is applicable of only antenna.

As indicated in Figure 78 and described in realization E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the Senalization PLP), then as shown in Figure 105, the sub-frame configuration according to the transmission antenna configuration can be provided.

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can provide a subframe configuration based on the transmission antenna configuration.

The sub-frame configuration described above based on the configuration of the transmission antenna enables the SP pilot pattern to be defined and allows the realization of a frame containing single antenna transmission and multi-antenna transmission combined.

A transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above is illustrated in Figures 76 and 85. However, in addition to the points described in embodiment E1, the configurator of frame 7610 also generates the subframe based on the configuration of the transmission antenna as described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A reception device corresponding to the transmission method and transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above is illustrated in Figures 86 to 88. However, in addition to the points described in embodiment E2, the sub-frame configuration based on the configuration of the transmission antenna enables channel fluctuation estimators (705_1, 705_2, 707_1, 707_2) to properly estimate channel fluctuations, even though the single antenna transmission and the multi-antenna transmission are combined in a single frame.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and transmission

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[Realization F2]

Embodiment F1 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration. In contrast to embodiment F1, the present embodiment describes a transmission frame configuration that enables the receiver to improve the channel estimate.

Figure 106 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the subframe configuration based on the transmission antenna configuration illustrated in Figure 104 of embodiment F1, the present embodiment describes a transmission frame configuration in which, for each subframe , a subframe start symbol is applied as the start OFDM symbol and a subframe close symbol is applied as the end OFDM symbol. However, a selection is possible as to whether or not the sub-frame start symbol and sub-frame close symbol are independently provided for each sub-frame, and as to whether the sub start symbol - frame and the sub-frame closing symbol are independent of each other in each sub-frame.

Figure 107 illustrates an example of a sub-frame start symbol and a sub-frame close symbol. As shown, the sub-frame start symbol and the sub-frame close symbol have higher SP density than other OFDM symbols. Specifically, SP in the subframe start symbol and subframe close symbol are located in all subcarrier positions where SP is possible.

Another sub-frame, a P2 symbol, or a P1 symbol takes place before the sub-frame start symbol and after the sub-frame close symbol. These use a different SP pilot pattern (symbol P1 does not use SP pilot pattern at all). Therefore, the process of estimating the transmission path (channel fluctuation) by means of the receiving device cannot perform an interpolation process that crosses different sub-frames in the time direction (i.e. the symbol direction of OFDM ). Therefore, when the SP pilot pattern is defined for the other OFDM symbols according to the same rule as the start and end OFDM symbols of the sub-frame, the interpolation accuracy of the start portion and the final portion of the subframe gets worse.

As shown in Figure 107, providing the sub-frame start symbol and the sub-frame close symbol allows the OfDm symbols to have SP in all sub-carrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 105 and described in embodiment F1, the serialization PLP (7801) is provided and the transmission is transmitted. control information required by the standard that is not the DVB-T2 standard (in whole or in part, that is, it is transmitted as the data of Post-Serialization L1 and the PLP of Senalization).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second data of signaling (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and F1, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The receiving device that corresponds to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and F1, the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol enables the estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2)

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more accurately estimate the fluctuations of the channel for the start portion and the end portion of the subframe, although the single antenna transmission and the multi-antenna transmission in the frame are combined.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

[Realization F3]

Embodiment F1 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration. The present embodiment describes a situation where the polarization of the transmitting antenna is taken into account, in addition to its configuration.

Figures 108A to 108D illustrate various types of broadcast networks. Figure 108A, in particular, illustrates a real DVB-T2 service network (SISO) currently used in the United Kingdom. The transmitting and receiving antennas are each unique antennas that have polarization V (vertical).

Figure 108B illustrates a distributed MISO system that employs an existing transmission antenna. In contrast to the SISO broadcast network using V polarization from Figure 108A, Figure 108B illustrates a MISO broadcast network using V polarization in which different transmission antennas are paired. This configuration also supports SISO.

Figure 108C illustrates a co-located MIMO configuration. In contrast to the SISO broadcast network using V polarization from Figure 108A, Figure 108C illustrates a MIMO broadcast network that uses V-H polarization in which an H (horizontal) antenna is added to serve as A transmitting or receiving antenna. This configuration supports MISO as! as SISO.

Figure 108D illustrates a configuration in which MISO distributed and MIMO co-located are combined.

As above, it is likely that future broadcast networks incorporate polarization in various ways. Preferably, each broadcast service provider can freely choose between these forms and implement them at any time. Therefore, future broadcast rules must support all forms of broadcast networks mentioned above.

Incidentally, as indicated by Figure 108D, the V / H transmission and the V / V transmission involve different channel characteristics, although the multi-antenna transmission takes place with the same number of transmission antennas. Therefore, when identical OFDM symbols are combined, a problem arises in which the receiver cannot perform channel estimation.

To solve this problem, Figure 109 illustrates a sub-frame based on the configuration of the transmission antenna (taking into account polarization). As shown in Figure 109, each frame is provided with a V / H-MIMO sub-frame, a V / V-MISO sub-frame and a V-SISO sub-frame. Specifically, the PLPs (for example, the Common PLPs) for V / H-MIMO meet and a V / H-MIMO sub-frame is provided, so that a pilot pattern of SP V / H-MIMO is applicable. Similarly, PLPs (for example PLP # 1) for V / V-MiSo meet and a V / V-MISO sub-frame is provided, so that a pilot pattern of SP V / V-MISO is applicable . Similarly, PLPs (for example, PLP No. 2 to PLP No. N) for V-SISO meet and a V-SISO subframe is provided, so that an SP pilot pattern is applicable V-SISO.

As indicated in Figure 78 and described in embodiment E1, when the serialization PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Serialization data and the Senalization PLP) then the sub-frame configuration can be provided according to the transmission antenna configuration (taking into account the polarization).

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can provide a sub-frame configuration based on the configuration of the transmission antenna (taking into account polarization).

The sub-frame configuration based on the transmission antenna configuration (taking into account the polarization) described above allows the receiver to perform channel estimation.

A transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (taking into account polarization) is illustrated in Figures 76 and 85. However, in addition to the points described in embodiment E1, frame configurator 7610 also generates the subframe based on the configuration of the transmission antenna as described above (having

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consider polarization).

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals thus processed are output as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A reception device corresponding to the transmission method and transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (taking into account polarization) is illustrated in Figures 86 to 88. However, in addition to the points described in embodiment E2, the sub-frame configuration based on the configuration of the transmission antenna (taking into account polarization) enables channel fluctuation estimators (705_1, 705_2, 707_1 , 707_2) properly estimate channel fluctuations, although transmission methods that use different polarizations in the frame are combined.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figure 109 illustrates a specific example of sub-frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization F4]

Embodiment F3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (taking into account polarization). In contrast to embodiment F3, the present embodiment describes a transmission frame configuration that enables the receiver to improve channel estimation.

Figure 110 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the sub-frame configuration based on the configuration of the transmission antenna (taking into account the polarization) illustrated in Figure 109 of the F3 embodiment, the present embodiment describes a transmission frame configuration in the that, for each sub-frame, a sub-frame start symbol is applied as the start OFDM symbol and a sub-frame close symbol is applied as the end OFDM symbol. However, a selection is possible as to whether the sub-frame start symbol and the sub-frame close symbol are provided or not independently provided for each sub-frame, and as to whether the sub start symbol -frame and the sub-frame closing symbol are independent of each other in each sub-frame.

As shown in Figure 107 and described in embodiment F2, providing the sub-frame start symbol and the sub-frame close symbol enables OFDM symbols to have SP in all subcarrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 105 and described in embodiment F1, the signaling PLP (7801) is provided and the transmission is transmitted. control information required by the standard that is not the DVB-T2 standard (in whole or in part, that is, it is transmitted as the L1 Post-Senalization data and the Senalization PLP).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second data of signaling (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and F3, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

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At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The receiving device that corresponds to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and F3, the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol enables estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2) more accurately estimate the channel fluctuations for the start portion and the end portion of the subframe, although transmission methods using different polarizations are combined in the plot.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figure 110 illustrates a specific example of a transmission frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

Embodiments F1 through F4, described above, analyze sub-frame configurations corresponding to a frame. The contents of embodiments F1 to F4 can be applied similarly to frame configurations corresponding to a superframe, to shorten frame configurations corresponding to a long frame, and the like.

Although applying embodiments F1 to F4 to a superframe is surely evident to those skilled in the art, a specific example is provided at this point. Specifically, the T2 frames and future extension frames (hereinafter FEF) that make up the super-frames of the DVB-T2 standard are considered to be the subframes described in each of the embodiments F1 to F4, and the data transmitted in one of the T2 frames or one of the FEF frames is set as if it were one of SISO and MISO and / or MIMO. Then, the data transmitted by each of the frames is collected in data for SISO and data for MISO and / or MIMO, and the frames are generated accordingly.

Also, a start symbol and a close symbol are inserted between the sub-frames analyzed in embodiments F1 to F4, to clarify the distinction between sub-frames. At a frame-by-frame level, a symbol P1, which is easy to identify by the receiver in the frame header, is inserted into the frame header, and is followed by a symbol P2 having SP density greater than that of other OFDM symbols. As such, the start symbol is of course unnecessary as is evident in the field to which this disclosure applies. However, that the symbol is unnecessary only means that the distinction between frames is sufficiently evident to make the symbol unnecessary. There is no harm in inserting the symbol as a way to further clarify and stabilize the transmission. In such circumstances, the start symbol is inserted into the frame header (before the P1 symbol).

[Realization G1]

The F1 embodiment describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration. The present embodiment describes a situation where the transmission power of the transmission antenna is taken into account, in addition to its configuration.

As indicated in the lower right portion of Figure 111, situations arise where the otherwise identical multi-antenna transmission may imply that the antennas have each different transmission power. Different transmission power leads to different channel characteristics. Therefore, when these are combined into identical OFDM symbols, a problem arises in which the receiver cannot perform channel estimation.

To solve this problem, Figure 111 illustrates a sub-frame configuration based on the transmission antenna configuration (taking into account the transmission power). As shown, the frame includes a sub-frame for transmission (MISO, MIMO) -pwrl multi-antenna, a sub-frame for transmission (MISO, MIMO) -pwr2 multi-antenna, and a sub-frame for transmission (SISO ) of the only antenna. Specifically, the PLPs between the MISO and / or MIMO PLPs for which the power of both transmitting antennas 1 and 2 is P / 2 (for example, Common PLP) meet and a transmission sub-frame is provided. pwr1 multi-antenna, so that a pilot pattern of transmission SP-pwr1 multi-antenna is applicable (a common SP pilot pattern for MISO is usable and

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MIMO when the amount of transmission antennas is equal and the transmission power is uniform). Also, the PLPs between the MISO and MIMO PLPs for which the power of the transmission antennas is 3P / 4 for antenna 1 and P / 4 for antenna 2 (for example, PLP # 1) meet and a multi-antenna-pwr2 transmission sub-frame is provided, so that a multi-antenna transmission-pwr2 transmission SP pilot pattern is applicable. Meanwhile, the PLPs for SISO (for example, PLP No. 2 to PLP No.) meet and a single antenna transmission sub-frame is provided so that a pilot pattern of transmission SP SP is applicable. Unique antenna However, in this example, the PLPs for SISO all have identical transmission power. When the transmission power differs, a different sub-frame is required for each value.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the Senalization PLP) then the sub-frame configuration can be provided according to the transmission antenna configuration (taking into account the transmission power).

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can provide a sub-frame configuration based on the configuration of the transmission antenna (taking into account the transmission power).

The sub-frame configuration based on the transmission antenna configuration (taking into account the transmission power) described above allows the receiver to perform channel estimation.

A transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (taking into account the transmission power) is illustrated in Figures 76 and 85. However, in addition to the points described in embodiment E1, frame configurator 7610 also generates the subframe based on the configuration of the transmission antenna as described above (taking into account the transmission power).

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A reception device corresponding to the transmission method and transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (taking into account the transmission power) is illustrated in Figures 86 to 88. However, in addition to the points described in embodiment E2, the sub-frame configuration based on the configuration of the transmission antenna (taking into account the transmission power) enables channel fluctuation estimators (705_1 , 705_2, 707_1, 707_2) properly estimate channel fluctuations, even though the transmission methods that use different transmission power are combined in the frame for the same multi-antenna transmission or single antenna transmission.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

Also, although Figure 111 illustrates an example of a sub-frame configuration, no limitation is intended. [Realization G2]

Embodiment G1 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (taking into account the transmission power). In contrast to embodiment G1, the present embodiment describes a transmission frame configuration that enables the receiver to improve the channel estimate.

Figure 112 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the sub-frame configuration based on the transmission antenna configuration (taking into account the transmission power) illustrated in Figure 110 of embodiment G1, the present embodiment describes a transmission frame configuration in which, for each sub-frame, a sub-frame start symbol is applied as the start OFDM symbol and a sub-frame close symbol is applied as the end OFDM symbol. However, a selection is possible as to whether or not the sub-frame start symbol and sub-frame close symbol are independently provided for each sub-frame, and as to whether the sub start symbol -frame and the sub-frame closing symbol are independent of each other in each sub-frame.

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As shown in Figure 107 and described in embodiment F2, providing the sub-frame start symbol and the sub-frame close symbol enables OFDM symbols to have SP in all sub-carrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 78 and described in embodiment E1, the signaling PLP (7801) is provided and the transmission is transmitted. control information required by the standard that is not the DVB-T2 standard (in whole or in part, that is, transmitted as the L1 post-signaling data and the signaling PLP).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second data of signaling (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and G1, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The receiving device that corresponds to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and G1, the transmission frame configuration using the subframe start symbol and the subframe close symbol enables the estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2) more accurately estimate the channel fluctuations for the start portion and the end portion of the sub-frame, despite the transmission methods that use different transmission power are combined in the frame for the same multi-antenna transmission or single antenna transmission.

Although the present embodiment is described as based on the DVB-T2 standard, no limitation is intended. The embodiment is also applicable to support a transmission method in which each antenna has a different transmission power, in otherwise identical multi-antenna transmission or single antenna transmission.

Also, although Figure 112 illustrates an example of a transmission frame configuration, no limitation is intended.

[Realization G3]

Embodiment F3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (taking into account polarization). The present embodiment describes a situation where the transmission power of the transmitting antenna (along with the polarization) is taken into account, in addition to the configuration thereof.

As indicated in the lower right portion of Figure 113, situations arise where the otherwise identical V / V-MISO transmission may imply that the antennas each have a different transmission power. Different transmission power leads to different channel characteristics. Therefore, when identical OFDM symbols are combined, a problem arises in which the receiver cannot perform channel estimation.

To solve this problem, Figure 113 illustrates a sub-frame configuration based on the configuration of the transmission antenna (taking into account the polarization and transmission power). As shown in Figure 113, each frame is provided with a V / H-MIMO sub-frame, a V-SISO sub-frame, a V / V-MISO-pwr1 sub-frame, and a V / sub-frame V-MISO-pwr2. Specifically, the PLPs between the V / V-MISO PLPs for which the power of both transmitting antennas 1 and 2 is P / 2 (for example, PLP # 2) are brought together and a sub-frame V is provided / V-MISO-pwr1, so that an SP pilot pattern V / V-MISO-pwr1 is applicable. Similarly, PLPs between V / V-MISo PLPs for which the power of transmitting antennas 1 and 2 is 3P / 4 and P / 4, respectively (for example, PLP # 3 to PLP n . ° N) meet and provide a sub-frame V / V-MISO-

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pwr2, so that a pilot pattern of SP V / V-MISO-pwr2 is applicable. Similarly, PLPs (for example, Common PLPs) for the V / H-MIMO meet and a V / H-MIMO sub-frame is provided, so that a V / H-MIMO SP pilot pattern is applicable. Also, the PLPs (for example, PLP # 1) for V-SISO meet and a V-SISO sub-frame is provided, so that a pilot pattern of SP V-SISO is applicable. However, these examples are provided for cases where only one PLP is available for V / H-MIMO and V-SISO. Additional and different sub-frames are necessary when PLPs are available for multiple different transmission powers.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the Senalization PLP) then the sub-frame configuration can be provided according to the transmission antenna configuration (taking into account the polarization and the transmission power) .

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can provide a sub-frame configuration based on the configuration of the transmission antenna (taking into account the polarization and transmission power).

The sub-frame configuration based on the transmission antenna configuration (taking into account the transmission power and polarization) described above allows the receiver to perform channel estimation.

A transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (taking into account the transmission power and polarization) is illustrated in Figures 76 and 85. However, in addition Of the points described in embodiment E1, frame configurator 7610 also generates the subframe based on the configuration of the transmission antenna as described above (taking into account the transmission power and polarization).

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A reception device corresponding to the transmission method and transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (taking into account the transmission power and polarization) is illustrated in Figures 86 to 88. However, in addition to the points described in embodiment E2, the sub-frame configuration based on the configuration of the transmission antenna (taking into account the transmission power and polarization) enables the estimators channel fluctuation (705_1, 705_2, 707_1, 707_2) properly estimate channel fluctuations, even though transmission methods that use different transmission power are combined in the frame for the same multi-antenna transmission or single transmission antenna using identical polarization.

Although the present embodiment is described as based on the DVB-T2 standard, no limitation is intended. The embodiment is also applicable to support a transmission method in which each antenna has a different transmission power, in otherwise identical multi-antenna transmission or transmission of a single antenna using identical polarization.

Also, although Figure 113 illustrates an example of a sub-frame configuration, no limitation is intended.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization G4]

Embodiment G3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (taking into account the transmission power and the polarization). In contrast to embodiment G3, the present embodiment describes a transmission frame configuration that enables the receiver to improve channel estimation.

Figure 114 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the sub-frame configuration based on the transmission antenna configuration (taking into account the transmission power and polarization) illustrated in Figure 113 of the G3 embodiment, the present embodiment describes a configuration of transmission frame in which, for each subframe, a subframe start symbol is applied as the start OFDM symbol and a symbol is applied

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Sub-frame closing as the OFDM termination symbol. However, a selection is possible as to whether or not the sub-frame start symbol and sub-frame close symbol are independently provided for each sub-frame, and as to whether the sub start symbol -frame and the sub-frame closing symbol are independent of each other in each sub-frame.

As shown in Figure 107 and described in embodiment F2, providing the sub-frame start symbol and the sub-frame close symbol enables OFDm symbols to have SP in all sub-carrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when; As illustrated in Figure 78 and described in embodiment E1, the serialization PLP (7801) is provided and the necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, transmitted as post-serialization data L1 and signaling PLP).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second data of signaling (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and G3, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The reception device corresponding to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and G3, the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol enables estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2) more accurately estimate the channel fluctuations for the start portion and the end portion of the subframe, despite the transmission methods that use different transmission power are combined in the frame for the same multi-antenna transmission or single antenna transmission using identical polarization.

Although the present embodiment is described as based on the DVB-T2 standard, no limitation is intended. The embodiment is also applicable to support a transmission method in which each antenna has a different transmission power, in otherwise identical multi-antenna transmission or transmission of a single antenna using identical polarization.

Also, although Figure 114 illustrates an example of a transmission frame configuration, no limitation is intended.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

Embodiments G1 to G4, described above, analyze sub-frame configurations corresponding to a frame. The contents of embodiments G1 to G4 can be applied similarly to frame configurations corresponding to a superframe, to shorten frame configurations corresponding to a long frame, and the like.

Although applying embodiments G1 to G4 to a superframe is surely evident to those skilled in the art, a specific example is provided at this point. Specifically, the T2 frames and future extension frames (hereafter FEF) that make up the super-frames of the DVB-T2 standard are considered to be the subframes described in each of the embodiments G1 to G4, and the data transmitted in one of the T2 frames or in one of the FEFs is set as if it were one of SISO and MISO and / or MIMO. Next, the data transmitted in each frame is one of: collected as SISO data in a frame generated for uniform transmission power

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when transmitted through the antenna; and collected as MISO and / or MIMO data in a frame generated for uniform transmission power when transmitted by the antenna.

Although embodiments G1 to G4 describe the start symbol and the closing symbol as being inserted to clarify the distinction between sub-frames, at a frame-by-frame level, a symbol P1, which is easy to identify by the receiver in The frame header is inserted into the frame header, and is followed by a P2 symbol that has SP density greater than that of other OFDM symbols. As such, the start symbol is of course unnecessary as is evident in the field to which this disclosure applies. However, that the symbol is unnecessary only means that the distinction between frames is sufficiently evident to make the symbol unnecessary. There is no harm in inserting the symbol as a way to further clarify and stabilize the transmission. In such circumstances, the start symbol is inserted into the frame header (before the P1 symbol).

[Realization H1]

Embodiment F1 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration. The present embodiment describes an additional arrangement of appropriate sub-frames in the frame.

Figure 115 illustrates a sub-frame configuration based on the transmission antenna configuration, in a particular case where the provision of appropriate sub-frames in the frame is taken into account. Comparison to Figure 104 of embodiment F1 reveals that the order of the multi-antenna transmission subframe (MISO, MIMO) and the single antenna transmission subframe (SISO) is changed. At this point, the P2 symbol carrying the L1 signaling data is for single antenna transmission (SISO), and the subsequent sub-frame is a single antenna transmission sub-frame (SISO) similar to the P2 symbol.

When the number of transmitting antennas is changed to half a frame, the power received for each antenna changes greatly instantaneously, for the receiver. At the instant when the received power changes, the automatic gain control process (hereinafter AGC) is difficult to change instantaneously in accordance with the change in power. Consequently, reception performance experiences deterioration.

The sub-frame configuration illustrated in Figure 104 of embodiment F1 implies a change in the number of transmission antennas at two points. However, in the sub-frame configuration of Figure 115, one of the changes in the number of transmission antennas has been deleted. Therefore, deterioration of reception performance is suppressed.

Also, in the sub-frame configuration of Figure 115, the sub-frame following the P2 symbol is a single antenna transmission sub-frame (SISO) similar to the P2 symbol. Therefore, SISO PLPs are transmitted in the remaining area of the P2 symbol. The sub-frame configuration illustrated in Figure 104 of embodiment F1 uses the remaining area of the P2 symbol as a fill, so that the multi-antenna transmission sub-frame (MISO, MIMO) takes place only as in the following symbol . As such, the tare that belongs to fill is susceptible to erasure.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the Senalization PLP) then as shown in Figure 116, the sub-frame configuration can be provided with an appropriate sub-frame arrangement in the frame.

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can provide a subframe configuration with an appropriate subframe arrangement in the frame.

Also, the sub-frame configuration of Figure 115 indicates an example in which the symbol P2 carrying the signaling data L1 is for single antenna transmission (SISO). However, when the P2 symbol is for multi-antenna transmission (MISo, MIMO), then as shown in Figure 117, the subsequent sub-frame is made in a multi-antenna transmission sub-frame (MISO, MIMO) similar to the P2 symbol. As such, identical results are obtained from those of the sub-frame configuration example shown in Figure 115.

In accordance with the provision of appropriate sub-frames in the frame based on the configuration of the transmission antenna described above, the frequency of changes in the amount of the transmission antennas is reduced, the deterioration of the reception performance is suppressed, and the tare that belongs to the filling is susceptible to erasure.

A transmission device configured to generate the subframe based on the antenna configuration of

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transmission (the appropriate subframe order) as described above is illustrated in Figures 76 and 85. However, in addition to the points described in realization E1, frame configurator 7610 also generates the subframe based on the transmission antenna configuration (the appropriate subframe order) as described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A reception device corresponding to the transmission method and transmission device configured to generate the sub-frame based on the configuration of the transmission antenna (the appropriate sub-frame order) as described above is illustrated in Figures 86. to 88. However, in addition to the points described in embodiment E2, in the structure of the subframe based on the configuration of the transmitting antenna (the appropriate subframe order), the processors related to OFDM (8600_X and 8600_Y) reduce the frequency of instantaneous changes to the received power, in particular for the received power that belongs to the AGC process.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

[Realization H2]

Embodiment H1 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order). In contrast to embodiment H1, the present embodiment describes a transmission frame configuration that enables the receiver to improve channel estimation.

Figure 118 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the sub-frame configuration based on the transmission antenna configuration (the appropriate sub-frame order) illustrated in Figure 115 of embodiment H1, the present embodiment describes a transmission frame configuration in which, for each sub-frame, a sub-frame start symbol is applied as the start OFDM symbol and a sub-frame close symbol is applied as the finalization OFDM symbol. However, a selection is possible as to whether or not the sub-frame start symbol and sub-frame close symbol are independently provided for each sub-frame, and as to whether the sub start symbol -frame and the sub-frame closing symbol are independent of each other in each sub-frame.

As shown in Figure 107 and described in embodiment F2, providing the sub-frame start symbol and the sub-frame close symbol enables OFDm symbols to have SP in all subcarrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 116 and described in embodiment H1, the signaling PLP (7801) is provided and the transmission is transmitted. control information required by the standard that is not the DVB-T2 standard (in whole or in part, that is, transmitted as the L1 post-signaling data and the signaling PLP).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second data of signaling (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H1, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the change in

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phase in the precoded signals (or precoded and changed) as indicated in Figures 6, 25 to 29 and 69. The so processed signals are output as the processed modulated signal 1 (7613_1) and the processed modulated signal 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The receiving device that corresponds to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and H1, the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol enables the estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2) more accurately estimate the channel fluctuations for the start portion and the end portion of the sub-frame, although the single antenna transmission and the transmission are combined multi-antenna in the plot.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

Also, although Figure 118 illustrates an example of a transmission frame configuration, no limitation is intended.

[Realization H3]

Embodiment F3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (taking into account polarization). The present embodiment describes an additional arrangement of appropriate sub-frames in the frame.

Figure 119 illustrates a sub-frame configuration based on the configuration of the transmission antenna (taking into account polarization), in a particular case where the provision of appropriate sub-frames in the frame is taken into account. Comparison to Figure 109 of embodiment F3 reveals that the order of the V / H-MIMO sub-frame and the V-SISO sub-frame is changed. At this point, the P2 symbol carrying the L1 signaling data is for V-SISO transmission, and the subsequent sub-frame is a V-SISO sub-frame similar to the P2 symbol.

When the number of transmission antennas is changed to half a frame, and when the polarization is changed for a constant amount of transmission antennas, the power received for each antenna changes greatly instantaneously, for the receiver. At the instant when the received power changes, the automatic gain control process (hereinafter AGC) is difficult to change instantly in accordance with the change in power. Consequently, reception performance experiences deterioration.

The sub-frame configuration illustrated in Figure 109 of embodiment F3 implies a change in the amount of transmission antennas, or in polarization, at three points. However, in the sub-frame configuration of Figure 119, one of the changes in the number of transmission antennas or in the polarization has been erased. Therefore, deterioration of reception performance is suppressed.

Also, in the sub-frame configuration of Figure 119, the subsequent sub-frame is a V-SISO sub-frame similar to the P2 symbol, and the remaining area of the P2 symbol can transmit the V-SISO PLPs. According to the sub-frame configuration indicated by Figure 109 of embodiment F3, the remaining area of the symbol P2 is used as a fill, so that a plurality of sub-frames V / H-MIMO take place only as that of the next symbol. As such, the tare that belongs to fill is susceptible to erasure.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the Senalization PLP) the sub-frame configuration can be provided with an appropriate sub-frame arrangement in the frame.

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can provide a subframe configuration with an appropriate subframe arrangement in the frame.

Also, the sub-frame configuration of Figure 119 indicates an example in which the symbol P2 carrying the signaling data L1 is for V-SISO. However, when the P2 symbol is, for example, for V / V-MISO transmission, then as shown in Figure 120, the subsequent sub-frame is a V / V-MISO sub-frame similar to the P2 symbol, and identical results are obtained to those of the sub-frame configuration from Figure 119.

In accordance with the provision of appropriate sub-frames in the frame based on the configuration of the transmission antenna (taking into account the polarization) described above, the frequency of changes in the amount of the transmission antennas or in the transmission is reduced. polarization, the deterioration of reception performance is suppressed, and

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the tare that belongs to the filling is susceptible to erasure.

A transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (taking into account the polarization and the appropriate sub-frame arrangement) is illustrated in Figures 76 and 85. Without However, in addition to the points described in embodiment E1, frame configurator 7610 also generates the subframe based on the configuration of the transmitting antenna (the appropriate subframe order, taking into account polarization) as it has been previously described.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A reception device corresponding to the transmission method and transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (the appropriate sub-frame order, taking into account polarization) It is illustrated in Figures 86 to 88. However, in addition to the points described in embodiment E2, in the structure of the sub-frame based on the configuration of the transmission antenna (the appropriate sub-frame order, taking into account polarization counts), OFDM-related processors (8600_X and 8600_Y) reduce the frequency of instantaneous changes to the received power, in particular for the received power that belongs to the AGC process.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figures 119 and 120 illustrate specific examples of sub-frame configurations, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization H4]

Embodiment H3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account polarization). In contrast to embodiment H3, the present embodiment describes a transmission frame configuration that enables the receiver to improve channel estimation.

Figure 121 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the sub-frame configuration based on the configuration of the transmission antenna (the appropriate sub-frame order, taking into account the polarization) illustrated in Figure 119 of embodiment H3, the present embodiment describes a transmission frame configuration in which, for each subframe, a sub-frame start symbol is applied as the start OFDM symbol and a sub-frame close symbol is applied as the OFDM symbol of finalization However, a selection is possible as to whether or not the sub-frame start symbol and sub-frame close symbol are independently provided for each sub-frame, and as to whether the sub start symbol -frame and the sub-frame closing symbol are independent of each other in each sub-frame.

As shown in Figure 107 and described in embodiment F2, providing the sub-frame start symbol and the sub-frame close symbol enables OFDm symbols to have SP in all subcarrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 78 and described in embodiment E1, the serialization PLP (7801) is provided and the transmission is transmitted. control information required by the standard that is not the DVB-T2 standard (in whole or in part, that is, transmitted as the L1 post-serialization data and the signaling PLP).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second data of signaling (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

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The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H3, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The receiving device that corresponds to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and H3, the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol enables estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2) more accurately estimate the channel fluctuations for the start portion and the end portion of the subframe, although transmission methods using different polarizations are combined in the plot.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figure 121 illustrates a specific example of a transmission frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization H5]

Embodiment H1 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order). The present embodiment describes an additional arrangement of appropriate sub-frames in the frame, taking into account the pattern of transmission power change.

Figure 122 illustrates two examples of transmission power change patterns for SISO and MISO / MIMO. Portion (a) of Figure 122 illustrates a sample pattern in which there is a difference in transmission power between SISO and MISO / MIMO. In this pattern, the transmission power P for SISO is assigned to the transmission antenna 1 only, while the transmission power P / 2 for MISO / MIMO is assigned to the transmission antennas 1 and 2.

Portion (b) of Figure 122 illustrates an example in which there is no difference in transmission power between SISO and MISO / MIMO. In this pattern, the transmission power of P for SISO is assigned to the transmission antenna 1 and the transmission power P / 4 to the transmission antenna 2, while equal transmission power is assigned for MISO / MIMO. For SISO, the transmission antenna 2 can transmit, for example, an identical signal to that transmitted by the transmission antenna 1. Alternatively, identical data can be transferred by the flows s1 (t), and s2 (t) (or by flows s1 (i) and s2 (i)), and a phase change can be applied as illustrated in Figures 6, 25 to 29 and 69. In such circumstances, signals as! processed are the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2), as shown in Figures 76 and 85.

In the example illustrated in the transmission power pattern of portion (a) of Figure 122, equal power is assigned for the transmission antennas 1 and 2 for MISO / MIMO. As such, this configuration can widely employ MISO / MIMO performance. However, when switching between SISO and MISO / MIMO, the transmission power of transmission antennas 1 and 2 also changes.

Conversely, in the example illustrated in the transmission power pattern of portion (b) of Figure 122, different power is assigned to the transmission antennas 1 and 2 for MISO / MIMO. As such, there is some deterioration in MISO / MIMO performance. However, when switching between SISO and MISO / MIMO, the transmission power of transmission antennas 1 and 2 can be preserved. Also, in an existing SISO transmission station, for SISO, the added power that accompanies the addition of the Transmission antenna 2 can be restricted to approximately 1 dB while maintaining the transmission power of an existing transmission antenna 1.

The following describes a situation where, as in portion (b) of Figure 122, there is no particular difference in transmission power.

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Figure 123 illustrates a sub-frame configuration from Figure 115 of embodiment H1. Obviously, the transmission power does not change despite the change from a SISO sub-frame to a MISO / MISO sub-frame.

In contrast to Figure 123, Figure 124 illustrates a situation where the SISO sub-frame and the MISO / MISO sub-frame are changed. Also, obviously there is no change in the transmission power in the frame. Therefore, arranging sub-frames that have no difference in transmission power before and after the sub-frame, regardless of whether the sub-frame is for SISO or for MISO / MIMO, is effective in preventing the process of AGC has an effect upon receipt. Therefore, the sub-frame order obtains a degree of freedom.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is to say, it is transmitted as the Post-Senalization data and the Senalization PLP) the sub-frame configuration can be provided with an appropriate sub-frame arrangement in the frame and also taking into account the power change pattern of transmission.

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can a subframe configuration is provided with an appropriate subframe arrangement in the frame, also taking into account the transmission power change pattern.

Figure 125 illustrates a situation where a sub-frame configuration is used from Figure 117 of embodiment H1. Obviously, the transmission power does not change despite the change from a MISO / MIMO sub-frame to a SISO sub-frame.

In contrast to Figure 125, Figure 126 illustrates a situation where the MISO / MISO sub-frame and the SISO sub-frame are changed. Also, obviously there is no change in the transmission power in the frame. Therefore, arranging sub-frames that have no difference in transmission power before and after the sub-frame, regardless of whether the sub-frame is for SISO or for MISO / MIMO, is effective in preventing the process of AGC has an effect upon receipt. Therefore, the sub-frame order obtains a degree of freedom.

In accordance with the provision of appropriate subframes in the frame based on the configuration of the transmission antenna (taking into account the transmission power change pattern) described above, the frequency of changes in transmission power is reduced , and the deterioration of reception performance is suppressed. Also, the sub-frame order obtains a degree of freedom.

A transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (an appropriate sub-frame order, taking into account the transmission power change pattern) is illustrated in the Figures 76 and 85. However, in addition to the points described in embodiment E1, frame configurator 7610 also generates the subframe based on the transmission antenna configuration (the appropriate subframe order, taking into account the transmission power change pattern) as described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A receiving device corresponding to the transmission device and transmission method configured to generate the sub-frame based on the configuration of the transmission antenna as described above (an appropriate sub-frame order, taking into account the pattern of transmission power change) is illustrated in Figures 86 to 88. However, in addition to the points described in embodiment E2, in the structure of the subframe based on the configuration of the transmission antenna (the order of sub -appropriate frame, taking into account the transmission power change pattern), OFDM-related processors (8600_X and 8600_Y) reduce the frequency of instantaneous changes to the received power, in particular for the received power that belongs to the AGC process .

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

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[Realization H6]

Embodiment H5 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the transmission power change pattern). In contrast to embodiment H5, the present embodiment describes a transmission frame configuration that enables the receiver to improve channel estimation.

Figure 127 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the sub-frame configuration based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the transmission power change pattern) illustrated in Figure 124 of the embodiment H5, the present embodiment describes a transmission frame configuration in which, for each sub-frame, a sub-frame start symbol is applied as the start OFDM symbol and a sub-frame close symbol is applied as the OFDM symbol of finalization. However, a selection is possible as to whether or not the sub-frame start symbol and sub-frame close symbol are independently provided for each sub-frame, and as to whether the sub start symbol -frame and the sub-frame closing symbol are independent of each other in each sub-frame.

As shown in Figure 107 and described in embodiment F2, providing the sub-frame start symbol and the sub-frame close symbol enables OFDM symbols to have SP in all sub-carrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 78 and described in embodiment E1, the serialization PLP (7801) is provided and the transmission is transmitted. control information required by the standard that is not the DVB-T2 standard (in whole or in part, that is, transmitted as the L1 post-serialization data and the signaling PLP).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second data of signaling (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H5, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The reception device corresponding to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and H5, the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol enables estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2) more accurately estimate the channel fluctuations for the start portion and the end portion of the subframe, even though the single antenna transmission and the transmission are combined multi-antenna in the plot.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

Also, although Figure 127 illustrates an example of a transmission frame configuration, no limitation is intended.

[Realization H7]

Embodiment H3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account polarization). The

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This embodiment describes an additional arrangement of appropriate sub-frames in the frame, taking into account the pattern of transmission power change.

Figure 128 shows examples of transmission power change patterns for SISO and MISO / MIMO (also taking into account polarization). Portion (a) of Figure 128 illustrates an example in which there is a difference in transmission power between SISO and MISO / MIMO. In this pattern, the transmission power P for SISO is assigned to the transmission antenna 1 only, while the transmission power P / 2 for MISO / MIMO is assigned to the transmission antennas 1 and 2.

Portion (b) of Figure 128 illustrates an example in which there is no difference in transmission power between SISO and MISO / MIMO. In this pattern, the transmission power of P for SISO is assigned to the transmission antenna 1 and the transmission power P / 4 to the transmission antenna 2, while equal transmission power is assigned for MISO / MIMO. For SISO, the transmission antenna 2 can transmit, for example, an identical signal to that transmitted by the transmission antenna 1. Alternatively, identical data can be transferred by the flows s1 (t), and s2 (t) (or by flows s1 (i) and s2 (i)), and a phase change can be applied as illustrated in Figures 6, 25 to 29 and 69. In such circumstances, signals as! processed are the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2), as shown in Figures 76 and 85.

In the example illustrated in the transmission power pattern of portion (a) of Figure 128, equal power is assigned for the transmission antennas 1 and 2 for MISO / MIMO. As such, this configuration can widely employ MISO / MIMO performance. However, when switching between SISO and MISO / MIMO, the transmission power of transmission antennas 1 and 2 also changes.

Conversely, in the example illustrated in the transmission power pattern of portion (b) of Figure 128, different power is assigned to the transmission antennas 1 and 2 for MISO / MIMO. As such, there is some deterioration in MISO / MIMO performance. However, when switching between SISO and MISO / MIMO, the transmission power of transmission antennas 1 and 2 can be preserved. Also, in an existing SISO transmission station, for SISO, the added power that accompanies the addition of the Transmission antenna 2 can be restricted to approximately 1 dB while maintaining the transmission power of an existing transmission antenna 1.

The following describes a situation where, as in portion (b) of Figure 128, there is no particular difference in transmission power.

Figure 129 illustrates a sub-frame configuration from Figure 119 of embodiment H3. Obviously, the transmission power and polarization do not change despite the change from a V-SISO sub-frame to a V / V-MISO sub-frame.

In contrast to Figure 129, Figure 130 illustrates a situation where the V-SISO sub-frame and the V / V-MISO sub-frame are changed. Subsequently, a change in polarization or transmission power in the frame takes place only when it is changed to a V / H-MIMO sub-frame. Therefore, having sub-frames that have no difference in polarization or transmission power before and after the sub-frame, regardless of whether the sub-frame is for SISO or for MISO / MIMO, is effective in preventing the AGC process has an effect upon receipt. Therefore, the sub-frame order obtains a degree of freedom.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the Senalization PLP) the sub-frame configuration can be provided with an appropriate sub-frame arrangement in the frame and also taking into account the power change pattern of transmission.

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can a subframe configuration is provided with an appropriate subframe arrangement in the frame, also taking into account the transmission power change pattern.

Also, Figure 131 illustrates a sub-frame configuration like that of Figure 120 of embodiment H3, in which the V / VMISO sub-frame and the V / H-MIMO sub-frame are changed. Therefore, a change in polarization or transmission power in the frame takes place only when it is changed to a V / H-MIMO sub-frame.

In contrast to Figure 131, Figure 132 illustrates a situation where the V / V-MISO sub-frame and the V-SISO sub-frame are changed. Subsequently, a change in polarization or transmission power in the frame takes place only when it is changed to a V / H-MIMO sub-frame. Therefore, providing sub-frames that have no difference in polarization and transmission power before and after the sub-frame, regardless of whether the sub-frame is for SISO or for MISO / MIMO, is effective in preventing the AGC process has an effect upon receipt. Therefore, the sub-frame order obtains a degree of freedom.

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In accordance with the provision of appropriate sub-frames in the frame based on the transmission antenna configuration (taking into account the transmission power change pattern and polarization) described above, the frequency of changes in the transmission is reduced. transmission and polarization power, and the deterioration of reception performance is suppressed. Also, the sub-frame order obtains a degree of freedom.

A transmission device configured to generate the sub-frame based on the configuration of the transmission antenna as described above (an appropriate sub-frame order, taking into account the pattern of transmission power change and polarization) is illustrated in Figures 76 and 85. However, in addition to the points described in realization E1, frame configurator 7610 also generates the subframe based on the configuration of the transmission antenna (the appropriate subframe order, taking into account the pattern of transmission power change and polarization) as described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

A receiving device corresponding to the transmission device and transmission method configured to generate the sub-frame based on the configuration of the transmission antenna as described above (an appropriate sub-frame order, taking into account the pattern of transmission power change and polarization) is illustrated in Figures 86 to 88. However, in addition to the points described in embodiment E2, in the structure of the subframe based on the configuration of the transmission antenna (the appropriate sub-frame order, taking into account the polarization and the transmission power change pattern), OFDM-related processors (8600_X and 8600_Y) reduce the frequency of instantaneous changes to the received power, in particular for the received power which belongs to the AGC process.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figures 129 to 132 illustrate specific examples of sub-frame configurations, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization H8]

Embodiment H7 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the pattern of transmission power change and polarization). In contrast to embodiment H7, the present embodiment describes a transmission frame configuration that enables the receiver to improve channel estimation.

Figure 133 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the subframe configuration based on the transmission antenna configuration (the appropriate subframe order, taking into account the pattern of transmission power change and polarization) illustrated in Figure 130 of embodiment H7, the present embodiment describes a transmission frame configuration in which, for each subframe, a subframe start symbol such as the start OFDM symbol is applied and a sub close symbol is applied. plot as the OFDM symbol of finalization. However, a selection is possible as to whether or not the sub-frame start symbol and sub-frame close symbol are independently provided for each sub-frame, and as to whether the sub start symbol - frame and the sub-frame closing symbol are independent of each other in each sub-frame.

As shown in Figure 107 and described in embodiment F2, providing the sub-frame start symbol and the sub-frame close symbol enables OFDM symbols to have SP in all sub-carrier positions where SP is possible, that is, in all subcarrier positions where the time direction interpolation process is applicable. Therefore, the precision of the interpolation of the start portion and the end portion of the sub-frame is improved.

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as illustrated in Figure 78 and described in embodiment E1, the serialization PLP (7801) is provided and the transmission is transmitted. control information required by the standard that is not the DVB-T2 standard (in whole or in part, that is, transmitted as the L1 post-serialization data and the signaling PLP).

The sub-frame start symbol and the sub-frame close symbol can also be provided when, as

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It is illustrated in Figure 83 and described in embodiment E1, the first signaling data (8301) and the second signaling data (8302) are used in the frame configuration.

The transmission frame configuration using the subframe start symbol and the subframe close symbol described above enables improvements to the channel estimation by the receiver.

The transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above is as described in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H7, frame configurator 7610 also generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The receiving device that corresponds to the transmission method and the transmission device that generates the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol described above are as described in the Figures 86 to 88. However, in addition to the points described in embodiments E2 and H7, the transmission frame configuration using the sub-frame start symbol and the sub-frame close symbol enables the estimators of channel fluctuation (705_1, 705_2, 707_1, 707_2) more accurately estimate the channel fluctuations for the start portion and the end portion of the subframe, although the single antenna transmission and the transmission are combined multi-antenna in the plot.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figure 133 illustrates an example of a transmission frame configuration, no limitation is intended.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization H9]

Embodiment H1 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order). In contrast to embodiment H1, the present embodiment describes a transmission frame configuration that particularly enables high-speed AGC tracking for the receiver in an instantaneous change in received power.

Figure 134 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, in contrast to the sub-frame configuration based on the transmission antenna configuration (the appropriate sub-frame order) illustrated by Figure 115 of embodiment H1, the present embodiment describes a transmission frame configuration in the that an AGC synchronization preamble is applied to the start symbol of the subframe in which the change in the amount of transmission antennas takes place.

The next four points are desired characteristics to produce the AGC synchronization preamble.

(1) A signal of short duration of time (to erase tare)

(2) A signal that includes components from as many frequency bands as possible, with respect to the sub-frame

(3) A signal in which the time domain amplitude is as uniform as possible (for high speed AGC synchronization

(4) A highly correlative signal (for high relative coincidence in a multi-route environment).

A buzz signal is a suggested example of a signal that satisfies the above. Specifically, in the buzz signal, the phase characteristics are represented as a quadratic function of frequency and time. However, the AGC synchronization preamble is not limited to a buzzer signal.

Through this AGC synchronization preamble, high-speed AGC monitoring is possible despite the change in the number of transmission antennas.

As indicated in Figure 116 and described in embodiment H1, when the signaling PLP is provided

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(7801) and the necessary control information is transmitted through the standard that is not the DVB-T2 standard (in whole or in part, that is, it is transmitted as the Post-Senalization data L1 and the Senalization PLP) AGC synchronization preamble can also be provided.

Also, as indicated by Figure 83 and described in realization E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can also provide the AGC synchronization preamble.

Also, the transmission frame configuration of Figure 134 is an example in which the symbol P2 carrying the signaling data L1 is for single antenna transmission (SISO). However, when the P2 symbol is for multi-antenna transmission (MISO, MIMO), identical results can also be obtained from the example of the transmission frame configuration from Figure 134. Specifically, in contrast to the configuration of subframe based on the transmission antenna configuration (the appropriate subframe order) illustrated by Figure 117 of embodiment H1, the present embodiment describes a transmission frame configuration in which a timing synchronization preamble is applied. AGC to the start symbol of the sub-frame in which the change in the amount of transmission antennas takes place. Figure 135 illustrates such a case.

In accordance with the transmission frame configuration used by the AGC synchronization preamble described above, AGC performance improvements are made available to the receiver.

The configuration of a transmission device that generates the transmission frame configuration using the AGC synchronization preamble described above is shown in Figures 76 and 85. However, in addition to the points analyzed in embodiments E1 and H1, the configurator of frame 7610 also generates the transmission frame configuration using the AGC synchronization preamble described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a reception device corresponding to the transmission method and the transmission device that generates the transmission frame configuration using the AGC synchronization preamble described above is shown in Figures 86 to 88. However, in addition to the points analyzed in embodiments E2 and H1, the transmission frame configuration using the AGC synchronization preamble enables high-speed AGC tracking through OFDM-related processors (8600_X and 8600_Y) when mixing single antenna transmission and transmission multi-antenna in the frame, and when the amount of transmission antennas has changed.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

Also, although Figures 134 and 135 illustrate examples of transmission frame configurations, no limitation is intended.

[Realization H10]

Embodiment H3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account polarization). In contrast to embodiment H3, the present embodiment describes a transmission frame configuration that particularly enables high speed AGC tracking for the receiver in an instantaneous change in received power.

Figure 136 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, in contrast to the sub-frame configuration based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the polarization) illustrated by Figure 119 of embodiment H3, the present embodiment describes a configuration of transmission frame in which an AGC synchronization preamble is applied to the OFDM symbol of the start of the sub-frame in which the change in the amount of transmission or polarization antennas takes place. As mentioned in embodiment H9, the AGC synchronization preamble may be a buzzer signal, although no limitation is intended.

Through this AGC synchronization preamble, high-speed AGC monitoring is possible despite the change in the amount of transmission antennas or the change in polarization.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP is provided

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(7801) and necessary control information is transmitted by the standard that is not the DVB-T2 standard (in whole or in part, that is, transmitted as Post-Senalization data and Serialization PLP) can also be provided AGC synchronization preamble.

Also, as indicated by Figure 83 and described in realization E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can also provide the AGC synchronization preamble.

Also, the transmission frame configuration of Figure 136 is an example in which the symbol P2 carrying the signaling data L1 is for V-SISO transmission. However, when the P2 symbol is for V / V-MISO transmission, for example, identical results can also be obtained from the example of the transmission frame configuration from Figure 136. Specifically, in contrast to the sub configuration -frame based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the polarization) illustrated by Figure 120 of embodiment H3, the present embodiment describes a transmission frame configuration in which it is applied an AGC synchronization preamble to the OFDM symbol of the start of the subframe in which the change in the number of transmitting or polarizing antennas takes place. Figure 137 illustrates such a case.

In accordance with the transmission frame configuration used by the AGC synchronization preamble described above, AGC performance improvements are made available to the receiver.

The configuration of a transmission device that generates the transmission frame configuration using the AGC synchronization preamble described above is shown in Figures 76 and 85. However, in addition to the points analyzed in embodiments E1 and H3, the configurator of frame 7610 also generates the transmission frame configuration using the AGC synchronization preamble described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a reception device corresponding to the transmission method and the transmission device that generates the transmission frame configuration using the AGC synchronization preamble described above is shown in Figures 86 to 88. However, in addition to the points analyzed in embodiments E2 and H3, the transmission frame configuration using the AGC synchronization preamble enables high-speed AGC tracking through OFDM-related processors (8600_X and 8600_Y) when mixing single antenna transmission and transmission multi-antenna in the frame, and when the amount of transmitting or polarization antennas have changed.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figures 136 and 137 illustrate a specific example of a transmission frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization H11]

Embodiment H5 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the transmission power change pattern). In contrast to embodiment H5, the present embodiment describes a transmission frame configuration that particularly enables high-speed AGC tracking for the receiver in an instantaneous change in received power.

Figures 123 to 126 indicate a sub-frame configuration based on the transmission antenna configuration from the H5 embodiment (the appropriate sub-frame order, taking into account the transmission power change pattern). These figures clearly indicate that no transmission power change takes place. Therefore, an AGC synchronization preamble as discussed in embodiments H9 and H10 is obviously not usable.

According to the above, the AGC synchronization preamble does not need to be applied when no transmission power change takes place. However, when a transmission power change takes place, the

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AGC synchronization preamble is applicable.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the PLP of Senalization) the AGC synchronization preamble does not need to be provided when no transmission power change takes place. However, when a transmission power change takes place, the AGC synchronization preamble is applicable.

Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same applies so that the AGC synchronization preamble does not need to be provided when there is no change in transmission power. However, when a transmission power change takes place, the AGC synchronization preamble is applicable.

The configuration of a transmission device that generates the transmission frame configuration as described above is shown in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H5, the frame configurator 7610 does not You need to apply the AGC synchronization preamble when there is no change in the number of transmission antennas. However, when a transmission power change takes place, the AGC synchronization preamble is applicable.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a reception device corresponding to the transmission method and the transmission device that generates the transmission frame configuration using the AGC synchronization preamble described above is shown in Figures 86 to 88. However, in addition to the points analyzed in embodiments E2 and H5, the transmission frame configuration using the AGC synchronization preamble enables high-speed AGC tracking through OFDM-related processors (8600_X and 8600_Y) when mixing single antenna transmission and transmission multi-antenna in the frame, and when the transmission power has changed.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

Also, although Figures 123 to 126 illustrate an example of a transmission frame configuration, no limitation is intended.

[Realization H 12]

Embodiment H7 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the pattern of transmission power change and polarization). In contrast to embodiment H7, the present embodiment describes a transmission frame configuration that particularly enables high-speed AGC monitoring for the receiver in an instantaneous change in received power.

Figure 138 illustrates a transmission frame configuration belonging to the present embodiment. Specifically, and in contrast to the sub-frame configuration based on the configuration of the transmission antenna (the appropriate sub-frame order, taking into account the pattern of transmission power change and polarization) illustrated in Figure 129 of embodiment H7, the present embodiment describes a transmission frame configuration in which, the AGC synchronization preamble is applied to the OFDM symbol of the start of the sub-frame in which the transmission power or polarization is changed . As mentioned in embodiment H9, the AGC synchronization preamble may be a buzzer signal, although no limitation is intended.

Through this AGC synchronization preamble, high-speed AGC monitoring is possible despite the change in transmission power or the change in polarization.

As indicated in Figure 78 and described in embodiment E1, when the signaling PLP (7801) is provided and necessary control information is transmitted by the standard other than the DVB-T2 standard (in whole or in part , that is, it is transmitted as the Post-Senalization data and the Senalization PLP) the AGC synchronization preamble can also be provided.

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Also, as indicated by Figure 83 and described in embodiment E1, when the frame configuration uses both the first signaling data (8301) and the second signaling data (8302), the same is applied so that it can also provide the AGC synchronization preamble.

Also, and in contrast to the sub-frame configuration based on the configuration of the transmission antenna (the appropriate sub-frame order, taking into account the pattern of transmission power change and polarization) illustrated in Figures 130 at 132 of embodiment H7, identical results can be achieved to those of the transmission frame configuration from Figure 138 by the transmission frame configuration in which the AGC synchronization preamble is applied to the OFDM symbol of beginning of the sub-frame in which the transmission power or polarization is changed. Figures 139 to 141 respectively illustrate each such situation.

In accordance with the transmission frame configuration used by the AGC synchronization preamble described above, AGC performance improvements are made available to the receiver.

The configuration of a transmission device that generates the transmission frame configuration using the AGC synchronization preamble described above is shown in Figures 76 and 85. However, in addition to the points analyzed in embodiments E1 and H7, the configurator of frame 7610 also generates the transmission frame configuration using the AGC synchronization preamble described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a reception device corresponding to the transmission method and the transmission device that generates the transmission frame configuration using the AGC synchronization preamble described above is shown in Figures 86 to 88. However, in addition to the points analyzed in embodiments E2 and H7, the transmission frame configuration using the AGC synchronization preamble makes it possible to track AGC at high speed by means of OFDM-related processors (8600_X and 8600_Y) when mixing single antenna transmission and transmission multi-antenna in the frame, and when the transmission power or polarization have changed.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figures 138 to 141 illustrate specific examples of transmission frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MlMo sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

Embodiments H1 to H12, described above, analyze sub-frame configurations that correspond to a frame. The contents of embodiments H1 to H12 can be applied similarly to frame configurations corresponding to a superframe, to shorten frame configurations corresponding to a long frame, and the like.

Although applying H1 to H12 embodiments to a superframe is surely evident to those skilled in the art, a specific example is provided at this point. Specifically, the T2 frames and future extension frames (hereinafter FEF) that make up the super-frames of the DVB-T2 standard are considered to be the subframes described in each of the embodiments H1 to H12, and the data transmitted in one of the T2 frames or in one of the FEFs is set as if it were one of SISO and MISO and / or MIMO. Next, the transmission device provides and transmits a control symbol and the subsequent data symbol that make up each frame so that each symbol is equal in terms of (1) the number of antennas, (2) the polarization characteristics of antenna, (3) the transmission power of the antenna, or (4) the polarization characteristics and the transmission power of the antenna, regardless of whether the frame is a SISO frame in which SISO data is collected, or It is a MISO / MIMO plot in which MISO and / or MIMO data is collected.

Also, a start symbol and a close symbol are inserted between the sub-frames as discussed in embodiments H1 to H12, to clarify the distinction between frames. At a frame-by-frame level, a symbol P1, which is easy to identify by the receiver in the frame header, is inserted into the frame header, and is followed by a symbol P2 having SP density greater than that of other OFDM symbols. As such, the start symbol is of course unnecessary as is evident in the field to which this disclosure applies. However, that the symbol is unnecessary only means that the distinction between frames is

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evident enough to make the symbol unnecessary. There is no harm in inserting the symbol as a way to further clarify and stabilize the transmission. In such circumstances, the start symbol is inserted into the frame header (before the P1 symbol).

[Realization J1]

As shown in Figure 103B of embodiment F1, the following is desirable for future standards:

• Select independently if each PLP is transmitted using single antenna transmission or multi-antenna transmission, and

• In addition, select whether the L1 signaling data is carried by the P2 symbol using single antenna transmission or multi-antenna transmission

To perform the above, the L1 signaling data that carries the control information is required again. In contrast to embodiment F1, the present embodiment describes the newly required signaling data L1.

As indicated by Table 2 of embodiment E1, in the DVB-T2 standard, the following is defined by the control information S1 (3 data bits) of the symbol P1:

• Single antenna transmission in the whole frame (T2_SISO)

• Multi-antenna transmission throughout the frame (T2_MISO)

• Signals not conforming to DVB-T2 (NO_T2)

To pass smoothly from the current standard to a future standard, DVB-T2 and future standards (for example, DVB-T3, DVB-T4) must enable transmission through time division multiplexing and identify what is using symbols P1. For example, DVB-T3 differs from DVB-T2 definitions in that, in order to satisfy the transmission method indicated in Figure 103B of embodiment F1, the control information S1 cannot indicate the amount of transmission antennas for all the plot.

To solve this problem, Figure 142A indicates the control information S1 (3 data bits). In addition to Table 2 of embodiment E1, DVB-T3 can further define, for example:

• Single antenna transmission for L1 signaling data (T3_L1_SISO)

• Multi-antenna transmission (MlSO) for signaling data L1 (T3_L1_MISO)

• Multi-antenna transmission (MIMO) for signaling data L1 (T3_L1_MIMO)

Next, as described by Tables 3 to 5 of embodiment E1, the signaling data L1 conveys an appropriate transmission method (SISO, MIMO, MISO) for each PLP.

Additionally, Figure 142B also indicates control information pertaining to the sub-frame configuration indicated by Figures 104 and 105 of embodiment F1. The L1 signaling data conveys the number of sub-frames (NUM_SUB-FRAME), the type of each sub-frame (TYPE_SUB-FRAME), the number of OFDM symbols for each sub-frame (SUB-FRAME_NUM_SIMBOLOS), and the SP pilot pattern for each sub-frame (SUB-TRAMA_PATRON_PILOTO). Therefore, the subframe configuration is indicated.

According to the control information described above S1 and the definitions of signaling data L1, the transmission of a single antenna and multi-antenna transmission in the frame can be combined.

The configuration of a transmission device that generates the previously described control information S1 and signaling data L1 is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and F1, the signal generator of P2 symbol 7605 (and the control symbol signal generator 8502), the control signal generator 7608, and the symbol inserter P1 7622 also generate the control information S1 and the signaling data L1 described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a receiving device corresponding to the transmission method and the transmission device that generates the control information described above S1 and signaling data L1 is indicated in Figures 86 to 88. However, the following points are added to the explanations of embodiments E2 and F1. Although the single antenna transmission and multi-antenna transmission are combined in the frame, the P1 8601 symbol detector and demodulator decodes the control information S1, and the transmission method for the L1 signaling data is obtained (SISO , MISO, MIMO). According to the transmission method obtained from

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of the signaling data L1, the signaling data L1 is decoded, and the symbol demodulator P2 8603 (which can also be applied to the signaling PLPs) obtains information pertaining to the transmission method (SISO, MISO, MIMO) for each PLP already sub-frame configuration. According to the signaling data L1 as! obtained, the PLPs are decoded by demodulation and channel selection.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

[Realization J2]

Embodiment F2 describes a transmission frame configuration using a sub-frame start symbol and a sub-frame close symbol. In contrast to embodiment F2, the present embodiment describes the newly required signaling data L1.

Figure 143 indicates control information pertaining to a sub-frame configuration using the sub-frame start symbol and the sub-frame close symbol as shown in Figure 106 of embodiment F2. The L1 signaling data conveys the amount of sub-frames (NUM_SUB-FRAME), the presence of a sub-frame start symbol in each sub-frame (SUB-FRAME_SYMBOL_START), and the presence of a sub closing symbol -frame in each sub-frame (SUB-TRAMA_SIMBOLO_CIERRE). Therefore, the sub-frame configuration using the sub-frame start symbol and the sub-frame close symbol is indicated.

According to the definition of the L1 signaling data described above, improvements to the precision of channel estimation for the receiver are possible.

The configuration of a transmission device that generates the signaling data L1 described above is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and F2, the signal signal generator P2 7605 (and the control symbol signal generator 8502) and the control signal generator 7608 also generate the signaling data L1 described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a receiving device corresponding to the transmission method and the transmission device that generates the control information described above S1 and signaling data L1 is indicated in Figures 86 to 88. However, the following points are added to the explanations of embodiments E2 and F2. The P2 8603 symbol demodulator (which can also be applied to the signaling PLP) decodes the L1 signaling data, and obtains information pertaining to the presence of the sub-frame start symbol and the sub-frame close symbol in Each sub-frame According to the signaling data L1 as! obtained, the channel fluctuation estimators (705_1, 705_2, 707_1, 707_2) employ the sub-frame start symbol and the sub-frame close symbol and can therefore more accurately estimate the channel fluctuation in the start and end portions of the sub-plot.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

[Realization J3]

The F1 embodiment describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration. In contrast to embodiment F1, the present embodiment describes L1 signaling data that make changes in the amount of transmission antennas easily detectable by the receiver.

In contrast to the sub-frame configuration shown in Figure 104 of embodiment F1, Figure 144 illustrates an additional point where the amount of transmission antennas is changed. According to Figure 144, the header of the multi-antenna transmission sub-frame (MISO, MIMO) and the header of the single antenna transmission sub-frame (SISO) are the points where the number of antennas is changed of transmission.

Figure 145A indicates corresponding L1 signaling data. The L1 signaling data (L1_TODAPLP_XIXO_MEZCLA), indicates that the L1 signaling data and all PLPs are as follows.

• when only SISO is available (= 0)

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• when only MISO / MIMO is available (= 1)

• when SISO and MISO / MIMO are both available (= 2)

Therefore, reading data "L1_TODAPLP_XIXO_MEZCLA = 0, 1" indicates that there is no change in the amount of transmission antennas.

For the sub-frame configuration shown in Figure 144, the data is read "TODAPLP_XIXO_MEZCLA = 2" and as such indicates the existence of a point at which the number of transmission antennas changes. In such a situation, according to the control information pertaining to the sub-frame shown in Figures 142B of realization J2, the positions of the points at which the amount of transmission antennas changes is known to be the header of the multi-antenna transmission sub-frame (MISO, MIMO) and the header of the single antenna transmission sub-frame (SISO).

The L1 signaling data described above (L1_TODAPLP_XIXO_MEZCLA) can also be carried by means of the control information S1 (3 data bits) of the P1 symbol. For example, situations are preferred where the transmission method for signaling data L1 (ie, SISO, MisO, MIMO) is uniquely selected. Figure 145B indicates the corresponding control information S1 (3 data bits). In addition to Table 2 of embodiment E1, DVB-T3 can, for example, further define:

• Single antenna transmission for L1 signaling data and all PLPs (T3_SISO_only)

• Multi-antenna transmission (MISO / MIMO) for L1 signaling data and all PLPs (T3_MIXO_only)

• A combination of single antenna transmission and multi-antenna transmission (MISO / MIMO) for L1 and PLP signaling data (T3_SISO AND MIXO_mixed)

Therefore, the data is read only T3_SISO_ or T3_MIXO_ to indicate that the number of transmission antennas does not change. For the sub-frame configuration shown in Figure 144, the data is read "T3_SISO Y MIXO_mzlado" and as such indicates the existence of a point at which the number of transmission antennas changes.

In accordance with the definitions provided above for signaling data L1 and control information S1, a change in the amount of transmission antennas is more easily detected by the receiver.

The configuration of a transmission device that generates the signaling data L1 described above and control information S1 is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and F1, the signal generator of control 7608, the symbol signal generator P2 7605 (and the control signal signal generator 8502) or the symbol inserter P1 7622 generates the signaling data L1 described above or control information S1.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a reception device corresponding to the transmission method and the transmission device that generates the previously described control information S1 and signaling data L1 is indicated in Figures 86 to 88. However, the following points are added to the explanations of embodiments E2 and F1. The P2 8603 symbol demodulator (which can also be applied to signaling PLPs) decodes the L1 signaling data, or alternatively the P1 8601 symbol detector and demodulator decodes the control information S1 to obtain information pertaining to the change in the number of transmission antennas. When changes in the number of transmission antennas take place, the P2 8603 symbol demodulator (which can also be applied to signaling PLPs) additionally obtains the control information belonging to the sub-frame indicated in Figure 142B and can by therefore detect (timing of) changes in the amount of transmission antennas. The (timing of) changes in the number of transmission antennas as! obtained can also particularly accelerate the AGC process through OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y).

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

[Realization J4]

Embodiment F3 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (taking into account polarization). In contrast to the F3 embodiment, the

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This embodiment describes L1 signaling data that make changes in the number of transmitting antennas or polarization easily detectable by the receiver.

In contrast to the sub-frame configuration shown in Figure 109 of realization F3, Figure 146 illustrates an additional point where the amount of transmission antennas or polarization is changed. According to Figure 146, the points at which the number of transmitting antennas or polarization change are the header of the V / H-MIMO sub-frame, the header of the V / V-MISO sub-frame and the V-SISO sub-frame header.

Figure 147A indicates corresponding L1 signaling data. The L1 signaling data (L1_TODAPLP_Y / Z_XIXO_MEZCLA) indicates that the L1 signaling data and all PLPs are as follows.

• when only SISO is available (= 0)

• when only V / V-MIXO is available (= 1)

• when only V / H-MIXO is available (= 2)

• when two or more of SISO are combined, V / V-MiXO and V / H-MIXO (= 3)

At this point, MIXO represents MISO and / or MIMO. Therefore, the reading of data "L1_TODAPLP_Y / Z_XIXO_MEzClA = 0, 1, 2" indicates that there is no change in the amount of transmission or polarization antennas.

For the sub-frame configuration shown in Figure 146, the data is read "TODAPLP_XIXO_Y / Z_MEZCLA = 3" and as such indicates the existence of a point at which the number of transmission antennas or polarization changes. In such circumstances, when the L1 signaling data includes control information pertaining to the sub-frame configuration, the positions of changes in the amount of transmission antennas or in the polarization are known to be in the sub-header. V / H-MIMO frame, the header of the V / V-MISO subframe and the V-SISO subframe header.

The L1 signaling data described above (L1_TODAPLP_Y / Z_XIXO_MEZCLA) can also be carried by the control information S1 (3 data bits) of the P1 symbol. For example, circumstances in which the transmission method is uniquely selected (VSISO, H-SlSO, V / V-MISO, V / H-MISO, V / V-MIMO, V / H-MIMO) are preferred for L1 signaling data. Figure 147B indicates the corresponding control information S1 (3 data bits). In addition to Table 2 of realization E1, DVB-T3 can further define, for example:

• Single antenna transmission for L1 signaling data and all PLPs (T3_SISO_only)

• Multi-antenna V / V transmission (MISO / MIMO) for L1 signaling data and all PLPs (T3_V / V-MIXO_only)

• Multi-antenna V / H transmission (MISO / MIMO) for L1 signaling data and all PLPs (T3_V / H- MIXO_only)

• A combination of single antenna transmission, multi-antenna V / V transmission (MISO / MIMO), and multi-antenna V / H transmission (MISO / MIMO) for L1 and PLP signaling data (T3_SISO YV / V-MIXO YV / H- MIXO_mixed)

Therefore, the data is read T3_SISO_ only, T3_V / V-MIXO_ only or T3_V / H-MIXO_ only to indicate that the number of transmission antennas and polarization do not change. For the subframe configuration shown in Figure 146, the data is read "T3_SlSO YV / V-MIXO YV / H-MIXO_mixed" and as such indicates the existence of a point at which the number of transmission antennas or The polarization changes.

In accordance with the definitions provided above for signaling data L1 and control information S1, a change in the amount of transmission antennas or polarization is more easily detected by the receiver.

The configuration of a transmission device that generates the signaling data L1 described above and control information S1 is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and F3, the signal generator of control 7608, the symbol signal generator P2 7605 (or the control signal signal generator 8502) or the symbol inserter P1 7622 generates the signaling data L1 described above or control information S1.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a reception device corresponding to the transmission method and the transmission device that generates the control information described above S1 and signaling data L1 is indicated in

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Figures 86 to 88. However, the following points are added to the explanations of embodiments E2 and F3. The symbol demodulator P2 8603 (which can also be applied to signaling PLPs) decodes the signaling data L1, or alternatively the symbol detector and demodulator P1 8601 decodes the control information S1, to obtain information pertaining to the change in the amount of transmission or polarization antennas. When there is a change in the number of transmission antennas or polarization, the P2 8603 symbol demodulator (which can also be applied to signaling PLPs) additionally obtains data belonging to the sub-frame configuration, and can detect ( timing of) the change in the amount of transmission antennas or polarization. The (timing of) changes in the number of transmitting antennas or in the polarization thus obtained can also particularly accelerate the AGC process by means of OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y).

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figure 146 illustrates a specific example of sub-frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization J5]

Embodiment H5 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the transmission power change pattern). In contrast to embodiment H5, the present embodiment describes signaling data L1 that make changes in transmission power easily detectable by the receiver.

In contrast to the sub-frame configuration shown in Figure 124 of embodiment H5, Figure 148A illustrates pattern 1 from the portion (a) of Figure 122 (where there is a difference in transmission power between SISO and MISO / MIMO) with an additional point where the transmission power is changed. According to Figure 148A, the header of the multi-antenna transmission sub-frame (MISO, MIMO) and the header of the single antenna transmission sub-frame (SISO) are the points where the transmission power is changed .

Also, Figure 124 of embodiment H5 illustrates pattern 2 from the portion (b) of Figure 122 (where there is no difference in transmission power between SISO and MISO / MIMO) with the addition of a point at which the power Transmission changes. Figure 148B illustrates such a case. Figure 148B clearly indicates that the transmission power does not change in the header of the multi-antenna transmission subframe (MISO, MIMO), or in the header of the single antenna transmission subframe (SISO) .

Portion (a) of Figure 149A indicates corresponding L1 signaling data. The L1 signaling data (L1_TODAPLP_XIXO_PWRDIF) indicates that the L1 signaling data and for all PLPs are as follows.

• when only SISO is available (= 0)

• when only MISO / MIMO is available (= 1)

• when SISO and MISO / MIMO are combined (no difference in transmission power) (= 2)

• when SISO and MISO / MIMO are combined (with a difference in transmission power) (= 3)

Therefore, reading data "L1_TODAPLP_XIXO_PWRDIF = 0, 1, 2" indicates that there is no change in transmission power. In the situation indicated by the sub-frame configuration in Figure 148B, the data is read "L1_TODAPLP_XIXO_PWRDIF = 2".

Also, the sub-frame configuration illustrated in Figure 148A indicates that the data is read "TODAPLP_XIXO_PWRDIF = 3", and therefore a change in transmission power takes place. In such a situation, according to the control information pertaining to the sub-frame shown in Figures 142B of embodiment J2, the positions of the points at which the transmission power changes are known to be the header of the multi-antenna transmission sub-frame (MISO, MIMO) and the header of the single antenna transmission sub-frame (SISO).

Portion (b) of Figure 149A indicates control information pertaining to the subframe. Comparison with Figure 142B of embodiment J1 reveals that the type of each sub-frame (SUB-FRAME TYPE) differs. Specifically, the data pertaining to the transmission power is included with the TIPO_SUB-FRAME data and transferred as the L1 signaling data. Therefore, the starting position of each sub-frame is identifiable as to whether or not the transmission power changes.

The signaling data L1 (L1_TODAPLP_XIXO_PWRDIF) of Figure 149A can also be carried by the control information S1 (3 data bits) of the symbol P1. For example, situations where the method is preferred

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of transmission for the L1 signaling data (ie, SISO, MISO, MIMO) is selected only. Figure 149B indicates the corresponding control information S1 (3 data bits). In addition to Table 2 of embodiment E1, DVB-T3 can, for example, further define:

• Single antenna transmission for L1 signaling data and all PLPs (T3_SISO_only)

• Multi-antenna transmission (MISO / MIMO) for L1 signaling data and all PLPs

(T3_MIXO_only)

• A combination of single antenna transmission and multi-antenna transmission (MISO / MIMO) combined (without

difference in transmission power) for signaling data L1 and PLP (T3_SISO Y

MIXO_mixed_nopwrdif)

• A combination of single antenna transmission and multi-antenna transmission (MISO / MIMO) combined (with a

difference in transmission power) for signaling data L1 and PLP (T3_SISO Y

MIXO_mezclado_pwrdif)

Therefore, reading data T3_SISO_unicamente, T3_MIXO_unicamente or T3_SISO Y

MIXO_mezclado_nopwrdif indicates that there is no change in transmission power. In the subframe configuration indicated by Figure 148B, the data is read T3_SISO Y MIXO_mezclado_nopwrdif.

For the sub-frame configuration shown in Figure 148A, the data is read "T3_SISO Y MIXO_mezclado_pwrdif" and as such indicates the existence of a point at which the transmission power changes.

In accordance with the definitions provided above for signaling data L1 and control information S1, a change in transmission power is more easily detected by the receiver.

The configuration of a transmission device that generates the signaling data L1 described above and control information S1 is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H5, the signal generator of control 7608, the symbol signal generator P2 7605 (or the control signal signal generator 8502) or the symbol inserter P1 7622 generates the signaling data L1 described above or control information S1.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a receiving device corresponding to the transmission method and the transmission device that generates the control information described above S1 and signaling data L1 is indicated in Figures 86 to 88. However, the following points are added to the explanations of embodiments E2 and H5. The P2 8603 symbol demodulator (which can also be applied to signaling PLPs) decodes the L1 signaling data, or alternatively the P1 8601 symbol detector and demodulator decodes the control information S1 to obtain information pertaining to the change in power of transmission. When there is a change in (the timing of) the transmission power, the P2 8603 symbol demodulator (which can also be applied to signaling PLPs) additionally obtains data belonging to the subframe configuration of the portion (b) in Figure 149A, and can detect (timing of) the change in transmission power. The (timing of) changes in the transmission power as! obtained can also particularly accelerate the AGC process through OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y).

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

[Realization J6]

Embodiment H7 describes a situation where a sub-frame configuration is applied based on the transmission antenna configuration (the appropriate sub-frame order, taking into account the pattern of transmission power change and polarization). In contrast to embodiment H7, the present embodiment describes signaling data L1 that makes changes in transmission power or polarization easily detectable by the receiver.

In contrast to the sub-frame configuration shown in Figure 130 of embodiment H7, Figure 150A illustrates pattern 1 from the portion (a) of Figure 128 (where there is a difference in transmission power between SISO and MISO / MIMO and polarization is taken into account) with an additional point where the transmission power or polarization is changed. According to Figure 150A, the points at which the transmission power or polarization change are the header of the V / V-MISO subframe, the V-SISO subframe header, and the header of the V / H-MIMO sub-frame.

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Also, Figure 130 of embodiment H7 illustrates pattern 2 from portion (b) of Figure 128 (where there is no difference in transmission power between SISO and MISO / MIMO and polarization is taken into account) with the addition of a point at which the transmission power or polarization changes. Figure 150B illustrates such a case. Figure 150B clearly indicates that the transmission power and polarization change only at the head of the V / H-MIMO subframe.

Figure 151A indicates corresponding L1 signaling data. The L1 signaling data (L1_TODAPLP_Y / Z_XIXO_PWRDIF) indicates that the L1 signaling data and all PLPs are as follows.

■ when only SISO is available (= 0)

■ when only V / V-MIXO is available (= 1)

■ when only V / H-MIXO is available (= 2)

■ when SISO and one of V / V-MIXO and V / H-MIXO are combined (no difference in transmission power) (= 3)

■ when combining SISO and one of V / V-MIXO and V / H-MIXO (with a difference in transmission power) (= 4)

■ When at least one of V / V-MIXO and V / H-MIXO is combined (= 5)

At this point, MIXO represents MISO and / or MIMO. Therefore, the data is read L1_TODAPLP_XIXO_PWRDIF = 0, 1, 2, 3 to indicate that there is no change in transmission power or polarization.

However, reading the TODAPLP_XIXO_PWRDIF = 4, 5 data indicates that a change in transmission power or polarization takes place. For the sub-frame configuration of Figures 150A and 150B, the data is read TODAPLP_XIXO_PWRDIF = 5. In such circumstances, when the L1 signaling data includes control information pertaining to the sub-frame configuration, the position of the point in which the transmission power or polarization changes is known to be at the head of the subframe.

The signaling data L1 (L1_TODAPLP_Y / Z_XIXO_PWRDIF) of Figure 151A can also be carried by the control information S1 (3 data bits) of the symbol P1. For example, circumstances in which the transmission method (V-SIsO, HSISO, V / V-MISO, V / H-MiSO, V / V-MIMO, V / H-MIMO) are preferred for the data are preferred. L1 signaling uniquely. Figure 151B indicates the corresponding control information S1 (3 data bits). In addition to Table 2 of embodiment E1, DVB-T3 can, for example, further define:

• Single antenna transmission for L1 signaling data and all PLPs (T3_SISO_only)

• Multi-antenna V / V transmission (MISO / MIMO) for L1 signaling data and all PLPs (T3_V / V- MIXO_only)

• Multi-antenna V / H transmission (MISO / MIMO) for L1 signaling data and all PLPs (T3_V / H- MIXO_only)

• A combination of unique and one V / V-MIXO and V / H-MIXO modulated signal transmission for L1 and PLP signaling data, with no difference in transmission power (T3_SISO Y V / V or V / H- MIXO_mixed_nopwrdif)

• The following is possible through the L1 and PLP signaling data:

Transmission with one of (1) at least two of V / V-MIXO and V / H-MIXO being combined, and (2) unique modulated signal transmission and one of V / V-MIXO and V / H-MIXO with a difference in transmission power (T3_V / V- YV / H-MIXO_mixed OR T3_SISO YV / V- or V / H-MIXO_mixed_pwrdif) Therefore, the data is read T3_SISO_ only, T3_V / V-MIXO_only, T3_V_H-MIXO YV / V or V / HMIXO_mezclado_nopwrdif to indicate that there is no change in transmission power or polarization.

For the sub-frame configurations shown in Figures 150A and 150B, the data is read T3_V / V- YV / H- MIXO_mixed OR T3_SISO YV / V- or V / H-MIXO_mixed_pwrdif, thus indicating that a change takes place in the transmission power or polarization.

In accordance with the definitions provided above for signaling data L1 and control information S1, a change in transmission power is more easily detected by the receiver.

The configuration of a transmission device that generates the signaling data L1 described above and control information S1 is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H7, the signal generator of control 7608, the symbol signal generator P2 7605 (or the control signal signal generator 8502) or the symbol inserter P1 7622 generates the signaling data L1 described above or control information S1.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal

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The configuration of a receiving device corresponding to the transmission method and the transmission device that generates the control information described above S1 and signaling data L1 is indicated in Figures 86 to 88. However, the following points are added to the explanations of embodiments E2 and H7. The P2 8603 symbol demodulator (which can also be applied to signaling PLPs) decodes the L1 signaling data, or alternatively the P1 8601 symbol detector and demodulator decodes the control information S1 to obtain information pertaining to the change in power of transmission or polarization. When there is a change in transmission power or polarization, the P2 8603 symbol demodulator (which can also be applied to signaling PLPs) additionally obtains data belonging to the sub-frame configuration, and can detect (timing of ) the change in transmission power or polarization. The (timing of) changes in the transmission power as! obtained can also particularly accelerate the AGC process through OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y).

Although the present realization is based on the DVB-T2 standard, no limitation is intended. The realization is applicable to any transmission method that supports different polarizations.

Also, although Figures 150A and 150B illustrate specific examples of sub-frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization J7]

Embodiment H9 describes a transmission frame configuration in which the AGC synchronization preamble is applied. In contrast to embodiment H9, the present embodiment describes the newly required signaling data L1.

Figure 152 illustrates control information pertaining to the transmission frame configuration applied by the AGC synchronization preamble, as shown in Figures 134 and 135 of embodiment H9. The presence of the AGC synchronization preamble (AGC_PREAMBULO) is transported using the L1 signaling data. Therefore, the transmission frame configuration that has the AGC synchronization preamble applied can be indicated.

In accordance with the L1 signaling data defined as described above, high-speed AGC tracking is available despite changes in the amount of transmission antennas.

The configuration of a transmission device that generates the signaling data L1 described above is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H9, the symbol signal generator P2 7605 (and the control symbol signal generator 8502) and the control signal generator 7608 also generate the signaling data L1 described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a receiving device corresponding to the transmission method and the transmission device that generates the L1 signaling data described above is indicated in Figures 86 to 88. However, the following points are added to the explanations of the embodiments E2 and H9. The P2 8603 symbol demodulator (which can also be applied to signaling PLPs) decodes the L1 signaling data, and thus obtains! data pertaining to the presence of the AGC synchronization preamble in each sub-frame. According to the signaling data L1 as! obtained, the OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y) make use of the AGC synchronization preamble and can therefore track AGC at high speed according to (the timing of) changes in the amount of transmission antennas

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

[Realization J8]

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Embodiment H10 describes a transmission frame configuration in which the AGC synchronization preamble is applied (taking into account polarization). In contrast to embodiment H10, the present embodiment describes the newly required signaling data L1.

Similar to embodiment J7, Figure 152 illustrates control information pertaining to the transmission frame configuration applied by the AGC synchronization preamble (and taking into account polarization), as shown in Figures 136 and 137 of embodiment H10. The presence of the AGC synchronization preamble (AGC_PREAMBULO) is transported using the L1 signaling data. Therefore, the transmission frame configuration that has the AGC synchronization preamble applied (and taking into account polarization) can be indicated.

In accordance with the L1 signaling data defined as described above, high-speed AGC tracking is available despite changes in the number of transmission antennas and polarization.

The configuration of a transmission device that generates the signaling data L1 described above is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H10, the signal signal generator P2 7605 (and the control symbol signal generator 8502) and the control signal generator 7608 also generate the signaling data L1 described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a reception device that corresponds to the transmission method and the transmission device that generates the L1 signaling data described above is indicated in Figures 86 to 88. However, the following points are added to the explanations of the embodiments E2 and H10. The P2 8603 symbol demodulator (which can also be applied to signaling PLPs) decodes the L1 signaling data, and thus obtains! data pertaining to the presence of the AGC synchronization preamble in each sub-frame. According to the signaling data L1 as! obtained, the OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y) make use of the AGC synchronization preamble and can therefore track AGC at high speed according to (the timing of) changes in the amount of transmission and polarization antennas.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission method that supports different polarizations.

Also, although Figures 136 and 137 illustrate specific examples of a transmission frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended thereto.

[Realization J9]

Embodiment H11 describes a transmission frame configuration in which the AGC synchronization preamble is applied (taking into account the transmission power change pattern). In contrast to embodiment H11, the present embodiment describes the newly required signaling data L1.

As shown in Figures 123 to 126, there is no change in transmission power in embodiment H11, and the AGC synchronization preamble need not apply to that example. However, when a transmission power change takes place, the AGC synchronization preamble is necessarily applied. Similar to embodiment J7, Figure 152 illustrates control information pertaining to the transmission frame configuration applied by the AGC synchronization preamble (and taking into account the transmission power change pattern). The presence of the AGC synchronization preamble (AGC_PREAMBULO) is transported using the L1 signaling data. Therefore, the transmission frame configuration that has the AGC synchronization preamble applied (and taking into account the transmission power change pattern) can be indicated.

In accordance with the L1 signaling data defined as described above, high-speed AGC tracking is available despite changes in transmission power.

The configuration of a transmission device that generates the L1 signaling data described above

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is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H11, the P2 7605 signal signal generator (and the control signal signal generator 8502) and the signal signal generator control 7608 also generates the L1 signaling data described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a receiving device corresponding to the transmission method and the transmission device that generates the L1 signaling data described above is indicated in Figures 86 to 88. However, the following points are added to the explanations of the embodiments E2 and H11. The P2 8603 symbol demodulator (which can also be applied to signaling PLPs) decodes the L1 signaling data, and thus obtains! data pertaining to the presence of the AGC synchronization preamble in each sub-frame. According to the signaling data L1 as! obtained, the OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y) make use of the AGC synchronization preamble and can therefore track AGC at high speed according to (the timing of) changes in the power of transmission.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The embodiment is applicable to any transmission and reception of a combination of single antenna transmission and multi-antenna transmission.

Also, although Figures 123 to 126 illustrate examples of a transmission frame configuration, no limitation is intended.

[Realization J10]

Embodiment H12 describes a transmission frame configuration in which the AGC synchronization preamble is applied (taking into account the polarization and the transmission power change pattern). In contrast to embodiment H12, the present embodiment describes the newly required signaling data L1.

Similar to embodiment J7, Figure 152 illustrates control information pertaining to the transmission frame configuration applied by the AGC synchronization preamble (and taking into account the polarization and the transmission power change pattern), as shown in Figures 138 to 141 of embodiment H12. The presence of the AGC synchronization preamble (AGC_PREAMBULO) is transported using the L1 signaling data. Therefore, the transmission frame configuration that has the AGC synchronization preamble applied (and taking into account the transmission power change pattern and polarization) can be indicated.

In accordance with the L1 signaling data defined as described above, high-speed AGC tracking is available despite changes in transmission power or polarization.

The configuration of a transmission device that generates the signaling data L1 described above is indicated in Figures 76 and 85. However, in addition to the points described in embodiments E1 and H12, the signal signal generator P2 7605 (or the control symbol signal generator 8502) and the control signal generator 7608 also generate the signaling data L1 described above.

At this point, the characteristic feature is that when the transmission method is selected to perform the phase change in precoded (or precoded and changed) signals, the signal processor 7612 makes the phase change in the precoded (or precoded) signals and changed) as indicated in Figures 6, 25 to 29 and 69. The signals as! processed are emitted as the modulated signal processed 1 (7613_1) and the modulated signal processed 2 (7613_2). However, this method of transmission does not necessarily need to be selected.

The configuration of a receiving device corresponding to the transmission method and the transmission device that generates the L1 signaling data described above is indicated in Figures 86 to 88. However, the following points are added to the explanations of the embodiments E2 and H12. The P2 8603 symbol demodulator (which can also be applied to signaling PLPs) decodes the L1 signaling data, and thus obtains! data pertaining to the presence of the AGC synchronization preamble in each sub-frame. According to the signaling data L1 as! obtained, the OFDM-related processors (8600_X, 8600_Y, 8601_X, 8601_Y) make use of the AGC synchronization preamble and can therefore track AGC at high speed according to (the timing of) changes in the power of transmission and polarization.

Although the present embodiment is based on the DVB-T2 standard, no limitation is intended. The realization is

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Also, although Figures 138 to 141 illustrate specific examples of a transmission frame configuration, no limitation is intended. The configuration may include any of an H-SISO sub-frame, a V / V-MIMO sub-frame and a V / H-MISO sub-frame.

Also, although polarization V and polarization H are described as the opposite polarizations, no limitation is intended.

Embodiments J1 to J10, described above, analyze sub-frame configurations that correspond to a frame. The content of embodiments J1 to J10 can be applied similarly to frame configurations corresponding to a superframe, to shorten frame configurations corresponding to a long frame, and the like.

Although applying embodiments J1 to J10 to a superframe is surely evident to those skilled in the art, a specific example is provided at this point. Specifically, the T2 frames and future extension frames (hereafter FEF) that make up the super-frames of the DVB-T2 standard are considered to be the subframes described in each of the embodiments J1 to J10, and the data transmitted in one of the T2 frames or one of the FEF frames is set as if it were one of SISO and MISO and / or MlMO. Then, the data transmitted by each of the frames is collected in data for SISO and data for MISO and / or MIMO, and the frames are generated accordingly.

Industrial applicability

The present invention is widely applicable to wireless systems that transmit different modulated signals from a plurality of antennas, such as an OFDM-MIMO system. Also, the present invention is also applicable to a wired system that has multiple connections (for example, a power line communication system, an optical fiber system, a digital subscriber line system and so on) when transmission of transmission is used. MIMO and the modulated signals described in this document are applied. Modulated signals can also be transmitted from a plurality of transmission locations.

List of reference signs

302A, 302B 304A, 304B 306A, 306B 314

308A, 308B 310A, 310B 312A, 312B 317A, 317B 402 404

encoders

interleavers

mappers

signal processing scheme information generator weighting composers wireless units antennas

phase changers

encoder

distributor

504 # 1, 504 # 2

505 # 1, 505 # 2 600

701_X, 701_Y

703_X, 703_Y

705_1

705_2

707_1

707_2

709

711

803

805A, 805B 807A, 807B 809A, 809B 811A, 811B 813A, 813B 815 819 901 903

1201A, 1201B

transmission antennas reception antennas weighting unit antennas

wireless units channel fluctuation estimator channel fluctuation estimator channel fluctuation estimator channel fluctuation estimator control information decoder signal processor internal MIMO detector interlocking logarithmic probability calculators

Logarithmic probability ratio calculator flexible input decoders / flexible output memory interleavers

coefficient generator

flexible input decoder / flexible output

distributor

OFDM related processors

1302A, 1302A 1304A, 1304B 1306A, 1306B 1308A, 1308B 5

Serial to parallel converters reorganizers

Fast Fourier Reverse Transform Units Wireless Units

Claims (4)

5 10 fifteen twenty 25 30 35 40 Four. Five fifty 55 60 65 1. A transmission device for Single-Input, Single-Output, SISO, Multiple-Input, Single-Output, MISO, and Multiple-Input, Multiple-Output, MIMO, comprising the transmission device: a signal generator (7608) that generates: a control signal that includes a first signal indicating at least one transmission scheme to be used to transmit flow data by physical layer pipes, PLP, and a second signal indicating a transmission scheme to be used to transmit the control signal; a signal processor (7612) that generates data symbols by modulating the flow data and that generates a control symbol by modulating the control signal; a frame configurator (7610) that configures at least one SISO frame and at least one MISO / MIMO frame; a pilot symbol inserter that inserts pilot symbols in each SISO frame and in each MISO / MIMO frame; a preamble inserter (7622) that generates a preamble that carries the second signal and that inserts the preamble in each SISO frame and in each MIMO / MISO frame; and a transmitter (7262) that performs transmissions; where The frame configurator (7610) is adapted to configure the at least one SISO frame from the control symbol and the data symbols resulting from the modulation of data for SISO in the flow data and the at least one MISO / MIMO from the control symbol and the data symbols resulting from the modulation of any or both data for MISO and data for MIMo in the flow data; the pilot symbol inserter is adapted to insert pilot symbols in each MISO / MIMO frame in which common pattern pilots are inserted into data for MISO and data for MIMO; The transmitter (7262) can be operated to switch between a first transmission process to transmit the at least one SISO frame and a second transmission process to transmit the at least one MISO / MIMO frame; the second signal identifies the transmission signal transmission scheme from among a plurality of schemes that include at least SISO, MISO and MIMO; Y The first signal identifies, for the PLPs in each SISO frame and in each MISO / MIMO frame, if a transmission scheme for flow data transmitted by each PLP is SISO, MISO or MIMO. 2. A reception device for Single-Input, Single-Output, SISO, Multiple-Input, Single-Output, MISO, and Multiple-Input, Multiple-Output, MIMO, comprising the receiving device: a receiver (701) that receives a transmission frame that is constituted by a preamble, a control symbol and data symbols; a preamble demodulator that detects the preamble to thereby obtain a signal indicating a transmission scheme used to transmit a control signal, the control signal including a signal indicating at least one transmission scheme used to transmit flow data by physical layer pipes, PLP; a control symbol demodulator that demodulates the control symbol based on the signal indicating the transmission scheme used to transmit the control signal to thereby obtain the control signal; Y a data symbol demodulator that demodulates data symbols based on the control signal and emits the flow data; where The receiver is adapted to receive, as the transmission frame, at least one SISO frame transmitted by executing a first transmission process, the at least one SISO frame being configured from the data symbols resulting from the modulation of data for SISO in the flow data, the control symbol and the preamble, and at least one MISO / MIMO frame transmitted by executing a second transmission process, the at least one MISO / MIMO frame being configured to from data symbols resulting from the modulation of any or both data for MISO and data for MIMO, and it being possible to switch between the first transmission process and the second transmission process; the signal indicating the transmission scheme used to transmit the control signal identifies the transmission scheme of the control signal from among a plurality of schemes that include at least SISO, MISO and MIMO; the signal indicating the transmission scheme used to transmit the flow data identifies, for the PLPs in each SISO frame and in each MISO / MIMO frame, if a transmission scheme for flow data transmitted by each PLP is SISO, MISO or MIMO; Y each SISO frame is configured with pilot symbols inserted in it, each MISO / MIMO frame is configured with pilot symbols inserted in it, and each MISO / MIMO frame uses common pattern pilots for PLP data for MISO and PLP data for MIMO. 3. A transmission method for Single-Input, Single-Output, SISO, Multiple-Input, Single-Output, MISO, and Multiple-Input, Multiple-Output, MIMO, comprising the transmission method: 5 10 fifteen twenty 25 30 35 40 Four. Five fifty generate: a control signal that includes a first signal indicating at least one transmission scheme to be used to transmit flow data via physical layer pipes, PLP, and a second signal indicating a transmission scheme to be used to transmit the control signal; generate data symbols by modulating the flow data, and generate a control symbol by modulating the control signal; configure at least one SISO frame and at least one MISO / MIMO frame; insert pilot symbols in each SISO frame and in each MISO / MIMO frame; generate a preamble that carries the second signal and insert the preamble in each SISO frame and each MIMO / MISO frame; and perform transmission; where the at least one SISO frame is configured from the control symbol and data symbols resulting from the modulation of data for SISO in the flow data, and the at least one MISO / MIMO frame is configured from the symbol of control and data symbol resulting from the modulation of any or both data for MISO and data for MIMO in the flow data; Pilot symbols are inserted into each MISO / MIMO frame whereby common pattern pilots are inserted into data for MISO and data for MIMO; the transmission stage changes between a first transmission process to transmit the at least one SISO frame and a second transmission process to transmit the at least one MISO / MIMO frame; the second signal identifies the transmission signal transmission scheme from among a plurality of schemes that include at least SISO, MISO and MIMO; Y The first signal identifies, for the PLPs in each SISO frame and in each MISO / MIMO frame, if a transmission scheme for flow data transmitted by each PLP is SISO, MISO or MIMO. 4. A reception method for Single-Entry, Single-Output, SISO, Multiple-Entry, Single-Output, MISO, and Multiple-Entry, Multiple-Output, MIMO, comprising the reception method: receive a transmission frame that is constituted by a preamble, a control symbol and data symbols; detecting the preamble to thereby obtain a signal indicating a transmission scheme used to transmit a control signal, the control signal including a signal indicating at least one transmission scheme used to transmit flow data via physical layer pipes , PLP; demodulate the control symbol based on the signal indicating the transmission scheme used to transmit the control signal to obtain the control signal in this way; and demodulate the data symbols based on the control signal and issue the flow data; wherein the reception stage receives, as the transmission frame, at least one SISO frame transmitted by executing a first transmission process, the at least one SISO frame being configured from data symbols resulting from the modulation of data for SISO in the flow data, the control symbol and the preamble, and at least one MISO / MIMO frame transmitted by executing a second transmission process, the at least one MISO / MIMO frame being configured to from data symbols resulting from the modulation of any or both data for MISO and data for MIMO, and it being possible to switch between the first transmission process and the second transmission process; the signal indicating the transmission scheme used to transmit the control signal identifies the transmission scheme of the control signal from among a plurality of schemes that include at least SISO, MISO and MIMO; the signal indicating the transmission scheme used to transmit the flow data identifies, for the PLPs in each SISO frame and each MISO / MIMO frame, if a transmission scheme for flow data transmitted by each PLP is SISO, MISO or MIMO; Y each SISO frame is configured with pilot symbols inserted in it, each MISO / MIMO frame is configured with pilot symbols inserted in it and each MISO / MIMO frame uses common pattern pilots for PLP data for MISO and data of PLP for MIMO.