ES2334532B1 - METHOD OF OPTIMIZATION OF THE DESIGN OF INTEGRATED MAGNETIC COMPONENTS AND INTEGRATED MAGNETIC COMPONENT OBTAINED BY SUCH METHOD. - Google Patents

Abstract

Optimization method of component design integrated magnetic and integrated magnetic component obtained by Said method

Method of optimizing the design of a integrated magnetic component formed by a plurality of discrete magnetic components, by increasing degrees of freedom of design of at least a first magnetic component discrete, where at least said first discrete magnetic component and a second discrete magnetic component share a core and at less a winding of said integrated magnetic component. He method comprises the steps of: choosing a magnetic component discrete forming said integrated magnetic component; Obtain a equivalent electrical circuit that reproduces the behavior electrical of said integrated magnetic component so that, in at least a portion of said core, the circulating magnetic flux it comes substantially only from said magnetic component Discreetly chosen. The method is characterized by the stage of: dividing the at least one winding shared by said component Magnetic discrete chosen whose degrees of design freedom are increased and at least a second discrete magnetic component, in at least two sub-windings, where at least one of said sub-windings is placed in a column of said core in which the winding was original and where at least one of said sub-windings is placed in a second column of said core through which the magnetic flux that circulates is due substantially to the discrete magnetic component chosen, whose degrees of design freedom are increased.

Description

Optimization method of component design integrated magnetic and integrated magnetic component obtained by Said method

Field of the Invention

The present invention applies to the field of integrated magnetic components, and more specifically, to the design optimization of integrated magnetic components.

       \ vskip1.000000 \ baselineskip
    

Background of the invention

The magnetic integration technique for Magnetic component manufacturing has been used since over 20 years ago For example, Cuk, S., in "New magnetic structures for switching converters "(IEEE Transactions on Magnetics; Volume 19, Issue 2, Mar 1983 Pages: 75-83), describes a topology that allows to achieve a null curl at the output of a DC / DC converter, and in which in addition the different magnetic components involved are integrated in the same magnetic core. The integration technique magnetic seeks to optimize the design of magnetic components by integrating several of them in the same core, of so that global energy efficiency is increased, and the weight and size are reduced.

Magnetic integration is used to design  magnetic components used in a great diversity of applications, such as high frequency DC / DC converters or Signal filtering applications.

For example, the patent application US5783984 describes the application of this technique to the integration of an LCL filter and a transformer. For it, only a magnetic core is necessary to constitute the two inductances of the LCL filter and the transformer, instead of the three independent magnetic cores that would be necessary in the case of constructing the filter using discrete components or individual.

In this way, the fact of using less Magnetic material and less copper is reflected in a decrease in weight and size, and an increase in energy efficiency.

However, magnetic integration Conventional presents a number of problems: On the one hand, in some cases the design of the integrated components cannot be Do independently. This not only increases the difficulty of design, but limits the ability to optimize as to size and weight, one of the fundamental aims sought when applying magnetic integration techniques. This problem gets from manifested in US5783984, where transformer e is integrated inductance using only two windings.

Making use of this magnetic topology integrated, if a voltage transformer is required relatively high and relatively low inductances, is necessary, or use a core with a column section very large, or very large air gaps. The first measure incurs in high losses in the core, while the second one reflects in a high electromagnetic emission due to the effect "fringing flux", which is increased when using large air gaps

This results in a limitation in the optimization capacity and greater difficulty in the process of design.

There are jobs that have been done distribution of windings in order to have more freedom to the when designing the integrated magnetic components, about certain integrated magnetic component topologies for Certain applications of power electronics. For example, at work "Integrated magnetic full wave converter with flexible output inductor ", Transactions on Power Electronics, Volume 18, Issue 2, March 2003 Page (s): 670-678 of Liang Yan; Dayu Qu; and Lehman, it is proposed a specific topology that solves this problem for a application of a specific DC / DC converter. Another concrete solution  is proposed in the work "Transformer and Series Inductance Integration for Harmonic Filtering in PWM Inverters Based in a Simple Design Procedure ", ISIE 2007 proceedings, IEEE, by Jorge Pleite, Virgilio Valdivia, Carlos González and Pablo Zumel.

       \ vskip1.000000 \ baselineskip
    

Summary of the Invention

The present invention seeks to solve the problems mentioned above by a method of design optimization of integrated magnetic components that add enough degrees of freedom so that they can be designed all components integrated independently and through simple equations.

Thus, in one aspect of the present invention, provides a method of optimizing the design of a component integrated magnetic formed by a plurality of components discrete magnetic, by increasing degrees of freedom of the design of at least a first discrete magnetic component, where at least that first discrete magnetic component and a second discrete magnetic component share a core and at least one winding of the integrated magnetic component. The method It comprises the steps of: choosing a discrete magnetic component that forms that integrated magnetic component; get a circuit electrical equivalent that reproduces the electrical behavior of that integrated magnetic component so that, in at least one core portion, the circulating magnetic flux proceeds substantially only of the discrete magnetic component chosen one. The method is characterized by the stages of: dividing the al minus a winding shared by the magnetic component discrete chosen whose degrees of design freedom are increased and at least a second discrete magnetic component, in at least two sub-windings, where at least one of those sub-windings is placed in a core column in which the winding was original and where at least one of the sub-windings is placed in a second column of the nucleus through which the magnetic flux that circulates is due substantially to the discrete magnetic component chosen, whose degrees of design freedom are increased.

Optionally, the winding shared by the two discrete magnetic components are divided only into one first sub-winding and a second sub-winding.

In another aspect of the present invention, provides an integrated magnetic component that integrates a plurality of discrete magnetic elements, obtainable through the method mentioned above.

Preferably, that magnetic component integrated is formed by an integrated LCL filter with a transformer.

In a particular embodiment, that component it comprises a magnetic core that in turn comprises a plurality of columns, each of which is connected to each column adjacent by a first perpendicular section of nucleus magnetic and a second perpendicular section of magnetic core. In addition, two of those columns each comprise at least one air gap. A column that does not comprise air gaps comprises a first winding, another column that does not include air gaps it comprises a second winding and the two perpendicular sections which connect the two inner columns comprise a total of three windings

In another particular embodiment, the component it comprises a magnetic core that in turn comprises a plurality of columns, each of which is connected to each column adjacent by a first perpendicular section of nucleus magnetic and a second perpendicular section of magnetic core. A of the two leftmost columns of the nucleus magnetic comprises at least a first air gap and one of the two columns located more to the right of the magnetic core comprises At least a second air gap. The first column that comprises at least one first air gap also comprises a first and a second winding and the second column comprising at least another first air gap also comprises a third and a fourth winding A third column comprises a fifth and a sixth windings and a fourth column comprises a seventh and a eighth windings.

       \ vskip1.000000 \ baselineskip
    

Brief description of the figures

In order to help a better understanding of the features of the invention according to an example preferred practical implementation of the same and to complement This description is accompanied as an integral part of it a game of drawings, whose character is illustrative and not limiting. In these drawings:

Figure 1a shows an inductance and a Discrete transformer according to the state of the art.

Figure 1b shows the integration of a inductance and a transformer with core sharing magnetic.

Figure 1c illustrates the integration of a transformer and an inductance with core sharing and windings

Figure 2a shows a series inductance and a integrated transformer with core sharing and windings

Figure 2b shows the reluctance model star of the integrated component of figure 2a, while Figure 2c shows the triangle reluctance model of the same. Figure 2d illustrates an electrical circuit equivalent that can be deducted immediately from the representation of Figure 2c

Figure 3a shows a PWM inverter connected to network through a series inductance of filtering and a transformer Low frequency that provides galvanic isolation.

Figure 3b illustrates a second circuit Electrical equivalent for the solution shown in the figure 2nd.

Figure 3c shows the flow distribution of the magnetic component 2a, associated with the electrical circuit of the figure 3b.

The 3d figure shows a magnetic component integrated resulting from applying the method of the present invention on the topology of figure 2a.

Figure 3e shows the reluctance model of the magnetic component of figure 3d.

Figure 3f illustrates an electrical circuit that models the electrical behavior of the magnetic component of the 3d figure

Figure 3g shows the flow distribution of the magnetic component of figure 3d, associated to the circuit electric figure 3f.

Figure 3h shows an alternative application of the method of the present invention.

Figure 3i shows the reluctance model of the magnetic component of figure 3h.

Figure 3j shows a circuit equivalent electrical modeling the electrical behavior of the component magnetic figure 3h.

Figure 3k shows the flow distribution of the magnetic component of figure 3h, associated to the circuit electric figure 3j.

Figure 4a shows a grid connected inverter through an LCL filter and a single phase transformer, second example of a system whose magnetic components can be integrated and for which the application of the method of the present invention.

Figure 4b shows a magnetic component that  integrates all the magnetic elements that appear in the system of figure 4a.

Figure 4c shows the reluctance model of the magnetic component of Figure 4b.

Figure 4d illustrates an electrical circuit that models the electrical behavior of the component of the figure 4b

Figures 4e represents the distribution of flux of the magnetic component of figure 4b, associated with electrical circuit of figure 4d.

Figure 4f illustrates a component topology  integrated magnetic resulting from applying the method of the present invention on the magnetic component of figure 4b. It is included the flow distribution on it, associated with the equivalent electric figure 4h.

Figure 4g illustrates the reluctance model of the magnetic component of Figure 4f.

Figure 4h illustrates an electrical circuit that models the electrical behavior of the magnetic component of the figure 4f.

Figure 4i illustrates the flow distribution simplified of the magnetic component of figure 4b, associated with electrical circuit of figure 4d.

Figure 4j illustrates another alternative topology integrated magnetic component that integrates the components magnetic elements that appear in the system of figure 4a. It is included on it the flow distribution associated with the equivalent electric of figure 41.

Figure 4k illustrates the reluctance model of the magnetic component of Figure 4j.

Figure 4l illustrates an electrical circuit that models the electrical behavior of the magnetic component of the figure 4j.

Figure 4m illustrates a component topology  integrated magnetic resulting from applying the method of the present invention on the magnetic component of figure 4j. It is included the flow distribution on it, associated with the equivalent electric figure 4o.

Figure 4n illustrates the reluctance model of the magnetic component of figure 4m.

Figure 4 illustrates an electrical circuit that models the electrical behavior of the magnetic component of the figure 4m.

Figures 5a, 5b and 5c illustrate the same. example of figures 4f, 4g and 4h.

Figures 6a, 6b and 6c illustrate the same example of figures 4m, 4n and 4o.

Detailed description of the invention

Then a series of definitions located in the field of magnetic integration, which  they will be useful to understand the purpose of the invention.

It is understood by "magnetic component discrete "to a conventional conventional passive magnetic component of an electrical circuit, such as a magnetic coil (or inductance) or a transformer. The figure is illustrated by a discrete transformer and a discrete inductance conventional.

It is understood by "magnetic component integrated "to that magnetic component that encompasses a plurality of discrete magnetic components, which are designed so that they are integrated by sharing some of their common elements, such as the core, windings, or both of them.

In this way, it is understood by "sharing of magnetic core "to the fact that the same magnetic core is shared by two or more discrete magnetic components that They integrate. As an example, Figure 1b shows the integration of an inductance and a transformer in which it is only shared magnetic core The numerical references NT1 and NT2 represent to the discrete transformer, while the reference NL represents to discrete inductance. Reference 1 represents a air gap.

On the other hand, it is understood as "sharing of windings "to share at least one winding, this winding being therefore associated with more of an integrated discrete magnetic component. For example, the Figure 1c illustrates a transformer integration solution e inductance, in which both core sharing occurs Like windings. At the top, reference NL10 It represents a discrete inductance. One of the arms comprises an air gap 2. NT10 and NT20 represent a transformer discreet. At the bottom, the integrated component is represented by the NT10 'and NT20' transformation ratios. He air gap is represented by 2 '.

Also, remember that, in a transformer, dispersed flow is one whose energy is not transferred from a wound to another.

Furthermore, in the context of the present invention, the terms "approximately", "substantially" and "practically" should be understood as indicating very close values to which said term accompanies. The person skilled in the art will understand that a small deviation from the indicated values, within reasonable terms, is inevitable. For example, when it is indicated that the magnetic flux that circulates through a part of a magnetic core is practically due to a single discrete magnetic component, it should be understood that magnetic fluids can be circulated through said part of the magnetic core that can be considered negligible with respect to the first, because the magnitude of the former is much greater than that considered considered negligible. The analysis of the solutions and the application of the method of the invention are greatly simplified if these simplifications due to negligible flows are taken into account. These effects are negligible or not depending on the topology on which the
method.

The present invention provides a method of Improved design of integrated magnetic components based on the winding distribution. Using this technique, they can be increased degrees of design freedom of a given given magnetic integration topology. This technique allows to have the sufficient number of degrees of freedom to perform the design of all discrete magnetic components that integrate from independently, so that the capacity of optimization thereof and simplifies its procedure of design.

       \ vskip1.000000 \ baselineskip
    

The design method is detailed below and its scope:

The application of the design method that presents is useful to improve a certain topology of Magnetic integration that meets the following:

1) Core sharing occurs.

2) Sharing of windings

3) You cannot adjust the number of turns of windings associated with discrete components and, because of that, you cannot award the optimal number of turns for each component.

If these three premises are met, it is necessary also be able to develop an electric circuit equivalent of such that at least a portion of the core circulates only flow associated with the discrete magnetic component for the which one wishes to increase the degrees of design freedom. He example presented below may serve to clarify For this point. If this electrical equivalent cannot be reached  The method is not applicable.

       \ vskip1.000000 \ baselineskip
    

Once the electrical circuit is obtained equivalent required, the application of the method can be performed systematically following the following steps:

1) Selection of the component or components discrete for which degrees of design freedom must be increased

2) Identification of a winding that is shared between the discrete component of the desired increase the degrees of design freedom and others components.

3) The winding subdivision is carried out in sub-windings. The first sub-winding must be placed where the original winding was placed (refers to the location of the winding before proceeding to subdivision). The rest of sub-windings must be placed each in a portion of the core through which only circulate flow associated with the discrete magnetic component of the equivalent Electrical circuit, whose degrees of design freedom are desired increase.

       \ vskip1.000000 \ baselineskip
    

The process of the invention allows increase the optimization capacity of topologies of magnetic integration presenting core sharing and of windings, increasing their degrees of design freedom.

The results obtained by the method are directly applicable, for example, and not limitatively, in power electronics and power conditioning electric

Examples of implementation of the method of the invention on two topologies that can be used to design integrated passive filters with transformer. These systems are useful, for example, and of non-limiting way, in converter connection applications of power to the power grid.

       \ vskip1.000000 \ baselineskip
    

Example 1

Figure 2a presents a topology of integrated series inductance with transformer. In this case the integration is obtained from the creation of a path magnetic for the dispersed flow through the core. Specific, through the central column which, as shown in figure 2a, comprises an air gap 3. Therefore, the primary windings and secondary transformer are also associated with the series inductance, thereby obtaining both sharing of core like windings N_ {N} {2}. Thanks to the sharing of windings less amount of copper, and this results in less equipment size and weight, more energy efficiency, and less costs.

The reluctance model of this solution is shown in figures 2b and 2c: The one in figure 2b is a model of star reluctance while performing a star-triangle transformation, you get the equivalent of figure 2c (triangle reluctance model). The reluctance values of the model in Figure 2c are They relate to those in Figure 2b as follows:

       \ vskip1.000000 \ baselineskip
    

one

       \ vskip1.000000 \ baselineskip
    

The circuit equivalent can be extracted classical electrical of a transformer (figure 2d), whose equations  Definitive are the following:

       \ vskip1.000000 \ baselineskip
    

2

The defining equations show how The entire magnetic core is shared by the elements discrete present in the integration, and how the windings N_ {1} and N_ {2} are shared (N_ {1} per transformer and coil of input, and N2 per transformer and output coil). The flow dispersed through the air is neglected (the dispersion effect at through the central column is much larger), and the transformer is represents by its magnetizing inductance and a coupling ideal magnetic.

An application example is now proposed industrial for which this type of solution may be useful: An inverter 32 with PWM modulation connected to the power grid 33, to supply it with the energy generated from panels photovoltaic 31 (figure 3a).

In this case, the L1 series inductance allows the  electronic system behave as a source of current, in addition to allowing high harmonic content to be filtered PWM modulation frequency. The Tr transformer provides of galvanic isolation and allows an adjustment of the amplitude of the output voltage.

To avoid having an excessive voltage drop in the series reactance, it is advisable to calculate it so that its value is not excessively high, to avoid a high fall from voltage to grid frequency in it.

Using the solution in Figure 2a allows Save the amount of copper and core needed. But nevertheless, since, according to the design of figure 2a, both the transformer as the series inductance require to be defined by it number of turns, are necessary, or very air gaps large (which produces a lot of EMI electromagnetic emission), or Well very large core column sections.

It is therefore desirable to increase the number of  degrees of freedom of these components, so that both are You can assign different numbers of turns.

It is then interesting to apply the method of the invention. We check that the requirements are met necessary:

The first two requirements are met, put that both core and windings are shared. However it is necessary, in order to apply the method of the invention, obtain a electrical circuit that also meets the third requirement, since, analyzing the most immediate electrical circuit (figure 2d), it notes that, in view of the equations that define it, The entire core is shared by all the elements.

       \ vskip1.000000 \ baselineskip
    

The electrical behavior of a component magnetic can be studied by more than one circuit electric. The analysis of these elements as quadrupoles, and the obtaining its Z parameters is very useful for deducing new equivalent electrical circuit topologies. Analyzing the reluctance model of Figure 2b through the laws of Kirchoff, in combination with Lenz Faraday's law (system of equations 1), you get the system of equations 2. On the other part, the Z parameters obtained from the circuit analysis electrical of Figure 2d are shown in the system of equations 3. The secondary magnetizing inductance is referred to by the  following transformation:

       \ vskip1.000000 \ baselineskip
    

       \ vskip1.000000 \ baselineskip
    

3

4

Equating the systems of equations (2) and (3), we see how we have a degree of freedom to give value to one of the four elements that make up the electrical circuit, put that we have 4 elements and only 3 linear equations independent. In this way, if we do L_ {l2} = 0, you can obtain the electrical circuit that appears in figure 3b, and in the that only one inductance series L_ {sa} appears. The equations definitions, expressed in terms of the reluctance model in star of figure 2b, are the following:

5

Magnetic flux distribution corresponding to this new electrical representation is shown in Figure 3c, where \ Phi_ {Lm2a} represents the magnetic flux associated with the magnetizing inductance of the transformer and \ Phi_ {Lsa} represents the magnetic flux associated with the series inductance.

From figure 3c, as well as from the observation of the defining equations of the elements of this new electrical circuit, it is checked that the right column only is associated to the transformer, that is, no flow circulates related to the coil through it. This means that this column is not shared by the two magnetic elements integrated discrete (the series inductance L_ {sa} on the one hand, and the transformer composed of the coupling tr_a and the magnetizing inductance referred to the secondary L_ {m2a} by another part. The asymmetric flow distribution is due to the new and also asymmetric electric circuit representation. Although in figure 2d two series inductances appear, physically, the only path for dispersed flow within the nucleus is represented by the central column, whereby these two series inductances are not independent and its effect can be presented from a single inductance).

       \ vskip1.000000 \ baselineskip
    

Now the three requirements are met necessary and sufficient for the method to be applied. its Application is simple and systematic.

1) The discrete component whose degrees of freedom you want to increase. In our case we choose The transformer For the aforementioned, an optimal design It requires more turns than the series inductance.

2) A winding shared by the Two discrete elements. In our case "N_ {1}" meets this  condition.

3) "N_ {1}" is subdivided into two sub windings "N_ {11}" is placed where "N_ {1}" it was previously placed, and "N_ {12}" is placed in the right column, since in this one there is only flow associated with the transformer. This new model is represented in the 3d figure, obtained after applying this step 3) to the magnetic component integrated of figure 2a.

       \ vskip1.000000 \ baselineskip
    

The new reluctance model is shown in the  figure 3e. Lenz Faraday's law is formulated on this occasion as and as it appears in equations (4), and the new Z parameters are those presented in the system of equations (5). Notice that the serial connection of the two sub-windings It should be done in such a way that the magnetomotive forces associated with both are additive, as shown in the circuit Figure 3e

       \ vskip1.000000 \ baselineskip
    

6

       \ vskip1.000000 \ baselineskip
    

Matching (5) and (3), and doing again L_ {12} = 0, the same electric circuit equivalent is obtained and the following defining equations (figure 3f):

       \ vskip1.000000 \ baselineskip
    

7

       \ vskip1.000000 \ baselineskip
    

Notice how now the defining equation of transformer has changed, and now its primary depends so much on N_ {11} as of N_ {12}. However, the series inductance It only depends on N_ {11}. We have gained a degree of freedom extra, and now you can assign different turns to the primary of the transformer and inductance, although the winding sharing N_ {11}. This is easily observed. in figure 3g, which shows the flow distribution for the component resulting from applying the method. He despises himself in said flow distribution flow due to magnetizing inductance of the transformer of the central column.

You can also apply the method with the same objective increasing the degrees of design freedom of the inductance. For that specific case, it should be taken into account that the flow associated with the magnetization of the transformer that circulates through the central column may be negligible with respect to circulating through the outer columns. Performing this simplification, winding N 12 is placed in the column central, connecting it so that the total turns of the series inductance are due to the difference between N_ {11} and N_ {12}. This alternative application of the methodology presented to the integrated magnetic component of figure 2a is presented in figure 3h. For its application, it is also part of a electrical equivalent as used in the previous case, and it apply step 3. Following an analysis similar to the one already described in detail, a reluctance model is obtained as illustrated by the Figure 3i and an electric circuit equivalent as shown in the figure 3j. The defining equations are the following:

8

The modification of the sense in which it coils the winding of the intermediate column would make N_ {11} and N_ {12} they joined the defining equation of L_ {sa} ''.

These solutions do not present difficulties added to the manufacturing process and yet the their optimization capacity is increased in a way considerable.

       \ vskip1.000000 \ baselineskip
    

Example 2

A second example is illustrated below. (Figure 4a) of application of the method of the invention, applied to the integration of an LCL filter and a transformer in the same magnetic core, useful for connection of inverter 42 to mains 43, as in the case of example 1. Regarding the solution from example 1, the LCL filter allows filtering of a order three times higher, which can increase the quality of the current signal injected into the network for the same total inductance value.

Figure 4b shows a solution composed of Four columns and three windings, presented in US5783984. He tertiary winding allows the capacitor to be connected, and the columns extremes, to which an air gap is practiced, allow passage of the flow associated to the series inductances through the nucleus. The equations that define the electric model are obtained in a simple way obtaining the Z parameters from a study of the electric and magnetic models of figures 4c and 4d as a hexapolo (system of equations 6), following a procedure analogous to that followed in example 1.

9

The problem with this solution is the same as the of the one raised in example 1 for the application we cover. In fact, to achieve the same level of attenuation of harmonics of the current at the output, in this case they are required minor inductances due to the higher order of the filter. The defining equations of the electrical circuit elements of Figure 4d are as follows:

10

Tertiary inductance may refer to primary or secondary (see figure 4d) using the following transformations:

eleven

For this circuit representation, the flow distribution corresponds to that presented in the figure 4e.

From the analysis of figures 4c, 4d and 4e, and of the defining equations shown above, it is extracted that the windings N_ {1} and N_ {2} are shared, and that also It produces sharing of part of the magnetic core. In addition, there is a part of the core through which only magnetizing flow circulates associated to the transformer (note that now associated to the transformer there is a magnetizing inductance and two couplings ideal magnetic, due to the existence of the third winding  N_ {3}). Therefore, the method is applicable. In this case we will also increase the degrees of design freedom of the transformer.

       \ vskip1.000000 \ baselineskip
    

The steps followed are:

1) Subdivision of the shared windings N_ {1} in N_ {11} and N_ {12}, and N_ {2} in N_ {21} and N_ {22}. N_ {1} is shared by coil N_ {1} and the transformer, and N_ {2} is shared by coil N_ {2} and the transformer.

2) N_ {11} and N_ {22} are placed where previously N1 and N2 were located, respectively.

3) N 12 and N 21 are placed in the core portions where there is only flow associated with the transformer, that is, in any of the core portions that exist above and below the central window.

       \ vskip1.000000 \ baselineskip
    

The resulting topology, and the circuits The resulting magnetic and electrical are presented in figures 4f, 4g and 4h (flow distribution is included on figure 4f, which presents the component). The defining equations are the following:

       \ vskip1.000000 \ baselineskip
    

12

       \ vskip1.000000 \ baselineskip
    

Notice that any of the windings N_ {12}, N_ {3} and N_ {21} could be placed in the core part existing under the central window, without modifying the operation nor the defining equations of the model.

Figures 5a, 5b and 5c illustrate this example, whose numerical references have been renamed for facilitate your understanding Specifically, Figure 5a corresponds to 4f, figure 5b corresponds to 4g and figure 5c to 4h. In Figure 5a the four columns of the magnetic core are represented by references c1 c2 c3 c4, while the respective core sections that join these columns are represented by references d1 d2 d3 d4 d5 d6. Both air gaps, which in the present example are located, not limiting, in columns c1 and c4, they are represented by e1 and e2.

Taking into account the change of nomenclature, The defining equations are as follows:

13

As can be seen in Figure 5a, this LCL filter integrated with a transformer consists of a magnetic core which in turn comprises a plurality of columns c1 c2 c3 c4. Each of these columns is connected to each adjacent column by a first upper perpendicular section of magnetic core d1 d2 d3 and a second perpendicular section Magnetic core bottom d4 d5 d6. Of these four columns c1 c2 c3 c4, two of them (the two that do not include windings) each comprise at least one e1 e2 air gap. In this case, these two columns c1 c4 each comprise a single air gap e1 e2. Column c2 that does not comprise air gaps comprises a first winding N_ {11d1}. In addition, another column (in this case c3) that does not include air gaps comprises a second winding N_ {22d1}. The two upper perpendicular sections and lower d2 d5 connecting the two inner columns c2 c3 They comprise a total of three windings N_ {12d1} N_ {3d1} N_ {21d1}. These three windings can be organized in different ways, as long as the three are located in those two central sections d2 d5. In the example in Figure 5a, the three windings are in the same section d2. Alternatively, the three windings can be on the opposite section d5. Similarly, two windings can be in a section (d2 or d5) and the third in the opposite section (d5 or d2).

       \ vskip1.000000 \ baselineskip
    

Example 3

A second integration solution is attached LCL filter and transformer. This solution is also based on a Four-column core with two air gaps. However, it can be easier from a manufacturing point of view, since that all windings are placed in the columns vertical

Your deduction starts from making a simplification on the distribution of solution flows shown in the example of figure 4b, shown in the figure 4e. Given that the reluctance of the columns extremes is much greater than that of internal (due to air gaps), the flow associated with the magnetizing inductance, circulating through said extreme columns, it can be despised This simplified representation is shown in the figure 4i. You can, in this way, divide the tertiary and reposition it in such a way that the magnetomotive force generated in its terminals are null (figure 4j). The circuit equivalents corresponding magnetic and electrical are shown in the figures 4k and 4l, respectively. It should be noted that it can be done N_ {31} = N_ {32} = 0 and N_ {33} = N_ {34} \ neq0, or N_ {33} = N_ {34} = 0 and N_ {31} = N_ {32} \ neq0, and would not be seen altered the operation of the solution. The equations Simplified definitions are as follows:

14

As explained in the case of the example 1, the modification of the winding direction on the core of N_ {33} as of N_ {34} would make in the defining equations of t_ {r2c} and L_ {m3c} the elements that make up both numerators were added. This change would imply that the direction of rotation of N_ {34} is also modified, to achieve desired magnetic coupling and decoupling effects.

On this solution you can also apply the method object of the invention, so that they are only placed windings in the vertical columns. In this case, they divide primary and secondary as it appears in figure 4m. The idea it consists in increasing the degrees of design freedom of the two series inductances of the filter.

Following an analysis similar to that already described in detail, a reluctance model is obtained as illustrated by the Figure 4n and an electric circuit equivalent as shown in the figure 4o. The defining equations are the following:

       \ vskip1.000000 \ baselineskip
    

fifteen

       \ vskip1.000000 \ baselineskip
    

For this and for all the previous cases, they must make the connections of the windings so that the senses of the magnetomotive forces remain as It appears in the magnetic circuits associated with each figure. A change in the winding direction of N_ {12} and N_ {21} implies that the elements of the denominators of L_ {sc1 '} and L_ {sc2 '}, respectively, become added. In case of change the winding direction of N_ {32} or N_ {33}, those of N_ {31} and N_ {34} should also be modified, respectively, so that the coupling effects are achieved and Magnetic decoupling desired.

Figures 6a, 6b and 6c illustrate this example, whose numerical references have been renamed for facilitate your understanding Specifically, Figure 6a corresponds to 4m, figure 6b corresponds to the 4n and figure 6c to the 4th. In Figure 6a the four columns of the magnetic core are represented by references c10 c20 c30 c40, while the respective core sections that join these columns are represented by references d10 d20 d30 d40 d50 d60. Both air gaps, which in the present example are located, not limiting, in columns c10 and c40, they are represented by e10 and e20.

Taking into account the change of nomenclature, The defining equations are as follows:

       \ vskip1.000000 \ baselineskip
    

16

       \ vskip1.000000 \ baselineskip
    

As can be seen in Figure 6a, this LCL filter integrated with a transformer consists of a magnetic core which in turn comprises a plurality of columns c10 c20 c30 c40. Each of these columns is connected to each adjacent column by a first upper perpendicular section of magnetic core d10 d20 d30 and a second perpendicular section Magnetic core bottom d40 d50 d60.

Of these four columns c1 c2 c3 c4, one of the two columns further to the left of the magnetic core (or either, c10 or c20) comprises at least a first air gap (e10) and one of the two columns to the right of the nucleus magnetic (that is, c30 or c40) comprises at least one second air gap (e20). In the case of this example, the column plus the left (c1) comprises a single air gap the and the column of more to the right (c4) comprises a single air gap e2.

In addition, the first column comprising the minus a first air gap (in this case, column c10) it also comprises a first and a second winding (N_ {31d2}, N_ {12d2}) and the other column comprising at least one other first air gap (in this case, column c40) further comprises a third and fourth winding (N_ {34d2}, N_ {21d2}). The others two c20 c30 columns each comprise two windings (the column c29 comprises N_ {32d2} and N_ {11d2} and column c30 it comprises N_ {33d2} and N_ {22d2}).

In short, the present invention provides a systematic and general method of optimizing the design of integrated magnetic components that adds degrees of freedom enough to be designed for all components integrated independently and through equations simple. This allows the solution provided to be useful for any topology or application, as long as it meets the requirements that have already been exposed.

In sum, although there are previous works in the which are presented solutions with core sharing and windings and sufficient degrees of freedom, in any of These cases propose a generalized design method.

The presented invention describes a systematic and general procedure, which can be applied to any topology oriented to any application, provided and when it meets the requirements that were already exposed.

In view of this description and game of figures, the person skilled in the art will understand that the invention has been described according to some preferred embodiments of the same, but that multiple variations can be introduced in said preferred embodiments, without leaving the object of the invention as claimed.

Claims (6)

1. Method of optimizing the design of a integrated magnetic component formed by a plurality of discrete magnetic components, by increasing degrees of freedom of design of at least a first magnetic component discrete, where at least said first discrete magnetic component and a second discrete magnetic component share a core and at minus a winding of said integrated magnetic component, where said method comprises the steps of: - choose a discrete magnetic component that said integrated magnetic component forms; - obtain an equivalent electrical circuit that reproduce the electrical behavior of said component magnetic integrated so that, in at least a portion of said core, the circulating magnetic flux proceeds substantially only of said discrete magnetic component chosen; characterized by the stage of: - divide the at least one winding shared by said discrete magnetic component chosen whose degrees of design freedom are increased and at least one second discrete magnetic component, in at least two sub-windings, where at least one of said sub-windings is placed in a column of said nucleus in which the original winding was and where at  minus one of said sub-windings is placed in a second column of said core through which the magnetic flux that circulates is due substantially to the magnetic component Discreetly chosen, whose degrees of design freedom are increased 2. Method according to claim 1, wherein said  winding shared by said first and second components discrete magnetic is divided into at least a first sub-winding and a second sub-winding. 3. Integrated magnetic component that integrates a plurality of discrete magnetic elements, obtainable by the method of claim 1. 4. Magnetic component integrated according to the claim 3, formed by an LCL filter integrated with a transformer. 5. Integrated magnetic component according to the claim 4, comprising a magnetic core which in turn it comprises a plurality of columns (c1, c2, c3, c4), each of which is connected to each adjacent column by a first perpendicular section of magnetic core (d1, d2, d3) and a second perpendicular section of magnetic core (d4, d5, d6), and where two of said columns (c1, c4) each comprise at least an air gap (e1, e2), characterized in that a column (c2) that does not comprise air gaps comprises a first winding (N 11d1), a column (c3) that does not comprise air gaps comprises a second winding (N 22d1) and the two perpendicular sections (d2, d5) that connect the two inner columns (c2, c3) comprise a total of three windings (N_ {12d1}, N_ {3d1}, N_ {21d1}). 6. Integrated magnetic component according to the claim 4, comprising a magnetic core which in turn it comprises a plurality of columns (c10, c20, c30, c40), each of which is connected to each adjacent column by a first perpendicular section of magnetic core (d10, d20, d30) and a second perpendicular section of magnetic core (d40, d50, d60), and where one of the two leftmost columns of the magnetic core (c10, c20) comprises at least a first air gap (e10) and one of the two columns located further to the right of the magnetic core (c30, c40) comprises at least one second air gap (e20), characterized in that the first column (c10) comprising at least a first air gap (e10) further comprises a first and a second winding (N 31d2, N 12d2), the second column (c40) comprising at least one other first air gap (e20) further comprises a third and fourth winding (N_ {34d2}, N_ {21d2}), a third column (c20) comprises a fifth and a sixth windings (N_ {32d2}, N_ {11d2}) and a fourth column (c30) comprises a seventh and an eighth windings (N_ {33d2}, N_ {{d2})).