EP2360776A1

EP2360776A1 - Microwave directional coupler

Info

Publication number: EP2360776A1
Application number: EP20100153670
Authority: EP
Inventors: Gwarek Wojciech
Original assignee: Whirlpool Corp
Current assignee: Whirlpool Corp
Priority date: 2010-02-16
Filing date: 2010-02-16
Publication date: 2011-08-24
Anticipated expiration: 2030-02-16
Also published as: EP2360776B1

Abstract

The present invention discloses a directional coupler comprising a main transmission line (110) comprising an input port (111) and an output port (112) for transmitting a first electromagnetic wave from the input port to the output port and a second electromagnetic wave in the opposite direction, a coupler line (120) arranged proximate to the main transmission line for electromagnetic coupling between the two lines, and a linear electronic means (130). The coupler line comprises a first port (121) at a first one of its ends and a second port (122) at its opposite end (or second end) between which the electronic means is arranged. The electronic means is configured to reduce the output signal induced by the first electromagnetic wave at the second port and the output signal induced by the second electromagnetic wave at the first port. For this purpose, the electronic means is configured to reduce the difference between the oppositely directed signals induced at the second port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the first electromagnetic wave and also configured to reduce the difference between the oppositely directed signals induced at the first port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the second electromagnetic wave.

Description

FIELD OF THE INVENTION

The present invention relates to a directional coupler, and more particularly to a microwave directional coupler for power monitoring applications.

BACKGROUND OF THE INVENTION

Directional couplers are generally employed in the field of radio technology for coupling energy flowing in a primary (or main) transmission line into a secondary transmission line. In the following, the secondary transmission line may also be referred to as the coupler line.
A typical directional coupler, whose basic structure is shown in Figure 1, is a four-port device comprising a main transmission line 110 with an input port 111 and an output port 112 and a coupler line 120 with a first port 121 and a second port 122. The coupler line 120 is placed in proximity to the main transmission line 110 such that electromagnetic coupling is established between the main transmission line 110 and the coupler line 120. Preferably, most of the incident signal transmitted from the input port 111 into the main transmission line 110 exits at the output port 112. Generally, in prior art directional couplers, either the first port 121 or the second port 122 of the coupler line is isolated (i.e. terminated) from the input port 111, while the other port of the coupler line, i.e. the second port 122 or the first port 121, respectively, is the coupled port, where a fraction of the input signal appears due to electromagnetic coupling. All four ports are preferably matched.
Both forward and backward directional couplers, depending on whether the coupling mode is forward or backward, respectively, are known. However, in the following, only backward directional couplers are considered.
Referring again to Figure 1, in a backward directional coupler, for an incident wave transmitted from the input port 111 to the output port 112 of the main transmission line 110, the coupled port is the first port 121 of the coupler line 120, i.e. the port of the coupler line located nearest the input port 111 of the main transmission line 110, and the isolated port is the second port 122 of the coupler line 120. If the signal is reversed so that it enters the main transmission line 110 at the output port 112, most of the signal exits the input port 111, but the coupled port is now the second port 122 that was previously regarded as the isolated port.
The ratio of the signal appearing at (or measured at) the first port 121 with respect to the incident signal at the input port 111 is called the coupling coefficient (or simply the coupling) and may be expressed in decibels (dB). The coupling depends mainly on the coupler (or coupling) geometry. For a specific distance, denoted as "d" in Figure 1, between the main transmission line 110 and the coupler line 120, the coupling reaches a maximum if the length, denoted as "I" in Figure 1, of the region or section at which there is coupling between the main transmission line and the coupler line is equal to a quarter-wavelength. Quarter-wave directional couplers such as e.g. the directional coupler disclosed in US4433313 have therefore been preferred for practical applications.
As there may be some signal appearing at the second (isolated) port 122 of the coupler line 120, another relevant parameter of a directional coupler is the level of parasitic transmission to the second port 122, which is called the isolation and may also be expressed in dB. Isolation corresponds to the ratio between the signal at the input port 111 and the signal at the isolated port 122, expressed in dB.
Another relevant parameter for a directional coupler is the directivity, which corresponds to the ratio of the signals at the first port 121 and the second port 122 of the coupler line 120. Directivity may also be expressed in dB and corresponds to the difference between isolation and coupling, both measured in dB.
Directional couplers have been improved for various purposes over the years with focus on optimization of the above mentioned parameters or characteristics, i.e., isolation, directivity and coupling. There is however still a need for further improvements of such characteristics.
Further, in American patent application US2004/0119559 , a microwave directional coupler of a small size in comparison with other prior art directional couplers is disclosed. The microwave directional coupler comprises a main transmission line and a coupler line such as shown in Figure 1 and comprises also four capacitors, one capacitor coupled at each port of the four ports of the directional coupler, and a fifth capacitor coupled between the output port of the transmission line and the terminated port of the coupler line. According to US2004/0119559 , the values of the capacitors, the transmission line impedances and lengths are selected such that it approximately simulates a quarter-wavelength transmission line at the operating frequency, thereby providing a directional coupler having a rather small size.
Generally, although prior art directional couplers physically have four ports, they are used as three-port devices, such as disclosed in US2004/0119559 , since one of the ports of the coupler line is internally terminated with a passive resistive matching. Thus, for measuring the energy of waves travelling in opposite directions in the main transmission line, i.e. both the energy of a high-frequency electromagnetic wave travelling from the input port to the output port and the energy of a high-frequency electromagnetic wave travelling from the output port to the input port of the main transmission line, two separate directional couplers need to be employed. The two separate directional couplers may be assembled in different manners, e.g., one facing the other one on opposite sides of the main transmission line or one after the other on the same side of the main transmission line. However, a directional coupler made of two separate directional couplers is cumbersome and therefore may be difficult to use for applications where space is limited such as for instance in a radio transmitter. In addition, the need of two separate directional couplers increases the cost of the final product for the intended application.
Thus, there is a need for providing new directional couplers that would alleviate at least some of the above-mentioned drawbacks and/or further improve at least some of the above-mentioned characteristics.

SUMMARY OF THE INVENTION

It is an object of the present invention to alleviate at least some of the above-mentioned drawbacks, and to provide a directional coupler enabling monitoring (or measurement) of waves travelling in opposite directions while still providing good values for its characteristics (such as isolation, directivity and coupling).
According to a first aspect of the invention, this and other objects are achieved by means of a directional coupler as defined in claim 1. The directional coupler comprises a main transmission line, a coupler line and an electronic means. The main transmission line comprises an input port and an output port for transmitting a first electromagnetic wave from the input port to the output port and a second electromagnetic wave from the output port to the input port. The coupler line is arranged proximate to the main transmission line for electromagnetically coupling the coupler line with the main transmission line. The coupler line comprises a first port at a first one of its ends and a second port at its opposite end (or second end) between which the electronic means is connected. The electronic means is linear (i.e. operates linearly) and configured to reduce the output signal induced by the first electromagnetic wave at the second port and the output signal induced by the second electromagnetic wave at the first port. For this purpose, the electronic means is configured to reduce the difference between the oppositely directed signals induced at the second port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the first electromagnetic wave and also configured to reduce the difference between the oppositely directed signals induced at the first port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the second electromagnetic wave.
According to a second aspect of the present invention, the object of the present invention is achieved by means of the method as defined in claim 16. The method for directional coupling of an electromagnetic wave comprises the steps of providing a main transmission line having an input port and an output port for transmitting a first electromagnetic wave from the input port to the output port and a second electromagnetic wave from the output port to the input port. The method further comprises the step of providing a coupler line comprising a first port at a first one of its ends and a second port at its opposite end, the coupler line being electromagnetically coupled with the main transmission line. The method further comprises the steps of providing a linear electronic means connected between the first port and the second port for reducing the output signal induced by the first electromagnetic wave at the second port and reducing the output signal induced by the second electromagnetic wave at the first port. More specifically, the method comprises the step of reducing the difference between the oppositely directed signals induced at the second port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the first electromagnetic wave and the step of reducing the difference between the oppositely directed signals induced at the first port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the second electromagnetic wave.
The present invention makes use of an understanding that the signals induced by the magnetic and electric components of an electromagnetic wave (or high-frequency electromagnetic field such as microwaves) travelling from the input port to the output port of the main transmission line added at one of the ports of the coupler line (i.e. the first port located nearest from the input port of the main transmission line) and subtracted at the opposite port of the coupler line (i.e. the second port located nearest from the output port of the main transmission line). The present invention is based on the idea that an electronic means may be connected between the first port and the second port of the coupler line for electrically isolating the second port from the first electromagnetic wave and for electrically isolating the first port from the second electromagnetic wave. In other words, the electronic means is configured to reduce the output signal induced at the first port by the second electromagnetic wave and to reduce the output signal induced at the second port by the first electromagnetic wave. For a specific electromagnetic wave (i.e. travelling in a specific direction), the electronic means has the effect of reinforcing the output signal at one end of the coupler line and reducing (preferably minimizing and even more preferably cancelling) the output signal at the opposite side. More specifically, the electronic means is configured to reduce, at the second port, the difference between the signal induced by the magnetic component and the oppositely directed signal induced by the electric component of an electromagnetic field coupled to the coupler line and originating from the first electromagnetic wave (e.g. incident microwaves). Similarly, the electronic means is configured to reduce, at the first port, the difference between the signal induced by the magnetic component and the signal induced by the electric component of an electromagnetic field coupled to the coupler line and originating from the second electromagnetic wave (e.g. reflected microwaves). As a result, the energy of the first electromagnetic wave can be measured (or monitored) at the first port (i.e. the port nearest the input port) and the energy of the second electromagnetic wave can be measured (or monitored) at the second port (i.e. the port nearest the output port) with good directivity.
The electronic means is linear (or operates linearly), i.e. not using frequency conversion of the signals detected at the first and second ports. In the present application, the term linear refers generally to the class of electronic components maintaining the frequency of the signals unchanged, in contrast to other components or electronic circuitry based on frequency conversion. The electronic means may be at least one electric component, i.e. it may be a single electric component (such as a capacitor). The operation of the linear electronic means does not, within a typical operating range, depend on the level of power in the main transmission line. The term linear electronic means is intended to cover passive electronic components such as, e.g., capacitors and resistances.
The directional coupler of the present invention is original and advantageous over traditional directional couplers in that it provides a directional coupler with four active ports. In other words, the directional coupler of the present invention provides, at the first port of the coupler line, a signal representative of the electromagnetic wave travelling from the input port to the output port (i.e. typically a signal representative of incident microwaves) and, at the second port of the coupler line, a signal representative of the electromagnetic wave travelling from the output port to the input port (i.e. typically a signal representative of reflected microwaves). Thus, as compared to prior art devices, there is no permanently isolated or terminated port and a single directional coupler is sufficient for monitoring both incident and reflected microwaves, which is less cumbersome and less expensive than with prior art directional couplers. In addition, assembly of a final device (e.g. a microwave oven or a mobile phone) including such a monitoring function of two electromagnetic waves travelling in opposite directions in a main transmission line is facilitated since a single directional coupler may suffice.
The present invention is also advantageous in that a microwave directional coupler with relatively small dimensions, i.e. typically smaller than the quarter-wavelength of the microwaves, may be achieved.
In the present invention, it has been realized that the above mentioned advantages may be achieved by connecting an electronic means between the first port and the second port of the coupler line. For connection of the electronic components of the electronic means, the coupler line may be bent or arranged such that the distance between the first port and the second port is smaller than a quarter-wavelength of the first electromagnetic wave (i.e. such that the first and second ports are arranged close to each other). One end of the electronic means may then be connected to a first portion of the coupler line comprising the first port and another end of the electronic means may be connected to a second portion of the coupler line comprising the second port. Any discontinuities introduced by a swift bending of the coupler line is then compensated by adjusting the values of the (lumped) components of the electronic means.
As will be illustrated in more detail in the following description, the present invention is also advantageous in that it provides a directional coupler with improved characteristics (directivity, isolation and coupling).
In particular, the electronic means may be configured to reduce (and preferably to minimize), at the first port, the difference between the phases and the difference between the amplitudes, in absolute values, of the oppositely directed signals induced by the magnetic and electric components of the coupled field originating from the second electromagnetic wave and, at the second port, the difference between the phases and the difference between the amplitudes, in absolute values, of the oppositely directed signals induced by the magnetic and electric components of the coupled field originating from the first electromagnetic wave. In other words, the electronic means connected between the first and second ports of the coupler line is configured to balance the amplitudes and phases of the output signals. Connection of electronic components between the two ports result in new degrees of freedom for balancing the amplitudes and phases of the output signals.
According to an embodiment, the electronic means may comprise a capacitor (or shunt capacitor), which provides the ability to design a directional coupler providing improved characteristics (in particular isolation and directivity) for a chosen bandwidth at high frequencies, for instance in the ISM band of 2.45 GHz. Examples of values for the shunt capacitor will be provided in the following detailed description.
According to another embodiment, the electronic means may further comprise a resistance (or shunt resistance) connected in parallel to the shunt capacitor, which is further advantageous in that it provides a design (with an adjustable shunt resistance) for further improving the characteristics of the directional coupler.
According to yet another embodiment, the directional coupler may further comprise a first resistance coupled between the first port and a reference potential, a second resistance coupled between the second port and the reference potential, a first capacitor coupled between a first portion of the coupler line and the reference potential, and a second capacitor coupled between a second portion of the coupler line and the reference potential. The first portion of the coupler line (also called first arm of the coupler line below) comprises the first port while the second portion of the coupler line (also called second arm of the coupler line below) comprises the second port. In particular, the first resistance may be configured to provide a voltage output representative of the power of the first electromagnetic wave (e.g. incident waves) and the second resistance may be configured to provide a voltage output representative of the power of the second electromagnetic wave (e.g. reflected waves). For example, the first, second and shunt capacitors may include at least one of a group comprising discrete capacitors, distributed capacitors, trimmer capacitors and open circuit stubs.
Such embodiments are advantageous in that the electronic means (or electronic circuitry) is relatively simple and thereby cheap.
Generally, the present invention provides a directional coupler wherein the design of the electronics is of low complexity. Further, the present invention facilitates assembly of the direction coupler as a limited number of components are needed. In addition, all components may be connected on the same printed circuit board.
According to a further embodiment, the directional coupler may comprise a first attenuator coupled to the first resistance and a second attenuator coupled to the second resistance. The attenuators are advantageous in that they provide impedance matching for connection of the directional coupler with a further detecting device (external or integrated in the directional coupler) configured to measure the respective voltage outputs at the first and second ports.
According to an embodiment, the second electromagnetic wave may correspond to a part of the first electromagnetic wave reflected back into the main transmission line via the output port of the main transmission line. Indeed, part of the first electromagnetic wave (e.g. incident wave) transmitted out of the transmission line via the output port may return into the transmission line (as so called reflected waves) via the same (output) port.
According to an embodiment, the main transmission line may be a waveguide transmission line and the electronic means may be mounted on a printed circuit board adapted to constitute part of a wall of the waveguide transmission line, which is advantageous in that the assembly and/or mounting of a final device including the inventive directional coupler may be facilitated. In the present embodiment, the directional coupler may be integrated in the wall of the waveguide transmission line (of for instance a microwave oven), which also provides a compact solution.
According to an embodiment, the coupler line may comprise a section (longitudinally) facing the main transmission line for electromagnetic coupling and two arms (or portions) departing from each ends of the section. The first arm (or first portion) comprises the first port and the second arm (or second portion) comprises the second port.
The geometry of the section of the coupler line longitudinally facing the main transmission line determines the level of electromagnetic coupling between the coupler line and the main transmission line. Preferably, the length of the section is smaller than quarter-wavelength but may be in the order of a quarter-wavelength for improving the coupling. Further, the distance between the first port and the second port is smaller than a quarter-wavelength for enabling connection of the electronic means.
As will be further illustrated in the following detailed description, the coupler line may have a generally U-shaped outline (or layout) or an outline corresponding substantially to the shape of the Greek upper case letter Omega or the Greek upper case letter Pi, thereby providing a section longitudinally facing the main transmission line for electromagnetic coupling and two arms extending from the section. In these non-limiting specific examples, the two arms depart from the section at a substantially right angle. However, it is not necessary that the two arms form a right angle with the section of the coupler line longitudinally facing the main transmission line. It will be appreciated that other angles may be envisaged, the purpose being that the first and the second ports are arranged close to each other, thereby facilitating the connection of the electronic means (e.g. the shunt capacitor or the shunt capacitor connected in parallel with the shunt resistance) between the first port and the second port of the coupler line. Using a bent coupler line, the size of the directional coupler is reduced, which is advantageous for applications wherein space is limited, such as in mobile phones for examples.
The main transmission line and the coupler line may include at least one of a group comprising microstrip transmission lines, stripline transmission lines and waveguide transmission lines.
The directional coupler of the present invention may be used for monitoring power of incoming and outgoing electromagnetic waves in, e.g., a radio transmitter, a radio receiver, a transceiver (operating for example as a base station or terminal in a telecommunication system) or a microwave heating device.
It is noted that the invention relates to all possible combinations of features recited in the claims.

BRIEF DESCRIPTION OF THE DRAWINGS

This and other aspects of the present invention will now be described in more detail, with reference to the appended drawings showing various exemplifying embodiments of the invention.

Fig. 1 is a schematic view of the basic structure of a prior art directional coupler;
Fig. 2a-2c are schematic views of directional couplers in accordance with exemplifying embodiments of the present invention;
Fig. 3 is a schematic view of an equivalent circuit for a short segment of the coupler line of a directional coupler according to an exemplifying embodiment of the present invention;
Fig. 4 is a schematic view of an equivalent circuit for a short-loop directional coupler according to an exemplifying embodiment of the present invention;
Fig. 5 is a schematic view of an equivalent circuit for a short-loop directional coupler and exemplifying electronic means according to an exemplifying embodiment of the present invention;
Fig. 6 is a schematic view of a directional coupler according to an exemplifying embodiment of the present invention implemented for a short-loop microstrip coupler;
Fig. 7 is a graph showing the results of simulations for the short-loop microstrip coupler of Figure 6 assuming absence of the shunt capacitor and the shunt resistance;
Fig. 8 is a graph showing the results of simulations for the short-loop microstrip coupler of Figure 6 assuming presence of the shunt capacitor and absence of the shunt resistance;
Fig. 9 is a graph showing the results of simulations for the short-loop microstrip coupler of Figure 6 assuming presence of both the shunt capacitor and the shunt resistance;
Fig. 10 is a graph showing the results of simulations for the short-loop microstrip coupler of Figure 6 optimized for a band centred at 3.5 GHz;
Fig. 11 is a schematic view of a directional coupler according to an exemplifying embodiment of the present invention implemented for a short-loop microstrip coupler with attenuators;
Fig. 12 is a graph showing the results of simulations for the short-loop microstrip coupler of Figure 11;
Fig. 13 is a schematic view of a directional coupler according to an exemplifying embodiment of the present invention implemented for a quarter-wavelength microstrip coupler;
Fig. 14 is a graph showing the results of simulations for the directional coupler of Figure 13;
Fig. 15 is a schematic view of a directional coupler according to an exemplifying embodiment of the present invention implemented for a small-size microstrip short-loop directional coupler;
Fig. 16 is a graph showing the results of simulations for the directional coupler shown in Figure 15;
Fig. 17 is a schematic view of a directional coupler according to an exemplifying embodiment of the present invention implemented for a stripline short-loop directional coupler;
Fig. 18 is a schematic view of the printed circuit board (or electronic circuitry) of the stripline short-loop directional coupler shown in Figure 17 and a schematic view of the loop;
Fig. 19 is a graph showing the results of simulations for the directional coupler shown in Figures 17 and 18;
Fig. 20 is a further graph showing the results of simulations for the directional coupler shown in Figures 17 and 18;
Fig. 21 is a graph showing the results of simulations and measurements for a directional coupler according to an exemplifying embodiment of the present invention implemented for a waveguide short-loop directional coupler using the same type of structure as the stripline directional coupler shown in Figure 18;
Fig. 22a and 22b are schematic views showing possible ways of realization of capacitors for directional couplers of the present invention; and
Fig. 23a and 23b are schematic views showing possible ways of realization of the shunt capacitor for directional couplers of the present invention.

DETAILED DESCRIPTION

Referring to Figures 2a-2c, schematic views of directional couplers according to exemplifying embodiments of the present invention are described.
Figures 2a-2c show directional couplers 200, 220 and 240, each comprising a main transmission line 110 and a coupler line 120 arranged proximate to the main transmission line 110 such that energy flowing in the main transmission line 110 can be coupled to the coupler line 120. The main transmission line 110 comprises an input port 111 and an output port 112 such that an electromagnetic wave can be transmitted from the input port 111 to the output port 112. In the following, the first electromagnetic wave will refer to an electromagnetic wave travelling from the input port 111 of the transmission line 110 to the output port 112 of the transmission line (e.g. incident microwaves) and the second electromagnetic wave will refer to an electromagnetic wave travelling from the output port 112 of the transmission line 110 to the input port 111 of the transmission line (e.g. reflected microwaves).
The coupler line 120 comprises two ports or terminals 121 and 122, namely a first port 121 at which energy representative of the first (incident) electromagnetic wave is detected and a second port 122 at which energy representative of the second (reflected) electromagnetic wave is detected.
The electronic means 130 is linear, i.e. not using frequency conversion. The electronic means 130 is connected between the first port 121 of the coupler line 120 (or a first portion 127 of the coupler line 120 comprising the first port 121) and the second port 122 of the coupler line 120 (or a second portion 128 of the coupler line 120 comprising the second port 122). The electronic means is configured to reduce the output signal induced at the second port by the first electromagnetic wave and to reduce the output signal induced at the first port by the second electromagnetic wave. More specifically, the electronic means 130 is configured to reduce, at the second port 122, the difference between the oppositely directed signals induced by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line 120 and originating from the first electromagnetic wave. The electronic means 130 is further configured to reduce, at the first port 121, the difference between the oppositely directed signals induced by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line 120 and originating from the second electromagnetic wave.
In particular, the electronic means 130 is configured to reduce, at each of the first and second ports 121 and 122, the difference between the phases and the difference between the amplitudes, in absolute values, of the signals induced by the second and the first electromagnetic waves, respectively.
The coupler line 120 comprises a first portion or arm 127 including the first port 121 and a second portion or arm 128 including the second port 122.
Generally, the coupler line 120 comprises a section 129 (longitudinally) facing the main transmission line 110 for electromagnetic coupling between the coupler line 120 and the main transmission line 110. The section 129 of the coupler line 120 is arranged proximate to the main transmission line 110. The coupler line 120 further comprises two portions 127 and 128 departing from the section 129. The first portion 127 of the coupler line 120 comprises the first port 121 while the second portion 128 of the coupler line 120 comprises the second port 122.
In this configuration, the level of coupling is determined by the length of the section 129 and the distance between the section 129 and the main transmission line 110. Generally, electromagnetic coupling reaches a maximum if the length of the section 129 is in the order of the quarter-wavelength of the electromagnetic wave. Further, the distance between the first port 121 and the second port 122 affects the connection of the electronic means 130 between the two portions 127 and 128 of the coupler line 120. It is then preferable if the length of the section 129 is smaller than or in the order of a quarter-wavelength and the distance between the first port 127 and the second port 128 is smaller (or even much smaller) than a quarter-wavelength of the first electromagnetic wave.
Examples of directional couplers with such a coupler line are illustrated in Figures 2a-2c, wherein the coupler line 120 may have an outline substantially shaped as the Greek upper case letter Pi (Figure 2a), the letter U (Figure 2b, inverted U) or the Greek upper case letter Omega (Figure 2c). Such configurations facilitate the connection of the electronic means 130 between the first port 121 and the second port 122 of the coupler line 120. In these examples, the coupler line 120 comprises a section 129 longitudinally facing the main transmission line 110 for electromagnetic coupling and two arms 127 and 128 extending away from the section 129. The two arms 127 and 128 form a substantially right angle (close to 90°) with the longitudinal section 129. However, it is not necessary that the two arms 127 and 128 depart at a right angle from the section 129 of the coupler line. Other angles may be envisaged, in particular for the configurations shown in Figures 2a and 2b in order to reduce the distance between the first port 121 and the second port 122 for facilitating the connection of the electronic means 130. However, a too swift bending of the coupler line may affect transmission of the electromagnetic wave in the coupler line. The configuration shown in Figure 2c is advantageous in that the length of the section 129 of the coupler line 120 may be selected independently of the distance between the first port 121 and the second port 122. Any discontinuity introduced by the bending of the coupler line 120 in these configurations and any other equivalent configurations can be compensated by adjusting the values of the (lumped or quasi-lumped) components of the electronic means 130.
In the contrary to traditional prior art directional couplers in which the main transmission line and the coupler line are of equal width (with impedance values of about 50 Ω for typical microstrip couplers), there is no need for equal widths of the main transmission line 110 and the coupler line 120 in the directional coupler as defined by the present invention. A change of the width of the coupler line 120 may be compensated by adjusting the values of the (lumped or quasi-lumped) components of the electronic means 130 of the directional coupler.
Further, referring again to Figures 2a-2c, it is not necessary that the main transmission line 110 and the section 129 of the coupler line 120 are strictly parallel. Any deviation (such as e.g. with the U-shaped outline of Fig. 2b or an even more pronounced deviation) may be compensated by adjusting the values of the components of the electronic means 130.
Referring to Figure 3, assuming that an electromagnetic wave travels in the main transmission line 110 from left (input port 111) to right (output port 112), an equivalent circuit 300 of the coupler line 120, which may be a loop or a short section of a lossless TEM transmission line, is represented. The loop 120 may be represented, or modelled, by an equivalent circuit 300 comprising a series inductance 323 (equal to unit inductance of the coupler line 120 multiplied by the length of the section or segment 129 of the coupler line 120 at which there is coupling) in connection with a parallel capacitance 324 (equal to unit capacitance of the coupler line 120 multiplied by the length of the section or segment 129 of the coupler line 120 at which there is coupling). The signal coupled from the main transmission line 110 to the loop 120 by electromagnetic coupling may be represented by a current source 325 and a voltage source 326. The parallel-connected current source 325 models the electric coupling and the series-connected voltage source 326 models the magnetic coupling. In this model, the effects of the electric coupling and the magnetic coupling are added on the left-hand side of the equivalent circuit 300 and are subtracted from each other on the right-hand side of the equivalent circuit 300.
Referring to Figure 4, a schematic view of an equivalent circuit for a short-loop directional coupler 400 according to an exemplifying embodiment of the present invention is shown. The coupler line 120 is represented by an equivalent circuit 420 similar to the equivalent circuit 300 described with reference to Figure 3 except that the inductance and voltage source 323 and 326 are distributed to the left-(323a and 326a) and right-(323b and 326b) hand sides of the equivalent circuit 420 in a symmetrical manner. Further, the coupler line 120 (i.e. the equivalent circuit 420) is connected to an electronic circuit or electronic means 130 configured to reduce the signal coming from (or read at) its right-hand side, i.e. at the branch wherein the output signals induced by the electric and magnetic coupling are subtracted. Thus, referring to e.g. Figures 2a-2c, the electronic means 130 is configured to reduce, for the first electromagnetic wave travelling from the input port 111 to the output port 112 of the main transmission line 110, the output signals at the second port 122 of the coupler line 120.
In particular, the signal at the right-hand side of the equivalent circuit 420 will be reduced if the electric coupling and the magnetic coupling are equal in magnitude and phase.
Similarly, the electronic means 130 is configured to reduce, for the second electromagnetic wave travelling from the output port 112 to the input port 111 of the main transmission line 110, the output signal induced at the first port 121 of the coupler line 120.
According to an embodiment of the present invention, the electronic means 130 of the directional coupler comprises a shunt capacitor 131 connected between the first port 121 and the second port 122 of the coupler line 120 (or between the first portion 127 and the second portion 128 of the coupler line 120).
In addition, the electronic means 130 may further comprise a shunt resistance 132 connected in parallel to the shunt capacitor 131. The effects and influences of these two components on the characteristics of the directional coupler will be further illustrated in the embodiments described in the following with reference to Figures 6-10.
Referring to Figures 5 and 6, the directional coupler, or more specifically its electronic circuitry, may further comprise a first resistance 433 connected between the first port 121 and a reference potential (denoted 428 in e.g. Figure 6), a second resistance 434 connected between the second port 122 and the reference potential, a first capacitor 435 connected between the first portion 127 of the coupler line 120 and the reference potential 428, and a second capacitor 436 connected between the second portion 128 of the coupler line 120 and the reference potential. Further, as mentioned above, the electronic circuit 130 comprises a shunt capacitor 131 connected between the first portion 127 and the second portion 128 and a shunt resistance 132 connected in parallel to the shunt capacitor 131 between the first branch 127 and the second branch 128.
The reference potential 428 is typically a ground potential but may also be any other reference potential.
In the present embodiment, the output signals, i.e. the signals representative of the energy level of the first electromagnetic wave and the energy level of the second electromagnetic wave, are read at (or further transmitted from) the first and second resistances 433 and 434, respectively.
For the purpose of explanation of the present invention, in the following, the first and second resistances 433 and 434 are referred to as R_1L and R_1R, respectively; the first and second capacitors 435 and 436 are referred to as C_1L and C_1R, respectively; and the shunt capacitor 131 and the shunt resistance 132 are referred to as C_sh and R_sh, respectively. These denotations will also be used in the following examples for specifying the values selected for these components. Further, the current density provided by the current source 325 is denoted J_E, the electromotive force at each one of the voltage sources 326a and 326b is denoted 0.5×E_M while the inductances 323a and 323b are denoted L_L and L_R respectively.
The voltage drop at the first and second resistances R₁ (433 or 434) due to the source J_E will be denoted as U_E and is a function of J_E, C_L L_L R₁ and C₁ (due to the symmetry R_sh and C_sh has a limited effect on U_E). It will be appreciated that U_E has the same direction (in vector form) at R_1L and R_1R. The voltage drop at the resistance R₁ (L or R, i.e. 433 or 434) due to the sources 326a and 326b (each equal to 0.5xE_M) will be denoted as U_M and is a function of E_M, L_L, R₁, C₁, R_sh and C_sh (due to the symmetry C_L has a limited effect on U_M). U_M has opposite directions at R_1L and R_1R. The voltage drop U1_L measured at the resistance R_1L corresponds to the sum of the signals induced by the electric coupling (U_E) and the magnetic coupling (U_M), such that U_1L=U_E + U_M, while the voltage drop U_1R measured at the resistance R_1R corresponds to the difference of both signals such that U_1R=U_E - U_M since the signals induced by the electric and the magnetic coupling are oppositely directed (i.e. the voltage vector representative of the contribution of the electric coupling at R_1R is oppositely directed to the voltage vector representative of the contribution of the magnetic coupling at R_1R). The voltages U_1R, U_1L, U_E and U_M are complex numbers and, thus, U_LR=0 when the voltages U_E and U_M are equal in both amplitude and phase. In electromagnetic modelling, the properties of U_E and U_M can be monitored and they can be expressed as U_E=0.5×(U_1L + U_1R) and Um=0.5×(U_1L - U_1R). As U_E and U_M can be considered to have clear physical properties, their behaviour versus changing parameters can be predicted by simulation. In the following, improvement (or optimization) of component values for improved directivity and isolation will be described in more detail for specific examples. Using simulation, components values can be optimized (or at least improved) by balancing the amplitudes and the phases of the output signals. In particular, the amplitude ratio (|U_E|/|U_M|) is preferably maintained close to unity and the phase difference (Arg(U_E)-Arg(U_M)) is preferably reduced (and even more preferably, minimized to about zero).
In Figure 4, the first and second capacitors C_1L and C_1R are connected in parallel to the first and second resistances R_1L and R_1R. As compared to prior art directional couplers in which resistances are coupled at the respective ports of the coupler line, the first and second resistances R_1L and R_1R are mounted in parallel to each other in the present invention such that at least a shunt capacitor C_sh, and preferably also a shunt resistance R_sh, can be connected between the resistances 433 and 434 or, the arms 127 and 128 of the coupler line 120. The shunt resistance 132 (R_sh) and the shunt capacitor 131 (C_sh) are advantageous in that they affect principally the magnetic coupling, i.e. the level of U_M, and very little the electric coupling, i.e. the value of U_E. Thus, these two components R_sh and C_sh provide two degrees of freedom in the process of balancing the amplitude and the phase of U_E and U_M.
As compared to prior art directional couplers in which resistances connected to the ports of the coupler line are configured to be purely resistive and equal to the characteristic impedance of the coupler line, the resistances (R _1L 433 and R_1R 434) of the directional coupler of the present invention may be subject to a significant parasitic series inductance (which is typical in the case of lumped packaged resistances). In addition, these resistances can be connected to the coupler line by a segment of a transmission line of the characteristic impedance different from the impedance of the coupler line. As a result, four degrees of freedom, namely the value of the first and second resistances 433 and 434 (R_1L =R_1R), the value of the first and second capacitors 435 and 436 (C_1L =C_1R), the value of the shunt resistance 132 (R_sh) and the value of the shunt capacitor 131 (C_sh) can be used to optimize the directional coupler from the point of view of its centre frequency, bandwidth of application and directivity. It will be appreciated that these electronic components have a negligible (or at least little) effect on the coupling coefficient, which is mainly determined by the geometry of the coupler line 120.
For each specific application or example, the goal is then to reduce the difference of U_E and U_M in amplitude (amplitude ratio (|U_E|/|U_M|) close to 1) and in phase (Arg(U_E)-Arg(U_M) close to 0) for the intended frequency band of interest. Normally, for specific values of R_1L, R_1R (with R_1L = R_1R) and R_sh, the value of the third component (for example C₁L=C_1R) is used for adjusting the amplitude ratio and the value of the fourth one (in the present example C_sh) is used for adjusting the difference between the phases, i.e. Arg(U_E)-Arg(U_M).
Referring to for instance Figures 5 and 6, in the directional coupler of the present invention, the electrical components of the electronic means 130 are symmetrically distributed between the first port 121 and the second port 122 for adjusting the level of the output signals induced by the magnetic and electric components of the coupled electromagnetic field.
In the following description, a number of examples for various implementations of the present invention are described.

Example 1: A short-loop microstrip directional coupler

With reference to Figures 6-10, a short-loop microstrip directional coupler according to an exemplifying embodiment of the present invention is described.
Figure 6 shows a microstrip directional coupler 600 wherein the transmission line 110 and the coupler line 120 are fabricated using printed circuit board (PCB) technology. The PCB consists of a dielectric forming a substrate having two conducting layers, one on each side (typically referred to as a top layer and bottom layer). The coupler line 120 (or conducting microstrip line) may then be made in the top conducting layer and separated from a ground plane, the bottom conducting layer, by the substrate. The electronic means 130 (represented by the shunt capacitor 131 and the shunt resistance 132 in Figure 6) is arranged on the same PCB. In Figure 6, the white-filled areas represent the pattern of metallization on the substrate (i.e. principally the transmission line 110 and the coupler line 120) while the black-filled areas represent the electronic components (e.g. the shunt capacitor 131 and the shunt resistance 132). The circuitry of the microstrip directional coupler 600 shown in Figure 6 is identical to the circuitry already described above with reference to Figure 5. The ground plane corresponding to the reference potential is denoted 428. The same type of representation will be used in Figures 11, 13, 15, 18 and 21.
The microstrip directional coupler 600 is further characterised in that it comprises a transmission line 110 comprising an input port 111 and an output port 112 and a coupler line 120 comprising a first port 121 (actually not denoted in Figure 6 since the first resistance 433 is connected to it) and a second port 122 (actually not denoted in Figure 6 since the second resistance 434 is connected to it). The coupler line 120 comprises a section 129 longitudinally facing the transmission line 110 for electromagnetic coupling and a first portion 127 comprising the first port 121 and a second portion 128 comprising the second port 122. The coupler line 120 has an outline substantially shaped as the Greek letter Pi such as in the embodiment described above with reference to Figure 2a. However, the thickness of the coupler line 120 at the first and second portions 127 and 128 has been adjusted (in the present example increased) to facilitate the connection of the electronic means 130, represented as a shunt capacitor 131 and a parallel-connected shunt resistance 132, between the two portions 127 and 128.
Based on the directional coupler 600 of the present embodiment, for the purpose of simulation, a number of assumptions and specific geometrical data (or design selections) are taken into account. It will of course be appreciated that the present invention, and in particular the present embodiment based on the directional coupler 600 shown in Figure 6, is not limited to such assumptions and specific geometrical data. Such assumptions and specific geometrical data may vary in function of the application in which the directional coupler is used.
For instance, the microstrip lines are assumed to be arranged on a 1.54 mm thick substrate having a relative permittivity equal to 10, which corresponds to a rather difficult environment as these conditions produce high values of parasitic components. Further, a 50 Ω main transmission line (about 1.5 mm wide) is electromagnetically coupled to a 0.5 mm wide loop having an external length equal to 4 mm. Moreover, it is assumed that the packages of the lumped components (resistances and capacitors) of the electronic circuit have a series inductance of 0.7 nH and that a pure incident wave travels from the input port 111 (left) to the output port 112 (right) of the main transmission line 110.
The simulation results obtained for different values of the lumped components are shown in Figures 7-10. In the figures, the scale of the y-axis is expressed in dB. The continuous lines represent the directivity (wherein the thick and the thin lines are used for distinguishing different cases) and the dashed and dotted lines represent the coupling (also used for distinguishing two cases, the dashed line corresponding to the same case as the thick line and the dotted line corresponding to the same case as the thin line).
In a first simulation, for the purpose of illustration of the present invention (by comparison with further simulations), the circuitry of the directional coupler 600 is only designed with resistances 433 and 434 (R_1L=R_1R) and capacitors 435 and 436 (C₁L =C_1R) but without the components 132 (R_sh) and 131 (C_sh). The results of the simulation (optimization with R_1L=_R1R=48Ω and C_1L=C_1R=0.4 pF) are shown in Figure 7 wherein the continuous line represents the directivity and the dashed line represents the coupling, both expressed in dB, as a function of the frequency of the first electromagnetic wave (in GHz). As can be seen, using these component values, the directional coupler 600 (without the components 132 and 131) provides relatively high directivity for low frequencies but significantly lower directivity at frequencies above 2 GHz.
In a further simulation, only the shunt resistance 132 (R_sh) is removed and the shunt capacitor 131 (C_sh) may be varied so one degree of freedom is added for optimizing the directional coupler. Two cases (i.e. the results for two different sets of component values) are shown in Figure 8. The first case, wherein the thick continuous line represents the directivity and the dashed line represents the coupling in Figure 8, corresponds to the result of optimization for low frequencies with R_1L=R_1R=48 Ω, C_1L=C_1R=0.36 pF and C_sh=0.28 pF. The second case, wherein the thin continuous line represents the directivity and the dotted line represents the coupling in Figure 8, corresponds to the result of optimization for the ISM band (i.e. at 2.45 GHz) with R_1L=R_1R=48 Ω, C_1L=C_1R=0.1 pF and C_sh=0.32 pF. As can be seen, the capacitance C_sh allows an effective design of the directional coupler for high frequencies. The directivity is higher than 30 dB in a band of 13.8%.
In a further simulation, the shunt resistance R_sh is reintroduced in the design of the directional coupler 600, i.e. the simulated directional coupler now corresponds identically to the design shown in Figure 6. The result of the simulation is shown in Figure 9 wherein the result of the second case shown in Figure 8 (i.e. the case for optimization at the ISM band with the thin line representing the directivity and the dotted line representing the coupling) has been reproduced for comparison. With the shunt resistance 132, the optimization is obtained with the following component values: R_1L=R_1R=70 Ω, C_1L=C_1R=0.245 pF, C_sh=0.345 pF and R_sh=125 Ω. As can be seen in Figure 9, wherein the thick continuous line represents the directivity and the dashed line represents the coupling, the possibility of modifying the value of the shunt resistance 132 (R_sh), i.e. the addition of a degree of freedom for balancing the signals in the directional coupler 600, results in a slightly wider and more symmetric characteristic of directivity. Moreover, a much more flat characteristic of the coupling is achieved. Thus, the use of both the shunt capacitor 131 and the shunt resistance 132 results in an improved directivity and an improved coupling.
In yet a further simulation, the directional coupler 600 was optimized for operation at even higher frequencies, for instance 3.5 GHz. In Figure 10, the thick line represents the directivity and the dashed line represents the coupling for a first case with all four degrees of freedom wherein R_1L=R_1R=48 Ω, C_1L=C_1R=0.175 pF, C_sh=0.2 pF and R_sh=125 Ω. Figure 10 shows also a thin line representing the directivity and a dotted line representing the coupling for a second case without the shunt resistance 132 (R_sh) wherein R_1L=R_1R=48 Ω C_1L=C_1R=0 pF and Csh=0.2 pF. As can be seen, the shunt resistance 132 (R_sh) provides a much wider band of high directivity and an almost completely flat characteristic of coupling. Connection of an adjustable shunt resistance 132 (R_sh) in parallel to the shunt capacitor 131 provides the possibility of designing a directional coupler with improved characteristics (in particular a wider and more symmetric band of high directivity and flatter characteristic of coupling), especially for high frequencies.
It will also be appreciated that, despite the difficult environment selected for the simulation (microstrip arranged on a thick substrate having a high permittivity), the directional coupler 600 according to the present embodiment of the present invention allows optimization for a wide range of frequencies. Thus, the present invention is advantageous in that it provides a directional coupler which is robust, i.e. applicable in different technologies, and flexible, i.e. different optimization targets may be found.
With reference to Figures 11 and 12, another embodiment of the present invention is described.
Figure 11 shows a schematic view of a directional coupler 1100 according to another exemplifying embodiment of the present invention implemented for a short-loop microstrip directional coupler with attenuators. The microstrip directional coupler 1100 is identical to the microstrip directional coupler 600 described with reference to Figure 6 except that it also comprises four additional resistances or impedances, as further explained below, to form attenuators. Indeed, the performance of a four-port directional coupler depends on the particular values of the first and second resistances 433 and 434 (R_1L and R_1R). Normally, these resistances are treated as input resistances of the next stage device of the circuit to which the direction coupler is connected. These next stage devices may correspond to detectors or amplifiers and their performance may depend on DC polarization and amplitude of the signal. The sensitivity of the directional coupler to the changes of the input impedance of the next stage devices can be reduced by supplementing (at the left and right sides of the circuit such as described with reference to Figure 6) the first resistance 433 with two resistances, denoted as resistances 437 and 439 (also referred to as R_2L and R_3L in the following) in Figure 11, thereby forming a first TT-type attenuator. Similarly, the second resistance 434 on the right-hand side of the circuit is supplemented with two other resistances, denoted as resistances 438 and 440 (also referred to as R_2R and R_3R in the following) in Figure 11, thereby forming a second TT-type attenuator. Alternatively, the resistances 437 and 438 (R_2L and R_2R) may be replaced by an impedance (like an RL circuit, i.e. a non-purely resistive component). If the relative changes of the impedance seen from the circuit connected to the directional coupler 1100 need to be reduced by a factor of 10 (for example from 10 % to 1 %), a 10 dB attenuator may be applied.
Normally, the attenuator is not purely resistive since it incorporates parasitic elements like connecting pads, bonding wires and component packages. After connecting an attenuator, a correction of the values of the electronic components of the directional coupler 1100 may be needed. The results of simulations run for two cases are shown in Figure 12. In a first case, a TT-type attenuator as resistive as possible (although with some realistic parasitic components of the lumped elements) is assumed and optimization is obtained for the following values: R_1L=R_1R=70 Ω, C_1L=C_1R=0.14 pF, R_sh=125 Ω, C_sh=0.06 pF, R_2L=R_2R=250 Ω (in series with an inductance equal to 1 nH), and R_3L=R_3R=50 Ω. In Figure 12, the results are shown by a thick line for representing the directivity and a dashed line for representing the coupling. As can be seen, a wide-band directivity with a relatively high level (about 600 MHz bandwidth with a directivity higher than 30 dB) is achieved. Thus, the use of an attenuator further improves the directivity. However, the attenuators have reduced the coupling to about 32 dB, thereby reducing the sensitivity to changes at a level of 50 Ω output impedance accordingly.
In a second case, a significant inductance connected in series with the resistances 437 and 438 (R_2L and R_2R) is assumed and optimization is obtained for R_1L=R_1R=70Ω, C_1L=C_1R=0.32 pF, R_sh=125 Ω, C_sh=0.1 pF, R_2L=R_2R=250 Ω (in series with an inductance equal to 20 nH) and R_3L=R_3R=50 Ω. In Figure 12, the result of the second case is represented by a thin line for the directivity and a dotted line for the coupling. As can be seen, a very flat coupling versus frequency (at the level of about 37dB) is obtained. However, the band of high directivity is rather narrow (about 240 MHz with a directivity higher than 30 dB).
In conclusion, although a lower output signal is obtained from the directional coupler (due to a lower coupling), a TT-type attenuator reduces the sensitivity of the directional coupler to changes of the input impedance of the next stage device connected to the circuit. Additionally, if the TT-type attenuators are not purely resistive, improvement of the characteristics of the directional coupler, for example an increase of its bandwidth or a reduction of the variation of the coupling with frequency, may be achieved as a result of the four degrees of freedom (R₁,C₁,Rsh, C_sh) available in the directional coupler of the present invention.

Example 2: Quarter-wavelength microstrip directional coupler

As already mentioned above, a typical backward directional coupler has a length equal to a quarter of the wavelength for optimizing the electromagnetic coupling. However, it is commonly known that, in order to obtain a high directivity in a quarter-wavelength coupler (based on a structure such as that shown in Fig.1), it is preferable to have equal velocity for the even and odd mode waves travelling in the section (or segment) of the coupler line 120 at which there is electromagnetic coupling with the main transmission line 110. Such a condition is automatically obtained in a homogeneously filled transmission line (like a stripline) but not in an non-homogeneously filled transmission line such as a microstrip line. Various methods are traditionally used for correcting the difference in speed of the odd and even modes of the coupled sections of a microstrip line such that the directivity of the directional coupler is improved. In the following, an alternative solution according to an exemplifying embodiment of the present invention is provided.
With reference to Figures 13-14, a quarter-wavelength microstrip directional coupler 1300 according to an exemplifying embodiment of the present invention is described.
The quarter-wavelength microstrip directional coupler 1300 shown in Figure 13, and in particular its circuitry, is identical to the directional coupler 600 described with reference to Figure 6 except that the outline of the coupler line 120 is different. The coupler line 120 of the directional coupler 1300 is bent in an Ω form such that the first port 121 and the second port 122 (i.e. the ends) of the coupler line 120 are close enough to connect lumped components between them. The coupler line 120 has therefore an outline substantially shaped as the Greek letter Omega such as shown in Figure 2c such that the length of the section 129 of the coupler line (section longitudinally facing the transmission line 110) is in the order of a quarter-wavelength and the distance between the first port 121 and the second port 122 of the coupler line 120 may be much smaller than a quarter-wavelength. As for the embodiment described with reference to Figure 6, the thickness of the coupler line 120 at the first and second portions 127 and 128 has been adjusted (in the present example increased) to facilitate the connection of the electronic means 130, represented as a shunt capacitor 131 and a parallel-connected shunt resistance 132.
For the purpose of electromagnetic simulation, the following assumptions and geometrical data have been taken into account and selected. In particular, the substrate is selected to be alumina with a relative permittivity of 9.7 and a thickness of 1.54 mm. The main transmission line 110 is assumed to be 1.5 mm wide while the coupler line 120 is assumed to be 0.8 mm wide. The distance or the gap between the main transmission line 110 and the coupler line 120 is assumed to be equal to 0.5 mm. These specific assumptions and geometrical data are provided for illustration of the present example but are not meant to limit the present invention and in particular the quarter-wavelength directional coupler described here in accordance with an exemplifying embodiment of the present invention.
Figure 14 shows the result of the simulation wherein optimization has been obtained for the following values of the components of the circuitry of the directional coupler: R_1L=R_1R=40 Ω, C_1L=C_1R=0.05 pF, R_sh=500 Ω and C_sh=0.13 pF. In Figure 14, the directivity is represented by a continuous line and the coupling is represented by a dashed line. As can be seen, the directional coupler 1300 of the present embodiment provides high directivity, at least higher than 30 dB in a frequency bandwidth centred around 2.45 GHz.
In the present example and also in the example described with reference to Figure 6, the signal representative of the energy level for a first electromagnetic wave travelling from the input port 111 to the output port 112 of the main transmission line 110 is read out at the first resistance 433 while the signal representative of the energy level for a second electromagnetic wave travelling from the output port 112 to the input port 111 of the main transmission line 110 is read out at the second resistance 434.

Example 3: Small-scale microstrip short-loop directional coupler

With reference to Figures 15 and 16, a small-scale (sub-millimetre range) microstrip short-loop directional coupler 1500 according to another exemplifying embodiment of the present invention is described.
The microstrip directional coupler 1500 shown in Figure 15 is identical to the directional coupler 1300 described with reference to Figure 13 except that the outline of the coupler line 120 is slightly different in size. The coupler line 120 of the directional coupler 1500 also has an outline of the type shown in Figure 2c, i.e. an Ω form, such that the first port 121 and the second port 122 (i.e. the ends, not denoted in Figure 15) of the coupler line 120 are close enough to connect lumped components between them. However, as can be seen in Figure 15, another design of the first portion 127 of the coupler line 120 and the second portion 128 of the coupler line 120 is represented wherein the respective thickness of the first portion 127 and the second portion 128 varies stepwise (in the present example two steps) towards the first port 121 and the second port 122, respectively. The present example therefore illustrates that the present invention is not limited to one design of the coupler line 120 and that many variations are possible within the scope of the present invention. The present example is also meant to illustrate that the present invention may be applied for a small-size (in the sub-millimetre range) directional coupler.
For the purpose of simulation, the following assumptions and geometrical data have been considered: a relative permittivity of the substrate equal to 10, a substrate height of 154 µm, a width of the main transmission line 110 of 150 µm, a length of the section 129 of the coupler line 120 of 2.4 mm, a spacing between the main transmission line 110 and the coupler line 120 of 50 µm and a width of the coupler line 120 of 350 µm. The present example is selected to simulate a directional coupler of relatively small size.
The results of the simulation obtained for the directional coupler 1500 with the above mentioned geometrical data are shown in Figure 16 wherein the continuous line represents the directivity and the dashed line represents the coupling. The simulation has been performed for the following values of the electronic components: R_1L=R_1R=30 Ω, C_1L=C_1R=0.32 pF, R_sh=100 Ω and C_sh=0.08 pF. Proximity of the coupler arms 127 and 128 introduces an additional distributed shunt capacitance in parallel to C_sh. Thus, the quoted value C_sh=0.08 pF is only a part of the physical value of the shunt capacitance connected between arms 127 and 128. It should be appreciated that placing the coupler arms even closer together may cause that the distributed capacitance replaces completely the lumped capacitance and the need for placing a lumped capacitor vanishes. As can be seen, a high directivity, higher than 35 dB from 0 to 3.8 GHz, has been achieved. Further, the coupling appears to be proportional to the frequency and is equal to about 25 dB at 2 GHz. The bandwidth of coupling changing by 1 dB is 12%. However, in many applications, coupling changes versus frequency can be allowed since they can be compensated in the next stage devices connected to the circuit. The present embodiment therefore provides an extremely wide-band directional coupler, which is advantageous for a number of applications including multi-band transmitters and receivers.

Example 4: Stripline short-loop directional coupler

With reference to Figures 17-20, a stripline short-loop directional coupler according to an exemplifying embodiment of the present invention is described.
Figure 17 shows a three-dimensional view and a side-view (or cross-section) of a stripline short-loop directional coupler 1700. The directional coupler comprises a substrate, denoted as 1720 in Figure 17, in which a main transmission line 110 is embedded. The printed circuit board 1750, on which the circuitry of the directional coupler 1700 is arranged, is placed at one side of the substrate 1720 such as represented in Figure 17. The coupler line 120 is a loop inserted in (or extending in) the substrate 1720 as shown in Figures 17 and 18 for electromagnetic coupling with the transmission line 110. As further shown in Figure 18, the coupler line (or loop) 120 of the directional coupler 1700 is connected via its extremities, i.e. the first port 121 and the second port 122, to the board 1750 at contacting points denoted 121' and 122'.
The circuitry of the directional coupler 1700 of Figure 17 is identical to the circuitry of the directional coupler 1100 described with reference to Figure 11. The resistance 439 at the left-hand side of the circuitry is connected between the resistance 437 (R_2L) and the reference potential and the resistance 440 at the right-hand side of the circuitry or board 1750 is connected between the resistance 438 (R_2R) and the reference potential 428. In this configuration, two sets of external probes may be connected to these two resistances 439 and 440, respectively, for reading the signal representative of the energy level of the electromagnetic wave travelling in the main transmission line 110 from left to right and from right to left, respectively, with reference to the representation shown in Figure 18. Optionally, the circuitry may be supplemented by two additional reading resistances (typically in the order of 50 Ω) for connecting internal detectors used to read the output signals representative of the energy levels of the electromagnetic waves travelling in the main transmission line 110.
In the present example, a symmetrical stripline is considered and a rather large size is assumed (in the millimetre range), which may for instance be used in high power transmitter circuits for base station or radar applications.
For the purpose of electromagnetic simulation, the following assumptions and geometrical data are taken into account and selected. The total height of the main transmission line 110 is assumed to be 16 mm and the width for a 50 Ω strip is assumed to be 23 mm. The loop 120 may for instance be made of a wire having a diameter of 0.8 mm, a length of 3 mm and a height of 2 mm. The loop 120 is inserted into the stripline space through openings made in the upper metal wall of the substrate 1720. The openings may for example be square with a side size in the order of 2.5 mm but the opening may also be circular. The PCB may be placed directly on the upper wall with the coupler ground plane soldered to that wall.
Figures 19 and 20 show the results of simulations for such a stripline directional coupler.
Figure 19 shows the results of simulations for two cases. The thick continuous line represents the directivity and the dashed line represents the coupling for a first case wherein R_1L=R_1R=66 Ω, C_1L =C_1R=0.5 pF, R_sh=160 Ω, C_sh=0.38 pF, R_2L =R_2L =650 Ω and R_3L=R_3R=50 Ω. The thin continuous line represents the directivity and the dotted line represents the coupling for a second case wherein R_1L=R_1R=66 Ω, C_1L=C_1R=0.115 pF, R_sh=300 Ω, C_sh=0.11 pF, R_2L=R_2L=650 Ω and R_3L=R_3R=50 Ω. The first case corresponds to optimization of the values of the electronic components for optimization at lower frequencies, and in particular in a band from 0.3 GHz to 1.5 GHz, while the second case corresponds to values of the electronic components obtained for optimization for the ISM band centred at about 2.45GHz.
Based on another series of simulations (not reported in the present patent application), it has also been concluded that it is possible to obtain high directivity in the ISM band for a variety of values of R_sh. However, different values of R_sh result in a different bandwidth of the directional coupler. It appears that high directivity and large bandwidth have been obtained for a value of the shunt resistance R_sh in the order of 300Ω leading to a directivity higher than 30 dB in a band of almost 500 MHz.
Figure 20 shows variations of the amplitude ratio ((|U_E|/|U_M|)-continuous lines) and the phase difference ((Arg(U_E)-Arg(U_M))-dashed and dotted lines) versus frequency for the two above mentioned cases, wherein the thick continuous line and the dashed line correspond to the first case and the thin continuous line and the dotted line correspond to the second case.

Example 5: Waveguide short-loop directional coupler

In the following, the case of a waveguide short-loop directional coupler is described.
The short-loop waveguide directional coupler considered in the present example may be constructed in the same way as the stripline directional coupler 1700 described with reference to Figure 17. Thus, in the present example, the configuration of the waveguide directional coupler is considered to be identical to the configuration of the stripline directional coupler 1700 described with reference to Figure 17. In the present example, the main transmission line 110 is a waveguide and the coupler line 120 is a loop inserted in the waveguide.
Referring in particular to Figure 18, it can be seen that the coupler line 120 may be mounted as part of a wall of the main transmission line. In particular, in a waveguide with a dominant mode of propagation TE01, the coupler loop 120 is preferably inserted into the waveguide 110 in the middle of the wider waveguide wall. The loop is preferably oriented along the wave propagation. The waveguide 110 operates above its cut-off frequency and the wave impedance (the ratio of the transverse electric and magnetic field) is frequency dependent and, in a TE01 mode, the wave impedance is always bigger than in free space.
The electronic components of the electronic means 130 are selected such that, in the frequency band of interest, the signals due to the magnetic and electric field components are equal in amplitude despite the frequency-dependent wave impedance in the waveguide.
Figure 21 shows both the simulation results and measurement values of the directivity for a waveguide directional coupler according to an exemplifying embodiment of the present invention wherein the waveguide directional coupler has a structure and design identical to the structure and design of the directional coupler 1700 described with reference to Figure 18. The simulation results are shown as a continuous (smooth) line while the measurements are shown as a continuous noisy line. It is to be noted that, in the middle of the band of interest at 2.45 GHz, the measurement setup has limited the measured directivity to about 35 dB, thereby explaining the discrepancy between the simulations results and the measurements in this frequency range while, otherwise, the agreement between the measurements and the simulation results is good.
Turning now to Figures 22 and 23, examples for realization of the capacitors used in the present invention will be described.
Figures 22a and 22b are schematic views showing possible ways of realization of the capacitors 435 and 436 (C_1R and C_1L). Those capacitors may be realized as open stubs between a strip and a ground plane of the PCB such as shown in e.g. Figure 22a or as open stubs connected by capacitors or trimmer capacitors such as shown in Figure 22b. Figures 22a and 22b also show contact areas 121' and 122' at which the extremities of the loop 120 may be connected.
Figures 23a and 23b are schematic views showing possible ways of realization of the shunt capacitor 131 (C_sh). The shunt capacitor 131 (C_sh) to be connected between two microstrip lines is usually of a relatively small value (usually between 0.05 pF and 1 pF). It may therefore be difficult to find lumped capacitors of that range of values. A capacitor with such a small capacitance may be realized by approaching the relevant microstrip lines of the printed circuit board close enough to create mutual capacitance between them. An alternative approach, however, consists in forming a shunt capacitor 1131 by connecting between the microstrip lines a piece of metalized PCB substrate 1150 with a transverse slot in the lower metallization layer 1149 thereby separating direct current flow between connected microstrip lines, such as shown in Figure 23a. Further, the value of such a shunt capacitor may be varied by connecting a trimmer capacitor 2133 (of the value of the order of 1 to 4 pF) in a slot in the upper metallization layer such as for the shunt capacitor 2133 shown in Figure 23b. In figures 23a and 23b, h represents the height or thickness of the substrate (distance between the two metal layers), w is the width of the capacitor and l is the length of the capacitor. The shunt capacitor 131 (C_sh) may also be formed with a "comb"-like shape.
The directional coupler may for instance be applied in applications wherein separate monitoring of the power of incident microwaves and the power of reflected microwaves travelling in the opposite direction than the travelling direction of the incident microwaves is required.
The present invention may in fact find applications in any technical fields wherein monitoring of microwave power is of interest such as for instance base stations for telecommunications, radars, microwave transceivers, and microwave heating devices.
In particular, the directional coupler of the present invention may be advantageous and especially useful in multi-band transmitters or receivers (for instance using the Omega-shaped directional coupler described with reference to Figures 15 and 16) as it provides very wide-band performance. Indeed, mobile terminals are nowadays intended to serve many frequency bands and, using prior art directional couplers, several couplers would be needed. However, in view of the relatively small size of mobile phones, it is preferable to use a single directional coupler, such as provided by the present invention.
The person skilled in the art realizes that the present invention by no means is limited to the preferred embodiments described above. On the contrary, many modifications and variations are possible within the scope of the appended claims. In particular, it will be appreciated that the above-mentioned examples and exemplifying embodiment are not limited to the above mentioned assumptions and geometrical data (e.g. the thickness of the substrate, its permittivity, the dimensions of the transmission lines, the impedances of the transmission lines) and that these assumptions and geometrical data may vary in function of the application and the manufacturing technology used for a specific application or a specific type of directional coupler. With the present invention, however, the various simulations show that adjustment of the values of the various electronic components of the circuitry of the directional coupler allow for compensation of those changes.
In addition, it will be appreciated that the various directional couplers described above for microstrip, stripline or waveguide applications need not be restricted to the specific values of the electronic components specified in connection to these examples. The specific values of the electronic components may be adjusted for a specific application.
In addition, it will be appreciated that the realization of the linear electronic means 130 may not be limited to a shunt capacitor but may also be realized by means of other passive electronic components or combination of such passive electronic components.
Further, as mentioned above, the use of the shunt capacitor 131 in the electronic means 130 provides the ability to design the directional coupler for a chosen bandwidth at high frequencies. The connection of such a capacitor between the two ports of the coupler line (or arms of the coupler line) is original and provides improvement of the characteristics of the directional coupler not providing by prior art directional couplers. Thus, the present invention may alternatively be defined as a directional coupler comprising a main transmission line, a coupler line and a capacitor. The main transmission line has an input port and an output port for transmitting an electromagnetic wave from the input port to the output port. The coupler line is arranged proximate to the main transmission line for electromagnetically coupling the coupler line with the main transmission line. The coupler line comprises a first arm including a first port at which a signal output representative of a first electromagnetic wave travelling from the input port to the output port is provided and a second arm including a second port at which a signal output representative of a second electromagnetic wave travelling from the output port to the input port is provided. The capacitor is connected between the first arm and the second arm and configured such that the signal output induced by the second electromagnetic wave at the first port is reduced and the signal output induced by the first electromagnetic wave at the second port is reduced. It will be appreciated that a directional coupler such as defined above may be combined with any one of the previously described embodiments.

Claims

A directional coupler (100) comprising:
a main transmission line (110) having an input port (111) and an output port (112) for transmitting a first electromagnetic wave from said input port to said output port and a second electromagnetic wave from said output port to said input port, and

a coupler line (120) arranged proximate to said main transmission line for electromagnetically coupling said coupler line with said main transmission line, said coupler line comprising a first port (121) at a first one of its ends and a second port (122) at its opposite end, and

a linear electronic means (130) connected between said first and second ports and configured to reduce

the output signal induced by the first electromagnetic wave at the second port by reducing the difference between the oppositely directed signals induced at the second port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the first electromagnetic wave and,

the output signal induced by the second electromagnetic wave at the first port by reducing the difference between the oppositely directed signals induced at the first port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the second electromagnetic wave.
A directional coupler according to claim 1, wherein the coupler line is arranged such that the distance between said first and second ports is smaller than the wavelength of the first electromagnetic wave.
A directional coupler according to claim 1 or 2, wherein said linear electronic means is configured to reduce
at the first port, the difference between the phases and the difference between the amplitudes, in absolute values, of the oppositely directed signals induced by the magnetic and electric components of the coupled field originating from the second electromagnetic wave, and
at the second port, the difference between the phases and the difference between the amplitudes, in absolute values, of the oppositely directed signals induced by the magnetic and electric components of the coupled field originating from the first electromagnetic wave.
A directional coupler according to any one of the preceding claims, wherein said linear electronic means (130) comprises a shunt capacitor (131).
A directional coupler according to claim 4, wherein said linear electronic means further comprises a shunt resistance (132) connected in parallel to the shunt capacitor.
A directional coupler according to any one of the preceding claims, further comprising:
a first resistance (433) connected between the first port and a reference potential (428),

a second resistance (434) connected between the second port and the reference potential,

a first capacitor (435) connected between a first portion of the coupler line, said first portion comprising the first port, and the reference potential, and

a second capacitor (436) connected between a second portion of the coupler line, said second portion comprising the second port, and the reference potential.
A directional coupler according to claim 6, wherein the first resistance is configured to provide a voltage output representative of the power of the first electromagnetic wave and the second resistance is configured to provide a voltage output representative of the power of the second electromagnetic wave.
A directional coupler according to claim 6 or 7, further comprising a first attenuator (437, 439) connected to the first resistance and a second attenuator (438, 440) connected to the second resistance for impedance matching with a detecting device measuring the respective voltage outputs at the first and second ports.
A directional coupler according to any one of the preceding claims, wherein the coupler line comprises a section (129) facing the main transmission line for electromagnetic coupling and two portions departing from each ends of said section, wherein a first portion (127) comprises the first port and a second portion (128) comprises the second port.
A directional coupler according to claim 9, wherein the length of the section (129) is smaller than a quarter wavelength or in the order of a quarter-wavelength and the distance between the first port and the second port is smaller than a quarter-wavelength of the first electromagnetic wave.
A directional coupler according to any one of the preceding claims, wherein the main transmission line and the coupler line include at least one of a group comprising microstrip transmission lines, stripline transmission lines and waveguide transmission lines.
A directional coupler according to any one of the preceding claims, wherein the second electromagnetic wave corresponds to a part of the first electromagnetic wave reflected back into the main transmission line via the output port of the main transmission line.
A directional coupler according to any one of the preceding claims, wherein the main transmission line is a waveguide transmission line and the electronic means is mounted on a printed circuit board (1750) adapted to constitute part of a wall of the waveguide transmission line.
A directional coupler according to any one of claims 4-13, wherein said first, second or shunt capacitors include at least one of a group comprising discrete capacitors, distributed capacitors, open circuit stubs and trimmer capacitors.
A radio transmitter, a radio receiver, a transceiver or a microwave heating device comprising a directional coupler defined in accordance to any one of the preceding claims.
Method for directional coupling of an electromagnetic wave comprising:
providing a main transmission line (110) having an input port (111) and an output port (112) for transmitting a first electromagnetic wave from said input port to said output port and a second electromagnetic wave from said output port to said input port,

providing a coupler line (120) comprising a first port (121) at a first one of its ends and a second port (122) at its opposite end, said coupler line being electromagnetically coupled with the main transmission line, and

providing a linear electronic means (130) connected between said first and second ports for reducing

the output signal induced by the first electromagnetic wave at the second port by reducing the difference between the oppositely directed signals induced at the second port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the first electromagnetic wave and,

the output signal induced by the second electromagnetic wave at the first port by reducing the difference between the oppositely directed signals induced at the first port by the magnetic and the electric components, respectively, of an electromagnetic field coupled to the coupler line and originating from the second electromagnetic wave.