EP2258021A1 - Résonateurs électromagnétiques monolithiques microminiatures - Google Patents

Résonateurs électromagnétiques monolithiques microminiatures

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Publication number
EP2258021A1
EP2258021A1 EP09806998A EP09806998A EP2258021A1 EP 2258021 A1 EP2258021 A1 EP 2258021A1 EP 09806998 A EP09806998 A EP 09806998A EP 09806998 A EP09806998 A EP 09806998A EP 2258021 A1 EP2258021 A1 EP 2258021A1
Authority
EP
European Patent Office
Prior art keywords
filter
resonator
resonator structures
substrate
structures
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP09806998A
Other languages
German (de)
English (en)
Inventor
Eric M. Prophet
Balam A. Willemsen
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Superconductor Technologies Inc
Original Assignee
Superconductor Technologies Inc
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Superconductor Technologies Inc filed Critical Superconductor Technologies Inc
Publication of EP2258021A1 publication Critical patent/EP2258021A1/fr
Withdrawn legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/203—Strip line filters
    • H01P1/20327—Electromagnetic interstage coupling
    • H01P1/20354—Non-comb or non-interdigital filters
    • H01P1/20381—Special shape resonators
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P7/00—Resonators of the waveguide type
    • H01P7/08—Strip line resonators
    • H01P7/082—Microstripline resonators

Definitions

  • the present inventions generally relate to microwave filters, and more particularly, to microwave filters designed for narrow-band applications.
  • Electrical filters have long been used in the processing of electrical signals.
  • such electrical filters are used to select desired electrical signal frequencies from an input signal by passing the desired signal frequencies, while blocking or attenuating other undesirable electrical signal frequencies.
  • Filters may be classified in some general categories that include low-pass filters, high-pass filters, band-pass filters, and band-stop filters, indicative of the type of frequencies that are selectively passed by the filter.
  • filters can be classified by type, such as Butterworth, Chebyshev, Inverse Chebyshev, and Elliptic, indicative of the type of bandshape frequency response (frequency cutoff characteristics) the filter provides relative to the ideal frequency response.
  • band-pass filters are conventionally used in cellular base stations and other telecommunications equipment to filter out or block RF signals in all but one or more predefined bands.
  • such filters are typically used in a receiver front-end to filter out noise and other unwanted signals that would harm components of the receiver in the base station or telecommunications equipment.
  • Placing a sharply defined band-pass filter directly at the receiver antenna input will often eliminate various adverse effects resulting from strong interfering signals at frequencies near the desired signal frequency. Because of the location of the filter at the receiver antenna input, the insertion loss must be very low so as to not degrade the sensitivity of the receiver as measured by its noise figure. In most filter technologies, achieving a low insertion loss requires a corresponding compromise in filter steepness or selectivity.
  • Microwave filters are generally built using two circuit building blocks: a plurality of resonators, which store energy very efficiently at one frequency, f 0 ; and couplings, which couple electromagnetic energy between the resonators to form multiple stages or poles.
  • a four-pole filter may include four resonators and five couplings between the signal input, resonators and signal output.
  • the strength of a given coupling is determined by its reactance (i.e., inductance and/or capacitance).
  • the relative strengths of the couplings determine the filter shape, and the topology of the couplings determines whether the filter performs a band-pass or a band-stop function.
  • the resonant frequency f 0 is largely determined by the inductance and capacitance of the respective resonator.
  • the frequency at which the filter is active is determined by the resonant frequencies of the resonators that make up the filter.
  • Each resonator must have very low internal resistance to enable the response of the filter to be sharp and highly selective for the reasons discussed above. This requirement for low resistance tends to drive the size and cost of the resonators for a given technology.
  • Microwave filters typically have multiple resonant frequencies, which allows microwave filters to be operated in different modes.
  • resonant frequencies include the fundamental frequency fo and multiples of the fundamental frequency fo (e.g., 2fo, 3fo, etc.) or multiples of a factor of the fundamental frequency f 0 (e.g., 2f o /n 3f o /n etc.).
  • filters have been fabricated using normal; that is, non- superconducting conductors. These conductors have inherent lossiness, and as a result, the circuits formed from them have varying degrees of loss. For resonant circuits, the loss is particularly critical.
  • the quality factor (Q) of a device is a measure of its power dissipation or lossiness. For example, a resonator with a higher Q has less loss.
  • Resonant circuits fabricated from normal metals in a microstrip or stripline configuration typically have Q's at best on the order of four hundred.
  • Q's at best on the order of four hundred.
  • HTS high temperature superconductor
  • the factors that drive the size of these kinds of filters are varied.
  • the filter size will generally increase if: the center frequency of the filter is decreased, the insertion loss target is decreased, the number of resonators required is increased, the power handling requirements (compression, intermodulation) requirements are increased, or if the stray coupling between non-nearest neighboring resonators is too large to be ignored. Any of these may lead a filter to be unrealizable due to the constraints imposed by finite, small substrate size. In order to preserve the high-quality performance of a filter, it is desirable to minimize as much as possible the peak current densities within the structure of the filter. As discussed in U.S. Patent No.
  • the peak current densities within a filter structure could be reduced by increasing the width of the microstrip lines and gaps between the lines relative to the thickness of the substrate. That is, wider microstrip lines could be used in the regions of the filter structure where high current is anticipated in order to minimize the current density within these regions, thereby increasing the power handling capability of the resulting filter.
  • the relatively high current flowing through the microstrips creates a relatively large electromagnetic field that interferes with surrounding structures.
  • box-like structures may be placed around the respective resonators in order to prevent the electrical fields generated at each of the resonators from interfering from each other. These box-like structures, however, add to the size and cost of the filter.
  • IMD intermodulation distortion
  • Intermodulation products are generated at various orders, with the order of a distortion product given by the sum of m + n.
  • a monolithic filter comprise a substrate (e.g., one composed of a dielectric material), and one or more resonator structures (which may be planar in nature) formed on a planar side of the substrate.
  • the filter takes the form of a microstrip filter, and thus, includes a continuous ground plane disposed on the other planar side of the substrate.
  • Each of the resonator structure(s) has a resonant frequency, e.g., in the microwave range (e.g., in the range of 800-2,200 MHz).
  • Each resonator structure comprises a folded transmission line (e.g., a spiral-in, spiral-out configuration) that is patterned to form a plurality of adjacent line segments and a plurality of gaps disposed between the adjacent line segments.
  • the folded transmission line is composed of a high temperature superconductor (HTS) material.
  • the filter further comprises an input terminal coupled to one end of the one or more resonator structures, and an output terminal connected to another end of the one or more resonator structures. The input terminal and output terminal may be coupled to the resonator structure(s) such that the filter can be operated as a narrowband filter.
  • the ratio of a sum of an average width of the adjacent lines and an average width of the gaps to a thickness of the substrate is equal to or less than 0.50. In one embodiment, the ratio is equal to or less than 0.30. In another embodiment, the ratio is equal to or less than 0.20. In still another embodiment, the ratio is equal to or less than 0.10.
  • Each of the resonator structures may have any shape, e.g., rectangular or circular. In yet another embodiment, each resonator structure has a nominal linear electrical length of a full wavelength at the resonant frequency of the respective resonator structure. If the filter comprises multiple resonator structures, they may be coupled to each other in series.
  • each of the resonator structures may have a nominal linear electrical length of a full wavelength at the resonant frequency of the respective resonator structure, and the input terminal and output terminal may be coupled to the resonator structures such that the filter can be operated in a higher order mode.
  • Fig. 1 is a plan view of a prior art spiral-in, spiral-out resonator filter
  • Fig. 2 is a cross-sectional view of the prior art spiral-in, spiral-out resonator of Fig. 1, taken along the line 2-2
  • Fig. 3 is a plan view of a basic spiral-in, spiral-out resonator structure constructed in accordance with the present inventions
  • Fig. 4 is a cross-sectional view of the spiral-in, spiral-out resonator structure of Fig. 3, taken along the line 4-4;
  • Fig. 5 is a magnified view of the spiral-in, spiral-out resonator structure of Fig. 4, taken along the line 5-5;
  • Fig. 6 is a plan view of the resonator constructed in accordance with the present inventions, wherein the resonator uses two of the spiral-in, spiral-out resonator structures illustrated in Fig. 3;
  • Fig. 7 is a plan view of the two-wavelength resonator of Fig. 6 as compared to a prior art two-wavelength spiral-in, spiral out resonator;
  • Fig. 8 is a plan view of another resonator constructed in accordance with the present inventions, wherein the resonator uses two circular spiral-in, spiral-out resonator structures;
  • Fig. 9 is a plan view of a single-resonator filter constructed in accordance with the present inventions, wherein the filter uses eight of the resonators illustrated in Fig. 6 coupled to each other to form a single higher-order resonator;
  • Fig. 10 is a plot of the computed frequency response of the filter of Fig. 9;
  • Fig. 11 is a plot showing the normalized intermodulation distortion plotted against normalized input power for two types of resonators constructed in accordance with the present inventions, wherein one type has mitered/rounded corners and the second type has non-mitered/rounded corners;
  • Fig. 12 is a plan view of a multi-resonator filter constructed in accordance with the present inventions, wherein the filter uses four of the resonators illustrated in Fig. 6;
  • Fig. 13 is a plot of the computed frequency response of the filter of Fig. 12;
  • Fig. 14 is a plan view of a multi-resonator filter constructed in accordance with the present inventions, wherein filter uses two of the resonators illustrated in Fig. 6;
  • Fig. 15 is a plot of the computed frequency response of the filter of Fig. 14;
  • Fig. 16 is a plan view of a multi-resonator filter constructed in accordance with the present inventions, wherein the filter uses ten of the resonators illustrated in Fig. 6;
  • Fig. 17 is a plan view of a multi-resonator filter that uses eight two-wavelength resonators constructed in accordance with the present inventions; and Fig. 18 is a plot of the computed frequency response of the filter of Fig. 17.
  • full-wavelength ( ⁇ ) spiral-in, spiral-out resonators are used due to their ability to reduce the peak current near the edges of the resonator lines.
  • the filters are used as band-pass filter having a pass band within a desired frequency range, e.g., 800-900 MHz or 1 ,800-2,220 MHz.
  • the RF filters are placed within the front-end of a receiver (not shown) behind a wide pass band filter that rejects the energy outside of the desired frequency range.
  • a conventional filter 10 will first be described.
  • the conventional filter structure 10 comprises a substrate 12 and a spiral-in, spiral-out (SISO) resonator structure 14 patterned on one planar side (top side) of the substrate 12.
  • the resonator structure 14 may be monolithically formed onto the substrate 12 using conventional techniques, such as photolithography.
  • the resonator structured may be composed of an HTS material, such as an epitaxial thin film Thallium Barium Calcium Cuprate (TBCCO) or Yttrium Barium Cuprate (YBCO).
  • the resonator structure 14 may be composed of superconductors such as Magnesium Diboride (MgB2), Niobium, or other superconductor whose transition temperature is less than 77K as these allow the designer to make use of substrates that are incompatible with HTS materials.
  • the resonator structure 14 may be composed of a normal metal, such as aluminum, silver or copper even though the increased resistive loss in these materials may limit the applicability of the invention.
  • the substrate 12 may be composed of a dielectric material, such as LaAl ⁇ 3, Magnesium Oxide (MgO), sapphire, or polyimide.
  • the conventional filter 10 has a microstrip architecture, and thus, further comprises a continuous ground plane 16 disposed on the other planar side (bottom side) of the substrate 12 opposite to the resonator structure 14.
  • the conventional filter 10 has a stripline architecture, in which case, the filter 10 may instead comprise another dielectric substrate (not shown), with the resonator structure 14 being sandwiched between the respective dielectric substrates.
  • the filter 10 further comprises an input terminal (pad) 18 and an output terminal (pad) 20 coupled to the resonator structure 14 in a manner that configures the filter 10 to have narrowband characteristics.
  • the resonator structure 14 includes a folded transmission line 22 that is patterned to form a SISO structure.
  • a SISO structure is a conductor that is folded over onto itself to form two parallel lines 24 that are connected to each other by a single 180° bend 26. The two lines 24 are then spiraled around the bend 26 together in the same direction, with the end of one line 24 exiting the structure in one direction to couple to the input terminal 18, and the end of the other line 24 exiting the structure in the opposite direction to couple to the output terminal 20.
  • one end of the transmission line 22 has a plurality of turns of lefthandedness, which when combined, turn through at least 360° and the other end of the transmission line 22 has a plurality of turns of righthandedness, which when combined, turn through at least 360°.
  • At least one turn of lefthandedness is disposed between at least two turns of righthandedness, and at least one turn of righthandedness is disposed between at least two turns of lefthandedness.
  • the transmission line 22 forms a plurality of line segments 32 and a plurality of gaps or spaces 34 between the line segments 32.
  • the transmission line 22 generates an electromagnetic field that has a field of influence 36 that tends to be of the same order as the widths of the line segments 32 and the gaps 34 between the line segments 32.
  • the currents in adjacent line segments 32 are unidirectional, which tends to reduce the peak magnitude of the current near the edges of the transmission line 22 within the resonator structure 14.
  • the ratio of the sum of the average width of the line segments 32 (in this case, 0.250 mm) and the average width of the gaps 34 (in this case, 0.250 mm) to the thickness 38 of the substrate 12 (in this case, 0.500 mm) is relatively great (in this case, 1), which generates an electromagnetic field that extends far beyond the resonator structure 14 itself, thereby resulting in a relatively large field of influence 36 between the resonator structure 14 and the ground plane 16 disposed on the substrate 12 below, and any metallic elements, including electrically grounded lids, above the substrate 12.
  • a filter 50 constructed in accordance with an embodiment of the present inventions will now be described.
  • the filter 50 comprises a substrate 52, a spiral-in, spiral-out (SISO) resonator structure 54 patterned on one planar side (top side) of the substrate 52, a continuous ground plane 56 disposed on the other planar side (bottom side) of the substrate 52 opposite the resonator structure 54, and an input terminal (pad) 58 and an output terminal (pad) 60 coupled to the resonator structure 54 in a manner that configures the filter 50 to have narrowband characteristics.
  • the resonator structure 54 includes a folded transmission line 62 that is patterned to form a SISO structure, and forms a plurality of line segments 72 and intervening gaps 74 between the line segments 72.
  • the ratio of the sum of the average width of the line segments 72 (in this case, 0.050 mm) and the average width of the gaps 74 (in this case, 0.025 mm) between the line segments 72 to the thickness of the substrate 52 (in this case, 0.500 mm) is relatively small.
  • this ratio is 0.15 in the illustrated embodiment, the ratio may be equal to or less than 0.50, preferably equal to or less than 0.30, and more preferably equal or less than 0.20.
  • widths of the line segments 72 and intervening gaps 74 are uniform, it should be noted that the widths of the line segments 72, as well as the widths of the gaps 74, may be non-uniform, as long as the ratio of widths and gaps to substrate thickness remains relatively small.
  • Direct capacitive coupling to the resonator structure 54 via the input terminal 58 and output terminal 60 can be achieved at the high-voltage ends of the resonator structure 54.
  • the lengths of the high-voltage ends of the resonator structure 54 may be adjusted according to the external loading, such that the current nodes occur at the geometric center of the resonator structure 54, giving rise to edge-current reduction at the edges of the transmission line 62 as well as the edges of the resonator structure 54, as described in U.S. Patent No. 6,026,311.
  • the current density of the resonator structure 54 was computed using the full- wave planar program Sonnet with cell sizes equal to the width of the line segments and gaps therebetween.
  • Sonnet uses red for the most intense current densities, while, as the current weakens, the colors vary with the rainbow down to blue for the weakest current densities.
  • the corresponding current densities will range from a fairly dark gray for the most intense current densities down to a very light gray or white for the mid-range current densities, on to nearly black for the very low current densities.
  • a relatively low current density region 80 is located in the center of the resonator structure 54 and at the ends of the transmission line 62, reflecting the three current nodes for a full-wave length structure (i.e., for a full- wavelength transmission line, the first zero-current node will be at the beginning of the transmission line, the second zero-current node will be in the middle of the transmission line, and the third zero-current node will be at the end of the transmission line), and a relatively high current density region 82 is located at the periphery of the resonator structure 54, reflecting the two current peaks for a full- wave length structure (i.e., for a full-wavelength transmission line, the first current peak will be at the half-way point of the spiral-in portion of the transmission line, and the second current peak will be at the half-way point of the spiral-out portion of the transmission line).
  • the single resonator structures constructed in accordance with the present inventions can be used as building blocks for the design of higher order resonators that are much smaller than and/or may be operated at significantly lower frequencies than, similar conventional monolithic resonators.
  • These resonators can be used in filters that are designed to operate the resonator at higher resonant modes. These higher order modes do not readily excite neighboring modes, resulting in a very clean broadband response with no-entrant moding and no signs of spurious modes out to three times the preferred resonance.
  • Such resonators may be operated in any full wave mode operation, (n ⁇ , where n is any integer)(e.g., second ( ⁇ ), fourth (2 ⁇ ), sixth (3 ⁇ ), etc.).
  • These higher order resonators also have higher power handling capabilities.
  • the resonators may be designed around the desired higher order mode. That is, the resonator may be tuned at the selected higher order mode, such that very little energy will couple into the other modes.
  • two of the basic resonator structures 54 are connected together in series to form a two-wavelength (2 ⁇ ) resonator 100.
  • 2 ⁇ two-wavelength
  • Direct capacitive coupling to the resonator structures 54 may be achieved at or near the central current node, such that other modes of the resonator structures 54 are not easily excited, since the local voltage at the central capacitive coupling node is nearly zero when the resonator would be resonating in any of its n ⁇ /2 modes.
  • the modeling suggests that the nearby (n ⁇ 1] ⁇ modes could be excited though the filter is often mistuned at those frequencies and the energy coupled in may in reality be quite small.
  • the size of foot print of the two wavelength (2 ⁇ ) resonator 100 is considerably smaller.
  • the relatively low current density region 80 is located in the center of each resonator structure 54, at the ends of the transmission line 62, and at the center of the transmission line 62 between the resonator structures 54, reflecting the five current nodes for a two-wave length structure, and two relatively high current density regions 82 are located at the peripheries of the resonator structures 54, reflecting the four current peaks for a two-wave length structure.
  • the reduced size of the higher-order resonator 100 results in lower costs due to the reduced substrate area and smaller microwave packaging, and gives rise to the possibility that normal metal (non-HTS) filters might be made small enough for use in cellular handset type applications.
  • the smaller size of the filter can also dramatically reduce the overall cryogenic head load, enabling the use of smaller, less power-hungry cryogenic coolers.
  • the enhanced length of the resonator helps to reduce some of the nonlinear effects in materials, such as HTS, by introducing multiple peaks along the length of the transmission line to reduce the peak current in the resonator. These higher order modes also radiate much less than do the lower modes, thereby allowing further reduction in the size of the filter.
  • each of the circular resonator structures 154 includes a folded transmission line 162 that is patterned to form a SISO structure, and forms a plurality of line segments 172 and intervening gaps 174 between the line segments 172.
  • the ratio of the sum of the average width of the line segments 172 and the average width of the gaps 174 between the line segments 172 to the thickness of the substrate is relatively small.
  • the current density of the resonator structures 154 was computed using the full-wave planar program Sonnet with cell sizes equal to the width of the line segments and gaps therebetween.
  • a relatively low current density region 180 is located in the center of each resonator structure 154, at the ends of the transmission line 162, and at the center of the transmission line 162 between the resonator structures 154, reflecting the five current nodes for a two-wave length structure, and two relatively high current density regions 182 are located at the peripheries of the resonator structures 54, reflecting the four current peaks for a two-wave length structure.
  • a single resonator 190 includes eight basic resonator structures 192 that are connected together between an input port 194 and an output port 196 in series to provide an improvement in power handling of 9dB (i.e., three successive doublings of resonators (2 3 )).
  • Each of the resonator structures 192 is similar to the previously described resonator structure 54 in that the widths of the line segments and intervening gaps are relatively small relative to the thickness of the substrate.
  • the resulting resonator 190 has an effective wavelength of 8 ⁇ that can be operated in many of the n ⁇ modes.
  • the computed frequency response (Sn and S 2 i) of the resonator 190 is shown in Fig. 10. Further details discussing techniques in increasing the power handling of filters using multiple basic resonator structures are disclosed in U.S. Patent Publication No. 2008-0278262 A1.
  • the corners of the resonators described herein can be shaped in order to effect a desired IMD slope.
  • the IMD for a resonator with rounded/mitered or chamfered corners and the IMD for a resonator with squared corners were measured against a normalized input power.
  • the resonator with the rounded corners exhibited an IMD slope of 3
  • the resonator with squared corners exhibited an IMD slope of 4.
  • the corners of the resonators may be advantageously shaped depending upon whether the filter is to be operated at relatively low power levels or relatively high power levels.
  • a band-pass filter 200 comprises four two-wavelength (2 ⁇ ) resonators 100, an input terminal 208 coupled to the first resonator 100(1) via a capacitive coupling 212, and an output terminal 210 coupled to the fourth resonator 100(4) via a capacitive coupling 214.
  • the second resonator 100(2) and third resonator 100(3) are coupled to together at their tops and bottoms via conductors 216 as a means to enhance the native coupling between the second resonator 100(2) and third resonator 100(3).
  • the current density of the resonators 100 was computed using the full-wave planar program Sonnet with cell sizes equal to the width of the line segments and gaps therebetween. As shown in Fig. 12, relatively low current density regions 220 are located in the centers of each resonator 100 and at the ends of the transmission line, and relatively high current density regions 222 are located at the periphery of the second and third resonators 100(2), 100(3).
  • the computed frequency response (Sn and S 2 i) of the resonator filter 200 is shown in Fig. 13.
  • a band-pass filter 250 comprises two sixteen-wavelength (16 ⁇ ) resonators 190, an input terminal 252 coupled to the first resonator 190(1 ), and an output terminal 254 coupled to the second resonator 190(2).
  • the computed frequency response (S 11 and S 21 ) of the resonator filter 250 is shown in Fig. 15.
  • a band-pass filter 300 comprises ten sixteen-wavelength (16 ⁇ ) resonators 190, an input terminal 302 coupled to the first resonator 190(1), and an output terminal 304 coupled to the tenth resonator 190(10).
  • the second resonator 100(2) and fifth resonator 100(5) are coupled to together at their tops and bottoms via cross-couplings 306, and the sixth resonator 100(6) and ninth resonator 100(9) are coupled to together at their tops and bottoms via cross-couplings 306, thereby creating transmission zeroes in the near stop-band, going from a Chebyshev-like response to a quasi-elliptical response. This is done to increase the near-band selectivity of the filter (slope of the rejection) at the expense of rejection further away from the pass band.
  • a band-pass filter 350 comprises eight two-wavelength (2 ⁇ ) resonators 352, an input terminal 358 coupled to the first resonator 352(1) via a capacitive coupling 362, and an output terminal 360 coupled to the eight resonator 352(8) via a capacitive coupling 364.
  • Each of the resonators 352 is identical to the resonator 100 illustrated in Fig. 6, with the exception that the line width of the line segments is 0.01 mm, and the width of the gaps between the line segments is 0.005 mm. Thus, the ratio of sum of the average width of the line segments and the average width of the gaps to the thickness of the substrate is .03.
  • the computed frequency response (Sn and S 2 i) of the resonator filter 350 is shown in Fig. 18.

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  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)
  • Filters And Equalizers (AREA)

Abstract

Filtre, comprenant un substrat et une ou plusieurs structures de résonateur formées sur une face plane du substrat. La ou chacune des structures de résonateur possède une fréquence de résonance et comprend une ligne de transmission repliée dotée de motifs formant une pluralité de segments de ligne adjacents et une pluralité d’espaces séparant les segments de ligne adjacents. Le rapport entre une somme d’une largeur moyenne des segments de ligne adjacents et d’une largeur moyenne des espaces et une épaisseur du substrat est inférieur ou égal à 0,50. Le filtre comprend en outre une borne d’entrée reliée à une extrémité de la ou des structures de résonateur, et une borne de sortie reliée à une autre extrémité de la ou des structures de résonateur.
EP09806998A 2008-03-25 2009-03-25 Résonateurs électromagnétiques monolithiques microminiatures Withdrawn EP2258021A1 (fr)

Applications Claiming Priority (2)

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US7063408P 2008-03-25 2008-03-25
PCT/US2009/038234 WO2010019285A1 (fr) 2008-03-25 2009-03-25 Résonateurs électromagnétiques monolithiques microminiatures

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TWI568203B (zh) * 2012-08-31 2017-01-21 Yong-Sheng Huang Harmonic Suppression Method of Radio Frequency Circuits
CN106329042A (zh) * 2016-10-25 2017-01-11 绍兴文理学院 一种级联式宽阻带超导带通滤波器及设计方法
CN111465947B (zh) * 2017-12-15 2023-12-05 谷歌有限责任公司 传输线谐振器耦合

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US4004257A (en) * 1975-07-09 1977-01-18 Vitek Electronics, Inc. Transmission line filter
US6347237B1 (en) * 1999-03-16 2002-02-12 Superconductor Technologies, Inc. High temperature superconductor tunable filter
JP3786031B2 (ja) * 2002-02-26 2006-06-14 株式会社村田製作所 高周波回路装置および送受信装置

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JP2011516005A (ja) 2011-05-19
KR20100135272A (ko) 2010-12-24
WO2010019285A1 (fr) 2010-02-18

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