EP1547192B1 - Device for transmitting or emitting high-frequency waves - Google Patents

Description

The present invention relates to a device for transmitting or radiating high-frequency waves according to the preamble of claim 1, as from BHATTACHARYYA ET AL. "ANALYSIS OF STRIPLINE-FED SLOT-COUPLED PATCH ANTENNA WITH VIAS FOR PARALLEL-PLATE MODE SUPPRESSION" IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, IEEE INC. NEW YORK, US, Vol. 46, No. 4, April 1, 1998 (1998-04-01), pages 538-545, XP000750734 ISSN: 0018-926X ,

A similar device is from the US 5,801,660 known.

From the JP 2000-174515A a coplanar waveguide is known, in which a strip line extends in a plane having a ground plane, wherein above this plane a guide plane is provided with an opening for coupling out high-frequency waves.

ITO M. ET AL: "MICROWAVE SYMPOSIUM DIGEST. Low cost multi-layer ceramic package for flip-chip MMIC up to W-band". 2000 IEEE MTT-S INTERNATIONAL BOSTON, MA, USA 11-16 JUNE 2000, PISCATAWAY, NJ, USA, IEEE, US, June 11, 2000 (2000-06-11), pages 57-60, XP010505923 ISBN: 0-7803 -5687-X disclose a substrate of a ceramic material, such as low-temperature co-fired ceramic (LTCC) for a microstrip line.

US 5,414,394 discloses a device having a microstrip line and an integrated impedance transformer provided at the end of the microstrip line.

The GB 2 007 919 A discloses a drainage structure for a microwave transmission device.

Devices for emitting electromagnetic waves, such as planar antenna elements, which are excited via a slot for oscillation and thus for the emission of high-frequency waves, are widely used, for example, in radio relay, satellite radio or radar technology. They are preferably used in the microwave range, since small sizes and therefore simple implementations using low costs are possible here.

A common planar antenna device is described with reference to FIG Fig. 6A in which a slot coupling via a microstrip line (MSL) 10 is excited. For this purpose, this microstrip line 10 has an abrupt end 10 'and thus forms an open-running line. At a distance d of about 1/4 of the line wavelength to this abrupt end 10 'of the microstrip line 10 is in a separated by a substrate 11 ground surface 12 perpendicular to the microstrip line 10, a slot 14, through which a penetration, ie, a coupling of the this point maximum magnetic field occurs. This also provided with an electric field component field stimulates planar emitter element 16, which is also called patch element, to a resonant oscillation and almost complete radiation of the high-frequency energy with the ground plane 12 orthogonal Hauptausbreitungsrichtung on. Fig. 6B shows the cross-sectional view of the device according to Fig. 6A in plan view.

A disadvantage of this arrangement is that microstrip line substrates 11 become very thin at higher frequencies, eg 254 μm in a short range radar application (SRR) at 24 GHz, and do not have sufficient structural strength for obstruction. Therefore, these substrates 11 must be provided with a rigid substrate 18, as in FIG Fig. 7A shown connected. This substrate 18 is not high frequency suitable for cost reasons. The support material 18 is mounted above the ground surface 12 with a fixed connection to the same, wherein to ensure the function of the antenna in the region of the coupling slot 14 and the radiator element 16 is a costly recess 19 in the substrate 18 is required, so on the coupling slot 14, the radiator element 16 can be coupled electromagnetically.

Another conventional embodiment of a slot-coupled antenna used to power the single radiator 16 is a so-called "buried", signal-carrying line 10 with abrupt cable end 10 ', which is designed in the form of a so-called triplate line and turn also via a slot 14 the single radiator 16 to Radiation stimulates. The signal line 10 is arranged substantially plane-parallel between two ground surfaces 12, 13, wherein in according to Fig. 8A respectively. Fig. 8B In this case, the microstrip line 10 is closer to one of the two ground planes 12, 13, which leads to an antenna arrangement with asymmetrical triplate supply. In contrast, there are also arrangements with symmetrical feeding, ie with equal spacing of the embedded signal line 10 to the outer ground planes 12, 13. The symmetrical or asymmetrical triplate arrangement has the advantage that larger line elements in a lower layer than Buried structures can be hidden, so that space can be saved. In particular, if larger antennas, which consist of a plurality of such individual emitters 16, are to be implemented in order to increase the directivity of the antenna, this shift of high-frequency line arrangements in lower layers can allow compact structures, since the feed network of an antenna array does not allow negligible share of the required space.

In addition, a buried feed network does not adversely affect the radiation characteristics of such an arrangement, in contrast to, especially at higher frequencies, "open" distribution or feed networks, which contribute significantly to parasitic radiation. Another advantage is the possibility to provide easily produced multi-layer or multilayer arrangements, since their individual layers or single layers have good high-frequency properties and carry the respective to be buried line structures. When using suitable layer or substrate materials, such as ceramics, can be dispensed with the connection with an additional mechanical support, since the multilayer arrangement on has sufficient structural stability. In particular, low temperature cofired ceramic (LTCC) substrates are suitable in this field.

The same with respect to 8A and 8B However, the antenna arrangement described has the disadvantage that at an abrupt end 10 'of the signal-carrying, central line 10 of the triplate structure, the separation of waves is greatly favored. A not inconsiderable power component of the signal can then undesirably propagate in the substrate material 11, for example in the form of parallel plate modes or waveguide modes. If the multilayer arrangement is enclosed laterally in a metallic carrier or housing, the excitation of waveguide modes is additionally promoted. The propagation of waveguide modes is determined by their cutoff frequency fg, the value of which depends directly on the distances of the bounding metallic walls.

In general, the relationship applies that the cut-off frequency f _{g of} a waveguide mode is shifted to lower frequencies when the distance between the electrically conductive, for example, metallic walls is increased. At the same time, the number of modes capable of propagation in a certain frequency band increases steadily. If such modes are excited in the substrate 11 by open-ended line ends, the power radiated via the radiator element 16 is reduced on the one hand, and on the other hand couplings with other circuit parts within the substrate 11, eg further antenna elements, are favored, which adversely affects the antenna characteristics and the overall system behavior impact.

ADVANTAGES OF THE INVENTION

The device according to the invention for transmitting or emitting high-frequency waves with the features of claim 1 has the advantage over the known approach that the excitation of substrate or waveguide modes in a slot-coupled antenna arrangement with symmetrical or asymmetrical triplate line prevents or on a for the Behavior of the antenna or the system is no longer relevant measure is reduced without negatively affecting the basic mode of action of a slit-coupled radiating device.

The device according to the invention makes it possible to provide a cost-effective improvement in the function of the antenna, since the suppression of the described excitation of substrate or waveguide modes contributes to improving the efficiency of the antenna and thus to improving the system behavior.

The idea underlying the present invention essentially consists in providing a shielding measure both in the area of the signal line and in the area of the coupling slot and adapting their dimensioning to both requirements.

In other words, an apparatus for transmitting high-frequency waves is provided, which has an end-provided microstrip line in a substrate for transmitting high-frequency useful signals, a first ground plane and a second ground plane, which are provided on opposite sides of the microstrip line, for shielding the microstrip line, providing an opening in the first ground plane at a predetermined distance from the end of the strip line for coupling out a high frequency signal, via means for electrically connecting the first ground plane to the second ground plane in the first ground plane Peripheral the microstrip line for shielding the same has (eg by so-called vias) and provides a planar coupling device for receiving and transmitting the high-frequency useful signal, wherein the via device is structured and / or dimensioned such that at a given frequency of the useful signal substantially no propagatable or resonant capable Waveguide modes occur in the substrate.

In the dependent claims are advantageous developments and improvements of the device specified in claim 1.

According to the invention, the structure of the plated-through device widens in the region of the coupling opening. This provides the advantage that the coupling to a radiating element (patch) is not hindered by the shielding through-connection device in the region of the coupling opening.

wobei C₀ für die Lichtgeschwindigkeit im Vakuum, ε_r für die dielektrische Permittivität des Substrats und f für die Frequenz eines Nutzsignals steht. Dadurch wird auf vorteilhafte Weise verhindert, dass ein erster ausbreitungsfähiger Hohlleitermode eines Rechteckhohlleiters, der hier näherungsweise vorliegt (TE₁₀-Mode), ein Mode mit transversal elektrischem (TE) Feld im Querschnitt gebildet wird.According to the invention, a distance a between opposing via devices in the region of the microstrip line is smaller than the quotient $c_{}$${}_0ö\left(2\cdot f\cdot \sqrt{{\epsilon }_r}\right).$

where C ₀ for the speed of light in vacuum, ε _r for the dielectric permittivity of the substrate and f stands for the frequency of a useful signal. This prevents advantageously such that a first propagation capable waveguide mode of the rectangular waveguide, which approximately is present here (TE ₁₀ mode), a mode with a transverse electrical (TE) field is formed in cross section.

wobei C₀ für die Lichtgeschwindigekeit im Vakuum, ε_r für die dielektrische Permittivität des Substrats und f_res für eine Resonanzfrequenz eines anregbaren Hohlleitermodes steht, welche oberhalb eines Nutzsignalfrequenzbandes vorzusehen ist. Dies ist ein Vorteil für die Dimensionierung der Durchkontaktierungs- bzw. Via-Wände im Bereich des Koppelschlitzes, da auf diese Weise vermieden wird, dass unerwünschte Resonanzfrequenzen Hohlraumresonanzen innerhalb der Schirmwände im Bereich des Koppelschlitzes bilden.According to a further preferred development, the following relationship exists between the width B between opposing via-contacting devices in the region of the coupling opening and the length L of the through-connection device in the region of the coupling opening $$L<\frac{1}{\sqrt{{\left(\frac{2\cdot f_{\mathit{res}}\cdot \sqrt{{\mathrm{<figure-callout\; id="0"\; label="\epsilon \; r\; c"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">\epsilon \; </figure-callout>}}_{\mathrm{r}}}}{c_{}}\right)}^2-{\left(\frac{1}{B}\right)}^2}}.$$

where C _{0 stands} for the speed of light in vacuum, ε _r for the dielectric permittivity of the substrate and f _res for a resonant frequency of a stimulable waveguide mode, which is to be provided above a useful signal frequency band. This is an advantage for the dimensioning of the via or via walls in the region of the coupling slot, since in this way it is avoided that undesired resonance frequencies form cavity resonances within the screen walls in the region of the coupling slot.

According to a further preferred development, the resonance frequency has a greater distance than about a few percent above the useful signal frequency band. In this way, a safe avoidance of resonance phenomena is ensured.

According to a further preferred development, the device is dimensioned for useful signals in a frequency band between 20 GHz and 30 GHz. For example, the device is suitable for use in an SRR (short range radar) application.

According to a further preferred development, the plated-through device consists of discrete through-connection elements, which are arranged laterally adjacent to one another, preferably forming an electromagnetic-shielding wall. This has the advantage of good shielding at cost-to-produce via elements, with the choice of the distance depends on the frequency.

According to a further preferred development, the discrete feedthrough elements are formed round and / or cylindrical. A simple production can be ensured.

According to a further preferred development, the through-connection device forms a continuous wall. This offers the advantage of a closed shielding device, for example in the form of a metallic layer, which allows almost no electromagnetic input or output couplings.

According to a further preferred development, the through-connection device is provided continuously in the region longitudinally adjacent to the end of the strip line.

The advantage here is a complete shielding of the stripline.

According to a further preferred development, the through-connection device is provided with a gap in the region longitudinally adjacent to the end of the strip line. As a result, little electromagnetic emission is emitted or absorbed at a slightly reduced production cost.

According to a further preferred refinement, the microstrip line is arranged closer to the ground area provided with the coupling opening than to the other ground area in the substrate or vice versa. This has the advantage of an asymmetric structure, which e.g. when coupling another microstrip line via the coupling opening is needed.

According to the invention, the microstrip line is arranged approximately equidistantly between the ground plane provided with the coupling opening and the other ground plane in the substrate. This provides the advantage of a simple arrangement.

According to a further preferred development, the planar coupling device forms a second microstrip line in another plane, which is provided with galvanic isolation for the electromagnetic coupling of this further microstrip line. In this way, a signal transmission device is advantageously provided with galvanic isolation.

According to a further preferred embodiment, both microstrip lines are formed substantially the same and overlap in the longitudinal direction by a two-fold predetermined distance, which preferably corresponds approximately to half the wavelength of the coupling useful signal. Thus, maximum electromagnetic coupling between the two microstrip lines is ensured.

According to a further preferred development, the coupling opening is provided in parallel with the ground surfaces in slot-shaped and / or rectangular fashion. This allows a simple cost-effective-to-produce coupling-hole layout in the ground plane and provides good coupling through the slot.

DRAWINGS

Embodiments of the invention are illustrated in the drawings and explained in more detail in the following description.

Show it:

Fig. 1

an oblique view of a section for explaining a first embodiment of the present invention;

Fig. 2

an oblique view for explaining the first embodiment of the present invention;

Fig. 3

a plan view of a schematic emitting device for explaining a second embodiment of the present invention;

Fig. 4

a simulation diagram for explaining the operation of with reference to Fig. 3 explained emitting device;

Fig. 5A, B

a schematic representation of a galvanically isolated coupling device for explaining a third embodiment of the present invention, wherein Fig. 5A a longitudinal section and Fig. 5B a cross section along the sectional plane A illustrates;

Fig. 6A, B

a schematic representation of a conventional slit-coupled Planarstrahlers, wherein Fig. 6A a longitudinal section and Fig. 6B a plan view illustrates;

Fig. 7A, B

a schematic representation of with reference to Fig. 6A , B illustrated arrangement with an additional mechanical reinforcement, wherein Fig. 7A a longitudinal section and Fig. 7B a plan view illustrates; and

Fig. 8A, B

a schematic representation of a conventional slot-coupled Planarstrahlers with a single-ended tripple-line supply, wherein Fig. 8A a longitudinal section and Fig. 8B a cross section along the sectional plane A illustrates.

DESCRIPTION OF THE EMBODIMENTS

In the figures, the same reference numerals designate the same or functionally identical components.

Fig. 1 shows a schematic oblique view of a slit-coupled radiating device for explaining a first embodiment of the present invention.

In Fig. 1 For example, a microstrip line 10 is embedded in a substrate 11. This substrate is preferably suitable for high frequency and has, for example, a low temperature cofired ceramic (LTCC), which has good dielectric properties at a low attenuation. Above the microstrip line 10, preferably parallel thereto, a first ground plane 12 is provided by the substrate 11 separately.

The lower end of the illustrated arrangement is formed by a second ground plane 13, which, like the first ground plane, consists of an electrically conductive material, preferably a metal. The first ground plane 12 has a coupling opening 14, which is preferably rectangular and / or slot-shaped, and which has a predetermined distance d (not shown) with respect to an abrupt end 10 'of the microstrip line 10. This coupling opening 14 is aligned in the Y-direction centered to the strip line 10 and the end of the strip line 10 'and at right angles thereto, similar to a cross. The predetermined distance in the X direction between the slot opening 14 and the end 10 'of the strip line 10 corresponds to approximately one quarter of the line wavelength, ie λ / 4, of the transmitted on the strip line 10 useful signal f, which in this example a bandwidth of the frequency band F in the range between 20 GHz and 30 GHz.

Between the upper ground surface 12, in which the coupling slot 14 is provided, and the lower ground surface 13, a feedthrough device 15 is provided, which consists of individual via elements 15 'according to the present embodiment. The individual via elements 15 'are preferably approximately round and / or cylindrical and provide a palisade wall-like shielding device.

A planar coupling device 16 serves in the present case as a planar radiator, which is excited by the decoupled by the coupling opening 14 electromagnetic field to a resonance. The planar coupling device 16 is preferably aligned parallel to the coupling opening 14. The side edges of the planar rectangular radiator 16 provided here are preferably aligned parallel to the edges of the coupling opening 14, ie in the X and Y directions. According to the present embodiment, the microstrip line 10 has an impedance transformer 17 in the region of the coupling slot 14 and in front of the abrupt end 10 'of the strip line, which is used as needed for impedance matching. In the region of the coupling slot 14, the plated-through device 15 expands so as to reconnect longitudinally adjacent the end section 10 'of the stripline 10 and thus represents a closed shielding device.

For shielding such triplate lines and consequently for avoiding propagatable or resonant waveguide modes in the substrate 11, a plated-through device 15 or also continuously closed shielding walls around the stripline 10 is suitable. Instead of providing massive walls, it is in practice advantageous to provide the via device 15 in the form of individual vias 15 '(vias) which represent a continuous electrically conductive wall to each other on the high frequency side by a sufficiently small lateral spacing of the vias. The maximum shielding effect is determined by the correct dimensioning of the distance and diameter of the individual via elements 15 '. In order to prevent propagatable waveguide modes, the distance of the walls from each other, i. For example, the distance between the lying on the one side of the strip line 10 via-hole device for the distance of lying in the Y direction on the other side of the strip line through hole 15, do not exceed a certain value.

wobei C₀ der Lichtgeschwindigkeit im Vakuum (C₀ = 3 · 10⁸ m/s), a dem Abstand der Durchkontaktierungseinrichtungen 15 bzw. Via-Wände und ε_r der dielektrischen Permittivität des Substratmaterials entspricht. Folglich muss die Ungleichung $$\mathrm{a}<\frac{c_0}{2\cdot f_g\cdot \sqrt{{\epsilon }_r}}$$

erfüllt sein, damit bis zur Frequenz fg kein Hohlleitermode angeregt wird. Der Abstand a ist je nach elektrischer Auswirkung der Formgebung der Vias bzw. deren Abständen sowie des zusätzlichen (vergleichsweise geringen) Einflusses der Signalleitung 10 zu reduzieren.The first propagatable waveguide mode of a rectangular waveguide, which is approximately present here, is the TE ₁₀ mode, a mode with transverse electric field (TE) viewed in cross section. The cutoff frequency of this mode is $${\mathrm{f}}_{\mathrm{G}}=\frac{{\mathrm{<figure-callout\; id="0"\; label="c"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">c</figure-callout>}}_0}{2a\sqrt{{\epsilon }_r}}$$

where C ₀ corresponds to the speed of light in vacuum (C ₀ = 3 × 10 ⁸ m / s), a the distance between the via devices 15 or via walls and ∈ _r the dielectric permittivity of the substrate material. Consequently, the inequality must be $$\mathrm{a}<\frac{{\mathrm{<figure-callout\; id="0"\; label="c"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">c</figure-callout>}}_0}{2\cdot f_G\cdot \sqrt{{\epsilon }_r}}$$

be satisfied, so that up to the frequency fg no waveguide mode is excited. The distance a is to be reduced depending on the electrical effect of the shape of the vias or their distances and the additional (relatively low) influence of the signal line 10.

If this via wall 15 was now guided at an appropriate distance a parallel to the signal line 10, then this wall 15 would intersect in the region of the coupling opening 14 with this orthogonally oriented coupling opening 14, whereby the mode of operation of the coupling slot 14 and thus of the antenna or transmission device would not more would be ensured. Therefore, it is necessary to significantly increase the distance of the via walls in the vicinity of the coupling slot 14 in order to be able to reduce it back to the original value only behind the slot 14 in the region of the open-ended signal line 10 '. Then behind the idle end 10 'of the microstrip line 10, a merging of the via walls 15 would be possible, but not necessarily required, since due to the existing there small distance of the via walls no stimulation substrate or waveguide modes would be possible. On the other hand, in order to achieve a maximum shielding effect and also to prevent electromagnetic couplings from the outside into the arrangement, the through-connection device 15, ie the walls, is preferably brought together longitudinally adjacent to the open-ended signal line 10 '.

With regard to the dimensioning or structuring of the via device 15 or the via walls in the region of the coupling opening 14, it must be taken into account that as the distance a of these walls increases, the limit frequency fg of the waveguide mode decreases, generally below the useful frequency f Antenna itself, so that the impairment of the function of the coupling opening 14 through the via walls 15 minimal or in a design of the arrangement is considered. On the other hand, this entails the danger that cavity resonances may form within these screen walls 15 with the greatly increased distance B in the region of the coupling opening 14, which greatly impair the function of the antenna if these possibly occurring undesired resonant frequencies lie in the useful frequency range. In order to prevent this now targeted, the length L of the via walls 15 in the X direction in the region of the coupling opening at the enlarged distance B of the screen walls 15 in the Y direction is to be selected accordingly.

, wobei p, m und n ganzzahlige Indizes sind, C₀ die Vakuumlichtgeschwindigkeit und ε_r die dielektrische Permittivität des nichtleitenden Füllmaterials darstellt. Für den hier relevanten TE₁₀-Mode gilt m = 1 sowie n = 0, so dass die möglichen Resonanzfrequenzen zwar von der Breite B, aber nicht von der Höhe H abhängen. Der ganzzahlige Index p muss bei TE-Moden größer als Null sein. Daraus ergibt sich die erste anregbare Hohlraumresonanz des TE₁₀-Modes gemäß $$f_{\mathit{res}}=\frac{c_0}{2\sqrt{{\epsilon }_r}}\sqrt{{\left(\frac{1}{L}\right)}^2+{\left(\frac{1}{B}\right)}^2}$$

In a completely closed, dielectrically filled, rectangular waveguide resonator of width B, height H and length L with ideally conducting electrical walls, possible discrete resonance frequencies result according to the following relationship: $$f_{\mathit{res}}=\frac{{\mathrm{<figure-callout\; id="0"\; label="c"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">c</figure-callout>}}_0}{2\sqrt{{\epsilon }_r}}\sqrt{{\left(\frac{p}{L}\right)}^2+{\left(\frac{m}{B}\right)}^2+{\left(\frac{n}{H}\right)}^2}$$

Wherein p, m and n are integer indices, C ₀ is the vacuum velocity of light and ε _{r is} the dielectric permittivity of the dielectric filling material. For the TE ₁₀ mode relevant here, m = 1 and n = 0, so that the possible resonance frequencies depend on the width B, but not on the height H. The integer index p must be greater than zero for TE modes. This results in the first excitable cavity resonance of the TE ₁₀ -mode according to $$f_{\mathit{res}}=\frac{{\mathrm{<figure-callout\; id="0"\; label="c"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">c</figure-callout>}}_0}{2\sqrt{{\epsilon }_r}}\sqrt{{\left(\frac{1}{L}\right)}^2+{\left(\frac{1}{B}\right)}^2}$$

In the design of the antenna with slotted coupling and via-shielding 15 of the signal line 10 is now to ensure that the cutoff frequency of the waveguide-like resonance according to equation (1), in which case a = B is set, may be below the Nutzsignalfrequenzbandes F, that but the first resonant frequency according to equation (4) must be above the Nutzsignalfrequenzbandes F to prevent interference with the operation of the transmission device 16 and / or antenna.

Moreover, according to the present embodiment FIG. 1 in the dimensioning of the shielding device or the through-connection device 15, the use of discrete via elements 15 'with a certain lateral distance from one another instead of closed metallic walls influences the cutoff frequency of the waveguide modes. In addition, it must be considered that the resonator in the region of the coupling slot does not have completely enclosed walls, as in the theoretical model, but large-scale coupling and uncoupling, for example in the region of widening of the via walls 15, which influence the resonant frequency accordingly. The coupling slot 14 itself also influences the resonance frequency, just as the signal line 10, 10 'running idle below the coupling opening 14 can itself change the resonance frequency.

Fig. 2 shows a schematic perspective view for explaining the first embodiment of the present invention.

In Fig. 2 is a section of the arrangement according to Fig. 1 shown. The microstrip line 10 is embedded in a dielectric substrate between a first ground plane 12 and a second ground plane 13. The two ground planes 12, 13 are interconnected via electrically conductive via elements 15 ', which form a via device 15 or a shield device. According to the illustrated embodiment, the stripline 10 is plane-parallel and symmetrical between two parallel ground planes 12 and 13, that is, in a symmetrical triplate arrangement. Preferably, the stripline 10 has an approximately rectangular cross-section, whereas the individual each laterally adjacent via-contacting elements 15 'are in particular cylindrically shaped.

Fig. 3 shows a schematic plan view of a radiating device for explaining a second embodiment of the present invention.

In Fig. 3 a radiating device according to the invention is shown, wherein this is substantially the same from the reference to Fig. 1 illustrated embodiment, that the feedthrough device 15 in the present case is not made of individual via elements 15 ', but from continuous electrically conductive walls, which are arranged between the first and the second ground surface electrically contacting. In this case, the usable frequency band F is preferably in the range of 22 GHz to 26 GHz.

The according to Fig. 3 illustrated triplate structure is asymmetrical, ie, the height of the substrate 11 on the signal line 10 to the first ground plane 12 is 150 microns, and the height of the substrate 11 below the signal line 10 to the second ground plane 13 is for example 450 microns (both ground surfaces in the Top view according to Fig. 3 not shown). The length of the coupling slot, ie its extension in the Y direction, is for example 2.6 mm, and the dielectric constant ε _{r of} the ceramic substrate material is ε _r = 7.7. Thus, the cutoff frequency of the waveguide mode TE ₁₀ in the signal line 10 at a small distance a of the via 15 and the via walls above the usable frequency band F, the distance a according to equation (2) must be less than 2.46 mm and is for example, selected to a = 1.9 mm.

So that the electromagnetic coupling is not impaired by the coupling opening 14 of the shielding device 15, in the region of the coupling slot 14, the distance of the via walls B is increased to, for example, 3.6 mm. The cut-off frequency fg of the TE ₁₀ -mode thereby decreases according to equation (1) to about 15 GHz. In order for the first resonance frequency f _{res of} this mode to be above, for example, 27 GHz in order to ensure a 1 GHz frequency spacing to the usable frequency band F, according to equation (4), the length L must be less than 2.4 mm. In order additionally to compensate for the above-mentioned influences on the resonance frequency f _res , L in the present exemplary embodiment is preferably chosen to be 1.2 mm.

In Fig. 4 is the amplitude profile of the reflection factor as a simulation result of a full wave analysis of the entire antenna array according to Fig. 3 shown. At about 27.7 GHz, a resonance clearly shows, since here the reflection factor has a high amplitude, which corresponds exactly to the described waveguide resonance of the TE ₁₀ -mode, which results from an analysis of associated field distribution images (not shown). At the same time, in the useful frequency band F between 22 GHz and 26 GHz a good reflection attenuation, which is greater than 12 dB, and, moreover, a very smooth course of the adaptation, from which the impairment can be excluded by other resonance-like effects in this frequency range. The course of the reflection factor can be as desired in large areas by appropriate dimensioning or structuring of planar coupling device 16 or

Planar radiator, coupling opening 14 or coupling slot, signal line 10 and impedance transformer 17 set.

In Fig. 5A a coupling device of an electromagnetic signal is shown under galvanic isolation. According to this third embodiment of the present invention, two microstrip lines 10 in a dielectric substrate 11 are separated by a ground plane 12 provided with a coupling opening 14. The lower stripline 10 extends in the illustration to the left and has in the region adjacent to the coupling opening 14 their idler end 10 ', whereas the upper stripline 10 extends in the drawing to the right and their free-running left end 10' in the region adjacent to the coupling slot 14 has. The arrangement is constructed point-symmetrical to the center of the coupling slot 14.

Essentially, the arrangement in the lower region corresponds to an asymmetrical triplate supply, which, however, does not transmit its coupled-out field to a planar radiator (16, not shown here) but into a further stripline 10. In this way, therefore, no antenna element, but a coupling device is provided, which transmits the signal via an electromagnetic coupling of a signal of a stripline in a plane the signal galvanically separated to a second stripline 10 in another plane. In the Fig. 5A Not shown Durchkontaktierungseinrichtung or screen walls are in the stripline and in particular in the region of the coupling opening 14, as described above, structured or dimensioned.

In Fig. 5B is the coupling device according to Fig. 5A shown in cross section, wherein the through-hole device is not illustrated for clarity, but still arranged as above.

Although the present invention has been described above with reference to preferred embodiments, it is not limited thereto but, within the scope of the claims, modifiable in a variety of ways.

In particular, the materials mentioned for the dielectric substrate, the ground areas and the strip line can be seen by way of example. In addition, the design of the coupling slots, the planar coupling device and the strip line is not necessarily rectangular, but may also have round, oval or polygonal cross-sections or plan views. In particular, the through-contact device or shielding walls do not have to run at right angles to each other, but may have rounded transitions.

Claims (18)

1. Apparatus for transmission or emission of radio-frequency waves, having:

   a micro stripline (10), which is provided with an end (10'), in a substrate (11) for transmission of a radio-frequency useful signal;

   a first ground plane (12) and a second ground plane (13), which are provided on opposite sides of the substrate (11), in order to form a TEM waveguide arrangement;

   a coupling opening (14) in the first ground plane (12) at a predetermined distance (d) from the end (10') of the stripline (10) for outputting the radio-frequency useful signal;

   a through-contacting device (15) for conductive connection of the first ground plane (12) to the second ground plane (13) in the side periphery of the coupling opening (14);
   wherein the through-contacting device (15) is designed such that, at a given frequency (f) of the useful signal, it prevents the propagation of waveguide modes and the excitation of waveguide mode resonances in a useful frequency band (f) of the useful signal;
   and

   a planar coupling device (16) for reception and transmission or emission of the output radio-frequency useful signal;
   characterized in that

   the through-contacting device (15), formed as two walls, extends from both sides and parallel to the micro stripline (10) over a specific length in each of the areas before, in the area of and at the end (10'), after the coupling opening (14), with the walls before and after the coupling opening (14) being at a distance (a) apart, where $\mathrm{a}<{\mathrm{c}}_{\mathrm{o}}/\left(2\mathrm{f}\sqrt{{\epsilon }_r}\right),$

   

   where c_o is the speed of light in a vacuum, and ε_r is the electrical permittivity of the substrate (11), and the walls broaden in the area of the coupling opening (14) up to a mutual separation (B).
2. Apparatus according to Claim 1,
   characterized in that
   the following relationship exists between the greater distance (B) between the walls of the through-contacting device (15) in the area of the coupling opening (14) and the length (L) of the through-contacting device in the area of the coupling opening (14): $$\mathrm{L}<\frac{1}{\sqrt{{\left(\frac{2\cdot f_{\mathit{res}}\cdot \sqrt{{\epsilon }_{\mathrm{r}}}}{c_0}\right)}^2-{\left(\frac{1}{B}\right)}^2}},$$

   

   
   where f_res is a resonant frequency of a waveguide mode which can be excited and can be provided above the useful signal frequency band (F).
3. Apparatus according to Claim 2,
   characterized in that
   the resonant frequency f_res is more than a few per cent above the useful signal frequency band (F).
4. Apparatus according to one of the preceding claims,
   characterized in that
   the apparatus is designed for useful signals in a useful signal frequency band (F) between 20 GHz and 30 GHz.
5. Apparatus according to one of the preceding claims, characterized in that
   the through-contacting device (15) is composed of discrete through-contacting elements (15') which are arranged laterally adjacent to one another, forming a wall.
6. Apparatus according to Claim 5,
   characterized in that
   the discrete through-contacting elements (15') are round and/or cylindrical.
7. Apparatus according to one of the preceding Claims 1-4,
   characterized in that
   the through-contacting device (15) forms a continuous wall.
8. Apparatus according to one of the preceding claims,
   characterized in that
   the through-contacting device (15) is provided in a continuous form in the area longitudinally adjacent to the end (10') of the stripline (10).
9. Apparatus according to one of the preceding claims,
   characterized in that
   the through-contacting device (15) is provided with a gap in the area longitudinally adjacent to the end (10') of the stripline (10).
10. Apparatus according to one of the preceding claims,
    characterized in that
    the micro stripline (10) is arranged closer to that ground plane (12) which is provided with the coupling opening (14) than to the other ground plane (13) in the substrate (11), or vice versa.
11. Apparatus according to one of the preceding Claims 1-9,
    characterized in that
    the micro stripline (10) is arranged approximately equidistant between that ground plane (12) which is provided with the coupling opening (14) and the other ground plane (13) in the substrate (11).
12. Apparatus according to one of the preceding claims,
    characterized in that
    the micro stripline (10) has an integrated impedance transformer (17) in the area of the coupling opening (14).
13. Apparatus according to one of the preceding claims,
    characterized in that
    the planar coupling device (16) forms a second micro stripline (10) on another plane, which is provided in a galvanically isolated form, for electromagnetic coupling of this further micro stripline (10).
14. Apparatus according to Claim 13,
    characterized in that
    the planar coupling device (16) can be caused to resonate by means of the coupling opening (14), and can therefore be excited for emission.
15. Apparatus according to Claim 14,
    characterized in that
    the coupling opening (14) can itself be made to resonate and can thus be excited for emission.
16. Apparatus according to Claim 15,
    characterized in that
    the two micro striplines (10) are designed to be essentially identical and overlap by twice a predetermined distance (d) in the longitudinal direction, which distance (d) preferably corresponds to approximately half the wavelength of the coupled useful signal.
17. Apparatus according to one of the preceding claims,
    characterized in that
    the coupling opening (14) is provided in the form of a slot and/or in the form of a rectangle parallel to the ground planes (12, 13).
18. Apparatus according to one of the preceding claims,
    characterized in that
    the substrate (11) has a ceramic material, preferably low temperature cofired ceramic (LTCC).

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