EP1421676B1 - Vorrichtung zur frequenzumsetzung von hochfrequenzsignalen unter verwendung von aperiodischen signalen und verfahren - Google Patents

Vorrichtung zur frequenzumsetzung von hochfrequenzsignalen unter verwendung von aperiodischen signalen und verfahren Download PDF

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EP1421676B1
EP1421676B1 EP02754073A EP02754073A EP1421676B1 EP 1421676 B1 EP1421676 B1 EP 1421676B1 EP 02754073 A EP02754073 A EP 02754073A EP 02754073 A EP02754073 A EP 02754073A EP 1421676 B1 EP1421676 B1 EP 1421676B1
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Prior art keywords
signal
mixer
signals
circuit
mixing
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English (en)
French (fr)
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EP1421676A2 (de
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Tajinder Manku
Yang Ling
William Kung
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Icera Canada ULC
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Sirific Wireless ULC
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • H04B1/26—Circuits for superheterodyne receivers
    • H04B1/28—Circuits for superheterodyne receivers the receiver comprising at least one semiconductor device having three or more electrodes
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1433—Balanced arrangements with transistors using bipolar transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1441—Balanced arrangements with transistors using field-effect transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1466—Passive mixer arrangements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1483—Balanced arrangements with transistors comprising components for selecting a particular frequency component of the output
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
    • H03D2200/0001—Circuit elements of demodulators
    • H03D2200/0025—Gain control circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
    • H03D2200/0041—Functional aspects of demodulators
    • H03D2200/0043—Bias and operating point
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
    • H03D2200/0041—Functional aspects of demodulators
    • H03D2200/0084—Lowering the supply voltage and saving power
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
    • H03D2200/0041—Functional aspects of demodulators
    • H03D2200/009—Reduction of local oscillator or RF leakage

Definitions

  • the present invention as it is defined in independent claim 1 relates generally to communications, and more specifically to a method and apparatus of modulating and demodulating RF (radio frequency) and baseband signals using aperiodic mixing signals.
  • the preferred embodiment of the invention satisfies the need for an inexpensive, high-performance, fully-integrable, multi-standard transceiver.
  • the original (or baseband) signal may be, for example: data, voice or video.
  • These baseband signals may be produced by transducers such as microphones or video cameras, be computer generated, or transferred from an electronic storage device.
  • the high frequencies provide longer range and higher capacity channels than baseband signals, and because high frequency (HF) radio frequency (RF) signals can effectively propagate through the air, they can be used for wireless transmissions as well as hard wired or fibre channels.
  • HF high frequency
  • RF radio frequency
  • RF signals are electromagnetic signals; that is, waveforms with electrical and magnetic properties within the electromagnetic spectrum normally associated with radio wave propagation.
  • Wired communication systems which employ such modulation and demodulation techniques include computer communication systems such as local area networks (LANs), point-to-point communications, and wide area networks (WANs) such as the Internet. These networks generally communicate data signals over electrically conductive or optical fibre channels.
  • Wireless communication systems which may employ modulation and demodulation include those for public broadcasting such as AM and FM radio, and UHF and VHF television.
  • Private communication systems may include cellular telephone networks, personal paging devices, HF radio systems used by taxi services, microwave backbone networks, interconnected appliances under the Bluetooth standard, and satellite communications. Other wired and wireless systems which use RF modulation and demodulation would be known to those skilled in the art.
  • the super-heterodyne receiver uses a two-step frequency translation method to convert an RF signal to a baseband signal.
  • Figure 1 presents a block diagram of a typical super-heterodyne receiver 10.
  • the mixers labelled M1 12 and M2 14 are used to translate the RF signal to baseband or to some intermediate frequency (IF).
  • IF intermediate frequency
  • the balance of the components amplify the signal being processed and filter noise from it.
  • US-A-5 862 173 discloses an example of such a receiver.
  • the RF band pass filter (BPF1) 18 first filters the signal coming from the antenna 20 (note that this band pass filter 18 may also be a duplexer).
  • a low noise amplifier 22 then amplifies the filtered antenna signal, increasing the strength of the RF signal and reducing the noise figure of the receiver 10.
  • the signal is next filtered by another band pass filter (BPF2) 24 usually identified as an image rejection filter.
  • BPF2 band pass filter
  • the signal then enters mixer M1 12 which multiplies the signal from the image rejection filter 24 with a periodic signal generated by the local oscillator (LO1) 26.
  • the mixer M1 12 receives the signal from the image rejection filter 24 and translates it to a lower frequency, known as the first intermediate frequency (IF1).
  • IF1 first intermediate frequency
  • a mixer is a circuit or device that accepts as its input two different frequencies and presents at its output:
  • the IF1 signal is next filtered by a band pass filter (BPF3) 28 typically called the channel filter, which is centred around the IF1 frequency, thus filtering out the unwanted products of the first mixing processes; signals (a) and (c) above. This is necessary to prevent these signals from interfering with the desired signal when the second mixing process is performed.
  • BPF3 band pass filter
  • the signal is then amplified by an intermediate frequency amplifier (IFA) 30, and is mixed with a second local oscillator signal using mixer M2 14 and local oscillator (LO2) 32.
  • the second local oscillator LO2 32 generates a periodic signal which is typically tuned to the IF1 frequency.
  • the signal coming from the output of M2 14 is now at baseband, that is, the frequency at which the signal was originally generated.
  • Noise is now filtered from the desired signal using the low pass filter LPF 38, and the signal is passed on to some manner of presentation, processing or recording device.
  • this might be an audio amplifier, while in the case of a computer modem this may be an analogue to digital convertor.
  • band pass and low pass filters 18, 24, 28 and 38 used in super-heterodyne systems must be high quality devices, so electronically tunable filters cannot be used.
  • the only way to use the super-heterodyne system in a multi-standard/multi-frequency application is to use a separate set of off-chip filters for each frequency band.
  • Direct-conversion transceivers attempt to perform up and down conversion in a single step, using one mixer and one local oscillator. In the case of down-conversion to baseband, this requires a local oscillator with a frequency equal to that of the input RF signal. If the LO signal of a direct conversion receiver leaks into the signal path, it will also be demodulated to baseband along with the input signal, causing interference. This LO leakage problem limits the utility of direct-conversion transceivers.
  • transceivers i.e. devices having both a receiver and a transmitter
  • ICs integrated circuits
  • Radio Frequency Integrated Circuits RFICs
  • MMICs monolithic microwave integrated circuits
  • RFICs Radio Frequency Integrated Circuits
  • MMICs monolithic microwave integrated circuits
  • PLD Programmable Logic Device
  • Other implementations are related to the ability to distinguish between and cope with very different modulation schemes.
  • United States Patent Serial No. 5,649,288 titled: “Dual-function double balanced mixer circuit”.
  • the invention provides a circuit topology and methodology which may be applied to up or down conversion of electrical signals.
  • This design employs a two-stage mixing process, the first mixing being performed with an active mixer, and the second using a passive mixer. Between these two mixers a high pass filter is placed.
  • the active mixer amplifies the incoming signal as well as mixing it.
  • the high pass filter reduces second order distortion (1/f noise) generated by the active mixer, to an acceptable level before passing the signal to the passive mixer.
  • the high pass filter also filters off low frequency common mode and differential signals from entering the passive mixer. These types of signals may arise from the DC offsets of the first mixer and second order distortion from the first mixer. Because the second mixer is a passive mixer, it does not introduce 1/f noise.
  • the result is an inexpensive, high-performance, multi-standard/multifrequency transceiver which can be fully integrated.
  • One aspect of the invention is defined as a modulation or demodulation circuit comprising: an active mixer for receiving an input signal and mixing the input signal with a first aperiodic mixing signal, the active mixer having adjustable performance; a high pass filter electrically connected to the output of the active mixer, generating a filtered, mixed signal; and a passive mixer connected to the output of the high pass filter, for receiving the filtered, mixed signal and a second aperiodic mixing signal, and mixing the filtered, mixed signal with the second aperiodic mixing signal.
  • Another aspect of the invention is defined as a method of up or down conversion comprising the steps of: active mixing an input signal with a first aperiodic mixing signal; high pass filtering the first mixed signal; and passive mixing the filter signal with a second aperiodic mixing signal.
  • FIG. 2 A circuit which addresses a number of the objects outlined above is presented as a block diagram in Figure 2 .
  • This figure presents a modulator or demodulator topology 50 in which an input signal x(t) is mixed with two aperiodic oscillator signals (labelled ⁇ 1 and ⁇ 2 ) which effect the desired modulation or demodulation. While this block diagram may seem similar to known topologies, it has a number of features which are quite distinct from that of the prior art.
  • the input signal x(t) is mixed with the first aperiodic oscillator signal ⁇ 1 using an active mixer 52.
  • the resulting signal then passes through a high pass filter (HPF) 54, which removes DC components generated by the active mixer 52.
  • HPF high pass filter
  • the filtered signal is then mixed with the second aperiodic oscillator signal ⁇ 2 by means of passive mixer 56.
  • Typical modulation and demodulation circuits use local oscillator signals which do not vary in frequency over time (i.e. they are mono-tonal, or periodic). Recently, however, there has been a growing realization that local oscillator signals whose frequencies vary over time, may effect modulation and demodulation far more effectively.
  • the preferred manner of such aperiodic local oscillator signals are described with respect to Figure 7 hereinafter, but the teachings of the invention may also be used with aperiodic signals described by others, including the following:
  • Figure 7 describes pairings of mixing signals which together emulate local oscillator signals, but do not have significant power at the frequencies of the local oscillators being emulated. In this way, the frequency translation is still effected, but self-mixing and unwanted mixing products are avoided.
  • Active mixers such as that used in Figure 2 are distinct from passive mixers in a number of ways:
  • active mixers are nonlinear devices, they generate what is referred to as "1/f" noise or second-order distortion.
  • This noise is called 1/f noise because its power spectra is generally inversely proportional to the frequency - in other words, the power of the noise signal is higher close to DC (direct current).
  • this second-order distortion is removed using the high pass filter (HPF) 54. Because the second mixer 56 is a passive mixer, it does not introduce new second-order distortion into the signal. Thus, this topology provides the benefits of active mixing, without introducing second-order distortion into the output signal.
  • HPF high pass filter
  • Figure 3 presents an exemplary schematic diagram of an implementation of the invention having the following characteristics:
  • the input to this circuit is a simple, single-ended RF signal labelled RFin (single-ended means there is a single potential with respect to ground), and the output is a differential RF or baseband signal labelled RFout+ and RFout-.
  • the frequency of the Rfin signal is modulated or demodulated, up or down, to RFout+ and RFout- in accordance with the parameters of the mixing signals.
  • each of the embodiments of Figures 3 , 5 and 6 four mixing signals are used, in two stages.
  • these pairs of mixing signals exhibit the synergy described above, that is, the pair of mixing signals ⁇ 1P and ⁇ 2P can emulate a local oscillator.
  • each of the circuits in Figures 3 , 5 and 6 is powered by a positive voltage source ( V DD ) and a negative voltage source ( V SS ).
  • the active mixer 70 is a single-balanced mixer, it will generate less noise than a double-balanced mixer (see the description of a double-balanced mixer with respect to Figure 5 ). This is simply because there are fewer noise contributors in the single-balanced design. However, the single-balanced mixer is less immune to external noise, particularly common mode noise.
  • the structure of the active mixer 70 is also presented in the block diagram of Figure 4 where the components are collected into three groups: a Mixer Block 120, a Gain Control Block 122, and a Current Source Block 124.
  • the Gain Control Block 122 is a gain-providing stage that consists of a number of input transistors, shown in Figure 3 as transistors 76 and 78. Each input transistor is fed with the same input signal RFin, and their outputs are fed to the Mixer Block 120 as amplified signals. The selection of the input transistor to be used at a particular time is accomplished using switches G 1 , G 2 ... G n .
  • the conditions of operation, and hence performance, of the Gain Control Block 122 are further alterable through the Current Source Block 124 which provides a variable amount of biasing current to the Gain Control Block 122.
  • the amount of biasing current is controlled by the application of various conditions to current control inputs C 1 , C 2 ... C n .
  • Figure 4 also presents the relationship of the Gain Control Block 122 to the Current Source Block 124.
  • the Current Source Block 124 provides a means to control the amount of current being fed to the Gain Control Block 122 so that this current is not drawn entirely from the Mixer Block 130.
  • the Current Source Block 124 comprises a number of controllable current sources. In Figure 3 , only one current source 80 is shown, but multiple sources and electronic switches could be employed. The control is limited to the ability to enable or disable a particular current source, which is accomplished using the current control inputs C 1 , C 2 ... C n .
  • the current source 80 is arranged to provide the current required by the RF amplifier transistors 76 and 78 thereby requiring the active mixer switches 82 and 84 to provide only a small proportion of the current required for the Gain Control Block 122.
  • the reduced contribution of current to the Gain Control Block 122 via the active mixer switches 82 and 84 results in less noise being referred to the inputs of the RF amplifier transistors 76 and 78 and also less noise being generated by the resistive load of the active mixer circuit 7 0, resulting in improved overall noise performance.
  • the total current flowing through the RF amplifier transistors 76 and 78 can be maintained at a level sufficient to ensure their operation at the required gain and linearity.
  • the linearity of the active mixer 70 is also improved by this current injection between the input amplifier and the mixer switching transistors 86 and 88 because the current flowing through the input amplifier (Gain Control Block 122) can be substantially independent of that flowing through the mixer switches transistors 86 and 88. This independence means that the switches can be biased optimally without prejudicing the noise and gain performance of the input amplifier stage (Gain Control Block 122 ).
  • the balanced mixer block 120 is effected by two separate transistor and resistor pairings, which receive the amplified RF signals from the Gain Control Block 122.
  • the amplified RF signal is passed to the sources of the switching transistors 82 and 84, and the drains of transistors 82 and 84 are connected independently to load resistors 86 and 88.
  • ⁇ 1P complementary mixing signals
  • a differential output signal is received at Out+ and Out-.
  • the value of the load resistors 86 and 88 is selected to provide the best bias conditions for the mixer transistors.
  • the RFin signal to the active mixer 70 may also by regulated using an input bias voltage and resistor 89.
  • the RF amplifier transistors 76, 78 in the Gain Control Block 122 may be degenerated by an impedance to improve its linearity.
  • Reactive degeneration generally has better noise performance than resistive degeneration, and inductive degeneration is more current efficient than both resistive and capacitive degeneration.
  • negative resistance may cause oscillation.
  • field effect transistors are used as shown in Figure 3 .
  • FETs generally have sufficient linearity that degeneration is not necessary.
  • Outputs from the two switching transistors 86, 88 in the active mixer 70 are then passed through a pair of high pass filters 72, each consisting of a capacitor 90, 92 and two resistors 94, 96, 98 and 100.
  • the use of the resistors 94, 96, 98 and 100 in the configuration of a voltage divider not only serves to drain the capacitors 90, 92 of the high pass filter, but also sets the common mode voltage for the next mixing stage (note that these resistors 94, 96, 98 and 100 could be active resistors).
  • the cut-off frequency of the pair of high pass filters 72 can be very low (either low with respect to the carrier frequency or close to DC, depending on the application and expected signals). As a result it may be considered to function almost entirely in the manner of a voltage divider.
  • the pair of high pass filters 72 might be effected in other manners, for example, in the form of an 'active resistor' network.
  • the outputs of the pair of high pass filters 72 are then passed to the inputs of the respective halves of a differential, passive mixer 74, whose other inputs are the aperiodic signals ⁇ 2P and ⁇ 2N, which work in concert with the ⁇ 1P and ⁇ 1N mixing signals used in the active mixer 70.
  • the passive mixer 74 comprises a known design having four transistors 102, 104, 106 and 108. Other architectures could also be used.
  • this second mixing stage completes the emulation of the local oscillator mixing, frequency translating the input RFin signal to the desired output signal Out+ / Out-. If this circuit is being used to demodulate a signal down to baseband, as it would in the case of a radio receiver, it may then be desirable to pass the outputs of the passive mixer 74 through a low pass filter 75 to remove any significant out-of-band signals.
  • This low pass filter 75 may be a simple resistor/capacitor filter as known in the art and as shown in Figure 3 .
  • One of the further benefits of this design is the use of simple resistive elements to fix the active mixer biasing voltages. This assists in the selection of optimal performance parameters for the second passive mixer.
  • the linearity of an active mixer is dependent on the biasing voltage of transistors, thus, a simple voltage divider is used as the load of mixer.
  • the optimum biasing must be found through simulation or other techniques. The bias voltage applied to each of the drains of the active mixer switches is thereby selected and fixed to that necessary for optimum linearity during design.
  • FIG. 5 Another embodiment of the invention is presented in Figure 5 in which the input signal is available as a differential signal (i.e. having positive and negative potentials with respect to ground).
  • This circuit is very much the same as that of Figure 3 , in fact, the high pass filter 72, passive mixer 74 and optional low pass filter 75 are identical.
  • the active mixer 130 of this embodiment offers the same functionality as the single-balanced active mixer 72 of Figure 3 (i.e. electronically adjustable gain and linearity controls), except that it is a double-balanced mixer which can receive a differential RF input.
  • the circuit receives a differential voltage, RFin+, RFin- as the radio frequency (RF) input signal, an input bias voltage Vbias, and two pairs of differential aperiodic mixing signals ⁇ 1P and ⁇ 1N, and ⁇ 2P and ⁇ 2N.
  • the output from this circuit is a modulated or demodulated differential output signal RFout+ / RFout-.
  • the double-balanced mixer 130 simply comprises two single-balanced mixers with the same structure as the single-balanced mixer 70 described with respect to Figure 3 above. Each of components in the single-balanced mixer 70 is mirrored in the second half of the double-balanced mixer as follows:
  • this circuit is thus the same as that described with respect to Figures 3 and 4 above, except that the double-balanced active mixer 130 modulates or demodulates the inputs RFin+ and RFin- separately before combining them.
  • the RF amplification portion of the active mixer can be degenerated by an impedance which could be a resistor, inductor, or capacitor.
  • An RF amplifier with reactive (inductive or capacitive) degeneration has lower noise factor than that with resistive degeneration since the degeneration reactance (apart from its loss resistance) does not introduce an additional noise source.
  • inductive degeneration the gain of the driver stage decreases with frequency.
  • capacitive degeneration is not recommended since it may cause oscillation.
  • degeneration may not be required as the inherent linearity of FETs is good enough for many applications.
  • the active mixer 160 is almost identical to the single-balanced active mixer 72 of Figure 3 (i.e. electronically adjustable gain and linearity controls), except that it offers the added functionality of receiving more than one RF input, which can be electronically selected.
  • the input signal RFin1 is amplified by the amplifying transistor 76 (subject to the potential from V1 bias and resistor 89 ), and is passed to the Mixer Block (consisting of transistors 82 and 84, and resistors 86 and 88 ).
  • the input signal RFin2 passes through an identical circuit consisting of amplifying transistor 162, V2bias and resistor 164, to the Mixer Block (consisting of transistors 82 and 84, and resistors 86 and 88 ).
  • the balance of the components of the active mixer 160 operate in the same manner as components 80, 82, 84, 86 and 88 described above.
  • the remaining systems in the circuit also operate in the same manner as described above, specifically: the high pass filter 72, passive mixer 74 and optional low pass filter 75.
  • This embodiment is quite significant however, as it presents an embodiment of the invention used for a multi-band/multi-standard receiver.
  • the various RF inputs ( RFin1, RFin2 ... RFinn ) could, for example, correspond to different frequency bands being received by a cellular telephone, allowing it to pass seamlessly from a geographic area using one standard, to a geographic area using another standard.
  • This multi-band/multi-standard receiver is fully integratable and does not require expensive off-chip components such as bandpass filters.
  • mixer and ancillary components optimised for each desired standard can be implemented as separate subsystems and selected appropriately.
  • the present invention not only permits optimisation to be carried out using substantially the same components in the signal path, but also allows the optimisation to be varied during operation, potentially dependent on real time measurement of performance and operating conditions such as interference and signal strength.
  • VLO virtual local oscillator
  • the mixing signal ⁇ 2 has some amount of power within the bandwidth of the up-converted RF (output) signal, and it leaks into the signal path, then if will be suppressed by the ⁇ 1 mixing signal which has minimal power within the bandwidth of the up-converted RF (output) signal.
  • This complementary mixing suppresses interference from the mixing signals ⁇ 1 and ⁇ 2 .
  • a synthesizer for generating the time-varying mixer signals ⁇ 1 and ⁇ 2 may be comprised of a single device, or multiple devices.
  • Direct-conversion transceivers suffer from LO leakage and 1/f noise problems which limit their capabilities, while heterodyne transceivers require image-rejection techniques which are difficult to implement on-chip with high levels of performance.
  • VLO signals are complementary in that one of the ⁇ 1 and ⁇ 2 signals has minimal power around the frequency of the output signal y(t) (which is around DC if conversion is to baseband), and the other has minimal power around the centre frequency, f RF , of the input signal x(t) .
  • virtual LO signals may be generated which provide the benefits of the invention to greater or lesser degrees. While it is possible in certain circumstances to have almost no LO leakage, it may be acceptable in other circumstances to incorporate virtual LO signals which still allow a degree of LO leakage.
  • FIG. 7 An exemplary set of acceptable waveforms is presented in Figure 7 , plotted in amplitude versus time.
  • Mixing signals are typically presented herein in terms of binary 1 s and 0s, however, bipolar waveforms, ⁇ 1, may also be used.
  • Bipolar waveforms are typically used in spread spectrum applications because they use commutating mixers which periodically invert their inputs in step with a local control signal (this inverting process is distinct from mixing a signal with a local oscillator directly).
  • the two mixing signals ⁇ 1 and ⁇ 2 in Figure 7 satisfy the above criteria. Any frequency component that one mixing signal might have around DC or around the centre frequency, f RF , of the input signal x(t) , is suppressed by the other. As well, the product of the two mixing signals is that of the desired LO frequency, so the input signal x(t) will be demodulated or modulated as desired.
  • Virtual local oscillator signals may also be generated in different forms, such as using three or more complementary signals rather than the two mixing signals shown above. These and other variations are described in the following co-pending patent applications:
  • ⁇ 1I which is 90 degrees out of phase with ⁇ 1Q
  • ⁇ 2I which is 90 degrees out of phase with ⁇ 2Q.
  • the pairing of signals ⁇ 1I and ⁇ 2I must meet the function selection criteria listed above, as must the signal pairing of ⁇ 1Q and ⁇ 2Q.
  • synthesizer components 172, 174, 176 and 178 to generate such signals would be clear to one skilled in the art from the description herein.
  • local oscillator 172 would be designed to operate at a frequency necessary to generate the aperiodic mixing signals ⁇ 1I, ⁇ 2I, ⁇ 1Q and ⁇ 2Q, but would avoid the carrier frequency of the input signal x(t) .
  • the IQPhase control block 174 simply generates separate I and Q local oscillator signals, 90 degrees out of phase with one another.
  • the sogen 176 and siggen 178 blocks then generate the aperiodic mixing signals in accordance with the restrictions described herewith.
  • the input signal x(t) is split between two signal streams, one comprising active mixer 180, high pass filter 182 and passive mixer 184, while the other stream comprises active mixer 186, high pass filter 188 and passive mixer 190.
  • active mixer, high pass filter and passive mixer components can be embodied as described above with respect to Figures 3 through 6 . These two streams are identical expect that they are fed with different mixing signals, one being fed with the pairing of signals ⁇ 1I and ⁇ 2I, and the other with the signal pairing of ⁇ 1Q and ⁇ 2Q.
  • mixer 180 receives the input signal x(t) and mixes it with ⁇ 1I, after which mixer 182 mixes signal x(t) ⁇ 1I with ⁇ 2I to yield the in-phase component of the input signal, that is, x(t) ⁇ 1I ⁇ 2I.
  • a corresponding process occurs on the quadrature side of the circuit where mixer 186 receives the input signal x(t) and mixes it with ⁇ 1Q; following which mixer 190 mixes signal x(t) ⁇ 1Q with ⁇ 2Q to yield the quadrature phase component of the input signal, that is, x(t) ⁇ 1Q ⁇ 2Q. If this circuit is being used to demodulate the input signal down to baseband then baseband in-phase (BBI) and baseband quadrature (BBQ) signals are output.
  • BBI baseband in-phase
  • BBQ baseband quadrature
  • a high level of integration results in decreased IC (integrated circuit) pin counts, decreased signal power loss, decreased IC power requirements, improved SNR (signal to noise ratio), improved NF (noise factor), and decreased manufacturing costs and complexity.
  • the invention provides the basis for fully integrated, multi-standard/multifrequency communications transmitters and receivers. Increasing levels of integration have been the driving impetus towards lower cost, higher volume, higher reliability and lower power consumer electronics since the inception of the integrated circuit. This invention will enable communications devices to follow the same integration route that other consumer electronic products have benefited from.
  • advantages from the perspective of the manufacturer when incorporating the invention into a product include:
  • the invention is applicable to those semiconductor technologies used in most consumer-intended communications devices, such as silicon.
  • silicon Those skilled in the art will also understand that the integrated circuit technology might be other than silicon. It is therefore the intent of the inventors to include integrated circuit technologies other than silicon, and higher frequency applications within the scope of this invention.
  • the electrical circuits of the invention may be described by computer software code in a simulation language, or hardware development language used to fabricate integrated circuits.
  • This computer software code may be stored in a variety of formats on various electronic memory media including computer diskettes, CD-ROM, Random Access Memory (RAM) and Read Only Memory (ROM).
  • electronic signals representing such computer software code may also be transmitted via a communication network.
  • Such computer software code may also be integrated with the code of other programs, implemented as a core or subroutine by external program calls, or by other techniques known in the art.
  • DSPs digital signal processors
  • FPGAs field programmable gate arrays
  • the invention may be applied to various communication protocols and formats including: amplitude modulation (AM), frequency modulation (FM), frequency shift keying (FSK), phase shift keying (PSK), cellular telephone systems including analogue and digital systems such as code division multiple access (CDMA), time division multiple access (TDMA) and frequency division multiple access (FDMA).
  • AM amplitude modulation
  • FM frequency modulation
  • FSK frequency shift keying
  • PSK phase shift keying
  • CDMA code division multiple access
  • TDMA time division multiple access
  • FDMA frequency division multiple access
  • wired communication systems include computer communication systems such as local area networks (LANs), point to point signalling, and wide area networks (WANs) such as the Internet, using electrical or optical fibre cable systems.
  • wireless communication systems may include those for public broadcasting such as AM and FM radio, and UHF and VHF television; or those for private communication such as cellular telephones, personal paging devices, wireless local loops, monitoring of homes by utility companies, cordless telephones including the digital cordless European telecommunication (DECT) standard, mobile radio systems, GSM and AMPS cellular telephones, microwave backbone networks, interconnected appliances under the Bluetooth standard, and satellite communications.
  • DECT digital cordless European telecommunication

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Superheterodyne Receivers (AREA)
  • Radar Systems Or Details Thereof (AREA)
  • Transceivers (AREA)

Claims (13)

  1. Modulations- oder Demodulationsschaltung, umfassend;
    einen aktiven Mischer (70) zum Empfangen eines Eingangssignals und Mischen des Eingangssignals mit einem ersten aperiodischen Mischsignal, wobei der aktive Mischer (70) eine Stromquelle (80) aufweist, deren Parameter eingestellt sein bzw. werden können, um Verstärkung und Linearität des aktiven Mischers (70) zu steuern bzw. zu regeln;
    einen Hochpassfilter (72), der elektrisch mit dem Ausgang des aktiven Mischers (70) verbunden ist und ein gefiltertes gemischtes Signal erzeugt; und
    einen passiven Mischer (74), der mit dem Ausgang des Hochpassfilters (72) verbunden ist, zum Empfangen des gefilterten gemischten Signals und eines zweiten aperiodischen Mischsignals, und Mischen des gefilterten gemischten Signals mit dem zweiten aperlodischen Mischsignal.
  2. Schaltung nach Anspruch 1, wobei das erste und das zweite aperiodische Mischsignal ein Lokaloszillatorsignal emulieren.
  3. Schaltung nach Anspruch 1, wobei der aktive Mischer (70) mehrere Treiberkomponenten (162) umfasst, und zwar jede zum Empfangen eines unterschiedlichen Eingangssignals in einem Multistandardverhältnis.
  4. Schaltung nach Anspruch 1, wobei die Eckfrequenz des Hochpassfilters (72) relativ zu den Eingangs- und Mischsignalen niedrig ist, wodurch er nur Gleichstromsignalkomponenten unterdrückt.
  5. Schaltung nach Anspruch 1, wobei die Eckfrequenz des Hochpassfilters (72) relativ zu den Eingangs- und Mischsignalen niedrig ist, wodurch er nur eine Verzerrung zweiter Größenordnung unterdrückt.
  6. Schaltung nach Anspruch 1, wobei der Hochpassfilter (72) ein Widerstandsteilnetzwerk bzw. -frequenzweiche (94, 96, 98, 100) umfasst, das bzw. die kalibriert ist, um die Gleichtaktspannungsausgabe festzusetzen.
  7. Schaltung nach Anspruch 1, wobei der aktive Mischer (70) Schalter (G1, G2) enthält, die erlauben, dass seine Verstärkungs- und Rauschzahl angepasst bzw. eingestellt wird, und der passive Mischer (74) rein passiv ist, wodurch er sein l/f-Rauschen verringert.
  8. Schaltung nach Anspruch 1, wobei der aktive Mischer (70) Schalter (G1, G2) umfasst, die erlauben, dass seine Rauschzahl und Linearität angepasst bzw. eingestellt werden.
  9. Schaltung nach Anspruch 2, ferner umfassend:
    einen Synthesizer (172, 174, 176, 178) zum Erzeugen von zeitvariierenden Signalen ϕ1 und ϕ2, wobei ϕ1 * ϕ2 eine signifikante Leistung bei der Frequenz eines Lokaloszillatorsignals aufweist, das emuliert wird, und weder ϕ1 noch ϕ2 eine signifikante Leistung bei der Frequenz des Lokaloszillatorsignals aufweisen, das emuliert wird.
  10. Schaltung nach Anspruch 1, wobei der aktive Mischer (70) umfasst:
    einen RF-Verstärker (76, 78) zum Verstärken des Eingangssignals; und
    einen symmetrischen Mischer (82, 84) zum Mischen des verstärkten Eingangssignals mit dem ersten aperiodischen Mischsignal.
  11. Schaltung nach Anspruch 10, wobei der symmetrische Mischer (82, 84) ein Paar aktiver symmetrischer Mmischer (82, 84, 136, 138) enthält, die in einer Totem-Pole-Anordnung mit dem RF-Verstärker (76, 78) konfiguriert sind.
  12. Schaltung nach Anspruch 10, wobei der RF-Verstärker zwei Transistoren (76, 78) umfasst, die angeordnet sind, um ein symmetrisches RF-Eingangssignal zu verstärken.
  13. Schaltung nach Anspruch 1, wobei die Stromquelle (80) mehrere konstante Stromquellen (124) umfasst, wodurch sie eine Variation der Verstärkung des RF-Verstärkers erlaubt und ebenfalls eine Anpassung bzw. Einstellung der Verstärkung und Linearität der RF-Verstärkerschaltung während und nach der Herstellung erlaubt.
EP02754073A 2001-08-28 2002-08-28 Vorrichtung zur frequenzumsetzung von hochfrequenzsignalen unter verwendung von aperiodischen signalen und verfahren Expired - Lifetime EP1421676B1 (de)

Applications Claiming Priority (3)

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CA2356077 2001-08-28
CA002356077A CA2356077A1 (en) 2001-08-28 2001-08-28 Improved apparatus and method for down conversion
PCT/CA2002/001316 WO2003019800A2 (en) 2001-08-28 2002-08-28 Method and apparatus for conversion of radio frequency (rf) signals using aperiodic mixing signals

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EP1421676A2 EP1421676A2 (de) 2004-05-26
EP1421676B1 true EP1421676B1 (de) 2008-11-05

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US (1) US7164897B2 (de)
EP (1) EP1421676B1 (de)
AT (1) ATE413722T1 (de)
AU (1) AU2002322934A1 (de)
CA (1) CA2356077A1 (de)
DE (1) DE60229755D1 (de)
WO (1) WO2003019800A2 (de)

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AU2002322934A1 (en) 2003-03-10
WO2003019800A2 (en) 2003-03-06
ATE413722T1 (de) 2008-11-15
DE60229755D1 (de) 2008-12-18
US20040246047A1 (en) 2004-12-09
US7164897B2 (en) 2007-01-16
CA2356077A1 (en) 2003-02-28
EP1421676A2 (de) 2004-05-26
WO2003019800A3 (en) 2003-10-09

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