EP1261883A1

EP1261883A1 - Radar device and method for coding a radar device

Info

Publication number: EP1261883A1
Application number: EP01989417A
Authority: EP
Inventors: Thomas Brosche
Original assignee: Robert Bosch GmbH
Current assignee: Robert Bosch GmbH
Priority date: 2001-01-08
Filing date: 2001-12-18
Publication date: 2002-12-04
Also published as: WO2002054107A1; US20030184469A1; US6822605B2; DE10100417A1

Abstract

The invention relates to a radar device with means (12), for generating a first code, means (18), for modulating a transmission signal in a transmission branch with the first code, means (32), for delaying the first code, means (20), for modulating a signal in a receiver branch with the delayed first code and means for mixing a reference signal with a received signal, whereby several receiver channels (111, 112, 11k) are provided. Receiver channels (111, 112, 11k) comprise means (1201, 1202, 120K), for the generation of further codes (C1, C2, Ck), receiver channels (111, 112, 11k) comprise means (131, 132, 13k), for demodulation with the further codes (C1, C2, Ck) and means (15). for modulating the transmitter signal with at least one of the further codes (C1, C2, Ck) are provided. The invention further relates to a method which may be carried out to advantage with said radar device.

Description

Radar device and method for coding a radar device

The invention relates to a radar device with means for generating a first code, means for modulating a transmission signal in a transmission branch with the first code, means for delaying the first code, means for modulating a signal in a reception branch with the delayed first code and means for mixing a reference signal with a received signal. The invention further relates to a method for coding a radar device comprising the steps of: generating a first code, modulating a transmission signal in a transmission branch with the first code, delaying the first code, modulating a signal in a reception branch with the delayed first code and mixing a reference signal with a receive signal.

State of the art

There are numerous applications for radar devices in the most varied fields of technology. For example, the use of radar sensors is possible for close-range sensors in motor vehicles. Basically, radar devices emit electromagnetic waves from a transmitting antenna. If these electromagnetic waves hit an obstacle, they are reflected and received again by another or the same antenna after the reflection. The received signals are then sent to signal processing and signal evaluation.

For example, radar sensors are used in motor vehicles for measuring the distance to targets and / or the relative speed with respect to such targets outside the motor vehicle. For example, vehicles that are driving in front or parked can be considered as targets.

Figure 1 shows a schematic representation of a radar device with a correlation receiver of the prior art. A transmitter 300 is caused by a pulse generation 302 to emit a transmission signal 306 via an antenna 304. The transmission signal 306 strikes a target object 308, where it is reflected. The receive signal 310 is received by the antenna 312. This antenna 312 can be identical to the antenna 304. After the reception of the received signal 310 by the antenna 312, it is transmitted to the receiver 314 and subsequently fed to a signal evaluation 318 via a unit 316 with low pass and analog / digital conversion. The special feature of the correlation receiver is that the receiver 314 receives a reference signal 320 from the pulse generation 302. The received signals 310 received by the receiver 314 are mixed in the receiver 314 with the reference signal 320. Receiver 314 may include an in-phase / quadrature (I / Q) de odulator. By correcting lation can be concluded, for example, on the distance of a target object on the basis of the time delay from transmission to reception of the radar pulses.

In principle, it is desirable to separate interference signals, which originate, for example, from other transmitting antennas, from signal components reflected at the targets. Interference is generated, for example, by other radar sensors, transmitters, consumers — on the vehicle electrical system, cell phones or by noise. Methods are already known which use an additional modulation of signals in order to separate interference signals from signal components reflected at targets. It has also already been proposed to use pseudo-noise coding (PN coding) to suppress interference signals. Coding is intended to minimize such interference, with the signal-to-noise ratio ("signal / noise" S / N) in the output signal of the radar device in particular being increased. Such an increase in the S / N ratio makes it possible either to recognize targets with a smaller reflection cross section or to reduce the pulse peak power at constant S / N. The advantages of recognizing targets with a smaller reflective cross-section are, for example, that a motor vehicle not only recognizes a motor vehicle driving in front, but is also more likely to recognize pedestrians or cyclists. The reduction in the pulse peak power has the consequence that less interference is caused by other systems, for example by radio relay systems; relieved in this context the reduction of the peak pulse power the approval of the sensors by the responsible regulatory authorities.

Furthermore, efforts are made to receive and evaluate the transmission signals of the other sensors when using several radar sensors. Therefore one wants to be able to distinguish between the signals of different radar sensors.

Advantages of the invention

The invention is based on the generic radar device in that a plurality of reception channels are provided, the reception channels have means for generating further codes, the reception channels have means for demodulation with the respective further codes, and that means for modulating the transmission signal with at least one of the further codes Codes are provided. In this way it is possible to distinguish between the signals of several radar sensors. Accordingly, the interference signal suppression is improved or the S / N ratio is increased by modulating the signals with a decoupling of different radar sensors by using different codes. In this way, the detection of false targets can be suppressed and the target geometry can be determined more precisely.

The modulation of one of the signals with the first code is preferably carried out by an amplitude modulation (ASK; "Amplitude Shift Keying"), and the modulation of the other Signals are carried out with the first code by phase modulation (PSK; "Phase Shift Keying"). It is therefore possible to combine an amplitude modulation with a phase modulation, so that different types of modulation can be used within the scope of the present invention. It is also possible to use frequency modulation (FSK; "Frequency Shift Keying").

The invention is particularly advantageous in that the modulation of the transmission signal with the first code is carried out by phase modulation (PSK) and that the modulation of the signal with the first code in the reception branch is carried out by amplitude modulation (ASK) or by frequency modulation (FSK). If other types of modulation than phase modulation (PSK) are used in the reception branch, phase modulation (PSK) is used in the transmission branch within the scope of the present invention.

However, it can also be advantageous for the transmission signal to be modulated with the first code by means of amplitude modulation (ASK), frequency modulation (FSK) or phase modulation (PSK) and for the signal to be modulated with the first code in the reception branch by phase modulation (PSK) he follows. If there is phase modulation (PSK) in the receiving branch, a wide variety of modulation types can be used in the transmitting branch.

The radar device is particularly advantageously further developed in that one of the combinations of modulation types mentioned is used for the further codes independently of the modulation types used for the first code. A low-pass filter is preferably provided for filtering the signals before demodulation. This makes it possible to use a low clock frequency for the further coding. This has the particular advantage that the coding in the reception channels does not have to be delayed. It is possible to implement a very large number of channels with only a small additional expenditure on components, these components being clocked at a relatively low frequency. An additional modulation only has to be provided on the radio frequency level, if necessary by an additional modulator. The realization of the reception channels on the low-frequency level also has the advantage that there is no deterioration in the S / N ratio.

It is particularly advantageous if the code is a pseudo-noise code (PN code). The use of PN codes for interference signal suppression has been described extensively in the literature, so that the invention can be implemented particularly well using PN codes.

The further codes and the modulation are preferably generated at a clock frequency which is an integral part of the pulse repetition frequency for generating the first code. In this way, the code generations and the demodulations with regard to the different codes are coordinated with one another.

It is preferred that k reception channels are provided, k means are provided for generating k further codes and that each of the k further codes is used for each incoming whose k-1 other codes are orthogonal. The orthogonality of the codes makes it possible, in the case of overlapping detection areas of the respective sensors, to evaluate only the corresponding sensors in a respective receiving channel. Furthermore, orthogonal codes can be produced in a simple manner in terms of circuitry.

A counter and a plurality of EXOR gates are preferably provided for generating the orthogonal codes. In this way, ideal decouplings of the respective radar sensors can be generated, for example by cyclical inversion.

In a particularly preferred case, a toggle flip-flop (TFF) and an EXOR gate are provided to generate the orthogonal codes. A TFF can be used to generate two orthogonal codes in a particularly simple manner.

In this context, it is particularly advantageous if an uncoded reception channel is additionally provided when using PSK in the reception branch. Using a TFF and an EXOR gate, it is therefore possible to implement three receive channels due to the additional uncoded receive channel.

Digital means for controlling the delay are preferably provided. Such digital means, for example a microcontroller or a digital signal processor, are capable of both the pulse repetition frequency and the PN code in a suitable manner delay, so that the signals in the receiving branch experience the required correlation.

However, it can also be advantageous if circuit means are provided for controlling the delay. In addition to controlling the delay with digital means, it is also possible to use hardware to implement the delay.

It is particularly advantageous if means for blanking out phase transitions are provided. Since the phase change is not instantaneous in the real setup, errors occur after the signal is integrated. However, if the phase-modulated signal is blanked out during the transition between the different phase positions, these errors can be minimized.

The invention is based on the generic method in that a plurality of reception channels are provided, that further codes are generated in the reception channels, that signals are modulated in the reception channels with the respective further codes, and that the transmission signal is modulated with at least one of the further codes. In this way it is possible to distinguish between the signals of several radar sensors. Accordingly, the interference signal suppression is improved or the S / N ratio is increased by modulating the signals with a decoupling of different radar sensors by using different codes. In this way, the detection of false targets can be suppressed and the target geometry can be determined more precisely. It is particularly preferred if one of the signals with the first code is modulated by an amplitude modulation (ASK; "Amplitude Shift Keying") and if the other signal is modulated with the first code by a phase modulation (PSK; "Phase Shift Keying "). It is therefore possible to combine an amplitude modulation with a phase modulation, so that different types of modulation can be used in the context of the present invention. It is also possible to use frequency modulation (FSK; "Frequency Shift Keying").

The transmission signal with the first code is preferably modulated by phase modulation (PSK), and the signal with the first code in the receiving branch is modulated by amplitude modulation (ASK) or frequency modulation (FSK; "Frequency Shift Keying"). If other types of modulation than phase modulation (PSK) are used in the receiving branch, a phase modulation (PSK) is used within the scope of the present invention in the transmitting branch.

It is particularly preferred if the modulation of the sensor signal with the first code by means of amplitude modulation

(ASK), frequency modulation (FSK) or phase modulation

(PSK) and when the modulation of the signal with the first code in the receiving branch by phase modulation

(PSK) takes place. So if there is phase modulation (PSK) in the receiving branch, there are a wide variety in the transmitting branch

Modulation types can be used. The method according to the invention is particularly advantageously further developed in that one of the combinations of modulation types mentioned is used for the further codes independently of the modulation types used for the first code.

The signals are preferably filtered in a low pass before demodulation. This makes it possible to use a low clock frequency for the further encodings. This has the particular advantage that the coding in the reception channels does not have to be delayed. It is possible to implement a large number of channels with only a small additional expenditure on components, these components being clocked at a relatively low frequency. An additional modulation only has to be provided on the radio frequency level, if necessary by an additional modulator. The realization of the reception channels on the low frequency level also has the advantage that there is no deterioration in the S / N ratio.

The codes are preferably pseudo-noise codes (PN codes). The use of PN codes for interference signal suppression has been described extensively in the literature, so that the invention can be implemented particularly well using PN codes.

The invention is particularly advantageous in that the generation of the further codes and the demodulation are carried out at a clock frequency which is an integral part of the pulse repetition frequency for generating the first PN code. In this way, the code generations and the Demodulations related to the different codes matched to each other.

It is particularly preferable that k reception channels are provided, k means are provided for generating k further codes and that each of the k further codes is orthogonal to each other of the k-1 further codes. The orthogonality of the codes makes it possible, in the case of overlapping detection areas of the respective sensors, to evaluate only the corresponding sensors in a respective reception channel. Furthermore, orthogonal codes can be produced in a simple manner in terms of circuitry.

The orthogonal codes are preferably generated by a counter and a plurality of EXOR gates. In this way, ideal decouplings of the respective radar sensors can be generated, for example by cyclical inversion.

In a particularly advantageous embodiment, the orthogonal codes are generated by a toggle flip-flop (TFF) and an EXOR gate. A TFF can be used to generate two orthogonal codes in a particularly simple manner.

When using PSK in the reception branch, an uncoded reception channel is preferably additionally provided. Using a TFF and an EXOR gate, it is therefore possible to implement three receive channels due to the additional uncoded receive channel. It is useful if the delay is controlled by digital means. Such digital means, for example a microcontroller or a digital signal processor, are able to delay both the pulse repetition frequency and the PN code in a suitable manner, so that the signals in the reception branch experience the required correlation.

However, it can also be advantageous if the delay is effected by switching means. In addition to controlling the delay with digital means, it is also possible to use hardware to implement the delay.

It is also advantageous if phase transitions are blanked out. Since the change of the phase position does not take place instantaneously in the real setup, errors occur after the signal is integrated. However, if the phase-modulated signal is blanked out during the transition period between the different phase positions, these errors can be minimized.

The invention is based on the surprising finding that it is possible to separate a plurality of reception channels by simple means. There is a separation between the coding of the radar sensor and the additional coding for the realization of several reception channels. The additional coding has no further tasks. This makes it possible to make them very simple and to use a relatively low clock frequency. The delay of the reference signal necessary for the function of the radar is specified in the first code noun. The additional coding does not have to be delayed due to the low clock frequency used for this in the reception channels. Even when implementing a large number of channels, only a small additional outlay is required for components which are clocked at a relatively low frequency. Only additional modulation is required at the RF level. It is also advantageous that the codes used, that is to say the first code and the further codes for separating the reception channels, can each be selected independently of one another in accordance with the respective requirements. The digital circuits for code generation and code shift as well as the switches and mixers can be easily integrated, for example in a "monolithic microwave integrated circuit" (MMIC).

drawings

The invention will now be explained by way of example with reference to the accompanying drawings using preferred embodiments.

It shows:

Figure 1 is a block diagram of a radar device of the prior art;

Figure 2 excerpts from PN codes;

FIG. 3 shows an autocorrelation function of a PN code over different value ranges; FIG. 4 shows sections of a PN code and of signals modulated with the PN code;

FIG. 5 autocorrelation functions of signals modulated in different ways;

Figure 6 is a schematic representation of an embodiment of a radar device;

FIG. 7 shows a block diagram of a first embodiment of a radar device according to the invention;

FIG. 8 shows a block diagram of a second embodiment of a radar device according to the invention;

FIG. 9 shows a circuit principle for the generation of orthogonal codes;

FIG. 10 shows a further circuit principle for the generation of orthogonal codes; and

Figure 11 is a block diagram of a third embodiment of a radar device according to the invention.

Description of the embodiments

Figure 2 shows sections of PN codes. In the upper part of Figure 2, a PN code is shown depending on the parameter v. The lower part of FIG. 2 shows the same PN code with a shift by v = 2. Derar- term PN codes and their shift are used in the context of the present invention to improve the S / N ratio and the ratio of useful signal to Doppler leak signal. Basically, such an improvement can be achieved by increasing the pulse repetition frequency f _PW . However, the maximum pulse repetition rate is limited by the range of the radar:

ⁱ -PW, max

2Rmax

with f _PW , ma: maximum pulse repetition frequency c: speed of light

R _max : range of the radar.

Targets with distances beyond R _max are not recognized. If the pulse repetition frequency is increased, the measurement for target distances between C / (2f _PW ) and R _{max is} no longer clear. In the context of the present invention, however, it is possible to achieve the S / N ratio by increasing the pulse repetition frequency, since the use of PN coding can increase the pulse repetition frequency without endangering the uniqueness of the distance measurement. The reason for the improvement of the S / N ratio by increasing the pulse repetition frequency is that if the transfer function of the low-pass filter used remains unchanged, it is integrated in the received signal over a larger number of pulses. If the pulse repetition frequency is increased by a factor m, for example, the coherent integration results in an S / N ratio increased by m. Accordingly, if the S / N ratio is _n before the pulse repetition frequency (Ξ / N) is increased, it is increase in pulse repetition frequency (S / N) _nm . However, if the original S / N ratio is sufficient for the integration of n pulses (S / N) _n , the peak pulse power P can also be reduced while maintaining the S / N ratio at an increased pulse repetition frequency f _PW , P _{t being} proportional to Reciprocal of the pulse repetition rate is. In the upper part of FIG. 2, a section of an 8-bit PN code selected as an example is shown. If the code used has a suitable auto-correlation function (AKF), the range of uniqueness for the distance measurement can be increased by coding the signal.

FIG. 3 explains the relationships in relation to the auto-correlation functions in more detail. The upper part of FIG. 3 shows the autocorrelation function of an 8-bit PN code over the value range from v = 1 to v = 255. Basically, the autocorrelation function of the PN code represents the result after a multiplication of the PN code with itself and subsequent summation depending on the shift of the code by v cycles. In the lower part of FIG. 3 is a section of the autocorrelation function of an 8 selected as an example -Bit PN code shown over shifts from 0 to 6 cycles. If a relative shift of 0 seconds is now set between the received and the reference signal, the maximum value of the autocorrelation function is obtained for an 8-bit PN code, which is 2 ⁸ -l = 255. If one increases the value of the shift, the value of the autocorrelation function decreases. From a shift of v> 1, a value of -1 is obtained for the example shown in FIG. 3. This German The pronounced maximum of the autocorrelation function of a selected PN code at v = 0 enables an exact measurement of the time delay of the received signal and thus an unambiguous determination of the target distance.

With a step size of the shift of Δv <0.5, a sufficient resolution of the autocorrelation function is obtained for an interpolation of the maximum. Due to the periodicity of the autocorrelation function, the uniqueness range of this code extends from v = 0 to v _ma χ = N - 1. Depending on the clock frequency with which the individual chips are modulated onto the radar signal, the "peak" of the autocorrelation function can be spatially determined Allocate extent. For a code with a frame length (period) of N chips and a chip clock frequency or pulse repetition frequency f _PW , there is a spatial uniqueness range, which ranges from 0 to Rein, whereby

applies.

The condition is on the area of uniqueness

Rein - Rmax

to deliver. Otherwise there would be an ambiguous measured value of for the target distances, which lie between R _a and R _max r = r -nRein

with n = 0, 1, and r> 0

in front.

A maximum code _{shift of} would then be for monitoring the range from 0 to R _ιns t instrumented for the radar sensor

_ ^R inst ^f PW ^v inst -

Clocking necessary.

4 shows basic types of modulation of a carrier frequency. The transmit signal of the radar sensor is generated by modulating the carrier frequency f ₀ generated with a local oscillator (LO) with the corresponding PN code. In principle, various types of modulation are available, for example PSK, QPSK, ASK, FSK and MSK. The present invention relates primarily to the types of modulation ASK and PSK and to a PSK modulation with blanking of phase transitions. In the upper part of FIG. 4, a section of an 8-bit PN code is shown. The middle part shows a signal ASK amplitude-modulated with the PN code. The lower part shows a signal PSK _A phase-modulated with the PN code, the phase transitions being blanked out. In the real construction of the sensor, transition times occur between the on and off state of the signal in the case of ASK and PSK modulation. These are for the transition between in the modeling of the example shown in FIG. 4 with a duration of, for example, 100 ps 0 ° and 180 ° have been taken into account. With PSK, the carrier frequency must be an integer multiple of the pulse repetition frequency or the chip clock frequency. In order to achieve optimal interference signal suppression, it is necessary to obtain the largest possible value after integrating the autocorrelation function over one or more frames at v = 0. With code shifts between v = 1 up to the maximum code shift i _{ns used} , the lowest possible values should occur. In this context, PSK modulation of the carrier frequency fo is expedient. The phase position of f ₀ is shifted with a PSK modulation between 0 ° and 180 °.

Since the phase change is not instantaneous in the real setup, errors occur after the signal is integrated. This reduces the ratio δ between the amplitude at v = 0 and the maximum amplitude at v> 1. If the PSK-modulated signal is blanked out during the transition time between different phase positions, as shown in FIG. 4 below, this error can be minimized.

5 shows examples of autocorrelation functions for different scenarios. The upper part of FIG. 5 shows the autocorrelation function of the ASK-modulated signal. The middle part shows an autocorrelation function of the phase-modulated signal PSK with non-ideal phase transitions, the phase transitions not being blanked out. The lower part shows a phase modulation of the signal PSK _A with blanking of the phase transitions. In the example shown, an ASK modulation results in δ ~ 2. With PSK modulation without blanking out the phase transitions, δ «10 results and with PSK _A modulation with blanking out of the phase transitions, δ« 255 results achieves a much better detection of the maximum or more effective interference suppression.

By combining ASK and PSK, whereby ASK is used in the transmission branch and PSK in the reception branch or vice versa, δ goes theoretically against infinity. This means that the autocof relation function is zero for v> 1. With the same chip clock frequency f _{PW, the} improvement in the S / N ratio is about 3 dB less than with a pure phase modulation PSK. With an amplitude modulation ASK in the transmission branch and with a phase modulation PSK in the reception branch, the average transmission power drops by the same factor. By blanking out the phase transitions, the width of the maximum of the autocorrelation function is reduced, or the bandwidth of the amplitude spectrum is increased. This improves the ability to separate between different targets at the same clock frequency f _PW , although a smaller step size is required for the code shift Δv. Alternatively, the pulse width τ can be increased while the spatial resolution, separability, step size and bandwidth remain approximately the same. FIG. 6 shows a schematic illustration of an embodiment of a radar device, on which basic principles can be explained. The schematic representation is greatly simplified. In particular, only the in-phase (I) channel is shown and the quadrature (Q) channel has been omitted, although in principle this can be constructed identically. The radar device comprises a clock generator 10 for generating a pulse repetition frequency PRF. The pulse repetition frequency is fed to a PN generator 12. A local oscillator 14 (LO) is also provided, which generates a carrier frequency f ₀ of, for example, 24 GHz. The carrier frequency is fed from the local oscillator 14 to a 3 dB power divider 16. The power divider supplies a transmission branch via a first phase modulator 18. The phase modulator 18 is shown here schematically as a switch. For example, it can be implemented as a mixer. Furthermore, the power divider 16 supplies a receiving branch via a second phase modulator 20, which can also be implemented as a mixer. The outputs of the phase modulators 18, 20 are each connected to switches 22, 24 for blanking out the phase transitions. The output of the switch 22 in the transmission branch supplies the transmission signal. The output of the switch 24 in the receiving branch is connected to a mixer 26. In addition to the output signal of the switch 24, the mixer receives the received signal. The output of the mixer 26 is connected to a low-pass filter 28, which supplies the in-phase (I) signal as an output signal. Furthermore, a microcontroller or a digital signal processor 30 is provided, which controls a delay 32. This delay serves to delay both the pulse repetition rate as well as the PN code in the receiving branch. While in the transmitting branch the pulse repetition frequency is given directly to a pulse shaping 34, the output signal of which switches the switch for blanking 22 of the phase transitions in the transmitting branch, the pulse repeating frequency in the receiving branch is given a pulse shaping 36 via the delay 32, which switches 24 for blanking the Phase transitions in the receiving branch switches. Furthermore, the PN code is used directly for phase modulation in the transmission branch by switching the switch 18 for phase modulation. In the receiving branch, the PN code is passed to the switch 20 for phase modulation with a delay. The reception branch ends in a reception antenna 38; the transmission branch ends in a transmission antenna 48.

The radar device according to FIG. 6 operates as follows. The local oscillator 14 generates a carrier frequency which is fed to the transmission branch via the power divider 16. Part of the power of the carrier frequency is fed to the receiving branch to generate a reference signal. In the transmission branch, the carrier frequency is phase modulated by the phase modulator 18. The phase modulation is carried out by the PN code, which is generated by the PN generator 12. The phase-modulated signal is fed from the phase modulator 18 to a switch 22 for blanking out the phase transitions. This switch is actuated by an output signal of pulse shaping 34, which generates a time window for blanking as a function of pulse repetition frequency 10. The pulse repetition frequency 10 is thus for the instantaneous establishment of a time window for the pulse shaping 34 and for the instantaneous provision of the PN code for the phase modulation 18 responsible. In the receiving branch, the output signal of the power divider 16 is also phase-modulated in the phase modulator 20. The output signal of phase modulator 20 is supplied to switch 24 for blanking, switch 24 being actuated by an output signal from pulse shaping 36. This pulse shaping 36 is controlled with a delay by the pulse repetition frequency. The phase modulator 20 is also switched by a delayed PN code.

In principle, the order of the phase switches 18, 20 and the respective switches for blanking 22, 24 is arbitrary. The blanking and / or phase switching can also be between the receiving antenna 38 and the mixer 26. The blanking can still be between mixer 26 and low pass 28.

FIG. 7 shows a block diagram of a first embodiment of a radar device according to the invention. A transmission unit T with means 18 for modulating a transmission signal is equipped with a transmission antenna 48. A PN code is supplied to the means 18 for modulating by a PN coder 12. This PN code is also transmitted to means 32 for delaying the code. The delayed code is fed to a receiver unit R with means 20 for modulating a signal in a receiving branch. The delayed code thus serves as a reference signal S _R. The receiver is equipped with a receiving antenna 38. The transmitter unit T is transmitted by a local oscillator 14 via means 15 for modulating the transmit signal, the carrier frequency f ₀ . This carrier frequency f ₀ is also input to the receiver unit R ben where this is mixed with the received signal. The output signal of the receiver unit is filtered and amplified by a low pass and amplifier .17. The signal is then fed to means 19 for forming the amount, the amount being formed here from the I and the Q signal: | s _± + js _Q |. This I / Q amount is supplied to means 13ι, 13 ₂ , ... 13] c, which each have a reception channel lli, 11 ₂ . ■ • • Hk are assigned. Each of these modulation means 13ι, 13 ₂ , ... 13 _k works with a specific code Ci, C ₂ , .... C _k , these being generated by the respective means 120ι, 120 ₂ , ... 120 _k of the codes are delivered. One of these codes, in the present case the code Ci, is used to modulate the local oscillator signal in the means 15 for modulating the transmission signal.

In the present general case according to FIG. 7, means 18 for modulating a transmission signal, the means 20 for modulating a signal in a receiving branch, the means 13ι, 13 ₂ , ... 13 _k for demodulating and the means 15 for additionally modulating the transmission signal Perform amplitude modulation ASK, frequency modulation FSK or phase modulation PSK. The whole system is controlled by a controller 21.

The code generation and demodulation in the receive channels III, H2 _/ • • • Hk preferably take place with a relatively low clock frequency, preferably with an integral part fp _{N /} m (m = 1, 2, ...) of the repetition frequency of a frame (period ) of the PN code f _PN . The additional coding and demodulation is based on the low frequency ne 29 possible, so that only a relatively small amount of hardware is required for this reason.

If the detection ranges of the respective sensors now overlap and if the transmission signal of each of these sensors is modulated with another suitable code Ci, C ₂ , ... C _k , then the transmission signal of the own sensor reflected at the targets and that reflected can be used in each sensor Transmission signal of the respective other sensors in the respective reception channel III, H2. • • • 11k can be evaluated.

FIG. 8 shows a further embodiment of a radar device according to the invention. Components which correspond to those from FIG. 7 are identified by the same reference symbols. The special feature of the circuit according to FIG. 8 is that three orthogonal codes are generated by the encoder 120. In the present case, the modulators 15, 20 can, for example, perform PSK modulation. Demodulator 18 performs ASK modulation or PSK modulation. The demodulators 13ι, 13 ₂ , 13 ₃ work as ASK modulators.

An example of the generation of the orthogonal codes is schematically explained with reference to FIG. 9. It is carried out with the aid of a counter 23 and several EXOR gates 25. An ideal decoupling of the respective radar sensors is obtained, for example, by cyclic inversion.

The following applies to the derivation of the 1st code Ci (n):

Cι (n) = (-l) ⁿ ; n = 1, 2, ... In general, the following applies to all other orthogonal codes Cι (n):

C, (n) = (-l) ^{round, n2l "1,} C ₁ _ ₁ (n); i = 2 ... k

with: d = -1.1, -1.1, -1, ...; C ₂ = 1, -1, -1,1,1, -1, -1, ...; C ₃ = 1,1,1, -1, -1, -1, -1,1,1,1,1, ... etc.

With each derivation i, the frame length N of the PN code doubles and then amounts to a total of 2 ^X N. Since the individual code elements (chips) must be integrated over at least one frame length, the limit frequency of the low pass used for integration may be around that To decrease factor 2. With, for example, k = 3 different channels, a frame length of N * 8 chips is obtained.

A simplified circuit for generating two orthogonal codes is shown schematically in FIG. 10. Instead of a counter, a toggle flip-flop (TFF) 27 is used here, which together with an EXOR gate provides the two codes.

FIG. 11 shows a circuit in which the circuit principle according to FIG. 10 can be used. In FIG. 11, elements that correspond to those from FIG. 7 are again identified by the same reference numerals. The encoder 120 supplies two orthogonal codes Ci, C ₂ in the channels lli, Ü2-

PSK modulation is used in the reception channels 11, 11 ₂ , which simplifies the circuit. In this case you can also add an uncoded channel ll _x and another sensor can be realized without additional coding of the transmission signal. The means 18 can use PSK modulation or ASK modulation for modulation in the transmission branch. It results from the same number of reception channels in the _. Compared to coding with ASK in the receiving branch half the frame length.

The present description largely explains the invention using the example of PN codes. Instead of PN codes, other codes can also be used, e.g. Gold codes, M-sequences, Kasami sequences, wavelets etc.

The preceding description of the exemplary embodiments according to the present invention is only for illustrative purposes and not for the purpose of restricting the invention. Various changes and modifications are possible within the scope of the invention without leaving the scope of the invention and its equivalents.

Claims

Expectations

1 . Radar device with

Means (12) for generating a first code,

Means (18) for modulating a transmission signal in a transmission branch with the first code,

Means (32) for delaying the first code, means (20) for modulating a signal in a reception branch with the delayed first code and means for mixing a reference signal with a reception signal, characterized in that a plurality of reception channels (lli, H2. ■ •• 11KL are provided, that the receiving channels (lli, A _2, ■ ■ ■ ll _k) means (120ι, 120 _2, ... 120 _k) for generating additional codes (Ci, C _2, ... C _k) which that the receiving channels (lli, A-2, • ■ ■ llk) have means (13ι, 13 ₂ , - .. 13 _k ) for demodulating with the respective further codes (Ci, C ₂ , ... C _k ) and that Means (15) for modulating the transmission signal with at least one of the further codes (Ci, C ₂ , ... C _k ) are provided.

2. Radar device according to claim 1, characterized in that - the modulation of one of the signals with the first code is carried out by an amplitude modulation (ASK; "Amplitude Shift Keying") and that the other signal is modulated with the first code by phase modulation (PSK; "phase shift keying").

3. Radar device according to claim 1 or 2, characterized in that the modulation of the transmission signal with the first code by phase modulation (PSK) and that the modulation of the signal with the first code in the receiving branch by amplitude modulation (ASK) or frequency modulation (FSK; "Frequency Shift Keying").

4. Radar device according to one of the preceding claims, characterized in that the modulation of the transmission signal with the first code is carried out by amplitude modulation (ASK), frequency modulation (FSK) or phase modulation (PSK) and that the signal is modulated with the first code in the receiving branch is done by phase modulation (PSK).

5. Radar device according to one of the preceding claims, characterized in that one of the combinations of modulation types mentioned is used independently of the modulation types used for the first code for the further codes (Ci, C ₂ , ... C _k ).

6. Radar device according to one of the preceding claims, characterized in that a low pass (17) for

Filtering of the signals before demodulation is provided.

7. Radar device according to one of the preceding claims, characterized in that the codes are pseudo-noise codes (PN codes).

8. Radar device according to one of the preceding claims, characterized in that the generation of the further codes (Ci, C ₂ , ... C) and the demodulation take place at a clock frequency which is an integer part of the pulse repetition frequency for generating the first PN code is.

9. Radar device according to one of the preceding claims, characterized in that - k reception channels (lli, 11 ₂ / ... llk) are provided, that k means for generating further codes (Ci,

C ₂ , ... C _k ) are provided and that each of the k further codes (C _x , C ₂ , ... C _k ) to each other of the k-1 further codes (Ci, C ₂ , ...

C _k ) is orthogonal.

10. Radar device according to one of the preceding claims, characterized in that a counter (23) and a plurality of EXOR gates (25) are provided for generating the orthogonal codes (Ci, C ₂ , .. • C _k ).

11. Radar device according to one of the preceding claims, characterized in that a toggle flip-flop (TFF) (27) and an EXOR gate (25) are provided for generating the orthogonal codes.

12. Radar device according to one of the preceding claims, characterized in that when using PSK in the reception branch an uncoded reception channel (11; -) is additionally provided.

13. Radar device according to one of the preceding claims, characterized in that digital means (30) are provided for controlling the delay.

14. Radar device according to one of the preceding claims, characterized in that switching means are provided for controlling the delay.

15. Radar device according to one of the preceding claims, characterized in that means (22, 24, 34, 36) are provided for blanking out phase transitions.

16. Method for coding a radar device with the steps - generating a first code,

Modulating a transmission signal in a transmission branch with the first code,

Delaying the first code,

Modulating a signal in a receiving branch with the delayed first code and

Mixing a reference signal with a received signal, characterized in that several receive channels (lli, H ₂ / • • • llk) are provided, - that in the receive channels (lli, Ü _{2 /} • ■ ■ ll _k ) further codes (Ci, C ₂ , • • • Ck) are generated that the reception channels (lli, H2 / • • ■ llk) signals are modulated with the respective further codes (Ci, C ₂ , ... C _k ) and that the transmission signal with at least one of the other codes (Ci, C ₂ , ... C _k ) is modulated.

17. The method according to claim 16, characterized in that the modulation of one of the signals with the first code by amplitude modulation (ASK; "Amplitude Shift Keying") and that the modulation of the other signal with the first code by phase modulation (PSK; "Phase Shift Keying") takes place.

18. The method according to claim 16 or 17, characterized in that the modulation of the transmission signal with the first code by phase modulation (PSK) and that the modulation of the signal with the first code in the receiving branch by amplitude modulation (ASK) or frequency modulation (FSK; "Frequency Shift Keying").

19. The method according to any one of claims 16 to 18, characterized in that the modulation of the transmitted signal with the first code by amplitude modulation (ASK), frequency modulation (FSK) or phase modulation (PSK) and that the modulation of the signal with the first code in the receiving branch is done by phase modulation (PSK).

20. The method according to any one of claims 16 to 19, characterized in that one of the combinations of modulation types mentioned is used for the other codes (Ci, C ₂ , ... C _k ) regardless of the modulation types used for the first code.

21. The method according to any one of claims 16 to 20, characterized in that the signals are filtered in a low pass (17) before demodulation.

22. The method according to any one of claims 16 to 21, characterized in that the codes are pseudo-noise codes (PN codes).

23. The method according to any one of claims 16 to 22, characterized in that the generation of the further codes (Ci, C ₂ , ... C _k ) and the demodulation is carried out at a clock frequency which is an integer part of the pulse repetition frequency for generating the first Code is.

24. The method according to any one of claims 16 to 23, characterized in that k reception channels (lli, Ü2 / ■ • • llk) are provided, - that k means for generating k further codes (Ci,

C ₂ , ... C _k ) are provided and that each of the k further codes (Ci, C ₂ , ... C _k ) to each other of the k-1 further codes (Ci, C ₂ , ... C _k ) is orthogonal.

25. The method according to any one of claims 16 to 24, characterized in that the orthogonal codes (Ci, C ₂ , ... C _k ) generated by a counter (23) and several EXOR gates (25).

26. The method according to any one of claims 16 to 25, characterized in that the orthogonal codes by

Toggle flip-flop (TFF) (27) and an EXOR gate (25) generated.

27. The method according to any one of claims 16 to 26, characterized in that when using PSK in the reception branch, an uncoded reception channel (II _x ) is additionally provided.

28. The method according to any one of claims 16 to 27, characterized in that the delay is controlled by digital means (30).

29. The method according to any one of claims 16 to 28, characterized in that the delay is controlled by circuit means.

30. The method according to any one of claims 16 to 29, characterized in that phase transitions are blanked out.