EP1206061A1 - Estimation de canal pour systèmes de communication de diversité d'espace - Google Patents

Estimation de canal pour systèmes de communication de diversité d'espace Download PDF

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Publication number
EP1206061A1
EP1206061A1 EP00310085A EP00310085A EP1206061A1 EP 1206061 A1 EP1206061 A1 EP 1206061A1 EP 00310085 A EP00310085 A EP 00310085A EP 00310085 A EP00310085 A EP 00310085A EP 1206061 A1 EP1206061 A1 EP 1206061A1
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EP
European Patent Office
Prior art keywords
training
communication
code
communication system
correlation
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Granted
Application number
EP00310085A
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German (de)
English (en)
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EP1206061B1 (fr
Inventor
Didier J. R. Van Nee
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Nokia of America Corp
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Lucent Technologies Inc
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Priority to DE60042408T priority Critical patent/DE60042408D1/de
Priority to EP00310085A priority patent/EP1206061B1/fr
Priority to US10/006,900 priority patent/US7099269B2/en
Publication of EP1206061A1 publication Critical patent/EP1206061A1/fr
Priority to US11/482,316 priority patent/US20070058694A1/en
Application granted granted Critical
Publication of EP1206061B1 publication Critical patent/EP1206061B1/fr
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Expired - Lifetime legal-status Critical Current

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0224—Channel estimation using sounding signals
    • H04L25/0226—Channel estimation using sounding signals sounding signals per se
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H04L1/02—Arrangements for detecting or preventing errors in the information received by diversity reception
    • H04L1/06—Arrangements for detecting or preventing errors in the information received by diversity reception using space diversity
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0204—Channel estimation of multiple channels
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2662—Symbol synchronisation

Definitions

  • the present invention relates to a communication system comprising first communication means, second communication means and a first transmission path as well as at least one further transmission path between said first and said second communication means, in which at least the first communication means are provided with transmission means for each of said transmission paths, which are capable of sending at least part of a communication signal to the second communication means, in which at least the second communication means comprise reception means for each of said transmission paths, which are capable of receiving at least part of said communication signal, and in which at least the first communication means comprise training means for generating a training code to be sent to the reception means enabling the reception means to match a received signal to a corresponding transmitted signal.
  • Communication systems especially of a non wired type, share a common faith in that they have to cope with the available bandwidth over the transmission path used.
  • Numerous techniques have been developed through the years to utilize the available band width as efficient as possible in order to enhance the bit rate over the transmission path.
  • One of these techniques is so called space division multiplexing (SDM) by which a communication signal is fed and divided over a number of separate transmission paths in parallel.
  • SDM space division multiplexing
  • the communication means of the type described in the opening paragraph employ this technique and for that purpose are equipped with transmission and reception means for every transmission path which is used for the exchange of the communication signal.
  • a problem encountered with SDM in an non-wired environment is that the signals sent by the different transmitters are likely to interfere with each other such that each receiver not only receives a signal from the associated transmitter but also from the other transmitters.
  • r i H i .x i , where H i is the appropriate column of a n-dimensional matrix containing the constants h j , said dimension n being equal to the number of transmission paths used.
  • H i the appropriate column of a n-dimensional matrix containing the constants h j , said dimension n being equal to the number of transmission paths used.
  • a straightforward training scheme would be to use a predetermined training symbol and to send at least one such symbol by each transmitter consecutively while the other transmitter(s) are inactive. In this manner the receiver may calculate the first column of the above matrix from the training symbol sent by the first transmitter, the second column from the training symbol sent by the second transmitter and so on.
  • n training symbols may be a minimum of n training symbols to recover the constants h j .
  • This training length is a serious problem for high rate wireless packet transmission links, because of the associated overhead which reduces the net data rate. For instance, at 100 Mbps, a 1000 byte packet has a duration of 80 ⁇ s.
  • the present invention has inter alia for its object to provide a communication system as referred to in the opening paragraph in which the training time may be reduced considerably compared to the above. It is a further object of the invention to provide such a communication system with improved frame detection and frequency synchronization.
  • a communication system of the kind described in the opening paragraph is according to the invention characterized in that the training means are capable of generating a training code with at least nearly ideal cyclic auto-correlation properties such that its cyclic auto-correlation function is at least nearly zero for all cyclic shifts, in that the transmission means are capable of concurrently sending said training code in a mutually shifted manner and in that the reception means are capable of performing a cyclic auto-correlation with respect to a received training signal. Because of the cyclic auto-correlation properties of the training code applied in accordance with the invention it is achieved that the auto-correlation performed on the received training signal leaves no or hardly no by-products.
  • the output of the auto-correlation operation at the reception side will be either n.h i or zero, where h j is one of the constants to be estimated and n is a known normalization factor.
  • the auto-correlation properties need not be absolutely ideal but may deviate to a small extend from the ideal situation, leaving sufficient certainty to derive the required constants. Because the transmitters send the training code concurrently in time, the invention, in theory, requires only one training symbol's duration to recover all constants h i .
  • the auto-correlation operation requires that the training code is multiplied by all itself and by all its cyclic shifts to render the above product. This may be effected by sending not only the training code but also its cyclic shifts to the reception means and multiplying these code's with the training code generated by the reception means.
  • the communication system according to the invention is characterized in that the reception means are capable of generating the cyclic shifts of a received training code and to correlate these with said training code. In this case only the training code has to be sent and this code as well as all cyclic shifts are generated at the reception side. Not only limits this embodiment transmission time and hence training time, it also avoids any distortion or other noise of the correlation products.
  • a specific embodiment of the communication system according to the invention is characterized in that the training code comprises a concatenation of the rows of a Fourier matrix.
  • Such a concatenations are generally referred to as a Frank and Zadoff-Chu sequences and happen to have ideal auto-correlation properties. This renders these sequences extremely suitable for use in a communication system in accordance with the present invention.
  • the length of the training code is preferably equal to the number of connection paths used or a integer multiple thereof.
  • a maximal length sequence has an auto-correlation of -1 for all cyclic shifts.
  • a further embodiment of the communication system according to the invention is characterized in that, during operation, the training codes are preceded and followed by a dummy code.
  • the dummy codes are taken sufficiently long to avoid substantial overlap between the auto correlation output signals during the training stage, so tha the constants may be derived unambiguously.
  • the communication system according to the invention is characterized in that the training means comprise a pre-correction filter for processing the training code.
  • a pre-correction filter as used in this embodiment is not absolutely necessary, but without it the reception means generally will have to perform a correction. The additional complexity and possible signal noise may be saved by pre-correcting the training signals before they are being send.
  • the communication system is characterized in that the training means comprise storage means for storing one or more training codes.
  • the transmitted training signal can be pre-calculated and stored in memory to avoid the complexity of a separate pre-correction filter which would otherwise be used merely during the training phase.
  • the pre-corrected training codes are for instance stored in a lookup table easily accessible for the training means. During the training stage, these codes merely need to be read out so that no complex circuitry is needed for generation or pre-correction of the codes concerned.
  • the training means during operation, issue a number of at least substantially identical training codes and in that the receiving means comprise summation means to average the received training codes.
  • a further specific embodiment of the communication system according to the invention is characterized in that the training means, at least during operation, issue at least substantial training codes at a substantially fixed interval and in that the reception means are provided with auto correlation means for correlating a received signal with one or more signals received after a delay corresponding to said interval or an integer multiple thereof. Because of the ideal auto-correlation properties of the code, the correlation outputs consist only of the sum of n t auto-correlation points, whereas all cross-correlation products between signals coming from different transmitters are zero.
  • the correlator output values are proportional to the total received power on each reception means.
  • the phase of the correlator outputs is equal to 2B f o T c , where f o is the frequency offset between transmission means and the reception means and T c the time interval between successive training codes.
  • phase of the summed correlator outputs is a measure for the carrier frequency offset that can be fed to a frequency correction circuit.
  • the frequency offset f o is given by the output phase divided by 2B T c .
  • the communication system depicted in figure 1 comprises first communication means 10 with transmission means which are capable of sending a communication signal in a wireless communication network as well as second communication means 20 which comprise reception means which are capable of receiving said communication signal.
  • the communication means are primarily intended for data exchange and operate based on a TCP/IP or any other suitable packaged data transmission protocol.
  • the transmission means comprise multiple transmitters 11..14 which are concurrently used for the transmission of the communication signal over multiple transmission carriers, whereas also multiple reception means 31..34 are available for receiving the signals sent.
  • the different transmission means 11.14 and reception means 31..34 each comprise their own antenna, which is depicted schematically in the drawing.
  • a division of the signal over several orthogonal frequency bands is effected in order to further enhance the capacity of the system, generally known as orthogonal frequency division multiplexing (OFDM).
  • OFDM orthogonal frequency division multiplexing
  • the communication signal is distributed over the different sub-carriers and frequencies to maximize the data throughput capacity of the connection.
  • SDM used in wireless data transmission presents a complication in that the signals of the different transmission means 11.14 will inevitably be received by all reception means 31.34.
  • h 1..16 represent transmission factors or channel coefficients which take into account the spatial displacement of the different transmitters and receivers as well as different environmental influences giving rise to for instance attenuation and distortion of the signal. Because these factors are not known and may change from time to time they need to be recovered for each data package being send. This process is generally referred to as channel training and consists of the transmission of a number of known data signals which enable the reception means to recover the transmission factors h 1 .. 16 . To this end, the transmission means are equipped with training means 21..24. Although these training means are indicated separately for all transmitters they may be shared among the transmitters in order to save circuitry.
  • the training means may comprise a processor unit capable of calculating the appropriate training signals, but in this example simply consist of a look-up table which has been filled in advance with suitable training codes.
  • these training codes are chosen to have at least nearly ideal auto-correlation properties, meaning that their auto-correlation product is zero for all cyclic shifts and non-zero for itself.
  • the so called Frank and Zadoff-Chu sequence is taken as training code.
  • a Frank code is obtained by concatenating all rows or columns of a discrete Fourier matrix.
  • the 4 by 4 discrete Fourier matrix F for example, is given by:
  • the auto-correlation function of this code renders zero for all its cyclic shifts and equals 16 for the non-shifted auto-correlation.
  • a code c with a length (16) equal to an integer multiple (4) of the number of sub-carriers (4).
  • the same training code c is supplied to all transmitters 11..14, but in a cyclically shifted fashion. This means that for instance the first transmitter 11 transmits the original code c, the second transmitter 12 the code cyclically shifted over one digit, the third transmitter 13 the code c shifted over two digits, and so on.
  • a pre-correction filter is applied to the code c so that its spectrum will be the same as that of the OFDM signal. It is not absolutely necessary to add a pre-correction but, without it, the receivers need to make a correction which would make the channel estimates more noisier for sub-carriers corresponding to low code spectral values of the transmitted code.
  • the corrected training signal is precalculated and stored in the look up table of the training means 21..24 to avoid the complexity of a separate pre-correction filter which would otherwise only be used for the training phase.
  • the second communication means 20 comprise separate receivers 31..34 with associated antennas for all transmitters 11..14 of the first communication means 10. These receivers 31..34 will each receive the training codes c coming from all transmitters 11..14.
  • the receivers are each coupled to correlation means 41..44 which perform a cyclic correlation of the code c with a part of the signal received by the associated receiver 31..34 of a length equal to the code c . If the training length is more than twice the code length, then the receivers may 31..34 first sum or average an integer number of parts of the training signal with a lengths equal to that of the code c . The averaged signal is then used to perform the cyclic correlation.
  • Figure 2 shows an example of the cyclic correlation output for two transmitters.
  • the cyclic correlation output shows separate impulse responses for different transmission antennas, which are separated by a delay equal to the cyclic shift applied in the subject transmitter.
  • the code in this example is a length 64 Frank sequence that is two times over sampled, so there are 128 samples in one code length. All receivers will see a different cyclic correlation function from which they can extract the channel information.
  • the reception means comprise a filter 51..54 which isolates the impulse response of the subject transmitter by multiplying the cyclic correlation output of the correlator 41.44 by a window function which is non-zero only at the desired impulse response.
  • the window has smooth roll-off regions in order to minimize errors in the estimated channel frequency response.
  • the roll-off factor is a compromise between frequency leakage and delay spread robustness; a larger roll-off region gives less frequency leakage but it also attenuates more impulse response components whose path delays fall within the roll-off region.
  • Figure 3 shows an example of a windowed correlation function based on the correlator output shown in figure 2.
  • the windowed impulse response, taken at the output of the filter 51..54, is cyclically shifted back by a second correlator 61..64 to compensate for the cyclic shift applied at the transmitter.
  • the channel frequency response is found by calculating a Fast Fourier Transform (FFT) over the windowed impulse response, taken at the output of the second correlator 61..64.
  • the reception means comprise a FFT-filter 71..74.
  • the FFT-filter 71..74 outputs resemble the channel coefficient values of all OFDM sub-carriers for one particular transmitter-receiver pair.
  • the estimated channel frequency response for all sub-carrier is drawn as curve a in figure 4, whereas curve b represents the actual values. At the lower sub-carriers there is a minor difference with the actual channel frequency response, which is assumed to be introduced by the windowing operation of the first filter 51..54.
  • the invention may as well be used for SDM on its own, i.e with merely a single carrier. If peak power is limited, which is often the case, then single carrier SDM training might require a separate training signal per transmit antenna in stead of a common signal source.
  • the training signal has to be long enough to get sufficient signal-to-noise ratio (SNR) at the receiver. Instead of sending separate training signals over the antennas one by one, one long the training signal may then be transmitted simultaneously on all antennas.
  • SNR signal-to-noise ratio
  • the same procedure outlined above can be followed up to the cyclic correlation. The outputs of the cyclic correlation directly resemble the desired channel coefficients that are needed for a single carrier SDM receiver.
  • the training time need not be longer than the duration of the training code c .
  • a repetition of the training code is used in order to realise sufficient SNR at the receiver's end.
  • the code c is preferably extended with a small dummy code preceding and following it. The addition of said dummy codes ascertains that the impulse responses associated with the different transmitters do not overlap mutually. This extended code is then stored in the look-up table of the training means 21..24. Even with this extension, the total training time need not be much longer than the duration of a training signal with sufficient SNR, whereas the prior art systems all require a multiple thereof.
  • the training signal consists of a repetitive pattern with at least nearly ideal auto-correlation properties may also advantageously be used for frame detection and the detection of the carrier frequency offset.
  • Figure 5 shows a block diagram of a system that detects the start of a transmission and that estimates the carrier frequency offset in a communication system like that of figure 1 but with two transmitters and receivers.
  • the reception means comprise delay means 81,82 to add a delay time T c to the received signal, where T c is equal to the duration of the training code c , and conversion means 91,92 to convert the delayed signal to a conjugated replica of it.
  • This T c seconds delayed and conjugated replica is fed to correlation means 101.102. Because of the ideal auto-correlation properties of the code c , the correlation outputs consist only of the sum of n t auto-correlation points, whereas all cross-correlation products between signals coming from other transmitters are zero.
  • the correlator output values are proportional to the total received power on each receiver.
  • the phase of the correlator outputs is equal to 2B,f o .T c , where f o is the frequency offset between transmitter and receiver. If all transmitters and receivers from one SDM terminal share the same reference oscillator, only a single frequency offset has to be estimated and corrected.
  • the correlator outputs are added at 105 and fed to summation means 110 to average the total in time. A start of a frame is detected by means of a comparator 120 which detects whether the magnitude of the absolute value 115 of the output signal of the summation means 110 exceeds some threshold *.
  • the value of the threshold * is a tradeoff between the probability of a false alarm and the detection probability. If a start-of-transmission is detected, a trigger is raised at the output 130 of the circuit and at the same time the actual phase ⁇ of the summed correlator outputs is a measure for the carrier frequency offset.
  • the frequency offset f o is given by the output phase ⁇ divided by 2B.f o .T c and may be derived at the output 140 of calculation means 125 in order to be fed to a frequency correction circuit.
  • FIG. 6 shows a block diagram of an SDM synchronization system that finds the symbol timing based on knowledge of the transmitted training codes c.
  • Each of the received signals is passed through a matched filter 151,152 with an impulse response equal to the conjugated and time-reversed code c.
  • the matched filter output shows the correlation between the received signal and c.
  • the power of the correlation outputs is calculated by taking the square of the output value with suitable means 160.
  • the power outputs of all receivers are added at 165 to improve the signal-to-noise ratio (SNR) of the detector output.
  • SNR signal-to-noise ratio
  • a further SNR enhancement is obtained by adding signal components which are spaced apart by T d seconds at 170, where T d is equal to the cyclic time shift applied to the codes c in the SDM transmitters.
  • the eventual output is fed to comparator 175 which selects the largest peak. The occurrence of this peak is a trigger that can be used for symbol timing.
  • the circuit of figure 7 moreover comprises an extra FIR filter 174 that sums over multiple multipath components with a spacing of one sample interval T s .
  • the number of taps L of this last FIR filter should preferably be such that L.T s is equal to the maximum tolerable delay spread that the receiver can tolerate. This structure gives an enhanced SNR performance in the case of multipath fading channels. In addition, it improves the maximum tolerable delay spread.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Synchronisation In Digital Transmission Systems (AREA)
EP00310085A 2000-11-13 2000-11-13 Estimation de canal pour systèmes de communication en diversité d'espace Expired - Lifetime EP1206061B1 (fr)

Priority Applications (4)

Application Number Priority Date Filing Date Title
DE60042408T DE60042408D1 (de) 2000-11-13 2000-11-13 Kanalschätzung für Raumdiversitätskommunikationssystemen
EP00310085A EP1206061B1 (fr) 2000-11-13 2000-11-13 Estimation de canal pour systèmes de communication en diversité d'espace
US10/006,900 US7099269B2 (en) 2000-11-13 2001-11-13 Methods and apparatus for generating and utilizing training codes in a space division multiplexing communication system
US11/482,316 US20070058694A1 (en) 2000-11-13 2006-07-07 Methods and apparatus for generating and utilizing training codes in a space division multiplexing communication system

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Application Number Priority Date Filing Date Title
EP00310085A EP1206061B1 (fr) 2000-11-13 2000-11-13 Estimation de canal pour systèmes de communication en diversité d'espace

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EP1206061A1 true EP1206061A1 (fr) 2002-05-15
EP1206061B1 EP1206061B1 (fr) 2009-06-17

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EP1681821B1 (fr) * 2005-01-13 2018-11-14 Samsung Electronics Co., Ltd. Séquences de préambule pour un système de communication multiporteuse employant un schéma à plusieurs entrées et sorties
EP2053816A1 (fr) * 2006-01-20 2009-04-29 QUALCOMM Incorporated Procédé et appareil pour le multiplexage d'ondes pilotes dans un système de communication sans fil
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US20070058694A1 (en) 2007-03-15
DE60042408D1 (de) 2009-07-30
US7099269B2 (en) 2006-08-29
US20020118635A1 (en) 2002-08-29

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