EP0608961A1

EP0608961A1 - Detection system for detecting resonance effects of a label in a frequency-swept interrogation field by means of single sideband demodulation and method for carrying out such detection

Info

Publication number: EP0608961A1
Application number: EP94200205A
Authority: EP
Inventors: Tallienco Wieand Harm Fockens
Original assignee: Nederlandsche Apparatenfabriek NEDAP NV
Current assignee: Nederlandsche Apparatenfabriek NEDAP NV
Priority date: 1993-01-28
Filing date: 1994-01-28
Publication date: 1994-08-03
Anticipated expiration: 2014-01-28
Also published as: EP0608961B1; ES2121139T3; DE69412872D1; NL9300180A; DE69412872T2

Abstract

A detection system for detecting or identifying a label comprising at least one resonant circuit. The system comprises a transmitter unit for generating a frequency-swept electromagnetic interrogation field and a detection unit for detecting resonance effects caused by a label located in the interrogation field. According to the invention the detection unit comprises a transmitter unit which detects signals coming from just one label frequency sideband of the instantaneous frequency of the interrogation field for detecting resonance effects which occur at least substantially in one sideband of a resonant frequency of the label.

Description

This invention relates to a detection system for detecting or identifying a responder, more specifically a label, comprising at least one resonant circuit, the system comprising a transmitter unit for generating a frequency-swept electromagnetic interrogation field and a detection unit for detecting resonance effects caused by a label located in the interrogation field.
An example of such a system is disclosed in Dutch patent application NL 8202951. This system comprises a receiver unit where a carrier wave with two sidebands of the interrogation field, together with an output signal of the transmitter unit, are applied to a mixer. An output signal of the mixer comprises the two sideband components transformed to a carrier wave of zero Hertz. A disadvantage of this system is that the signal to noise ratio is often not good enough to be able to detect a resonance effect with certainty. This is partly caused by noise and interfering components present in a sideband.
To meet this disadvantage, the system according to the Dutch patent application is provided with means for detecting noise and interfering components in a frequency band which does not coincide with the frequency band in which the resonance effects to be expected are detected. This, however, entails the disadvantage that the total radio-frequency bandwidth of the receiver unit must be enlarged to make it possible for both frequency bands to be detected. As a result, the sensitivity of the receiver unit to noise and interfering components is increased, so that the disadvantages referred to are not adequately removed.
The present invention is based on the insight that the resonance effect to be received occurs at least substantially in just one sideband of the interrogation field, whereas such effects are not present in the other sideband, which is complementary (for instance mirrored) with respect to the resonance frequency. Accordingly, a detection system according to the invention is characterized in that the detection unit comprises a receiver unit which detects signals coming from just one label frequency sideband of the instantaneous frequency of the interrogation field for detecting resonance effects which occur at least substantially in one frequency sideband of a resonance frequency of the label.
Because the receiver unit for receiving the resonance effects is tuned only to one sideband of the interrogation field, herein referred to as the label frequency sideband, no noise and interfering components coming from the other sideband occur in the further processing of a received signal, since signals from this last sideband are not mixed with signals from the label frequency sideband. This yields a considerable gain in the signal to noise ratio.
A frequency sweep will typically be implemented so as to ascend and descend alternately. The position of the label frequency sideband is dependent on this. According to a particular embodiment, the system accordingly selects an upper or lower sideband of the interrogation field for the label frequency sideband, depending on the frequency sweep. If a frequency sweep is performed which, for instance, is only ascending (saw tooth), the label frequency sideband may be set at a predetermined sideband.
For a label frequency sideband which is predetermined or set by the system, it holds in particular that an average frequency of the label frequency sideband at a time of a frequency sweep corresponds with the frequency of the interrogation field at a previous time of the frequency sweep.
According to a particular embodiment, the label frequency sideband is an upper sideband of the instantaneous frequency of the interrogation field during a period in which the frequency of the interrogation field decreases per unit time and/or the label frequency sideband is a lower sideband of the instantaneous frequency of the interrogation field during a period in which the frequency of the interrogation field increases per unit time. The label frequency sideband may accordingly have been set as such beforehand or selected as such by the system.
According to a highly advanced embodiment, the receiver unit comprises means for detecting spectral components of noise and interfering signals in an interfering frequency sideband of the frequency of the interrogation field, whilst the label frequency and interfering frequency sidebands are located on opposite sides of the instantaneous frequency of the interrogation field, and signals coming from these sidebands are detected separately from each other. As discussed hereinabove, the resonance effects only occur in one sideband, herein referred to as the label frequency sideband. In the complementary sideband associated with this sideband, hereinafter referred to as interference sideband, these resonance effects do not occur, so that only the noise and interfering components, if any, are present in this sideband. These interfering components are accordingly detected separately from any resonance effect and may further be used for determining and setting, for instance, noise threshold levels in the receiver unit. For detecting the noise and interfering components, the radio-frequency bandwidth of the receiver unit need not be enlarged, in contrast with the system according to Dutch patent application 8202951. As a result, the sensitivity of the system according to the invention is not reduced when the noise and interfering signals referred to are detected.
For the sake of clarity, it is noted that throughout the present description, the term label is understood to include the broader term responder. The resonance effects of a label can be caused by, for instance, coils in combination with a capacitor. The coils can be wound air-core coils or etched coils, such as for instance the coils used in adhesive labels, or coils wound onto a ferrite core. The term label as used herein is understood to include labels resonating in a different manner, for instance labels based on mechanical resonance, in combination with the magnetostriction effect, or labels based on ferroresonance.
The present invention further relates to a method for detecting or identifying a label comprising at least one resonant circuit, in which method a frequency-swept electromagnetic interrogation field is generated and resonance effects caused by a label located in the interrogation field are detected.
In accordance with the invention, the method is characterized in that said detection is carried out within just one label frequency sideband of the instantaneous frequency of the interrogation field.
The invention will be further explained with reference to the accompanying drawings. In these drawings:

Fig. 1 shows a detection system which is known per se;
Fig. 2 shows a resonance curve and phase diagram relating to the detection system of Fig. 1;
Fig. 3 shows a vector diagram relating to the detection system of Fig. 1;
Fig. 4 shows a vector diagram relating to the detection system of Fig. 1;
Fig. 5 shows a phase and frequency diagram of a detection system according to the invention;
Fig. 6 shows a phase and frequency diagram of a detection system according to the invention;
Fig. 7 shows frequency spectra for explaining the invention;
Fig. 8 shows frequency spectra for explaining the invention;
Fig. 9 shows a possible embodiment of a detection system according to the invention;
Fig. 10 shows a first particular embodiment of a receiver circuit 8, 9 according to Fig. 9;
Fig. 11 shows a second particular embodiment of a receiver circuit 8, 9 according to Fig. 9;
Fig. 12 shows a particular embodiment of a PN 22 or PN 23 according to Fig. 11;
Fig. 13 shows a vector diagram for clarifying the operation of the circuit according to Fig. 11;
Fig. 14 shows a particular embodiment of the switch 10, label signal processor 11, interference processor 12 and resonance detector 13 according to Fig. 9; and
Fig. 15 shows a particular embodiment of a detection system according to the invention.

An example of systems which are known per se and the prior art contained therein is described in applicant's Dutch patent application NL 89000658. The basic principle thereof is shown in Fig. 1. A transmittive circuit 1 controls a transmitting coil circuit 2. This circuit comprises an antenna coil L₁ and a tuning capacitor C₁. The electric losses in the antenna coil are represented by the resistor R₁. The label, whose circuit is indicated by 3, comprises in this example an air-core coil L₂ and a capacitor C₂. Here, too, the resistor R₂ represents the electric circuit losses. The current I₁(t) through the coil L₁ generates a primary magnetic alternating field H₁(t), also referred to as the interrogation field. As a result, in coil L₂ an induction voltage is generated, which is indicated with a voltage source V₂(t). The voltage V₂(t) can be written as:

$V₂(t) = - dφ/dt (1)$

wherein φ = the magnetic flux through the label coil L₂ as a result of the magnetic alternating field H₁(t). This also means that the induced voltage V₂(t) lags 90° in phase behind the magnetic alternating field H₁(t).
The voltage V₂(t) causes a current I₂(t) to flow in the series circuit L₂, C₂, R₂. The magnitude and the phase of the current I₂(t) with regard to the voltage V₂(t) depends on the (instantaneous) frequency f_c of the interrogation signal - in this example defined as the primary alternating field H₁(t) or the current I₁(t) - and on the resonance frequency f_o of the circuit 3. The following applies:

with ${v = f}_{c} {/f}_{o} {- f}_{o} {/f}_{c} (3)$
and

Herein v is also referred to as the normalized frequency and Q as the quality factor of the circuit 3. I₂ and V₂ are the known rotary vector notations of I₂(t) and V₂(t), respectively.
The absolute magnitude of the current I₂ can be defined as:

The phase angle between the current I₂ and the voltage V₂ can be defined as:

Relation (5) gives the known resonance curve, as shown in Fig. 2a. Fig. 2b shows the phase relation between I₂ and V₂. If the frequency of the interrogating signal f_c is equal to the resonance frequency f_o, i.e. for v = 0 according to relation (3), then the phase difference between I₂ and V₂ is zero. For f_c < f_o, i.e. for v < 0, the phase difference lies between 0 and 90 degrees, so that the current I₂ leads the voltage V₂ in phase. For f_c > f_o, i.e. for v > 0, the phase difference lies between 0 and -90 degrees, so that the current I₂ lags behind the voltage V₂ in phase. The current V₂ already lags 90 degrees behind the alternating field H₁, so that the phase difference between the current I₂ and the alternating field H₁ lies between 0° and
-180°. The scale for the phase difference between the current I₂ and the alternating field H₁ is shown on the right-hand side of Fig. 2b.
Fig. 3 shows a vector diagram of vectors H₁ V₂ and I₂. Here, the direction of H₁, V₂ and I₂ corresponds, respectively, with the phase of the alternating field H₁(t), the voltage V₂(t) and the current I₂(t). The magnitude of H₁, V₂ and I₂ corresponds with the amplitude of the alternating field H₁(t), the voltage V₂(t) and the current I₂(t). In Fig. 3 the direction of vector V₂ is fixed (relative to vector H₁), but the direction of I₂ is dependent on the frequency. The direction of I₂ coincides with that of V₂ if the frequency of the interrogating signal is equal to the resonance frequency of the circuit 3. Fig. 3 further shows a circle 6. This circle 6 is the geometrical position of all possible vectors I₂ as a function of the normalized frequency v. Arrow 7 indicates the direction in which the circle 6 is traversed if the normalized frequency v is varied from low to high. The points satisfying the equation $v = -1/Q$
and $v = 1/Q$
correspond, respectively, with the frequencies for which the phase angle arg I₂ / H₁ is - 45° and -135°. The amplitude for
$v = -1/Q$
and $v = +1/Q$
equals 1/√2 times the top value, that is, has the -3 dB value. The frequency difference between the two points $v = -1/Q$
and $v = +1/Q$
is called the -3 dB bandwidth and has a magnitude defined as f_o/Q.
The current I₁ through the coil L₂ of the circuit 3 causes a secondary magnetic alternating field H₂ which is in phase with the current I₂. This secondary alternating field H₂ in turn induces an induction voltage V₄ in the receiver coil L₃. These two induction voltages V₃ and V₄ each lag 90 degrees in phase relative to their respective generatory magnetic fields H₂ and H₁, so that the phase difference between the voltages V₃ and V₄ is equal to the phase difference between the secondary alternating field H₂ and the primary alternating field H₁. Thus the phase difference between the voltages V₃ and V₄ will also be between 0 and -180 degrees. Analogously to the vector diagram of I₂ and H₁, a vector diagram can also be constructed for V₃ and V₄ (see Fig. 4).
Hereinafter it is assumed that the frequency f_c of the interrogating signal is uniformly varied from low (f_min) to high (f_max). As long as f_c << f_o, the phase angle between V₃ and V₄ will be almost equal to zero. If the frequency f_c passes the resonance frequency f_o, the phase angle between V₃ and V₄ will shift from approx. 0° to approx. -180°. For the voltage V₃, this negative phase shift means that its frequency, while passing the resonance frequency f_o, is temporarily lowered somewhat, since the frequency is the first derivative of the phase of an alternating voltage, as is known from the signal theory.
Fig. 5 shows the phase shift φ and the instantaneous frequency f_v3 of V₃ as a function of time t. At time t = t_o, f_c = f_o. The lower portion of Fig. 5 shows the frequency difference of f_v3 and f_c. During the passage of the resonance frequency f_o of the label, this difference is negative.
Likewise, for the situation where the frequency f_c decreases uniformly from f_max to f_min, it can be derived that the frequency difference between f_v3 and f_c during the passage of the resonance frequency is temporarily greater than zero.
Fig. 6 gives the phase and frequency variation for this situation. The fact is the phase of the voltage V₃ must increase by 180 degrees during the passage of the resonance frequency, which has as a consequence that the instantaneous frequency of V₃ must be temporarily higher than the driving frequency f_c of the alternating field signal H₁ (overtaking effect).
The current I₂(t), and hence the voltage V₃(t), can also be regarded as the result of a double modulation process, in which the amplitude of the current I₂(t) arises through amplitude modulation of the voltage V₂(t) in accordance with the amplitude resonance curve according to relation (5), and the phase of I₂(t) through phase modulation of the voltage V₂(t) in accordance with the phase resonance curve according to relation (6).
The following applies:

${I₂(t) = A(t) * cos(2πf}_{c} .t + ø(t)) (7)$

where:

$ø(t)=-arctg(Qv(t)) ; and$

${v(t) = f}_{c} {.(t)/f}_{o} {- f}_{o} {/f}_{c} .(t).$

Here f_c.(t) represents the varying frequency of the interrogating signal, in this case the alternating field H₁ or the current I₁.
Relation (7) is known per se from the general theory of amplitude-modulated signals. It represents a so-called single sideband signal (SSB). A single sideband signal is an amplitude-modulated signal in which either of the two frequency sidebands, as well as the carrier wave, has been suppressed.
Fig. 7a shows the frequency spectrum of an amplitude-modulated signal consisting of a carrier wave component f_c and the usual two sidebands: lower sideband and upper sideband (abbreviated as LSB (Lower Side Band) and USB (Upper Side Band)).
Fig. 7b shows the spectrum of a single sideband signal, with the carrier wave and the upper sideband having been suppressed.
It has already been derived that when the frequency of the interrogating signal H₁(t), I₁(t) swings from a mininum value f_min to a maximum value f_max, the frequency of the label signal (I₂(t), H₂(t)) at the time of the passage of the resonance frequency is temporarily lower than f_c, hence temporarily lags behind the sweep of f_c. If this is regarded as a modulation of V₂(t), then that frequency of I₂(t) is temporarily located in the frequency range of the lower sideband. Conversely, if f_c swings from maximum to minimum, the frequency of the label signal upon resonance increases slightly temporarily, also lags slightly behind the sweep of f_c, and therefore falls into the frequency range of the upper sideband.
It has thus been rendered plausible that relation (7) could indeed represent a single sideband modulated signal. The extent to which the other sideband has dropped away, however, can either be determined by means of a quantitative analysis or must appear from an empirical investigation. Such an empirical investigation has demonstrated that the label signal I₂(t) is indeed strictly a single sideband signal, in which the other sideband does not occur. Accordingly, the signal energy of the label signal is located entirely in the sideband that lags behind with regard to the frequency sweep.
This means that if the interrogating frequency f_c oscillates continuously between the minimum and the maximum frequency, so that the direction of the frequency sweep is reversed all the time, the label signal occurs alternately in the lower sideband during the ascending sweep and in the upper sideband during the descending sweep. This is shown in Fig. 8. In this connection, the sideband in which the label signal occurs is also referred to as the label sideband, whilst the other sideband is referred to as the interference sideband.
The present invention is based on the above-described physical phenomenon.
Fig. 9 shows an exemplary embodiment of a schematic diagram of a detection and/or identification system according to the invention. A transmitter circuit 1 feeds a radio-frequency signal I₁(t) sweeping in frequency f_c to a transmitter coil L₁. The transmitter coil L₁ generates a magnetic alternating field H₁(t) which is directly proportional to the signal I₁(t). At the instant when the interrogating signal H₁(t) has a frequency which is equal to the resonance frequency f_o or is equal to one of the resonance frequencies of label 3, this label 3 will produce a label signal H₂(t), which signal induces a voltage V₃(t) in receiver coil L₃. The receiver circuit 8, 9 comprises a synchronous demodulation circuit 8, which may for instance comprise one or more multiplication circuits to enable a received signal to be multiplied by a reference signal Ref, produced by the transmitter circuit 1 via line 14, for demodulating the received signal according to the principle of direct conversion so as to obtain a demodulated signal D(t). The reference signal Ref is for instance directly proportional to the signal I₁(t) and comprises the frequency f_c. The demodulated signal D(t) is then provided to a circuit 9, in which the demodulated signal D(t) is split into a signal LSB(t) coming from the lower sideband LSB and a signal USB(t) coming from the upper sideband USB. Both signals are applied to a bipolar switch 10, which is controlled by a control signal 15 from the transmitter circuit 1, in such a manner that during the ascending frequency sweep the signal LSB(t), in which the label signal H₂(t) is possibly present, is transmitted to a label signal processor 11 and the signal USB(t), in which no label signal H₂(t) can be present but which may contain noise and interference signals, is transmitted to an interference processor 12. During the descending frequency sweep the switch 10 is controlled by the control signal 15, in such a manner that the signal LSB(t) is supplied to the interference processor 12 and the signal USB(t) is supplied to the label signal processor 11. The consequence of the switching operation by means of the switch 10 is that a signal potentially comprising the label signal H₂(t) is supplied in each case to the label signal processor (11), and that a signal which, except for noise and interfering signals, cannot comprise a label signal H₂(t) is supplied in each case to the interference processor (12). Thus, the receiver circuit 8-13 has been split into two channels, which makes it possible to measure the level of noise and interfering signals independently of the presence of a label signal.
Applicant's Dutch patent 8202951 discloses a system in which likewise a received signal is split for the purpose of obtaining a label signal channel and an interference channel. The splitting operation, however, is carried out in an entirely different manner, viz. by splitting the received signal into two different frequency bands. For use in a shoplifting detection system, the received signal is split into a label band (l.f. part) 3-15 kHz, and an interference band (h.f. part) 20-50 kHz. The label signal only comprises frequency components in the range of 3-15 kHz (originally also in the 0-3 kHz range, but that part is filtered out in the receiver to enable the sweep frequency itself with its harmonics to be sufficiently suppressed as well) Noise and interference signals - in particular interference signals resulting from interference with radio signals occurring in the radio-frequency band used and signals coming from other shoplifting detection systems - comprise frequency components that occur both in the frequency range of 3-15 kHz and in the range of 20-50 kHz.
In the detection and identification equipment according to the invention, the interference band of 20-50 kHz referred to above is replaced with an interference channel having at least substantially the same frequency range as the label band, for instance 3-15 kHz. In accordance with the invention, the label signal is separated from the received signal on the basis of a sideband separation. The output signals of the label signal processor 11 and the interference processor 12 are applied to a resonance detector 13. The label signal processor, together with the resonance detector 13, forms a first signal processing channel, and the interference processor 12, together with the resonance detector, forms a second signal processing channel. The label signal processor processes the label signal in a manner known per se. For that purpose the label signal processor may for instance comprise a matched filter adapted to a resonance circuit of a label. The interference processor 12 is likewise of a known type and determines, for instance, the amplitude of spectral components of the detected noise and interfering signals. On the basis of this amplitude, a detection threshold level is determined which is supplied to the resonance detector 13. The resonance detector 13 produces an output signal, for instance only when the amplitude of the signal generated by the label signal processor exceeds the detection threshold level. The output signal of the resonance detector 13 may then be a predetermined signal ('alarm') or, for instance, the signal generated by the label signal processor 11.
Thus the following improvements over the prior art systems are achieved:

1. The sensitivity to noise and interfering signals is reduced without the received label signal being weakened. The total radio-frequency bandwidth for which the receiver is sensitive is reduced from, for instance, 2*(15 - 3 + 50 - 20) = 84 kHz to 2*(15 - 3) = 24 kHz. Thus the sensitivity to noise and interfering signals is reduced by, for example, 10log(84/24) = 5.4 dB, without the label signal being weakened.
2. The signal to noise ratio of the received label signal is improved. The label channel no longer includes any noise and interfering components coming from the sideband where no label signal is present. This also yields a gain in the signal to noise ratio of, for instance, 3 dB.

For separating the two sidebands, a number of methods are known from the radio communications technique, such as for instance the filter method, the phase or quadrature method and the third or Weaver method. The filter method is useful only in combination with a superheterodyne receiver, which, however, is not preferred for practical reasons. The Weaver method is not preferred either because the radio-frequency reference signal to be used therein also does not have the same frequency as the frequency of the interrogating signal (= f_c). Preferably, however, the phase method is used, which is moreover entirely in line with the direct conversion technique already in use.
Fig. 10 shows the block diagram of a first preferred embodiment of a receiver/ demodulator 8, 9 according to the phase method. The label signal is received by antenna coil L₃ and passed to mixers 16 and 17. Both mixers 16, 17 also receive the reference signal Ref, comprising the frequency of the carrier wave, from the transmitter circuit 1. The reference signal Ref which is supplied to the mixer 17 has been phase-shifted 90° by means of a phase shifter 18. The output signal of either of the mixers, for instance mixer 17 in Fig. 10, is also phase-shifted 90° by means of a phase shifter 19. The sideband selecting operation will be evident from the following simple derivation.
Suppose that an input signal S generated by the coil L₃ and supplied to the mixers 16, 17 comprises two frequency components, viz. a first component in the upper sideband, having frequency f_usb, and a second component in the lower sideband, having frequency f_lsb. Then S can be defined as follows:

${S = Acos(2π(f}_{c} {+ f}_{usb} {)t) + Acos(2π(f}_{c} {- f}_{lsb})t) (8)$

In this example the reference signal Ref is written as ${Ref = sin(2π(f}_{c} t)$
. The output signals I and Q of the respective mixers 16 and 17, with omission of the high-frequency sum components, can be written as:

${I = S * sin(2πf}_{c} .t)$

and

${Q = S * sin(2πf}_{c} .t + 90°)$

Therefore

and

The additional phase shift by means of phase shifter 19 in the Q channel has the following result:

An adder circuit 20 then gives as output signal:

${I + Q}_{-90°} {= 0 + Asin(2πf}_{lsb} .t) (11)$

i.e., only the lower sideband signal.
A subtracter circuit 21 then gives as output signal:

${I - Q}_{-90°} {= -Asin(2πf}_{usb} .t) + 0 (12)$

i.e. only the upper sideband signal.
This method of splitting the sideband is known per se from the art of radio communications technique. An associated disadvantage is the phase shifter 19. This circuit must meet the requirement that it can provide very accurately a phase shift of a magnitude of 90 degrees over a relatively wide frequency range of, for instance, 3-15 kHz. This requires a circuit which must meet very high quality requirements with regard to accuracy.
A sideband splitting method derived from the phase method involves the use of 'Polyphase Networks' (PN), as disclosed in "Single Sideband Modulation using Sequence Asymmetric Polyphase Networks" by M.J. Gingell, Electrical Communication, vol. 48, nos. 1 and 2, 1973, pp. 21 - 25, and in British patent specifications 1,174,709 and 1,174,710.
Fig. 11 shows a block diagram of a second preferred embodiment of the receiver/ demodulator 8, 9, comprising two PNs 22, 23. The mixers 16 and 17 are symmetrically coupled to the PNs 22, 23. Since the signals from the mixers 16, 17 are shifted 90 degrees relative to each other, the combined four outputs of the mixers will form a ring to which the phases 0°, 90°, 180° and 270° can be assigned.
In Fig. 12 this is shown symbolically in a vector diagram. The reference numerals 24, 25, 26 and 27 indicate the input terminals of the PNs 22, 23, which are connected with the outputs of the mixers 16, 17, as shown in Fig. 11. At the input of PN 22, 23, a signal received by the coil L₃ gives a displacement vector 28 as shown in the vector diagram according to Fig. 12. However, the rotary direction of the vector 28 depends on the order in which the mixers 16, 17 are connected and on the frequency of the received signal. If the frequency of the received signal is greater than the carrier wave frequency f_c, i.e. a signal in the upper sideband, then the frequency of the output signal of the mixers is called positive (see also f_usb in the relations 9 and 10). If the frequency of the received signal is smaller than f_c, then a signal in the lower sideband is involved, and then the frequency of the mixer output signals are called negative, as is also evident with regard to f_lsb in the relations 9 and 10. This means that at the input of PN 22 vector 28 rotates counterclockwise, in the direction indicated by an arrow 29, for a positive frequency, i.e. for a received signal in the upper sideband, whereas for a negative frequency, i.e. a lower sideband signal, the vector 28 rotates clockwise, in the direction indicated by an arrow 30. At the input of PN 23 the outputs of mixer 17 are changed round. As a result, vector 28 will rotate in the reverse direction with regard to PN 22.
Fig. 13 shows an example of a Sequence Asymmetric Polyphase Network. The characteristic property of a PN is that a presented four-phase signal which, for instance, is presented in such an order that the vector 28 rotates counterclockwise, for instance, does propagate from the left to the outputs on the right through the circuit and that this signal does not as such propagate when it is presented in the reverse phase sequence, i.e. when the vector 28 rotates clockwise.
Thus the PN 22, 23 discriminates between a positive frequency and a negative frequency. Thus, for instance PN 22 will only transmit signals having positive frequencies, i.e. signals resulting from the detection with mixers 16, 17 of upper sideband signals with the carrier wave. PN 23, on the other hand, as a consequence of the reversal of the connections 25 and 27 of mixer 17, will only transmit signals having negative frequencies, i.e. signals resulting from the detection of signals in the lower sideband. This means that at the output of PN 22 the demodulation product of the upper sideband signal appears, indicated by USB(t), whilst at the output of PN 23 the demodulation product of the lower sideband appears, indicated by LSB(t).
The circuit of Fig. 11 is therefore equivalent to that of Fig. 10. However, the advantage of the circuit of Fig. 11 is that the tolerances of components 22, 23 in the PN need to meet considerably less strict requirements than the tolerances of components in the phase shift network 19 of Fig. 10.
Still further sophisticated embodiments of the invention can be obtained by using a Digital Signal Processor (DSP). A DSP can take over one or more functions from the previously mentioned analog function blocks in Figs. 9, 10 and 11.
Fig. 14 shows a particular embodiment of the invention in which a DSP is used. The circuit of Fig. 14 replaces the switch 10, label signal processor 11, interference processor 12 and resonance detector 13 according to Fig. 9. In the example of Fig. 14, the lower sideband signal LSB(t) and the upper sideband signal USB(t) are applied, respectively, to an analog/digital converter ADC1 and ADC2. Switch 10, label signal processor 11, interference processor 12 and resonance detector 13 are integrated into an algorithm of the DSP. For that purpose, the synchronization signal Ref 5 of the transmitter circuit 1 is supplied to the DSP.
In addition, yet more functions can be taken over by the DSP. The adder and subtracter circuits 20 and 21, respectively, of Fig. 10 can for instance be integrated into the algorithm of the DSP. The broadband 90° phase shifter 19 of Fig. 10 can be of digital design. This phase shift is called a Hilbert transformation in the signal theory, and DSP algorithms for this purpose are known per se. Thus, the entire signal processing after the mixers 16, 17 can be carried out in a DSP, as schematically shown in Fig. 15.
As decribed hereinabove, the invention relates to shoplifting dectection systems of the so-called radio-frequency type. In them, labels are used with an air-core coil, both in a design with a coil wound from wire and in a design with a coil etched on a support material. A different type of shoplifting detection system utilizes the mechanical resonance of a plate of magnetic material, the magnetostriction effect being used for coupling to the magnetic interrogation field H₁(t). This mechanism is described, for instance, in European patent 0096182 to Identitech Co. The invention can also be used for detection or identification of labels in which resonance effects according to the principle of magnetic ferroresonance are utilized, whereby the resonance is the result of the precession effect of the electron or core spin. This identification technique is disclosed in applicant's Dutch patent application 9101941. The invention can moreover be used in adsorption as well in transmission detection systems.
By incorporating into a label such vibration elements, both electromagnetic and mechanical, that a plurality of resonance frequencies occur, a code number can be assigned to a label by detecting both the number of resonances and the precise frequencies of these resonances. This renders such a label useful for identification applications such as, for instance, person admission control, livestock management systems and for the identification of goods. In these systems, too, where resonance effects are detected by means of a frequency-swept interrogating signal, the invention is applicable and can give rise to greater recognition distances and improved recognition reliability. Nor is the invention limited to the type of frequency sweep shown in Fig. 8. It is also possible, for instance, to implement a saw-tooth or triangular frequency sweep, whereby the frequency ascends or descends monotonously from a predetermined frequency value f_x to, respectively, a value f_x + Δf or f_x - Δf, only to return very quickly to the initial value f_x. Because in that case only a sweep ascending in frequency or descending in frequency is used, the label frequency sideband will always be, respectively, a lower or upper sideband of the frequency of the interrogation field, so that the switching unit 10 in Fig. 9 can be omitted.
The invention is applicable wherever one or more resonance effects are to be detected or measured by means of a frequency-swept interrogating signal ans all of these applications are understood to fall within the concept of the invention.

Claims

1. A detection system for detecting or identifying a label comprising at least one resonant circuit, the system comprising a transmitter unit for generating a frequency-swept electromagnetic interrogation field and a detection unit for detecting resonance effects caused by a label located in the interrogation field, characterized in that the detection unit comprises a transmitter unit which detects signals coming from just one label frequency sideband of the instantaneous frequency of the interrogation field for detecting resonance effects which occur at least substantially in one sideband of a resonant frequency of the label.

2. A detection system according to claim 1, characterized in that the system, depending on the frequency sweep, selects an upper or lower sideband of the interrogation field for the label frequency sideband.

3. A detection system according to claim 1 or 2,
characterized in that an average frequency of the label frequency sideband at a time of a frequency sweep corresponds with the frequency of the interrogation field at a previous time of the frequency sweep.

4. A detection system according to any one of the preceding claims, characterized in that the label frequency sideband is an upper sideband of the instantaneous frequency of the interrogation field during a period in which the frequency of the interrogation field decreases per unit time.

5. A detection system according to any one of the preceding claims, characterized in that the label frequency sideband is a lower sideband of the instantaneous frequency of the interrogation field during a period in which the frequency of the interrogation field increases per unit time.

6. A detection system according to any one of the preceding claims, characterized in that the label frequency sideband is located on a side of the instantaneous frequency of the interrogation field that corresponds with the side of the instantaneous frequency of the interrogation field in which the frequency is located at which the label resonates at at least substantially the moment when the frequency of the interrogation field passes a resonance frequency of the label.

7. A detection system according to any one of the preceding claims, characterized in that the receiver unit comprises means for detecting spectral components of noise and interfering signals in an interfering frequency sideband of the frequency of the interrogation field, whilst the label frequency and interfering frequency sidebands are located on opposite sides of the instantaneous frequency of the interrogation field and signals coming from said sidebands are detected separately from each other.

8. A detection system according to claim 7, characterized in that the label frequency and interfering frequency sidebands are located mirror-symmetrically relative to the instantaneous frequency of the interrogation field.

9. A detection system according to any one of the preceding claims 6-8, characterized in that the detection unit further comprises an interference processor to which the signals detected in the interfering frequency sideband are applied for determining the amplitude of said spectral components and for determining, on the basis of these amplitudes, a detection threshold level with the aid of which it can be decided for signals detected in the label frequency sideband whether or not a resonance effect is present in these signals.

10. A detection system according to claim 9, characterized in that the sigals detected in the label frequency sideband are applied to a resonance detector set at the detection threshold level, which resonance detector admits these signals and/or produces a different predetermined signal when the amplitude of these signals exceeds the threshold level.

11. A detection system according to any one of claims 7-10, characterized in that the receiver unit comprises two single sideband demodulators by which simultaneously signals from the label frequency sideband and signals, separated from said signals, from the interfering frequency sideband can be received.

12. A detection system according to claim 11, characterized in that the transmitter unit successively generates an interrogation field increasing in frequency and an interrogation field decreasing in frequency and that the detection unit further comprises a first processing channel for processing signals coming from the label frequency sideband, a second processing channel for processing signals coming from the interfering frequency sideband and a switching unit via which the output signals of the two single sideband demodulators are applied to the first and second processing channels, respectively, whilst the switching unit, in the case of an interrogation field increasing in frequency, assumes a first position so that signals obtained from the label frequency sideband with a first single sideband demodulator are applied to the first processing channel and signals obtained from the interfering frequency sideband with a second single sideband demodulator are applied to the second processing channel, and the switching unit, in the case of an interrogation field decreasing in frequency, assumes a second position so that signals obtained with the second single sideband demodulator are applied to the first processing channel and signals obtained with the first single sideband demodulator are applied to the second processing channel.

13. A detection system according to any one of the preceding claims, characterized in that the receiver unit comprises a single sideband demodulator of the phase type or a single sideband demodulator utilizing a polyphase network.

14. A dectection system according to any one of the preceding claims, characterized in that the detection unit comprises a digital signal processor.

15. A detection system according to any one of the preceding claims, characterized in that the detection system is designed as a shoplifting detection system.

16. A detection system according to any one of the preceding claims, characterized in that a label comprises a plurality of separate, combined or harmonic resonance frequencies, which can be detected by the dectection system and in combination form a code on the basis of which a label or group of labels can be identified.

17. A detection system according to any one of the preceding claims, characterized in that an electromagnetic resonant circuit of a label comprises at least one LC circuit.

18. A detection system according to any one of the preceding claims, characterized in that an electromagnetic resonant circuit of a label comprises means for generating mechanical resonance whereby a mechanical movement couples with the interrogation field through the magnetostriction effect.

19. A detection system according to any one of the preceding claims, characterized in that an electromagnetic resonant circuit of a label comprises means for generating an electron spin resonance.

20. A detection system according to any one of the preceding claims, characterized in that the detection system is designed as an absorption or a transmission detection system.

21. A method for detecting or identifying a label comprising at least one resonant circuit, in which method a frequency-swept electromagnetic interrogation field is generated and in which resonance effects caused by a label located in the interrogation field are detected, characterized in that said detection is carried out within just one label frequency sideband of the instantaneous frequency of the interrogation field.

22. A method according to claim 20, characterized in that, depending on the frequency sweep, an upper or lower sideband of the interrogation field is selected for the label frequency sideband.

21. A method according to claim 20, characterized in that a medium frequency of the label frequency sideband at a time of the frequency sweep is chosen such that it corresponds with the frequency of the interrogation field at a previous time of the frequency sweep.

22. A method according to claim 21, characterized in that the detection is carried out in an upper sideband of the instantaneous frequency of the interrogation field during a period in which the frequency of the interrogation field decreases per unit time and/or that the detection is carried out in a lower sideband of the instantaneous frequency of the interrogation field during a period in which the frequency of the interrogation field increases per unit time.

23. A method according to any one of the preceding claims 21 or 22, characterized in that components of noise and interference signals in an interfering frequency sideband of the frequency of the interrogation field are detected, the label frequency and interfering frequency sidebands being chosen on opposite sides of the instantaneous frequency of the interrogation field and signals coming from these sidebands being detected separately from each other.