EP0562868A2 - Procédé et appareil pour l'accès multiple entre des récepteurs-émetteurs dans la communication sans fil utilisant un spectre élargi OFDM - Google Patents

Procédé et appareil pour l'accès multiple entre des récepteurs-émetteurs dans la communication sans fil utilisant un spectre élargi OFDM Download PDF

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Publication number
EP0562868A2
EP0562868A2 EP93302325A EP93302325A EP0562868A2 EP 0562868 A2 EP0562868 A2 EP 0562868A2 EP 93302325 A EP93302325 A EP 93302325A EP 93302325 A EP93302325 A EP 93302325A EP 0562868 A2 EP0562868 A2 EP 0562868A2
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European Patent Office
Prior art keywords
transceiver
information
signal
frequency
portable
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German (de)
English (en)
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EP0562868A3 (en
Inventor
Michael Fattouche
Hatim Zagloul
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Quarterhill Inc
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WiLAN Inc
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Priority claimed from CA002064975A external-priority patent/CA2064975C/fr
Priority claimed from US07/861,725 external-priority patent/US5282222A/en
Application filed by WiLAN Inc filed Critical WiLAN Inc
Publication of EP0562868A2 publication Critical patent/EP0562868A2/fr
Publication of EP0562868A3 publication Critical patent/EP0562868A3/xx
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2602—Signal structure
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B1/707—Spread spectrum techniques using direct sequence modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03343—Arrangements at the transmitter end
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2626—Arrangements specific to the transmitter only
    • H04L27/2627—Modulators
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2649—Demodulators
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/02—Channels characterised by the type of signal
    • H04L5/023—Multiplexing of multicarrier modulation signals, e.g. multi-user orthogonal frequency division multiple access [OFDMA]
    • H04L5/026—Multiplexing of multicarrier modulation signals, e.g. multi-user orthogonal frequency division multiple access [OFDMA] using code division
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03375—Passband transmission
    • H04L2025/03414—Multicarrier

Definitions

  • This invention relates to voice and data transmission in wireless communications, and particularly between fixed and portable transmitters and receivers.
  • PCN Personal Communication Networks
  • PCN Personal Communication Networks
  • PCN typically include a number of transceivers, each capable of transmitting and receiving information (voice, data or video) in the form of electromagnetic signals.
  • the transceivers may be fixed or portable, and may be identical or one or more of them may be more complex.
  • the system must allow the transceivers to access each other to enable the exchange of information.
  • multiple access that is, access by more than one transceiver to another transceiver, must be allowed.
  • a transceiver, or portable radio unit must be small in size.
  • the small size of the units means only small and light-weight power sources can be used. If the portable is to be used for any length of time, it must therefore consume minimal power.
  • transceivers talk to each other over a fixed bandwidth. Because of the limited availability of the RF spectrum, the system must be bandwidth efficient yet at the same time maintain high quality exchange of information at all times in one of the most hostile channels known in communication. The new multiple access technique proposed here addresses all these issues.
  • the new access technique has a low Bit Error Probability (BER) as well as a low probability of dropped and blocked calls. This is due to the fact that the access technique is robust against multipath, Doppler shifts, impulse noise and narrowband interference. It has a low cochannel interference and little or no intersymbol interference.
  • BER Bit Error Probability
  • the new access technique can offer up to 38 times the capacity of analog FM. It includes in one aspect wideband orthogonal frequency division multiplexing of the information to be exchanged, and may include slow Frequency Hopping (FH).
  • FH Frequency Hopping
  • DSP Digital Signal Processors
  • the system operates with relatively small cells. In other aspects, dynamic channel allocation and voice activation may be used to improve the capacity of the system.
  • Operation of the system in accordance with the techniques described in this disclosure may permit compliance with technical requirements for spread spectrum systems.
  • a method for allowing a number of wireless transceivers to exchange information (data, voice or video) with each other.
  • a first frame of information is multiplexed over a number of frequency bands at a first transceiver, and the information transmitted to a second transceiver.
  • the second transceiver may be a base station with capacity to exchange information with several other transceivers.
  • the information is received and processed at the second transceiver.
  • the frequency bands are selected to occupy a wideband and are preferably contiguous, with the information being differentially encoded using phase shift keying.
  • a signal may then be sent from the second transceiver to the first transceiver and de-processed at the first transceiver.
  • the first transceiver transmits again.
  • the second transceiver may exchange information with another transceiver in a time duplex fashion.
  • the processing of the signal at the second transceiver may include estimating the phase differential of the transmitted signal and pre-distorting the transmitted signal.
  • the time intervals used by the transceivers may be assigned so that a plurality of time intervals are made available to the first transceiver for each time interval made available to the second transceiver while the first transceiver is transmitting, and for a plurality of time intervals to be made available to the second transceiver for each time interval made available to the first transceiver otherwise.
  • Frequencies may also be borrowed by one base station from an adjacent base station. Thus if one base station has available a first set of frequencies, and another a second set of distinct frequencies, then a portion of the frequencies in the first set may be temporarily re-assigned to the second base station.
  • each transceiver may be made identical except for its address.
  • the basic apparatus is a transceiver which will include an encoder for encoding information, a wideband frequency division multiplexer for multiplexing the information onto wideband frequency voice channels, and a local oscillator for upconverting the multiplexed information.
  • the apparatus may include a processor for applying a Fourier transform to the multiplexed information to bring the information into the time domain for transmission.
  • the transmitter includes a vocoder 110, an interleaver 112, a modulator 114, a filter 116, local oscillator 118, power amplifier (PA) 120 and antenna 122.
  • the receiver includes an LNA 124, a local oscillator 126, a filter 128, automatic gain control (AGC) 130 with an associated passband hardlimiter not separately shown, carrier recovery 132, sampler 134, clock recovery 136, adaptive (or fixed) equalizer 138, demodulator 140, deinterleaver 142 and decoder 144.
  • AGC automatic gain control
  • interleaver 112 deinterleaver 142
  • power amplifier 120 automatic gain control 130 with passband hard-limiter
  • clock recovery 136 and carrier recovery 132 clock recovery 136 and carrier recovery 132
  • equalizer 138 equalizer
  • a local area network will typically be a System of equal transceivers.
  • the invention may also be implemented with combinations of cellular and local area networks, or to a system with a number of equal transceivers and a master or controlling transceiver. "Equal" as used here means that the transceivers have more or less the same processing equipment and processing capabilities.
  • the system described here is primarily for the exchange of voice information.
  • the present invention proposes in one embodiment a wideband modulation scheme for exchange of information between transceivers such as portables and base stations.
  • W-OFDM Wideband-Orthogonal Frequency Domain Modulation
  • B K ⁇ f
  • the frame carries the information intended for transmission under the form of multilevel differential phase shift keying (MDPSK) symbols or differential quadrature amplitude modulated (DQAM) symbols.
  • MDPSK multilevel differential phase shift keying
  • DQAM differential quadrature amplitude modulated
  • the two tail slots act as guard bands to ensure that the out-of-band signal is below a certain power level. For example, when a pulse P(f) is selected for pulse shaping and the out-of-band signal has to be ydB or less relative to the in-band signal, K2 is selected such that 20.log10
  • the bit rate is equal to K1log2M/( ⁇ t + (1 + ⁇ )/ ⁇ f) where (1 + ⁇ )/ ⁇ f is the duration of the frame and ⁇ t is the guard time required to take into account the delay of arrival and the delay spread due to multipath.
  • the bandwidth efficiency which is defined as the ratio between the bit rate and the bandwidth, is equal to: log2M/((1 + ⁇ + ⁇ t ⁇ f)(1 + 2K2/K1))
  • both K and ⁇ f are selected sufficiently large to achieve a high throughput as well as to reduce the effects on the BER of the clock error, the Doppler shift and the frequency offset between the LO in the transmitter and the one in the receiver.
  • OFDM with a K and a ⁇ f large enough to be able to achieve a specific throughput and large enough to be able to avoid using either a clock or a carrier recovery device without substantially affecting the BER is referred to here as Wideband-OFDM.
  • Orthogonal Frequency Division Multiplexing (OFDM)) is preferred in which the information, for example encoded speech, is multiplexed over a number of contiguous frequency bands.
  • OFDM Orthogonal Frequency Division Multiplexing
  • the modulation is preferably phase modulation, while the effect of phase distortion is reduced by employing differential phase modulation.
  • the modulation may be referred to as Differential OFDM (DOFDM).
  • DOFDM Differential OFDM
  • neither pilot tones nor diversity are required in DOFDM.
  • quadrature amplitude modulation might be used, but amplitude modulation makes it difficult to equalize the distorting effects of the channel on the signal.
  • a 100MHz band is divided into 4096 points, as shown in Figure 2, plus two tail slots of 195.3 KHz each.
  • the 4096 points represent N voice channels (vc).
  • D8PSK Differential eight Phase Shift Keying
  • ⁇ (n) also takes an arbitrary value. ⁇ (n) may be used as a security key and will be known only to the transmitter and receiver. Information in the form of output bits from a vocoder are mapped onto ⁇ (n). Vocoders are well known in the art and do not need to be described in detail here. The focus here is to transmit the bits with an acceptable Bit Error Rate, i.e. with a BER ⁇ 10 ⁇ 2 for voice and ⁇ 10 ⁇ 8 for data.
  • the frame duration is 52 ⁇ s
  • the frame rate is 252 frames per 13.104 ms or equivalently
  • 126 full duplex frames may be transmitted/received every 13.104 ms.
  • the reason for pre-selecting an interval of 13.104 ms is to ensure a transmission delay to allow one transceiver to communicate with other transceivers at the same time, but must not be so long that the delay becomes unacceptable to the user. Delays longer than about 40 ms are too great for voice, and it is preferable to be lower. For data, the delay may be longer and still be acceptable.
  • Table I displays the structure of a vc slot and the number N of vc for each vocoder rate. The control symbols in each vc slot are required for handoff and power control.
  • Figure 2 shows that N vc can be transmitted simultaneously. This is known as Frequency Division Multiple Access.
  • Figure 3c shows that 126 full duplex frames can be transmitted every 13.104 ms in a Time Division Multiple Access fashion (TDMA). The total number of Full Duplex voice channels (FDvc) is therefore 126xN and is shown in Table I.
  • a portable travelling outdoor at 100 km/hr moves 1.44 cm, which leaves the outdoor channel largely unchanged.
  • a portable moving at 2 m/s moves 0.1 cm again leaving the channel unchanged.
  • a predistorted signal, transmitted by the BS should reach the portable undistorted.
  • the portable transmits/receives one FDvc every 13.104 ms
  • the BS can transmit/receive up to 21 frames or equivalently up to 21xN FDvc every 13.104 ms.
  • the frames 440 labelled frame 2...frame 21 are frames that may be transmitted to other portables. This implies that while one BS is processing its data over 520 ⁇ s, six other BS can communicate to their corresponding portables in a Time Division Multiple Access (TDMA) fashion using the same frequency bands. Also, during the 13.104 ms, or such other preselected time interval that is suitable, the BS may communicate with one or more other portables.
  • TDMA Time Division Multiple Access
  • the transmitted signal is preferably frequency hopped into a new vc slot within the same (frequency domain) frame.
  • This frequency hopping is ordered by the BS which is constantly monitoring the channel frequency response. Monitoring techniques, as well as frequency hopping, are known in the art, and not described here further.
  • the probation period is terminated if speech degradation has ceased. Frequency hopping is then ordered at the end of the probation period.
  • the period of 10 cycles is long enough to indicate the portable stationarity and is short enough to allow speech interpolation between unacceptable speech frames, hence maintaining good speech quality.
  • the BS ensures that no collisions take place between hopping portables.
  • FIG. 5a corresponds to step 1 in the protocol described above.
  • Speech is provided to a vocoder 510 where the speech is digitized and coded to create bits of information.
  • the bits are provided to the modulator 512 which turns them into D8PSK symbols, with three bits per symbol.
  • the D8PSK symbols are then processed in the processor 514 which is described in more detail in Figure 6a.
  • the output from the processor is then filtered in low pass filter 516, upconverted to RF frequencies using local oscillator 518 and transmitted by antenna 520.
  • Figure 5b corresponds to steps 2 and 3.
  • the received signal at the base station is filtered in a bandpass filter 522, and down converted by mixing with the output of a local oscillator 524.
  • the average power of the downcoverted signal is monitored by a power controller 525 that adjusts the average power to the specifications required by the sampler 526.
  • the adjusted downconverted bits are then sampled in sampler 526 to produce bits of information.
  • the bits are then processed in the deprocessor 528, described in more detail in Figure 6b.
  • An estimate of the phase differential of the received signal is taken in the channel estimator 530, as described in more detail in relation to Figure 7a and 7b below, and the estimated phase differential is supplied to a decoder-demodulator 532 to correct the received bits.
  • the estimated phase differential is also supplied to a pre-distorter 534 in the transmitter.
  • a pre-distorter is used to alter the phase of the D8PSK symbols to make the channel appear Gaussian (ideal) as opposed to a fading channel.
  • the pre-distorter 534 receives a signal corresponding to the estimated phase differential of the channel.
  • the signal being transmitted is predistorted with the estimated phase differential so that the received signal at the portable with which the BS is communicating will be corrected for any phase distortion over the channel.
  • the advantage of rendering the channel Gaussian is a large saving in the power required to achieve an acceptable BER.
  • the initial power control 525 also sends a signal to the pre-distorter 534 to adjust the transmitted power to an appropriate signal level for the sampler 526 in the portable's receiver depending on the average power of the received signal. Thus if the average power is too low, the transmitted power is increased and if the average power is too high, the transmitted power is decreased.
  • the power controller 525 may also be used in frequency hopping to monitor the average power of the received frequency and determine when frequency hopping need take place.
  • Figure 5c corresponds to step 4, and shows the receiver of the portable, which is the same as the receiver in the BS except it does not include an estimator or a power controller. These are not required in the portable on the assumption that the BS will carry out the phase estimation and the power control. However, if desired, the portable may include these functions.
  • Figures 6a, 6b and 6c illustrate the function and structure of the processor and the deprocessor respectively in the transmitter and receiver.
  • Software for modelling the function of the processor in a general purpose computer is attached hereto as Appendix A, and for modelling the function of the deprocessor as Appendix B.
  • Figure 6a shows that the processor is a DSP implementation of an RC pulse shaping filter with a 20% roll-off, followed by an inverse Fourier transform.
  • the processor first inverse Fourier transforms the 4096 D8PSK modulated symbols output from the modulator.
  • the transformed symbols are then triplicated as a group so that the total number of samples is tripled, with three consecutive groups each consisting of the 4096 transformed symbols.
  • the triplication of the signal is illustrated in Figure 6c, where the symbols are shown as first delayed and added together.
  • the three groups are windowed by a Raised Cosine window with a roll-off of 0.2 centered in the middle of the three groups.
  • the processor takes D8PSK symbols in, pulse shapes them and inverse Fourier transforms them.
  • the deprocessor undoes what the processor did, i.e. it removes the pulse shaping, then Fourier transforms the received signal to obtain the original D8PSK symbols.
  • the first two blocks in Figure 6b are similar to the second two blocks in Figure 6a except for two differences. The two differences are as follows. In the first block of the deprocessor, the repeated groups of symbols are partially overlapped as shown in Figure 6c on the right hand side. In the second block, a rectangular window is used instead of the Raised Cosine. In the preferred implementation, the blocks are repeated three times but other numbers of repetition may be used.
  • Figures 6a, 6b and 6c show that the DSP blocks used in the processor are identical to the ones used in the deprocessor, except for a small change in the two transforms and a small change in the shapes of the two windows. Thus the same hardware can be used by both the processor and the deprocessor.
  • Figure 7a shows a block diagram of an example of a preferred channel estimator
  • Figure 7b is a flow chart showing the operation of the phase estimator.
  • a computing means may be a special purpose computer or a general purpose computer programmed to carry out the digital signal processing described here, as for example with the software that is attached hereto as appendix C.
  • Other methods of estimating the channel may be used that obtain an estimate of the channel group delay or phase differential of the transmitted symbols.
  • a preferred implementation is described here.
  • the estimate A'(n) is the square-root of the sum of the squares of the quadrature (Q) and inphase (I) samples output from the deprocessor.
  • the third block performs a Hilbert transform operation H[ ⁇ ln(A'(t))] on the result of the second block.
  • H[ ⁇ ln(A'(t))] is an estimate of
  • for n 2, ..., 4096, where ⁇ (n) is the phase differential of the transmitted signal ( ⁇ is the phase of the signal).
  • the Hilbert transform is preferably carried out by taking the discrete fast Fourier transform of the data record, multiplying the positive frequency spectrum of the transform by -i (square root -1), and the negative frequency spectrum of the transform by i, and taking the inverse discrete fast Fourier transform.
  • the result is a set of symbols representing an estimate of the phase differential of the received signal, as determined from its sampled amplitude envelope.
  • a different estimation may be made to estimate the phase differential.
  • a series of data frames of a number of consecutive amplitude samples (A(t)) of the electromagnetic signal are constructed. These data frames are then segmented into segments [t1,t2], where the amplitude of the electromagnetic signal is at least a predetermined number of dB less than its running mean, for example, 10dB.
  • phase differential may be calculated from ⁇ (t) ⁇ -t0/(t02 + t'2).
  • the polarity of ⁇ (n) is extracted using the last block shown in Figure 7a.
  • the estimate so calculated does not provide the sign of the differential. This may be determined by known techniques, for example by adding the phase differential to and subtracting the phase differential from the received phase (tan ⁇ 1 (Q/I)) and taking the sign to be positive if the addition results in the smaller Euclidean distance to the expected value and negative if the subtraction results in the smaller Euclidean distance to the expected value.
  • the determination of the sign need only be carried out for phase differentials greater than a predetermined threshold. This will be in the vicinity of a fade and may be accomplished by segmenting the data record into a segment in which the phase differential is larger than a selected threshold and setting the remainder of the data record to zero. This computation may be carried out with a simple discrimination circuit or equivalent computing means in the estimator.
  • the estimates A'(n) and ⁇ '(n) are used directly in the predistortion filter in Figure 5b, while the estimates ⁇ (n) and ⁇ of the unbiased channel group delay and of the bias of the channel group delay respectively are used in the demodulator.
  • the complexity of the processor-deprocessor-channel estimator is displayed in Table II. Complexity is measured in Mega Instructions Per Second (MIPS) where one instruction is defined as one complex addition, one complex multiplication and a storage of one complex number. It does not include overhead.
  • MIPS Mega Instructions Per Second
  • the complexity of the processor-deprocessor-channel estimator in the BS is computed from the complexity of the Inverse Fast Fourier Transform (IFFT)/Fast Fourier Transform (FFT)/Hilbert Transform.
  • the complexity is 4096 x 12 x 4 x 21/13.104ms for the BS.
  • For the portable it is computed from the complexity of the FFT/IFFT per vc: (32 x 5 + 64 + 128 + 256 + 512 + 1024 + 2048 + 4096)2/13.104ms for the portable with a 6.18 Kbps vocoder.
  • Such a complexity assumes that the A/D converter operates at 100 MHz with 12 bit precision.
  • the portable has smaller complexity due to the fact that the portable transmits/receives one vc in 13.104 ms and the BS transmits/receives up to 21xN vc in 13.104 ms.
  • the interleaver-deinterleaver is not required since the signal is predistorted before transmission forcing the received samples to be independent.
  • the interleaver-deinterleaver is not required as a separate entity from the vocoder due to the fact that the channel is highly frequency selective, hence the interleaving/deinterleaving can be applied implicitly in the vocoder over one vc, without a need for a separate time domain interleaver/deinterleaver. This eliminates excess speech delays associated with interleaving/deinterleaving between frames.
  • the PA is not required since the cells can have, as shown later, a radius of up to at least 250 m outdoors and 30 m indoors, if the transmitted power is up to 6 dBm. Such a power can be generated by the Local Oscillator (LO) without a need for a PA. It is important to avoid using a PA since DOFDM generates a time domain signal with non constant envelope. A power efficient class C PA cannot be used without distorting the signal. A class A PA can be used at the expense of power efficiency.
  • LO Local Oscillator
  • a clock recovery device is not required since a sampling error in the time domain is equivalent to a phase shift in the frequency domain.
  • the phase shift is a linear function of frequency. It contributes to the bias in the channel group delay.
  • Such a bias can be easily estimated and removed as mentioned previously by averaging ⁇ '(n) over n in the frequency domain.
  • Such an estimate is accurate as long as the sampling error is less than 0.2 ⁇ s or equivalently less than 20 samples (since in this case, the corresponding phase shift is less than ⁇ ), and as long as the number of points in one vc is large enough as it is here.
  • a carrier recovery device is not required since a carrier offset in the time domain is equivalent to a sampling error in the frequency domain.
  • a sampling error of up to 10% of the duration of one pulse is acceptable. This implies that a frequency offset of up to 2.414 KHz is acceptable regardless whether it is due to carrier offset as low as 1 part in a million, i.e. as low as 1 KHz per 1 GHz.
  • a carrier frequency higher than 2.414 GHz is required, one can decrease in Figure 2 the number of points per 100 MHz or one can use an RC pulse with a rolloff larger than 20%.
  • the portable transmits a frame.
  • the BS receives the frame and predistorts a frame intended for transmission accordingly, assuming that the channel is reciprocal and stationary over 520 ⁇ s. This includes controlling the transmitted power according to the received power.
  • the BS transmits the predistorted frame and simultaneously orders the portable to control its power. The order is conveyed using the control symbol in the vc slot (See table I).
  • the degree of power control may be determined using the power controller 525, and the instruction for the inclusion of a power control symbol in the vc may be sent from the power controller 525 to the pre-distorter 534.
  • One advantage of wideband modulation over narrowband modulation is that the wideband signal does not experience short term fading the same way the narrowband one does.
  • the wideband signal is mainly affected by shadowing and other long term effects which vary slowly and are easily monitored from one frame to the other as long as the same vc slot is used by the portable to transmit and receive (i.e. as long as TDD is employed).
  • each received vc is predistorted by the BS.
  • the channel can be modeled approximately as an ideal memoryless Additive White Noise Gaussian (AWGN) channel, assuming channel reciprocity and stationarity over 520 ⁇ s. From the BS point of view, since the received signal is not predistorted by the portable prior transmission, the channel estimator is used to reduce the effect of the channel group delay.
  • AWGN Additive White Noise Gaussian
  • the LO generates a 6 dBm average power, hence the signal power transmitted by the BS over one vc slot is (6 dBm - 10log10N dB) while the signal power transmitted by the portable over one vc slot is 0 dBm. Also, since the noise power over a 100 MHz band is -94 dBm, it is (-94 dBm - 10log10N dB) over one vc. A typical noise figure at the receiver is 7 dB. The penalty for not using a matched filter in the receiver is 1 dB.
  • the radius of the urban cell can be 250 m while it can be 30 m for the indoor cell.
  • Such a path loss provides the portable with a 17 dB received SNR, while it provides the BS with an (11 dB + 10log10N dB) received SNR.
  • the channel can be modeled approximately as an ideal AWGN channel, hence the 17 dB received SNR results in a 2x10 ⁇ 3 BER.
  • the channel can be pessimistically modeled as a Rayleigh fading channel from the BS point of view. The corresponding BER are displayed in Table III which shows that the achieved BER is ⁇ 4x10 ⁇ 3. A BER ⁇ 10 ⁇ 2 is acceptable for voice.
  • the number of Full Duplex voice channels (FDvc) that can be transmitted/received per frame is 136 over 100 MHz, for a 6.18 Kbps vocoder. If the bandwidth is halved to 50 MHz, the number of FDvc per frame is reduced to 68, the noise floor is reduced by 3 dB and the number of full duplex frames that a BS can transmit/receive is doubled to 42, leaving the frame duration, the number of frames per 13.104 ms and the processor/deprocessor complexity unchanged.
  • FIGs 8a, b and c a shaded area is shown around the center of the pattern, indicating 19, 38 and 126 full duplex frames that the central BS can transmit/receive respectively.
  • Tables IVa, b and c show the number of cell layers in each cell pattern reuse, the coverage area in Km2 of the pattern reuse for both the indoor and the urban environments, as well as the carrier to interference ratio (CIR) in dB, for the 100 MHz, 50 MHz and 5.96 MHz bands, respectively. In all cases, the CIR is large enough to sustain a toll quality speech.
  • CIR carrier to interference ratio
  • the control of this protocol may use any of several known techniques.
  • the commonly used technique is to have the portable monitor the channel and determine which of several base stations it is closest to. It can then order the nearest BS to communicate with it.
  • Another technique is to use a master control which receives information about the strength of the signal on the channel used by the portable and controls the BS accordingly.
  • Such techniques in themselves are known and do not form part of the invention.
  • the location of the portable can be determined with high accuracy based on the received vc at the seven adjacent BS. Locating the portable can assist in the BS hand-off.
  • a BS hand-off and a portable hand-off do not necessarily occur simultaneously, contrary to other prior art systems.
  • a new vc is not required immediately. What is required is a BS hand-off, meaning that BS Y (associated with cell Y) must initiate transmission to the portable over the same vc, while the BS X (associated with cell X) must terminate its transmission to the portable.
  • a BS hand-off occurs without the knowledge of the portable and can occur several times before a portable hand-off is required.
  • a portable hand-off is required only when the CIR is below a certain level.
  • the Mobile Telephone Switching Office (not shown) calls for a portable hand-off in accordance with known procedures. Reducing the portable hand-off rate reduces the probability of dropped calls. This is because a dropped call occurs either because the portable hand-off is not successful or because there are no available channels in cell Y.
  • the present invention allows the use of post-detection diversity at the BS, and the use of dynamic channel allocation (DCA).
  • DCA dynamic channel allocation
  • DCA is made possible by each BS having capability to transmit/receive more than the number of FDvc allocated to its cell, namely seven times the number of FDvc for a 5.86 MHz band and up to twenty-one times the number of FDvc for a 100 MHz as well as a 50 MHz band.
  • the DCA protocol simply consists of borrowing as many FDvc as needed from the adjacent cells, up to a certain limit. The limit for the case when we employ a 6.18 Kbps vocoder, a 5.86 MHz band and 18 frames per cell is obtained as follows.
  • the cell reuse pattern consists of 7 cells. Each cell is preassigned 144 FDvc. Assuming that at peak hours, 75 FDvc are used on the average and 5 FDvc are reserved at all times, then we are left with 64 idle channels which represent the limit on the number of FDvc one can borrow from the cell.
  • BS Y does not need to borrow immediately a new channel from an adjacent cell. It can use the original channel as long as the level of CIR is acceptable. If on the other hand, a portable wants to initiate a call in cell Y where all preassigned channels are used, BS Y can borrow a channel from an adjacent cell up to a limit of 64 channels per cell.
  • FCA Fixed Channel Allocation
  • a 7 cell pattern each with a preassigned 144 Fdvc can carry a total traffic of 880.81 Erlang at 0.01 Blocking Probability (BP).
  • BP Blocking Probability
  • DCA a 7 cell pattern consists of 6 cells each with 80 FDvc that can carry a total traffic of 392.17 Erlang, combined with one cell with 528 FDvc that can carry 501.74 Erlang. The total traffic is therefore 893.91 Erlang. This increase appears to be marginal (1.5%).
  • Voice activation is controlled by the BS according to techniques known in the art. At any instant during a conversation between a BS and a portable, there are four possibilities:
  • the BS controls the voice activation procedure by allocating in cases 1, 3 and 4 three slots (frames 1.1, 1.2 and 1.3) to the BS and one slot the portable (frame 1) every four slots as shown in Figure 9a. Likewise up to 21 portables may communicate with the base station in like fashion.
  • the BS on receiving a signal from the portable, allocates three slots (frames 1.1, 1.2 and 1.3) to the portable and one slot (frame 1) to the BS every four slots as shown in Figure 9b.
  • the base station may communicate with the base station in like fashion. Consequently, instead of transmitting two full duplex voice frames over four slots as in Figure 4, voice activation allows us to transmit three full duplex voice frames over four slots.
  • voice activation provides a 50% increase in the number of available FDvc at the expense of increasing DSP complexity.
  • CDMA Code Division Multiple Access
  • Table IV displays the capacity of analog FM and of the present system with a 6.18 Kbps vocoder, 5.86 MHz band, 1 frame per cell and DCA.
  • the capacity of analog FM is 6 HDvc/1.25MHz/cell while for the present system it is 150 HDvc/1.25MHz/cell.
  • the 6.25 MHz band consists of 5.86 MHz plus two tail slots.
  • the invention may also be applied to produce a 48 Mbps wireless LAN, which also satisfies the technical requirements for spread spectrum.
  • the LAN will incorporate a plurality of transceivers, all more or less equal in terms of processing complexity, and possibly with identical components except for addresses.
  • a 26 MHz band is divided into 128 points, as shown in Figure 10, plus two tail slots of 1.48 MHz each within the 26 MHz band. Adjacent points are separated by 180 KHz and each point, as with the application described above for a portable-base station, represents a D8PSK symbol.
  • the transmitter components will be the same as shown in Figure 5b, with suitable modifications as described in the following, and will include an encoder. The output bits from the encoder are mapped onto the D8PSK symbols.
  • a rectangular time domain window corresponding to a RC frequency domain pulse has a 5.55 ⁇ s duration, and includes a 25% roll-off and excess frame duration of 0.26 ⁇ s, making a total 7.2 ⁇ s duration for the frame.
  • the Time Division Duplex protocol is as illustrated in Figure 12 (assuming there are at least a pair of transceivers):
  • Each transceiver has transmitter components similar to those illustrated in Figure 5b, with suitable modifications to the internal structure to allow the use of the particular frequency band and frame duration employed.
  • the transmitter/receiver functional and structural block diagrams are shown in Figures 13a, 13b and 13c for the exchange of data.
  • Data is provided to an encoder 810 where the data is digitized and coded to create bits of information.
  • the bits are provided to the modulator 812 which turns them into D8PSK symbols, with three bits per symbol.
  • the D8PSK symbols are then processed in the processor 814 which is described in more detail in Figure 14a.
  • the output from the processor is then filtered in low pass filter 816, upconverted to RF frequencies using local oscillator 818 and transmitted by antenna 820.
  • the received signal at the base station is filtered in a bandpass filter 822, and down converted by mixing with the output of a local oscillator 824.
  • the average power of the downcoverted signal is monitored by an initial power control 825 that adjusts the average power to the specifications required by the sampler 826.
  • the adjusted downconverted signal is then sampled in sampler 826 to produce bits of information.
  • the bits are then processed in the deprocessor 828, described in more detail in Figure 14b.
  • An estimate of the phase differential is taken in the channel estimator 830, as described in more detail in relation to Figure 7 above, and the estimated phase differential is supplied to a decoder/demodulator 832 to correct the received bits.
  • the estimated phase differential is also supplied to a pre-distorter 834 in the transmitter.
  • a pre-distorter is used to alter the phase of the D8PSK symbols to make the channel appear Gaussian (ideal) as opposed to a fading channel.
  • the initial power control 825 also sends a signal to the pre-distorter 834 to adjust the transmitted power to an appropriate signal level for the sampler 826 in the first transceiver. It will be appreciated that a pre-distorter will be included in the first transceiver's transmitter but that it will not be operable, except when the first transceiver is operating as a base station.
  • Figure 13c shows the functional blocks of the receiver of the first transceiver, which is the same as the receiver in the second transceiver except it does not include an estimator.
  • the processor is illustrated in Figure 14a and 14c and the deprocessor in Figure 14b and 14c.
  • the processor first inverse Fourier transforms the 128 D8PSK symbols output from the modulator.
  • the transformed symbols are then triplicated as a group so that the total number of samples is tripled (see the left side of Figure 4c), with three consecutive groups each consisting of the 128 transformed symbols.
  • the three groups are windowed by a Raised Cosine window with a roll-off of 0.25 centered in the middle of the three groups.
  • the processor takes D8PSK symbols in, pulse shapes them and inverse Fourier transforms them.
  • the deprocessor undoes what the processor did, i.e. it removes the pulse shaping, then Fourier transforms the received signal to obtain the original D8PSK symbols.
  • the first two blocks in Figure 14b are similar to the second two blocks in Figure 14a except for two differences as follows. In the first block shown in Figure 14b, the repeated groups of symbols are partially overlapped, as shown in Figure 14c. In the second block, a rectangular window is used instead of the Raised Cosine to produce 128 output samples corresponding to the 416 input samples.
  • phase estimator is the same as that shown in Figure 7, except that there are only 128 input samples, and the same description applies.
  • the present system is designed to operate as a spread spectrum system preferably over such bands as are permitted, which at present are the 902 - 928 MHz band, 2.4 - 2.4835 GHz and 5.725 - 5.85 MHz.
  • the carrier frequency in the local oscillator shown in Figures 5a, b and c will then be 915 MHz in the case of the 902 - 928 MHz band, and the frequencies used for modulation will be centered on this carrier frequency.
  • a system may consist of one or more central controllers (comparable to the Base Stations in the exemplary cellular system described) and some slave units (comparable to the portables).
  • the slave unit executes the commands it receives from the central controller.
  • the commands may be requesting the slave unit to transmit a receive acknowledge, a status code or information that the slave has access to.
  • the command may also be to relay the command or the information to another slave unit.
  • Vocoder rate received SNR BER 18.77 Kbps 27.8 dB 4 X 10 ⁇ 3 9.15 Kbps 30.8 dB 2 X 10 ⁇ 3 6.18 Kbps 31.33 dB 2 X 10 ⁇ 3 Table III

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  • Engineering & Computer Science (AREA)
  • Signal Processing (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Power Engineering (AREA)
  • Mobile Radio Communication Systems (AREA)
EP19930302325 1992-03-27 1993-03-25 Method and apparatus for multiple access between tansceivers in wireless communication using OFDM spread spectrum. Withdrawn EP0562868A3 (en)

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Application Number Priority Date Filing Date Title
CA2064975 1992-03-27
CA002064975A CA2064975C (fr) 1992-03-27 1992-03-27 Methode et dispositif pour mettre en communication les emetteurs-recepteurs d'un systeme de communication sans fil a etalement du spectre
US861725 1992-03-31
US07/861,725 US5282222A (en) 1992-03-31 1992-03-31 Method and apparatus for multiple access between transceivers in wireless communications using OFDM spread spectrum

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EP0562868A2 true EP0562868A2 (fr) 1993-09-29
EP0562868A3 EP0562868A3 (en) 1994-07-06

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EP0613267A2 (fr) * 1993-02-08 1994-08-31 Koninklijke Philips Electronics N.V. Système MDFO avec réduction de diaphonie
DE4330665A1 (de) * 1993-09-10 1995-03-16 Thomson Brandt Gmbh Verfahren zur ]bertragung von digitalisierten Fernseh- und/oder Tonrundfunksignalen
WO1995034149A1 (fr) * 1994-06-02 1995-12-14 Amati Communications Corporation Procede et appareil de coordination de communications multi-point a point dans un systeme de transmission de donnees multitonalites
GB2296407A (en) * 1994-12-22 1996-06-26 Roke Manor Research Frequency hopped cellular mobile radio system using orthogonal frequency division multiple access
WO1996041458A1 (fr) * 1995-06-07 1996-12-19 Deutsche Thomson-Brandt Gmbh Procede et circuit destines a ameliorer la separation des porteuses lors de la transmission de signaux ofdm
EP0810753A2 (fr) * 1996-05-27 1997-12-03 Sony Corporation Système de transmission et réception à accès multiple, utilisant multiplexage de signaux à porteuses multiples
EP0899923A1 (fr) * 1997-08-29 1999-03-03 Sony International (Europe) GmbH Transmission de signaux de réglage de puissance dans un système à modulation multiporteuse
EP0915586A1 (fr) * 1997-11-07 1999-05-12 Sony International (Europe) GmbH Transmission multiporteuse, compatible au système GSM existant
EP0930744A1 (fr) * 1998-01-02 1999-07-21 NewTec CY N.V. Procédé d'accès multiple employant une transmission multiporteuse
US5933454A (en) * 1994-06-02 1999-08-03 Amati Communications Corporation Multi-carrier data transmissions system using an overhead bus for synchronizing multiple remote units
EP0938208A1 (fr) * 1998-02-22 1999-08-25 Sony International (Europe) GmbH Transmission multiporteuse, compatible au système GSM existant
US6192068B1 (en) 1996-10-03 2001-02-20 Wi-Lan Inc. Multicode spread spectrum communications system
USRE37802E1 (en) 1992-03-31 2002-07-23 Wi-Lan Inc. Multicode direct sequence spread spectrum
DE10239810A1 (de) * 2002-08-29 2004-03-11 Siemens Ag Verfahren und Sendeeinrichtung zum Übertragen von Daten in einem Mehrträgersystem
US6937623B2 (en) 1997-10-22 2005-08-30 Texas Instruments Incorporated Method and apparatus for coordinating multi-point to point communications in a multi-tone data transmission system

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USRE37802E1 (en) 1992-03-31 2002-07-23 Wi-Lan Inc. Multicode direct sequence spread spectrum
EP0613267A3 (fr) * 1993-02-08 1996-12-04 Philips Nv Système MDFO avec réduction de diaphonie.
EP0613267A2 (fr) * 1993-02-08 1994-08-31 Koninklijke Philips Electronics N.V. Système MDFO avec réduction de diaphonie
DE4330665A1 (de) * 1993-09-10 1995-03-16 Thomson Brandt Gmbh Verfahren zur ]bertragung von digitalisierten Fernseh- und/oder Tonrundfunksignalen
US5933454A (en) * 1994-06-02 1999-08-03 Amati Communications Corporation Multi-carrier data transmissions system using an overhead bus for synchronizing multiple remote units
WO1995034149A1 (fr) * 1994-06-02 1995-12-14 Amati Communications Corporation Procede et appareil de coordination de communications multi-point a point dans un systeme de transmission de donnees multitonalites
US7079549B2 (en) 1994-06-02 2006-07-18 Texas Instruments Incorporated Method and apparatus for coordinating multi-point to point communications in a multi-tone data transmission system
US7068678B2 (en) 1994-06-02 2006-06-27 Texas Instruments Incorporated Method and apparatus for coordinating multi-point to point communications in a multi-tone data transmission system
AU695092B2 (en) * 1994-06-02 1998-08-06 Amati Communications Corporation Method and apparatus for coordinating multi-point-to-point communications in a multi-tone data transmission system
GB2296407A (en) * 1994-12-22 1996-06-26 Roke Manor Research Frequency hopped cellular mobile radio system using orthogonal frequency division multiple access
GB2296407B (en) * 1994-12-22 1999-10-06 Roke Manor Research Frequency hopped cellular mobile radio systems
US6088327A (en) * 1995-06-07 2000-07-11 Deutsche Thomson-Brandt Gmbh Method and circuit arrangement for improving carrier separation for the transmission of OFDM signals
WO1996041458A1 (fr) * 1995-06-07 1996-12-19 Deutsche Thomson-Brandt Gmbh Procede et circuit destines a ameliorer la separation des porteuses lors de la transmission de signaux ofdm
EP0810753A3 (fr) * 1996-05-27 2001-09-05 Sony Corporation Système de transmission et réception à accès multiple, utilisant multiplexage de signaux à porteuses multiples
EP0810753A2 (fr) * 1996-05-27 1997-12-03 Sony Corporation Système de transmission et réception à accès multiple, utilisant multiplexage de signaux à porteuses multiples
US6192068B1 (en) 1996-10-03 2001-02-20 Wi-Lan Inc. Multicode spread spectrum communications system
US6320897B1 (en) 1996-10-03 2001-11-20 Wi-Lan Inc. Multicode spread spectrum communications system
US6160791A (en) * 1997-08-29 2000-12-12 Sony International (Europe) Gmbh Transmission system for the transmission of power control information in an OFDM system
EP0899923A1 (fr) * 1997-08-29 1999-03-03 Sony International (Europe) GmbH Transmission de signaux de réglage de puissance dans un système à modulation multiporteuse
US6937623B2 (en) 1997-10-22 2005-08-30 Texas Instruments Incorporated Method and apparatus for coordinating multi-point to point communications in a multi-tone data transmission system
US7110370B2 (en) 1997-10-22 2006-09-19 Texas Instruments Incorporated Method and apparatus for coordinating multi-point to point communications in a multi-tone data transmission system
US6535501B1 (en) 1997-11-07 2003-03-18 Sony International (Europe) Gmbh Transmission method and transmission apparatus for transmitting signals on the basis of a OFDM/TDMA-system in a GSM/system
EP0915586A1 (fr) * 1997-11-07 1999-05-12 Sony International (Europe) GmbH Transmission multiporteuse, compatible au système GSM existant
EP0930744A1 (fr) * 1998-01-02 1999-07-21 NewTec CY N.V. Procédé d'accès multiple employant une transmission multiporteuse
EP0938208A1 (fr) * 1998-02-22 1999-08-25 Sony International (Europe) GmbH Transmission multiporteuse, compatible au système GSM existant
US6545997B1 (en) 1998-02-22 2003-04-08 Sony International (Europe) Gmbh Transmission method and transmission apparatus for transmitting signals on the basis of a OFDM/TDMA-system with pilot symbols
DE10239810A1 (de) * 2002-08-29 2004-03-11 Siemens Ag Verfahren und Sendeeinrichtung zum Übertragen von Daten in einem Mehrträgersystem
US7688797B2 (en) 2002-08-29 2010-03-30 Gigaset Communications Gmbh Method and transmission device for transmission of data in a multi-carrier system

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