EP0462120A4 - Circuit and method for driving and controlling gas discharge lamps - Google Patents
Circuit and method for driving and controlling gas discharge lampsInfo
- Publication number
- EP0462120A4 EP0462120A4 EP19900903566 EP90903566A EP0462120A4 EP 0462120 A4 EP0462120 A4 EP 0462120A4 EP 19900903566 EP19900903566 EP 19900903566 EP 90903566 A EP90903566 A EP 90903566A EP 0462120 A4 EP0462120 A4 EP 0462120A4
- Authority
- EP
- European Patent Office
- Prior art keywords
- control
- circuit
- circuitry
- brightness
- lamp
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Withdrawn
Links
Classifications
-
- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
- H05B41/14—Circuit arrangements
- H05B41/36—Controlling
- H05B41/38—Controlling the intensity of light
- H05B41/39—Controlling the intensity of light continuously
- H05B41/392—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor
- H05B41/3921—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations
- H05B41/3927—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations by pulse width modulation
-
- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
- H05B41/14—Circuit arrangements
- H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
- H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
- H05B41/288—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps without preheating electrodes, e.g. for high-intensity discharge lamps, high-pressure mercury or sodium lamps or low-pressure sodium lamps
- H05B41/2881—Load circuits; Control thereof
-
- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
- H05B41/14—Circuit arrangements
- H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
- H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
- H05B41/288—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps without preheating electrodes, e.g. for high-intensity discharge lamps, high-pressure mercury or sodium lamps or low-pressure sodium lamps
- H05B41/292—Arrangements for protecting lamps or circuits against abnormal operating conditions
- H05B41/2921—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
- H05B41/2925—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against abnormal lamp operating conditions
Definitions
- the fluorescent lamp which requires less energy than the incandescent lamp to produce the same amount of light, has enjoyed increasing popularity.
- fluorescent lamps are used to the complete exclusion of incandescent lamps.
- the energy efficient fluorescent lamp has not replaced incandescent lamps to the same extent in other applications.
- Dimming circuits for incandescent lamps have been well- known for many years, but dimming circuits for fluorescent lamps are more difficult to construct; previous efforts at producing effective fluorescent lamp dimmers have not been entirely successful. This is especially true when dealing with reliable lamp starting and maintaining optimum lamp life.
- a fluorescent dimming circuit should compensate for several inherent disadvantages of the fluorescent lamp relative to the incandescent lamp.
- fluorescent lamps must be “started” at full intensity. Generally, these lamps will not “start” at all at reduced intensity. In any case, low intensity startups will reduce their lifespan.
- fluorescent lamps glow as a result of continuous excitation. The level of excitation can be reduced after the initial startup, but even a momentary interruption in this reduced excitation will put the lamp out, so that it must be .restarted with greater excitation.
- the fluorescent lamp requires external ballast circuitry to "start” the lamp, so that it becomes important that the ballast uses minimal power and starts the lamp in a way that does not reduce the life of the lamp.
- excitation of fluorescent lamps requires storage of potentially dangerous charges, so that it becomes important that the controls for the lamps be isolated from the excitation circuitry.
- an ideal fluorescent dimming circuit should provide safeguards against electrical shock.
- An inverter's output terminal(s) typically are connected resistively or capacitively to ground or a grounded inverter case.
- a significant amount of high frequency current can flow between the inverter's output terminal (s) and ground. This can cause an electrical shock or a fire.
- a fluorescent dimming circuit that prevents or limits the amount of current that can flow between a power inverter output terminal(s) and ground.
- Another primary object of the invention is provide harmonic mode starting of a gas discharge lamp to facilitate firing thereof.
- Another primary object of the present invention to provide a novel and improved control method and circuit for controlling power supplies that are responsive to a control voltage input.
- Yet another primary object of the invention is to provide a novel and improved system for preventing electric shock in a solid-state lamp ballast circuit.
- a more specific object of the present invention is to provide a new and improved control system for dimming gas-discharge lamps by creating an initial blocking period for lamp starting and by generating a pulse train signal with pulses of variable duty cycle, wherein the duty cycle of the pulses determines the brightness of the lamp.
- Another object of the present invention is to provide a new and improved control system for dimming gas-discharge lamps by generating a pulse train signal with pulses of variable duty cycle, wherein the pulse train signal is integrated to produce a DC level signal which may be applied as a control signal to a solid- state dimming ballast.
- Yet another object of the present invention is to provide a new and improved control system for dimming gas-discharge lamps which starts the lamps at full intensity, then dims the lamps to the desired level.
- Another object of the present invention is to provide a new and improved control system for dimming gas-discharge lamps which detects the occurrence of lamp-extinguishing momentary power interruptions and restarts the lamps at full intensity, thereafter dimming the lamps to the desired level.
- a further object of the present invention is to provide a new and improved control system for dimming gas-discharge lamps, the system having a user- adjustable brightness control, wherein another control, separate from the brightness control, sets the maximum amount of dimming selectable through the brightness control to prevent inadvertent damage to the lamps and circuitry or excessive reduction of light levels.
- a final object of the present invention is to provide a new and improved control system for dimming gas-discharge lamps wherein low-voltage, solid-state control circuitry is provided, this circuitry being electrically isolated from the ballast circuitry driving the lamps.
- the present invention further provides sensing circuitry connected between earth ground and one of the two output terminals of the inverter voltage source to establish a current path from earth ground to the one output terminal of the voltage source to thus monitor any high frequency current which may flow through a person, for example, connected between one of the output terminals of the inverter and earth ground; and limiting circuitry responsive to the sensing circuitry for limiting the high frequency current through the person in response to the current through the person exceeding a predetermined limit.
- control circuit and method for controlling the resonant inverter solid-state lamp ballast produces a pulse train signal with pulses of variable duty cycle.
- the signal is integrated to provide a DC signal which the solid- state ballast uses to control the level of dimming of the lamp.
- the duty cycle of the pulses is increased, the level of the DC signal decreases, so that the ballast dims the lamp. Conversely, reduction of the pulse duty cycle increases the brightness of the lamp.
- the control circuit incorporates delay circuitry which suppresses the pulse output at powerup, so that the lamp starts at full intensity. Thereafter, the delay circuitry adjusts the pulse signal so that the lamp intensity is adjusted smoothly to the desired level.
- a reset circuit resets the delay circuitry in case of a momentary power failure so that the lamp will restart at full intensity and then smoothly dim to the desired intensity, rather than starting at a low intensity.
- a brightness control circuit allows the user to set the desired light intensity, and an adjustable pulse control circuit allows limitation of the maximum amount of dimming. Overcurrent circuitry dis-ables pulse output if excessive current is drawn from the circuit.
- Figure 1 is a combined schematic and block diagram of a resonant inverter in accordance with the prior art.
- Figure 2 is a combined block and schematic diagram of a resonant inverter for use with a gas discharge lamp or the like.
- Figure 3 is an equivalent schematic diagram of the resonant circuit and gas discharge lamp of Figure 2.
- Figure 4 is a schematic diagram of a first illustrative embodiment of the resonant inverter circuitry of the present invention utilizing resonance mode starting at the fundamental frequency of the excitation signal and parallel resonance mode operation also at the fundamental frequency of the excitation signal.
- Figure 5 is a schematic diagram of a first illustrative current sensing circuit for use with the circuitry of Figure 4.
- Figure 6 is a circuit diagram of a second illustrative current sensing circuit for use with the circuitry of Figure 4.
- Figure 7 is a circuit diagram of a further illustrative embodiment of the resonant inverter circuitry of the present invention utilizing harmonic mode starting and fundamental resonance mode operation.
- Figure 8 is a circuit diagram of a further illustrative embodiment of the resonant inverter circuitry utilizing resonance mode starting and series resonance mode operation.
- Figure 9 is a circuit diagram of a further illustrative embodiment of the resonant inverter circuitry of the present invention utilizing harmonic mode starting.
- Figure 10 is a graph of the ringing signal which will occur across the gas discharge lamp to effect the firing thereof in the circuitry of Figure 9.
- Figure 11 is a graph of the voltage occurring across the gas discharge lamp of Figure 9 during operation thereof — that is, after the firing thereof by the voltage waveform of Figure 10.
- Figure 12 is a circuit diagram of a further modification of the resonant inverter circuitry of the present invention incorporating illustrative sense circuitry for sensing the voltage across the gas discharge lamp of the circuitry of Figure 9.
- Figure 13 is a block diagram of an off-line power inverter.
- Figure 14A is a circuit diagram of a full-wave rectifier for use with the power inverter of Figure 13.
- Figures 14B and 14C are voltage waveforms occurring at different points in the rectifier of Figure 14A.
- Figure 15 is a block diagram of an off-line resonant inverter utilizing an integrated circuit controller circuit.
- Figure 16 is a schematic diagram indicating possible current flow paths in the power inverter of Figure 13 in response to a person inadvertently contacting one of the inverter output terminals and the inverter chassis with the load disconnected.
- Figure 17 is a simplified schematic diagram of the circuitry of Figure 16.
- Figure 18 is a schematic diagram corresponding to that of Figure 16 and including an output transformer for the power inverter.
- Figure 19 is a simplified schematic diagram of the circuitry of Figure 18.
- Figures 20 and 21 are schematic diagrams of illustrative sensing circuits for use with the current limiting circuitry of the present invention.
- Figure 22 is a block diagram of an illustrative connection of the sensing circuitry of Figure 21 with the power inverter of Figure 13 where possible locations of relays for disabling the inverter are illustrated.
- Figure 23 is a schematic diagram illustrating how the power inverter according to Figure 13 may be used to drive a fluorescent lamp.
- Figure 24 is a schematic diagram corresponding to Figure 23 and further illustrating the dangerous condition that may exist when a person is in contact with one of the output terminals of the inverter and earth ground while the fluorescent lamp is disconnected.
- Figure 25 is a schematic diagram corresponding to Figure 24 and further illustrating sensing circuitry for disabling, if necessary, the controller for the power inverter.
- Figure 26 is a circuit diagram showing direct connection of the control circuitry of the present invention to the resonant inverter circuit of Figure 2.
- Figure 27 is a circuit diagram corresponding to Figure 26 but showing the control circuitry of the present invention connected to the resonant inverter circuit by an optocoupler.
- Figure 28 is a circuit diagram of the control circuitry of the present invention.
- Figure 29 is a graph of the waveform applied to the current sense terminal of the UC2843 by circuitry associated with the control circuitry of the present invention.
- the present invention is made up of three major circuit sections which will be described in order with reference to their associated drawing Figures.
- the circuit sections are: (1) Resonant Inverter Circuitry, (2) Current Limiting Circuitry, and (3) Control Circuitry.
- Resonant Inverter Circuitry is the Resonant Inverter Circuitry
- FIG. 1 A block diagram of a resonant inverter utilizing the integrated circuit (IC) SG2525 is shown in Figure 1.
- the combination of CT2 and RT2 determines the oscillator frequency of the IC.
- a resistor R4 is usually required between the terminals P15 and P13.
- a resistor divider comprising resistors R5 and R6 determines the amount of DC voltage applied to the non inverted terminal (pin 2) of the operational amplifier contained in the SG2525 integrated circuit. This voltage, in turn, sets the magnitude of the duty cycle of the output pulses from pin 14 and pin 11 of the SG2525.
- an impedance Z2 is necessary between the inverted terminal (pin 1) and the compensation terminal (pin 9) of the SG2525 for loop stability of the IC.
- Output signals from pin 11 and pin 14 periodically and alternately turn Q2 and Q3 on and off.
- Q2 when Q2 is on, Q3 is off, and when Q2 is off, Q3 is on.
- Q2 when Q2 is off but Q3 is on, stored energy from CR flows back through LR and Q3.
- the pulse repetition frequency is identical with the resonance frequency of the LC (LR and CR) network, the circuit can be described as a resonant inverter.
- FIG. 2 An efficient and economical ballast configuration based on a resonant inverter technique is shown in Figure 2.
- LR and CR form a resonant circuit and the lamp Tl acts as a load across CR.
- Figure 3 is a circuit diagram equivalent to the Figure 2 connections of LR, CR, and Tl, where the impedance of load Tl is RL.
- the respective impedances of the circuit parameters of Figure 3 can be described as follows:
- XCR XLR.
- RL is replaced by the lamp Tl. Initially, before the lamp Tl fires, it offers an infinite impedance (that is, no current flow therethrough) and as a result the voltage across CR or Tl ( Figure 2) continues to grow. However, once the voltage across Tl reaches the lamp firing potential, the lamp Tl fires and offers much lower impedance. At this instance, due to the lamp chracteristic, the voltage across Tl clamps down to the normal lamp operating potential and stays there. This is a convenient and reliable mechanism for starting and operating a fluorescent lamp.
- the current through the resonant inductor LR is equal to the vector sum of the current through the resonant capacitor CR and the current through the load or the lamp T. This is true, because, during the normal operation the lamp T can be considered mostly a resistive load and, as a result, the current through the capacitor CR will have 90 degree phase difference, with respect to the lamp current.
- the current through LR which is also the total circuit current, can be described as,
- the voltage across the resonant capacitor is the same as the voltage across the lamp, ylamp.
- the current through the capacitor CR is determined by the ratio of the lamp firing potential to the impedance of CR. That is,
- j _CR firing equals the total load current, which is circulating between CR and LR through the power switches Q2 and Q3. For this reason, if the lamp firing potential is very high, depending on XCR, a very large amount of circulating current can flow through Q2 and Q3 before the lamp fires. This large circulating current during starting may exceed the maximum rated current through Q2 and Q3 and thereby, may destroy Q2 and Q3.
- the switches SI and S2 are closed by, for example, sensing current through the lamp and using this sense signal to activate a switch that will close SI and S2, for example, a relay.
- Current sensing can be accomplished conveniently by using a sense resistor (RS) that is placed in series with the lamp Tl as shown in Figure 5.
- Current through Tl can also be sensed by using a conventional current transformer (CT) as shown in Figure 6 where the Fig. 5 and Fig. 6 sensing circuits may also be used in the other embodiments of the invention.
- CT current transformer
- harmonic mode starting at a harmonic (fn) of the fundamental fr of the excitation signal
- parallel resonance mode operation at the fundamental frequency fr
- the natural resonance frequency of the circuit can be made equal to any higher harmonic frequency (fn) of the excitation frequency (fr) .
- the voltage Cl developed across Cl is dependent on the values of (LI + L2) and Cl and their quality. Thereby, the right value and quality components should preferably be selected. Examples of preferred components are polypropylene capacitors, as will be further discussed below.
- inductors LI and L2 with C2 form the resonance circuit that resonates at the excitation frequency.
- the switch SI closes, and LI and Cl forms the resonant network.
- the effect of C2 can now be ignored where, in this mode, the lamp Tl is in series with Cl and LI. Since C2 can be made very small in value, current flow through through C2 (and thus power switches Q2 and Q3) can be kept very small.
- the high impedance of C2 at fn is such that a firing voltage sufficient in magnitude to fire the lamp can readily be developed across this capacitor.
- Figure 8 can also be arranged for: 1) resonance mode starting but non-resonance series operation, 2) harmonic mode starting but series resonance mode operation and 3) harmonic mode starting and non-resonance series operation.
- harmonic mode starting and non-resonance operation are utilized as shown in Figure 9.
- Figure 9 In this embodiment,
- voltage across Cl can be increased to a very high level by choosing low loss LI and Cl and by resonating them at harmonics higher than the fundamental. That is, by keeping the excitation frequency (fr) fixed, the resonant network is so chosen that it resonates at the nth harmonic frequency, (fn) .
- this embodiment can be used in the circuit of Figure 1 where the sensing circuits of Figs. 5 or 6 are not required.
- Tl is a commercially available 250 watt High Pressure Sodium (HPS) lamp. It typically requires approximately 2,500 peak voltage to start. Once the lamp is fired, the operating potential across the lamp is only 100 volts. Lamp firing voltage and operating voltage waveforms are shown in Figures 10 and 11.
- fr 30,000 Hz
- Vin 360v.
- harmonic mode starting is advantageous because there is a rapid build-up of voltage such that at the natural (or resonant) frequency of the circuit, the lamp firing potential can be easily exceeded.
- the circuit impedance is typically such in harmonic mode starting that the average power flow can be kept within the maximum rating of the power switches Q2, Q3, for example.
- the impedances of LI and Cl are the same, namely, 245 ohms.
- the impedance of LI must be equal to the impedance of Cl so that they cancel each other.
- the amount of current flow and thereby the voltage growth across Cl can be further controlled by incorporating a sense network as shown in Figure 12. Accordingly, a high impedance resistor divider network (RI & R2) placed across Cl, senses voltage which is then rectified by the diode Dl. This rectified signal can now be used to interrupt the frequency generator (SG2525 in Fig. 1) which generates fr. The interruption of the frequency generator via the soft start pin is further described in the current limiting circuitry section below.
- the Q-factor or the quality of the inductors and the capacitors should be good in order for harmonic mode starting to be effective not only in the embodiment of Figure 9 but in the other harmonic mode starting embodiments.
- the quality of an inductor depends primarily on the magnetic core material, resistance of the winding, skin depth associated with the high frequency excitation, etc. Poorly designed high frequency inductors can cause core saturation, and excessive heat dissipation.
- the quality of a capacitor depends on its construction, such as, frequency response characteristic of the dielectric film, associated effective series resistance (ERS) , leakage current characteristics, high frequency ripple current capability, etc.
- ERS effective series resistance
- leakage current characteristics high frequency ripple current capability
- the voltage that can be applied across a capacitor without dielectric breakdown varies with frequency.
- a polypropylene capacitor would be preferred over a polyester capacitor, for example.
- Switch 17 is an input section for AC power.
- Power line protection circuitry section 18 includes a fuse, etc.
- EMI section 19 comprises conducted electro-magnetic interference (EMI) suppression circuitry.
- L3 is a common mode inductor
- L4a and L4b are differential mode inductors
- Yl and Y2 are equal value capacitors for limiting leakage current to earth ground where the values of Y are such the high frequency EMI generated by the inverter are shunted to earth ground while the lower frequency AC signal is not so shunted.
- Rectifier section 20 comprises half or full wave rectification circuitry. Ripple voltage is filtered in filer section 21. The DC voltage is converted to a high frequency by a high frequency inverter and control circuitry as shown in section 22. Isolation transformer section 23 is optional, and provides a high frequency transformer for voltage isolation, step-up, step-down or for multiple output secondary voltages.
- High frequency power inverter circuitry 22 may preferably comprise the SG2525-based resonant inverter circuitry previously described with reference to Figures 1 through 12.
- a block diagram of an off-line resonant inverter utilizing the integrated circuit (IC) SG2525 is shown in Figure 15.
- the off-line high frequency power inverter disclosed herein is not limited to use with the resonant inverter circuitry described previously.
- the off-line high frequency power inverter can also be constructed using power inverter topology other than resonant inverter topology. For example, push-pull topology, half bridge topology, and others can be used.
- the current flow will be determined by the construction of the transformer. For example, electrostatic shielding between primary and secondary winding, capacitive coupling (Cw) between these two windings, etc., will play major roles.
- Figures 16 and 18 Possible current flow paths between the terminals A or B and the ground G are shown in Figures 16 and 18 where Figure 18 includes an isolation transformer placed between the inverter output and the load.
- Figures 17 and 19 are simplified equivalent circuits of Figures 16 and 18, respectively. In these simplified cases, the effects of bridge rectifier diode drop voltages are neglected.
- a simple resistor sensor circuit is not suitable in the circuitry of Figure 13 because, during normal operation, a continuous high voltage pulsating DC potential exists between G and P (+) or, G and P (-) .
- the frequency of this pulsating high voltage DC is determined by the input AC, as discussed hereinbefore with respect to Figures 14A, 14B, and 14C. However, if such pulsating voltage is not present, a simple resistor circuit would be suitable.
- high frequency AC voltage sensing can be accomplished by a capacitor and resistors combinations as shown in Figures 20 and 21.
- the circuit of Figure 21 converts the high frequency AC signal into a DC signal.
- the amount of sense voltage VS is determined by: (1) the inverter frequency (fi) , (2) Rp, (3) CS, (4) RSI, and (5) RS2.
- XCS 1607 Ohms.
- XCS 803,500 Ohms, which is 500 times higher.
- the 60 Hertz AC input signal will be attenuated 500 times compared to a 30,000 Hertz AC signal.
- the high frequency signal that is detected by the circuits of Figures 20 and 21 is a function of the high frequency current that is flowing between power inverter output terminal and ground, when the resistor Rp is placed between them.
- the detected signal can be used, for example, to turn on a relay or some other switch to disable the power inverter output ( Figure 22) temporarily or permanently in order to avoid that current flow.
- the relay can be placed between any of the sections shown in the circuit, i.e. in any location indicated by an X.
- either of the sensing circuits of Figures 20 or 21, for. example may replace either the Yl or the Y2 capacitor of Figure 13 where preferably the Y2 capacitor would be replaced with one of the sensing circuits.
- the value of CS of Figures 20 or 21 could preferably be exactly the same as the remaining Yl capacitor.
- the sensing circuitry serves not only its sensing function as described above, but also serves the EMI suppression function formally performed by the replaced Yl capacitor.
- the detected signal via the circuits described by Figures 20 and 21 can also be used to regulate or limit current between the inverter output and ground when they are short circuited or connected by a resistor.
- a resistor As an example, refer to the resonant inverter of Figure 15.
- the resonant inverter can drive a load such as a fluorescent lamp Tl. This is shown in Figure 23.
- a person can accidentally be in contact with terminal A and the grounded inverter case simultaneously. In analyzing this situation, the person can be replaced by an equivalent resistor of 500 ohms, as shown in Figure 24.
- Figure 25 shows one such detection circuit connected to the resonant inverter circuit described previously.
- the values of CS, RSI and RS2 are preferably chosen such that, as soon as the peak voltage across Rp reaches 21.7 volts, the voltage that develops between the base and emitter of a transistor Q4 is enough to turn on the transistor where soft start capacitor 24 is connected across the collector and emitter of Q4.
- Turning Q4 on causes a pull-down of the soft start pin 8 of the SG2525 IC. This, in turn, causes an immediate shut-down of the output drive pulses emanating from pin 11 and pin 14. Power inverter switches Q2 and Q3 then stop functioning. In this situation no current flows through Rp, and Q4 turns off.
- control circuit provides a preferred alternative to the method previously described for controlling the brightness of the load Tl as shown in Figures 2, 15 etc. by adjusting the variable resistor R6.
- the new and improved pulse generating circuit is shown generally at 42 (this circuit will be described later in full detail) .
- the output 43 of this circuit is connected to the base of an output transistor 44.
- the collector of an output transistor 44 is connected to non-inverting input NI (pin 2) of the SG2525 IC, while the emitter of output transistor 44 is connected to ground.
- An integrating capacitor 46 is connected between the non-inverting input NI and ground.
- the pulse generating circuit 42 preferably generates a variable duty cycle, square wave pulse train at a fixed frequency greater than 1 kHz.
- the output pulses at output 43 control the charging of integrating capacitor 46.
- pulse generating circuit 42 produces a pulse at output 43
- the voltage applied to the base of transistor 44 turns on transistor 44, allowing current to flow from the collector to the emitter of the transistor 44.
- the collector of transistor 44 is connected to the capacitor 46 and the non-inverting input NI, and since the emitter of transistor 44 is connected to ground, a pulse from pulse generating circuit 42 effectively grounds the integrating capacitor 46, tending to discharge the capacitor 46.
- output 43 is not producing a pulse, transistor 44 is turned off, and integrating capacitor 46 tends to charge to the level of the voltage drop across variable resistor R6 as determined by the voltage divider comprising resistor R5 and variable resistor R6.
- the voltage at non-inverting input NI (pin 2) varies with the duty cycle of the pulses at output 43. Since the output 43 produces a series of pulses at high frequency, the pulses produce a periodic pull up and down of the DC level across integrating capacitor 46.
- the integrating capacitor 46 integrates over time the DC level shift produced by the pulsed output 43, so that for a given pulse duty cycle, a continuous DC voltage appears at non-inverting input NI (pin 2) .
- the DC voltage at non-inverting input NI (pin 2) will vary with the duty cycle of the pulsed output 43 in the following manner. As the duty cycle increases, the capacitor 46 will be grounded for a relatively greater portion of time, and the voltage at non-inverting input NI (pin 2) will be reduced. Conversely, as the duty cycle of pulses at output 43 is reduced, the voltage at non-inverting input NI (pin 2) will be increased.
- FIG 27 shows a preferred embodiment of the circuit of Figure 26 wherein the output transistor 44 is replaced by a conventional opto-isolator 48.
- the opto-isolator 48 comprises a light-emitting diode (LED) 50 and a phototransistor 52.
- the light-emitting diode 50 is connected between output 43 and ground.
- the phototransistor 52 has its collector connected to non- inverting input NI (pin 2) and its emitter connected to grou
- the phototransistor 52 turns on in response to light emissions from LED 50, which operates in response to the pulses from output 43. This embodiment thus operates in substantially the same manner as the embodiment shown in Figure 26.
- the opto- isolator 48 electrically isolates the resonant inverter circuitry from the pulse generating circuitry 42.
- the resonant inverter circuitry may contain large voltages and current, and as will be seen, controls for the pulse generating circuit 42 will be handled by human operators. Therefore, this electrical isolation provides a substantial safety benefit.
- the pulse generating circuit 42 comprises a power supply section 54, a reset section 56, a delay section 58, an overcurrent section 60, a pulse control section 62, a brightness control section 64, and a variable duty cycle frequency source 65.
- variable duty cycle frequency source 65 may preferably be an UC2843 integrated circuit manufactured by Motorola, although other integrated circuits could be used or a circuit could be constructed to perform the necessary functions.
- the operation of the frequency source 65 is described in detail in Motorola publications which will be familiar and accessible to those skilled in the art. However, the functions of the pins used in this circuit are described in Table 1 in sufficient detail to permit those skilled in the art to understand the circuit and to practice the invention disclosed.
- Compensation Voltage may be applied externally to vary the duty cycle of the pulses.
- OSC Provides sawtooth wave output with frequency depending on external circuitry.
- Output Produces variable duty cycle pulse output with frequency depending on external circuitry connected to OSC terminal and duty cycle depending on voltage applied to Compensation terminal.
- Vcc Power supply (+12v DC) Vcc Power supply (+12v DC) .
- the power supply section 54 comprises a transformer 66, a full-wave bridge rectifier 68, a capacitor isolation diode 70, and a smoothing capacitor 72.
- the power supply section 54 is preferably also provided with a conventional three-terminal 12 volt voltage regulator 84 and an associated capacitor 86.
- the voltage regulator 84 has an input terminal 88, an output terminal 90, and a ground terminal 92.
- Alternating current input from an AC source 74 is connected to the primary coil of transformer 66.
- the turns ratio of transformer 66 is selected with reference to the voltage of AC source 74 so that 12 volts AC is produced on the secondary coil.
- Full-wave bridge rectifier 68 is a conventional device.
- the rectifier 68 has two input terminals 75 and 78 and two output terminals 80 and 82.
- the two terminals of the secondary coil of transformer 66 are connected respectively to input terminals 75 and 78 of rectifier 68.
- Output terminal 80 of rectifier 58 is connected to circuit and Earth ground, while output terminal 82 is connected to the anode of isolation diode 70 and provides a rectified 12 volt DC output thereto.
- the cathode of diode 70 is connected to the input terminal 88 of regulator 84 and to the positive terminal of smoothing capacitor 72.
- the negative terminal of smoothing capacitor 72 is connected to both circuit ground and Earth ground.
- the output terminal 90 of regulator 84 is connected to Vcc (pin 7) of variable duty cycle frequency source 65, and ground terminal 92 is connected to ground.
- the capacitor 86 is connected between the output terminal 92 of regulator 84 and ground.
- the voltage regulator 84 compensates for variations in the voltage of AC source 74, thus stabilizing the 12 volt DC power provided to the integrated circuits of frequency source 65. A stable voltage supply for frequency source 65 is necessary to avoid variations in the pulse signal output 43 of the frequency source 65.
- the 12 volt DC regulated output at output terminal 90 of regulator 84 will be used as the DC source 24 connected to Vcc of the pulse width modulator 4 (shown in Figure 26) .
- the entire circuit may be controlled by a single power switch (not shown in the drawings) .
- This switch may be any conventional switch and may be installed in the power supply circuitry in a number of ways which are conventional and will be immediately apparent to those skilled in the art.
- the brightness control section 64 comprises a variable resistor 94 and a voltage divider resistor 96.
- the variable resistor 94 is connected between the compensation pin (pin 1) of frequency source 65 and ground.
- the voltage divider resistor 96 is connected between Vref (pin 8) of frequency source 65 and the compensation pin (pin 1) of frequency source 65.
- Vref (pin 8) of frequency source 65 provides a constant 5.1 volt DC signal.
- the variable resistor 94 and resistor 96 form a voltage divider so that, as the variable resistor 94 is adjusted, the voltage applied to the compensation pin (pin 1) of frequency source 65 will vary.
- the voltage on the compensation pin (pin 1) of frequency source 65 controls the duty cycle of the pulses produced at output 43, the duty cycle determining the brightness of the load Tl as described previously with reference to Figure 26.
- the power supply switch previously described may be integrated with the variable resistor 94 in a manner well known in the art.
- the delay section 58 comprises a PNP transistor 98, a resistor 100, capacitor 102, and resistor 104.
- the emitter of transistor 98 is connected to the compensation terminal (pin 1) of frequency source 65, while the collector of transistor 98 is connected to ground.
- the base of transistor 98 is connected to one terminal of resistor 104, the other terminal of the resistor 104 being connected to the output terminal 82 of bridge rectifier 68.
- the positive terminal of capacitor 102 is connected to the base of transistor 98, while the negative terminal of capacitor 102 is connected to ground.
- Resistor 100 is connected between the base of transistor 98 and ground.
- the delay section 58 provides novel and uniquely advantageous operation because, in operation, the delay section 58 suppresses transmission of the dimming signal 43 at power-up. With the dimming signal suppressed by delay section 58, the load Tl (shown in Figure 26) is started at full brightness. Full-brightness starting is essential for two reasons: First, full-brightness starting prolongs the life of the fluorescent tubes. Second, fluorescent tubes may not start at all if power is not provided for the full duty cycle.
- delay section 58 to suppress the dimming signal 43 will now be described in detail.
- the transistor 98 When no power is applied to the circuit 42 from AC source 74, the transistor 98 will conduct fully, thus effectively grounding the compensation terminal (pin 1) of frequency source 65.
- the compensation terminal When the compensation terminal is grounded in this manner, a zero duty cycle at output 43 is selected.
- the brightness of the load Tl (shown in Figure 26) varies inversely with the duty cycle of the pulsed output 43.
- a zero duty cycle of the pulsed output 43 corresponds to full brightness at the load Tl (shown in Figure 26) . Therefore, when the transistor 98 is fully conductive, the load Tl will be at maximum brightness.
- the capacitor 102 When power is applied to the circuit 42, the capacitor 102 will charge according to a time constant determined by the values of resistors 100 and 104 and capacitor 102. As the capacitor 102 charges, the transistor 98 will be rendered less conductive, until the transistor 98 ceases to conduct. When the transistor 98 ceases to conduct, the delay section 58 will have no effect on the voltage at the compensation pin (pin 1) of frequency source 65. The voltage at the compensation pin (pin 1) of frequency source 65 will then be controlled entirely by the brightness control section 64.
- the delay section 58 will initially inhibit any dimming of the load Tl (as shown in Figure 2) , regardless of the setting of variable resistor 94 (the brightness control) .
- the load Tl will "start” at full brightness.
- the delay section 58 will cease to inhibit dimming and the load Tl will dim to the level selected by means of variable resistor 94.
- An important feature of the present invention is that the fluorescent lamp Tl does not come on at full brightness and then suddenly become dim; the steadily increasing voltage across capacitor 102 as it charges reduces the conductance of transistor 98 steadily over a brief period of time.
- the voltage at the compensation pin (pin 1) of frequency source 65 will therefore increase steadily from zero to the level determined by the setting of variable resistor 94.
- the fluorescent lamp Tl will come on at full brightness, and then dim to the preset level in a smooth and pleasing manner.
- the length of the delay produced by delay section 58 can be adjusted by changing the value of resistors 100 and 104 and capacitor 102 in accordance with well-known time constant principles.
- Reset section 56 operates to reset the delay section 58 during a power failure, preparing delay section 58 to operate properly when power is returned to the circuit.
- Reset section 56 comprises a diode 106, resistor 108, PNP transistor 110, filter capacitor 112, and voltage divider resistors 114 and 116.
- the anode of diode 106 is connected to the base of delay section transistor 98, and the cathode of diode 106 is connected to one terminal of resistor 108.
- the other terminal of resistor 108 is connected to the emitter of transistor 110.
- Resistor 108 preferably has a small value, in the range of 5-7 Ohms.
- the collector of transistor 110 is connected to ground.
- the positive terminal of filter capacitor 112 is connected to the base of transistor 110, while the negative terminal of the capacitor 112 is connected to ground.
- One terminal of resistor 114 is connected to the output terminal 82 of full-wave bridge rectifier 68, while the other terminal of the resistor 114 is connected to the base of transistor 110.
- Resistor 116 is connected between the base of transistor 110 and ground.
- Resistors 114 and 116 together form a voltage divider which determines the voltage at the base of transistor 110.
- the values of resistors 114 and 116 are chosen with reference to the values of resistors 100 and 104 so that transistor 110 does not conduct while AC power source 74 is providing power to the circuit 42.
- the value of capacitor 112 is chosen with reference to the values of resistors 114 and 116 so that, if power is removed from the circuit, capacitor 112 will discharge through resistor 116 in about 1 millisecond.
- the reset section 56 operates as follows: The voltage at the base of transistor 110 falls to zero within one millisecond as the capacitor 112 discharges through resistor 116. Because delay section capacitor 102 is still charged, the voltage at the emitter of transistor 110 is considerably greater than zero. Therefore, transistor 110 begins to conduct, effectively shorting and discharging the delay section capacitor 102. Thus, the reset section 56 quickly prepares the delay section 58 so that the fluorescent tube Tl may be restarted automatically at full brightness as described previously.
- the diode 70 is provided in the power supply section 54 to isolate the reset section 56 from filter capacitor 72 so that, during a power interruption, filter capacitor 72 will not discharge through the reset section 56 and prevent proper operation of the reset section 56.
- Pulse control section 62 determines the frequency of the pulsed output 43 and limits the maximum duty cycle of said output pulses.
- Pulse control section 62 comprises NPN transistor 118, frequency set capacitor 120, frequency set resistor 122, resistor 124, variable resistor 126, and resistor 128.
- the base of transistor 118 is connected to the oscillator terminal (pin 4) of frequency source 65.
- the collector of transistor 118 is connected to Vref (pin 8) of frequency source 65, and the emitter of transistor 118 is connected to one of the two terminals of resistor 124.
- the other terminal of resistor 124 is connected to one of the two terminals of variable resistor 126.
- the other terminal of variable resistor 126 is connected to the current sense terminal (pin 3) of frequency source 65.
- the resistor 128 is connected between the current sense terminal (pin 3) of frequency source 65 and ground.
- the frequency set resistor 122 is connected between the oscillator terminal (pin 4) of frequency source 65, and Vref (pin 8) of frequency source 65.
- the frequency set capacitor 120 is connected between the oscillator terminal (pin 4) of frequency source 65 and ground.
- the oscillator terminal (pin 4) of the frequency source 65 will produce a ramp signal (sawtooth wave) with a DC offset, the frequency of the ramp signal depending on a time constant determined by the values of frequency set resistor 122 and frequency set capacitor 120.
- the resistor 122 and capacitor 120 will be chosen so that the frequency of the ramp signal is greater than 1 kiloHertz.
- the ramp signal from the oscillator terminal (pin 4) of frequency source 65 is transmitted by means of the transistor 118 to a voltage divider formed by resistors 124 and 128 and the variable resistor 126.
- the operation of these voltage divider resistors causes the signal on the current sense terminal (pin 3) of frequency source 65 to be at all times a percentage of the varying voltage at the oscillator terminal (pin 4) of frequency source 65.
- the percentage or fraction of the oscillator terminal output that will appear at the current sense terminal (pin 3) of frequency source 65 is determined by the setting of variable resistor 126.
- the peak voltage output of the oscillator terminal (pin 4) of an UC2843 integrated circuit is approximately 2.8 volts; the minimum voltage output (D.C. offset) is 1.2 volts.
- resistors 124 and 128 and the setting of variable resistor 126 are chosen so that the peak voltage applied to the current sense terminal (pin 3) of fre-quency source 65 will be approximately 1.4 volts.
- the frequency source 65 will inhibit generation of a pulse signal at output 43 whenever the voltage applied to the current sense terminal (pin 3) is greater than about one volt. Therefore, the effect of applying a high frequency ramp signal to the current sense terminal (pin 3) is to suppress pulse generation during a portion of each ramp cycle.
- the ramp signal 131 has a peak voltage Vmax.
- Vmax is a fraction of the peak voltage of the ramp signal at the oscillator terminal (pin 4) of frequency source 65.
- Vmax is preferably about 1.4 volts.
- a single ramp cycle 129 takes place over a time period encompassing a first time period 130 -and a second time period 132.
- the voltage of the ramp signal rises from 0.6 volts to one volt; during this period 130, the frequency source 65 is not inhibited form transmitting a pulse at output 43.
- the frequency source 65 is not inhibited form transmitting a pulse at output 43.
- whether a pulse is transmitted by frequency source 65 and the actual duration of any pulse transmitted are determined by brightness control section 64, delay section 58, and reset section 56 in the manner explained previously.
- the voltage of the ramp signal 131 applied to the current sense terminal (pin 3) exceeds one volt, and the frequency source 65 is inhibited from producing any signal at output 43.
- the application of the ramp signal 131 to the current sense terminal (pin 3) limits the maximum duty cycle of the pulses at the output 43.
- the maximum duty cycle of the pulsed output 43 can be adjusted by means of variable resistor 126, and may be set at a value other than 50% as dictated by the requirements of the consumer or the design parameters of the resonant inverter ballast (shown in Figure 27) .
- the overcurrent section 60 is a protective circuit that disables pulsed output 43 if excessive current is drawn from the output 43.
- Overcurrent section 60 comprises a resistor 134 and a diode 136.
- the anode of diode 136 is connected to an output reference 45 which may serve as the ground reference for the output signal 43.
- the cathode of diode 136 is connected to the current sense terminal (pin 3) of frequency source 65.
- the resistor 134 is connected between the anode of diode 136 and ground. The diode 136 prevents transmission of the ramp signal at the current sense terminal (pin 3) to the output reference 45.
- the output 43 of frequency source 65 is inhibited when more than one volt is applied to the current sense terminal (pin 3) .
- the voltage drop across diode 136 is approximately 0.6 volts; therefore the output 43 will be inhibited if the voltage at the anode of diode 136 is greater than 1.6 volts. This condition will occur when the voltage drop across resistor 134 is greater than 1.6 volts.
- resistor 134 may be a 4.7 Ohm resistor, so that when more than 0.34 Amperes of current is drawn from output 43, the voltage drop across resistor 134 will be greater than 1.6 volts and the output 43 will be disabled.
- the overcurrent section 60 prevents damage to the circuit of the present invention.
- each resonant inverter ballast connected to the pulse generating circuit 42 will draw current, so that there is a practical limit to the number of resonant inverter circuits that can be controlled by a single pulse generating circuit 42.
- the pulse generating circuit as disclosed will drive approximately 16 ballasts without exceeding 0.34 Amp current draw from output 43.
- an NPN power transistor can be used to increase the fanout capability of the circuit 42.
- the base of the power transistor may be connected to the output 43, while the collector of the power transistor is connected to a DC power source such as that provided at Vcc (pin 7) of frequency source 65.
- the pulse signal output to the ballasts is then taken at the emitter of the power transistor.
- the fanout capability of the circuit 42 can be expanded to allow control of almost any number of ballasts using well- known techniques.
Landscapes
- Circuit Arrangements For Discharge Lamps (AREA)
Abstract
Description
Claims
Applications Claiming Priority (6)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US308515 | 1989-02-10 | ||
US07/308,515 US4943886A (en) | 1989-02-10 | 1989-02-10 | Circuitry for limiting current between power inverter output terminals and ground |
US33205589A | 1989-04-03 | 1989-04-03 | |
US332055 | 1989-04-03 | ||
US410480 | 1989-09-21 | ||
US07/410,480 US5245253A (en) | 1989-09-21 | 1989-09-21 | Electronic dimming methods for solid state electronic ballasts |
Publications (2)
Publication Number | Publication Date |
---|---|
EP0462120A1 EP0462120A1 (en) | 1991-12-27 |
EP0462120A4 true EP0462120A4 (en) | 1992-12-30 |
Family
ID=27405320
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
EP19900903566 Withdrawn EP0462120A4 (en) | 1989-02-10 | 1990-02-12 | Circuit and method for driving and controlling gas discharge lamps |
Country Status (5)
Country | Link |
---|---|
EP (1) | EP0462120A4 (en) |
KR (1) | KR910700598A (en) |
AU (2) | AU642862B2 (en) |
CA (1) | CA2046278A1 (en) |
WO (1) | WO1990009729A1 (en) |
Families Citing this family (10)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
IL105564A (en) * | 1993-04-30 | 1996-06-18 | Ready Light Energy Ltd | Discharge dimmer lamp |
GB2279187A (en) * | 1993-06-19 | 1994-12-21 | Thorn Lighting Ltd | Fluorescent lamp starting and operating circuit |
US5363018A (en) * | 1993-09-16 | 1994-11-08 | Motorola Lighting, Inc. | Ballast circuit equipped with ground fault detector |
US5744913A (en) * | 1994-03-25 | 1998-04-28 | Pacific Scientific Company | Fluorescent lamp apparatus with integral dimming control |
US5596247A (en) * | 1994-10-03 | 1997-01-21 | Pacific Scientific Company | Compact dimmable fluorescent lamps with central dimming ring |
US5962988A (en) * | 1995-11-02 | 1999-10-05 | Hubbell Incorporated | Multi-voltage ballast and dimming circuits for a lamp drive voltage transformation and ballasting system |
AU2003902210A0 (en) | 2003-05-08 | 2003-05-22 | The Active Reactor Company Pty Ltd | High intensity discharge lamp controller |
DE10359882A1 (en) * | 2003-12-19 | 2005-07-14 | Patent-Treuhand-Gesellschaft für elektrische Glühlampen mbH | Circuit arrangement for operating electric lamps |
DE102010048755A1 (en) * | 2010-10-16 | 2012-04-19 | Hella Kgaa Hueck & Co. | Circuit arrangement for supplying power to LEDs, has selection element for selecting capacitor, and resistors for adjusting potential at input for soft start of direct current to direct current converter |
KR102154155B1 (en) * | 2018-08-24 | 2020-09-09 | 주식회사 솔루엠 | Planar transformer having y-capacitor |
Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4237403A (en) * | 1979-04-16 | 1980-12-02 | Berkleonics, Inc. | Power supply for fluorescent lamp |
EP0259646A1 (en) * | 1986-08-19 | 1988-03-16 | Siemens Aktiengesellschaft | Method and arrangement for supplying a gaseous discharge lamp |
DE3632272A1 (en) * | 1986-09-23 | 1988-04-07 | Erzmoneit Dorit | Electronic circuit arrangement for operating low-pressure discharge lamps in a series circuit |
Family Cites Families (9)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3611021A (en) * | 1970-04-06 | 1971-10-05 | North Electric Co | Control circuit for providing regulated current to lamp load |
US4053813A (en) * | 1976-03-01 | 1977-10-11 | General Electric Company | Discharge lamp ballast with resonant starting |
US4395660A (en) * | 1980-12-31 | 1983-07-26 | Waszkiewicz E Paul | Lamp dimmer circuit utilizing opto-isolators |
US4388563A (en) * | 1981-05-26 | 1983-06-14 | Commodore Electronics, Ltd. | Solid-state fluorescent lamp ballast |
US4399391A (en) * | 1981-06-10 | 1983-08-16 | General Electric Company | Circuit for starting and operating fluorescent lamps |
US4631450A (en) * | 1983-12-28 | 1986-12-23 | North American Philips Lighting Corporation | Ballast adaptor for improving operation of fluorescent lamps |
US4613934A (en) * | 1984-03-19 | 1986-09-23 | Pacholok David R | Power supply for gas discharge devices |
US4544863A (en) * | 1984-03-22 | 1985-10-01 | Ken Hashimoto | Power supply apparatus for fluorescent lamp |
US4949020A (en) * | 1988-03-14 | 1990-08-14 | Warren Rufus W | Lighting control system |
-
1990
- 1990-02-01 KR KR1019900702248A patent/KR910700598A/en not_active Application Discontinuation
- 1990-02-12 EP EP19900903566 patent/EP0462120A4/en not_active Withdrawn
- 1990-02-12 AU AU51066/90A patent/AU642862B2/en not_active Ceased
- 1990-02-12 WO PCT/US1990/000798 patent/WO1990009729A1/en not_active Application Discontinuation
- 1990-02-12 CA CA002046278A patent/CA2046278A1/en not_active Abandoned
-
1994
- 1994-02-04 AU AU54928/94A patent/AU674187B2/en not_active Expired - Fee Related
Patent Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4237403A (en) * | 1979-04-16 | 1980-12-02 | Berkleonics, Inc. | Power supply for fluorescent lamp |
EP0259646A1 (en) * | 1986-08-19 | 1988-03-16 | Siemens Aktiengesellschaft | Method and arrangement for supplying a gaseous discharge lamp |
DE3632272A1 (en) * | 1986-09-23 | 1988-04-07 | Erzmoneit Dorit | Electronic circuit arrangement for operating low-pressure discharge lamps in a series circuit |
Non-Patent Citations (1)
Title |
---|
See also references of WO9009729A1 * |
Also Published As
Publication number | Publication date |
---|---|
AU5106690A (en) | 1990-09-05 |
KR910700598A (en) | 1991-03-15 |
WO1990009729A1 (en) | 1990-08-23 |
CA2046278A1 (en) | 1990-08-11 |
AU642862B2 (en) | 1993-11-04 |
AU674187B2 (en) | 1996-12-12 |
AU5492894A (en) | 1994-04-14 |
EP0462120A1 (en) | 1991-12-27 |
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