EP0238903B1 - Reference current source - Google Patents
Reference current source Download PDFInfo
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- EP0238903B1 EP0238903B1 EP87103064A EP87103064A EP0238903B1 EP 0238903 B1 EP0238903 B1 EP 0238903B1 EP 87103064 A EP87103064 A EP 87103064A EP 87103064 A EP87103064 A EP 87103064A EP 0238903 B1 EP0238903 B1 EP 0238903B1
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- European Patent Office
- Prior art keywords
- transistor
- current source
- base
- resistor
- collector
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- G—PHYSICS
- G05—CONTROLLING; REGULATING
- G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
- G05F3/00—Non-retroactive systems for regulating electric variables by using an uncontrolled element, or an uncontrolled combination of elements, such element or such combination having self-regulating properties
- G05F3/02—Regulating voltage or current
- G05F3/08—Regulating voltage or current wherein the variable is dc
- G05F3/10—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics
- G05F3/16—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices
- G05F3/20—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices using diode- transistor combinations
- G05F3/30—Regulators using the difference between the base-emitter voltages of two bipolar transistors operating at different current densities
-
- G—PHYSICS
- G05—CONTROLLING; REGULATING
- G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
- G05F3/00—Non-retroactive systems for regulating electric variables by using an uncontrolled element, or an uncontrolled combination of elements, such element or such combination having self-regulating properties
- G05F3/02—Regulating voltage or current
- G05F3/08—Regulating voltage or current wherein the variable is dc
- G05F3/10—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics
- G05F3/16—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices
- G05F3/20—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices using diode- transistor combinations
- G05F3/26—Current mirrors
- G05F3/265—Current mirrors using bipolar transistors only
Definitions
- the invention relates to a reference current source according to the preamble of claim 1.
- the band gap stabilization attributable to RJ Widlar (IEEE Journal of Solid-State Circuits, Vol. SC-6, No. 1, 1971) relates to voltage stabilization. It achieves parameters similar to those of the Zener diode stabilization that was previously used, manages with smaller supply voltages and can advantageously be implemented within a bipolar semiconductor circuit.
- the core of the circuit consists of two transistors, the current densities of which are kept in a certain ratio by a circuit-technical trick.
- the resulting voltage difference between the base emitter diodes is proportional to the absolute temperature. He becomes a resistance supplied, which is arranged at the emitter of the transistor with the lower current density and this results in that the current consumption of the two transistors is proportional to the absolute temperature.
- the invention is based on the object of specifying a circuit for one or more stable output currents which is suitable for bipolar integration, the current or currents being dependent neither on the temperature nor on the supply voltage, the supply voltage passing through a large range can and should also allow small values of the supply voltage.
- Fig. 1 shows the known band gap voltage stabilization in principle.
- Fig. 1a shows the first form of stabilization, which is based on the aforementioned publication by Widlar.
- the second form comes from Ahmed's US PS, also mentioned, it is more independent of component fluctuations and has a higher internal reinforcement.
- the current through R1 also becomes proportional to the absolute temperature. Furthermore, the current through R1 is almost the same as current I2 in the circuit of FIG. 1a, and current I1 in the circuit of FIG. 1b. So the voltage drop across the resistors R2, R3 also becomes proportional to the absolute temperature.
- the compensation effect with regard to the generated voltage Vr is that the voltage drop across R2 which increases with temperature is added to the voltage drop across the emitter base diode of the first transistor Q1 which decreases with temperature.
- the first resistor R1 can also be inserted between the emitter of the second transistor Q2 and the reference point M, the base and the collector of the first transistor Q1 being connected to one another.
- the circuit shown in FIG. 1 with differential amplifier OA and resistors R2, R3 relates to the generation of temperature-stable voltages.
- the design of the amplifier circuit is not important. It is only essential that the ratio of the two currents I1, I2 is maintained regardless of their size and that the voltage difference between the base of transistor Q1 and the collector of transistor Q2 goes to zero.
- This model concept is called "controlled double current source".
- FIG. 3 A preferred embodiment of the controlled dual current source is shown in FIG. 3. It consists of a differential amplifier OA1, the input of which is connected to node A, B and two transistors Q3, Q4 with conductivity that is complementary to that of transistors Q1, Q2.
- the bases of the transistors Q3, Q4 are connected to the output of the differential amplifier OA1.
- the emitters of the transistors Q3, Q4 are optionally connected to a supply voltage Vs via resistors R6, R7.
- the collector of transistor Q3 is at node A and the collector of transistor Q4 is at Node B connected. If one can neglect the input currents of the differential amplifier OA1, the collector currents of the transistors Q3, Q4 are identical to the currents I1, I2 shown in FIG. 2.
- the ratio of the currents I1, I2 is determined by the design of the transistors Q3, Q4.
- the effect of tolerances and the noise contribution of the transistors Q3, Q4 can be reduced by additionally inserted emitter resistors R6, R7.
- 3 shows a further transistor Qp, the base of which is also connected to the output of the differential amplifier OA1 and the emitter of which is also connected to the supply voltage Vs, possibly via an emitter resistor Rp. It adds a third output to the controlled double current source, which carries the same or proportional output current Ir and is used in a consumer symbolically represented as load resistor R1.
- FIG. 4 shows a first embodiment of the differential amplifier OA1 introduced in FIG. 3. It consists of the differential amplifier with transistors Q5, Q6, whose bases are connected to nodes A, B, and whose emitters are connected to the reference point, and a resistor can also be inserted between the emitters and the reference point in order to supply the operating currents influence or reduce a common mode influence.
- the differential stage works on a current mirror from the transistors Q7 and Q8 which are complementary to the transistors Q5 and Q6 and whose emitters are connected to the supply voltage.
- the collector of transistor Q6 is connected to the collector and base of transistor Q8 and the base of transistor Q7, and the connection of the collectors of transistors Q5 and Q7 forms the output of differential amplifier OA1.
- the circuit Fig. 4 also shows the mentioned starting problem when there is no special starting circuit with the transistors Qs1 and Qs2 and the resistors Rs1, Rs2, Rs3. Since the nodes A and B are connected to the reference point via the resistors R4, R5, the base of the transistors Q1, Q2 remains at zero potential even after the supply voltage is switched on and the circuit is de-energized. However, if the resistor R4 is removed, an initial potential can build up at the node A due to residual currents, which leads to a first current in the transistor Q5.
- This current returns to the node A through the current amplification of the transistor Q3 with multiple values and leads to an avalanche-like increase in the total current until, due to the increasing voltage drop across the resistor R1, the current of the transistor Q2 is throttled, the potential at node B increases, the transistor Q6 is live and prevents the further increase in current via the current mirror Q8, Q7, whereby the circuit has entered the desired operating point. It is therefore crucial for this type of start that the temperature compensation can be carried out on one side with the resistor R5.
- FIG. 5 A significantly different embodiment of the differential amplifier OA1 is shown in FIG. 5.
- the potential of nodes A, B is not fed directly to a differential input.
- the mode of operation here is based on the fact that the same operating point is impressed on the transistor Q6 connected to the node B as the transistor Q1, so that the potentials of the nodes A and B must also be identical to one another.
- the current source is provided with transistor Q10, the base of which is connected to the base of the other current source transistors Q3, Q4 and the emitter of the same as with the current source transistors connected to the supply voltage Vs.
- the transistor Q10 determines the current in the transistor Q6 via the connection of the collectors of the transistors Q6, Q10.
- the downstream amplification transistor Q9 forms the output of the amplifier and controls the interconnected bases of the current source transistors.
- This configuration requires three transistors for the OA1 amplifier.
- the transistors Q9 and Q10 form an effective starting circuit of this circuit, so that both compensation resistors R4, R5 may be connected.
- Fig. 6 shows a configuration in which the current source transistors Qn1 ... Qni are of the same conductivity type as the transistors Q1, Q2 of the inner bandgap cell. It is similar to the circuit of FIG. 5 except for a transistor Q11 connected as a diode, which is connected in parallel with the base-emitter path of the other transistor current sources with a corresponding emitter resistor R10. As a result, the diode transistor draws a current that is the same as or proportionally equal to the other current sources. This current must be supplied from transistor Q9 together with the base currents of the current source transistors. Thus the stabilization effect now extends to the current through transistor Q9. Further transistors Qn1 ... Qni, which are arranged analogously to transistor Q9, serve as stabilized output current sources. For the reasons already mentioned, emitter resistors R9, Rn1 ... Rni inserted in the normal case are expedient.
- a starting aid which supplies a starting current which is only slightly dependent on the supply voltage Vs is shown in FIG. 4. It consists of two transistors Qs1, Qs2 and three resistors Rs1, Rs2, Rs3.
- the first transistor Qs1 forms a simple voltage stabilization with the resistors Rs1 and Rs2, in that the first resistor Rs1 is connected between the supply voltage and the base and the second resistor Rs2 is connected between the base and the collector of the transistor Qs1.
- the resistor Rs2 is relatively small compared to Rs1 and is designed so that the collector voltage of the transistor Qs1 changes as little as possible in the intended range of the supply voltage.
- the second transistor Qs2 receives this stabilized collector voltage between the base and the emitter, it being possible for a further shear resistor Rs3 to be connected upstream of the emitter.
- the current developed by transistor Qs2 flows into the bases of current source transistors Q3, Q4.
- the circuit enters the operating state when the current supplied by transistor Qs2 is so large that the amplified current flowing in transistor Q3 generates a sufficient voltage drop across resistor R4 to make transistor Q5 conductive.
- FIG. 5 Another method of starting assistance is shown in FIG. 5.
- a starting transistor Qs is provided, the base of which is connected to the supply voltage Vs via a capacitor Cs, the emitter of which is connected to the reference point and the collector of which is connected to the bases of the current source transistors Q3, Q4.
- the mode of operation is based on the fact that the charging current surge amplifies the transistor Qs when the supply voltage is switched on is passed to the bases of the current source transistors, which thus open the current flow of the circuit. After the capacitor Cs has been charged, Qs is de-energized.
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Description
Die Erfindung betrifft eine Referenzstromquelle gemäß dem Oberbegriff des Anspruchs 1.The invention relates to a reference current source according to the preamble of claim 1.
Während die Stabilisierung von Spannungen viel Aufmerksamkeit gefunden hat, wurde die Stabilisierung von Strömen bisher weniger beachtet. In einer Reihe von Anwendungen, z. B. bei der Versorgung aus Stromquellen innerhalb einer bipolaren integrierten Schaltung und bei gewissen Typen von DA- und AD-Umsetzern, ist aber primär ein stabiler Strom erforderlich. Zwar ist es möglich, stabile Ströme von einer Referenzspannungsquelle abzuleiten. Dies ist aber stets mit Mehraufwand und Genauigkeitsverlust verbunden. Deshalb besteht ein erhebliches technisches Interesse auch für Mittel und Methoden zur Stabilisierung von Strömen.While voltage stabilization has received much attention, less attention has been paid to current stabilization. In a number of applications, e.g. B. in the supply from current sources within a bipolar integrated circuit and with certain types of DA and AD converters, but primarily a stable current is required. It is possible to derive stable currents from a reference voltage source. However, this is always associated with additional effort and loss of accuracy. Therefore, there is considerable technical interest in means and methods for stabilizing currents.
Die auf R.J. Widlar zurückgehende Bandgap-Stabilisierung (IEEE Journal of Solid-State Circuits, Vol. SC-6, No. 1, 1971) betrifft die Spannungsstabilisierung. Sie erreicht ähnlich gute Parameter wie die bis dahin vorwiegend verwendete Zener-Dioden-Stabilisierung, kommt mit kleineren Versorgungsspannungen aus und kann vorteilhaft innerhalb einer bipolaren Halbleiterschaltung implementiert werden. Der Kern der Schaltung besteht aus zwei Transistoren, deren Stromdichten durch einen schaltungstechnischen Kunstgriff in einem bestimmten Verhältnis gehalten werden. Der sich daraus ergebende Spannungsunterschied der Basisemitterdioden ist proportional zur absoluten Temperatur. Er wird einem Widerstand zugeführt, der am Emitter des Transistors mit der kleineren Stromdichte angeordnet ist und dadurch ergibt sich, daß die Stromaufnahme der beiden Transistoren proportional zur absoluten Temperatur wird. In der US-PS 4 059 793 ist aufgezeigt, daß dieser Widerstand auch zwischen Basis und Kollektor des Transistors mit der höheren Stromdichte vorteilhaft angeordnet werden kann. Einen Hinweis, daß innerhalb dieser Grundanordnung ein Strom mit frei einstellbarem Temperaturkoeffizienten erzeugt werden kann, gibt J.E. Hanna in der US-PS 4 091 321. Dies wird dadurch erreicht, daß einem Transistor der Bandgap-Schaltung, der einen zur absoluten Temperatur proportionalen Strom führt, ein Widerstand parallel geschaltet wird. Dieser Widerstand zeigt eine Stromaufnahme proportional zur Basisemitterspannung, die einen negativen Temperaturkoeffizienten besitzt. Die Summe der beiden Ströme besteht somit aus einem temperaturabhängig ansteigenden und einem abfallenden Strom, durch Wichtung kann eine Temperaturunabhängigkeit erreicht werden. Da sich die erwähnte PS mit der Erzeugung temperaturstabiler Spannungen beschäftigt, sind keine Hinweise auf eine Ausnutzung dieses Effektes zur Schaffung temperaturstabiler Stromquellen enthalten.The band gap stabilization attributable to RJ Widlar (IEEE Journal of Solid-State Circuits, Vol. SC-6, No. 1, 1971) relates to voltage stabilization. It achieves parameters similar to those of the Zener diode stabilization that was previously used, manages with smaller supply voltages and can advantageously be implemented within a bipolar semiconductor circuit. The core of the circuit consists of two transistors, the current densities of which are kept in a certain ratio by a circuit-technical trick. The resulting voltage difference between the base emitter diodes is proportional to the absolute temperature. He becomes a resistance supplied, which is arranged at the emitter of the transistor with the lower current density and this results in that the current consumption of the two transistors is proportional to the absolute temperature. US Pat. No. 4,059,793 shows that this resistor can also be advantageously arranged between the base and collector of the transistor with the higher current density. An indication that a current with a freely adjustable temperature coefficient can be generated within this basic arrangement is given by JE Hanna in US Pat. No. 4,091,321. This is achieved in that a transistor of the bandgap circuit which carries a current proportional to the absolute temperature , a resistor is connected in parallel. This resistor shows a current consumption proportional to the base emitter voltage, which has a negative temperature coefficient. The sum of the two currents thus consists of a temperature-dependent rising and a falling current. Weighting enables a temperature independence to be achieved. Since the PS mentioned deals with the generation of temperature-stable voltages, there are no indications that this effect can be exploited to create temperature-stable current sources.
Der Erfindung liegt die Aufgabe zugrunde, eine Schaltung für einen oder mehrere möglichst stabile Ausgangsströme anzugeben, die sich für eine bipolare Integration eignet, wobei der oder die Ströme weder von der Temperatur noch von der Versorgungsspannung abhängig sein sollen, wobei die Versorgungsspannung einen großen Bereich durchlaufen kann und wobei auch kleine Werte der Versorgungsspannung zulässig sein sollen.The invention is based on the object of specifying a circuit for one or more stable output currents which is suitable for bipolar integration, the current or currents being dependent neither on the temperature nor on the supply voltage, the supply voltage passing through a large range can and should also allow small values of the supply voltage.
Diese Aufgabe wird bei einer Referenzstromquelle der eingangs erwähnten Art durch die kennzeichnenden Merkmale des Anspruchs 1 gelöst.This object is achieved in a reference current source of the type mentioned at the outset by the characterizing features of claim 1.
Die Erfindung wird im folgenden anhand von Beispielen erläutert. In den zugehörigen Zeichnungen zeigen:
- Fig. 1
- bekannte Formen der Spannungsstabilisierung,
- Fig. 2
- das Grundprinzip der Stromstabilisierung,
- Fig. 3
- die Ausführung der gesteuerten Stromquellen,
- Fig. 4
- eine erste Verstärkeranordnung,
- Fig. 5
- eine zweite Verstärkeranorndung mit pnp-Stromquellen,
- Fig. 6
- eine Anordnung mit npn-Stromquellen.
- Fig. 1
- known forms of voltage stabilization,
- Fig. 2
- the basic principle of current stabilization,
- Fig. 3
- the execution of the controlled current sources,
- Fig. 4
- a first amplifier arrangement,
- Fig. 5
- a second amplifier arrangement with pnp current sources,
- Fig. 6
- an arrangement with npn power sources.
In Fig. 1 ist die bekannte Bandgap-Spannungsstabilisierung in prinzipieller Form dargestellt. Fig. 1a zeigt die erste Form der Stabilisierung, die sich an die genannte Veröffentlichung von Widlar anlehnt. Die zweite Form entstammt der ebenfalls genannten US-PS von Ahmed, sie ist unabhängiger gegenüber Bauelementschwankungen und hat eine höhere innere Verstärkung.In Fig. 1, the known band gap voltage stabilization is shown in principle. Fig. 1a shows the first form of stabilization, which is based on the aforementioned publication by Widlar. The second form comes from Ahmed's US PS, also mentioned, it is more independent of component fluctuations and has a higher internal reinforcement.
Die an sich bekannte Wirkungsweise dieser Schaltung beruht darauf, daß den beiden Transistoren über die Widerstände R2, R3 Ströme I1, I2 zugeführt werden, die zueinander im umgekehrten Verhältnis dieser Widerstände stehen: I2/I1 = R2/R3. Mittels dieses Stromverhältnisses und weiter mittels des Verhältnisses der Emitterbasisfläche der beiden Transistoren wird ein bestimmtes Verhältnis der Stromdichten der Emitterbasissperrschicht der Transistoren Q1, Q2 festgelegt. In den Schaltungen der Fig. 1 ist angenommen, daß der zweite Transistor Q2 die kleinere Stromdichte erhalten hat. Seine Basis-Emitter-Spannung ist deshalb kleiner. Der Spannungsunterschied wird in beiden Varianten als Spannungsabfall über dem Widerstand R1 wirksam. Da, wie die Beschreibung des bipolaren Transistors zeigt, der Spannungsunterschied proportional zur absoluten Temperatur ist, wird der Strom durch R1 ebenfalls proportional zur absoluten Temperatur. Weiter ist in der Schaltung der Fig. 1a der Strom durch R1 dem Strom I2 nahezu gleich, in der Schaltung der Fig. 1b dem Strom I1. Also wird der Spannungsabfall über den Widerständen R2, R3 ebenfalls proportional zur absoluten Temperatur. Der Kompensationseffekt hinsichtlich der erzeugten Spannung Vr besteht darin, daß der mit der Temperatur zunehmende Spannungsabfall über R2 zu dem mit der Temperatur abnehmenden Spannungsabfall über der Emitterbasisdiode des ersten Transistors Q1 addiert wird.The known mode of operation of this circuit is based on the fact that currents I1, I2 are supplied to the two transistors via the resistors R2, R3, which currents are in the inverse relationship of these resistors: I2 / I1 = R2 / R3. Using this current ratio and further using the ratio of the emitter base area of the two transistors, a specific ratio of the current densities of the emitter base blocking layer of the transistors Q1, Q2 is established. In the circuits of Fig. 1 it is assumed that the second Transistor Q2 has received the lower current density. Its base-emitter voltage is therefore lower. The voltage difference is effective in both variants as a voltage drop across the resistor R1. Since, as the description of the bipolar transistor shows, the voltage difference is proportional to the absolute temperature, the current through R1 also becomes proportional to the absolute temperature. Furthermore, the current through R1 is almost the same as current I2 in the circuit of FIG. 1a, and current I1 in the circuit of FIG. 1b. So the voltage drop across the resistors R2, R3 also becomes proportional to the absolute temperature. The compensation effect with regard to the generated voltage Vr is that the voltage drop across R2 which increases with temperature is added to the voltage drop across the emitter base diode of the first transistor Q1 which decreases with temperature.
Um zu einem von der Temperatur unabhängigen Strom zu kommen, ist nach Fig. 2 vorgesehen, den durch Transistor Q1 und Transistor Q2 fließenden, mit der Temperatur zunehmenden Strömen je einen abnehmenden Strom hinzuzufügen. Dies erfolgt gemäß der Erfindung durch Parallelschaltung von Widerständen R4, R5, da, wie gesagt, der Spannungsabfall über dem Transistor einen negativen Temperaturgang aufweist. Durch geeignete Wahl dieser Widerstände erreicht man, daß der Temperaturkoeffizient der Ströme I1, I2 in Fig. 2 null wird. Es hat sich gezeigt, daß man bei der Wahl der Widerstände nicht auf das Verhältnis der in den Transistoren Q1, Q2 fließenden Ströme Rücksicht nehmen braucht. Es ist also nicht erforderlich, daß der durch den Widerstand R4 fließende Strom zu dem Strom durch den Widerstand R5 im selben Verhältnis steht, wie der durch den Transistor Q1 fließende Strom zu dem durch den Transistor Q2 fließende Strom. Insbesondere ist es möglich, einen der Widerstände R4, R5 wegzulassen und trotzdem den Punkt der Temperaturunabhängigkeit der Ströme I1, I2 einzustellen. Dieser Umstand erleichtert die Ausführung der Verstärkerschaltung besonders hinsichtlich des Startverhaltens.In order to arrive at a current that is independent of the temperature, it is provided according to FIG. 2 to add a decreasing current to the currents flowing through transistor Q1 and transistor Q2 and increasing with the temperature. According to the invention, this is done by connecting resistors R4, R5 in parallel, since, as said, the voltage drop across the transistor has a negative temperature response. A suitable choice of these resistors ensures that the temperature coefficient of the currents I1, I2 in FIG. 2 becomes zero. It has been shown that the ratio of the currents flowing in the transistors Q1, Q2 need not be taken into account when selecting the resistors. It is therefore not necessary that the current flowing through resistor R4 be in the same ratio to the current through resistor R5 as that flowing through transistor Q1 Current to the current flowing through transistor Q2. In particular, it is possible to omit one of the resistors R4, R5 and still set the point of the temperature independence of the currents I1, I2. This fact facilitates the execution of the amplifier circuit particularly with regard to the starting behavior.
Der erste Widerstand R1 kann auch zwischen dem Emitter des zweiten Transistors Q2 und dem Bezugspunkt M eingefügt sein, wobei die Basis und der Kollektor des ersten Transistors Q1 miteinander verbunden sind.The first resistor R1 can also be inserted between the emitter of the second transistor Q2 and the reference point M, the base and the collector of the first transistor Q1 being connected to one another.
Anstelle des Merkmals, daß zwischen der Basis des ersten Transistors Q1 und dem Bezugspunkt M ein Widerstand R4 und zwischen dem Kollektor des zweiten Transistors Q2 und dem Bezugspunkt M ein Widerstand R5 angeschlossen ist, kann auch nur das Merkmal erfüllt sein, daß nur zwischen der Basis des ersten Transistors Q1 und dem Bezugspunkt M ein Widerstand R4 angeschlossen ist oder daß nur zwischen dem Kollektor des zweiten Transistors Q2 und dem Bezugspunkt M ein Widerstand R5 angeschlossen ist.Instead of the feature that a resistor R4 is connected between the base of the first transistor Q1 and the reference point M and a resistor R5 is connected between the collector of the second transistor Q2 and the reference point M, only the feature can be fulfilled that only between the base of the first transistor Q1 and the reference point M, a resistor R4 is connected, or that only between the collector of the second transistor Q2 and the reference point M, a resistor R5 is connected.
Die in Fig. 1 gezeigte Schaltung mit Differenzverstärker OA und Widerständen R2, R3 bezieht sich auf die Erzeugung temperaturstabiler Spannungen. Für die Erreichung der Temperaturkompensation des Stromes kommt es auf die Ausführung der Verstärkerschaltung nicht an. Wesentlich ist nur, daß das Verhältnis der beiden Ströme I1, I2 unabhängig von ihrer Größe gewahrt bleibt und daß die Spannungsdifferenz zwischen Basis vom Transistor Q1 und Kollektor vom Transistor Q2 gegen null geht. Es soll also gelten I1 = Rt1 ` Uab und I2 = Rt2 ` Uab, wobei Uab die Spannung zwischen den Knoten A und B in der Schaltung der Fig. 2 bedeutet und wobei Rt1 und Rt2 Übertragungswiderstände sind, die einen möglichst hohen Wert aufweisen sollen, aber in einem festen Verhältnis zueinander stehen. Diese Modellvorstellung wird mit "gesteuerte Doppelstromquelle" bezeichnet.The circuit shown in FIG. 1 with differential amplifier OA and resistors R2, R3 relates to the generation of temperature-stable voltages. To achieve the temperature compensation of the current, the design of the amplifier circuit is not important. It is only essential that the ratio of the two currents I1, I2 is maintained regardless of their size and that the voltage difference between the base of transistor Q1 and the collector of transistor Q2 goes to zero. I1 = Rt1 `Uab and I2 = Rt2` Uab should apply, where Uab is the voltage between nodes A and B in the circuit of FIG. 2 and where Rt1 and Rt2 are transmission resistances that should have the highest possible value, but have a fixed relationship to each other. This model concept is called "controlled double current source".
Eine bevorzugte Ausführungsform der gesteuerten Doppelstromquelle wird in Fig. 3 gezeigt. Sie besteht aus einem Differenzverstärker OA1, dessen Eingang an den Knoten A, B angeschlossen ist und zwei Transistoren Q3, Q4 mit gegenüber den Transistoren Q1, Q2 komplementärer Leitfähigkeit. Die Basen der Transistoren Q3, Q4 sind mit dem Ausgang des Differenzverstärkers OA1 verbunden. Die Emitter der Transistoren Q3, Q4 sind gegebenenfalls über Widerstände R6, R7 mit einer Versorgungsspannung Vs verbunden. Der Kollektor des Transistors Q3 ist am Knoten A und der Kollektor des Transistors Q4 ist am Knoten B angeschlossen. Wenn man die Eingangsströme des Differenzverstärkers OA1 vernachlässigen kann, sind die Kollektroströme der Transistoren Q3, Q4 mit den in Fig. 2 eingetragenen Strömen I1, I2 identisch. Durch die Ausführung der Transistoren Q3, Q4 wird das Verhältnis der Ströme I1, I2 festgelegt. Dabei kann durch zusätzlich eingefügte Emitterwiderstände R6, R7 der Effekt von Toleranzen sowie der Rauschbeitrag der Transistoren Q3, Q4 reduziert werden. Die Fig. 3 zeigt einen weiteren Transistor Qp, dessen Basis ebenfalls mit dem Ausgang des Differenzverstärkers OA1 verbunden ist und dessen Emitter ebenfalls, gegebenenfalls über einen Emitterwiderstand Rp, mit der Versorgungsspannung Vs verbunden ist. Er fügt der gesteuerten Doppelstromquelle einen dritten Ausgang hinzu, der den gleichen oder verhältnisgleichen Ausgangsstrom Ir führt und in einem symbolisch als Lastwiderstand R1 dargestellten Verbraucher genutzt wird.A preferred embodiment of the controlled dual current source is shown in FIG. 3. It consists of a differential amplifier OA1, the input of which is connected to node A, B and two transistors Q3, Q4 with conductivity that is complementary to that of transistors Q1, Q2. The bases of the transistors Q3, Q4 are connected to the output of the differential amplifier OA1. The emitters of the transistors Q3, Q4 are optionally connected to a supply voltage Vs via resistors R6, R7. The collector of transistor Q3 is at node A and the collector of transistor Q4 is at Node B connected. If one can neglect the input currents of the differential amplifier OA1, the collector currents of the transistors Q3, Q4 are identical to the currents I1, I2 shown in FIG. 2. The ratio of the currents I1, I2 is determined by the design of the transistors Q3, Q4. The effect of tolerances and the noise contribution of the transistors Q3, Q4 can be reduced by additionally inserted emitter resistors R6, R7. 3 shows a further transistor Qp, the base of which is also connected to the output of the differential amplifier OA1 and the emitter of which is also connected to the supply voltage Vs, possibly via an emitter resistor Rp. It adds a third output to the controlled double current source, which carries the same or proportional output current Ir and is used in a consumer symbolically represented as load resistor R1.
In Fig. 4 ist eine erste Ausführungsform des in Fig. 3 eingeführten Differenzverstärkers OA1 dargestellt. Sie besteht aus dem Differenzverstärker mit den Transistoren Q5, Q6, deren Basen an den Knoten A, B, angeschlossen sind und deren Emitter mit dem Bezugspunkt verbunden sind, wobei zwischen den Emittern und dem Bezugspunkt auch ein Widerstand eingefügt sein kann, um die Arbeitsströme zu beeinflussen oder einen Gleichtakteinfluß zu vermindern. Die Differenzstufe arbeitet auf einen Stromspiegel aus den zu den Transistoren Q5 und Q6 komplementären Transistoren Q7 und Q8, deren Emitter an der Versorgungsspannung angeschlossen sind. Dabei ist der Kollektor des Transistors Q6 mit Kollektor und Basis des Transistors Q8 und der Basis des Tranistors Q7 verbunden und die Verbindung der Kollektoren der Transistoren Q5 und Q7 bildet den Ausgang des Differenzverstärkers OA1.FIG. 4 shows a first embodiment of the differential amplifier OA1 introduced in FIG. 3. It consists of the differential amplifier with transistors Q5, Q6, whose bases are connected to nodes A, B, and whose emitters are connected to the reference point, and a resistor can also be inserted between the emitters and the reference point in order to supply the operating currents influence or reduce a common mode influence. The differential stage works on a current mirror from the transistors Q7 and Q8 which are complementary to the transistors Q5 and Q6 and whose emitters are connected to the supply voltage. The collector of transistor Q6 is connected to the collector and base of transistor Q8 and the base of transistor Q7, and the connection of the collectors of transistors Q5 and Q7 forms the output of differential amplifier OA1.
Die Schaltung Fig. 4 zeigt auch das erwähnte Startproblem, wenn keine spezielle Startschaltung mit den Transistoren Qs1 und Qs2 und den Widerständen Rs1, Rs2, Rs3 vorhanden ist. Da die Knoten A und B über die Widerstände R4, R5 mit dem Bezugspunkt verbunden sind, bleibt die Basis der Transistoren Q1, Q2 auch nach dem Einschalten der Versorgungsspannung auf Nullpotential und die Schaltung stromlos. Entfernt man jedoch den Widerstand R4, so kann sich am Knoten A durch Restströme ein Anfangspotential aufbauen, das zu einem ersten Strom im Transistor Q5 führt. Dieser Strom kehrt durch die Stromverstärkung des Transistors Q3 mit mehrfachem Wert zum Knoten A zurück und führt zum lawinenartigen Anwachsen des Gesamtstromes, bis infolge zunehmenden Spannungsabfalls am Widerstand R1 der Strom des Transistors Q2 gedrosselt wird, das Potential am Knoten B ansteigt, der Transistor Q6 stromführend wird und über den Stromspiegel Q8, Q7 die weitere Stromzunahme verhindert, womit die Schaltung in den erwünschten Arbeitspunkt eingetreten ist. Für diese Art des Starts ist also entscheidend, daß die Temperaturkompensation einseitig mit dem Widerstand R5 ausgeführt werden kann.The circuit Fig. 4 also shows the mentioned starting problem when there is no special starting circuit with the transistors Qs1 and Qs2 and the resistors Rs1, Rs2, Rs3. Since the nodes A and B are connected to the reference point via the resistors R4, R5, the base of the transistors Q1, Q2 remains at zero potential even after the supply voltage is switched on and the circuit is de-energized. However, if the resistor R4 is removed, an initial potential can build up at the node A due to residual currents, which leads to a first current in the transistor Q5. This current returns to the node A through the current amplification of the transistor Q3 with multiple values and leads to an avalanche-like increase in the total current until, due to the increasing voltage drop across the resistor R1, the current of the transistor Q2 is throttled, the potential at node B increases, the transistor Q6 is live and prevents the further increase in current via the current mirror Q8, Q7, whereby the circuit has entered the desired operating point. It is therefore crucial for this type of start that the temperature compensation can be carried out on one side with the resistor R5.
Eine wesentlich andere Ausführung des Differenzverstärkers OA1 ist in Fig. 5 dargestellt. Bei ihr wird das Potential der Knoten A, B nicht direkt einem Differenzeingang zugeführt. Die Wirkungsweise beruht hier darauf, daß dem am Knoten B angeschlossenen Transistor Q6 der gleiche Arbeitspunkt aufgeprägt wird wie dem Transistor Q1, so daß auch die Potentiale der Knoten A und B untereinander gleich werden müssen. Zu diesem Zweck ist die Stromquelle mit dem Transistor Q10 vorgesehen, dessen Basis mit der Basis der übrigen Stromquellentransistoren Q3, Q4 verbunden ist und dessen Emitter ebenfalls wie bei den Stromquellentransistoren mit der Versorgungsspannung Vs verbunden ist. Über die Verbindung der Kollektoren der Transistoren Q6, Q10 bestimmt der Transistor Q10 den Strom im Transistor Q6. Der nachgeschaltete Verstärkungstransistor Q9 bildet den Ausgang des Verstärkers und steuert die miteinander verbundenen Basen der Stromquellentransistoren. In dieser Konfiguration kommt man mit drei Transistoren für den Verstärker OA1 aus. Weiterhin ist es ohne Nachteile möglich, auch eine größer Anzahl Transistoren Qp1 ... Qpi als Ausgangsstromquellen vorzusehen, da die hohe Schleifenverstärkung über die Transistoren Q6, Q9 eine größere Belastung zuläßt. Die Transistoren Q9 und Q10 bilden einen wirksamen Startkreis dieser Schaltung, so daß beide Kompensationswiderstände R4, R5 angeschlossen sein dürfen.A significantly different embodiment of the differential amplifier OA1 is shown in FIG. 5. The potential of nodes A, B is not fed directly to a differential input. The mode of operation here is based on the fact that the same operating point is impressed on the transistor Q6 connected to the node B as the transistor Q1, so that the potentials of the nodes A and B must also be identical to one another. For this purpose, the current source is provided with transistor Q10, the base of which is connected to the base of the other current source transistors Q3, Q4 and the emitter of the same as with the current source transistors connected to the supply voltage Vs. The transistor Q10 determines the current in the transistor Q6 via the connection of the collectors of the transistors Q6, Q10. The downstream amplification transistor Q9 forms the output of the amplifier and controls the interconnected bases of the current source transistors. This configuration requires three transistors for the OA1 amplifier. Furthermore, it is possible without disadvantages to also provide a larger number of transistors Qp1 ... Qpi as output current sources, since the high loop gain via the transistors Q6, Q9 permits a greater load. The transistors Q9 and Q10 form an effective starting circuit of this circuit, so that both compensation resistors R4, R5 may be connected.
Schließlich zeigt Fig. 6 eine Konfiguration, bei der die Stromquellentransistoren Qn1 ... Qni vom gleichen Leitfähigkeitstyp sind wie die Transistoren Q1, Q2 der inneren Bandgap-Zelle. Sie gleicht der Schaltung von Fig. 5 bis auf einen als Diode geschalteten Transistor Q11, der der Basis-Emitter-Strecke der übrigen Transistorstromquellen mit einem entsprechenden Emitterwiderstand R10 parallel geschaltet ist. Der Diodentransistor nimmt infolgedessen einen zu den übrigen Stromquellen gleichen oder verhältnisgleichen Strom auf. Vom Transistor Q9 muß dieser Strom zusammen mit den Basisströmen der Stromquellentransistoren zugeführt werden. Somit erstreckt sich der Stabilisierungseffekt nunmehr auch auf den Strom durch Transistor Q9. Weitere, zum Transistor Q9 analog angeordnete Transistoren Qn1 ... Qni dienen als stabilisierte Ausgangsstromquellen. Aus den schon erwähnten Gründen sind im Normalfall eingefügte Emitterwiderstände R9, Rn1 ... Rni zweckmäßig.Finally, Fig. 6 shows a configuration in which the current source transistors Qn1 ... Qni are of the same conductivity type as the transistors Q1, Q2 of the inner bandgap cell. It is similar to the circuit of FIG. 5 except for a transistor Q11 connected as a diode, which is connected in parallel with the base-emitter path of the other transistor current sources with a corresponding emitter resistor R10. As a result, the diode transistor draws a current that is the same as or proportionally equal to the other current sources. This current must be supplied from transistor Q9 together with the base currents of the current source transistors. Thus the stabilization effect now extends to the current through transistor Q9. Further transistors Qn1 ... Qni, which are arranged analogously to transistor Q9, serve as stabilized output current sources. For the reasons already mentioned, emitter resistors R9, Rn1 ... Rni inserted in the normal case are expedient.
In Fig. 4 und Fig. 5 sind noch Maßnahmen zur Absicherung eines zuverlässigen Schaltungsstarts dargestellt. Eine Starthilfe, die einen Startstrom liefert, der nur wenig von der Versorgungsspannung Vs abhängt, zeigt Fig. 4. Sie besteht aus zwei Transistoren Qs1, Qs2 und drei Widerständen Rs1, Rs2, Rs3. Der erste Transistor Qs1 bildet mit den Widerständen Rs1 und Rs2 eine einfache Spannungsstabilisierung, indem der erste Widerstand Rs1 zwischen Versorgungsspannung und Basis und der zweite Widerstand Rs2 zwischen Basis und Kollektor des Transistors Qs1 angeschlossen ist. Der Widerstand Rs2 ist verhältnismäßig klein gegenüber Rs1 und wird so ausgelegt, daß sich die Kollektorspannung des Transistors Qs1 im vorgesehenen Bereich der Versorgungsspannung möglichst wenig ändert. Der zweite Transistor Qs2 empfängt diese stabilisierte Kollektorspannung zwischen Basis und Emitter, wobei vor dem Emitter noch ein weiterer Scherungswiderstand Rs3 geschaltet sein kann. Der vom Transistor Qs2 entwickelte Strom fließt in die Basen der Stromquellentransistoren Q3, Q4. Die Schaltung tritt in den Betriebszustand ein, wenn der vom Transistor Qs2 gelieferte Strom so groß ist, daß der im Transistor Q3 fließende, verstärkte Strom einen ausreichenden Spannungsabfall über dem Widerstand R4 erzeugt, um den Transistor Q5 leitend zu machen.In FIG. 4 and FIG. 5, measures for securing a reliable circuit start are also shown. A starting aid which supplies a starting current which is only slightly dependent on the supply voltage Vs is shown in FIG. 4. It consists of two transistors Qs1, Qs2 and three resistors Rs1, Rs2, Rs3. The first transistor Qs1 forms a simple voltage stabilization with the resistors Rs1 and Rs2, in that the first resistor Rs1 is connected between the supply voltage and the base and the second resistor Rs2 is connected between the base and the collector of the transistor Qs1. The resistor Rs2 is relatively small compared to Rs1 and is designed so that the collector voltage of the transistor Qs1 changes as little as possible in the intended range of the supply voltage. The second transistor Qs2 receives this stabilized collector voltage between the base and the emitter, it being possible for a further shear resistor Rs3 to be connected upstream of the emitter. The current developed by transistor Qs2 flows into the bases of current source transistors Q3, Q4. The circuit enters the operating state when the current supplied by transistor Qs2 is so large that the amplified current flowing in transistor Q3 generates a sufficient voltage drop across resistor R4 to make transistor Q5 conductive.
Eine weitere Methode der Starthilfe ist in Fig. 5 dargestellt. Dabei ist ein Starttransistor Qs vorgesehen, dessen Basis über einen Kondensator Cs mit der Versorgungsspannung Vs, dessen Emitter mit dem Bezugspunkt und dessen Kollektor mit den Basen der Stromquellentransistoren Q3, Q4 verbunden ist. Die Wirkungsweise beruht darauf, daß der Ladestromstoß bei Einschalten der Versorgungsspannung vom Transistor Qs verstärkt auf die Basen der Stromquellentransistoren geleitet wird, die damit den Stromfluß der Schaltung eröffnen. Nach der Aufladung des Kondensators Cs wird Qs stromlos.Another method of starting assistance is shown in FIG. 5. A starting transistor Qs is provided, the base of which is connected to the supply voltage Vs via a capacitor Cs, the emitter of which is connected to the reference point and the collector of which is connected to the bases of the current source transistors Q3, Q4. The mode of operation is based on the fact that the charging current surge amplifies the transistor Qs when the supply voltage is switched on is passed to the bases of the current source transistors, which thus open the current flow of the circuit. After the capacitor Cs has been charged, Qs is de-energized.
Die stationäre Zündschaltung nach Fig. 4 hält den Arbeitspunkt der Stabilisierungsschaltung in allen Betriebszuständen aufrecht, benötigt aber einen Zusatzstrom. Die dynamische Zündschaltung nach Fig. 5 benötigt keinen Betriebsstrom. Kommt es jedoch bei angelegter Spannung aus irgendeinem Grunde zum Abbruch des Stromflusses, so bleibt die Schaltung im Aus-Zustand.4 maintains the operating point of the stabilization circuit in all operating states, but requires an additional current. 5 does not require any operating current. However, if the current flow is interrupted for any reason when the voltage is applied, the circuit remains in the off state.
In allen Schaltungen Fig. 3 bis Fig. 6 sind nicht mehr als zwei Transistorsysteme galvanisch in Reihe geschaltet. Das bedeutet, daß bei Verwendung von Silizium-Transistoren etwa 1 V Betriebsspannung für die Funktionsfähigkeit ausreicht.In all of the circuits in FIGS. 3 to 6, no more than two transistor systems are electrically connected in series. This means that when using silicon transistors, about 1 V operating voltage is sufficient for the functionality.
Claims (14)
- A reference current source with two transistors and a controlled double current source (CDCS), characterised in that the base of the second transistor (Q2) is connected to the collector of the first transistor (Q1), that the emitter of the first transistor (Q1) is connected to a datum point (M), that the first connection of the controlled double current source (CDCS) is connected to the base of the first transistor (Q1), that the second connection of the controlled double current source (CDCS) is connected to the collector of the second transistor (Q2), that either a first resistor (R1) is inserted between the base and the collector of the first transistor (Q1) and the emitter of the second transistor (Q2) is connected to the datum point (M) or that the first resistor (R1) is inserted between the emitter of the second transistor (Q2) and the datum point (M) and the base and the collector of the first transistor (Q1) are connected together and that between the base of the first transistor (Q1) and the datum point (M) a resistor (R4) is attached and/or between the collector of the second transistor (Q2) and the datum point (M) a resistor (R5) is attached.
- A reference current source according to claim 1, characterised in that the resistor (R4) value between the base of the first transistor (Q1) and the datum point (M) and/or the resistor (R5) value between the collector of the second transistor (Q2) and the datum point (M) is such that the currents (I1, I2) of the controlled double current source (CDCS) depend as little as possible on the ambient temperature.
- A reference current source according to claim 2, characterised in that in the predetermined range of the ambient temperature, there exists at least one value of the temperature close to which the temperature dependency of the currents (I1, I2) of the double current source disappears.
- A reference current source according to any one of claims 1 to 3, characterised in that the double current source consists of two transistors (Q3, Q4) which are of a conduction type which is complementary to the first and second transistors (Q1, Q2), the emitters of which are connected directly or via resistors (R6, R7 to a supply voltage (Vs) and the bases of which are connected to the output of an amplifier arrangement (OA1).
- A reference current source according to claim 4, characterised in that the amplifier arrangement (OA1) is a differential amplifier, the first input of which is connected to the base of the first transistor (Q1) and the second input of which is connected to the collector of the second transistor (Q2).
- A reference current source according to any one of claims 1 to 5, characterised in that the amplifier arrangement (OA1) consists of a differential stage comprising two transistors (Q5, Q6), which operates on a current mirror comprising two transistors (Q7, Q8) of complementary conductivity, in which the bases of the differential stage form the inputs, the emitters of the differential stage are connected via a resistor directly to the datum point, the outputs of the differential stage are connected to the input and output of the current mirror and the output of the current mirror forms the output of the amplifier arrangement (OA1).
- A reference current source according to any one of claims 1 to 6, characterised in that the amplifier arrangement (OA1) has an input transistor (Q6), the base of which is attached to the collector of the second transistor (Q2) and the emitter of which is connected to the datum point (M).
- A reference current source according to any one of claims 1 to 7, characterised in that the collector of the input transistor (Q6) is connected to the collector of a transistor connected as a current source, in which the base is connected to the bases of the transistors (Q3, Q4) and the emitter is connected directly or via a resistor (R8) to the supply voltage (Vs).
- A reference current source according to any one of claims 1 to 8, characterised in that the base of an output transistor (Q9) is connected to the collector of the input transistor (Q6), the emitter of the output transistor is connected to the reference point in some cases via a resistor (R9) and the collector of the output transistor forms the output of the amplifier arrangement (OA1) and is connected to the base of the current source transistors (Q3, Q4, Q10).
- A reference current source according to any one of claims 1 to 9, characterised in that the base of the current source transistors (Q3, Q4, Q10) is connected via a transistor (Q11) which is connected as a diode directly or via a resistor (R10) to the supply voltage (Vs).
- A reference current source according to any one of claims 1 to 10, characterised in that at least one further transistor (Qp1), which serves as an output current source, is connected, in which its base is connected to the base of the current source transistors (Q3, Q4) and its emitter is connected directly or via a resistor (Rp1) to a connection of the supply voltage (Vs).
- A reference current source according to claim 10, characterised in that at least one further transistor (Qn1) which serves as an output current source, is connected, in which its base is connected to the base of the output transistor (Q9) and its emitter is connected directly or via a resistor (Rn1) to the datum point (M).
- A reference current source according to any one of claims 1 to 12, characterised in that the base of the current source transistors (Q3, Q4) is connected to the collector of a starter transistor (Qs), its emitter being connected to the reference point and its base being connected via a capacitor (Cs) to the supply voltage (Vs).
- A reference current source according to any one of claims 1 to 11, characterised in that the collector of a second starter transistor (Qs2) is connected to the base of the current source transistors (Q3, Q4), that the base of the second starter transistor (Qs2) is connected to the collector of a first starter transistor (Qs1), that a series resistor (Rs1) leads from the supply voltage (Vs) to the base of the first starter transistor (Qs1), that a further resistor (Rs2) is attached to the base and to the collector of the first starter transistor (Qs1) and that the emitter of the first starter transistor (Qs1) is connected to the reference point and the emitter of the second starter transistor (Qs2) is connected directly or via a resistor (Rs3) also to the reference point.
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
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DE3610158 | 1986-03-26 | ||
DE19863610158 DE3610158A1 (en) | 1986-03-26 | 1986-03-26 | REFERENCE POWER SOURCE |
Publications (2)
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EP0238903A1 EP0238903A1 (en) | 1987-09-30 |
EP0238903B1 true EP0238903B1 (en) | 1991-05-08 |
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EP87103064A Expired - Lifetime EP0238903B1 (en) | 1986-03-26 | 1987-03-04 | Reference current source |
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US (1) | US4785231A (en) |
EP (1) | EP0238903B1 (en) |
DE (1) | DE3610158A1 (en) |
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IT1246598B (en) * | 1991-04-12 | 1994-11-24 | Sgs Thomson Microelectronics | BAND-GAP CHAMPIONSHIP VOLTAGE REFERENCE CIRCUIT |
IT1252324B (en) * | 1991-07-18 | 1995-06-08 | Sgs Thomson Microelectronics | HIGH STABILITY VOLTAGE REGULATOR INTEGRATED CIRCUIT AND LOW CURRENT CONSUMPTION. |
GB2262675A (en) * | 1991-12-20 | 1993-06-23 | Codex Corp | Comparator start-up arrangement |
DE4344447B4 (en) * | 1993-12-24 | 2009-04-02 | Atmel Germany Gmbh | Constant current source |
DE19530737A1 (en) * | 1995-08-22 | 1997-02-27 | Philips Patentverwaltung | Circuit arrangement for supplying a constant current |
DE19609831A1 (en) * | 1996-03-13 | 1997-09-18 | Philips Patentverwaltung | Circuit arrangement for supplying a direct current |
DE10033933B4 (en) * | 2000-07-05 | 2005-12-01 | Samsung SDI Co., Ltd., Suwon | Constant current source for providing small currents and multi-channel current source |
US6433621B1 (en) * | 2001-04-09 | 2002-08-13 | National Semiconductor Corporation | Bias current source with high power supply rejection |
DE10231175B4 (en) * | 2002-07-10 | 2004-08-12 | Infineon Technologies Ag | Temperature stable power source arrangement |
US11714444B2 (en) * | 2021-10-18 | 2023-08-01 | Texas Instruments Incorporated | Bandgap current reference |
Family Cites Families (11)
Publication number | Priority date | Publication date | Assignee | Title |
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US4059793A (en) * | 1976-08-16 | 1977-11-22 | Rca Corporation | Semiconductor circuits for generating reference potentials with predictable temperature coefficients |
US4091321A (en) * | 1976-12-08 | 1978-05-23 | Motorola Inc. | Low voltage reference |
JPS5659321A (en) * | 1979-08-09 | 1981-05-22 | Toshiba Corp | Constant-current regulated power circuit |
US4350904A (en) * | 1980-09-22 | 1982-09-21 | Bell Telephone Laboratories, Incorporated | Current source with modified temperature coefficient |
NL8103813A (en) * | 1981-08-14 | 1983-03-01 | Philips Nv | CURRENT STABILIZATION CIRCUIT. |
JPS5866129A (en) * | 1981-10-15 | 1983-04-20 | Toshiba Corp | Constant current source circuit |
US4399399A (en) * | 1981-12-21 | 1983-08-16 | Motorola, Inc. | Precision current source |
US4490670A (en) * | 1982-10-25 | 1984-12-25 | Advanced Micro Devices, Inc. | Voltage generator |
NL8301138A (en) * | 1983-03-31 | 1984-10-16 | Philips Nv | POWER SOURCE SWITCH. |
DE3321556A1 (en) * | 1983-06-15 | 1984-12-20 | Telefunken electronic GmbH, 7100 Heilbronn | BANDGAP SWITCHING |
NL8302458A (en) * | 1983-07-11 | 1985-02-01 | Philips Nv | CURRENT STABILIZATION CIRCUIT. |
-
1986
- 1986-03-26 DE DE19863610158 patent/DE3610158A1/en active Granted
-
1987
- 1987-03-04 EP EP87103064A patent/EP0238903B1/en not_active Expired - Lifetime
- 1987-03-25 US US07/029,908 patent/US4785231A/en not_active Expired - Lifetime
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DE3610158C2 (en) | 1990-01-25 |
DE3610158A1 (en) | 1987-10-01 |
US4785231A (en) | 1988-11-15 |
EP0238903A1 (en) | 1987-09-30 |
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