EA019047B1 - Method for controlling pulse voltage stabiliser - Google Patents

Abstract

The invention relates to electrical engineering, in particular, to converting technology and can be used in designing pulse stabilizers of direct voltage (PVS) with a power circuit of a step-up type and a discrete processing of information signals, having short duration of transient processes and a static error of outlet voltage stabilization. For which the measurement at the outlet Uand the inlet Uof a PVS, throttle Iand load Icurrent of the PVS is carried out at each period of transformation in fixed moments of time (mT+τ). The measured voltage values Uand U, currents Iand Iare memorized. A signal "capacitor current" I=I-I∙U/Uis calculated, as well as a mismatch signal ε=U-U, where U- reference voltage. From the value of the signal "capacitor current" I(mT+τ) the value of "capacitor current" signal is subtracted as I((m-1)T+τ), memorised at the previous stage, and the value of "capacitor current" signal is memorized as produced at the current stage of transformation. The mismatch signal ε is increased Ktimes, and its finite-different summation is carried out. The "capacitor current" signal is exposed to frequency correction, and summed up with a mismatch signal is sent to an inlet of a pulse-width modulator.

Description

The invention relates to the field of electrical engineering, in particular to converter technology, and can be used in the construction of pulsed DC voltage stabilizers (ISN) with a boost type power circuit and discrete processing of information signals having a short duration of transient processes and a static error in stabilizing the output voltage. Known [1] is a control method for an ISN made in the form of a controlled electric switch (UEC) and an IS filter (inductive-capacitive filter with a closing diode), connected in series between the input and output of the ISN, which consists in measuring the current 1 _{C of the} PVS capacitor -filter and _O and the voltage at the output of SRI, and the voltage calculated _{C. n} IS on capacitance capacitor filter by integrating a signal obtained by summing the current of the capacitor taken with a coefficient 1 / C, where C - PS-capacitor filter and the error signal voltage and _{C. n} from and _out , taken with a coefficient of K ₀ = (0.05-0.0005) / (K _s -C), where K _C is the internal active resistance of the capacitor of the PSA filter, a voltage mismatch signal ε is generated by comparing the voltage and _{S n} with a reference voltage and. ,,, multiply the current 1 _C and the error signal ε by the coefficients K _p and K _{and 1,} respectively, and

where L - inductance OS-filter and _Rin - input voltage SRI, 1 _and - static control impulse UEC, T - conversion period and _n - rate of change of the sawtooth voltage pulse width modulator (PWM) generating a control signal by summing the signals obtained as a result of multiplication and the control signal form the UEC control pulses according to the PWM principle with the modulator locked at the moment of formation of the modulated pulse front.

This method provides a power circuit SRI down type (SRI PN) short duration transients effects on the output (load current _and 1) and a terminal (the voltage change at the input _Rin and IOS).

The disadvantages of this method include the fact that it does not provide high static accuracy of voltage and _output stabilization, since the voltage mismatch signal ε is multiplied by the coefficient K _and1 , the value of which, determined through the SPI parameters, is relatively small and cannot be increased without deteriorating dynamic characteristics ISN. In addition, this method is not applicable directly to stabilize the output voltage of an ISN with a boost type power circuit (ISN PV) containing a inductor and a diode connected in series between the input and output of the stabilizer, a controlled electrical switch connected between the common wire of the stabilizer and the junction point between the inductor and the diode , and a capacitor connected between the output and the common wire of the stabilizer.

There is a known [2] control method for the ISN PV, which consists in measuring the output voltage in the interval of the on state of the UEC, storing the output voltage at the end of the interval of the on state of the UEC, receiving a voltage mismatch signal by comparing the stored value of the output voltage and the reference voltage and generating the mismatch signal according to voltage pulse control UEC on the principle of PWM. This method allows you to provide ISN PV high static accuracy of stabilization of the output voltage, but does not provide a short duration of transients.

Known control method SRI Mo [3], according to which the measured current of 1 _{C (1)} capacitor ISfiltra and voltage and _O at the output of stabilizer obtained error signal voltage ε (ΐ), subtracting the reference voltage and the _floor of the AND _O (1) receive the first signal by multiplying the voltage mismatch signal ε (ΐ) by the coefficient K _p <1 / (K _s C + T), where K _C and C, respectively, the internal resistance and capacitance of the PSA filter capacitor, receive the signal capacitor current, counting its equal to the current 1 _C (1) of the capacitor of the PSA filter, multiply the signal t ok capacitor by a factor of 1 / C and, summing with the first signal, receive the second signal, integrate the second signal, and the range of possible changes in the integral of the second signal is limited by the values of its maximum deviations in dynamic modes that do not lead to interruption of the modulation, receive the third signal, counting equal to the integral of the second signal, a control signal is obtained by multiplying the signal by the capacitor current and the third signal by the coefficients K _p and K _{and 1,} respectively, and summing the inverse values of the signals obtained as a result of At the same time as multiplication, UEC control pulses are generated by the control signal according to the PWM principle with the modulator locked at the moment of formation of the modulated pulse front. In this case, Кп К _и1 correspond to Кп and К _и1 from [1].

This method allows you to provide ISN PN small duration of transients in dynamic modes of operation and a small amount of static error of the output voltage. However, when it is used to control ISN PV, a short duration of transients in dynamic operating modes is not achieved due to the difference in power circuits of ISN PN and ISN PV.

- 1 019047

In the well-known ISN PVs, as a rule, they solve the problem of reducing either the magnitude of the static error of the output voltage or the duration of transients. However, there are areas of technology in which both the short duration of transients and the small static error of the output voltage are required from the ISN PV. For example, such requirements for ISS PV are presented when they are used in power supply systems of spacecraft. It is quite difficult to provide a short duration of transient processes and a small static error of the output voltage in the SID, since an increase in the gain of the voltage mismatch signal or the integration of this signal can reduce the static error of the output voltage, but, as a rule, leads to an increase in the duration of transient processes in dynamic modes work of ISN. In the known solution [3], both a short duration of transient processes and a small static error of the output voltage are provided. However, this solution is applicable to the ISN PN and, when used in the ISN PV, does not provide a short duration of transient processes due to the difference in the power circuits of the ISN.

As a prototype, a control method for an ISN with a PWM [4] was selected, according to which, in a switching regulator containing a choke with inductance b and a diode, connected in series between the input and output of the stabilizer, a controlled electrical switch connected between the common wire of the stabilizer and the junction point of the inductor and diode , a capacitor with capacitance C, connected between the output and the common wire stabilizer, and the measured voltage _O at the output of the stabilizer is formed by the voltage error signal by subtracting the reference voltage ix from the voltage at the output of the stabilizer produces a first signal by multiplying the error signal voltage by a factor K _p, integrating the second signal by interrupting integration limit the range of the integral of the second signal from the values of its maximum deviations in the dynamic conditions that do not lead to a modulation of interruption is obtained the total signal, multiplying the current signal of the capacitor and the third signal by the coefficients K | and K respectively and summing the inverted values of the signals resulting from the multiplication control signal is formed pulses controls the electric wrench according to the principle of pulse width modulation modulator blocking at the time of forming the modulated wavefront, the invention further measure the input voltage and _Rin, currents throttle 1 _b and a load _1N, calculated signal current of the capacitor _C 1 _s = 1 s _n -1 _EXIT / _Rin and receive a third signal by integrating a fourth signal; calculated by subtracting of the signal of the third capacitor current signal taken from a coefficient K ₀ << 1 / T where T - conversion period, the second signal is determined to be the first one, the resultant signal obtained by subtracting the integral of the second signal from the sum signal, and in the case of the front pulse edge modulation by controlling the controlled electric key, the control signal is obtained by inverting the resulting signal, and in the case of modulation of the trailing edge of the control pulses of the controlled electric key, the control signal is taken equal to the result the signal, while K ₁ = 2,4-b-and _l (T) / (and _output -T); K ^ K / T; K _p <4 _K-1 and _C-out / _[Rin and T (2B _S S + T)], where Bc - internal resistance of the capacitor, form a non-linear voltage reference modulator and _l (1) = 2,4 and _l (T) - (1 ² /4-T ² +1 ³ /6-T ³ ), where 0 <1 <T. and _l (T) is the amplitude value of the nonlinear reference voltage of the modulator.

In the well-known solution [4], both a short duration of transient processes and a small static error of the output voltage as applied to the ISN PV are provided. At the same time, the implementation of such a solution on analog elements is difficult both in design and in production. Structural schemes of specialized integrated circuits manufactured by the industry differ significantly from the structure of the control device that implements the solution [4]. The hardware implementation of the control law on such microcircuits causes the use of a large number of external elements, which entails the complication and decrease in the reliability of the control device. Therefore, there is a need to develop new solutions that allow the transition to a microelectronic element base while maintaining a short duration of transient processes and the static error of the output voltage of the ISN.

The basis of the invention is the task of improving the quality of the output voltage of the ISN PV in dynamic and static modes of its operation with discrete processing of its information signals, which allows for the practical implementation of the ISN with a control device based on microcontroller technology. Moreover, by improving the quality of the output voltage in dynamic modes is meant a reduction in the duration of transient processes, and in static modes - ensuring the astatism of the output voltage.

The problem is solved in that in the method of controlling a pulse stabilizer containing a inductor with an inductance b and a diode connected in series between the input and output of the stabilizer, a controlled electrical switch connected between the common wire of the stabilizer and the junction point of the inductor and diode, a capacitor with a capacitance C included between the output and the common wire stabilizer, consisting in the fact that the measured voltage and the output _O and the inlet and _Rin stabilizer current throttle _b 1 and a load _1N, calculating a current signal capacitor 1 _s = 1 _b -1 _n- th _output / and _input ,

- 2 019047 generate a voltage mismatch signal by subtracting the reference voltage from the voltage at the output of the stabilizer, calculate the first signal by multiplying the voltage mismatch signal by the coefficient K _p , where K _p <4-K! -C-and _out / [and _in - T (2 _C-B + C T)] K _C - internal resistance of the capacitor, the control signal form the control pulses of controlled electric wrench according to the principle of pulse width modulation according to the invention the measurement of stresses and _O and _Rin and the outlet and inlet of the stabilizer throttle currents 1 _b and the narrow 1 _n each transform at each time period at fixed time points (tT + τ), where t = 0, 1, 2, 3, ..., τ - in the general case, an arbitrarily chosen fixed time interval | τ | <Τ, the same remote from the start of each conversion period T, produce storing measured values of voltages and _O and and _Rin at the outlet and inlet stabilizer choke currents 1 |, and a load _1N, calculating the first signal K _p e (mT + t) produced at time instants ( tT + τ), the second signal is obtained by subtracting the capacitor current 1 _С (tT + t) from the signal value If the capacitor current is 1 _C ((t-1) T + t), stored in the previous period, the signal value of the capacitor current obtained in the current conversion period is stored, the third signal is calculated, and ₃ , by multiplying the second signal by the coefficient K ₁ = b- and _l / (and _output -T), where and _{l are the} amplitude value of the sawtooth linear reference signal of the pulse-width modulator, the value of the third signal is stored, the fourth signal is calculated υ ₄ (тТ + τ) = 2υ ₃ (тТ + τ) -υ ₃ ((т1) Т + т), the fourth signal is stored, the fifth signal is calculated υ5 (тТ + τ) = υ ₄ (тТ + τ) + υ5 ((т-1) Т + τ),

and. (tT ₊ ^ = K _r ^^ T + x), the sixth signal is calculated, the control signal is formed, summing the fifth and sixth signals.

In FIG. 1 shows the power circuit ISN PV. In FIG. 2 shows the time diagram of the voltages and currents of the elements of the power circuit ISN PV. In FIG. 3 shows a structural diagram of a control device ISN PV. In FIG. Figure 4-9 shows the time diagrams of the output voltage, inductor current, and load current of the ISN PV, confirming the solution of the problem.

The power circuit of the ISN PV (Fig. 1) contains a control element 1, consisting of UEC 2 and diode 3, inductor 4, capacitor 5, input capacitor 6, input 7, output 8, common wire 9.

Referring to FIG. 2 time diagrams of the currents and voltages of the power circuit elements of the ISN and the load current are necessary for understanding the concepts of the adjustable component and the signal current of the capacitor and explaining the changes in the currents and voltages of the elements of the power circuit of the ISN during the operation of the stabilizer. The time diagrams show: 1 _b - current of the inductor; 1 _n - load current; 1 _С - signal capacitor current; 1 _{C. St} is the stationary component of the signal capacitor current; 1 _{C. p} is the stationary component of the signal capacitor current; and _y is the PWM control signal; and _GLIN - PWM reference signal; 1 _and the control signal of the power key ISN PV with the allocation of a stationary 1 _and . _Art and adjustable 1 _and . _p constituents.

To prove the solution of the above problem by means of the proposed control method, in particular, to ensure high quality of the output voltage of the ISN PV, we will use the approach described in [5], where the control law is synthesized, which ensures the minimum final duration of transient processes in the ISN with PWM and lower power circuit type. The control law is synthesized by the case of small deviations of the pulse duration of the power key control ISN & _uy Т T, (1) where T is the transformation period. It consists in converting a system with pulse-width modulation (PWM) to a system with amplitude-pulse modulation (AIM), synthesizing a sequential correction device using the third polynomial synthesis equation [6] and reversing the transition from a system with AIM to a system with PWM, taking into account specifics introduced by PWM. Application in the synthesis of the third polynomial equation allows you to achieve the minimum final duration of transients in the presence of deviation of the parameters of the correction device and the power circuit of the ISN from the nominal ones. The synthesis itself is carried out according to the adjustable components of the state variables, which are understood as the deviations of the state variables from their values in the stationary mode.

In [5], an implementation variant of this control law was also found for such an ISN, in which only discrete values of the adjustable voltage component of the capacitance of the output filter capacitor of the ISN are used. According to this embodiment, the adjustable component of the input signal of the pulse-width modulator has the form = - ^ - [2 ^ ( _т т) -и _Сг (( _т -1) т)], (2) where б ₀ = LС / Т, B and C - inductance and capacitance of the output filter ISN; T is the conversion period, and _in - the voltage at the input of the ISN; To _m = D1 _and . _y / DI _y (tT) = T / and _t ; D1 _and . _y is the increment of the pulse duration of the power key control ISN relative to the time tT controlled switching; and _{C. p} (tT) - discrete values of the adjustable component of the voltage across the capacitor of the output filter capacitor; and _in - the amplitude of the sawtooth PWM voltage.

- 3 019047

Determine the adjustable voltage component and _s . _p on the capacitance of the output filter capacitor ISN, included in (2), by performing computational operations with discrete or continuous values of the output voltage and the _output stabilizer is not possible. This is because the equivalent circuit of the capacitor of the output filter ISN can be represented as a series-connected capacitance Cf of the capacitor and its internal active resistance K _s . Therefore, in the output voltage of the ISN - the voltage on the capacitor of the output filter of the ISN, in addition to the voltage and _c on the capacitor capacitance, there is a voltage And _kc on the internal active resistance K _{s of the} capacitor. Since the internal active resistance K. _{from the} capacitor is subject to significant changes under the influence of temperature and time factors, the voltage b | < _c will also change, and therefore, determine the voltage across the capacitance and _c , for example, by subtracting the undefined voltage b | < _c from output voltage and _{output is} not possible.

Calculate the adjustable voltage component and _s . _p on the capacitance of the capacitor of the output filter of the ISN down-type can be by integrating the current of this capacitor. For a boost type SID, this solution is not applicable, because, unlike a step-down type SID, during operation of the step-up type SID, the connection between the inductor and the capacitor of its output filter is periodically broken at intervals of the non-conducting state of the diode. Therefore, when the duration of the control pulses deviates from the stationary upstream ISN output filter capacitor, not only the adjustable, but also part of the stationary component of the inductor current is transmitted, which for the purpose of generating the control signal is an information noise. To eliminate such information interference, you can use the solution used in [4], according to which instead of the actual current of the capacitor ISN, an information signal is used, the capacitor current, calculated by the expression

1c 1b ⁼ _bm ~ 1a and 1 and _t, (B) 1 where _L - current throttle power SRI chain _1N - load current IOS and _O - the output voltage of SRI and _Rin - input voltage SRI.

Multiplication by a factor of load current and _O / _Rin and resulting load current level to the level of the constant component current choke. Therefore, according to (3), the information signal of the capacitor current does not contain a constant component, i.e. similar to the current of the capacitor ISN step-down type, and the integration of this current allows you to calculate the adjustable component of the voltage and _{C. p} on the capacitor of the output filter included in (2).

The adjustable voltage component acting on the inductor ISN, is calculated as the voltage difference on the throttle at the stages of the conductive and non-conductive states of the power switch. For a down-type SID, the adjustable component of the voltage at the inductor included in (2) is equal to _in - the voltage at the input of the SID, and for a boost-type SID, the adjustable component of the voltage at the inductor is equal to the _output - the voltage at the output of the SID. Therefore, the law of variation of the adjustable component of the input signal of a pulse-width modulator in a boost type IMS according to the discrete values of the adjustable component of the voltage across the capacitance of the output filter capacitor can be obtained from (2) by replacing both _input and _output and ^ (тТ) ~ - Л— [ _{2 and p (t} T) _Cp s ((t -1) 7)]. (4) ^ uL ™

The current discrete values of the input signal of the pulse-width modulator, which determine the duration of the control pulse over the period T, which is the time interval from tT to (t + 1) T, according to [5] are determined as _ul (tT) = and _y ^ t -1) 7) + C7 _V4 > 7). (5)

In the practical implementation of the ISN, it is undesirable for the switching times of the TT time of the power switch and the sampling points of the information signals, since at the moments of switching (at the switching intervals) of the power switch, current changes occur in the power circuits of the power switch with high speeds and disturbances in the electromagnetic field are generated, leading to distortions in the information values signals during their measurement due to electromagnetic interference in the measuring circuits. Therefore, it is desirable to shift the moments of measuring and fixing the discrete values of information signals by a certain time interval τ, where τ is, in the general case, an arbitrarily chosen fixed time interval | τ | <Τ (see Fig. 2). In order to have time for the computational procedures necessary to determine the input signal of a pulse-width modulator by the moment of formation of the adjustable front of switching the power switch, it is advisable to choose τ so that the time moments (tT + τ) are maximally removed from the moments of the adjustable switch of the power switch . For ISN MF and when modulating the leading edge of the pulse, the time instants (tT + τ) should be selected immediately after the moment of turning off the power switch of the ISN, i.e. τ << Τ.

- 4 019047

In view of the above expression, expressions (4) and (5) can be rewritten in the form υ _ν „(тТ + т) = -_ ^ - [2and _c . (ТТ + т) -и _Сг ((т - 1) Т + g)] (6) and _9t & _m and

u ^ (mT + r) = and _y, ((m - 1) 7 + r) + and _y , (mT + r). (7)

Calculate the increment over the period T of the regulated voltage component DL _s . _p on the capacitor capacitance can be achieved by integrating the increment of the adjustable component D1 _s on the time interval equal to the period T. _p of the capacitor current ι тТ & и _Ср ( _т т) = - / d / _s . _p (( _t -1) D) L. (8)> (Л | 1) Г

Since the increment of the adjustable component of the signal, the capacitor current in the interval between the adjustable switching times tT of the power switch of the ISN remains unchanged (Fig. 2), to determine it, it is sufficient to calculate the first difference of the signal, the capacitor current p ( ^tT ) = ⁷ with p ( ^tT > ~! <:.? (( ^t ~ \) ^t ) or taking into account the shift by τ

Д7 _Ср ("7 + r) = 1 _{С р} (тТ + т) - 1 _С „ ((/ "- 1) 7 + т). (nine)

Since the values of the first difference of the signal, the capacitor current in the interval between the adjustable switching times of the power switch of the SPI is constant, (8) taking into account the shift by the interval τ can be written in the form

T

D17 _Cp (t7 + g) = —D / _Cp (7 + g). (10)

Determine the discrete values of the adjustable component of the voltage across the capacitance of the output filter capacitor by the expression and _with? (tT + r) = and _cr ((t -1) 7 + τ) + & and _cp (tT + g). (eleven)

Thus, replacing the integration procedure according to (8) by determining the area of the rectangle according to (10) allows us to determine the increment of the adjustable voltage component on the capacitor capacitance Di _s . _p (tT + g) and the adjustable voltage component itself and _s . _p (tT + t) in the vicinity of the instant tT + τ, i.e. earlier than the moment of time at which the adjustable front of the power switch control pulse is formed. Accordingly, the calculation of the input signal of the pulse width modulator using (6) and (7) can also be performed in the vicinity of the time instant τT + τ, i.e. earlier than the moment of adjustable switching of the power switch.

In FIG. 3 is a structural diagram of a control device that implements the discrete law (2) of the formation of the input PWM signal.

Devices for sampling and storage of UVX1-UVX4 provide sampling of input signals at time instants (tT + τ) and storage of selected signal values at subsequent time intervals with a duration of conversion period T. The meter of the first difference PR ensures the fulfillment of (9), calculator B1 performs calculations according to (10) and (11), and calculators B2 and B3 - according to (4) and (5), respectively.

To ensure astatism of the output voltage of the ISN, a method similar to that used in the prototype [4] is used, according to which the input signal _{from the} PWM is formed as the sum of the dynamic control signal and _y . _q and signal and _y . _st , setting the static level of the output voltage. Signal and _y . _{st is} calculated as the integral of the voltage mismatch signal, taken with a certain coefficient K _p , and the value of this coefficient is chosen small enough so that the signal increment and _y in the interval of the transient process. _Art was much smaller than the increment of the dynamic control signal and _y . _d . This eliminates the influence of the signal and _y . _article on the dynamic characteristics of the ISN. In the proposed solution with a discrete method of generating the input signal of the modulator, the integration procedure is replaced by the procedure of summing the discrete values of the mismatch signal e (kT) and the signal and _y . _{Art is} calculated by calculator B4 according to the expression

I and _y ^ K ^ _£ (tT ₊ t), (12) where e (kT) = and _o (tT) -and ₀ are the discrete values of the voltage error signal, and ₀ is the reference voltage, and 1 = T, 2T , 3T, ...

To confirm the operability of the ISN with a discrete method of generating the input signal of the modulator, the ISN PV model in the 8r1se format has been developed. The control device model is made in accordance with the structural diagram shown in FIG. 3, using analog devices that simulate digital processing of incoming information. The ISN power circuit model has the following parameters: inductor inductance b = 150 μH, capacitance of the output filter capacitor

- 5 019047 ra C = 1000 μF, conversion period T = 25 μs, output voltage and _output = 100 V. Information signals are sampled at times tT + τ, where τ = 1 μs. Computational procedures in the model, taking 1 μs in duration, are performed immediately after the selection of information signals in accordance with the structural diagram of the control device (Fig. 3).

Switching of the load can occur at arbitrary points in time relative to the beginning of the conversion period T. Therefore, the accuracy of calculating the increment of the adjustable component of the voltage across the capacitor of the output filter of the ISN will depend on the moment of switching the load. In FIG. Figures 4 and 5 show time diagrams of the load currents of 1 _n and the inductor C, the output voltage and the _output , input and reference PWM voltages, for the case when a stepwise change in the load current leads to a change in the duration of the control pulse, which does not violate condition (1).

The diagrams differ in the input voltage level of the ISN: in FIG. 4 and _in = 80 V, in FIG. 5 and _in = 90 V.

In FIG. Figure 6-8 shows the time diagrams of the load currents of 1 _n and the inductor 1 _b , output voltage and _output , input and reference PWM voltages, for and _in = 50 V at different moments of load switching, when a stepwise change in the load current leads to a change in the duration of the control pulse, not violating condition (1). So, switching the load immediately after the start of the conversion period (Fig. 6) leads to the equality of the values of the regulated component of the voltage across the capacitor capacitance obtained in the calculation according to formulas (8) and (10). In the case of switching the load in the middle or at the end of the conversion period (Fig. 7, 8), the adjustable voltage component of the capacitor capacitance obtained by (10) does not coincide with the value of the adjustable component obtained by (8). This leads to inaccurate formation of the duration of the power switch control pulse, which is expressed in an increase in the duration of the transient process up to 3-5 conversion periods and the amplitude of the deviation of the output voltage of the ISN. Achieving the duration of the transient process in 3-5 conversion periods follows from the return of the inductor current, the output voltage and the input signal of the pulse-width modulator to stationary values after 3-5 conversion periods after the moment of switching current.

In FIG. Figure 9 shows the timing diagrams for the case of a stepwise change in the load current of a larger value, leading to a violation of condition (1). In this case, the duration of the transition process increases slightly, but remains finite.

A study of the processes in the ISN model showed the operability of the ISN with the proposed control device and the achievement of the minimum final duration of transients in 3-4 conversion periods with a stepwise change in the load current that does not violate condition (1). With a significant value of the switched component of the load current, the duration of the transition process increases to 3-5 conversion periods, however, the final nature of the transition process is preserved, which is explained by the use of the third polynomial equation for the synthesis of the control law and an adequate transition from a system with AIM to a system with PWM.

The need for a single, during the period T, digitization and computational operations frees up a significant time interval in the operation of a digital computing device, which can be used to diagnose SIDs, solve problems of distributing the load current between individual SPSs when they work parallel to the total load, which is an additional advantage of the proposed method for managing ISN.

Literature.

1. A.S. No. 1403037 USSR, cl. C05P 1/56. A method of stabilizing the output voltage of a pulse stabilizer / V.I. Ivanchura, A.V. Manakov, Yu.V. Krasnobaev, B.P. Sustin. - Publ. 06/15/88, bull. Number 22.

2. A.S. No. 440659 USSR, cl. C05P 1/56. DC voltage stabilizer / Yu.A. Mordvinov and P.P. Gursky. - Publ. 10/15/92, bull. Number 38.

3. Patent No. 2025764 of the Russian Federation, cl. C05P 1/56. A method of controlling a pulse stabilizer / V.I. Ivanchura, A.V. Manakov, Yu.V. Krasnobaev, B.P. Sustin. - Publ. 12/30/94, bull. Number 24.

4. Patent No. 2238583 of the Russian Federation, cl. C05P 1/56. A method of controlling a pulse stabilizer / Yu.V. Krasnobaev, I.V. Alatov, Yu.A. Vtorushin, B.N. Mumlin. - Publ. 10/20/2004, bull. Number 34.

5. Soustin B.P., Ivanchura V.I., Chernyshev A.I., Islyaev Sh.N. Spacecraft power supply systems. - Novosibirsk: Science, Siberian Publishing Company, 1994. - 318 p.

6. Tsypkin Ya. Z. Theory of linear impulse systems. - M .: Fizmatgiz, 1963 .-- 968 p.

Claims (1)

CLAIM A method of controlling a pulsed stabilizer containing a inductor with an inductance B and a diode connected in series between the input and output of the stabilizer, a controlled electrical switch connected between the common wire of the stabilizer and the junction point of the inductor and diode, a capacitor with a capacitance C connected between the output and the common wire of the stabilizer, It consists in the fact that the measured voltage and the output _O and the inlet and _Rin stabilizer inductor currents 1 _s and load _1N, calculated capacitor current signal _C = 1 1b-1 _n s _O / _Rin and forming dissolved voltage error signal by subtracting the reference voltage from the voltage stabilizer at the output, the first signal is calculated by multiplying the voltage error signal by a factor K _p, where K _p <4-K ₁ -C-and _out / [and _in -T (2-K _c -C + T)], K _c is the internal resistance of the capacitor, the control signal forms the control pulses of the controlled electric key on the basis of the width width modulation, characterized in that the measurement voltages and _O and and _Rin at the outlet and inlet of stabilizer the current inductor 1 _s and load _1N produce in each period conversion at fixed times (mT + τ), where m = 0, 1, 2, 3, ..., τ - in the general case, an arbitrarily chosen fixed time interval | τ | <Τ, equally remote from the beginning of each th period transform T produce memorization of measured values of voltage and _O and and _Rin at the outlet and inlet of stabilizer the current inductor 1 _s and load _of 1 _N, the first signal calculating K _p e (mT + t) produced at the times (mT + τ) a second signal is obtained by subtracting the capacitor current 1 _C (tT + t) from the signal value, the capacitor current signal ^ (^ - Ι ^ + τ) stored in the previous period, the capacitor current signal received in the current conversion period is stored, calculate the third signal and ₃ , multiplying the second the signal by the coefficient K ₁ = E-I _l / (and _output -T), where and _l is the amplitude value of the sawtooth linear reference signal of the pulse-width modulator, the value of the third signal is stored, the fourth signal υ ₄ (тТ + τ) = 2υ is calculated ₃ (tT + τ) -υ ₃ ((t-1) Τ + τ), remember the fourth signal, calculate the fifth signal υ5 (tT + τ) = υ ₄ (tT + τ) + υ5 ((t-1) T + τ), calculate I (7 ₆ (^ 7 '+ τ) = ^ _ρ Σε (/ »7' + τ) _{} the} sixth signal ^{t = 0} form the control signal by summing the fifth and sixth signal