DE917260C - Magnetic or dielectric modulator - Google Patents

Description

The invention relates to a magnetic or dielectric modulator for amplifying or modulating an electrical signal with the frequency f _a , whereby vibrations with the frequencies k · f j, ± f _a {k - 0, I, 2, 3 ...) generated, which can be taken from a pair of terminals.

So lead z. B. the primary windings of a conventional magnetic modulator, which is composed of two non-linear transformers with the same gear ratios and the same magnetization curves, whose primary windings are in the same direction in series, whose secondary windings are connected in series in opposite directions, a polarizing direct current and at the same time just like the secondary windings under the influence of a on them, mostly simple, harmonic supply voltage with the fundamental frequency f _v a distorted alternating current, namely the primary windings an alternating current with the frequencies 2 / "j ,, ^ f ₉ etc., the secondary windings an alternating current with the frequencies f _v , ^ f ₁ , «0 etc.

If a small signal with the frequency f _{Q is} superimposed on the polarizing direct current, each component of the alternating currents is modulated with the frequency f _q. »5

Intermodulation between the modulation components produced in this way and components with the supply frequency f _v or a harmonic thereof results in components with the in the signal circuit, ie in the primary windings

Frequency f _a , ie there is a reaction of the output signals on the signal source.

In the case of a magnetic amplifier, which consists of a magnetic modulator and one behind them switched demodulator exists, there is also such a reaction on the signal source.

In the case of dielectric modulators and amplifiers, which are based on the same principle, in which but instead of non-linear inductances, the same

ίο capacities are used, a similar one arises Impact on the signal source.

This reaction on the signal source is generally undesirable. The circuit according to the invention largely avoids this disadvantage and has the characteristic that on the pair of terminals at which the desired oscillation Wf ₂ , Jl · f _a occurs, where η is one of the values of k with the exception of k - ο, a Impedance, which together with the internal output impedance of the further

ao direction forms an impedance Z according to

C «f, + fa) ^a

where ZJ ₂ , ^ _{1 is} the complex impedance Z for the frequency nf _p + f _q and Z _n . _p ^ _{a represents} the complex conjugate value of the impedance Z for the frequency nf _P - fq , where furthermore «= 1 if it is a dielectric modulator, and a = - 1 if it is a magnetic modulator, and where Z is the equivalent impedance of the parallel connection of the external impedance and the internal output impedance if all voltage components with frequencies kf ^ -j- f _a (k - ο and n) are vanishingly small compared to the voltage components with frequencies f _a and nf _p ± f _g , or where Z is the equivalent impedance of the series connection of the external impedance and the internal output impedance, if all current components with frequencies kf _p ± f ₅ (k = 0 and n) vanish compared to the current components with frequencies f _a and nf _v ± f _a are small. The invention is explained with reference to the figures shown in the drawing.

Fig. Ι shows a magnetic modulator of the usual type,

Fig. 2 is a circuit diagram according to the invention for a magnetic modulator, in which the feed frequency very high compared to the signal frequency and the fundamental oscillation modulated with the signal the desired output signal is

3 shows a variant of the secondary side of a magnetic modulator in which the feed frequency is not very high compared to the signal frequency and a sideband of the fundamental oscillation modulated with the signal is the desired output signal; a dielectric modulator of the usual type,
6 shows a circuit diagram according to the invention for a dielectric modulator in which the feed frequency is very high compared to the signal frequency and the fundamental frequency modulated with the signal is the desired output signal the signal frequency is not very high and a sideband of the fundamental oscillation modulated with the signal is the desired output signal, FIG. 8 shows the equivalent diagram of the circuit according to FIG. 7, FIG. 9 shows a circuit diagram according to the invention for a magnetic modulator of the type shown in FIG , at which the feed frequency is very high compared to the signal frequency and the second harmonic modulated with the signal is the desired output signal,

FIG. 10 shows a circuit diagram according to the invention for a dielectric modulator of the type shown in FIG Type in which the feed frequency is very high compared to the signal frequency and that with the signal modulated second harmonic is the desired output signal,

11 shows a circuit diagram according to the invention for a magnetic modulator in which the feed frequency not very high compared to the signal frequency and a sideband of that modulated with the signal second harmonic is the desired output signal, and

Fig. 12 is a circuit diagram according to the invention for a dielectric modulator, in which the feed frequency not very high compared to the signal frequency and a sideband of that modulated with the signal second harmonic is the desired output signal.

For a better understanding of the invention we consider the already mentioned example of a magnetic modulator of the type shown in Fig. 1, which has two non-linear transformers T ₁ and T ₂ with cores 5 and 6, which in a non-linear part of the induction as a function of the magnetic The curve representing the field strength can be operated. The transformers have the same curves and the same gear ratios, their primary windings ι and 2 are in series and their secondary windings 3 and 4 in opposite directions in series, whereby a periodic supply voltage at the terminals C, D with the fundamental frequency f _v only current or flux components with the circular frequencies 2p, 4p, 6p etc. at the terminals A, B and exclusively current or. Flux components with circular frequencies p, 3p, $ p etc. at terminals C, D result. _{A direct current J 0} flowing through the primary windings at the same time generates an adjustable premagnetization. Here p = 2 π f j is assumed.

If no signal is initially superimposed on the direct current J ₀ , then the following applies if the instantaneous values of the flows and currents at terminals A, B Φ _χ or J ₁ and at terminals C, D Φ ₂ or J ₂ are disregarded of hysteresis and eddy current losses in the cores,

J ₁ = J ₁ ((P ₁ , Φ ₂ ), J ₂ = 4 (Φι, Φ2).

If a sufficiently small signal is introduced at terminals A, B , the first applies to the changes

Order of I ₁ , I ₂ , Φ ₁ and Φ ₂ compared to the initial state

oil ^AT

Ai ₁ = -jr- + aa

^δΙ *

With the help of energy considerations it can be demonstrated that γ- = Yjg-. Since AI ₁ and A Φ ₁ only components ^s

with the angular frequencies 2p, 4p etc. and AI ₂ and A Φ _% only contain components with the angular frequencies p, 2p etc., the changes to I ₁ , I ₂ , Φ _χ and Φ _{2 can be} written

AI ₁ = [«" + a ₂ cos (2pt + OC ₂ ) + ...] A Φ ₁ + [O ₁ cos pt + a ₃ cos (3 ^ + oc ₃ ) + ...] A Φ ₂ , AI ₂ = Ia ₁ cos pt + «3 cos (3pt + a ₃ ) + ...] Λ Φ _χ + [δ ₀ + b ₂ cos (2 ^ + ^ ₂ ) + ...] A Φ ₂ ,

k - o, i, 2, 3 etc. Let the external system connected to the primary and secondary windings be such that only flux components and therefore voltage components with the frequencies f _Q and fv i fq occur, i.e. when q = 2 π f _q ,

where, for the sake of simplicity, the phase angle of cos pt is assumed to be zero.

If AI ₁ , Δ I ₂ , Δ Φ _χ and A Φ _{2 are} generated by a small signal with the frequency f _q , then these changes generally contain components with the frequencies k · f _v ± f _Q , where

»Ο Δ Φ _χ = cp _q cos {qt + 0 _β ),

Δ Φ ₂ = <p _v _ _q [cos (/> -?) * + © "-J + <p _{v + q} cos [(p + q) t + 6 _{1) + q} ].

These A Φ _λ and A Φ ₂ then result in components with the frequency f _q in AI ₁ , which are given by AI _q = « _O g? _e cos (qt + θ _α ) + τ / 2 Ci ₁ ψ _ν _ _Q cos (qt - θ _ν -α) + ¹ I ^{2 α} ι Ψ ν + α ^cos ii ^t + ^ v + a)

or

i _q ej {at + y> _Q ) = a _o <p _Q ej (qt + 0 _e ) + 1/2 α _ιΨρ _ _a ε / {qt - θ _ρ _ _β ) + τ / 2 a _{l (} p _{v + Q} ε j {qt + Θ _{ν + (Ι} )

be conditioned.

In AI _{2 there} are no components with the frequency f _q .

A transition according to directed values, where k _a ε j θ _χ = K _x and k _x - j θ _χ = K% and K * is therefore the conjugate complex value of K _x , results

In a similar way, the result for the current components with the frequencies f _v + f _a and fv - / "« in / 11 ₂

In Δ I _{1 there} are no components with the frequencies f “ + f _Q and f _v - f _q .

During the transition from magnetic fluxes to voltages and after converting I ₁ , _ _q into Ip- _q , the last three equations become

ί q - «0 ·

v% - _a V.

V + Q

¹ i (P + q) '

iq

γ Iν +1 = 1/2 «1 77 - ^x / 2 ^ ₂ ε / ^ ₂

According to the invention, the external impedance

dimensioned in such a way that together with the internal output impedance of the further device, as measured between the terminals C ₁ D , it forms an equivalent impedance Z which meets the condition

(/ υ Ta) 'Zp-q = (fj, + f ") -

corresponds, where Z _{P + Q represents} the equivalent impedance for the parallel connection of the external impedance and the internal output impedance for the frequency f _B + f _q and Zp _ j represents the conjugate complex value of this equivalent impedance for the frequency f _v - f _q .

As will be demonstrated below, this condition corresponds

V * V ,

'V - 3 ¹ D + Q

p-q-p + q

In this case, as follows from the formula for I _{q, it} follows that components with the frequencies f _v + f _q and f _v - f _{q then} have no influence on components with the frequency f n, in other words that then a repercussion on the signal source is avoided.

If the load connected to terminals C, D has an impedance Z ' or an admittance Y'

= - ^ ₇ - then holds

τ, V <V,

j * γι * γ *

■ * ■ P - Q - * ■ P - Q "P - Qt

where Y ' _P + _{a is} the value of Y' for the frequency f _v ~ \ - f _a and Y'p-q is the complex conjugate value of Y ' for the frequency / "- f _a .

After eliminating I ₂ , _{+ a} and IJ _ _a by means of this admittance _, Fp-ja and Fp _ _B can be determined from the equations for I ₃₁₊ ,, and Ip _ a. The substitution in the condition for freedom from retroactive effects - _ - gives the following formula:

where the asterisk again indicates that the complex conjugate value of the formula between accolades must be taken.

The equation for I ₃ , _{+ a} can be written as follows:

From this it follows that the inner

Represents output admittance for frequency f _v + ^ _a. Also, from the equation for Ip _ _a detectable that

- 11 V.

represents ^inner output admittance for the frequency f _% - f _g . The value (δ ₀ - iJ2b _z sjß ₂ ) - ¹ has the dimension of an inductance. If Y is defined as the equivalent admittance for the externally connected admittance Y ' and the internal output impedance lying parallel to this, one can write for the condition of freedom from interference

or

(fv fa)

-q- (fn +

+0 where Z ₁ , _{+ e is} the equivalent impedance for the parallel connection of the external impedance Z ' and the internal output impedance for the frequency f _p + f _a and Zp _ _{Q represents} the complex conjugate value of this equivalent impedance for the frequency f _v - f _a . It is

♦ 5 demonstrable that the derived condition for freedom from retroactive effects is not limited to the case in which only even-numbered ones are present on the input terminals Harmonics of the supply frequency and at the output terminals only odd harmonics occur like

jo in the example chosen, but applies to any magnetic modulator that z. B. from a single non-linear inductance or a non-linear transformer or from two non-linear transformers of the same, but in which the separation between even and odd harmonics is effected in a different way, z. B. in which the windings 1 and 2 of Fig. 1 are not connected in series, but in parallel. One must also sufficiently fulfill the prerequisite on which the analysis was based, namely that the flux components and therefore also the stress components with the frequencies kfj, ± f _Q for k ~> 2 equal to zero or at least almost equal to zero ν + a T

j (P + q)

are. In the selected example according to FIG. 1, currents with the frequencies f _q , 2f "± f ' _Q , 4f _P ± f _a etc. flow through the input terminals A, B. This is the external impedance between the terminals, B for the frequencies 2 / ^ ± f _a , 4f _v ± f _a etc. very small compared to the external impedance for the frequency f _t , then the voltage components at these terminals with the frequencies 2f _v ± f _a , \ f _p ± f _a etc. are compared to the voltage components with the frequency f _a vanishingly small. If this condition is not automatically met, e.g. B. if the signal source is an impedance-free voltage source, this can be achieved by connecting a suitable capacitance between the terminals A, B. At the output terminals one must also ensure that the impedance Z ', which is intended to contain the impedance of the supply generator, has a high value for the frequencies f _v ± f _a compared to the impedance for the others at the terminals C, D occurring frequencies, i.e. sf _v ± f _q , 5 fm db fq etc.

If the signal has a frequency range between fq-Äfq and f _Q + Af _Q , the derived condition must be fulfilled, at least approximately, for all frequencies between f _Q - Af _a and f _q + Af _a.

Similar considerations apply to a magnetic amplifier, but the load generated by a demodulator must then be taken into account in the external impedance Z '.

If the signal frequency f _{a is} very small compared to the supply frequency f _v , the condition for freedom from interference can be met by switching on a capacitance between the output terminals C and Ό, possibly in series with or in parallel with a load resistor depending on the application. The circuit for this case is shown in FIG. The quadrupole M can contain a magnetic modulator of any type. The parallel resonance circuit, which consists of the internal output impedance with an inductive character in a parallel _{branch and the combination of a capacitance C 2} and a resistor R and, if the supply generator is not impedance-free, the impedance of the supply generator in the generally other branch, must then be set to the frequency f _{p be} matched. The feed generator is shown in FIG. 2 in series with the external impedance C ₂ -R , but can also be connected in parallel to this, in which case the internal impedance must also be taken into account. The capacitance C ₂ simultaneously causes

that components with frequencies higher than fp ± fa · which are already low compared to the components with frequencies f _p ± f _q , are attenuated even further. The capacitance C ₁ in the input circuit fulfills such a task for components which are above the frequency f _{q of} the input signal. The resistor R can here, for. B. be the load formed by a demodulator, which in this case, so when the signal frequency f _q ίο is very low compared to the feed frequency f _v , must be a phase demodulator, because the expression

V j.
__ - p _ + _ Q corresponds to the expression

P + q

V * Pi

^j p + q>

as follows from the formula for I _q , expressed in Φ _α , _{Φρ_ 8} and 0 “ _{+ e} .
So this means that

so <Pp-q ε —jöji-a = - ψρ + q ψρ-q = 9? j, _{+ i} ; Vp- _q = ~

Using these equations results for O ₂ , ie the currents occurring at the terminals C, D

or

A Φ ₂ = - 9P, _{+ e} cos [(# -?) * - 0, + J + <p _{P + q} cos [(p + q) t Λ Φ ₂ = −2 (p _{p + q} sin pt ■ sin (qt +

The voltage at terminals C, D therefore results in

Vcd = -2p (p _{P + q} cospt- sin (^ + φ _{Ρ + α} ) -2q <p _{P + q} sin ^ i cos (qt + θ _{ν + α} ).

If p is large compared to q , one can write for Vcd:

= -2p (p _{v + q} cos

sin

From the expressions for Δ I ₁ and Zl I ₂ as a function of A Φ ₁ and Zl Φ _{2 it} can be deduced that the component with the fundamental frequency of Φ _{2 is} proportional to cos pt, i.e. the supply voltage is proportional to sin pt . Since the voltage Vcd is proportional to cos pt , the supply voltage is therefore phase-modulated with the signal of frequency f _q and free from amplitude modulation. If f _{q is} not very low compared to f _P , there is no pure phase modulation, since in Vcd the term - 2q <p _P + _q sin pt cos (qt + 0 _{P + q} ) is no longer negligible, so that amplitude modulation also occurs . If the feed frequency is no longer very high compared to the signal frequency, for example if an intermediate frequency band between the frequencies f _q - Af _q and f _q - \ - Af _{q is to be} amplified or modulated from which z. If, for example, the lower sideband f _p - (f _q ± Δ f _q ) is required, a circuit such as that shown for the secondary side in FIG. 3 can be used. Here the internal output impedance is shown schematically as inductance Zi. This sixth order network contains three inductances and three capacitances in a delta connection and can therefore be replaced by the series connection of two parallel circuits L _a -C _a and L _t -C _b according to FIG. The resonance frequencies of these two circles must be at the frequencies f _v + f _Q and f _p - f _q . In order to meet the condition for freedom from feedback, L _a and L ₆ must correspond to the following relationship:

L _n U

f I f f. f '

I ρ \ I q Ip Iq

which also means that the absolute bandwidth of the two parallel circuits remains the same if the Coils roughly the same ratio of row

resistance: have inductance. This naturally defines the capacities C _a and C _6. If, in accordance with the preceding, the circle L _a -C _a is tuned to the frequency f ^ + f _q , this means that in FIG. 3 the circles Zj-C ₁ and go L ₂ -C _{2 are} also tuned to the frequencies f _P + f _{q are} matched, with a degree of freedom then remaining for the circle X ₃ -C ₃ , which can serve to give the transmission of the frequencies f _v - (( ₉ ± Δ f _q ) a desired size. For components with higher The circuit connected to the terminals C, D forms a short circuit with frequencies as f "+ f _q , so that undesired voltage components cannot occur at these terminals. The voltage component occurs with the frequency f _v -j- (f _q i Zl f") on circuit L ₂ -C ₂ so that only voltages with frequencies f _v - (/ " ₄ ± Zl f _q ) occur at terminals E, F. If the supply generator is not a pure impedance-free voltage source, its internal resistance must be taken into account Furthermore, the circles mentioned may also contain parallel or series resistances, which are then used at d he vote must be taken into account accordingly. So z. B. a parallel to the circle L ₃ -C ₃ lying resistor represent the load formed by a demodulator.

It is evident that every network found by transformation from the network between the terminals C, D of FIG. 3 also corresponds to the set conditions.

But if you want to continue to use the frequency range / "" + (f _q ± Δ f _Q ) , then the two circles Zj-C ₁ and L ₂ -C ₂ must be tuned to the frequency f _v - f _q , and it starts terminals E, F only have voltage components with frequencies f _P + (f _q ± Zl f _q ) .

In the foregoing it has been assumed that the external system connected to the primary and secondary windings is such that only flux components and therefore voltage components with frequencies f _q and f _p ± f _q occur. That

However, the external system can also be such that only current components with the frequencies f _a and fp ± fa occur.

Serves z. B. as a signal source a device that is to be understood as a current source with high internal resistance, currents with the frequencies 2 f _p ± f _a , 4f “± f _a etc. cannot occur in the circuit according to FIG. If care is also taken that the external impedance Z 'connected to the terminals C, D for the frequencies f _v + _ f _{a is} low compared to the impedance for the frequencies 3 f _v + f _Q , Sfviifa , etc., then im generally satisfies the requirement that only current components with
step.

the frequencies f "and f _v

= [c ₀ + c ₂ cos (2 pt + ?> ₂ ) + .. ·] Δ I ₁ + [C ₁ COsPt + c ₃ cos {3 pt + γ ₃ ) + ....] 4J ₂ , = t ^c i cosp t + C ₃ cos (3 pt + γ ₃ ) + ...] Λ J ₁ , + [d ₀ + d ₂ cos (2 pt + <5 ₂ ) + · ·.] A J ₂ ,

For the relationship between smaller changes in J ₁ , I ₂ , Φ _χ and Φ ₂ one can write in a similar way to the previous one:

for the sake of simplicity, the phase angle of cos pt is again selected to be zero.

If again ΔI ₁ , Δ J ₂ , Δ (P ₁ and Δ Φ ₂ arise under the influence of a small signal with the frequency f _g , then these changes generally contain components with the frequencies kf _p + _ f _a Acceptance then applies

ZlJ ₁ = J ₄ cos (qt + A ₀ ),

Δ J ₂ = J ₃₁ -JCOs [^ - q) t + A ₃₁ -J + i _{P + 8} cos [{p + q) t +

This leads to the following equations in time-independent complex values, with the flows through 25 voltages are replaced.

Ll

= c ₀ J ₈ + 1/2 C ₁ I * _ _e + 1/2 C ₁ J _{31 + 5} -

ν *

^V PQ

j (P - ?) - jd ₂ J _{31 + 4}

V j ⁼

-Q +

From this equation it follows that a reaction on the signal source of components with the frequencies fp + fq and f _v - f _{a is} avoided if the following applies

T * T

· * »- (T - ^± ι

Ip + q ■ ·

By introducing the terminals C, D

• [Z ' _{P + a}

q) (d ₀ - 1/2 d _z ε j O ₂ )] = closed external impedance Z 'can be derived from the

Equations for ^

and

V,

4-j

again

3 (■ P - i) """ 3 (P + 1) Vp-g, and F _{21 + 8 are} eliminated and these are resolved for Jp_ _? And J _{31 + 8} .

Substitution of the values found in this way for Ip-q and J _{31 + 8} in the equation J £ _ ₂ = -J _{31 + 8} results

p - q ■ [Z ' _P _

- q) (d ₀ - 1/2 d ₂ ε j <5 ₂ )].

It can again be demonstrated that / (p + q) (d ₀ -1/2 d _z ε j δ ₂ ) is the internal output impedance for the frequency f _B + f _a and / (p - q) (d ₀ - ΐ / 2ίΖ ₂ ε j <5 ₂ ) represents the internal output impedance for the frequency f j, -— fq . The size (d ₀ - 1/2 ίϋ ₂ ε / O ₂ ) has the dimension of an inductance. If Z is defined as the total impedance of the external impedance Z ' and the internal output impedance in series with it, the condition for freedom from interference can be written as

p- q

or

p-q

(/ D Γ Ql ^ PI ~~ \ tρ "Γ Tq)

where Z _{31 + 8 represents} the equivalent impedance for the series connection of the external impedance Z 'and the internal output impedance for the frequency f _v + f _Q and Z £ _j the conjugate complex value of this equivalent impedance for the frequency f _P -f _a .

In the special case that the feed frequency is very high compared to the signal frequency, the feed current is purely phase-modulated and therefore free of amplitude modulation. However, one cannot be satisfied with connecting a capacitance between the terminals C, D as the external impedance, since this does not prevent current components from occurring with frequencies higher than fj, + f _Q. In this case, the components with higher frequencies suppressing trap circuits must be connected in series with the capacitance.

If f% is not very high compared to f _q , then amplitude modulation also occurs. In this case, too, it is not sufficient to use a circuit according to FIG. 3 on the secondary side, but components with higher components must also be used here

Connect frequencies as f _v - \ - fq suppressing trap circuits in series with the internal output impedance. So far only magnetic modulators and amplifiers have been discussed. Dielectric modulators are to be regarded as dual circuits magnetic modulators, and when the rivers Φ φ in the rules applicable to the latter equations by charges and the coil currents are replaced by capacitor voltages V I, we arrive at similar equations.

In Fig. 5, a dielectric modulator of the usual type is shown, wherein C ₁ and C _{2 represent} non-linear capacitances. The circuit is again chosen in such a way that a periodic supply voltage with the fundamental frequency f _o exclusively current or voltage components with the angular frequencies ij>, 4 ft etc. at the terminal ends, B and exclusively current or voltage components with the angular frequencies ft, 2 > ft etc. at terminals C, D results. Here, too, p = 2 π f _{P is} assumed.

The direct voltage F ₀ serves as an adjustable bias voltage. This is not necessary when using a dielectric with remanent polarization, a so-called electret. If the outer network is chosen in such a way that no other

As charge components, i.e. current components, which occur with the frequencies f _g and f _v ± f _a , where f _{t is} again the frequency of the input signal, such an impedance must be connected to the pair of terminals C, D in order to avoid any effect on the signal source that the total impedance of this external impedance and the internal output impedance in series with it, which in this case is capacitive, corresponds to the following condition:

(/ 3> + Iq) Zp + j = (fp f _q ) Z _p _ q,

where Z _{11 + 5 represents} the value of this equivalent impedance for the frequency f _v + f _a and Zp _ _{q represents} the complex conjugate value of the equivalent impedance for the frequency fj, - f _Q. Here too, in the case of a dielectric amplifier, the load formed by a demodulator in the external impedance Z ' must be taken into account. Furthermore, the formula given here applies again to a dielectric modulator or amplifier of any type and is not restricted to a type according to FIG. 5.

It should also be noted that with a frequency range of the signal between f _q -Af _a and f _Q + Af _g, the above-mentioned condition for all frequencies between f _g - Af _g and f _g - \ - Af _g must be at least approximately fulfilled if one Impact on the signal source should be avoided.

If the feed frequency is very high compared to the signal frequency, the condition for freedom from reverse action can be met by placing an inductance between terminals C, D , possibly in series with or parallel to a resistor serving as a real load, depending on the application , or a resistor formed by a demodulator. The series circuit consisting of the internal output impedance and the series or parallel connection of the inductance and the resistor must then be matched to the frequency f _v , the possible impedance of the supply generator also having to be taken into account. In addition, the requirement is then sufficiently fulfilled that only currents with the frequencies f _v - \ - f _g and f _v - f _a flow through the terminals C, D. Current components with higher frequencies, which are already weak compared to the current components with the frequencies f _v + f _Q and f _p - f _q , are suppressed even further by the inductance. So that no currents with frequencies higher than f _a flow through the terminal, B , an inductance can also be connected in series with the signal source, if this does not already have an internal impedance that is relatively high at least for frequencies above that of the signal. In Fig. 6 a circuit is shown for this case, in which the quadrupole E is any dielectric modulator. The series connection of L ₂ and R represents the external impedance and L _{1 represents} the inductance, which suppresses the undesired current components occurring at terminals A, B.

As in a similar case with the magnetic modulator, a phase-modulated voltage occurs at terminals C, D. If the frequency f _{Q is} no longer low compared to the supply frequency f _v , a more complex network must be selected for the output impedance, and no pure phase modulation is obtained either. 7 shows a circuit for the secondary side of a dielectric modulator in the event that a frequency band fqa: Af _{a is to be} modulated again and only one of the sidebands is to be used again, the internal output impedance, which is capacitive in dielectric modulators, being shown schematically by the capacitance Z _t is. The circuit according to FIG. 7 is dual to the circuit according to FIG. 3. The sixth order network contains three inductors and three capacitances in a star connection and can therefore by connecting two series circuits L _a -C _a and L _r C _b according to FIG. 8 in parallel be replaced. Here, too, the resonance frequencies must be at the frequencies f _v + f _a and f _B - f _a , and the following relationship between L ₀ and L _{6 must} exist:

T ν 'T q Td Tq

In addition, the ratios between series resistance and inductance of the coils must be approximately the same so that the two series circuits have the same absolute bandwidth. Here, too, the circuits mentioned may contain parallel or series resistances, which must be taken into account when coordinating the circuits, as well as the possible impedance of the supplying generator. A resistor in circle X ₃ -C ₃ can again represent the load generated by a demodulator.

Each network which can be transposed from the network shown in FIG. 7 also corresponds to the iao requirements.

If the desired frequency band is again f ₉ - (f _a ± Af q), the series circles Z ₁ -L ₁ and L ₂ -C ₂ must be tuned to the frequency f " + f _a , and the band f" - (fq ± Af _a ) can be taken from terminals E, F.

But it is also possible to choose the external system in such a way that only voltage components with the frequencies f _q and f _v ± f _a occur. Is the signal source z. B. to be understood as an impedance-free voltage source, voltage components of frequencies 2, fp ± f _q , 4 / ρ uz fg etc. cannot occur at terminals A ₁ B in a circuit according to FIG. One must then ensure that the external impedance connected to the terminals C ₁ D for

to the frequencies f _v + f _q and f _v - f _{q is} high compared to the impedance for the frequencies 3 / \ p ± fq, 5fvazfq etc., so that voltage components with the frequencies $ f _v ± f _q , 5f _v ± fa compared to stress components with the frequencies f ^ + f. _q and f _P - f _{q are} vanishingly small.

The external impedance connected to terminals C, D must then be such that the equivalent impedance of the parallel connection of this external impedance and the internal output impedance corresponds to the following condition in order to avoid any repercussions on the signal source:

(/ ν "t" / β

where Z ₃ , _ | _ _{s is} the value of this equivalent impedance for the frequency f _p -f- f _a and Z * _v ^ _g the conjugate complex

The equivalent impedance value for the frequency f _v - f _q is.

It also applies again that in a frequency range

of the signal between f _v - Af _a and f _q + Af _q the condition, at least approximately, must be met for all frequencies between f _a - Λ f _a and f _Q + Af _q .

In the special case that the feed frequency is very high compared to the signal frequency, the feed voltage is again purely phase-modulated. As with the magnetic modulator, in which the current components must meet certain conditions, it is not sufficient to connect an inductance between the terminals C, D as the external impedance; for similar reasons, elements for higher frequencies must be provided parallel to the output impedance have approximately zero impedance as f _p. In a circuit according to FIG. 7, too, this measure must be taken to avoid undesirable voltage components.

So far, the fundamental oscillation of the supply voltage with the modulation present on it has been considered as the desired output signal. A suitable choice of the external system ensures that only current or voltage components with the frequencies f _a , f _v - f _q and fp + fq occur.

It is possible, however, that in a certain case a higher harmonic of the supply voltage with the modulation present on it is desired as the output signal. With the help of the external system, one must then ensure that only current or voltage components with the frequencies f _q , nf'Jf- ³ T f Q and η f _v - η f _Q (where η is an integer) occur, or at least that the current or voltage components with other frequencies compared to those with the frequencies f _q , η f _v + f _a and »/ \ p— f _{q are} vanishingly small. In this case, the condition that must be met to avoid any reaction on the signal source is

where for dielectric modulators and amplifiers a = ι and for magnetic modulators and amplifiers = - 1 applies. Ζ _ην - ^. _α is the complex value of the impedance Z for the frequency refj, + / "«, and Z% j, - _q is the complex conjugate value of the impedance Z for the frequency nf _v - f _q .

Z represents the equivalent impedance of the parallel connection of the external impedance and the internal output impedance if the voltage components with the frequencies kf _v ± f _a (k = 0 and n) are vanishingly small compared to the voltage components with the frequencies f _a and nf _v ± f _q are or is Z the equivalent impedance of the series connection of the external impedance and the internal output impedance, if the current components with the frequencies kf _v 3zf _Q (k = 0 and n) are vanishingly small compared to the current components with the frequencies f _q and nfv-nfq.

If the desired output signal is an even harmonic modulated with the input signal of the supply oscillation can, if desired, be applied to a bias of the dielectric modulator be dispensed with, provided that this does not entail any disadvantages for certain applications, since an oscillation with the carrier wave frequency is then also missing in the output signal.

In the conventional types shown in FIGS. 1 and 5, even harmonics must be taken from terminals A, B. It is evident that in the case of magnetic modulators and amplifiers this can also be done by means of separate windings lying parallel to the primary windings. Likewise, the fundamental frequencies and the higher odd harmonics can be supplied by separate windings lying parallel to the secondary windings.

If the desired output signal is an even harmonic modulated with the input signal and the feed frequency is very high compared to the signal frequency, simple circuits can be specified under certain conditions in which the occurrence of an effect on the signal source is avoided if modulators are used in which only frequencies f _a , 2, f _v ± f _q , 4f _P ± f _a etc. occur on one pair of clamps and only frequencies f _B i f _a , Zfvzhfq etc. occur on another pair of clamps, as is the case with conventional embodiments according to FIGS. 1 and 5 is the case. Assuming that, in the case of a magnetic modulator, the supply generator is an impedance-free voltage source, the condition for freedom from interference is fulfilled in that between the terminal end, B, a capacitance C ₁ , possibly in series with or in parallel with a resistor R, as shown in Fig. 9 is switched, which is matched with the internal output impedance to the frequency 2 f _P , again also taking into account the impedance of the signal source. The signal source is shown in FIG. 9

parallel to the external impedance, but can also be connected in series with it or only with R , which causes a more effective short-circuit function of C ₁ for frequencies higher than f _q . The capacitance forms a short circuit for higher harmonics, ie ^ f ₁ , etc., so that no voltage comporients with the frequencies 4 f _v ± f _Q , 6fpuzf _q etc. occur on the primary side. Since the supply source on the secondary side is an impedance-free voltage source, at least for frequencies other than f _v , no undesired frequencies occur there either. The desired second harmonic is again purely phase-modulated by the signal. So that the direct voltage source can supply a polarizing direct current without this source forming a short circuit for desired components, a second winding magnetically parallel to the primary winding can be used, which of course can also be carried out in all other cases. Instead, a permanent magnet can also be used for the premagnetization.

If only the desired current components are to occur, notch filters must again be used for the undesired components. If, in the case of the dielectric modulator, the supply generator has a very high impedance, at least for frequencies higher than f ₉ , the condition for freedom from interference is met by an inductance L between terminals A, B , possibly in series with or parallel to a resistor R, if desired, is switched via an isolating transformer as in FIG. 10, the series connection of which is matched to the frequency 2 f _{v with the internal output impedance.} Here too, undesirable current components are greatly weakened. The desired second harmonic current component is again purely phase modulated.

If only desired voltage components are to occur in the dielectric modulator, see above short circuits must be used again.

If f _{P is} not very high compared to f _a and you want to continue to use a sideband of the second harmonic modulated by the signal, then in the case of a magnetic modulator, if the supply generator is an impedance-free voltage source at least for frequencies other than f _{v, as shown in Fig.} 3 between the terminals C, D or a circuit resulting therefrom by transformation can be connected between the terminals A, B , as shown in FIG. 11. For L _a and L ₆ must then apply

L _n U

₂ faf ~~~ 2f f '

and the circles Z ₁ -C ₁ and L ₂ -C ₂ must be matched to the undesired sideband, again taking into account the impedance of the signal source.
Similarly, one can in the case of the dielectric modulator when the supply generator having a very high impedance at least for frequencies other than f _v, one of these resulting from transformation circuit between terminals A in FIG. 7 between the terminals C, D lying or , B as shown in FIG. This must also apply here

L _a U

and the circles Zj-L ₁ and L ₂ -C ₂ must be matched to the undesired sideband. In the case of amplifiers, the load generated by a demodulator must again be taken into account.

Claims (10)

Patent claims: i. Magnetic or dielectric modulator for modulating or amplifying an electrical signal with the frequency f _g by means of a supply oscillation with the fundamental frequency f _p , which generates oscillations with the frequencies kf _v ± f _g (where k = 0, i, 2, 3 ...) which can be taken from a pair of clamps, characterized in that the pair of clamps at which the desired oscillation nf _v ± / " _e occurs, where η is one of the values of k with the exception of k = ο, has an impedance which forms an impedance Z together with the internal output impedance of the further device, according to FIG where Zn 3, + _{s is} the complex impedance Z for the frequency η f „ + f _a and Ζ _η3 , _ _{β is} the complex conjugate value of the impedance Z for the frequency nfj, - f _a , furthermore a = 1 if it is a dielectric modulator, and a = -1 if it is a magnetic modulator, and where Z is the equivalent impedance of the parallel connection of the external impedance and the internal output impedance, if all voltage components with the. Frequencies kfvnzfq (k = ο and n) are vanishingly small compared to the voltage components with the frequencies f _Q and nf _v ± f _Q , or Z is the equivalent impedance of the series connection of the external impedance and the internal output impedance, if all current components with the frequencies kf _P ± f _q (k = 0 and n) are vanishingly small compared to the current components with the frequencies / "" and nf _v ± f _q . 2. Modulator according to claim 1, characterized in that the feed frequency f _{v is} so high compared to the signal frequency f _a that the desired output oscillation is free of amplitude modulation. 3. Magnetic modulator according to claim 2, wherein the desired output oscillation is f _P ± f _g and a signal voltage source is used with an at least for frequencies above that of the signal voltage relatively small internal impedance, characterized in that the external impedance contains a capacitance and with the internal output impedance forms a parallel resonance circuit i »5 tuned to the frequency f _p. 4. Dielectric modulator according to claim 2, in which the desired output oscillation is fvazfq and a signal current source is used with an internal impedance that is relatively high at least for frequencies above that of the signal current source, characterized in that the external impedance contains an inductance and the internal output impedance has one forms a series resonant circuit tuned to the frequency f _p. 5. Magnetic modulator according to claim 2, in which the desired output oscillation is zfvuzfa and the components with the frequencies f _a , 2 f "± f _a , 4fv ± fa etc. on one set of terminals and components with the frequencies f _{v on another set of terminals} ± f _Q , 3 f D ± f β etc. and in which a supply voltage source with a relatively small internal impedance is used at least for other than the supply frequency, characterized in that the external impedance contains a capacitance and with the internal output impedance a frequency 2 f _{v forms a} tuned parallel resonance circuit. 6. Dielectric modulator according to claim 2, in which the desired output oscillation is zfvdzfa and the components with the frequencies f _q , 2f _v uz f _Q , 4 fB db fq etc. and on another set of terminals components with the frequencies f _v ± f _a , 3fpuzfa ^etc. generated and in which a supply current source with a relatively high internal impedance is used at least for frequencies other than the supply frequency, characterized in that the external impedance contains an inductance and, with the internal output impedance, forms a series resonant circuit tuned to the frequency 2 f _v. 7. Magnetic modulator according to claim i, at which the desired output oscillation is a sideband of the fundamental oscillation modulated by the signal is and a signal voltage source with at least one for frequencies above the the signal voltage uses a relatively small internal impedance, characterized in that that the parallel connection of the internal output impedance and the external impedance are two parallel resonance circuits in series corresponds, one of which to the unwanted sideband and the other to the desired one Sideband is matched. 8. Dielectric modulator according to claim 1, wherein the desired output oscillation Sideband of the fundamental wave modulated by the signal and a signal current source with one at least for frequencies above that of the signal current source, which is relatively high internal impedance is used, characterized in that the series connection of the internal Output impedance and the external impedance of two parallel series resonance circuits corresponds, one of which to the unwanted sideband and the other to the desired one Sideband is matched. 9. Magnetic modulator according to claim 1, wherein the desired output oscillation is a sideband of the second harmonic modulated by the signal and the components with the frequencies f _a , 2 fj, ± f _a , 4 / 'vif Q ^etc. - and on another set of terminals components with the frequencies f _v 4 ^ f _a , 3 f & if α etc. and in which a supply voltage source is used with an internal impedance that is relatively low at least for frequencies other than the supply frequency, characterized in that the parallel connection of the internal output impedance and the external impedance are two in series. Corresponds to parallel resonance circles, one of which is tuned to the undesired sideband and the other of which is tuned to the desired sideband. 10. Dielectric modulator according to claim 1, wherein the desired output oscillation is a sideband of the second harmonic modulated by the signal and the components with the frequencies f _Q , zf ^ if ,, 4 f j> dz f α etc. and on another set of terminals components with the frequencies f _v ± f _a , 3fj> dzfa etc. are generated and in which a feed current source is used with an internal impedance that is higher at least for frequencies other than the feed frequency characterized in that the series connection of the internal output impedance and the external impedance corresponds to two parallel series resonance circuits, one of which is tuned to the undesired sideband and the other is tuned to the desired sideband. 1 sheet of drawings © 9542 8.54