DE4209305C2 - Method and device for field-oriented control of asynchronous machines - Google Patents

Description

Speed-controllable electrical systems are becoming increasingly popular Drives both as feed drives and as main spindles drives in modern machines such as B. in machining centers for machining materials or in automa tized transport facilities used. The stand Today's users mainly three basically under different basic types of electrical machines are available. These are the DC machine on the one hand, and the other Synchronous machine and - last but not least - the asynchronous machine. The Asyn is characterized by these three basic types chron machine (if, as usual, as a squirrel-cage machine is carried out) in particular by its simple mechanical Construction and its associated, low manufacturing costs as well as their robustness during overload operation. Considered if one looks at these advantages, it is obvious that the lion's share both on the main spindle drive market and on to that of the feed drives in the asynchronous machine suspect.

The advantages of the asynchronous machine are, however, a major disadvantage opposite part: The direction of the so-called rotor flow space pointer of an asynchronous machine is neither stator-fixed (like this in the case of a DC machine) is still rotor-proof (like this with the modern, permanently excited synchronous machines of the Case is given). Rather, this rotor flow space rotates pointer when the machine is loaded slowly over its rotor path. As a result, it is the asynchronous machine both in their treatment in a stator-related as well when treating them in a rotor-related coordinate system  around a meshed controlled system. This in turn increases the Difficulty with control-technical treatment of the Asyn chron machine compared to that of synchronous and direct current machines considerably.

In the following the prior art will be described with reference to the drawing Fig. 1-3. It shows

FIG. 1 is a suitable for the viewing of the asynchronous machine in field-oriented coordinate system equivalent circuit diagram;

FIG. 2 shows a control block diagram of the controlled system derived from the equivalent circuit in FIG. 1 with the components of the field-oriented voltage vector (u _d , u _q ) as input variables and the components of the field-oriented stator current vector (i _d , i _q ) as output variables;

Fig. 3 is a space vector diagram to illustrate the effects of misorientation of the field-oriented coordinate system.

A major advance in the control engineering of asynchronous machines is the so-called "field-oriented control" of asynchronous machines, which was first introduced by Blaschke [1]. Here, use is made of a so-called "field-oriented coordinate system" with the so-called longitudinal axis d and the transverse axis q orthogonal to this. This is characterized in that - assuming an optimal function of the field-oriented control system - its d-axis always points in the direction of the rotor flux space pointer _R. Conversely, this also means that the rotor flow space pointer _{R of} the asynchronous machine always has only one component in the d-direction of the coordinate system in question, but not in the q-direction of the same.

If we now also consider the stator current space vector of the asynchronous machine in this field-oriented coordinate system, it can be split into a component i _d in the direction of the d-axis and a component i _q in the direction of the q-axis. The component i _{q of} the stator current space vector is usually referred to as cross current i _{q of} the asynchronous machine and directly affects the internal torque of the asynchronous machine, but not its rotor flux. It therefore acts like the armature current of a DC machine. In a corresponding manner, the component i _{d of} the stator current space vector, which is usually referred to as the longitudinal current i _{d of} the asynchronous machine, directly affects only the rotor flux of the asynchronous machine. The internal torque of the machine in question is influenced only indirectly by changing its rotor flux. The longitudinal current i _d thus acts like the excitation current of a DC machine. In the manner just described, a decoupling of the meshed control paths for the rotor flux and the internal torque of the asynchronous machine is achieved. The control-related treatment of the asynchronous machine in the field-oriented coordinate system is as simple as that of a DC machine.

With all these explanations, however, was always an optimal one Function of such a field-oriented control system pre- puts. Unfortunately, this optimal function is in practice mostly not given. The main causes for this are in following explained.

In the following, that part of a field-oriented rule systems for asynchronous machines, which regulates the asyn Chron machine up to the specification of the setpoints for the sta gate currents in field-oriented coordinates as well as a possibly subsequent transformation of these target values into another Ko ordinate system accomplished, referred to as guidance system. This is therefore part of a field-oriented rule systems to understand.

A very important problem area is that of hiring desired stator currents of the asynchronous machine, d. H. the current regulation. In the case of a field-oriented management system Output quantities are only target values for the currents to be set, which, moreover, are usually still in one specified with respect to the stator rotating coordinate system are. To set the desired from the guidance system On the one hand, currents is a functional unit for transformation of the field-oriented quantities in the stator-fixed coordinate system and on the other hand a functional unit for current regulation  required. The latter unit can be used for Current regulation either with variables of the stator-oriented coordinate system or with sizes of the field-oriented coordinate system work.

A very simple solution to the problem at hand is to first transform the field-oriented current setpoints i _d, setpoint and i _q, setpoint into a stator-oriented coordinate system and the stator-oriented current setpoints obtained in this way are transformed into a three-phase converter with a current control unit, which works on the principle of time-discrete Switching state change works, feed [2]. In this way, a dynamic, high-quality current control can be implemented, which automatically uses the full available reserve capacity of the three-phase converter used if required.

In many applications, however, such a solution comes not considered. Often the user sets a constant pulse frequency of the three-phase converter required, a requirement best by using a modulation scheme according to the principle of pulse width modulation (PWM) can [3]. The innermost control engineering functional unit at Such a PWM process is always the controlled on position of the average over a pulse period output voltages of the three-phase converter. In other applications case is the direct connection of the stator terminals Asynchronous machine with the output terminals of the converter wishes and there will be the interposition of a three-phase LC output filters with regulated output voltage required. In both cases, the regulation of the stator currents of the Asyn chron machine using the steady (when operating with LC Output filter) or quasi-continuous (when using PWM) Stator voltages of the asynchronous machine that can be entered as manipulated variables respectively.

Manufacturers are prepared to set up such a regulation modern control systems for asynchronous machines still great headache. There are two key points here in the foreground:

• - The frequency response of the guide transfer function of the current control unit as well
• - the inevitable interference of the people in question Controlled system through the internal voltages of the asynchronous machine or through related sizes.

Are the current regulators with sizes in a stator-oriented Coordinate system work, so already the frequency response the guide transfer function of such a current controller massive problem. It is desired to cover the whole Frequency range, which range from 0 to some 100 Hz may, a leadership transfer function with an amount of almost 1 and a phase shift of almost 0. Also they act over a wide range in amplitude and frequency variable, induced voltages of the asynchronous machine as disturbances on the current control loops. The fault transfer The carrying function should therefore be covered in the entire Fre quenz range have an amount of almost 0. This extreme Men requirements show that this is an extreme difficult problem to solve. Even well-known, on the Experienced companies in the field of control technology operate up to today an immense effort to control the current from Asyn chron machines with controllers, which with stator-oriented size work well to get a satisfactory grip [4].

Somewhat more favorable conditions exist if the current regulator works with variables in the field-oriented coordinate system or with such in a coordinate system which has only a low rotational speed compared to the field-oriented one. If the controllers work with field-oriented variables, this has the advantage that both the guide variables and the control and disturbance variables that occur in the control loops in question are always constant in the electrically steady state of the asynchronous machine acts. This statement applies regardless of the speed and the load condition of the machine. In the other case, the use of a coordinate system, which has a low rotational speed compared to the field-oriented, the frequencies of the quantities occurring in the electrically steady state of the asynchronous machine are so low that the frequency response of the guide transfer function of a current controller is not significant in this case either. Such a coordinate system can e.g. B. be a rotor-oriented. In both cases, the control transmission functions of the current regulator do not have to be as extreme as those when the current regulator is designed with stator-oriented variables. However, the problem of influencing the current control loops through disturbance variables remains. The dynamic behavior in this regard can, however, be improved by means of a voltage feedforward control such as, for. B. from [6] (pp. 81-97, especially Fig. 4.11 and Chap. 4.2.2) is known. To understand such a pilot control are the relationships with reference to FIG. 1 and FIG. 2 explained. For this purpose, it should first be specified that the component u _{d of} the stator voltage pointer of the asynchronous machine pointing in the direction of the d-axis of the field-oriented coordinate system is referred to below as the longitudinal voltage u _{d of} the asynchronous machine. In the same way, the component of the stator voltage space vector of the asynchronous machine pointing in the direction of the q axis is referred to below as the transverse voltage u _{q of} the asynchronous machine.

Fig. 1 shows the electrical equivalent circuit of the asynchronous machine for its treatment in the field-oriented coordinate system. It consists in part of the series connection of a choke with the inductance L _σ ( 1 ), which represents the subtransient inductance of the asynchronous machine [5] and a voltage source with the voltage u _{d, stat} ( 2 ). The voltage u _{d, stat} is the longitudinal voltage of the asynchronous machine required in the electrically steady state. The longitudinal voltage u _{d of} the asynchronous machine is located at the terminals of this series connection, the series current i _{d of} the asynchronous machine is formed in the inductor L _σ ( 1 ). The other part of the electrical equivalent circuit of the asynchronous machine consists in a corresponding manner of the series connection of another choke with inductance L _σ ( 3 ) and a voltage source with voltage u _{q, stat} ( 4 ). The voltage u _{q, stat} is the transverse voltage of the asynchronous machine required in the steady state. The transverse voltage u _{q of} the asynchronous machine is at the terminals of this series connection, and the transverse current i _q of the asynchronous machine is formed in the further choke with inductance L _σ . The voltages u _{d, stat} and u _{q, stat} are referred to below as stationary voltages.

This is not the usual AC equivalent circuit diagram the asynchronous machine. The currents and tensions are equal sizes. At the inductance falls in the stationary Condition no voltage off. Represent the voltage sources the stationary voltages including the voltage drops on the leakage inductance and the ohmic stator resistance.

This results directly in the FIG. 2, control block diagram of the current control paths with the longitudinal voltage u _d and the transverse voltage u _{q of} the asynchronous machine as input variables and the longitudinal current i _d and the cross current i _{q of} the asynchronous machine as output variables. The difference between the longitudinal voltage u _{d of} the asynchronous machine and its longitudinal voltage u _{d, stat} required in the electrically steady state, acts on the input of an integrator with the integration time constant L _σ ( 5 ), which outputs the longitudinal current i _{d of} the asynchronous machine at its output. Similarly, the difference between the transverse tension acts u _q of the asynchronous machine, and their in electrically steady to stand required cross-voltage u _{q, stat} on the input of a further integrator with integration time constant L _σ (6) which at its output the cross-current i _q of Asynchronous machine outputs.

As FIG. 2 shows, the voltages required in the steady state can be interpreted as disturbance variables. Their effect is compensated according to [6], [7], [8] or [9] by simulating them in a model and adding them to the current regulator outputs. This is also the state of the art on which the preamble of claims 1 and 14 is based. The model processes the target components or the actual components of the stator current vector and the magnetizing current. It works e.g. B. according to the formulas (4.26a) and (4.26b) given in [6] on page 87:

Size R _S is the ohmic stator resistance of the asynchronous machine, size L _H is its main field inductance related to the stator side, size L _R is its rotor inductance related to the stator side, size ω₁ is the angular velocity, with which the field-oriented coordinate system of the guide system rotates compared to a stator-oriented coordinate system, for size i _m around the value of the magnetizing current of the asynchronous machine assumed by the guide system, for size i _sq * (t) around the setpoint of the cross current and for size i _sd * (t) around the nominal value of the longitudinal current of the asynchronous machine. Instead of the setpoints i _sq * (t) and i _sd * (t) of the cross current and the longitudinal current, their actual values can be used to simulate the stationary voltages.

These assumed values of the disturbance variables (u _sd0 * (t) and u _sq0 * (t)) are then added to the voltage _setpoints output by the current regulators. In this way, the influence on the controlled system from FIG. 2 is compensated for by the disturbance variables u _{d, stat} and u _{q, stat} - assuming an optimal function of the model and an exact knowledge of the machine parameters R _S , L _σ , L _H and L _R . Since these requirements are not always met, problems arise in practice.

First, the requirement for exact knowledge of these machine parameters shows that even a current control that works with variables of the field-oriented coordinate system (or a coordinate system that rotates very slowly compared to the field-oriented coordinate system) and also uses a voltage pre-control according to the principle described above, remains sensitive to fluctuations in the machine parameters R _S , L _s , L _H and L _R. Every incorrect assumption of one of these parameters leads to faulty voltage pre-control and thus to a renewed effect of disturbance variables on the current control loops.

Second, the "field-oriented coordinate system" which the management system (see definition of the management system on page, paragraph), field-oriented current setpoints and all other, from the model processed sizes, just by an assumed, field-oriented coordinate system. But this is by no means true necessarily with the actual, field-oriented Coordinate system.

Fig. 3 shows a snapshot of the space vector diagram of the asynchronous machine in question both with a rotor-oriented and with a field-oriented coordinate system. The axes x and y shown in dashed lines are the axes of a rotor-related coordinate system, the axes d and q are the axes of the Management system adopted, field-oriented coordinate system. Finally, the space vector is the current stator current space vector of the asynchronous machine. It is clearly identified by its components i _d and i _q . In the guidance system, the angle ε, which the axis d includes with the axis x, is continuously calculated according to the equation

formed and used to determine the field-oriented coordinate system based on the rotor-oriented coordinate system. In the simulation rule for ε, instead of the cross-flow setpoint i _q, the actual value i _{q of} the cross-flow _should also be used. If the _rotor time constant T _{Rq, FS} , which is assumed by the guidance system and is effective in the direction of the axis q, deviates from the actual value of this time constant, then an error angle Δε arises between the field-oriented coordinate system with the axes d and q and the actual field-oriented coordinate system with the Axes d _m and q _m . However, it is the components i _{d, m} and i _{q, m of} the stator current space vector in this last-mentioned coordinate system, the control of which would allow an ideal decoupling between the influencing of the rotor flux of the asynchronous machine and the direct influencing of its internal torque. The management system, however, is only the sizes

i _d = i _{d, m} · cos Δε - i _{q, m} · sin Δε

such as

i _q = i _{q, m} · cos Δε + i _{d, m} · sin Δε

accessible.

For Δε ≠ 0 there is therefore a decoupling of the controlled systems for the rotor flux and for the internal torque of the asynchronous machine not given. This in turn affects time the course of the misalignment Δε. The consequence of this are in the best case undesirable pendulum moments. With unfavorable loading drive states of the asynchronous machine can even become one Instability of the entire control system.

This problem also sheds new light on a further weakness of the compensation of the disturbance variables u _{d, stat} and u _{q, stat} mentioned above in the current control loops with the aid of a model based on the principle given in [6-9]. This compensation takes place in a controlled manner, starting from variables in the field-oriented coordinate system adopted by the guidance system. The occurrence of an error angle Δε, which is unavoidable during operation, immediately leads to incorrect compensation of the disturbance variables in the current control loops and thus to a renewed effect of disturbance variables on the same. In extreme cases, this can even cause these current control loops to become unstable. If the current actual values are processed in the model, those consequences of misorientation, which result from the direct influence of the stator current components, are canceled out, since this influence has a symmetrical effect on the d and q components of the stationary voltage vector. The effects of the incorrectly oriented (and possibly also incorrectly calculated) magnetizing current remain.

As these comments show, the often-made claim Field-oriented asynchronous machines offer a dy Named behavior, that of DC machines would be comparable, with the prior art only limited durability.

The object of the present invention is based on from the prior art described above [6-9] First a dynamic, extremely high quality and against misorientation the field-oriented coordinate system and Parameter fluctuations insensitive detection of the electrical steady state required, stationary voltages to create. By using the then stationary voltages are said to be extremely dynamic high quality and insensitive to the aforementioned influences Current regulation arise. This object is achieved through the characterizing features of the claims 1 and 14 solved.

The so on their Setpoint values set stator currents result in a misorientation of the field-oriented coordinate system due to the then deviating current setpoints still at an amount other than the desired amount the rotor flux and to a different one than desired internal torque. Therefore, it is the purpose of claims 11-13 and 22-24 the occurrence of a significant misorientation of the field-oriented Coordinate system using the invention to prevent detected stationary voltages.

All of the above, so far unavoidable weak points of field-oriented regulations occur when using the invention presented herewith not. Overall, a system for extremely high dynamic quality is being created and extremely insensitive to parameter fluctuations field-oriented control of asynchronous machines. Such a way regulated asynchronous machine actually offers a dynamic one Behavior similar to that of a DC machine in nothing is inferior.

The invention will now be described with reference to FIGS. 4-16. It shows

Fig. 4 is a diagram for explaining the sizes of the rotating reference coordinate system used in the inventive process;

Fig. 5-7 technical control block diagrams of different embodiments of the device for determining the components of the invention of the stationary voltage vector (u _{δ 0,} u _{u 0);}

Fig. 8 is a block diagram for explaining a way to capture the components of the stator current vector (i _δ, i _u) in the reference coordinate system;

Fig. 9 is a block diagram for explaining a possibility for detecting the components of the voltage vector (u _δ, u _u) in the reference coordinate system;

Fig. 10 is a block diagram of a power control according to the invention;

11 shows the block diagram of the current regulator (17, 18) of Figure 10 at which runs as a P controller..;

FIG. 12 shows the block diagram of the current regulator ( 17, 18 ) from FIG. 10 when it is implemented as a PI regulator;

FIG. 13 shows the block diagram of the current regulator ( 17, 18 ) from FIG. 10 when it is designed as a bypass I regulator;

Fig. 14 shows the block diagram of a device according to the invention for replication of the vector of the induced voltages (u _{δ i,} u _{i u);}

Fig. 15 is a block diagram of a device for the inventive determination of the slip angular frequency ω _R using the vector of the induced voltages (u _{i δ,} u _{i u)} of FIG. 14, while the rotating Bezugskorrodinatensystem is (δ, u) identical to that of the guide system assumed, field-oriented coordinate system (d, q);

Fig. 16 is a block diagram of a device for the inventive determining the actual value Ψ _n of Rotorflußverkettung, which then for the formation of the longitudinal current setpoint value i _{δ, should} be used, also there is the rotating reference coordinate system (δ, u) identical to that accepted by the guidance system, the field-oriented coordinate system (d, q).

The present invention makes a so-called reference coordinate system with the axes δ and u use. In this Reference system, the current regulator and the acquisition work for the voltages required in the steady state. The guidance system itself continues to work in the field-oriented Coordinate system.

FIG. 4 serves to illustrate the explanations that now follow . The reference coordinate system rotates with the reference angular velocity ω in relation to a stator-related coordinate system with the axes a and b. All components of space pointers pointing in the direction of the axis δ of the reference coordinate system are identified below with the index δ, all components of space pointers pointing in the direction of the axis u of the reference coordinate system are identified with the index u. An optimal functioning of the invention presented herewith is achieved when ω = ω _s .

This applies regardless of the position of the reference coordinate system in relation to the field-oriented coordinate system. The decisive factor for the optimal function is the fact that for ω = ω _s the setpoints supplied to the current regulators are constant quantities. If, on the other hand, a reference coordinate system with ω ≠ ω _{s is} selected, deviations from the optimal functioning must be accepted. In this case, the setpoints supplied to the current controllers have an angular frequency ω _s -ω different from zero. For ω≈ω _s , that is, when the reference coordinate system is at a low rotational speed compared to the field-oriented coordinate system, the deviations from the optimal functioning are so small that they practically do not appear.

Consequently there is some in the choice of the reference coordinate system Free space. A field-oriented coordinate system, a rotor-oriented Coordinate system or a mixture of both are only examples here as options for execution of the reference coordinate system.

A field-oriented management system ( 7 ) is used in all forms of training of the invention presented herewith. This first determines the field-oriented current setpoints i _d, target and i _q, target. These are then transformed into the reference coordinate system. The setpoints i _{δ ,} setpoint and i _{u ,} setpoint thus obtained for the components i _w and i _{u of} the stator current space vector in the reference coordinate system are finally output at the outputs of the guide system.

Fig. 5 shows a first embodiment of the invention presented herewith . First, the difference between the component of the stator voltage space vector, referred to as delta voltage u _δ , of the asynchronous machine to be controlled and a correction signal u _{w , b is} formed and fed to the input of a first integrator ( 8 ), whose Integration time constant corresponds to the subtransient inductance L _σ . The delta current i _{δ of} the asynchronous machine to be controlled is subtracted from the output variable i _{w , b of} this integrator ( 8 ). The quantity i _{δ , b} -i _{w obtained in} this way is fed to the input of a proportional amplifier with the amplification factor G _δ ( 9 ). Its output variable u _{δ , bP} is identical to the correction signal u _{δ , b} . The correction signal u _{δ , b} is now used as a signal u _{δ 0} for the delta voltage u _{w , stat} required in the electrically steady state. The difference between the component of the stator voltage space vector of the asynchronous machine to be controlled and a correction signal u _{u , b} , which is referred to as theta voltage u _{u , is} formed in a corresponding manner and is fed to the input of a further integrator ( 10 ) whose integration time constant also corresponds to the subtransient inductance L _σ . The theta current i _{u of} the asynchronous machine to be controlled is subtracted from the output variable i _{u , b of} this integrator ( 10 ). The quantity i _{u , b} -i _{u obtained in} this way is fed to the input of a proportional amplifier with the amplification factor G _u ( 11 ). Its output variable u _{u , bP} is identical to the correction signal u _{u , b} . The correction signal u _{u , b} is now used as the signal u _{u 0} for the theta voltage u _{u , stat} required in the electrically steady state.

In the arrangement according to FIG. 5, the transfer functions result

The stationary voltages u _{δ , stat} and u _{u , stat} are therefore _detected here with a first-order delay. Only knowledge of the subtransient inductance L _{σ is} necessary for this detection. No other machine parameters are needed. Furthermore, a misorientation of the field-oriented coordinate system has no effect on the function of the arrangement in question, since neither the amount nor the position of the magnetizing current space vector in relation to the reference coordinate system is processed in it.

In the embodiment of FIG. 6 is expanded, the arrangement shown in Fig. 5 to the effect that the difference signal _{δ i, b} i _δ addition of an integrator with integration time constant T _δ is supplied (12) to the input. The output variable u _{δ , bI of} this integrator ( 12 ) and the output variable u _{δ , bP of} the proportional amplifier ( 9 ) are now added. The resulting sum u _{δ , bP} + u _{δ , bI} is used as a correction signal u _{δ , b} instead of the quantity u _{δ , bP} . In a corresponding manner, the difference signal i _{u , b} -i _{u is} additionally fed to the input of an integrator with the integration time constant T _u ( 13 ). The output variable u _{u , bI of} this integrator ( 13 ) and the output variable u _{u , bP of} the proportional amplifier ( 11 ) are now added. The resulting sum u _{u , bP} + u _{u , bI} is used instead of the quantity u _{u , bP} as a correction signal u _{u , b} .

If the modified in Fig. Education shape shown 6 in that instead of the signal u _{δ, b,} the output u _{δ, bi} of the integrator (12) u _δ as signal _{0 δ} u for the required in the electrically steady state delta _{voltage, stat} of the controlled Asynchronous machine is used and that instead of the signal u _{u , b,} the output variable u _{u , bI of} the integrator ( 13 ) is used as a signal u _{u 0} for the theta voltage u _{u , stat} required in the electrically steady state, of the asynchronous machine to be controlled, this creates the arrangement shown in FIG. 7.

FIG. 8 shows one possibility for detecting the delta current i _δ and the theta current i _u . First, two stator currents of the asynchronous machine ( 14 ) which clearly identify the stator current space vector in a stator-oriented coordinate system are recorded there. In the arrangement shown in FIG. 8, these are the currents i _R and i _S. These currents are then fed to the associated inputs of a first coordinate converter ( 15 ) [6]. Furthermore, the angle referred to below as the reference angle ϕ between the axis δ of the reference coordinate system and a freely selectable but fixed axis of the stator-oriented coordinate system is output by the guide system ( 7 ) and fed to the associated input of the first coordinate converter ( 15 ). This first coordinate converter outputs the delta current i _δ and the theta current i _{u of} the asynchronous machine ( 14 ) at its outputs.

In a concrete example of this, the axis R is selected as the above-mentioned, freely selectable but fixed axis of the stator-oriented coordinate system. Furthermore, the axes δ and u form an orthogonal coordinate system. In this case, the output variables of the first coordinate converter ( 15 ) are formed according to the following equations:

FIG. 9 shows one possibility of detecting the delta voltage u _δ and the theta voltage u _u . First, two stator voltages of the asynchronous machine ( 14 ) which clearly identify the stator voltage space vector in a stator-oriented coordinate system are detected. In the arrangement shown in FIG. 9, these are the voltages u _RS and u _ST . These voltages are now fed to the associated inputs of a second coordinate converter ( 16 ). Furthermore, the reference angle ϕ is output by the guide system ( 7 ) and fed to the associated input of the second coordinate converter ( 16 ). This second coordinate converter outputs the delta voltage u _δ and the theta voltage u _{u of} the asynchronous machine ( 14 ) at its outputs.

In a concrete example, the axis R is again selected as the above-mentioned, freely selectable but fixed axis of the stator-oriented coordinate system. Furthermore, the axes δ and u form an orthogonal coordinate system. In this case, the output variables of the second coordinate converter ( 16 ) are formed according to the following equations:

The entire current control of the asynchronous machine with the intervention of the voltage precontrol is shown in FIG. 10. The setpoint value i _δ output by the control system ( 7 ) _{is intended} for the delta current i _{δ of} the asynchronous machine ( 14 ) and a signal i _{δ is} for its actual value the associated values Inputs of a delta current controller ( 17 ) supplied. This delta current controller outputs a variable u _{δ , corr} at its output. Now the sum of the output variable u _{δ , corr of} the delta current controller ( 17 ) and the signal u _{δ 0} for the delta voltage required in the steady-state condition is formed and as the setpoint u _{δ ,} the delta voltage u _{δ of} the asynchronous machine ( 14 ) used. Similarly, the output from the guidance system (7) Reference is i _u, of the asynchronous machine (14) and a signal _is i _u for the Thetastrom i _u, the corresponding inputs of a Thetastromreglers (18) _is supplied to the actual value. This theta current regulator outputs a quantity u _{u , corr} at its output. Now the sum of the output variable u _{u , corr of} the theta current regulator ( 17 ) and the signal u _{u 0} for the theta voltage required in the steady state is formed and as the setpoint u _{u , should be used} for the theta voltage u _{u of} the asynchronous machine ( 14 ). As a signal i _{δ, is} for the actual value of the delta current i _δ is the measured current i _δ Delta used itself. Likewise, the measured theta current i _u itself is used as signal i _{u and is used} for its actual value.

The setpoints u _{δ , set} and u _{u , set} for the delta and theta voltage of the asynchronous machine ( 14 ) are now fed to the associated inputs of a setpoint coordinate converter ( 19 ). The reference angle ϕ is in turn given by the guide system ( 7 ) and the associated input of this setpoint coordinate converter ( 19 ) supplied. This setpoint coordinate converter ( 19 ) outputs at its outputs two quantities u _1, setpoint and u _{2, set} out which clearly identify the set point of the stator voltage space vector of the asynchronous machine ( 14 ). These are fed to the setpoint inputs of a power electronic actuator ( 20 ) with associated control device ( 21 ), which outputs a voltage space vector with components u ₁ and u ₂ at its output terminals. The output terminals of the power electronic actuator ( 20 ) are connected to the associated stator terminals of the asynchronous machine ( 14 ).

In a concrete example of this, the axis R is again chosen as a freely selectable but fixed axis of the stator-oriented coordinate system. The axes δ and u form an orthogonal coordinate system. In the embodiment in question, the setpoint coordinate converter ( 19 ) is designed so that it outputs the setpoints u _{RS, should} and u _ST, for the stator voltages u _RS and u _{ST of} the asynchronous machine ( 14 ). These setpoints are formed in the setpoint coordinate converter in the exemplary embodiment in question according to the following equations:

The additive connection of the signals u _{δ 0} and u _{u 0} to the current controller outputs compensates for the interference of the current control loops by the stationary voltages u _{w , stat} and u _{u , stat} (see Fig. 2) with high quality, both statically and dynamically. The relationships u _{δ 0} = u _{w , stat} u _{u 0} = u _{u , stat} for the steady-state condition remain valid even if the integration time constant of the integrators ( 8, 10 ) deviates from the value corresponding to the subtransient inductance L _σ . Only the transient response of the signals u _{δ 0} and u _{u 0} after changes in the stationary voltages u _{δ , stat} and u _{u , stat} is impaired by an incorrectly set integration time constant.

The overall arrangement thus has a low sensitivity to parameters with respect to L _σ and a complete insensitivity to all other machine parameters and to a misorientation of the field-oriented coordinate system used by the guide system ( 7 ).

The complete insensitivity to a misorientation of the field-oriented coordinate system stems from the fact that neither the amount nor the position of the magnetizing current space vector with respect to the reference coordinate system find their way into the arrangement described. The complete insensitivity to all machine parameters with the exception of L _σ stems in a corresponding manner from the fact that no other machine parameters except L _σ find their way into the arrangement described.

If the correction signals u _{δ , b} and u _{u , b} , as shown in FIG. 6, are formed with the aid of the two integrators with the integration time constants T _δ ( 12 ) and T _u ( 13 ), then the delta current controller ( 17 ) can be used instead of the measured delta current i _δ the output quantity i _{δ , b of} the first integrator ( 8 ) as signal i _{δ , is} supplied _and the output variable i _{u , b of} the further integrator ( 10 ) can be sent to the theta current regulator ( 18 ) instead of the measured theta current i _u as signal i _{u , is} supplied.

Fig. 11 shows the execution of the current controller ( 17, 18 ) as a P controller. There, the difference i _{w ,} target -i _{δ , is} formed in the delta current controller ( 17 ) between its two input variables. This difference is now fed to the input of a proportional amplifier with the gain factor Ki _δ ( 22 ). Its output variable, designated u _{δ , corr, P,} is output at the output of the delta current controller ( 17 ) as a variable u _{δ , corr} . In a corresponding manner, the difference i _{u , should} -i _{u , is} formed in the theta current regulator ( 18 ) between its two input variables. This difference is fed to the input of a proportional amplifier with the gain factor Ki _u ( 23 ). Its output variable, designated u _{u , corr, P,} is output at the output of the theta current regulator ( 18 ) as variable u _{u , corr} .

Fig. 12 shows the execution of the current controller ( 17, 18 ) as a PI controller. There the difference i _{δ ,} target -i _{δ , is} formed in the delta current controller ( 17 ) between its two input variables. This difference is now fed to the input of a PI amplifier with the proportional gain factor Ki _δ * and the integration time constant Ti _δ ( 24 ). Its output variable, designated u _{δ , corr, PI,} is output at the output of the delta current controller ( 17 ) as a variable u _{δ , corr} . In a corresponding manner, the difference i _{u , should} -i _{u , is} formed in the theta current regulator ( 18 ) between its two input variables. This difference is fed to the input of a PI amplifier with the proportional gain factor Ki _u * and the integration time constant Ti _u ( 25 ). The output variable designated u _{u , corr, PI} is output at the output of the theta current regulator ( 18 ) as a variable u _{u , corr} .

Fig. 13 shows the design of the current controller ( 17, 18 ) as a bypass I controller. Here too, the difference i _{δ ,} target -i _{δ , is} formed in the delta current controller ( 17 ) between its two input variables. This difference is now fed to the input of an integrator with the integration time constant Ti _{δ , BI} ( 26 ). The output variable of this integrator ( 26 ) _, designated i _{δ , corr} , and the setpoint i _{δ ,} for the delta current i _{w of} the asynchronous machine ( 14 ) are then added. In this way, the so-called modified delta current setpoint i _{δ , set} * = i _{δ , set} + i _{δ , corr} . Now, the difference _{δ i, soll} * -Ci1 _δ · i _{δ is δ} i between the modified Delta current target _value, * and the weighted signal _δ by the factor Ci1 _{to δ i, is} formed for the delta current i _δ of the induction machine (14) and fed to the input of a proportional amplifier with the amplification factor Ci _δ ( 27 ). Its output variable, designated u _{δ , corr, BY,} is output at the output of the delta current controller ( 17 ) as a variable u _{δ , corr} . In a corresponding manner, the difference i _{u , soll} - i _{u , is} formed in the theta current regulator ( 18 ) between its two input variables. This difference is now fed to the input of an integrator with the integration time constant Ti _{u , BI} ( 28 ). The output quantity of this integrator ( 28 ) _, designated i _{u , corr} , and the setpoint i _{u , shall be added} for the theta current i _{u of} the asynchronous machine ( 14 ). In this way, the so-called modified theta current setpoint i _{u , set} * = i _{u , set} + i _{u , corr} . Now, the difference i _u, is _{intended to} * - Ci1 _u · i _{u is} between the modified Thetastromsollwert i _{u, to} * and _is the factor Ci1 _u weighted signal i _u, formed for the Thetastrom i _u of the asynchronous machine (14) and fed to the input of a proportional amplifier with the amplification factor Ci _u ( 29 ). The output variable designated u _{u , corr, BY} is output at the output of the theta current regulator ( 18 ) as variable u _{u , corr} .

In a further embodiment of the present invention, the difference u _δ - u _{δ , b} between the delta voltage u _{δ of} the asynchronous machine ( 14 ) and the quantity u _{δ ,} is not the input of the first integrator ( 8 ), as in the previously described embodiments _{. b} , but the difference u _{δ , nb} - u _{δ , b} between a quantity u _{δ , nb} and the quantity u _{δ , b is} supplied. Likewise, the input of the further integrator ( 10 ) is not, as in the training forms described so far, the difference u _u - u _{u , b} between the theta voltage u _{u of} the asynchronous machine ( 14 ) and the quantity u _{u , b} , but the difference u _{u, nb} - u _{u, b} u _u between a _size, and the size _nb u _{u, b} supplied. The quantities u _{δ , nb} and u _{u , nb} are simulated signals for the delta voltage u _δ and the theta voltage u _u .

The simulation of the voltage space vector by information from the feeding pulse inverter is known. In [12] z. B. the components of the voltage space vector with the help of an inverter model from the Inverter switching functions and the measured DC link voltage educated.

An exemplary embodiment, not shown, for this purpose is described below. In this exemplary embodiment, a three-phase, pulse-width-modulated converter is used as the power electronic actuator ( 20 ) with control device ( 21 ). From the setpoints u _{δ ,} for the delta voltage u _{w of} the asynchronous machine ( 14 ) and u _{u ,} for the theta voltage u _{u of} the asynchronous machine ( 14 ) and the reference signal for pulse width modulation [3], the signals u _{δ , nb} for the delta voltage u _{δ of} the asynchronous machine ( 14 ) and u _{u , nb} for the theta voltage u _{u of} the asynchronous machine ( 14 ). In this way, in the exemplary embodiment just described, the effort for detecting the stator voltages and their subsequent transformation into the reference coordinate system can be saved.

A special case of the embodiment of the present invention in question represents the second exemplary embodiment described below. Here, the setpoint u _{w , is used} for the delta voltage u _{δ of} the asynchronous machine directly as a quantity u _{δ , nb} and the setpoint u _{u , is to be used} directly for the theta voltage u _{u of} the asynchronous machine as the quantity u _{u , nb} .

Already at the beginning of the description it was mentioned that the simulation of the slip angle ε and the magnetizing current i _m only accurately reflects the real conditions if the rotor time constant T _{R of} the machine is exactly known. However, an exact knowledge of this time constant can often not be assumed. However, methods are known for obtaining additional information about ε and i _m about the stator voltages of the machine and then using this to correct the slip angle and magnetization current simulation [10], [11] and [6] (at [6] pages 107 to 117 and in particular Figure 4.28 on page 114). All these solutions have in common that the space pointers of the stator voltages and the stator currents form the space pointers of the voltages induced in the stator by the rotor flux. This space pointer is always perpendicular to the rotor flux linkage space pointer, its amount is proportional to the stator angular frequency ω₁ and the amount of the rotor flux linkage space pointer.

The main disadvantage of these solutions is explained below using an example in the field-oriented coordinate system simulated by the guidance system ( 7 ).

The following generally applies:

and

With

u _{d, stat} = R _S · i _d - L _σ · ω₁ · i _q + u _id

and with

u _{q, stat} = R _S · i _q + L _σ · ω₁ · i _d + u _iq

Here u _id and u _{iq are} the components of the space vector of the voltages induced by the rotor flux in the stator. Since a differentiation of the components of the stator current space vector is not practicable, the terms which neglect the time derivatives of i _d and i _q are used in the simulation of u _id and u _iq and it is used in a simplified manner:

u _id = u _d -R _S · i _d + L _σ · ω₁ · i _q

and

u _iq = u _q -R _S · i _q - L _σ · ω₁ · i _d

Such a simulation of u _id and u _iq is based directly or indirectly on the methods according to [6] (see the explanations there on page 113) and also according to [10], [11] (see there in each case FIG. 2). However, u _id and u _iq are only correctly reproduced if

actually zero if the current control loops are in the steady state Condition.

This limitation is completely eliminated when using the method according to the invention, since then basically dynamically high-quality and stationary exact replicas (u _{δ 0} , u _{u 0} ) of the components of the space vector of the stationary voltages required (u _{δ , stat} , u _{u , stat} ) in the case of the reference coordinate system (δ, u) on which the invention is based.

In the embodiment described below with reference to Fig. 14, there is a limitation that the reference coordinate system must be an orthogonal coordinate system. In this embodiment, both the signal u _{δ 0} for the delta voltage u _δ required in the electrically steady state and the signal u _{u 0} for the theta voltage required in the electrically steady state u _{u are} fed to the associated inputs of the guide system ( 7 ). Furthermore, the following four sizes are formed in this guide system ( 7 ):

• - A variable u _{w , R0} , which is a measure of the delta voltage u _{δ , R, stat} = R _S · i _δ which drops in the electrically curved state at the ohmic stator resistance R _{S of} the asynchronous machine ( 14 ).
• - A variable u _{u , R0} , which is a measure of the theta voltage u _{u , R, stat} = R _S · i _u that drops in the electrically curved state at the ohmic stator resistance R _{S of} the asynchronous machine ( 14 ).
• - A variable u _{w , L σ 0} , which is a measure of the delta voltage u _{δ , L σ , stat} = -L _σ · ω _S · i _u which drops at the subtransient inductance L _{σ of} the asynchronous machine ( 14 ) in the electrically steady state .
• - A quantity u _{u , L σ 0} , which is a measure of the theta voltage u _{u , L σ , stat} = L _σ · ω _S · i _δ that drops at the subtransient inductance L _{σ of} the asynchronous machine ( 14 ) in the electrically steady state.

Now, in the guide system (7) has a size ui _δ according to the equation u _δ = u _{δ 0} - u _{δ, R0} - u _{δ, L s 0} and a size ui _u according to the equation u _u = u _{u 0} - u _{u, R0} - u _{u , L s 0} formed. The size ui _δ is referred to below as the induced delta voltage ui _{δ of} the asynchronous machine ( 14 ), the size ui _u as the induced theta voltage ui _{u of} the asynchronous machine ( 14 ). These sizes ui _δ and ui _u are now used in the guide system ( 7 ) to support the flow model [1] of the asynchronous machine ( 14 ) which is always present in the same.

In the guide system ( 7 ), the quantities u _{δ , R0} , u _{u , R0} , u _{w , L σ 0} and u _{u , L σ 0 can be formed} as follows from the target values of the delta current and the theta current:

• - u _{δ , R0} = R _S · i _{δ , should} ,
• - u _{u , R0} = R _S · i _{u , should} ,
• - u _{δ , L σ 0} = -L _σ · ω · i _{u ,} and as well
• - u _{u , L σ 0} = L _σ · ω · i _{δ , should} .

Of course, instead of the target values of delta current and theta current, their actual values or also the output variables i _{δ , b} and i _{u , b of} the two integrators ( 8, 10 ) can be used.

The special features of a development of the embodiment of FIG. 14, FIG. 15. In this embodiment, the shape adopted by the guide system (7) field-oriented Koordi natensystem used as a reference coordinate system. The axis δ is thus identical to the longitudinal axis d, the axis u identical to the transverse axis q of the field-oriented coordinate system adopted by the guide system ( 7 ). The setpoint i _{δ ,} target for the delta current i _{δ of} the asynchronous machine ( 14 ) is first fed in the control system ( 7 ) to the input of a delay element of the first order with the proportional gain factor 1 and the delay time constant T _{R δ , FS} ( 30 ). The size T _{R w , FS} is the rotor time constant of the asynchronous machine ( 14 ) currently assumed by the guidance system ( 7 ) and effective in the direction of the longitudinal axis δ of the reference coordinate system. The size T _{R δ , FS} is variable over time. The output variable i _{m, FS of} this 1st order delay element ( 30 ) is referred to below as the magnetization current i _{m, FS of} the asynchronous machine ( 14 ) initially assumed by the control system ( 7 ). Now a quantity ω _{R, FS} according to the equation

educated. This size is the angular velocity of the rotor flux space pointer of the asynchronous machine ( 14 ) initially assumed by the guiding system ( 7 ) compared to a rotor-oriented coordinate system, with T _{R u , FS being adopted} by the guiding system ( 7 ) in the direction of the transverse axis u effective rotor time constant of the reference coordinate system. Furthermore, the difference ω _R = ω _{R, FS} - ω _{R, corr is formed} between the quantity ω _{R, FS} and a quantity ω _{R, corr} .

This variable ω _R is used as the angular velocity of the rotor flux space vector of the asynchronous machine ( 14 ) compared to a rotor-oriented coordinate system in the guide system ( 7 ). The formation of the quantity ω _{R, corr} occurs depending on a Boolean variable Crit _ω , which is referred to below as the supporting criterion Crit _ω for the reference angle ϕ. If this support criterion Crit _{ω has} the value logic 1, then ω _{R, corr is formed} according to the equation

If, on the other hand, Krit _{ω is} equal to 0, then the quantity ω _{R, corr has} the constant value 0.

The decisive characteristics of a further form of training are shown in FIG. 16. In this form of training, use is made in the guidance system ( 7 ) of a magnetization state replica ( 31 ) and of a magnetization state controller ( 32 ). The magnetization state emulator outputs the rotor flux linkage Ψ _n simulated by the guide system ( 7 ) at its output. This size Ψ _n and Ψ size _to which are denotes the target value of the magnetization state of the asynchronous machine (14) fed to the corresponding inputs of the magnetization state regulator (32). At its output, this outputs the target value i _{δ ,} target for the delta current of the asynchronous machine ( 14 ). The functioning of the magnetization state emulator ( 31 ) differs depending on a Boolean variable Crit _m . This is referred to below as the support criterion Crit _m for the magnetization state of the asynchronous machine ( 14 ). If this support criterion Crit _{m has} the value logically one, the magnetization state emulator ( 31 ) is supplied with a quantity u _i and the quantity ω _S at its associated inputs, and its output quantity Ψ _n is formed in accordance with the equation

The size u _i obeys the relationship:

On the other hand, if the support criterion Crit _{m has} the value 0, the magnetization state replica ( 31 ) is supplied with the setpoint i _δ at its associated input _{, should be} supplied to the asynchronous machine ( 14 ) for the delta current i _δ and the formation of the output variable Ψ _{n of} the magnetization state replica is carried out in the same only because of the guide system (7) which EXISTING flux model of the asynchronous machine (14) from the time profile of the desired value _{w i, is intended} for the delta current i _δ of Asyn chronmaschine (14).

In a first exemplary embodiment, the induced theta voltage ui _{u of} the asynchronous machine ( 14 ) is used as the quantity u _i .

A second embodiment of this arises from the first when in the case of _m = 1 Crit the Magnetisierungszustandsnachbildner (31) in place of the induced Thetaspannung ui _{u is} the so-called main field induced voltage of the induction machine ui u _{i h} is supplied as a size (14). This induced main field voltage ui _{h of} the asynchronous machine ( 14 ) results from its induced delta and theta voltage ui _δ and ui _u according to the equation

Finally, it should be noted that the functions of the two integrators ( 8, 10 ), whose integration time constant corresponds to the subtrasient inductance L _σ , of the proportional amplifier with the gain factor G _δ ( 9 ), of the proportional amplifier with the gain factor G _u ( 11 ), des Integrator with the integration time constant T _δ ( 12 ), the integrator with the integration time constant T _u ( 13 ), the first coordinate converter ( 15 ), the second coordinate converter ( 16 ), the delta current controller ( 17 ), the theta current controller ( 18 ), the setpoint Coordinate converter ( 19 ), the power electronic actuator ( 20 ) with associated control device ( 21 ) and the guide system ( 7 ) can be combined in whole or in part in terms of device technology.

literature

Blaschke, F .: The method of field orientation for controlling the asynchronous machine. Siemens Research and Development Report Vol. 1 (1972), pp. 184-193
[2] Boehringer, A .; Brugger, F .: Transformerless transistor pulse converters with output powers up to 50 kVA. Electrical engineering and mechanical engineering, 1979, H. 12, pp. 538-545
[3] Leonhard, W .: Control of electrical drives. Springer, Berlin 1985
[4] Hügel, H .; Scheswig, G .: New current control method for three-phase asynchronous motors. Drive Technology Vol. 30 (1991), H. 12, pp. 36-42
[5] Boehringer, A .: The start-up of converter motors with a DC link. Dissertation, University of Stuttgart, 1965
[6] Zimmermann, W .: Dynamic high-quality guidance of converter-fed asynchronous machines with almost purely sinusoidal voltages. Dissertation, University of Stuttgart, 1987
[7] Fetz, J .; Horstmann, D .: Comparison of different field oriented control methods for an induction machine fed by a PWM-modulator controlled inverter. EPE Proceedings, Aachen, 1989, pp. 1079-1085
[8] Pollmann, AJ: Software Pulsewidth Modulation for µP Control of AC Drives. IEEE Transactions on Industry Applications, vol. IA-22, H. 4, 1986, pp. 691-696
[9] Viola, R .; Grotstollen, H .: Influence of valve switching times on the behavior of pulse inverters. etzArchiv 10 (1988), H. 6, pp. 181-187
[10] Hentschel, F .; Hussels, P .; Büsch, G .; Hambach, J .: Method for controlling an asynchronous machine. DE 32 21 906 A1
[11] Hussels, P .; Hentschel, F .: Method for controlling an asynchronous machine. DE 35 23 665 A1
[12] Prinz, M .: Process for the formation of the flow area pointer of an asynchronous machine fed via a self-commutated inverter. DE 39 26 994 A1

Claims (25)

1. Method for field-oriented control of asynchronous machines, whereby in a rotating reference coordinate system (δ, u)

• a) the stator voltages and stator currents are detected, and the components of the corresponding vectors are formed, and
• b) the components of the stationary voltage vector (u _{δ 0} , u _{u 0} ) required in the electrically steady state are determined from the components of the detected current vector (i _δ , i _u ) and voltage vector (u _δ , u _u ),
  characterized,
• c1) that the differences between the components of the detected voltage vector (u _δ , u _u ) and the components of a correction signal vector (u _{δ , b} , u _{u , b} ) are formed, and
• c2) with an integration time constant, which corresponds to the subtransient inductance (i _σ ), to be integrated into the respective component of the simulated current vector (i _{δ b} , i _{u b} ), and
• d1) that the differences between the components of the detected and the simulated current vector are formed and fed to the input of controllers ( 9, 11-13 ) which output the components of the correction signal vector (u _{δ , b} , u _{u , b} ) at their outputs , and
• d2) that the components of the correction signal vector (u _{δ , b} , u _{u , b} ) are used as the respective components of the stationary voltage vector (u _{δ 0} , u _{u 0} ).

2. The method according to claim 1, characterized in that P controllers are used as controllers. ( Fig. 5) 3. The method according to claim 1, characterized in that PI controllers are used as controllers. ( Fig. 6) 4. Modification of the method according to claim 3, characterized in that instead of the components of the correction signal vector (u _{w , b} , u _{u , b} ) the output _signals (u _{δ , bI} , u _{u , bI} ) of the integrators contained in the PI controllers ( 12, 13 ) can be used as components of the stationary voltage vector (u _{δ 0} , u _{u 0} ) ( Fig. 7). 5. The method according to any one of claims 1 to 4, characterized featured, that it is the rotating reference coordinate system (δ, u) around a field-oriented coordinate system acts. 6. The method according to any one of claims 1 to 4, characterized featured, that it is the rotating reference coordinate system (δ, u) around a rotor-oriented coordinate system acts.   7. The method according to any one of the preceding claims, characterized in that the components of the current vector are regulated in the reference coordinate system and the components of the output signal vector (u _{δ , corr} , u _{u , corr} ) of the current regulator ( 17, 18 ), the components of the stationary voltage vector (u _{δ 0} , u _{u 0} ) are added, whereby the components of the setpoint vector (u _{δ , should} , u _{u , should} ) of the stator voltages arise ( FIG. 10). 8. The method according to claim 7 and according to one of claims 3 or 4, characterized in that for current control instead of the measured current vector (i _w , i _u ) the simulated current vector (i _{δ b} , i _{u b} ) as the actual current vector (i _{δ ist} , i _{u ist} ) is used. 9. Modification of the method according to one of the preceding claims, in which a pulse-width-modulated converter is actuated as a power electronic actuator by the components of the set vector of the stator voltage (u _{δ , set} , u _{u , set} ), characterized in that

• a) _{that, to} the components of the target vector for the stator voltage (u _w, u _{u, soll)} the components of a voltage vector (u _{δ, nb,} u _u, are formed _nb),
• b) that the differences between the components of the voltage vector formed (u _{δ , nb} , u _{u , nb} ) and the components of the correction signal vector (u _{δ , b} , u _{u , b} ) are formed, and
• c) with an integration time constant which corresponds to the subtransient inductance (L _σ ) to the respective component of the simulated current vector (i _{δ , b} , i _{u , b} ).

10. The method according to any one of the preceding claims, characterized in that the components of the vector for the induced voltage (Ui _δ , Ui _u ) from the components of the stationary voltage vector (u _{δ 0} , u _{u 0} ) by subtracting the voltage drops at the leakage inductance (u _{δ L σ 0,} u _{u L σ 0)} and the stator resistance (u _{δ R0,} u _{u R0)} are determined, the voltage drops from the output of a guide system (7) components _to the desired current vector (i _δ, i _{u should} ) or from the components of the measured current vector (i _δ , i _u ) or from the components of the simulated current vector (i _{δ , b} , i _{u , b} ) and the parameter values for the leakage inductance and the stator resistance ( FIG. 14) . 11. The method according to claim 10, characterized in that the rotating reference coordinate system is a field-oriented coordinate system output by the guidance system, the angular velocity of which is determined by adding the measured rotor speed and a slip speed, the slip speed being determined as follows:

• a) The field-forming component of the target current vector (i _{δ , soll} ) is delayed by the rotor time constant (T _{r δ} ) effective in the longitudinal direction and then multiplied by the rotor time constant (T _{r u} ) effective in the transverse direction,
• b) the torque-forming component of the target current vector (i _{u soll} ) is divided by the product obtained, and gives the uncorrected slip speed (ω _{R, FS} ),
• c) A correction variable for the slip speed (ω _{R, corr} ) is formed by dividing the component (u _{i δ} ) weighted by a factor K _ω <0 by the field-parallel component (u _{i u} ) of the induced voltage vector and by the uncorrected slip frequency (ω _{R, Fs} ) subtracted,
• d) the correction quantity (ω _{R, corr} ) is set to zero if a binary support criterion (crit _w ) has the logical value zero. ( Fig. 15)

12. The method according to any one of claims 10 or 11, characterized in that

• a) that the rotating reference coordinate system is a field-oriented coordinate system output by the guide system ( 7 ), and
• b) the amount of the induced voltage vector (u _{i δ} , u _{i u} ) is divided by the angular velocity (ω _s ) of the field-oriented system, and the quotient is used as the actual flow chaining value (Ψ _n ), in particular for flow control, and
• c) that instead of the amount of the induced voltage vector (ui _δ , ui _u ), its component perpendicular to the field (ui _u ) can also be used as a counter of the quotient introduced in b).

13. The method according to claim 12, characterized in that the Flußverkettungsistwert (Ψ _n) is a model with the magnetic target current component (i _{δ soll)} modeled as an input variable, when a binary support criterion (crit _m) the logical value zero (FIG. 16). 14. Device for field-oriented control of asynchronous machines

• a1) with measuring devices for the stator currents and stator voltages,
• a2) with coordinate converters (KW1, KW2), which form the components of correspondingly detected vectors in a rotating reference coordinate system (δ, u),
• b) with a model circuit which, from the components of the detected voltage vector (u _δ , u _u ) and the detected current vector (i _δ , i _u ), the components of the stationary voltage vector required in the steady state (u _{δ 0} , u _{u 0} ) determined

characterized in that the model circuit has:

• c) integrator ( 8, 10 ) with an integration time constant which corresponds to the subtransient inductance to which the differences between the components of the detected voltage vector (u _δ , u _u ) and a correction signal vector (u _{δ , b} , u _{u , b} ) are supplied and whose output signals represent the components of a simulated current vector (i _{δ , b} , i _{u , b} ), and
• d) Controllers ( 9, 11-13 ), to which the differences between the components of the detected (i _δ , i _u ) and the simulated current vector (i _{δ , b} , i _{u , b} ) are supplied and the respective ones at their outputs Output components of the correction signal vector (u _{δ , b} , u _{u , b} ), which are then used as components of the stationary voltage vector (u _{δ 0} , u _{u 0} ).

15. The apparatus according to claim 14, characterized in that P controllers are used as controllers ( Fig. 5). 16. The apparatus according to claim 14, characterized in that PI controllers are used as controllers ( Fig. 6). 17. Modification of the device according to claim 16, characterized in that instead of the components of the correction signal vector (u _{w , b} , u _{u , b} ) the output _signals (u _{δ , bI} , u _{u , bI} ) of the integrators contained in the PI controllers ( 12, 13 ) can be used as components of the stationary voltage vector (u _{δ 0} , u _{u 0} ) ( Fig. 7). 18. Device according to one of claims 13-17, characterized by, current regulator ( 17, 18 ), which will regulate the components of the current vector in the reference coordinate system and for their output signals (u _{δ , corr} , u _{u , corr} ) the components of the stationary voltage vector (u _{δ 0} , u _{u 0} ) are added, whereby the components of the setpoint vector (u _{δ , should} , u _{u , should} ) of the stator voltages arise ( FIG. 10). 19. The apparatus of claim 18 and according to one of claims 16 or 17, characterized in that for the current control instead of the measured current vector (i _δ , i _u ), the simulated current vector (i _{δ b} , i _{u b} ) as the actual current vector (i _{δ ist} , i _{u ist} ) is used. 20. Modification of the device according to one of claims 14-19, with a pulse-width modulated converter as a power electronic actuator which _is controlled by the components of the target vector of the stator voltage (u _{δ , should} , u _{u , should} ), characterized in that

• a) components (u _{δ , nb} , u _{u , nb} ) of a voltage vector are formed from the components (u _{δ , should} , u _{u , should} ) of the target vector for the stator voltage,
• b) that the integrators ( 8, 10 ) are supplied with the differences between the component of the simulated voltage vector (u _{δ , nb} , u _{u , nb} ) and the components of the correction signal vector (u _{δ , b} , u _{u , b} ).

21. Device according to one of claims 14-20, characterized by subtracting elements, the components of the stationary voltage vector (u _{δ 0} , u _{u 0} ), the voltage drops across the stator resistance (u _{δ , R0} , u _{u , R0} ) and leakage inductance ( u _{δ L σ 0,} u _{u L σ 0),} subtract the voltage drops from the output of a guide system (7) components _to the setpoint current vector (u _δ, u _{u soll)} or for the components of the measured current vector (i _w, i _u ) or from the components of the simulated current vector (i _{δ , b} , i _{u , b} ) and the parameter values for the leakage inductance and the stator resistance ( FIG. 14). 22. The apparatus according to claim 21,

• a) a field-oriented coordinate system output by the guidance system ( 7 ) being used as the reference coordinate system (δ, u),
  marked by
• b) a delay element (30) of the first order with one of the active in the longitudinal direction of the rotor time constant corresponding time constant, the field-forming component of the desired current vector (i _{δ,) is to} be supplied to corresponding one of the delay element (30) downstream proportional amplifier with one of the active in the transverse direction of the rotor time constant Gain factor,
• c) with a divider to which the torque-forming component of the target current vector (i _{u soll} ) is supplied as a divisor and the output of the proportional amplifier as a dividend, the output of the divisor representing the uncorrected slip speed (ω _{R, FS} ),
• d) with a further divider to which the field-parallel component (ui _δ ) of the induced voltage vector weighted by a factor K _ω <0 and the field-perpendicular component (ui _u ) of the induced voltage vector as a divisor is supplied, its output quantity being a correction value ( ω _{R, corr} ) represents the slip speed,
• e) with a subtractor which subtracts the correction variable (ω _{R, corr} ) from the uncorrected slip speed (ω _{R, FS} ) and thus simulates the slip speed (ω _R ),
• f) with a switch in the output line of the further divider, which switches the correction value (ω _{R, corr} ) to zero if a binary support criterion (crit _ω ) has the logical value zero ( FIG. 15).

23. The device according to one of claims 21 or 22,

• a) a field-oriented coordinate system output by the guidance system ( 7 ) being used as the reference coordinate system (δ, u),
  marked by
• b) a divider to which the amount (u _i ) of the induced voltage vector as a dividend and the angular velocity (ω _s ) of the field-oriented coordinate system as a divisor is supplied, the output of which as the actual flow chaining value (Ψ _n ), in particular for the flow control ( 32 ), is used, whereby instead of the amount u _i , only the component perpendicular to the field (ui _u ) of the vector of the induced voltages may be supplied ( FIG. 16).

24. The device according to claim 23, characterized by a model with the magnetizing component (i _{δ soll} ) of the target current vector as an input variable which simulates the actual flow chaining value (Ψ _n ), and by a switch which switches the actual flow value from the divider to the model, if a binary support criterion (crit _m ) has the logical value zero.