DE2543706A1 - Quadraphonic transmission over F.M. source - provides amplitude has phase correlation using auxiliary carrier to provide phase relationships - Google Patents

Abstract

The transmission of a quadraphonic programme over a frequency modulated radio transmitter uses a quadraphonic coder to process the four signals over preamplifiers consisting of impedance converters, level regulators, frequency dependent amplifiers with an RC time constant of 50 microseconds and a phase inverter. 180 deg. outputs relative to the input signal are generated, summed and transmitted over an intermediate amplifier. After delay the signals are processed in a passive summing matrix. The four 0 deg. outputs are modulated with a 38kHz carrier but with different phases. This auxiliary carrier is generated by doubling the output of a 19 kHz oscillator and is introduced over transistor stages ensuring phase relationships of O (plus-or-minus) 45 deg. and 180 deg. (plus-or-minus) 45 deg..

Description

COMPATIBLE METHOD OF TRANSFERRING QUADROFONER PRO-GRAMS FREQUENCY MODULAR BROADCASTING TRANSMITTERS.

The invention relates to a compatible method for the transmission of quadrophones Programs on frequency-modulated radio stations using one of the four original channels composed of a main signal, a directional signal with suppressed subcarrier and a pilot signal for regeneration on the receiver side of the undressed subcarrier0 With quadrophone transmission, everyone should be in the recording room Sound impressions affecting a listener, be it coming from the sound sources direct sound, be it indirect sound caused by reflections, if possible projected true to the direction in the playback room.

Since the sound coming from above and below gives little musical impression contributes, it is sufficient to set up one loudspeaker in each side edge of the playback room. Are these four speakers with correlated low frequency alternating voltages fed, so phantom sound sources can be in the plane of the four loudspeakers produce.

The transmission of the required for each individual sound source to be reproduced Amplitude and phase correlations between the four playback speakers the purpose of the procedure.

Such a transmission method is subject to requirements relating to a) its compatibility and b) its quadrophone properties: aa) Each mono receiver should have all four channels as well as all intermediate positions of the quadrophones Play broadcast with equal rights.

ab) Every stereo receiver should transmit the quadrofone in the usual way Reproduce quality with full base width stereophonic.

ac) The quadrofone transmission should not have a larger bandwidth in the radio frequency spectrum as a stereophonic broadcast of the same program material. Given of the 100 kHz grid used in Europe for the broadcasting station frequencies in the area II would otherwise result in an increase in interference with reception from adjacent channel transmitters the quadrophone modulated transmitter as well as an increase in interference with the quadrophone Reception by adjacent channel transmitters is unavoidable. In addition, the signal-to-noise ratio is basically worse with increasing reception bandwidth.

ad) In particular, frequencies around 57 kHz must not be used in the multiplex signal occur because those frequency-modulated radio stations in Germany that Scatter traffic announcements into their programs, always a special pilot tone of 57 kHz with a deviation of + 3.75 kHz so that they can be found by a Optical display or similar on the vehicle receiver that responds to this frequency to facilitate.

ae) In the multiplex signal, apart from that for monaural reception Any sum signal that is necessary anyway, only difference signals occur, since these are relative are low in energy and therefore of the available lift range (without pilot tone approx. + 65 kHz) only take up a small part, which is necessary for mono reception (remote reception) get lost.

ba) The reproducible sound incidence devices must cover the entire Include angle range from Oo to 3600 without any gaps.

bb) In all points of an imaginary circular line around the listener Phanton sound sources not only one after the other, but also simultaneously can appear, otherwise spatial shifts when new sound events occur appear.

bc) Phantom sound sources should also be used in as many points as possible Connection surface of the four loudspeakers through their simultaneous interaction can be located.

bd) The reproduction of sound sources that is not related to the room in only one loudspeaker at a time, i.e. a four-fold monaural reproduction not the intentions of the Quadrofonie, but is intended for special effects within quadrophone productions are also possible.

be) With regard to crosstalk attenuation, all channels and the acoustic centers between them behave point-symmetrically to the center, although the Ability of the human hearing to recognize direction only axially symmetrical to Line connecting the nose and the back of the head and less pronounced to the rear than to the front is. If you adjust the crosstalk attenuation of this physiological one with the quadrophone transmission Imperfection, the acoustic impression of the room collapses as soon as the Listener turns head 600 or more.

bf) The crosstalk attenuation between front-center and back-center as well between left-side and right-side must be constant and large (20 dB), i.e., if any two adjacent speakers have an in-phase and equal emit a strong signal (middle impression), the same signal must be in the other two Loudspeakers must be attenuated by at least 20 dB at all times. From this criterion The width of the reproducible spatial impression depends crucially on O Should a sound impression be created in a corner, so the crosstalk attenuation can increase the other three speakers will be lower (approx. 10 dB) because of the acoustic fixation a perceived sound source to a loudspeaker is not aimed at anyway will.

bg) The processing of the low frequency signal voltages must neither frequency-dependent in the transmitter and in the receiver within the transmission range take place, for example, when using 90 ° phase shifters in the Low frequency paths would be inevitable. Otherwise there will be point-like sound sources reproduced widened and unstable.

bh) The method is to offer four signals for reproduction that are used for Radiation through loudspeakers installed in the side edges of the playback room are determined.

In the attempts made since then, the radio transmitter quadrofon To modulate, one can use quadrofoner just as with storage and duplication Performances on records distinguish between two different groups of procedures: the discrete methods and the "matrix" methods.

In the case of the "discrete" (from English discrete = separate, individual) method - according to what is necessary for reception with mono and stereo receivers Matrices - four channels of information independently transmit the are available again with good channel separation after dematrixing. In the USA is using the system developed by Louis Dorren and named after him experimented in Japan with the one developed by JVC and Matsushita "CD-4" system (Funkschau 7/1973 p. 219f.) A Both methods require a bandwidth of the multiplex signal of 91 kHz and meet hence the requirements ac), which almost excludes their use in Europe. The Japanese met furthermore, the requirement ad) does not, and both methods meet the requirement ab) only partially, as according to a negative opinion published above of the Institute for Broadcasting Technology in Hamburg contributed a number of stereo receivers Quadrophone transmissions would be impermissibly disturbed "(because of the additional signals above the normal stereo multiplex range).

The expansion of the bandwidth of the multiplex signal should be avoided only the application of a so-called matrix method remained. It is on the existing modulators and demodulators have not changed anything, the transmission remains so two-channel. "Matrix" is understood here to mean an analog computing circuit, which the four recorded channels by multiplying and adding or subtracting their instantaneous values are reduced to two transmission channels, from which one "Dematrix" working with the same factors behind the receiver demodulator die four playback channels can be derived.

This so-called "4-2-4 system" is explained below using the example of the "SQ system" (SQ = stereo quadraphonic) developed by Columbia Records, USA and Sony, Japan, which is currently the most widely propagated example. Radio stations are also experimenting with in the USA. In all transfer equations, the current value of the left rear receiving channel is denoted by w, that of the left front with x, that of the right front with y and that of the right rear with z. w ', x', y ', z' mean the instantaneous values of the corresponding derived playback channels. The factor + j or -j means a broadband phase shift around 1900. The factor 0.7 is an abbreviation for L and R are the instantaneous values of the alternating voltages in the transmission channels.

The two front channels remain unchanged during SQ encoding. The two rear channels become after a reduction of 3 dB and after a 90 ° phase shift one of the rear channels mixed with each of the front channels: L = -j017w + x + 0.7z R = -0.7w + y + j0.7z The stereo reproduction of a program encoded in this way sounds satisfactory. All channels are reproduced with the same overall power0 Due to the phase shift, the rear channels appear to the left or right.

to the right of the center of the base. All the impulses in the rear Channels occur, but are due to the frequency-dependent delay times of the phase shifter more polished than with a bad dynamic speaker.

However, in mono playback, there is a cancellation.

Sound sources in the middle at the back do not result in a sum signal: w = z = 0.7a (instantaneous value) L = - j0.5w + 0.5z R = - 0.5w + jO, 5z L + R = o Da in the middle In the back of many SQ recordings, an instrument is placed or reverberation mixed in the procedure does not meet requirement aa) 0 This applies to all known 4-2-4 matrix systems.

The dematrix in the SQ "decoder" routes the transmitted signals L and R continues unchanged to the front playback channels: x '= L or x '= -j0.7w + x + 0.7z y' = R or yt = -0.7w + y + j0.7z The rear channels w ' and z ', on the other hand, are removed from the transmission channels by attenuation by 3 dB and phase rotation derived: w '= - (- jO, 7L) - 0.7R or w' = w + jO, 7x - 0.7y z '= 0.7L - jO, 7R or z '= 0.7x - j0.7y + z It can be seen that the crosstalk attenuation is only between the front Channels x and y is large. Between the rear canals w and z it is after the Equations just as large, but this only applies to the frequencies at which the Phase rotation is exactly 900. Since each of the rear channels twice (with encoding and decoding) passes through a phase shifter and there a phase shifter with exactly 900 shift over 9 octaves is not feasible, occurs between the rear channels always have frequency-dependent crosstalk. Also arise frequency-dependent due to the different processing of the front and rear channels Runtime differences. As can also be seen from the equations, the is Crosstalk attenuation between front and rear channels as well as diagonally opposite channels Channels only 3 dB. With a signal arising on the left side on the recording side the instantaneous value w = x = 0.7b, the following components get into the right channels: Left: = = -0.5b w '= 0.7b + j0.5b z' = 0.5b x '= -0.5b + 0.7b crosstalk attenuation between left side and right side: 4.8 dB.

With a signal arising in the middle in front on the recording side the instantaneous value x = y = 0.7c, the following proportions get into the rear channels: w '= jO, 5c-- 0.5c z' = 0.5c - jO, 5c Add the two parts that are talking about vectorially to a part of c shifted by +1350 and a part of -450 the amplitude value 0.7. Apart from the phase shift, it is between Center front and center back the crosstalk attenuation O dB.

Other matrix methods bring it up to 8 dB here, but is then the crosstalk attenuation between the front channels only 3 dB (Scheiber "dimension 4 ", Sansui" QS "=" RM'M.

To improve these extremely bad values, included the more expensive SQ "decoders" behind the dematrix still so-called "logic circuits" (Four-Channel Sound "," High-Fidelity ", special issue, p. 42), which are intended to to pick out the currently unwanted channels during playback and continue to dampen.

The quieter useful signals (e.g.

Room reverberation), which are contained in the currently unwanted channels, of course with affected. If there are several primary sound sources, jump when playing the quieter when using the loudest apparent to others Places. This effect is so strong that it is different from the well-known Haas effect of the human Auditory, according to which the hearing is a sound source according to the direction of incidence of the first Wavefront locates, cannot be concealed sufficiently. Through this "logical Circuits ", the matrix" decoders "become very complex.

In summary it must be stated that the matrix method "SQ" does not meet the requirements aa), bb), bc), be), bf) and bg)! All other Matrix methods produce similar results.

Furthermore, a suggestion by Gerd Mallon is known that is practical is a "4-3-4" matrix method.

The third information channel is using quadrature modulation transmitted on 38 kHz. This method is much more useful than that previously mentioned procedure. Its main disadvantages are that in addition In addition to the stereo signals in the multiplex signal, another sum signal (w + z) occurs and must be transferred, which naturally leads to a noticeable reduction of the travel available for the mono signal and thus to a narrowing the coverage radius of such a modulated transmitter must lead. Through that If the transmitter power is to be increased, the interference from neighboring and co-channel transmitters increases but on again.

In addition, playback channels are generated in the said method, one installation of the loudspeakers in the middle of the side wall of the playback room This means that the impression of space is narrowed. The crosstalk attenuation to the each neighboring speaker is only 6 dB, to the opposite one Loudspeaker, however, it is great. Overall, the process therefore meets the requirements ae), bd) and bh) not and the requirement bf) only partially.

Finally, the variant of the Cooper'scsn UMX method (designated by Nippon Columbia as UD-4), which is suitable for radio broadcasts, does not meet the nTMX "requirements bd) and bg) and offers above 3000 Hz. The object of the invention is to provide a To create transmission methods for quad-financial programs that meet all of the requirements listed.

This object is achieved according to the invention in that the phase angle each component of the direction signal during the positive half-waves of this component causing low frequency AC voltage relative to the constant phase angle the subcarrier oscillation regenerated in the receiver with the relative to a fixed one Reference direction measured acoustic angle of incidence of the sound, which this part of the direction signal coincides, and that the ratio of the amplitude this proportion of the directional signal to the amplitude of the same sound source caused portion of the main signal a measure of the distance of the causative Sound source from the center of the acoustically transmitted sound source local area is.

To implement the angle of incidence of the emanating from a sound source Sound relative to the assumed reference direction "Left Sideways" in the equally large phase angle of the part of the directional signal that is correlated to this sound relative to the phase position of the subcarrier oscillation assumed to be 0 °, according to a further embodiment of the invention, the four diagonal directions of sound incidence LV, RV, LH, RH, as in the connection level of all four microphones compared to the as 00 assumed direction left Laterally crooked, perpendicular to each other in the Coordinates intersecting the center of the room and components of the subcarrier oscillation assigned which has the same phase angle compared to the phase angle assumed as Oo of the original subcarrier oscillation like the spatial coordinates.

That means the output voltage of the front left microphone the amplitude of one of the subcarrier oscillation leading by 45 # Component, the output voltage of the front right microphone the amplitude of a of the subcarrier oscillation by 1350 leading component, the output voltage of the rear left microphone the amplitude of one of the subcarrier oscillation lagging by 45 ° Component and the output voltage of the rear right microphone is the amplitude modulates one of the subcarrier oscillation lagging component by 1350 so that the subcarrier components are suppressed in the modulation results.

The angular frequency of the subcarrier oscillation 211 is denoted. 38000 s-1 with #, then applies to the chronological sequence of the four modulation results mx, my, mw and m z due to the phase positions of the assigned subcarrier components: m (t) = x.sin shows + 450) my (t) = -y.sin (, zt - 450) mw (t) = w.sin (#t - 450) mz (t) = -z.sin (c.J t + 45 °) Since the subcarrier component for the modulator of the front right channel or the modulator of the rear right channel a phase shift from 1800 versus the subcarrier component for the modulator of the rear left Channel or the modulator of the front left channel, you come with only two modulators produce the same result by adding the modulator of the rear left Channel or the modulator of the front left channel also the output voltage of the front right or rear right microphone with 1800 turned Phase supplies. These phase rotations occur at the inputs of the two Modulators two differential signals from the output voltages of two each other diagonally opposite microphones.

The expressions for the time course of the modulation results of the two modulators are then: m (w-y) (t) = (w - y) sin (#t - 45 °) m (x-z) (t) = (x-z) sin (# t + 45 °) m (x-z) (t) = (x - z) sin (, t + 45 Die Summation of the modulation results delivers the directional signal, which with its relative amplitude is the amount of displacement of the causative sound source from the center of the connecting plane of the four microphones, with its relative phase angle defines the direction of this shift. This direction signal therefore defines the location of the sound source in the manner of a vector with polar coordinates.

The combined polar coordinate-modulated direction signal R, after an increase to be explained by 3 dB, has the amount: With the phase position assumed to be 0 ° for the left-side direction, the direction signal forms the phase angle x, which is calculated according to m (xz) tan (α + 45 °) = m (xy). It lies. for m (xz)>0> 0 and m (wy)> 0 <0 in the range -45 ° <α <+ 45 ° + 45 ° <α <+ 135 ° for | m (xz) 6 0 - and | m (wy) <0> 0 in the range | + 135 ° <α <+ 225 ° 1350% -45 (= -135 °) The sum signal and pilot tone must now be added to the direction signal R in order to obtain the complete, compatible quadraphonic multiplex signal Q.

The pilot tone P with half the carrier frequency, the phase position 0 ° and the constant amplitude c has the time course P (t) = c sin 2t The entire quadraphonic multiplex signal Q has (taking into account a compatibility-related increase in the directional component by 3 dB, which will be explained below ) now the following temporal delay: Q (t) = w + x + y + z + c sin + (wy) / 2 sin (# t-45 °) + (xz) 2 sin tJt + 450) Furthermore it is It is still possible to use two electronic changeover switches synchronized by the subcarrier components instead of the modulators, each of which has a diagonal pair of channels connected to its inputs and whose output voltage sum together with the pilot tone already forms the complete multiplex signal, from which the mixed products from the low-frequency voltages and the generated by the Switching harmonic waves of the subcarrier components still have to be filtered out by a phase-pure low-pass filter. The conceptual derivation of this possibility of obtaining the multiplex signal by carrier-synchronous switching is given in the description of exemplary embodiments of the invention; If sound sources in the middle of each side contain the same proportions of the alternating voltages corresponding to these sound sources in two transmission channels in phase, and if the amplitudes of these proportions are in both channels, then they add up in the sum signal M due to their in phase The two carrier-frequency modulation results generated by the low-frequency AC voltage components with the same level and phase, however, always differ from one another in their phase position by 900, since the carrier components also have this phase difference.

As a result, geometrical addition occurs when the direction signal is generated from the two modulation results: Since the amplitude maxima of the modulation results are equal to the amplitudes of the modulating low-frequency alternating voltages due to the sine functions, one can also describe, if only the amount of the received directional signal is of interest: R = = a It can be seen that the sum signal in relation to the direction signal around is weaker. Although this does not affect the reciprocal to the coding, the quadrofone decoding, it does, however, affect the compatibility for two-channel reception. If there is a sound source on the left or

on the right side, the stereo receiver should only take it out the left or right speaker. Needed for this However, it is a directional signal with the same amplitude as the corresponding sum signal.

For this reason, according to a further embodiment of the invention the amplitudes of the difference signals before their modulation by 3 dB compared to the amplitude of the sum signal increased, with which the for sound sources in the middle of the page by the geometric addition of the modulation results with simultaneous linear addition of the associated sum signal components, the relative direction signal amplitudes are reduced by 3 dB, which would reduce the crosstalk attenuation with two-channel reception, is counteracted. In a time-division multiplex coder in which the difference signals do not occur alone, the same effect is achieved by the fact that before the synchronous To switch each channel, an anti-phase component attenuated by a factor of 6 (15.6 dB) is added from the diagonally opposite channel. This will be in-phase components weakened to 5/6 times, out-of-phase components to the 7 times raised. This results in an increase in the difference component by 7/6 20 log = 2.9 dB 5 6 Although this is in front of the switch inputs of a time division multiplex coder a matrix is required, the typical disadvantage of the actual matrix coder, process different parts of the overall multiplex signal differently and consequently a runtime compensation for the Sum signal to need, nevertheless bypassed, since the here required Matrix for reducing the total proportions acts on all channels in the same way.

At the receiving location, each component present in the sum signal is to be used with the help the polar coordinates transmitted in the direction signal for this as well as any other part as a phantom sound source in the point of the reproduction plane defined by the coordinates be moved. Since the summing circuits required for this only have amplitudes put together, the polar coordinates of the direction signal must first be in a linear ## es coordinate system can be transformed. This is done by following a further development of the invention in the receiver by simultaneous demodulation of the direction signal with two opposite the phase position of the regenerated subcarrier by -45 ° or

+ 45 ° phase-shifted subcarrier components the two from each other independent difference signals (w-y) and (x-z) are recovered, respectively belong to pairs of sound directions of incidence whose connecting lines are in the Cut the middle of the room at a right angle, and the four summing circuits with the Sum signal can each be put together with such factors that the four original signals largely separated from each other again.

The composition of the individually present in the decoder after demodulation Transmission signals M = w + x + y + z D-45 ° = z (w - y) D + 45 ° = # 2 (x - z) to the playback channel signals w ', x', y 'and z' are done by four simple arithmetic operations: w '= 1/3 (M + D-45 ° # 2) x '= 1/3 (M + D + 45 ° # 2) y' = 1/3 (M - D-45 ° # 2) z '= 1 / 3 (M - D + 45 ° # 2) By inserting the original signals w, x, y, z into them Expressions one obtains the transfer equations of the method without taking into account the compensation to be described for the remaining crosstalk: = w + x - y + z 3 3 3 x '= w / 3 + x + y / 3 - z / 3 y' = w + x 3 z '= w / 3 - x / 3 + y / 3 + z Das the same result is achieved with a time division multiplex decoder in which two electronic changeover switch by components of the regenerated subcarrier oscillation are switched synchronously, which lag or lead this oscillation by 450 each. The multiplex signal is connected to the inputs of the switch; in each of its four output voltages, the alternating voltage is predominantly one of the four transmission channels included. The same crosstalk attenuation as with decoding To obtain according to the matrix principle, the difference between the four Output signals can be increased by 3 dB, which is achieved here by adding a factor 6 reduced anti-phase proportion of the respective diagonal channel to each of the four output alternating voltages of the electronic switch is achieved. Even though thus a matrix is required behind the switch outputs of a time division multiplex decoder is, the typical disadvantage of the actual matrix decoder is different proportions of the entire multiplex signal to be processed differently and consequently at to require a delay compensation for the sum signal to the highest quality standards, nevertheless circumvented, since the matrix required here to reduce the sum shares acts on all channels in the same way.

When using the invention as indicated so far lie the simultaneously reproducible phantom sound sources on the circumference and within the area of a circle, its diameter with the distance between neighboring loudspeakers matches. In order to also meet the requirement bd), you can according to a further embodiment of the invention in the receiver between the four summing circuits and four additional compensation stages are added to the four low-frequency amplifiers, the DC voltages derived from the low frequency signals as required opened or closed individually.

Each compensation stage leads the adjacent channels in phase opposition, the diagonally opposite channel in-phase components of the assigned Summing circuit output low-frequency voltage with such a dimensioned Amplitude to that the remaining talk about to the other three channels is fully compensated.

It is important that each compensation level is always except Operation is set if the assigned channel does not currently have any concise sound information transmits, because otherwise only talking from other channels over the compensation lines would be passed on to other channels.

To achieve this closing and opening of the compensation stages, the levels of the channel signals supplied by the decoder are displayed in four comparators continuously with the level of the expected crosstalk, represented by the decreased channel sum, compared. If the level of one or more channels is not high clearly above the level caused by the crosstalk in these channels, so the assigned comparators switch over and set the compensation levels their channels out of order. Thereby the compensation conditions for the other channels significantly improved.

The effect of the compensation stages is based on the fact that the transfer equations can be influenced dynamically in such a way that they are always optimal for the current program situation are adapted.

If one assumes the channel signals supplied by the decoder w '= w + x / 3 - y / 3 + z / 3 (1) x '= w / 3 + x + y / 3 - z / 3 (2) y' = - w / 3 + x / 3 + y + z / 3 (3) z '= w / 3 - x / 3 + y / 3 + z (4) and forms from it to compensate for crosstalk the following compensation signal components for the individual channels: kw = (5) 3 x 'kx = (6) 3 k @ = (7) 3 z 'kz = (8) 3 then these components become the channel signals as follows superimposed, which gives the "compensated" channel signals: wk = w '- kx + ky - kz (9) xk = x '- kw - ky + kz (10) yk = y' + kw - kx - kz (11) zk = z '- kw + kx - ky (12) After inserting equations (1) ... (4) and (5) ... (8) into (9) ... (12) you can see that the crosstalk attenuation has not yet been changed: wk = 2 / 3w + 2 / 9x - 2 / 9y + 2 / 9z (13) xk = 2 / 9w + 2 / 3x + 2 / 9y - 2 / 9z (14) 2 +2 2 2 15 yk = w + x - y - z (1 @) 9 9 3 9 zk = 2 / 9w - 2 / 9x + 2 / 9y + 2 / 3z (16) + makes sense the compensation only if it does not work between all channels at the same time. Sinks in one of the channel signals supplied by the decoder, the level is almost equal to the crosstalk from the other channels, it means that this channel does not currently contain any distinctive modulation of its own and consequently only can cause no noticeable crosstalk into the other channels. Consequently the compensation signal emanating from this channel can also be switched off without further ado will. However, this switch-off has a beneficial effect on playback in all four channels.

Example: Right in front briefly rest (w = a; x = y = 0; z = d) playback without compensation: w '= a + b / 3 + d / 3 x' = a / 3 + b - d / 3 y '= a + b + d 3 3 3 z' = a / 3 - b / 3 + d During the compensation, the signal component ky = y / 3 is now switched off, because it cannot contain any useful signal because of y = 0. One obtains: + To the volume drop To prevent this, in-phase signal components are also added from each compensation stage added to own channel. This also has an effect with only one active channel still positive, because the lack of crosstalk also subjectively means the perceived Volume drops.

wk = w '- kx - kz wk = 7 / 9a + 1 / 9b + 1 / 9d xk = x' - kw + kz xk = 1 / 9a + 7 / 9b - 1 / 9d yk = y '+ kw - kx - kz yk = -2 / 9a + 2 / 9b + 2 / 9d zk = z' - kw + kx zk = 1 / 9a - 1 / 9b - 7 / 9d It can be seen that the crosstalk attenuation between all three information-carrying channels from 9.6 dB to the remarkable value of 17 dB increases! The crosstalk attenuation is equal to the channel that is currently not modulated now 11 dB. "Artificially" set this value by lowering or even switching off this channel improvement is not recommended. With every transmission of natural sound events This is because even a channel that is currently not "present" contains low-level sound information, their intermittent attenuation leads to the destruction of the natural sound field would have to lead. The best example of this are those with modulation-controlled automatic level controls (so-called "logic circuits") equipped more expensive "decoders" for SQ disks.

In the practice of musical performances, times come much more frequently before, to which only two or even only one of the four channels one for the auditory Direction determination contains decisive concise part of the program.

Example for two dominant signals: w = 0 x = b y = c z = 0 (fade in a stereo base from the front.) The decoder delivers: w '= b / 3 - c / 3 y' = b / 3 + c x ' = b + c / 3 z '= -b / 3 + c / 3 3 3 During the compensation, the signal components are now kw and kz switched off because they do not contain any useful signals, but now crosstalk: You get: wk = w '- kx + ky wk = 1 / 9b - 1 / 9c xk = x' - ky xk = 8 / 9b 8 yk - y kx yk - c 9 zk = z '+ kx - ky zk = -1 / 9b + 1/9 between the two channels carrying information at the moment, the crosstalk compensation is therefore complete (in practice # 30 dB), the crosstalk attenuation is equal to both currently inactive channels 18 dB.

If there is only one dominant single signal at the moment, this only appears in the associated loudspeaker0 Example: w = O x = O y = 0 z = 0 The decoder delivers: w '= b / 3 y' = b / 3 x '= b Z' During compensation only the signal component kx is allowed to pass; kw, ky and kz are switched off because they only contain crosstalk.

wk = w '- kx wk = 0 xk = x' xk = x 'yk = y' - kx yk = 0 zk = z '+ kx zk = 0 The improvement in the crosstalk attenuation will be emphasized again is by no means achieved by lowering the level of currently unwanted channels, what always directions shifts of phantom sound sources when using an additional loud sound source would have to result.

Rather, there is real compensation through out-of-phase signal components, at the expense of the crosstalk attenuation - not the level - between and voh at the moment low-level channels. But there is a channel that has little its own modulation contains, anyway, no interfering crosstalk signals to others Channels, this remains inaudible. A high crosstalk attenuation is after the described method is always achieved where it is needed. At the Then only the directional perception is added to several new, dominant sound sources a temporally preceding phantom sound source a little more blurred without being changes its direction of incidence even in the slightest. Because of the Haas effect, this blurring will by no means be noticed.

The conditions are much more favorable than, for example, when locating the voice of a natural speaker in a room, with more than 80% of the sound energy come from other directions. In summary, it should be noted that the transfer Quadrophone sound recordings of all kinds over the entire described facility in no case audibly differentiated from the transmission via four completely separate channels can be. Without the compensation levels, this distinction is in use a four-microphone arrangement in the recording room is also not possible. Only with "synthetic Quadrofonie1, with completely incoherent channel signals, is for the experienced listener a noticeable difference, especially when you are at very unequal distances from the speakers.

The advantages achieved with the invention are in particular: that within the already needed for two-channel stereo transmission Frequency range a four-channel transmission is made possible, which is not possible with the human hearing noticeable deterioration in quality compared to the direct Listening to the broadcast four-channel sound carrier brings with it. The necessary Increase in the flow of information compared to stereo without simultaneous bandwidths increase is through the redundancy of the FCC stereo transmission method allows, which consists in the fact that two symmetrical to the suppressed subcarrier lying sidebands carry the same directional information.

Since the Dorren Quadraplex method and its Japanese variant after Opinion of the Institute for Broadcasting Technology, Hamburg, not used in Central Europe can be offered as alternatives to the described invention for quadrophones Broadcasts only use the matrix method (e.g. SQ), the Mallon method and TMX at. Compared to this method, the invention has three possible types of reception the following advantages: There are no compatibility problems with monaural reception. In SQ broadcasts, the rear channels would be greatly weakened here, rear center not reproduced at all by adding the SQ encoding equations can easily be proven to a sum signal.

In the case of stereophonic reception, the invention has the advantage that that the perceptible base width is independent of any deviations in the phase position of the subcarrier regenerated in the receiver compared to the original phase position of the subcarrier at the transmitter before it is suppressed. Such deviations occur in many stereo decoders due to aging of frequency-determining components and lead to a reduction in the channel separation. The associated reduction the perceptible base width disappears immediately when the transmitter according to the invention is modulated with a Quadrofonie program. The reason is that the multiplex signal now not only as with stereo components with the phase angles 0 ° and 1800 relative to the subcarrier phase position, but portions with all possible interphase angles contains, not only from arbitrarily placed primary sound sources, but from everything also from the sound reflections coming from all directions caused. In such a multiplex signal there are always parts the phase position relative to any subcarrier occurring in the stereo receiver the phase angle 0 ° or 1800 have and therefore good channel separation, so full Base width, result.

The two-channel reproduction of a transmission modulated according to the invention one must imagine that the circular area occupied by phantom sound sources into a line that coincides with the horizontal axis of the original Circular area forms an angle as large as the phase angle of the regenerated Subcarrier relative to the phase position of the original subcarrier before its suppression. In the case of phase-accurate subcarrier regeneration, the horizontal axis is thus obtained of the transmitted sound source circular area as a stereo basis.

The Mallon method and TMX have the property just described not, as it quadrature modulates two different signal combinations in the multiplex signal contain, but not a uniform amplitude- and phase-modulated directional signal with polar coordinate character. The matrix processes naturally have these properties not; the flow of information is no greater than with stereo broadcasts. About that In addition, the rear channels sound unnatural when playing an SQ broadcast on two channels, since they have a wider audio frequency range from both speakers constant phase shift of about 900. this is because of the frequency-dependent sound wavelengths with constant head dimensions not possible with natural hearing.

In the case of quadphone reception, the invention is within the limits maximum possible crosstalk attenuation of the available signal bandwidth if one assumes that there should be no preferred direction. The effort for the decoder without compensation steps corresponds roughly to the effort for a stereo decoder with downstream SQ dematrix. The achieved crosstalk attenuation is the lowest between the corners, where it is 9.6 dB, the diagonally opposite one Corner by antiphase radiation an acoustic correction effect like a Exercises base broadening. The matrix methods offer 3 dB to two corners and above 20 dB towards the rest of the corner. The sound sources wander from the corners to the center of the lake down, the crosstalk attenuation increases to the opposite sides from 9.6 dB to over 30 dB. With the SQ method, they decrease to 0 dB.

If the electronic effort is increased, the Invention, as already stated, very high crosstalk attenuation between the corners achieve.

In the case of the matrix method, the ratios can be the same Improve effort as well, but only for one currently dominant one at a time Sound source, with the remaining reproduced sound sources to run away ("shifting") tend. These disadvantages become apparent when using the compensation stages according to the invention avoided.

On the transfer equations realized with the invention without compensation stages w '= w + x / 3 - y / 3 + z / 3 (1) x' = w / 3 + x + y / 3 - z / 3 (2) w x z x '= - + + @ + (3) 3 z '= w / 3 - x / 3 + y / 3 + z (4) is shown in the following as the important center impressions can be reproduced simultaneously and without crosstalk.

The alternating voltage generated by a sound source in the front center, whose instantaneous amplitude value is to be referred to as Cf (center front) - as usual with stereo to achieve a mid-range impression - with 3 dB reduced Strength fed into the front channels x and y: x = y = 0.7 cf (5) The corresponding The value for a middle rear sound source (cb = enter back) is also included in the Return channels w and z fed in: w = z = 0.7 cb (6) The same applies to a left side (cl) and a right side (cr) to be localized sound source: w = x = 0.7 c1 (7) y = z = 0.7 cr (8) Finally, the one in the The sound source in the middle of the room caused alternating voltage with the instantaneous value c fed into the four channels in equal parts, to the same extent here Achieve overall performance when playing as from any other position using amplitude reduced by 6 dB: w = x = y = z = 0.5 c (9) The four recording channels now contain the following signals: w = 0.7 cb + 0.7 c1 + 0.5c (10) x = 0.7 Cf + 0.7 c1 + 0.5c (11) y = 0.7 Cf + 0.7 cr + 0.5c (12) z = 0.7 cb + 0.7 cr + 0.5c (13) Substituting the four recording channel state equations (10) through (13) into the Transmission equations (1) to (4) give the reproduction channel state equations (14) to (17): 4 4 4 x '= 3. (0.7 cb + 3.0.7 cl + 4.0.5c (15) y ' = 4/3 (0.7 cf + 0.7 cr + 0.5 c) 4 z '= 34 (0t7 cb + 0.7 cr + 0.5 c) 4 The factor 34 is irrelevant because it affects all signals 3 equally. One recognises, that neither the relative amplitudes of the center signals nor their distribution on the four channels have experienced any interference on the transmission path. This means a completely separate ("discrete") transmission of the for presence and spatial impression decisive center signals, one of the greatest advantages of the method. The sound tends to separate from the speakers.

Some embodiments of the invention are shown in the drawing and are described in more detail below. 1 shows the block diagram of the Quadrofonie coder with a separate amplitude modulator for each channel; 2 shows the vector diagram of the modulation result with simultaneous modulation with a signal left in front (solid line), a signal right side (dotted Lines), a signal center back (dashed lines) and a signal center Above (dash-dotted lines), where of all modulating signals your positive one Peak value is present; 3 shows the block diagram of an equivalent quadraphonic coder, with only two amplitude modulators that are controlled with differential signals; 4 shows the block diagram of a time division multiplex quadraphonic coder with two synchronized ones electronic switches; 5 shows the block diagram of a quadraphonic decoder according to the matrix principle with separate recovery of the two difference signals the encoder from the direction signal; 6 shows the block diagram of a Quadrofonie decoder using two integrated stereo decoders; Fig 7 the vector diagram of the sampling of the modulation results from FIG. 2 with the subcarrier components the decoder of Figure 5; 8 shows the vector diagram of the sampling of the modulation results from FIG. 2 with the subcarrier regenerated in the correct phase in a stereo decoder; 9 shows the block diagram of the compensation stages with automatic mono-stereo-quadro switching.

Fig. 1 shows the basic form of the Quadrofonie coder. From the entrances w, x, y, z (LH, LV, RV, RH) the four signals go to the preamplifiers (1). Each preamplifier consists of an impedance converter, a level adjuster, a Frequency-dependent negative feedback via an RC element with a time constant of 50 μs Amplifier, with which the standard pre-emphasis is achieved, and a phase inversion stage. Each preamplifier has two outputs, the alternating voltages with the phase positions Output 0 ° and 180 ° relative to the AC input voltage.

The output voltages come from the outputs, which are phase-rotated by 1800 to a passive summing matrix (2) to which an intermediate amplifier (3) with 12 dB gain and 180 ° phase shift follows. This eliminates the loss when adding up of four signals, which is 12 dB, balanced again. Then runs through the sum signal is one equipped with the integrated TCA 590 bucket chain control Runtime compensation stage (13), which by adjusting the runtime-determining clock frequency allows an exact compensation of the longer signal propagation time in the case of the difference signals. After the delay, the sum signal arrives at the passive multiplex matrix (4).

The output voltages arrive separately from the OO outputs four amplitude modulators (5). They are designed in such a way that only the sum and difference signals however, carrier and low frequency alternating voltages do not appear at their outputs.

The four modulators are all the same with carrier waves Frequency of 38 kHz, but operated with different phase positions, namely the modulator for the w-channel with -450, for the x-channel with +450, for the y-channel with +1350 and for the z-channel with +2250 = -1350 compared to the phase position of the original Subcarrier.

To generate these subcarrier components, the sinusoidal AC output voltage of a 19 kHz crystal oscillator (6), which has an adjustable Attenuator (7) also delivers the pilot tone to the multiplex matrix at the same time Full-wave rectification frequency doubled (8), the frequency doubler in turn has two outputs with the phase positions 00 or 0 1800. At each of these exits two transistor stages (9) are connected, which have an RC or CR element a phase shift of the subcarrier oscillation by +450 between the collector and emitter or -450. This results in subcarrier components with the phase positions 0 ° # 45 ° and 180 ° # 45 °.

Because of the better harmonic distortion, intermodulation and carrier suppression values it is recommended to use symmetrical switching modulators for the modulators.

In this case, the four sub-carrier components are limited by limitation and inversion formed into carrier-frequency symmetrical square waves, which serve to reverse the polarity of the applied low-frequency oscillations. However must then the non-suppressed sidebands from harmonics of the switching frequency and Low frequency can be suppressed in a filter.

The carrier-frequency output signals of the four modulators w, x, y and z (5) have positive instantaneous values of the modulating low-frequency oscillations the phase positions -45 °, +450, +1350 and -135 °, for their negative instantaneous values the Phases + 135 °, -135 °, -45 ° and + 45 °. They are in a summation matrix (10) summarized and then without phase shift by 15 dB to the finished directional signal reinforced (11). Of this, 12 dB are used to compensate for the summation losses and 3 dB for the Compensation for the addition loss is required, which arises because of side-center sound sources the two components of the directional signal generated by the modulators in question are not in phase, but with the same amplitudes to an oscillation medium phase position and only 3 dB increased amplitude. The direction signal now also arrives at the multiplex matrix (4), which is added to the main signal by adding and pilot tone supplies the finished quadraphonic multiplex signal. The broadband amplifier (12) still has to compensate for the summation losses (9.6 dB) on the passive multiplex matrix.

As a result of the selected phase angle of the carrier components Completely assembled directional signal, differential signal character. The ones too diagonal Modulators belonging to opposite channels work with antiphase Carrier components. For all in-phase signal components at their inputs the modulation results are therefore in opposite phase and stand out in the Summing up. Sound components to be reproduced in phase by all speakers therefore do not generate any direction signal at all. The phase angles are taken from the table the direction signals for positive AF instantaneous values and the relative amplitudes the main and directional signals for different directions of sound incidence.

Fig. 2 shows the creation of the direction signal in the vector diagram. For all modulating signals, as Instantaneous value the positive Peak value assumed. For compatibility with stereo signals, the Left Direction Lateral Reference direction from which the phase angle is clockwise Increase to + 180 ° and decrease counterclockwise to -180 °.

The left front signal (LV = x) is generated because of the increase in the direction signal a vector of length x \ l2 with a phase angle of +450 by 3 dB.

The Right Side (RV = RH = 0.7 cr) signal generates two vectors with the lengths cr and the phase angles +1350 and -1350, which become a vector with the length cr J and the phase angle add 1800.

The middle rear signal (LH = RH = 0.7cm) generates two vectors with the lengths cb and the phase angles -45 ° and -135 °, which result in a vector with Add the length cb # 2 and the phase angle -90 °.

The middle signal (LV = RV = RH = LH = 0.5 c) generates four vectors with the lengths 0.7 c and the phase angles + 45 °, + 135 °, -135 ° and -45 °, which are annihilate each other completely.

Due to the difference signal character of the output signals of two modulators belonging to diagonally lying channels one arrives at the same result, if instead of the modulators w and y (or x and z) only the modulator w (or x), to which the 180 ° phase-rotated low-frequency signal is also used of the channel y (or z) is supplied. The difference signal formation is thus before relocated the modulators. Two modulators (y and z) are thus omitted, as is the 1800 output of the 38 kHz frequency doubler and the two phase shifter stages connected to it. In addition, the summation matrix for the direction signal is superfluous, since the difference signals can be increased by 3 dB before the modulators. There are two additional Matrix circuits including the associated 6 dB amplifier necessary, because of the differences in the signal pathways, it is also recommended here to use the Let the sum signal go through a time-of-flight correction circuit o In this way one arrives at the matrix variant of the Quadrofoniecoders, which is shown in Fig. 3 in the block diagram is shown, the preamplifier as well as the formation and the path of the sum signal remain as described above up to the multiplex matrix. To form the difference signals On the one hand, the output AC voltages of the 0 ° output from the w preamplifier and the 180 ° output from the y-preamplifier, on the other hand the output alternating voltages the 0 ° output from the x preamplifier and the 180 ° output from the z preamplifier each added in a summing matrix (1), (2) and each in an amplifier (3) without phase rotation amplified to the raised difference signals vz (w-y) and ff (x-z). The reinforcement is 9 dB, because in addition to the compatibility-related increase by 3 dB, there is also a total loss of 6 dB must be compensated.

The generation of the subcarrier components for the two modulators (w-y) and (x-z) take place in the same way as described above for the modulators w and x. The use of symmetrical switching modulators is of course also recommended here, which, however, produce harmonic sidebands and therefore a phase-pure low-pass filter require.

A different method of generating the necessary for switching symmetrical carrier-frequency square wave components is described below described for the time division multiplex quadraphonic encoder. The multiplex matrix (4) receives the two modulation results, the sum signal and the pilot tone are supplied to the broadband and phase-pure output amplifier (5) compensates for the summation loss of 12 dB and makes the output low-resistance.

There is no separate presence of the directional signal. Is it needed alone, need it only sum and pilot input to be short-circuited in the multiplex matrix.

With a switching modulator, the input is constantly switched through to the output, only the low-frequency signal passing through is for the duration of each negative carrier half-wave reversed. The switching modulator is therefore a carrier-synchronous electronic switch, its output for the duration of the positive carrier oscillation half-waves with the not phase-shifted AC input voltage and for the duration of the negative carrier oscillation half-waves is connected to the AC input voltage rotated by 180 ° 0 The mean value the AC output voltage of such a switch is independent of the input amplitude always 0, i.e. the low frequency is not let through by the switch. That's why a separate path, the sum channel, must be provided in the matrix coder for the low frequency If, on the other hand, the electronic switches are interpreted in such a way that their inputs only during the positive subcarrier half-waves with the outputs without phase rotation are connected, while the negative subcarrier half-waves are connected to zero potential, thus a chopped low frequency oscillation is obtained, the mean value of which corresponds to the low frequency signal itself is proportional "So the low frequency has not disappeared behind the switch; it can be fully regenerated by sieving out the carrier frequency components. With four electronic switches, a coder is possible that does not have a master channel needs, because in this case the input w (or x) because of the 1800 phase shift the associated subcarrier components are switched through to the output for just as long is how the input y (resp.

x) is blocked, only two electronic switches can be used and design them as changeover switches that operate during the blocking time of the input w (or x) the entrance Switch y (or z) through to the output.

In this way one arrives at the time-division multiplex variant of the quadrofonie coder (Fig. 5). It is easier to master in practice than the matrix variant, since they have all the proportions of the four channels, regardless of their degree of correlation (i.e. the amount of their total shares), processed in exactly the same way, which means differences in runtime 0 modulators in the real sense with their distortion and Intermodulation problems are absent; the whole circuit consists only of linear ones Amplifiers and fast switches, however, the condition is the presence of one extremely high-quality low-pass filter that let through phase-pure up to 53 kHz and off 61 kHz must be blocked effectively, because this is where the lower sideband begins because of the mixing of the 2nd harmonic of the subcarrier with the low frequency Increase of the directional signal required for compatibility with stereo receivers however, 3 dB is more laborious to achieve here than with the matrix coder. In four summing circuits (2) (each with a 6 dB loss), the output becomes 0 a channel signal t taken from a preamplifier (1); from the 180 ° exit the channel signal taken from the diagonally assigned preamplifier is added. Afterward the total loss is compensated for with a 6 dB amplifier stage (3). After that, the total component in each channel signal is due to the out-of-phase admixture reduced to 5/6, the difference portion, however - because of the in-phase for him Admixture - increased to 7. However, 7: 5 corresponds to the ratio with sufficient accuracy 3 dB (more precisely 7.07: 5.00).

The four channel signals are sent from the amplifier outputs to the four inputs of the two changeover switches (4).

Your output signals are with the pilot tone in the multiplex matrix (5) summed to the quadraphonic multiplex signal, filtered in the low-pass filter (6) and reduced by 10 dB to compensate for the summation losses and by x dB to compensate for the attenuation of the filter with broadband end amplification.

The preparation of the 38 kHz square wave voltages required for synchronous switching can be done as described for the matrix coder, with the sinusoidal oscillations could be limited, but here is another possibility as an exemplary embodiment shown.

It is assumed that a much cheaper 76 kHz crystal is used, and the exactness of the rectangular symmetry, which is decisive for the crosstalk attenuation, Rise time and phase correlation of the two subcarrier components become rigid given by the circuit 0 Phase adjustment is only required for the pilot tone required, but this is off when deriving the sinusoidal pilot tone a square wave is indispensable anyway0 The 76 kHz crystal oscillator (7) has also a 0 ° and a 1800 output. The 76 kHz oscillations at both outputs are differentiated (8) and by frequency division in a T-flip-flop (9), (10) converted into the desired 38 kHz switching voltages with a steep edge. Because the flip-flops are differentiated from the rising edges of the 76 kHz oscillations are implemented, the symmetry is also rigidly given. The phase shift of the The output voltage of the two flip-flops after frequency division is exactly 900, however, it is still undetermined which of the oscillations leads 0 Da the x / z switch always requires a leading switching voltage, it must still be ensured that the flip-flop, which is used to generate the w / y switching voltage, only ever into the one Can tilt position in which it is used to generate the x / z switching voltage Flip-flop is already located at this moment. For this purpose the two are generated Switching voltages to an EXCLUSIVE-OR circuit (anti-equivalence -Circuit) (11), which only indicates a logical H "as long as both switching voltages do not have the same logical state. The output of this circuit is with connected to the preparatory input of the T-flip-flop that supplies the w / y switching voltage generated. As long as the above condition is met, the preparatory one is Input of the w / y flip-flop at the moment of the arrival of a rising edge on Trigger input is high so that the flip-flop can toggle 0 but has im The moment such a rising edge occurs, the w / y switching voltage already occurs the same logic state as the x / z switching voltage, it does not rush it at 900 after but before. By the logical 'kl ", which the EXCLUSIVE-OR circuit in this case applies to the preparatory input T of the w / y flip-flop, this retains Flip-flop remains in its position until the next 76 kHz rising edge occurs In the meantime, however, the x / z flip-flop has a constant on at its preparatory input H, received a rising edge and is flipped, the EXCLUSIVE-OR circuit returns a logical "H"; the next rising edge on the w / y flip-flop can do this So implement flip-flop. The correct phase position is thus restored. the EXCLUS IV-OR circuit is made up of the four NAND gates of an integrated circuit CD 4011 A, the two T flip-flops are the two JK flip-flops of the integrated Circuit CD-4027 A, whose preparatory J and K inputs are each controlled together will.

With the help of a simple, equipped with two transistors and with two diode-controlled AC flip-flops (12) is still the 38 kHz x / z switching voltage divided down to a 19 kHz square-wave voltageç The subsequent amplifier (13) contains a 19 kHz low-pass filter and an adjustable one Phase shifter, with the pilot phase as the reference phase exactly between the rigidly coupled subcarrier component phases can be placed like an adjustable attenuator. With the largely digital Signal processing ensures that the coder is operationally reliable with just a few adjustment points and is easy to use.

For the decoding of the multiplex signal in the receiver, Art the processing of this signal in the coder is irrelevant; the matrix and the time division multiplex principle can optionally be used for decoding.

With the decoder operating reciprocally to the matrix coder (Fig. 5), the sum signal M from the multiplex signal is obtained immediately after a low-pass filter - the deemphasis element (1), which is required anyway - is obtained: M = w + x + y + z (a ) By demodulating the directional signal, increased by 3 dB from the coder, in two switching demodulators (2), (3), two of four signal pairs are initially obtained, one of which is in phase with the corresponding portion in the main signal and one in opposite phase, with a component lagging by 45 ° compared to the regenerated subcarrier oscillation, the signal pair D ~ 450 = w # 2 - y 712 (b) with a component leading by 450, the signal pair D + 45o = x # 2 - z 72 (c) instead With two further demodulators with the subcarrier components -135 ° and +1350, the other two signal pairs can also be obtained by simple phase reversal in two reversal stages (4): D + 135o = y # 2 - w F2 (d) D ~ 1350 = z ff2 - x F2 (e) To reclaim the channel By multiplying equations (k) to (n) with the attenuator (5), the main signal is attenuated by 3 dB: M '= w / 2 # 2 + x / 2 # 2 + y / 2 # 2 + z / 2 # 2 (f) In four simple summing circuits (6) the four directional signal components are added to the reduced main signal, the summing losses (6 dB) as a factor of 0.5 when inserting equations b) to f) into the matrix equations g) to j) are taken into account: w '= M' + D-45 ° (g) x '= = M' + D + 450 (h) y '= M' + D + 135 ° (i) z '= M' + D-135 ° (j) w '= 3 / 4w # 2 + 1 / 4x # 2 - 1 / 4y # 2 + 1 / 4z # 2 (k) x' = 1 / 4w # 2 + 3 / 4x # 2 + 1 / 4y # 2 - 1 / 4z # 2 (l) y '= -1 / 4w # 2 + 1 / 4x # 2 + 3 / 4y # 2 + 1 / 4z # 2 (m) z' = 1 / 4w # 2 - 1 / 4x # 2 + 1 / 4y # 2 + 3 / 4z # 2 (n) factor the equations change into the desired form w '= w + x / 3 - y / 3 + z / 3 (o) x' = w / 3 + x + y / 3 - z / 3 (p) 3 3 3 y '= -w / 3 + x / 3 + y + z / 3 (q) wxyz' = - + + z (r) 3 3 3 In the practical implementation of the decoder, this multiplication is of course not necessary. In any case, no amplifier stages are required in the decoder to compensate for the summation losses. When designing the stages for regenerating the subcarrier and separating the components, it is advisable to provide a rigid phase coupling of the subcarrier components, as the crosstalk attenuation depends on this. This can be done either by a phase-locked loop operating at 76 kHz, from whose output voltage the subcarrier components are derived digitally, as already described for the time division multiplex coder, or simply by means of a supercritically coupled band filter at 38 kHz, which always has a Phase shift of 900 generated.

Since the development of elegant methods for a calibration-free subcarrier component generation however not the object of this invention is shown in the block diagram of the matrix decoder only the conventional carrier recovery method known from conventional stereo decoders used. The pilot tone is generated from the multiplex signal with a 19 kHz resonant circuit (7) sieved out, frequency doubled by two-way rectification (9), amplified (8), in two phase shifter stages (10), (11) each with an adjustable CR or RC element rotated by 450 or -450 in phase and used twice for demodulation (23). After the simple dematrixing (6), the 4 channels are already available for a quadrofone Playback available.

The peripheral functions of a decoder, such as the pilot tone display and the automatic mono-stereo switching correspond to the usual technology in stereo decoders and are therefore omitted from the block diagram.

A display of Quadrofonie broadcasts as well as an automatic switch from stereophonic to quadrofonic are in the compensation levels described below and can be arranged after the matrix decoder in the same way. They work depending on the modulation, as it is within the meaning of the invention there is no own continuous pilot signal for quadrofonie.

If the time division multiplex principle is used to decode the multiplex signal, so you can build the circuitry and balance by using integrated Stereo decoders, which work according to the time division multiplex principle, greatly simplify. Achieved for crosstalk attenuation, channel equality, distortion and intermodulation factor you get particularly good values with it.

One embodiment of a time division multiplexed decoder (Fig. 6) works with two integrated stereo decoders of the type MC 1310 (1), (2). This guy has for use in quadraphonic decoders for polar coordinate-modulated directional signals the particular advantage that between the output of a preamplifier and the inputs the phase discriminators, in which the pilot tone with internally generated subcarrier frequencies is compared in terms of phase, the pilot voltage via an externally connected Capacitor must be performed. At this point you can easily adjust the settings Passive RC phase shifters can be looped in (3), (4), which reduce the pilot tone by -22.50 or +22.50 shift. The source resistances of the preamplifier RQ and the input resistances RE of the phase detector stages are included in these phase shifters. All Modulation components in the multiplex signal are not affected by this phase shift affected as they are within the integrated circuit directly to the demodulator stages to get redirected. On the switching demodulator of an integrated stereo decoder the regenerated subcarrier thus appears by -45 °, on that of the other by + 45 ° shifted compared to its normal position.

The signal at output L of a correctly set stereo decoder is mathematically equal to half the sum of the sum and difference part of the multiplex signal: L. = 1/2 (M + D) (s) The signal at output R is equal to half the difference from the sum and demodulated difference component of the multiplex signal: R = 1 - D) (t) The sum component M of the multiplex signal is already in the LF position. The difference portion In contrast, D is demodulated with the regenerated subcarrier. For carrier frequencies Difference components with the phase position of the subcarrier component used in each case for demodulation the demodulation result is maximum. Carrier-frequency difference components with around +900 or -90 ° from the subcarrier component used deviating phase position on the other hand not demodulated at all. So for the -45 ° subcarrier phase shift working stereo decoder the difference signal component D-45 ° = # 2 (w-y) (b) for the with + 45 ° subcarrier phase shift the difference signal component D + 45o = 72 (x-z) (c) the demodulation result.

Substituting equations (a) and (b) or

(a) and (c) into the dematrixing equations (s) and (t) of the two stereo decoders one obtains the equations (w) and (y) or (x) and (z), which are the channel-wise Reproduce the composition of the output voltages of both stereo decoders. For 1/2 + 1/2 # 2 is written 1.2 for short, for 1/2 - 1/2 # 2 -0.2 is short.

On the decoder with -45 ° subcarrier shift: L-45 ° = 1.2w + 0.5x - 0.2y + 0.5z (w) R-45 ° = -0.2w + 0.5x + 1.2y + 0.5z (y) To the Decoder with +450 subcarrier shift: L + 450 = 0.5w + 1.2x + 0.5y - 0.2z (x) R + 450 = 0.5w - 0.2x + 0.5y + 1.2z (z) With these four output signals is already a satisfactory quadrophone reproduction is possible. However, the important thing is crosstalk attenuation between the middle of the pages is not yet optimal, as the following calculation shows. Given a left-side signal, that is, w = x = 0.7c; y = z = O. This gives itself: L-45 ° = w '= 1.2 c1 L + 45 ° = x' = 1.2 c1 R-45 ° = y '= 0.2 c1 R + 45o = Z1 = 0.2 c1 crosstalk attenuation L / R = 20 log 1.2 = 15.6 dB. To get out of the 0.2 through the equations (w) to (z) signals represented the optimal system equations (o) to (r) only needs a sixth of the output voltage of each decoder output added in phase opposition to the output voltage of the other output of the same decoder to become. After that there would still be a to achieve the system equations arithmetically It is necessary to multiply by the factor 1.233 = 0.811, which of course in the decoder is superfluous, as there is a lowering of approx.

2 dB would correspond. For this purpose, amplifier stages can be used to compensate for the summation losses do not apply because only a residual loss of 4 dB remains. In future Integrated decoders can use the phase inverters (5) and summing elements (6) anyway completely omitted; one in the common emitter line to each switching modulator The two current impressions belonging to the crosstalk compensation are already present Resistance only needs to be increased in its ohmic value, or for the purpose of being more universal Using the integrated decoder one could lead out this connection. Of the entire Quadrofonie decoder is required not a single oscillating circuit; it could be fully integrated without further ado. You would then come up with a single phase loop for subcarrier regeneration off; leave the voltage controlled oscillator oscillate to 152 kHz instead of 76 kHz, the derivation of the required subcarrier components can be carried out digitally without adjustment with the highest precision. The inexpensive quadraphonic decoder without adjustment elements is thus possible.

The reconversion of the polar coordinate modulated direction signal in two as linear coordinates for the definition of the direction of the sound sources Channel signal pairs (or differential signals) can thus be illustrated and Detect quantitatively that the corresponding to the individual direction signal components Vectors (e.g. from FIG. 2) by projection onto the directional axes of the subcarrier components are broken down into their components belonging to these directions (FIG. 7). Under the requirement that the regenerated subcarrier components in the decoder on exactly the same phase positions are set as before they were suppressed in the coder This means that the same signal components behind the demodulation outputs as before the modulator inputs were present.

The modulated direction signal R, which for a certain part of the program the amount R and the phase angle α in turn into the components assigned to the fixed phase angles -450 and +450 m (w-y) and m (x-z), whose magnitudes become m (w-y) = | R | cos (α + 45 °) m (x-z) = | R | sin (α + 45 °).

The demodulation results in these components again raised difference signals D ~ 450 = F2 (w-y) and D + 450 =? 2 (x-z).

When receiving the polar coordinate-modulated quadphone transmission with a stereo receiver only receives one component of the directional signal in its decoder exploited, namely that which lies in the direction of the 0 ° / -180 axis (Fig. 8th). If the phase position of the direction signal is 0 °, its amount is positive Maximum, from # 90 ° its zero crossing and from 180 ° its negative maximum; he obeys thus the cosine function of the direction signal phase angle mOo = | R | cos O through the demodulation is created from this by each of the two transmitters, each raised by 3 dB Difference signals D-45 ° = # 2 (w-y) and D + 450 = 72 (x-z) a component whose amplitude is related to that of the full difference signal as cos (+ 45 °) = 1/2 # 2: 1. Both demodulated components add up to the stereo difference signal D: D = 1/2 # 2. # 2 (w-y) + 1/2 # 2. # 2 (x-z) D = (w-y) + (x-z) With the help of the sum signal M = w + x + y + z the channel pairs L = 1 + D) = w + x and -2 (M R = 2 (M - D) = y + z formed separately.

The compensation stages (Fig. 9) have the task of dominating Signal components in one or more corners (effect recordings in radio plays, etc.) to prevent the other loudspeakers from emitting signal components that are attenuated by 9.6 dB. As a result, e.g. for any currently dominant corner signal, a large Crosstalk attenuation () 30 dB) to all other corners and with two dominant corners at the same time Signals from any corner a large one crosstalk attenuation (, 30 dB) between these corners and at the same time from 18 dB to the other two corners achieved.

The output signals w ', x', y 'and z' of the quadraphonic decoder are first amplified by 20 dB in four counter-coupled operational amplifiers (1). From their outputs, the channel signals go on the one hand to the lighting-controlled input voltage dividers (2) of the four compensation stages (3) and on the other hand to four peak value rectifiers (4) equipped with germanium diodes. The threshold voltages of these diodes are already exceeded by low-frequency signals that are 30 dB below the full level of a channel. The positively directed output voltages of the rectifiers, which contain the channel level information, control four comparators 710 C (5) at their inverting inputs, 50 so that their outputs normally assume a negative potential. A further positive directional voltage is used as a common reference voltage for all four comparators. has been obtained, and which represents information about the total amplitude of the crosstalk to be expected in each channel according to the transfer equations. If a channel is currently not modulated, it only contains the crosstalk, which in the case of incoherent remaining channel signals consists of only a third of the sum of the remaining channels. This means that the voltage falls below the reference voltage (directional voltage of 3 <times the channel sum) and the output of the associated comparator becomes positive.

Control the output voltages of the four comparators above four switching transistors (8) four light-emitting diodes (9) with series resistors (10) and smoothing capacitors (11).

These LEDs illuminate (only if the comparator output is positive) four photo resistors that are the variable parts of the input voltage divider (2) of the compensation levels (3). via the upper and fixed partial resistances of this Voltage dividers are used to send the preamplified channel signals to the compensation stages (3) laid. When photoresistors are illuminated, they become low-resistance and close the associated input signals briefly, which means that the compensation stage concerned is without any shift in the operating point is rendered ineffective. This will cause switching interference avoided.

The four compensation stages (3) are phase inversion stages with non-inverting ones (0 °) and inverting (180 °) outputs, from their output AC voltages Adjustment potentiometers (13) for precise compensation adjustment of partial voltages in One third of the amplified channel voltages are used by each compensation stage the other three channels in passive summing circuits (14) so that their Phase position of the phase position indicated by the signs in the transmission calibrations the crosstalk signals from the relevant channel is in the opposite direction. Additionally Another signal component reaches the summing circuit from each compensation stage of the own channel to read the volume loss from the compensation equations from the same. The outputs of the quadro / stereo switching stages are also connected to these summing circuits (15) connected. The total loss amounts to approx. 14 dB, so that the overall order has a through gain of 20 dB - 14 dB = 6 dB. Four impedance converters (16) can be arranged in front of the outputs, some of them can also be used for stereo / quadro switching.

If a front panel display (17) is currently in operation If compensation levels are desired, those at the collectors of the switching transistors (8) inverted information available from the comparators (5) four further switching transistors (17), in their emitter lines in series with series resistors four more LEDs (18) are located. These light-emitting diodes are now in phase opposition to the light-emitting diodes (9) controlled, so that the compensation lines are displayed on the front panel are currently open.

When receiving a two-channel stereo broadcast with an inventive equipped Quadrofonie receivers, both right loudspeakers receive the right one and both left speakers fed to the left channel. That can be undesirable since you are used to stereo playback, the base along a wall and not to be had in the middle of the room Stereo or mono reception 1. mute the rear speakers, 2. the modulation content of the rear channels supplied by the decoder have the same level as the assigned front channels mix in, which means that the phase is perpendicular to the stereo base in the multiplex signal delete contained noise components; 3. The three operating modes Mono / Stereo / Quadro with lamps or light emitting diodes.

While for the distinction between mono on the one hand and stereo / quadro on the other hand, the pilot tone display voltage supplied by the decoder in the usual way is used, the important decision between Stereo and Quadro must be made out of the modulated program can be derived. It is to be demanded that when occurring a level difference between one or both of the front and the associated rear channel output signals of the Quadrofonie decoder the switchover on four-channel playback takes place quickly, but that when both difference signals disappear switch back to two-channel playback only after a waiting period (approx. 30 s) can be done to bridge modulation pauses within a broadcast.

In connection with high-resistance MOS-FET gates, one can be used for this Realize a suitable pointer very easily.

In two operational amplifiers (19) the other non-inverting Inputs are low-frequency pre-amplified signals from the two front ones Channels with the associated signals at the inverting inputs rear canals compared. If there are differences between the front and back - due to quadphone modulation - These differential signals are amplified by the opeiationseamplifiers and then by peak value rectification (20) into positive level DC voltages reshaped. The response threshold is set by means of variable negative feedback of the operational amplifier adjusted (21). The charging capacitors ier two rectifiers form in connection with adjustable high-ohmic load resistances timing elements (22) with adjustable Fall time constant for switching back from Quadro to Stereo.

The input resistances of the following NOR circuit (23) are due to of the isolated field effect inputs are extremely large and influence the discharge times not. With quadrophone modulation, due to the front / rear difference components of the of the capacitors or both are constantly being recharged, the NOR gate supplies logic L level. To ensure that the device is in operation in the event of malfunctions during tuning or a mono broadcast cannot switch to Quadro made this switch dependent on the presence of the 19 kHz pilot tone, This is done in a second NOR gate (24) the output information of the first gate with the information provided by the decoder about the presence linked to the pilot tone.

A switching transistor is activated in the decoder when the pilot tone is present opened, whereby its collector potential due to the voltage drop across the load resistor (Pilot tone indicator lamp) drops from H to L. This L together with the in the case of quadrphone modulation, the L supplied by the first NOR gate at the output of the second NOR gate an H.

When receiving a weakly incident stereo modulated transmitter the noise, which also contains front / rear difference components, deceives a quadrofone Modulation before. Therefore, the switch should only be made if the signal-to-noise ratio is sufficiently large on Quadro. A hand-operated switch is used for this purpose (25) a logical H (field strength too low; incorrect switchover to Quadro) or a Logical L (field strength sufficient for sufficient signal-to-noise ratio; Quadro switching only for Quadro transmission) in a third NOR gate (26) with the inverted (27) Output information of the second NOR gate linked. Only with L at both entrances the output of the third NOR gate can go high.

This prevents switching to Quadro in switch position H, while in switch position L the automatic system is enabled. The information of the switch (25) can also be controlled by an intermediate frequency level-controlled automatic be delivered in the manner of an automatic muting.

The output information of the third NOR gate (L for mono and stereo, H for Quadro) controls two inverters (28) and one input of the fourth NOR gate (29), the other input of which with the information provided by the decoder about the Presence of a pilot tone controlled will. Only with stereo reception both inputs of the fourth NOR gate receive an L, causing the output to go high goes and a light emitting diode (31) turns on via a switching transistor (30), which therefore only lights up with stereo reception, but goes out again with Quadro reception.

The two inverters (28) deliver an H level for mono and stereo reception and can each be loaded with 0.8 mA. These H provide the base current for two Emitter follower (15), which the output alternating voltages intended for the rear channels the compensation adding networks (14) for the associated front channels, whereby phase-perpendicular to the stereo base noise components from the two Decoders wipe each other out.

Furthermore, three more of the outputs of the two inverters Inverter (32) controlled, at the outputs of which an H level appears on Quadro reception. Two of them open the output emitter followers (16) for the rear channels, the third one switches on a light-emitting diode (34) via a switching transistor (33), which enables Quadro reception indicates. The H level supplied by the decoder when the pilot tone is missing switches over a further switching transistor (35) a light-emitting diode (36) that shows the operating state Indicates mono. The four NOR gates are, for example, on an integrated circuit CD 4001 A, which contain six inverters on an integrated circuit CD 4049 A.

Claims (16)

Patent claims: (1) Compatible procedure for the transmission of quadrophone programs via frequency-modulated radio transmitter using one of the four recording channels composite main signal, a direction signal with suppressed subcarrier and a pilot signal for the receiver-side regeneration of the suppressed subcarrier, characterized in that the phase angle of each component of the directional signal during the positive half-waves of the low-frequency AC voltage causing this component relative to the constant phase angle of the subcarrier oscillation regenerated in the receiver with the acoustic angle of incidence measured relative to a fixed reference direction of the sound that causes this part of the directional signal matches, and that the ratio of the amplitude of this component of the directional signal to the amplitude the proportion of the main signal caused by the same sound source is a measure for the distance of the causing sound source from the center of the one to be transmitted Sound source local area is. 2. The method according to claim 1, characterized in that the implementation the angle of incidence of the sound emanating from a sound source relative to a with 0 ° assumed reference direction in the equally large phase angle of to this Sound correlated portion of the directional signal relative to the with 00 assumed phase position of the subcarrier oscillation is achieved by adding the four diagonal sound incidence directions in the connecting plane of all four microphones opposite to the reference direction assumed as Oo as oblique, perpendicular to each other coordinates intersecting in the middle of the room and components of the subcarrier oscillation be assigned which the same phase angle compared to that assumed as 0 ° Have the phase angle of the original subcarrier oscillation like the spatial one Coordinates. 3. The method according to claim 1, characterized in that the output AC voltage of the front left microphone the amplitude of one of the subcarrier oscillations by 450 leading component, the AC output voltage of the front right microphone the amplitude of a component leading the subcarrier by 1350, the AC output voltage of the rear left microphone is the amplitude of one of the subcarrier oscillation by 450 lagging component and the output AC voltage of the rear right Microphone shows the amplitude of a component lagging 1350 after the subcarrier oscillation modulated so that the subcarrier components are suppressed in the modulation results are. 4. The method according to claim 1, characterized in that the amplitude modulation can also be done in just two modulators by increasing the amplitude of one of the subcarrier oscillation by 450 leading components with the difference of the output alternating voltages of the front left and rear right microphones, as well as the amplitude of a the Subcarrier oscillation by 450 lagging components with the difference between the output alternating voltages of the rear left and front right microphones is modulated so that the Subcarrier components are suppressed in the modulation results. 5. The method according to claim 1, characterized in that the sum the modulation results together with the sum of the four modulating signals and a pilot tone with half the subcarrier frequency and reduced amplitude forms the complete multiplex signal. 6. The method according to claim 1, characterized in that the complete Multiplex signal excluding the pilot tone to be added, also by synchronous subcarrier Switching can be generated, namely by means of an electronic switch, which is actuated by a component leading by 450 of the subcarrier oscillation becomes the output alternating voltage during the positive half-waves of this component of the front left microphone, during the negative half-waves of this component the output AC voltage of the rear right microphone is switched through to the output is, and by simultaneously by a second electronic switch, which is actuated by a component lagging by 45 ° to the subcarrier oscillation, during the positive half-waves of this component the output alternating voltage of the rear left microphone, during the negative half-waves of this component the AC output voltage of the front right microphone switched through to the output will. 7. The method according to claim 1, characterized in that the complete Multiplex signal without pilot tone also with instead of two electronic switches four electronic circuit breakers can be generated, with microphone output alternating voltages and sub-carrier components as in claim 3 are assigned 8. The method according to claim 1, characterized in that sub-carrier components with other phase positions, e.g. 00, +900, 1800, -900, as indicated in claims 2, 3, 4, 6 and 7 can be used if the AC output voltages are not individual Microphones, or their differentiation, but sums of components of several output alternating voltages or differences in such sums can be used for modulation. 9. The method according to claim 1, characterized in that in the event compared to the modulation according to claim 4, the amplitudes of the difference signals by 3 dB the amplitude of the sum signal of all channels that make up the direction signal are increased otherwise, due to the geometric addition of its components, not the amplitude of the sum signal could reach. 10. The method according to claim 1, characterized in that in the event the modulation according to claim 3 or 4 also the sum of the modulation results 3 dB can be increased. 11. The method according to claim 1, characterized in that in the event the modulation according to claim 3, 6 or 7 also to the output alternating voltage of a each microphone one sixth of the output alternating voltage rotated by 1800 in phase each diagonally opposite microphone can be added to turn to achieve the correct amplitude composition of the multiplex signal. 12. The method according to claim 1, characterized in that in the receiver by simultaneous demodulation of the directional signal with two opposite the phase position of the auxiliary carrier regenerated with the help of the pilot tone leading or lagging by 45 ° Subcarrier components two independent differences in channel alternating voltages which each belong to diagonal pairs of sound directions of incidence are recovered, whose connecting lines intersect at right angles, and those in four summing circuits are combined with the also recovered sum signal so that the four original AC channel voltages are largely separate again. 13. The method according to claim 1, characterized in that in the receiver also two electronic changeover switches that operate through components of the regenerated subcarrier oscillation operated, which of these lead or lag by 450, the entire multiplex signal can alternately switch through to their outputs. By synchronizing with the subcarrier components is this changeover switch in each of the four output voltages mainly contain the alternating voltage of one of the four original channels. 14. The method according to claim 1, characterized in that in one Receiver according to claim 13 the channel separation by adding one sixth that inverted AC voltage of each switch output to the AC voltage the other output of the same switch can be optimized. 15. The method according to claim 1, characterized in that in the receiver also components of the regenerated subcarrier oscillation with different phase positions, e.g. 00, +900, 1800, -900, for demodulation or for switching the multiplex signal can be used, whereby sums of alternating voltages in each case neighboring Channels are obtained, which again largely through combination on the audio frequency side can be separated. 16. The method according to claim 1, characterized in that in the receiver to further increase the channel separation between the summing stages and the four Low-frequency amplifiers can still have compensation stages inserted as required by direct voltages derived from the alternating channel voltages opened or closed individually. The compensation level of each channel leads the remaining three channels components of the AC channel voltage with that phase position and an amplitude dimensioned in such a way that the remaining crosstalk to these channels is fully compensated. It is important that each compensation level always then is put out of operation if the assigned channel is currently not a concise one Sound information transmits because in this case only crosstalk would be passed on from other channels via the compensation lines. compensation only takes place when necessary, i.e. to prevent crosstalk to other channels.