DE2308788A1 - CURRENT METER - Google Patents

Description

The invention relates to a device for measuring electrical current and other variables which can be related to a current or expressed in current values. Although the current meter of the present invention is generally applicable, it is particularly suitable for measuring very small currents. The measurement involves the generation of an output variable that is directly proportional to an input current. The output variable is calibrated or is displayed directly in current units. The measuring instrument according to the invention may further comprise known or desired size are used for display, measurement or control of variables corresponding to changes in current and to generate currents.

According to a method for measuring electrical currents which can be used in a wide range of current values the measurement by means of an integrator. The current is fed to the integrator. After a period of time, the

309835/0542

The value of the integrated current is checked, and the current value can be determined from the result of this test. For example the integration current is allowed to reach a predetermined value. Then the mean current is a function of time, which is necessary to achieve this value. Alternatively, the mean current is a puncture of the time it takes to decrease of the integrated current to zero at a reference current is necessary.

With the measuring device according to the invention, in particular very small currents, for example in the order of magnitude of some pA can be displayed by integrating the current, the integrated current being fed to a differentiator. The desired indication is preferably generated from the output signal of the differentiator by means of devices for periodically reducing the output signal of the integrator to zero. The output signal can, however, also from the Integrator are generated, the output signal of the differentiator being fed back in order to equalize the input signal and maintain a substantially constant input signal at the input.

The present invention has for its object to provide an ammeter with the very low Currents can be displayed very precisely.

The current meter according to the invention for measuring or displaying the value of a small input current includes a Integrator, devices for supplying the current to the integrator, inclination measuring devices responsive to the output signal of the integrator for generating an output signal representing the value of the input signal, and means for periodic resetting of the output signal of the integrator.

The invention is explained in more detail below on the basis of the exemplary embodiments shown in the accompanying drawing. Ee show:

-3 -309835/0542

Fig. 1, 2, and 4 Fig. 5 Fig. 6th Fig. 7th Fig. 8th Fig. 9 Fig. 10 Fig. 11 Fig. 12th Fig. 13th

Sub-circuit diagrams of the current responsive

Facility;

the circuit diagram of a differentiation device;

the circuit diagram of an integrator-differentiator

Arrangement;

a waveform diagram;

the circuit diagram of a circuit arrangement;

the waveform in the circuit of FIG. 8;

a more detailed circuit diagram;

the block diagram of a circuit to complete the circuit of FIG. 10;

a waveform diagram;

a more detailed circuit diagram of a unit

a difference responsive system; 14 shows the block diagram of a current difference

responsive system; and
15 shows the circuit diagram of a further exemplary embodiment

of the differentiator.

Fig. 1 shows a so-called operational amplifier A1 with high gain starting at frequency 0, with a inverting and direct input - or +. The on. input current i to be measured occurring at a terminal 10 fed to the inverting input of the amplifier A1 via a resistor R. Between the output and the inverting A capacitor C1 is connected to the input of the amplifier, so that the amplifier has an integrating effect. The capacitor C1 can be discharged using a switch S1.

The output voltage V1 of the amplifier A1 is fed to a further and similar amplifier A2, which is equipped with a Input capacitor C2 and a feedback resistor R1, which is variable, works as a differentiator. The capacitor " C2 is shunted by means of a switch S2 which is coupled to the switch S1.

_ 4 ~ 309835/0542

-Μ-

The output voltage V1 of the amplifier A1 results from the equation _

V1 = j ^ / i.dt. (1) The voltage V2 is given by the equation

V2 = R1 C2 (2)

so _{d (1} f _idt)

V2 = -R1 C2 C1 7

German

= - (C2R1 / C1) (3)

In this way, a current can be measured, the magnitude of which can be determined by the voltage V2.

Integration cannot go on indefinitely. Therefore, switches S1 and S2 are operated periodically, see above that the work cycles start over and over again. In practice, for example, the switches with a switching frequency working from 10 Hz. When the switches are open, there is a continuous current. If by closing the Switch caused discontinuity can be neglected, the switches only need during a very short time to be closed every cycle. This is beneficial under certain conditions.

An alternative arrangement is shown in Fig. 2, in which the voltage V1 across a capacitor C2 a Switch S3 is supplied, which is coupled to switch S1. The switch S3 either leads the signal to the output or by mass. With this arrangement there is a continuous output signal, while the switch S1 is open and a signal zero when the switch is closed.

With a reset frequency of 10 Hz, the closing time of switches S1 and S3 can be 1 ms, i.e. it results in a signal: gap ratio of 99: 1 for the integration.

- 5 -309835/0542

However, other ratios can also be used. Fig. 3 shows a circuit for generating a continuous Current with a signal: l corner ratio of 1: 1.

In the circuit of FIG. 3, an input DC voltage is fed to a stage at terminal 11, which is a Includes amplifier A2 with feedback to the inverting input. The inverting input - is each to one Resistors R3 and R4 connected so that two output signals + ν and -v are generated. These output signals can be selected using a switch S5 and are derived from a resistor R5, an amplifier A3 and a capacitor C3 existing integrator, the output signal of which is a triangular Has course. This signal is sent via the capacitor C4 directly to a switch S6 and via the inverters R6, R7, A4 and a Capacitor Cf> fed to switch S6. By combination of the output signals, a continuous output current can be achieved.

The circuit of FIG. 1 can be modified in various ways. For example, it may be useful to have a adjustable intermediate damping stage (Fig. 4) to be provided. The output voltage V1 of the amplifier A1 is fed to a stage which has an input resistor R8, an amplifier A5 and a variable feedback resistor R9. In this arrangement, the gain of the intermediate stage is the same the ratio of R9 / R8. V2 results in

V2 = (i R1 C2 R9) / (C1 R8) (4)

To avoid malfunctions due to disturbances or interference, the one shown in FIG. 5 can be used in practice Better execution form of the differentiator can be used in which the feedback resistor R1 is replaced by a small Capacitor C6 is shunted. An input capacitor C2 is in series with a resistor R1O.

- 6 309835/0542

If the input signal is constant, the output voltage exists V1 of the circuit of FIG. 1 from a series of oblique or sawtooth-shaped pulses, the inclination of which is a Function of the input current. The change in voltage that occurs within a given time interval during the period of the rise or fall taking place can also be used as an indication of the input current.

If very small currents are to be measured, the input current may contain two components, their one is constant or subject to relatively long-term changes, while the other, more important component is subject to faster changes. Under these circumstances it is possible to use an arrangement to reduce or Suppression of the first component by means of a high resistance between the output and the input of this circuit to use, although it has certain disadvantages.

Fig. 6 shows a possible arrangement of the integrator differ Ent iator circuit. In this circuit, input 12 is connected to the inverting input of amplifier A6 connected, which has a capacitor C7 for negative feedback. The output of the amplifier is to an amplifier stage connected, consisting of resistors R11 and R12, a Capacitor C8 and an amplifier A7.

The output 13 of this amplification stage is fed back to the input 12 via a feedback path, which has an integrator R13, C9, A8, which, in the feedback path switched, works as a differentiator. In one position of a switch S6, the effective position, the feedback is completed by a capacitor C1O, while in the other position of the switch, the reset position, the integrator is switched off and the capacitor C10 is connected to ground. The circuit of FIG. 6 can also be designed in such a way that instead of the current generator of FIG a power generator of the type shown in Fig. 3 is used.

• - 7 -

309835/0542

Instead of the usual high resistances, the current generators of the type shown in Figs. 2 and 3 for supplying a control terminal leakage bias current is more conventional Field effect transistor electrometers can be used.

Other methods can also be used in conjunction with an input current integrator to determine the rate of change the output signal of the integrator and thus the value of the input current can be used. For example There are many methods by which the output signal can be queried at certain times, with the slope of the Curve is determined directly or indirectly.

Fig. 7 shows in a diagram an example of an output voltage curve as it can be obtained by the integrator. In the case of an input current that is constant over the integration time - the integration time can be a few ms or a few 10 ms, for example - the signal has, for example, between times tQ and t 1. a voltage level e, .. Between the times t * and tp there is a voltage level from e * to e. From tp through t-, and tr to t, - the voltage increases linearly over the values βρ, e ,, e ^ and er. Between the times te to t / - there is a voltage drop e, - to eg (which can be equal to e * ) and the cycle repeats itself with the values e ", e ₈ , e ~ at times t ₇ , to and tg, etc. The times t ^ to tp and te to tg are very short.

The slope of the integration curve can be determined by measuring and deriving the differential voltages:

e ₅ ~ e ₇

^e 4 ^e 8

^e 5 ~ ^e 8
e ₄ ~ e ₇

or less accurate:

309835/0542

^e 5 ~ ^e 6

^e 5 ~ ^e O e ₅ ~ e ₂

e _K ^ e,

e ₅ - _βι e ₄ ~ e ₆

e ₄ - e ₃ e ₄ ~ e ₂ β /. ^ e ^

A difficulty with practical embodiments of the described Circuit has its reason in the fact that the device for testing or other response to the integrator voltage or other significant voltages circuits with high or extremely high impedance has, through which undesired changes in the measurement voltage occur. When using a switching device, for example of an insulated gate field effect transistor, the transistor itself forms an input capacitance which, though extremely small, can be of influence.

This difficulty will be explained with the aid of the example shown in FIGS. Fig. 8 shows a circuit with an integrator A1, C1 and one connected to it Transistor Q1, which corresponds to switch S1. The gate of this A control signal is applied to transistor Q1 as shown in waveform W1. The switching pulse signal can in practice have an amplitude of -10V.

When the pulse occurs, switch Q1 closes and capacitor C1 is discharged. The output voltage falls on

- 9 309835/0542

a value near zero. The switch then opens and starts a new integration measuring cycle. Does C1 have a Capacity of 1pF and the closed switch has a capacity of 0.7 pF, for example, a pulse of -7 V am Integrator output generated (Fig. 9). The set period of the integration cycle is, for example, 100 ms. Thereon the cycle is repeated.

This effect can be overcome or avoided by applying a compensation pulse of opposite polarity to a specific point in the circuit, and although it has a corresponding capacity C11. The capacity of the Capacitor can be equal to the capacitance of the switch, for example 0.7 pF. This capacitor becomes a suitable one Control signal voltage of opposite polarity supplied.

The compensation pulse amplitude is preferably controlled automatically, so that the undesired voltage level is exactly eliminated.

Another way to avoid the difficulty is to avoid the stage during the Integration period to provide a suitable distant signal. This creates an output signal during the reset period generated with a very high voltage, the effect of which, however, by simultaneous switching of the resistor R2 of FIG can be reduced. The difficulty is also overcome by using a switching device, for example by means of a reed relay switch that does not transfer the charge, or by means of a photosensitive interrupter.

An embodiment of the measuring device according to the invention is shown in FIG. 10, the value of the individual Elements are noted next to the same. The amplifier in the upper part of the figure containing transistors Q1 to Q9 and having an inverting input A and a non-inverting input B, a terminal C generates Output signal. This output signal C feeds the main return

309835/0542

- 10 -

coupling capacitor C12, which is switched back to input A. is. The output signal is passed from output terminal C to devices for further amplification and differentiation. The output C is also connected to an amplifier, generally designated as A9, which the transistors Q10, Q11 and Q 12 contains and at terminal D a constant Generates electricity. Four switches S7, S8, S9 and S1O consisting of "insulated gate" field effect transistors are also provided. During the reset period, switch S7 connects input A to ground and switch S8 is closed, so that it connects terminal D to the direct input B of the amplifier. The appropriate control voltages are at CA, CB and CC supplied.

At the end of the reset period, switches S7 and S8v are opened, with the amplifier in the balanced state remains. Terminal D is then connected to another integrator via switch S9. Has the switch S7 supplied Control voltage disturbed the compensation of the amplifier capacitively, the output signal changes, so that at the Terminal D creates a current. This current flows via switch S9 to the integrator and changes the output voltage am Output F. During the reset period, switch S1O connected to the positive rail (+12 V) is closed. At the end of the reset period, when switch S7 opens, switch S10 is also opened. The change in voltage across the Capacitor C1A neutralizes the voltage change due to of the gate S7. If the neutralization is incomplete, there is a further improvement in the output of the constant current source A9 and through switch S9 into the integrator.

In view of the component values detailed in FIG. 10, a further description is unnecessary.

- 11 -

309835/0542

Fig. 11 shows the circuit of a differentiator. A sawtooth-shaped input signal is fed to terminal 14. It reaches the direct input of the amplifier A1O via a switch S11, which has an adjustable feedback with resistors R15, R16, R17 and R18. The same amplifier input can be connected to ground via a switch S12. The output of the amplifier is connected to a variable attenuation stage with an amplifier A11, V / iderständen R19, R20, R21 and R22 and switches S13 and S14 connected _t S15.

The differentiator contains an input capacitor C16, an amplifier A 12 with an input resistor R23 and selectable feedback resistors R24 to R28, and a selector switch ¹ S16. The input signal is also fed to an amplifier A13. A capacitor C17 can be switched by means of a switch S17 which serves ^{as an input capacitor during a sawtooth input signal and a feedback capacitor during the reset P 1 S.}

Another embodiment of the integrator is shown in FIG. The circuit includes inputs M and N, the signal to be integrated is fed to the M input. A step correction voltage can be applied via a capacitor C18 are fed to an input P.

The input signal is fed from input M to the direct input of amplifier A 13. The inverting input is connected to ground via a capacitor C21 and a resistor R30. The amplifier input is controlled by a Switch S 21 switched. The output signal switched by switches S22 and S23 is fed to an amplifier A14, which is fed back inverting via a capacitor C22 and a resistor R31. The capacitor C22 is by means of a switch S24 can be bridged. A capacitor C23 serves as a feedback capacitor.

- 12 309835/0542

The input M is also to a reduction controller connected. The voltage from a decrease or attenuation potentiometer R32 is fed to a sawtooth generator R23, C24, A15 supplied, which is switched by a switch S25 will. The sawtooth voltage is fed in via a capacitor C25. The measuring cycle of the measuring device is as follows:

Operating mode Duration S21 S22 S2 £ S24 S25

Reset 0.001 s ON ON OFF ON ON

Increase 0.099 s OFF OFF ON OFF OFF

Reset 0.001 s ON ON OFF ON ON

Increase 0.099 s OFF OFF ON OFF OFF

and so forth.

According to the input M to be measured Signal current, a sawtooth voltage is generated at output N, which is subsequently amplified and differentiated, for example by means of the circuit shown in FIG. 11 in the case of the Differentiation capacitor is periodically reset by means of a switching system in synchronism with the resetting of the system of FIG will.

During the reset, the output signal is at zero (a reference potential corresponding to zero) and the switch S21 is connected to ground. The output signal of the amplifier A13 is equal to zero or equal to a reference potential.

At the end of the reset, switch S21 consisting of an insulated gate field effect transistor turns on the gate potential is switched off, which rises from -15 V to 0 V. The capacitive coupling in the field effect transistor leads to a Transfer of the charge to the measuring circuit. A similar voltage waveform of opposite polarity appears in the capacitor C16 supplied. If the amplitude is set precisely, the next charge transfer into the measuring circuit will cause the internal

- 13 309835 / 05A2

follow the capacitance of the field effect transistor exactly balanced. If the setting is imperfect, the A residual charge is fed to capacitor C23, which leads to a stage. This results in signal curves as they are in the 12a or 12b are shown, and not the desired waveform of Fig. 12c.

During the rise period the diminution sawtooth generator conducts A15 the capacitor C25 a sawtooth Signal to, so that a decrease in the current flowing through the capacitor C25 into the measuring circuit takes place. This stream will added to the input signal stream. The entire stream flows through capacitor C23 regulated by amplifier A13 and A 14. The output signal is thus a sloping (sawtooth) signal, the slope of which is proportional to the input current and the reduction or additional flow. The reduction or compensation regulation can also be used for neutralization a constant detector current can be used, which results for example as a result of a superimposed direct current signal.

The rise period is followed by another period during which the various capacitors react to those previously described Tensions are brought back. The voltage changes on the capacitors lead to changes in the stored ones Charge of each capacitor. By changing the overall charge on capacitors C23, C18, C25 and C19 there is a movement of the charge through switch S21.

The switch S22 and the capacitor C21 are used to reset the amplifier A 13 to zero, while the Integrator capacitors are reset. To save the switch S22 and the capacitor C21, an ideal amplifier which does not require an automatic zero control to measure the input current and input voltage to bring to zero. However, suitable devices, such as vibrating capacitor electrometers, are more expensive.

309835/0542

-K-

If the difference between two input signals Ax and Ay is to be measured, two are similar or identical Units 16 and 18 according to FIG. 14 are used. The signal Ax is fed to the input M of the unit 16, which has a corresponding Emits sawtooth output signal. The second input N can be connected to ground.

The second signal Ay is the first input M. Unit 18 supplied, the second input N ^ to the output the unit 16 is connected. The desired output signal is tapped from the output of the unit 18.

For example, if the input signal Ax of the unit 16 is 64 pA, the output signal of the unit Change 16 by 5 V in 0.1 s. This signal is fed to the input N ^ of the unit 18, so that 64 pA through the Capacitor C19 flow into the input of amplifier A13. For example, if the input signal Ay of the unit 18 80 pA, there is a difference of 80 - 64 = 16 pA and an output signal is tapped at the output of the unit 18, which changes by 1.25 V in 0.1 s. ,

The output signal is amplified, inverted or forwarded directly, differentiated / reset, depending on the choice, amplified or recorded on a recording device. The two gain levels are controlled in such a way that that the gain can be chosen precisely. In a practical embodiment, binary steps are from 1 to 28, i.e., 1, 2, 4, 8, 16, 32, 64, 128.

In this embodiment, the capacitance of capacitor C19 was 1.28 pF, the full scale of the instrument was chosen in binary steps between 128 and 1 pA, i.e.

Full scale deflection (pA) 1 gain (G) 128

The capacitor can be switched for higher current values, for example to 163.84 pF (256 pA on X64 and 16384 pA on XI) or 20971.52 pF (32768 pA on X64 and

- 15 309835/0542

2 4th 8th 16 32 64 128 64 32 16 8th 4th 2 1

2 097 152 pA on XI).

Similarly, the capacitor C25 can be changed from 0.6 pF to 10 pF, 150 pF or 2400 pF for greater reduction or superimposed currents are switched.

To feed in the current, the amplification section can be used instead of A15 and the circuit part containing the associated components uses the circuit of Fig. 2 or Fig. 3, will. For the capacitor with the capacitance of 20 971.52 pF, it is advisable to use a capacitor of approximately 0.022 μF and to use a preset tuning potentiometer to dampen the voltage feedback to the capacitor, so that the effective capacitance is about 20,971.52 pF. All important capacitors can be used on this Way to be set.

The amplifier A13 has a high impedance output so that it operates as a constant current source. Ever With an input voltage of 1 mV, it emits a charge of 40 μΑ over a frequency range from 0 to 5 kHz, whereupon at 2 MHz a reduction of 20 dB per decade during the reset occurs to 0.1 μΑ per mV at 2 MHz and a gain of 1.

The direct input of amplifier A13 was on this one Embodiment 25 times more sensitive at all frequencies. Since the gain characteristics of amplifier A14 controlled by resistor R31 and capacitor C22 was during the rise and sawtooth periods the frequency at which a gain of 1 occurs is also 2 MHz.

The instruments described have a frequency response from 0 to 1 Hz with a switching frequency of 10 Hz. However, other frequencies can also be used. be turned. It was found that with a sinusoidal

- 16 309835/0542

f irm input signal of 1 Hz is the output signal of the Differentiator is sinusoidal and the integrator output is cosine.

In the manner described, very small currents can be generated and, through integration and differentiation, very small currents Currents are measured, if desired with automatic cyclical repetition. The integrated sawtooth tension can be measured with regard to the rate of change by other methods than described, for example through an analog / digital conversion for digital measurement, or through sampling and storage.

It is also possible to provide an automatic zero setting of the system input in the manner described, see above that even with a drift of the comparison amplifier input, the potential difference between the one carrying the signal input current Conductor and the ground connection can be kept at a low value and constant. The same goes for that Potential at the reset switches.

The facility to neutralize the impact of the Capacitor charges on the circuit is also beneficial.

Fig. 15 shows a differentiator circuit which instead of of the right part of Fig. 11 can be used. The resistance R4O gives this, similar to resistor R23 in FIG. 11 desired frequency behavior. The units or components within the boxes shown in dashed lines are identical to one another. With the arrangement of FIG. 15, with an input signal of up to 10 V per 100 ms, a complete charge of approximately 10 μΑ can be achieved. With a feedback resistance of 1 ΜΩ, the output signal at terminal F is 10 V.

In the circuit of FIG. 11, a current of 100 μA is supplied to the amplifier via the capacitor C16 with a feedback resistor of 100 kn with an output signal of 10 V

- 17 309835/0542

leads.

A possible source of inaccuracy in the circuits described can be that due to dielectric Absorption, for example in the field effect transistor switch with insulated gate, distortion will occur. To avoid this inaccuracy, the correction voltage and / or a compensating distortion change can be brought about in the supply voltage. Appropriate compensation distortion can be brought about that the switch is supplied with a square-wave control voltage, the square-wave voltage has a lowering for correction. A square wave with an overload can also be used for correction can be used as control voltage for the switch. as well combinations of these corrections are possible.

Claims

309835/0542

Claims (7)

PATENT APPLICATION 1.y current measuring device, especially for the display of small input currents, characterized by an integrator, by a device for supplying the current to the integrator, by a device responsive to the output signal of the integrator Inclination measuring device for generating an output signal which represents the value of the input current, and by means for periodically resetting the integrator output signal. 2. Measuring device according to claim 1, characterized by a differentiation device, the input of which is supplied with the output signal of the integrator, and by Means for providing an output signal from the differentiator. 3. Measuring device according to claim 1 or 2, characterized by switching devices connected to the input of the integrator, which send an unwanted input signal to the integrator generate, and by means for generating a correction signal, the effect of which is equal and opposite that of the undesired signal and which supplies the correction signal to the input during operation of the switching device. -'19 309835/0542 -Vi- 4. Measuring device according to claim 3, characterized in that that the switching device consists of a field effect transistor exists, the undesired signal arises due to the capacitance of the transistor, and that the device a capacitor to generate the correction signal and contains a further switching device, through which the integrator input from the change in charge of the Condenser deriving current is supplied. 5. Keßgerät according to claim 1, characterized by an integrator with a device for supplying the current to be measured, with the output of the integrator an output signal is generated by a feedback from an output of the integrator to the input of the integrator, wherein the feedback contains a second integrator which has a signal component at the input of the first integrator generated, which is supplied opposite to the input current to be measured, the one containing the two integrators Circuit path has such a gain that the input signal to the first amplifier substantially remains constant, and wherein the output signal of the device is tapped from the output of the first integrator. 6. Measuring device according to claim 1, characterized by a device for the constant supply of a bias current - 20 309835/0542 to the input of the integrator. 7. Measuring device according to claim 1, characterized by means of generating an inverted current of equal amplitude and opposite polarity from the input current, whereby the input and the inverted current are fed alternatively to the integrator, through a phase reversal connection to the output of the integrator, and by a selection device for selecting the signal at the input or output of the phase reversing stage. 309835/0542 L e r s e i t e