DE202021106321U1 - Three-phase connection converter and its control system for standalone hybrid AC-DC microgrid operation - Google Patents

Abstract

A three phase connection converter (102) comprising:
a DC sub-grid connection port (201, 202), an AC sub-grid connection port (212, 213, 214), a voltage source converter (205), a current detection circuit (209, 210, 211), a voltage detection circuit (203), a controller unit (207), a gate driver circuit (206) and a filter unit (204, 208);
a DC sub-grid connection port configured to be coupled to a DC sub-grid (101) that receives or transmits DC power;
an AC sub-grid connection port configured to be coupled to an AC sub-grid (103) that receives or transmits AC power;
a controller unit configured to continuously receive the measured electrical parameters from the voltage and current sensing circuits and calculate an operating condition based on determining the operating condition to send pulses to the voltage source converter through gate driver circuits;
a gate driver circuit configured to receive the pulses from the controller circuit and convert them into a form that drives the switches of the voltage source converter;
a filter circuit configured to smooth the electric voltage and electric current flowing through the connection converter.

Description

FIELD OF THE INVENTION

The present disclosure relates to a development of a three-phase interconnection converter (IC) and its control method for standalone hybrid AC-DC microgrid (HMG) operation. The control method of the disclosed invention is for smooth power transfer between the AC and DC subgrids of a standalone hybrid AC-DC microgrid using a three-phase connection converter as a power transfer channel with a reduced number of sensors.

BACKGROUND AND PRIOR ART

Microgrid (MG) is a small power system formed on the consumer side by networking distributed generators (DG), loads, storage devices, along with power conversion, protection and monitoring equipment. MG operates in a controlled and coordinated manner as an autonomous system in standalone mode or as a controllable integrated unit when connected to the mains power grid. Depending on the supply voltage, microgrid systems are divided into three groups: AC microgrid, DC microgrid, hybrid AC-DC microgrid. The advantage of the AC microgrid is that the DG units can be connected directly to the existing power grid with minimal modifications. The advantage of DC-based distribution grids or microgrids is that contemporary DC loads can be efficiently powered by photovoltaic (PV) sources, fuel cells and energy storage. The hybrid AC-DC microgrid (HMG) includes the attributes of AC and DC microgrids. HMG enables AC and DC based generals, energy storage systems and loads to be easily and efficiently connected in their respective sub-grids without the need for multiple power conversions.

Power exchange between the AC and DC subgrids occurs through an interconnect converter (IC), with the control techniques using the DC subnet bus voltage (Vdc) and the AC subnet bus frequency (f) as inputs to generate a reference power command. The necessary power is transmitted between the sub-grids by controlling the corresponding current shares using a vector control strategy. Vector control technique is famous for AC machine control, its aim is to independently control the torque and flux in the AC machines. In power control with a pulse-width modulated (PWM) rectifier, vector control takes place for the decoupled control of active and reactive power. Similarly, in HMGs, vector control of ICs is performed to achieve decoupled control of the active and reactive power flowing between the sub-grids. The entire control process includes sensing the three-phase AC bus voltages, the currents on the AC side of the IC, and the DC link voltage. The same can also be done without detecting the AC voltages. AC current sensing is essential for performing current regulation and overload protection. DC voltage sensing is required for controlling the IC and also for overvoltage monitoring. However, the measurement of the AC voltage is only required to arrive at the frequency value f, which can also be calculated with a voltage sensorless technique as disclosed in the present invention.

Operating the IC without sensing AC voltage reduces system size and associated costs, increases system reliability by acting as a backup controller in the event of a sensor failure. This extension of the IC control also improves the robustness of the system as it is not affected by problems such as offset in the sensor's output, noise, resolution limitation. It is therefore proposed according to the present invention that the AC bus voltage sensor has a vectorless control of the IC which is free from the above-mentioned problems of the prior art. The embodied IC and control method constructs the required AC phase voltages by filtering the IC's corresponding gate switching pulses using second-order bandpass filters.

credentials

• T.V. Tran and K. Kim, "Frequency Adaptive Grid Voltage Sensorless Control of LCL-Filtered Inverter Based on Extended Model Observer," IEEE Trans. Ind. Electron., vol. 67, no. 9, pp. 7560-7573, Sept. 2020.
• M. Mehreganfar et al., "Sensorless Predictive Control of AFE Rectifier With Robust Adaptive Inductance Estimation," IEEE Trans. Ind. Inf., vol. 15, no. 6, pp. 3420-3431, June 2019.
• H. Zhang et al., "Study on PWM Rectifier Without Grid Voltage Sensor Based on Virtual Flux Delay Compensation Algorithm," IEEE Trans. Power Electron., vol. 34, no. 1, pp. 849-862, Jan. 2019.
• Y Cho and K Lee Virtual-flux-based predictive direct power control of three-phase PWM recti fiers with fast dynamic response,” IEEE Trans. Power Electron., vol. 31, no. 4, pp. 3348-3359, Apr. 2016.
• X. Xiao et al., "Virtual flux direct power control for pwm rectifiers based on an adaptive sliding mode observer," IEEE Trans. Ind. Appl., vol. 54, no. 5, pp. 5196-5205, Sep./Oct. 2018
• Y Zhang, Z Wang, J Jiao, and J Liu, "Grid-voltage sensorless model predictive control of three-phase PWM rectifier under unbalanced and distorted grid voltages," IEEE Trans. Power Electron., vol. 35, no. 8, pp. 8663-8672, Aug. 2020.
• J Liang, H Wang and Z Yan, "Grid Voltage Sensorless Model-Based Predictive Power Control of PWM Rectifiers Based on Sliding Mode Virtual Flux Observer," IEEE Access, vol. 7, pp. 24007-24016, Feb. 2019.
• Guo L, Li Y, Jin N, Dou Z and Wu J, "Sliding mode observer-based AC voltage sensorless model predictive control for grid-connected inverters," IET Power Electron., vol. 13, no. 10, pp. 2077-2085, Aug. 2020.
• J Kukkola and M Hinkkanen, "State Observer for Grid-Voltage Sensorless Control of a Converter Under Unbalanced Conditions," IEEE Trans. Ind. Appl., vol. 54, no. 1, pp. 286-297, Jan.-Feb. 2018
• A Mallik, W Ding, C Shi and A Khaligh, "Input voltage sensorless duty compensation control for a three-phase boost PFC converter," IEEE Trans. Ind. Appl., vol. 53, no. 2, pp. 1527-1537, Mar./Apr. 2017

Groupings of alternative elements or embodiments of the invention disclosed herein are not to be construed as limitations. Each group member may be identified and claimed individually or in any combination with other members of the group or other elements found herein. One or more members of a group may be included in or removed from a group for reasons of convenience and/or patentability. Where such inclusion or deletion occurs, the description herein is deemed to include the group as modified and thus satisfy the written description of all Markush groups used in the appended claims.

As used in the description herein and in the following claims, the meaning of "a", "an" and "the" includes plural references unless the context clearly dictates otherwise. Also, as used in the specification herein, the meaning of "in" includes "in" and "on" unless the context clearly dictates otherwise.

The reference to ranges of values herein is for convenience only, for individual reference to each particular value that falls within the range. Unless otherwise noted herein, each individual value is included in the description as if it were individually listed here. All methods described herein can be performed in any suitable order, unless otherwise noted herein or otherwise clearly contradicted by context.

The use of any and all examples or exemplary language (e.g., "such as") provided with respect to particular embodiments herein is intended only for further explanation of the invention and is not intended to limit the scope of the invention otherwise claimed. No language in the specification should be construed to indicate unclaimed elements essential to the practice of the invention.

The above information disclosed in this section is only for enhancement of understanding of the background of the invention and therefore it may contain information that does not form the prior art that is already known in this country to a person skilled in the art.

SUMMARY OF THE INVENTION

In accordance with an embodiment of the present invention, the connection converter is a device connected between a DC sub-grid and an AC sub-grid to form a hybrid AC-DC microgrid. Wherein the disclosed invention discloses the power circuit of the link converter and its attendant sensor circuits, filter circuits and controllers.

The control method embodiment discloses a decoupled vector control method with the reduced number of sensors for optimal and reliable operation of a standalone hybrid AC-DC microgrid.

• • Dass die fundamentalen Komponenten der Schaltspannungen (VI, V2 and V3) mittels einer einfachen Berechnung berechnet werden.
• • AC-Netzspannungen werden nicht geschätzt, sondern die Grundschwingungskomponenten der Schaltspannungen V1, V2 und V3 des Wandlers werden extrahiert, für die der Wert der Filterinduktivität nicht benötigt wird.
• • Die Schätzung der Netzwechselspannung erfordert den Filterinduktorwert, und der IC-Betrieb wird beeinflusst, wenn sein Wert ungenau ist. Dabei ist das offenbarte Steuerverfahren von solchen Problemen nicht betroffen, da kein Induktorwert beteiligt ist.
• • Bei dem offenbarten Steuerverfahren wird ein Bandpassfilter verwendet, um den Wert von f zu erhalten, um V1, V2 und V3 zu erzeugen, aber keine Wechselstromnetzspannungen, so dass die Phasenverzögerung des Bandpassfilters keine Auswirkung hat.

The disclosed invention provides a phase-locked loop (PLL) and PI controller design of three-phase link converters that supports the control method for optimal operation. In addition, the disclosed control method offers the following advantages:

• • That the fundamental components of the switching voltages (VI, V2 and V3) are calculated using a simple calculation.
• • AC line voltages are not estimated, but the fundamental harmonic components of the converter switching voltages V1, V2 and V3 are extracted, for which the value of the filter inductance is not needed.
• • Estimating the AC line voltage requires the filter inductor value, and IC operation will be affected if its value is inaccurate. In doing so, the disclosed control method is not affected by such problems since no inductor value is involved.
• • In the disclosed control method, a bandpass filter is used to obtain the value of f to produce V1, V2 and V3, but no AC line voltages, so the phase lag of the bandpass filter has no effect.

OBJECT OF THE INVENTION

The invention is intended to provide an optimal control method for ICs and to reduce the cost and size of the HMG system by eliminating AC voltage sensors. In addition, for the HMG system with AC voltage sensors, the invention aims to increase the reliability of the system by providing support in a situation such as sensor failures.

character list

• In order to further explain the various aspects of some exemplary embodiments of the present invention, a more detailed description of the invention will be given on the basis of exemplary embodiments thereof which are illustrated in the attached figures. It should be understood that these figures represent only illustrated embodiments of the invention and are therefore not to be considered as limiting its scope. The invention will be described and illustrated with additional specificity and detail using the accompanying figures.
• In order to be able to better understand the advantages of the present invention, a detailed description of the invention is given below with reference to the attached figures, which, however, should not be regarded as limiting the scope of the invention to the attached drawings, in which:
  • 1 shows: Schematic of a standalone hybrid AC-DC microgrid.
  • 2 shows: power circuit of the connection converter and its auxiliary circuit with controller, voltage sensors, current sensors and filters.
  • 3 shows: control method of the connection converter for the optimal operation of a hybrid AC-DC microgrid.
  • 4 Figure 1 shows: overall flow chart of the operation of the connection converter with the disclosed control method.
  • 5 shows: Results of the implementation of the AC-DC partial network generations P _ac in kW, Q _ac in kVAR and P _dc in kW for both load-varying events.
  • 6 shows: results of implementation of power flowing through IC Pic in kW, Q _IC in kVAR for both load varying events.
  • 7 shows: Realization results of (a) DC sub-grid voltage V _dc , DC sub-grid current I _dc and current flowing in the DC side of the IC I _IC,dc for both load varying events. (b) AC sub-grid voltage V _ac , AC sub-grid current I _ac and IC AC side current Iic,ac for all load cases.

DETAIL DESCRIPTION

The present disclosure provides the detailed description of the embodiment of the three-phase connection converter (IC) and its control method for standalone hybrid AC-DC microgrid operation.

The present disclosure also provides the details of the embodiment of the IC (102) and the control of the IC (207) in a stand alone HMG and the design of its closed loop control parameters. 1 shows the schematic of an HMG that connects separate AC and DC microgrids (referred to as subgrids of HMG) via an IC.

Connection Converter (IC):

The IC is basically a voltage source converter (205) that has bi-directional power flow capability as in 2 shown. Consisting of filters (204, 208), voltage sensors (203), current sensors (209, 210, 211), digital controller (207) and a gate driver (206). The measured output current values from the sensors are given to the digital signal controller via the analog-to-digital converter input pins. The control logic is implemented via the digital signal controller and the output pulses are generated via EPWM/GPIO pins. The generated pulses are then fed to the converter's gate drive circuitry to generate the actual control pulses at voltage levels corresponding to the Turning on and off the power electronic devices (switches of the IC) are required.

Droop Control Technique:

$$V_{dc}^{*}=V_{dc}^{\text{'}}+v{P}_{meas}$$

In the droop control of the AC-DC sub-grids, the preset droop characteristics according to equations (1) and (2) are used to determine the frequency f* and the voltage amplitude V* of the AC sources in the AC sub-grid; and the voltage V _dc * of the DC sources in the DC sub-grid. Thus, in the steady state, all sources operate at the common frequency and voltage amplitudes, with each source delivering power proportional to its rated power and meeting the total load demand. $$f*=f\text{'}+m{P}_{meas};V*=V\text{'}+n{Q}_{meas}$$

$$V_{\mathrm{i.e}c}^{*}=V_{\mathrm{i.e}c}^{\text{'}}+v{P}_{meas}$$

Where Pmess and _Qmess are the measured active and reactive power values at each of the source's terminals, f', V', V _dc ' are open-circuit frequency and voltage amplitudes, and m, n and v are the droop coefficients.

In this way, communication between the power sources using droop control is done indirectly without dedicated communication links.

AC voltage sensorless vector control of the IC (207):

The reference currents are determined and the current control is carried out in the IC control for the power transmission between the sub-grids. To generate reference currents, the value of f is required and the AC sub-grid voltages are measured to obtain the value of f. In the disclosed scheme, f is obtained from the line-to-line converter AC side switching voltages and not from the AC sub-line voltages. Even the IC's 3-leg switching voltages are not measured but calculated using the gate drive signals and the measured DC link voltage V _dc . Furthermore, the switching voltages obtained are filtered using bandpass filters in order to achieve the input-side three-phase fundamental frequency voltages V ₁ , V ₂ and V ₃ required for the PLL. The strategy for generating the PLL input voltages V ₁ , V ₂ and V ₃ is explained below.

• • Das Produkt aus V_dc und den Gate-Schaltimpulsen der Phasen a (S₁) und b (S₃) des IC werden berechnet, um Schaltspannungen zu erhalten, die zwischen 0 und V_dc der beiden Phasen schwanken.
• • Die Schaltspannungen der Phasen a und b werden mit Bandpassfiltern zweiter Ordnung mit einer Mittenfrequenz von 50 Hz gefiltert, um Gleichstrom und andere Frequenzelemente zu entfernen. Dadurch erhält man ihre Grundfrequenzkomponenten V1 und V2.
• • Im ausgeglichenen dreiphasigen System beträgt die Summe aller Phasenspannungen 0, daher wird der Grundanteil der Schaltspannung V3 der Phase c aus V1 und V2 berechnet, wie in Gleichung (3) dargestellt. $${\mathrm{<figure-callout\; id="3"\; label="V"\; filenames="DE202021106321U1\_0023.png,DE202021106321U1\_0024.png"\; state="\{\{state\}\}">V</figure-callout>}}_3=-\left({\mathrm{<figure-callout\; id="1"\; label="V"\; filenames="DE202021106321U1\_0021.png,DE202021106321U1\_0023.png"\; state="\{\{state\}\}">V</figure-callout>}}_1+V_2\right)$$
  
• • Die erzeugten V₁, V₂ und V₃ werden für die weitere Steuerung von IC verwendet.
• • Die Übertragungsfunktion F(s) des verwendeten Bandpassfilters (BPF) ist in Gleichung (4) angegeben, sie wird einen Einheitsgewinn in ihrem Zentrum oder ihrer natürlichen Frequenz haben und hat bei anderen Frequenzen eine geringere Verstärkung. $$F\left(s\right)=\frac{2\xi {\omega }_{n}s}{s^2+2\xi {\omega }_{n}s+{\omega }_n^2}$$
  
• • Worin, ω_n = 2*π*50 ist die Eigenfrequenz und ξ ist das Dämpfungsverhältnis von BPF. ξ wird auf der Grundlage der berücksichtigten Bandbreite (BW) von BPF ausgewählt und liegt im Allgemeinen im Bereich von 0 bis 1 für unterdämpfte Systeme, da BW von BPF = 2ξ (ω_n so ξ = 0.2 gilt als Bandbreite von 20Hz.

The strategy to generate V1, V2 and V3:

• • The product of V _dc and the gate switching pulses of phases a (S ₁ ) and b (S ₃ ) of the IC are calculated to obtain switching voltages that vary between 0 and V _dc of the two phases.
• • The switching voltages of phases a and b are filtered with second-order bandpass filters centered at 50 Hz to remove DC and other frequency elements. This gives their fundamental frequency components V1 and V2.
• • In the balanced three-phase system, the sum of all phase voltages is 0, so the fundamental part of the switching voltage V3 of phase c is calculated from V1 and V2 as shown in Equation (3). $${\mathrm{<figure-callout\; id="3"\; label="V"\; filenames="DE202021106321U1\_0023.png,DE202021106321U1\_0024.png"\; state="\{\{state\}\}">V</figure-callout>}}_3=-\left({\mathrm{<figure-callout\; id="1"\; label="V"\; filenames="DE202021106321U1\_0021.png,DE202021106321U1\_0023.png"\; state="\{\{state\}\}">V</figure-callout>}}_1+V_2\right)$$
  
• • The generated V ₁ , V ₂ and V ₃ are used for further control of IC.
• • The transfer function F(s) of the Band Pass Filter (BPF) used is given in Equation (4), it will have unity gain at its center or natural frequency and will have less gain at other frequencies. $$f\left(s\right)=\frac{2\xi {\omega }_{n}s}{s^2+2\xi {\omega }_{n}s+{\omega }_n^2}$$
  
• • Where, ω _n = 2*π*50 is the natural frequency and ξ is the damping ratio of BPF. ξ is chosen based on the considered bandwidth (BW) of BPF and generally ranges from 0 to 1 for underdamped systems since BW of BPF = 2ξ (ω _n so ξ = 0.2 is considered bandwidth of 20Hz.

$$I_{IC,d}^{*}=\frac{Q*}{V_{IC,d}}$$

The vector control of the IC with the disclosed approach without AC voltage measurement is as in 3 shown. The generated V1, V2 and V3 (304) are given to PLL instead of sampled voltages and the reference currents for which P* and Q* generation are achieved. The V ₁ , V ₂ and V ₃ are transformed into a dq reference frame to obtain the d and q axis components V _IC,D , V _IC,q . V _IC,d is made equal to 0 by the PI controller of the PLL. The θ _I needed to perform the transformations and the fundamental frequency f are calculated from the output ω _I of the controller. With V( _IC,d ) = 0, P* and Q* are decoupled and the reference currents I( _Ic,q )* and I( _IC,d ) ^* are calculated as shown in equations (5), (6). $$I_{IC,q}^{*}=\frac{P*}{V_{IC,q}}$$

$$I_{IC,\mathrm{i.e}}^{*}=\frac{Q*}{V_{IC,\mathrm{i.e}}}$$

Phase-locked loop design method (307):

$$V_q=V_m\text{cos}\left({\theta }_0-\theta \right)$$

The PLL (307) estimates the rotation frequency ω of the voltage vector and locks it to the line frequency. Consider the phase estimated by PLL to be θ and actual phase of the AC vector of magnitude V _m to be θ ₀ . The instantaneous values of the voltage vector resolved in orthogonal dq phases can be determined as given in equations (7), (8). $$V_{\mathrm{i.e}}=V_m\text{sin}\left({\theta }_0-\theta \right)$$

$$V_q=V_m\text{cos}\left({\theta }_0-\theta \right)$$

The voltage vector _Vac is aligned with the q-axis when θ ₀ = 0 and V _d becomes 0. For controlling the phase angle to make V _d = 0, the non-linear expression of V _d is linearized about the point θ = θ ₀ as shown in equation (9). $$V_{\mathrm{i.e}}=V_m\left({\theta }_0-\theta \right)$$

The closed loop transfer function (TF) of the PLL control system is given in equation (10). $$\frac{\theta \left(s\right)}{{\theta }_0\left(s\right)}=\frac{\frac{V_m{K}_{PLL}}{{\tau }_{PLL}}\left(1+s{\tau }_{PLL}\right)}{s^2+\left(V_m{K}_{PLL}\right)s+\frac{V_m{K}_{PLL}}{{\tau }_{PLL}}}$$

The correlation of the above TF with the standard closed loop results from the second order transfer function (CLTF) with the natural frequency ω _n,pLL of the PLL controller system and the damping coefficient ξ _PLL as shown in equation (11). $${\omega }_{n,PLL}=\sqrt{\frac{V_m{K}_{PLL}}{{\tau }_{PLL}}}\text{and}{\xi }_{PLL}=\frac{V_m{K}_{\mathrm{<figure-callout\; id="2"\; label="P\; L\; L"\; filenames="DE202021106321U1\_0022.png,DE202021106321U1\_0023.png"\; state="\{\{state\}\}">P</figure-callout>}LL}}{2{\omega }_{n,PLL}}$$

The controller parameters K _PLL , τ _PLL are obtained from equation (11) as shown in equation (12). $$K_{PLL}=\frac{2{\xi }_{PLL}{\omega }_{n,PLL}}{V_m}\text{and}{\tau }_{PLL}=\frac{V_m{K}_{PLL}}{{\omega }_{n,\mathrm{<figure-callout\; id="2"\; label="P\; L\; L"\; filenames="DE202021106321U1\_0022.png,DE202021106321U1\_0023.png"\; state="\{\{state\}\}">P</figure-callout>}LL}{}^2}$$

The KPLL= 0.0544 and τPLL= 0.1414 are determined from equation (12) by choosing §PLL= 0.707 and ω _n,PLL = 10 rad/s. The bandwidth of the PLL is kept small in order to have a good filter effect, since the PLL only contains the basic component. The magnitude of the AC voltage V _m =260V.

Design of the proportional-integral (PI) control and its parameters (312):

In the disclosed control method, the difference between f _pu and V _pu is passed to the PI controller, which generates the P*. Furthermore, I _q ^* is obtained from P* and is specified for current control.

The open loop transfer function (OLTF) between V _pu and f _pu is as given in equation (13). $$\frac{V_{p\mathrm{and}}\left(s\right)}{f_{p\mathrm{and}}\left(s\right)}=\frac{K_P{K}_{a}K{K}_b\left(1+s{\tau }_P\right)}{s^{2}C{\tau }_P\left(1+s{T}_{\mathrm{i.e}}\right)V_q}$$

Where K _P , τ _P are the controller parameters: K _a = rated power of both sub-rasters =10kW and K _b = 1/V _dc,rated = 1/600 are the gains in the P* and V _dc,norm generation parameters, respectively. V _q =260, C = DC link capacitance = 3.2mF. K = V _m /V _(dc,rated) = 260/600, the gain between I _q and I _dc is obtained by equating the IC's input and output power. T _d = control delay = 10 ^-5 s (one sampling time).

The controller parameters are selected taking into account the stability of the system. The TF shown in equation (13) has two poles at the origin, hence its magnitude diagram decay with a slope of -40 dB/decade initially. There is a zero at 1/ _τP and a pole at 1/T _d , and τ _L is the controller parameter, the zero 1/ _τP must be properly located for the system to be stable. The two ways to set 1/τ _P are 1/τ _P > 1/T _d and 1/τ _P < 1/T _d . With the assumption 1/τ _P > 1/T _d the phase margin (PM) is negative and the system is unstable, but for the case 1/τ _P < 1/T _d the PM is positive and the system is stable. Hence the zero 1/τ _P will precede the 1/T _d for the controller design.

The cross-over frequency ω _c is given as the geometric mean of 1/τ _p and 1/T _d as shown in equation (14). $${\omega }_c=\frac{1}{\sqrt{{\tau }_P{T}_{\mathrm{i.e}}}}$$

The value of τ _P is determined using equation (15) where "a" is an integer greater than 1. $${\tau }_P=a^2{T}_{\mathrm{i.e}}$$

The gain of the OLTF at the crossover frequency ω _c is unity as shown in equation (16). $${\left|\frac{V_{\mathrm{i.e}c,nO\mathrm{right}m}\left(s\right)}{f_{nO\mathrm{right}m}\left(s\right)}\right|}_{\omega ={\omega }_c}=1=\frac{K_P{K}_a\mathrm{<figure-callout\; id="1"\; label="K\; K\; b"\; filenames="DE202021106321U1\_0021.png,DE202021106321U1\_0023.png"\; state="\{\{state\}\}">K</figure-callout>}{K}_b{}_{}\sqrt{1+{\left({\omega }_c{\tau }_P\right)}^2}}{{\omega }_c{}^{2}C{\tau }_{\mathrm{<figure-callout\; id="1"\; label="P\; V\; q"\; filenames="DE202021106321U1\_0021.png,DE202021106321U1\_0023.png"\; state="\{\{state\}\}">P</figure-callout>}}{V}_q{}_{}\sqrt{1+{\left({\omega }_c{T}_{\mathrm{i.e}}\right)}^2}}$$

Using Equations (14), (15) and (16), the value of KP can be _determined as shown in Equation (17). $$K_P=\frac{C{V}_q}{K_{a}K{K}_{b}a{T}_{\mathrm{i.e}}}$$

The phase margin (PM) can be calculated as shown in equation (18). $$PM={\text{tan}}^{-1}\left({\omega }_c{\tau }_P\right)-{\text{tan}}^{-1}\left({\omega }_c{T}_{\mathrm{i.e}}\right)={\text{tan}}^{-1}\left(a\right)-{\text{tan}}^{-1}\left(\frac{1}{a}\right)$$

For a good phase margin and less overshoot, a=9 and the controller parameters are determined as τ _P = 81 × 10 ^(-5) , _KP = 1280, and PM = 77.3°.

Overall flow of the disclosed control method:

4 shows the general control method of the IC. The DC-Link current voltage and the AC-side currents of the IC are recorded with voltage and current sensors. The product of the DC voltage with the switching pulses of any two phases of the converter is calculated to get the switching voltages.

The resulting switching voltages are second-order bandpass filtered to remove the DC and higher frequency components, and the fundamental voltages are obtained. The extracted fundamental voltages are given as input to the phase locked loop (PLL) to estimate the frequency.

Normalized values of frequency and DC voltage are then calculated and the error between the two is fed to the PI controller which generates the reference power value required to cancel the error. The reference values of the currents in the dq frame are calculated from the reference power.

The pulses for controlling the IC are generated by current control of the IC, which is performed using the error between calculated reference currents and the measured currents.

The control pulses obtained are used together with the measured DC voltage to calculate the switching voltages.

Case Studies:

The HMG system is implemented with an RTDS hardware setup and three load cases are analyzed. Initially, for the 10kW of load connected to the DC sub-grid and 6+j3 kVA load of the AC sub-grid, the 8kW of active power is generated by both sub-grids, so 2kW of active power transfer through the IC from the AC to DC side. In addition, at the first load change event, the load of the DC sub-grid is changed to 2kW, therefore, active power generated by the AC sub-grid, Pac and DC sub-grid, P _dc became 4kW each single power 2kW of the power to transmit through IC, P _IC from the DC side to the AC side. At the next load change event, the active load on the AC side is changed to 8kW causing both P _ac and P _dc to become 5kW, increasing P _IC to 3kW flows from DC to AC sub-grid. Since the 3kVAR reactive load is covered by the AC sub-grid Q _ac , the reactive power supplied by IC Q _IC is 0. 5 shows the P _ac , Q _ac , and P _dc 6 shows P _IC , Q _IC for both load varying events.

The DC voltage V _dc DC subgrid the current I _dc and the current flowing in the DC side of the IC I _IC,dc , dc for both load changing events are in 7(a) shown, the I _IC,dc reverses its direction for the change in power transfer direction. The AC voltage Vac, the AC of the subgrid I _ac and the current flowing on the AC side of the IC I _IC,ac are in for all three load cases 7(b) shown. The change in direction and magnitude of I _IC,ac can be seen for all load variations.

The figures and the preceding description provide exemplary embodiments. Those skilled in the art will recognize that one or more of the elements described may well be combined into a single functional element. Alternatively, certain elements can be broken down into multiple functional elements. Elements from one embodiment may be added to another embodiment.

For example, the order of the processes described herein may be changed and is not limited to the manner described herein. In addition, the actions of any block diagram need not be implemented in the order shown; nor are all actions necessarily required to be performed. Actions that are not dependent on other actions can also be parallel to the others actions are carried out. The scope of the embodiments is in no way limited by these specific examples.

Although implementations of the invention have been described in language specific to structural features and/or methods, it should be understood that the appended claims are not necessarily limited to the specific features or methods described. Rather, the specific features and methods are disclosed as examples of implementations of the invention.

Claims (4)

A three phase connection converter (102) comprising: a DC sub-grid connection port (201, 202), an AC sub-grid connection port (212, 213, 214), a voltage source converter (205), a current detection circuit (209, 210, 211), a voltage detection circuit (203), a controller unit (207), a gate driver circuit (206) and a filter unit (204, 208); a DC sub-grid connection port configured to be coupled to a DC sub-grid (101) that receives or transmits DC power; an AC sub-grid connection port configured to be coupled to an AC sub-grid (103) that receives or transmits AC power; a controller unit configured to continuously receive the measured electrical parameters from the voltage and current sensing circuits and calculate an operating condition based on determining the operating condition to send pulses to the voltage source converter through gate driver circuits; a gate driver circuit configured to receive the pulses from the controller circuit and convert them into a form that drives the switches of the voltage source converter; a filter circuit configured to smooth the electric voltage and electric current flowing through the connection converter. Three-phase connection converter according to claim 1 wherein the controller circuit performs AC sensorless vector control of the link converter, wherein the AC voltages (304) are constructed using DC voltage and pulses of the link converter (302), the control method generating the reference current values using power control and current control loops (305) and a phase locked loop (307) is used to control the operation of the link converter. Three-phase connection converter according to claim 2 wherein the controller circuitry includes a phase locked loop (307) whose parameter escalation method calculates the frequency of the AC subgrid bus voltage without actually sensed AC voltages. Three-phase connection converter according to claim 2 wherein the controller circuit consists of a PI controller (312) to perform closed-loop control of the link converter, its parameter design method for smooth and controlled power transfer between the sub-grids.

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