DE19781627C2 - Interference cancellation system for global, satellite-based location receivers - Google Patents

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention

The invention relates to a zeroing and cancellation system for Receiver for global position detection (GPS = Global Positioning System), suppress the inband interference signals and / or interference signals in the L1 and L2 Lock GPS frequency bands. In particular, the invention relates to the Reception of orthogonally polarized electric field vectors and on Method of receiving these components with high cross polarization Separation, and on methods of damping interference and / or Interference signals, whereby adaptive polarization mismatching of the target GPS Received antenna feed signal is used. The invention suppresses interference and / or interference by significantly reducing the Interference signal to useful signal ratio (J / S = jammer-to-signal) from the point of view of the GPS receiver.

2. Explanation of the state of the art

The global position detection system (GPS) is a satellite-based one Navigation system that transmits digitally encoded data, the two and three dimensional location determination used on the receiving antenna. Its purpose is the cost-effective supply of users with highly accurate position, Speed and universal time recording worldwide. Therefore is a reliable operation of the GPS in a noisy environment for both valuable for both military and civil purposes.

The key to precise navigation lies in the processing of a very weak GPS spread spectrum signal that is digitally encoded and encrypted Rough acquisition (C / A) and precision (P (Y)) data, typically -120 dBm to -136 dBm (isotropic). The GPS signal spectrum uses two L- Band frequencies, namely L1 at 1575.42 MHz and L2 at 1227.60 MHz, with  Bandwidths of either 2.05 MHz for the C / A code or 20.46 MHz for the P (Y) code, and uses right-hand circular polarization (RHCP = Right Hand Circular Polarization) for both L1 and L2 to reflect the user dependency of the Simplify receiving antenna orientation. The C / A and P (Y) codes are on L1, the P (Y) code is on L2. The theoretical Process gain for the C / A and P (Y) codes are 43 dB and 53 dB, respectively. The critical GPS receiver reception states are: C / A code acquisition; P-code Direct acquisition; P code track; and P-code carrier-assisted track.

The digital GPS data can be acquired and processed even if the Radio frequency carrier reception is prevented by interference; however one becomes Get higher accuracy when the signal carrier is available. This is generally possible because the GPS method has an inherent anti-interference (AJ) has; however, the low received signal level makes the GPS for low energy interference and / or disorder vulnerable. For a local Inband source, it is relatively easy to completely mask what the GPS signal prevents successful processing of the digital data. So the GPS System identified multiple vulnerabilities and vulnerabilities Interference. From a military and civilian perspective, it is important for the GPS systems to establish an adequate anti-interference ability and to ensure that this Property is available in all environments. This was from the fellow military recognized and led to the development of several spatial, zeroing and / or beam-shaping antennas and digital filter concepts.

GPS receivers showed different levels of vulnerability to each other Types of interference and jammer wave types that include: broadband Gaussian noise, undamped wave (CW), breakover frequency CW, pulsed CW, amplitude modulated (AM) CW, phase shift modulated (PSK) Pseudo noise, narrowband and broadband frequency modulated signals, etc. The vulnerability is highly dependent on the environment and the Receiver Mode. Broadband Gaussian noise is the most critical type of interference in the group above because of the difficulty in filtering broadband noise without the GPS becoming quieter and because of the extraordinarily high  Costs and the influence on the working behavior, which of spatial filter, d. H. Zero adjustment solutions are accompanied on a moving platform.

The use of phase zero balancing is widely known in the art known, although the use of phase-shifted polarization for GPS interference zeroing is not known.

There is a need for an interference cancellation system for GPS systems that can can deal with complex interference environments arising from different Interference and / or interference waveform types, L1 and / or L2 interference, multiple Interference sources and different interference polarizations composed are. There is still a need for a high interference interference cancellation system Extinction levels for one or the other or both of the GPS working frequencies and the ability to adapt to changes in the orientation of the receiver antennas and / or to adapt the interference source.

An interference cancellation system is disclosed in U.S. Patent 5,298,908 described, in which the data to be received from a circular polarizing transmitter antenna with the same polarization as that Data reception port of the receiving antenna are transmitted. The Receiving antenna also has a receiving port, its circular polarization is orthogonal to the data receiving antenna port, so that the latter Channel contains only interference signals. The interference signals in this channel are correlated with the inference components in the data channel. A control loop is used for adaptive adjustment of the phase and amplitude of the interference signal and the adaptively adapted interference signal is subtracted from the data signal which contains some of the interference signals.

US Pat. No. 4,283,795 describes an adaptive cross-polarization Extinguishing arrangement described in which a first desired polarized Signal and a second orthogonally polarized interference signal Cross-polarized components received simultaneously by an antenna become. The orthogonally polarized components of the received signal are separated and transmitted along separate paths and combined after the  Suitable phase and amplitude of the separated polarized interference signal Way to maximum cross-polarization components in the other path have been discontinued. A feedback path has a circuit for Obtaining a residual interference signal in the recombined Output signal and generates a signal that is representative of the envelope and then generates appropriate control signals depending on such representative envelope signals to improve amplitude adjustment and phase of the separated polarized interference signal.

In contrast, the invention has for its object an improved To create interference cancellation system for GPS.

To this end, the invention provides a system for suppressing interference and Interference signals for a receiver of a satellite-based global Position detection system (GPS) before, the features of which are specified in claim 1.  

It is achieved with the invention that the differences in the apparent polarization the right circularly polarized GPS signals and that of the interference sources exhausted and the inband interference and interference signals in the GPS-L1 and L2 frequency bands can be suppressed. Furthermore, the invention enables that the orthogonal elements of the interference signals and the GPS signals with a high degree of cross polarization separation processed and in the antenna system adaptively cross-polarized and the interference zeroed out. Finally brings the invention has the advantage of receiving the interference signals a port of an adaptive antenna feed circuit with a High-frequency polarimeter structure and sampling of the interference signal enable so that the combined interference signals and the GPS signals modulated and zeroed out the interference signal in the port to the GPS receiver can be. It is also advantageous that an orthogonally polarized Receiver antenna structure created in a compact form and with a small profile which is able to process the L1 and L2 GPS signals independently. Furthermore, the invention brings a zero adjustment system, the multiple Can eliminate sources of interference with a coherent relationship, which one Vector summation enables and multiple sources of interference with similar ones Extinguishes polarizations. Another advantage of the invention is that the Interference signals and the control of adaptive cross polarization Zero adjustment system without the need to process the GPS signal be recorded. Furthermore, the invention brings the advantage of the antenna and to divide adaptive cross-polarization zeroing circuits such that  the antenna subsystem can be located remotely and powered can and that the electrical interface between these functional elements from a minimal number of high frequency coaxial or fiber optic cables and lines can be composed. Advantageously, the multiple implementation configurations and system modularity exploited, dealing with individual requirements for interference processing only in L1, only in L2, in L1 and L2, in L1 with bypassed L2. The Finally, the invention enables an improvement in the operating range of the GPS Receiver through an installed loss / gain insertion in the GPS Receiver.

Advantageous embodiments of the invention are in the subclaims specified. In particular, the invention creates a highly qualified one Orthogonal polarization receiving antenna system, which the received L- Band environment into the sham orthogonally polarized signals that make up the GPS signal and represent the inband sources of interference. The orthogonal components the received environment are filtered, amplified and by the antenna system to the zeroing system in each GPS band using separate ones Transfer cables. In the case of the L2 bypass configuration, this can right circular polarized signal can be developed and transmitted on the antenna. A sample of the interference signal in each band of the GPS channel is acquired and processed to identify interference conditions and control signals generate the adaptive antenna feed circuits in each of interest Band are fed, showing the effective angle of inclination and the ellipticity (or axis ratio) of the combined antenna system. The effective one Polarization property of the antenna system is controlled so that the antenna is cross-polarized or mismatched for the interference source and thus the Interference signal in the channel containing the GPS signals zeroed or is suppressed. In configurations where L1 and L2 bands are separate processed, they are recombined after independent zeroing and the target GPS receiver. The capture and Control loops are optimized to identify the interference signals and  to acquire and the inclination and ellipticity properties of the adaptive Systems to quickly modulate to zero. Adaptation includes variation of Polarization properties, polarization orientation, shrinkage, Maneuvering changes etc. If there is no interference, any adaptive L1 / L2 loop can be configured so that the effective Polarization property of the antenna system to the preferred right circular Polarization for optimal reception of the GPS signal using the GPS Recipient.

BRIEF DESCRIPTION OF THE DRAWINGS

Figure 1 shows a top level block diagram with the adaptive cross-polarization interference cancellation system for GPS signals;

Fig. 2A-2F show various alternative implementation concepts of the invention based on repäsentativen applications for GPS receivers and GPS accuracy requirements. The figures illustrate three categories of implementation: a single orthogonal antenna configuration; dual orthogonal antenna configurations; and dual antenna orthogonal bifrequency antenna system configurations;

Figure 3 shows a preferred embodiment of a single channel dual orthogonal antenna configuration for zeroing only L1 (or L2) interference;

Fig. 4 shows a second preferred embodiment using a dual orthogonal antenna Bifrequenzkonfiguration nulling of interference L1 and L2 bypass;

Fig. 5 shows the structure of the orthogonal dualfrequency correction antenna with two rectangular microstrip shows correctors (not drawn to scale) in an orthogonal arrangement with independent L1 and L2 Orthogonaleinspeisungen;

Fig. 6 shows the structure of an ortho-dualfrequency correction antenna with two rectangular microstrip correctors (not drawn to scale) in an orthogonal arrangement with frequency-multiplexed L1 and L2 Orthogonaleinspeisungen;

Figure 7 illustrates the method for locating the optimal 50 Ω impedance feed ports for the ortho bifrequency right angle correction antennas for L1 and L2 orthogonal feeds;

Figure 8A shows the radio frequency block diagram and receiver processing scheme for a channel or loop of interference cancellation;

FIG. 8B illustrates an alternative interference receive detection circuit that can be used in the embodiment of FIG. 8A;

Fig. 9 explains the modulator scheme for the polarimeter or gamma / phi modulator part of the microwave section according to the invention;

Fig. 10 illustrates the down converter scheme for an interference heterodyne / detection circuit;

Fig. 11 illustrates the intermediate frequency amplifier and video detection scheme for an interference-heterodyne receiver / detection circuit;

Fig. 12 illustrates a schematic of a logarithmic amplifier for the interference receiver / detector circuit of Fig. 8B;

FIG. 13 shows the varactor-controlled phase shifters for the polarimeter modulator from FIG. 9; and

Fig. 14, 15 and 16 illustrate the GPS interference extinction control algorithm for detecting and for canceling interference.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Figure 1 shows a top level block diagram of the adaptive cross-polarization interference cancellation system for GPS signals. The diagram shows a channel or a band of the invention with the cancellation concept and explains the received signal, which is composed of the combined GPS signals 1 and the interference or interference signal 3 . The received signal 1 , 3 consisting of the GPS signals and the interference signals is received by the antenna system 5 , which effectively divides the received signals into orthogonal components VP and HP (vertically polarized and horizontally polarized) for the adaptive antenna feed system 7 . A power coupler 9 scans the differential port 11 of the network, which supplies the differential signal to the GPS receiver, not shown, via line 13 . An interference reception detection circuit 15 receives and processes the difference signal and supplies the signal to adaptive control circuits 17 , which control the phase of the modulators within the adaptive antenna feed system 7 for inclination and ellipticity. The loop closes in the event of interference for cross-polarizing the feed and for canceling out the interference for the GPS receiver. The GPS receiver may wish an activation signal 19 for the fault display via a GPS interface 21 deliver in order to activate the receiver circuit 15 or to deactivate. When activated, a fault detection signal 23 can be sent back to the GPS receiver.

To explain the operating behavior of the invention for zeroing a signal, it is assumed that all received signals, GPS signals 1 and interference signal 3 are composed of vertically and horizontally polarized waves. The theory that supports the ortho-polarization zero balance concept used in the invention is based on the property that any wave of any polarization can be synthesized or decomposed from or into two waves that are orthogonally polarized to each other. For example, a circularly polarized wave can be generated by the coexistence of a vertically and a horizontally polarized wave, each of which has the same amplitude and a 90 ° phase difference. A linearly polarized wave can be generated by the coexistence of a vertically and a horizontally polarized wave with a 0 ° phase difference. Thus, orthogonal polarization antennas can be used to match or mismatch a transmitted signal, using the relative phase and amplitude modulations to combine the orthogonal components. In the event of an interference signal, orthogonal polarization antennas can be used in the antenna system 5 to mismatch the receiving system and effectively nullify the signal. This mismatch 0 would affect other signals in the environment, including the desired GPS signals, by creating conditions in the antenna that could range from a perfect match to a complete mismatch.

The loss achievable by polarization mismatch, the zero or the Mismatch can be a value between infinity and zero. The Theoretical polarization mismatch loss für can be used for two widely separated elliptically polarized antennas in free space calculated with the following relation become.  

in which:
γ = the ellipticity ratio, the signed voltage ratio of the main axis of the polarization ellipse to its minor axis (1 ≦ | γ | ≦ ∞),
β = the polarization mismatch angle (0 ° ≦ β ≦ 90 °), and
T = send and R = receive mean.

The polarimeter implementation used according to the invention effectively modulates the orthogonal receiving components of both the interference signals and the GPS signals and generates a polarization Mismatch against the interference signal in the signal path to the GPS receiver.

Fig. 2A-2F show various alternative approaches to implementation of the invention, the accuracy requirements of GPS based on representative applications for GPS receivers and. Figs. 2A-2F illustrate three categories of antennas and interface implementations:

• a) simple ortho antenna configuration ( Fig. 2A) that can be applied to the L1 or L2 band where a single zeroing channel is implemented in one of the L1 or L2 bands;
• b) the dual ortho antenna configurations ( Figs. 2B and 2C), in which separate L1 and L2 band antennas process the orthogonal receive signals in each band for L1 for nullification / cancellation and L2 bypass ( Fig. 2B), and an implementation, with the separate L1 and L2 band antennas, the orthogonal receive signals in each band for individual zeroing of L1 and L2 ( FIG. 2C); and
• c) the dual antenna ortho bifrequency antenna system configurations ( Fig. 2D-2F) with implementations in which a pair of bifrequency antennas provide the orthogonal electrical components for the L1 and L2 bands that have higher cross polarization separation , and wherein the orthogonal receive signals in each band are used for zeroing / wiping L1 and bypassing L2 ( Fig. 2D), a second implementation in which the orthogonal L1 and L2 band signals in each band for the individual L1 and L2 zeroing are processed ( Fig. 2E), and a third implementation (minimum interface) in which a pair of bi-frequency antennas provide the combined sum of the orthogonal components of the L1 and L2 bands, thereby passing the signals in each correction antenna through the Antenna feed position are frequency multiplexed ( Fig. 2F).

Fig. 3 shows a preferred embodiment of a single-channel dual Ortho antenna configuration for zero adjustment only the L1-band (or L2-band) interference. The antenna system 5 delivers vertically and horizontally polarized signals VP, HP to the adaptive antenna feed system 7 . The antenna system 5 comprises an antenna or antennas 25 , which can either be a pair of L1 dipoles oriented for orthogonal operation, or an L1 microstrip correction antenna with vertical and horizontal ortho feed. Co-correction antenna configurations are generally known in the art. Alternatively, antenna 25 may be the new bifrequency antenna shown in FIGS . 5 and 6, which provides the L1 vertical and horizontal components for each channel. The antennas 25 receive or acquire the L1 (or L2) GPS signals along with any in-band interference signals. The detected vertical and horizontal components pass through band filters 27 , 29 and preamplifiers 31 , 33 to provide the VP and HP signals.

Fig. 3 shows the polarimeter architecture (sometimes referred to as a gamma / phi) modulator 35 of the Einspeissystems 7, whereby the unequal phase (or delay) of the received Ortho signals VP, HP initially to tilt (phi) the relative quadrature by variable phase control circuits 37 , 39 , which are put into action by the adaptive phase controller 41 of the adaptive control circuit 17 , are first set and then combined in the first hybrid node 43 . The unequal phase (or delay), or phase shift, of the received ortho signals VP, HP results from the relationship between the two received signal components and from unequal delays in the transmission lines and circuits between the antennas 25 and the polarimeter 35 . The output signals of the first hybrid circuit 43 are theoretically of the same amplitude. The outputs 43 a, 43 b of the first hybrid circuit 43 are set in relative phase by variable phase control circuits 45 , 47 , on which an adaptive phase controller 49 of the adaptive control circuit 17 acts, and then combined in the second hybrid circuit 51 by a minimum zero at the differential output port 53 which is effectively or desirably the zero of the interference signal. The second output 55 of the hybrid circuit 51 is a summing port 55 and at the same time generates a maximum output. Balanced varactor phase shifter assemblies, described below, are used in each branch of the gamma ( 45 , 47 ) and phi ( 37 , 39 ) modulation process to obtain a matched operation on frequency and power. The differential (or delta) and summing (or sigma) outputs 53 , 55 of the second hybrid link 51 are sensed, processed by the interference detection receiver circuit or receiver processing circuit 15 , and used to provide loop control signals for the tilt and ellipticity ( or gamma / phi) modulations to be generated adaptively by the adaptive control circuit 17 . The control loop compensates for device fluctuations, sham interference signal changes, and component imbalances for the system. The zero or differential output 53 of the second hybrid device 51 is also supplied to the GPS receiver as an input 13 with a suppressed interference signal via a power coupler 9 .

FIG. 5 shows the structure of a dual ortho bifrequency correction antenna with two right-angled microstrip correctors 61 , 63 in an orthogonal arrangement. The dimensions D1 and D2 of the two microstrip correctors 61 and 63 are the same for each corrector and are chosen such that they optimally receive the L1 and L2 bands, each with orthogonal linear polarization, ie D1 is approximately equal to λ _1d / 2 and D2 is approximately equal to λ _2d / 2, where λ _1d and λ _{2d are} the signal wavelengths for L1 and L2 in the antenna dielectric, and where D1 is one dimension for each corrector and D2 is the second orthogonal dimension for each corrector, as further is explained below. The correction antennas 61 , 63 can be located on a single printed circuit board. Correction antenna configurations are known in the art and typically include a conductive portion 65 , 67 which overlies an electrically thin dielectric support member 69 , 71 which in turn overlies a conductive ground plane 80 with leads or probes connected to the conductive portions are. According to the invention, the feed or probe locations for the correctors are selected so that an optimal linear signal coupling and cross polarization separation is obtained. In this embodiment, four ( 4 ) feed points 73 , 75 , 77 , 79 are used to independently provide the L2 / V, L1 / H, L1 / V, and L2 / H polarization (P) electrical signals ,

FIG. 6 shows an alternative structure for a dual ortho bifrequency correction antenna with the two right-angled microstrip correctors 81 , 83 in an orthogonal arrangement. The dimensions and orientation of the two microstrip correctors 81 , 83 are the same as discussed above, using a new feed arrangement for frequency division multiplexing. The feed or probe locations 85 , 87 are located along the diagonals 89 , 91 or diagonal areas of each corrector and selected so that there is an optimal combined signal coupling and cross polarization separation for L1 and L2 signals. The two feed points 85 , 87 serve for the simultaneous supply of frequency-multiplexed electrical L2 / V and L1 / H, and L1 / V and L2 / H polarized signals. This arrangement shows a simpler cable interface.

The antenna design for the preferred embodiment of the GPS interference cancellation system uses the half-wavelength microstrip right angle element (current microstrip antenna element technology includes half-wavelength, quarter-wave and full-wavelength elements). The lengths D1 and D2 of the antenna correctors 61 , 63 (as well as the correctors 81 , 83 ) are critical dimensions and slightly less than a half wavelength in the dielectric substrate material 69 , 71 :

D ≈ 0.49λ _d = 0.49λ ₀ / (ε _r ) ^1/2

where D (D1 and D2) = the length of the microstrip element, ε _r = the relative dielectric constant of the substrate and λ ₀ = the free space wavelength for each frequency of interest. Variations in the dielectric constant and the feed inductance make it difficult to predict the exact dimensions so that the exact microstrip length is determined empirically.

The radiation source for a right angle microstrip antenna is the electric field that is excited between the edges of the microstrip element and the ground plane (excitation of an almost infinitesimal slot with a uniform E field). The fields are excited 180 ° out of phase between opposite edges. The input impedance of the antenna can be tuned using either a coaxial feed or an edge feed with a quarter-wave converter. The approximate input edge impedance of a microstrip element is R _in ≈ 60λ ₀ / W, where W is the slot width. The input impedance in the design is matched to 50 Ω impedance using a coaxial feed. The 50 Ω point for the infeed is obtained by varying the distance between the infeed point and the edge of the element. The impedance of the element at the frequency and polarization to which it is designed is essentially zero at approximately the middle line of symmetry of the element. Thus, strategically determining the feed points so that they are in one dimension near the element's zero impedance point, while in the second dimension they are at the 50 Ω point, the result is a pair of orthogonal feeds. Every possible entry position for 50 Ω impedance is calculated according to material properties and roughly localized for the element. These values serve as starting points, however exact dimensions are set empirically. The manufacturing accuracy, consistency of material and mutual coupling result in small changes within a group of units.

According to FIG. 7, the dual polarization microstrip rectangular element has dimensions selected such that D ₁ is tuned to the half-wavelength of the resonance frequency L1 and D ₂ is tuned to the half-wavelength in the dielectric of the second resonance frequency L2. Each rectangular element in the embodiment can be powered using either one ( 1 ) feed ( Fig. 6) or two ( 2 ) independent 50 Ω impedance coaxial feeds ( Fig. 5) near the center of each element. In the case of two ( 2 ) feeds per element, the feed 77 will receive vertically radiated polarization for L1 and the feed 79 will receive the horizontally radiated polarization for L2 and vice versa for the other element. In the case of a single feed per element, the feed 87 will receive a multiplex signal which is composed of the sum of the vertically polarized L1 and horizontally polarized L2 and vice versa in the other element.

The two antenna feed arrangements which are preferred according to the invention are:

• a) one with a dual coaxial approach for linear polarization of the two orthogonal modes of the right-angled correctors that are resonant at two different frequencies ( Fig. 5), and
• b) a second arrangement with a single coaxial frequency-multiplexed approach for the two orthogonally linear-polarized modes of the right-angled correctors ( Fig. 6).

The first approach of FIG. 5 is a four-port solution that independently optimizes the input impedance for each frequency and each polarization. The second approach according to FIG. 6 is a two-port frequency-multiplexed solution that optimizes the input impedances for the two frequencies. Dual frequency multiplexed operation can be achieved by locating the feed for each corrector along the area on the diagonal of the rectangular element.

The exact dimensions of each element and the feed points defined empirically using an iterative process. The procedure consists of the structure of the elements using defined materials down to the dimensions of the design equations for a right angle Microstrip element. The resonance frequency and impedance are measured usually slight differences from the theoretical predictions are used because of the combined effects of: Dielektrizitätskonstantenvariation; Impedance variation for non-resonant Coupling elements; Feed probe inductance and mutual coupling. Settings on the microstrip element sizes and feed points are made executed to correct the resonance frequency or the feed impedance. Multiple iterations may be required. If they are optimized, the microstrip dimensions and feed probe locations from unit to unit be consistent based on material uniformity and manufacturing variance.

The feed points are localized so that the one-dimensional current distribution of the element is used at resonance frequency. The feed input impedance of the antenna varies proportionally with the corrector current and the proofreader. Resonance frequency and pattern of the microstrip element are essentially independent of the feed point. The dimensions of the rectangular Corrector are mechanically tuned so that they are with the L1 and L2 Frequencies are resonant. The corrector current distribution is almost sinusoidal in Direction of current and almost uniform except near the edges in a direction orthogonal to the current. In the practical implementation a single-voltage source is used to excite the corrector Probe used that is orthogonal to the dimension corresponding to the corrector the wavelength-emitting edges is moved until a point is found which is compliant with the complex apparent conductance of the current for 50 Ω. Two symmetrical 50 Ω solutions for each wavelength exist between the middle  and the edges. The coupling between the feeds is minimal due to the orthogonality of the modes.

The polarization of the multiplexed or diagonally localized feeds is more tricky to position. The exact polarization at resonance changes slightly with the digit and the 50 Ω impedance digit must be symmetrical be compared.

The measurements can use one or the other of the two test runs: one standardized measuring line or an automatic network analyzer. The Penetration reflection coefficient is measured over frequency. The standing one Wave ratio is relative to the size and minimum position Corrector recorded and entered in a Smith chart. Of this The reflection coefficient becomes the resonance frequency and the power factor of the corrector using graphic methods. The power factor (the Reciprocal of Q) is suitable for resonance circuit diagrams and analysis. The Smith chart shows the location of the complex apparent conductance of the Feed against frequency for the resonance circuit. To measure errors minimize by impedance transformation due to the transition from Coaxial to microstrip and the transmission line are caused the input impedance is generally for individual discrete frequencies in one Band around the wavelength of interest measured after calibration of the Smith diagram by providing a small circuit on the plane which the line is connected to the corrector.

Thus, the probe or Anlötstellen 77 (L1-polarization 1) and 79 (L2-Poalrisation 2) with respect to Figure 7 for the antenna 63 of Figure 5 are determined as follows:.. Is The probe body 77 along the 0 Ω site near the centerline for the D2 length and the probe site is moved orthogonal to the D2 direction until the 50 Ω impedance is located as shown. Similarly, probe location 79 is along the 0 Ω location near the centerline for D1 length and the probe location is moved orthogonal to the D1 direction until the 50 Ω impedance is located as shown. The probe or soldering points 73 , 75 of the antenna 61 is determined in the same way.

For the only coaxial frequency division multiplexed antenna of FIG. 6, for the corrector 83, the probe or solder point 87 is at the location where the 50 Ω impedance has been calculated for each of the L1 and L2 frequencies and essentially on the diagonal or in the diagonal area.

Reference is now made to FIG. 8A, which schematically shows the radio frequency block diagram and the processing receiver for a single channel implementation of GPS interference cancellation at frequency L1. The input shown is a pair of orthogonal RF signals from the antennas 25 or from the antennas according to FIGS. 5 and 6. As shown in FIG. 1, the cancellation is functionally composed of the following: The antenna system 5 ; the adaptive antenna feed system 7 ; a power coupler 9 ; the interference reception and detection circuit 15 ; and the adaptive control circuit 17 . Fig. 8 further explains the hardware part and the manufacturing method of the invention in the following physical units: a microwave section 101 ; a down converter section 103 ; a receiver / detection section 105 ; and a system control section 107 . As shown, the down converter 103 and the receiver acquisition channel 105 monitor a coupled delta or difference port 109 of the microwave section 101 . The microwave section 101 shown consists of bandpass filters 27 , 29 and preamplifiers 31 , 33 , which form interfaces with the ortho antennas, a solid-state polarimeter or gamma / phi modulator 35 , a delta port power divider / coupler 9 and a delta monitoring port HF -Amplifier 10 . The channel bandwidth and the noise pattern are set by the arrangement of band filters 27 , 29 and low-noise RF preamplifiers 31 , 33 . The filter and preamplifier are located generally at the antenna (25, Fig. 5, Fig. 6, for example) in order to compensate for interface separation losses and to enable the arrangement of a remote antenna. The polarimeter modulator or gamma / phi modulator 35 uses a 90 ° hybrid architecture, as will be described. The two groups of gamma and phi modulator controllers 111 , 113 serve to control the tilt and ellipticity of the polarimeter 35 , and the polarimeter supplies a delta 53 and a sigma 55 output port. The sigma output 55 of the polarimeter is terminated and is not used in the system. The Deltaport output 53 of the polarimeter 35 is sampled in an RF power divider / coupler 9 . The output port 13 of the power divider serves as the input to the GPS receiver and includes the interference-suppressed GPS reception signals of interest. The second output port 110 is the zero monitoring port and is amplified in an L1 band RF amplifier 10 , is provided for signal processing and detection and is used to adaptively generate the zero loop control signals for the inclination and ellipticity modulation. As shown, the interference reception detection circuit 15 consists of a down converter 103 and a receiver / detector 105 . The signal acquisition and processing section of the invention provides dynamic range control via automatic gain control and video-acquired zero signals for processing by the system control section. The adaptive system control section 107 is digital signal processing consisting of the signal A / D converters (ADC) 115 or encoders, via multiplexers 117 , a microcontroller 119 with signal processing and with loop control algorithm, and control signal D / A - Converters (DAC) 121 for the analog control of the modulator 35 .

The output of the microcontroller and control program / algorithm consists of iterative settings for the AGC amplifier and (four) gamma / phi phase shift control signals. These signals are D / A converted at 121 and applied to the respective devices as analog control signals to complete the loop. The control loop and signal processing algorithm compensate the system for dummy interference signal polarization orthogonality, interference signal properties, polarization changes and component imbalance as described. The zero output 53 of the second hybrid circuit is fed to the GPS receiver at 13 as an input with a suppressed interference signal.

Fig. 8B shows an alternative interference reception detection circuit that is much simpler and therefore preferred. It is essentially a logarithmic amplifier with a log video output, which is fed directly to the A / D converter 115 . The alternative circuit reduces the need for AGC.

Fig. 9 shows the details of the polarimeter modulator 35, which is used in the adaptive antenna feed system. 7 The modulator shown uses a 90 ° hybrid architecture. The polarimeter is composed of two 90 ° hybrid couplers 43 , 51 and a pair of gamma phase shift modulators 45 , 47 and a pair of phi phase shift modulators 37 , 39 . Each phase shifter can be set over a minimum range of 0 ° -180 °. The first pair of matched RF phase shifters 37 , 39 is located in the ortho lines from the antenna in front of the first 90 ° hybrid circuit 43 and sets the tilt angle, or phi, of the polarimeter 35 . For zero voltage or signal minima, these phase shifters are set to relative quadrature and compensate for phase and delay imbalances in each arm of the antenna's orthogonal paths, as well as imbalances and incompleteness in downstream hybrid circuits and modulators. The phase shifter output signals from 37 , 39 are combined in the first hybrid coupling 43 . The output signals of the first hybrid coupling 43 are theoretically the same in amplitude (ie the amplitude difference is at a minimum). The outputs 43 a, 43 b of the first hybrid circuit 43 are set to relative phase by the second pair of balanced phase shifter modulators 45 , 47 and combined in the second 90 ° hybrid coupling 51 in order to achieve a voltage zero or signal minima at an output port 53 , delta port called produce, which is effectively the zero voltage of the interference signal. The second pair of RF phase shifters 45 , 47 sets the ellipticity, or gamma, of the polarimeter 37 . The other output of the second hybrid circuit, called Sigmaport, simultaneously generates a voltage peak or maximum output. Balanced varactor phase shifter assemblies to be described are used in each arm of the gamma and phi modulation process to provide a balanced modulator operation over frequency and signal level. The delta port output 53 of the second hybrid connection 51 is divided in terms of power into an RF power divider / coupler 9 .

For the further description of the polarimeter or gamma / phi modulator, it is assumed that the orthogonal VP and HP components are represented by E ₁ cos ωt and E ₂ cos ωt + α. The phase shift α between E ₁ and E ₂ represents the net phase difference that is introduced into an orthogonally polarized system by the relation of the two received signal elements and by unequal delays in the transmission lines and networks between the antennas and the polarization unit. The phi phase shifters 37 , 39 adjust the components so that α is extinguished. The outputs of the first hybrid circuit 43 at 43 a and 43 b are essentially of the same amplitude with the opposite phase angles with respect to (E ₁ - 90 °) and E ₁ . The gamma phase shifters 45 , 47 adjust the signals at 43 a and 43 b so that they are 90 ° apart, and the second hybrid connection 51 also shifts and combines the inputs so that they are 180 ° out of phase, making zero is produced. In an actual system, the phase shifts phi and gamma are adjusted to compensate for imbalances in the system to produce a zero signal at the delta port. Setting the phi from its nominal value effectively compensates for the mismatch condition to generate a minimum at delta port 53 . The setting of the angle gamma effectively compensates for amplitude imbalances in order to generate a minimum zero at the delta port 53 .

Fig. 10 schematically shows the down-converter 103rd The diagram shows the local oscillator 131 , a configuration consisting of a single sideband mixer 133 and IF amplifier 135 , an IF band filter 137 , a second IF amplifier 139 and a low-pass filter 141 . The downconverter uses a local oscillator (LO) 131 of 1,586 MHz and converts L1 to an intermediate frequency of 10 MHz with a bandwidth of 2 MHz using a single sideband mixer configuration. The IF output runs through a bandpass filter 137 and is amplified in a second IF amplifier stage 139 and low-pass filtered at 141 in order to adjust the dynamic range of the receiver.

Referring to FIG. 11, the receiver / detector 105 has, shown schematically, a 2 MHz band filter 143 having 10 MHz, whose input is the downward converter signal, a linear IF amplifier 145 with automatic gain control (AGC), a gain / Driver amplifier 147 , a video acquisition stage 149 with separate broadband and narrowband video filters 151 , 153 . AGC is used in the linear scheme to get enough dynamic range to handle the amount of expected interference signals. Both the detected broadband video signals 151 a as well as the detected narrow-band video signals 153 a are converted and processed by the acquisition and tracking algorithm in a micro-control unit 119 A / D.

The alternative logarithmic or exponential IF amplifier approach shown in FIG. 12 (see FIG. 8B) is advantageous because it enables the required dynamic range without AGC or a narrow AGC range. This arrangement comprises three cascade stages of amplifiers 110 , 112 , 114 with three detectors 116 , 118 , 120 , which are summed by the summer 122 and fed directly to the A / D converter 115 . This logarithmic amplifier improves the performance of interference reception and detection by expanding the dynamic operating range for detecting interference and for zero detection.

Modifications of the invention can make an asymmetric gamma / phi Phase shifter organization for a simplified modulator arrangement and Include 180 ° / 90 ° hybrid polarimeter architecture.

Fig. 13 shows the partial schematic of an analog variable, varactorgesteuerten phase shifter of the RF polarimeter 35th The variable phase shifter structure is used for each of the four gamma / phi modulators 37 , 39 , 45 , 47 in the polarimeter 35 which are adjusted from 0-180 °. The schematic diagram illustrates a hybrid reflective implementation that uses two varactor-tuned phase shift diodes 120 , 122 that produce a variable transmission line phase shift with constant time delay. The phase shift is twice the electrical length through the varactor to ground. The control voltage inputs are the gamma and phi voltages from the digital / analog converter 121 .

Figs. 14 to 16 show flow diagrams which illustrate the steps 119 used in the microprocessor control unit for detecting and canceling interference and / or noise. As shown, the basic system steps include an initial built-in test loop, a phase scan to determine the presence of an interference signal, a coarse and fine loop to clear and close in the event of interference, and a maintenance procedure to detect the zero of interference on changes and adjust. Interference detection is based on exceeding a disturbance or interference threshold. The algorithm roughly sets the polarimeter to zero the interference signal, followed by a fine adjustment of the polarimeter to maximize the zero for the GPS receiver. The coarse sampling uses a phase resolution correspondence with the shape and size of the achievable zero phenomena and the spectral / temporal properties of the interference signal.

The microprocessor 119 monitors the difference or delta port 53 via the power divider 9 and the RF amplifier 10 . This delta-zero port signal via line 109 is processed by the receiver detection circuit 15 and converted into digital form by the analog / digital converter 115 . As shown in Figure 14, the first step, as usual, will consist of an initialization and a BIT (built-in test) step to verify that the DC voltages are applied to the various circuit cards and to determine that the analog / digital and the digital / analog converter are ready for operation and all other initiations that may be required have been triggered, as indicated functionally in block 301 . The system then determines whether there is an interference or interference signal (block 303 ). The presence of an interference or interference signal is determined by looking at the magnitude of the voltage of the Deltaport zeros signal (over line 109 as processed) to determine whether the voltage is above a predetermined voltage threshold for normal GPS signals. If the voltage is above the predetermined threshold value, the presence of an interference or interference signal is assumed. If interference and / or interference has been determined, the phi 1 and phi 2 phase shifts for phase shifters 39 , 37 are set to zero degrees (block 305 ). This is done by setting the control voltage for the phase shifters 39 , 37 (cf. FIG. 13) to a predetermined voltage which corresponds to zero degrees. Typically there is a linear relationship between the control voltage and the amount of phase shift that is determined empirically.

After phi 1 and phi 2 have been set to zero degrees for phase shifters 39 and 37 , the gamma 1 signal for phase shifters 47 is sequentially set to 0 °, 45 °, 90 °, 135 ° and 180 ° and is set for each setting the gamma 2 input for phase shifter 45 is scanned from 0 to 180 ° in 2.5 ° steps. The delta port zero signal is monitored at each sample point and the output voltage is stored at each sample point (block 309 ). After completing this scan, the gamma 1 and 2 control voltages are applied to phase shifters 47 , 45 to set the control voltages to levels that produced the minimum output at the delta port (block 311 ).

Then the phi 2 voltage signal for the phase shifter 37 is set repetitively to 0 °, 45 °, 90 °, 135 ° and 180 ° and with each setting the phi 1 control voltage for the phase shifter 39 is set from 0 ° to 180 ° in FIG. 2 , 5 ° increments scanned (block 313 ). Again, the Deltaport zero signal is monitored each time it is set and the output voltage is stored (block 315 ). Then the phi 1 and phi 2 control voltages are applied to phase shifters 39 , 37 at voltage levels that produced the minimum output at the delta zero port (block 317 ).

Referring to FIG. 15, the system proceeds by an alpha and rho-value of 22.5 and 0.35 is set (Block 319). A fine tuning routine is called (block 321 ) as shown in FIG. 16. The fine tuning routine scans gamma 1, ie supplies control voltages to the phase shifter 47 over a range of alpha degrees that is from below to above the current setting of gamma 1 in rho degree steps (block 401 ). During this scan, the delta zero port is monitored and the output voltages are stored at each scan point (block 403 ). The system then determines the control voltage settings for gamma 1 that produce the minimum output at the delta port (block 405 ). Gamma 1 is then set to this control voltage (block 407 ). The system then samples or phi 1 from alpha degrees below to alpha degrees above the current phi 1 setting and samples or incrementally sets the control voltage in rho degree increments (block 409 ). During this scan, the delta zero port is monitored and the output voltage is stored for each scan point (block 411 ). The system then determines the phi 1 control voltage setting that produced the minimum output at the delta zero port (block 413 ). Phi 1 is then set to this control voltage (block 415 ). The fine tuning routine is then completed and the flow returns to block 321 of FIG. 15.

At this point, the values for alpha and rho are reset to 11 and 0.044, respectively (block 323 ). Then the fine tuning routine of FIG. 16 is called again and the fine tuning routine is repeated for the new values of alpha and rho (block 325 ) in the same manner as described above.

At this point it is understandable that the phase shifters have been set to produce a minimum output signal at the delta port which is representative of the GPS signal, with the interference eliminated or suppressed. The system continues to monitor the zero port output to determine the signal level (block 327 ). The system determines whether the interference or interference signal has been canceled by determining whether the delta zero port output voltage is higher than the set minimum by a predetermined amount (decision block 329 ). If a voltage change has occurred, the system then determines whether a new noise or interference signal is present or has changed (decision block 331 ). If it has changed, return to block 305 of FIG. 14 to begin a new search for a minimum exit at the delta port.

If the noise or interference signal has not been canceled (decision block 329 ), but a voltage change has occurred, fine tuning continues by returning to decision block 323 . For example, let's assume that the delta zero port output voltage is at a minimum value of 100 mV, which is typical for GPS signals. Assume further that the system and GPS receiver are on a moving vehicle, such as a truck. As the truck moves, the phase shift can occur due to the truck movement, which will result in the delta zero port signal changing due to the phase shifts. However, the change is usually not larger than a predetermined value. For example, the output voltage can change from 100 mV to 1 V when the receiver moves. If only this "small" change occurs (decision block 329 ), the fine-tuning routine is called again, but at the "finer" alpha and rho values (block 323 ) to continuously change the phase shift signals and a minimum or zero voltage on that Maintain Deltaport. On the other hand, if the level of the delta port signal becomes significant or "large" (as determined in decision blocks 329 , 331 ), such as an increase in a 100 mV signal to, for example, 5 V, it is determined that a new noise or interference signal is present and the overall system starts a new beginning at decision block 305 .

The specific numerical values for the increments of the phase shifts and of alpha and rho can be changed from those described above become. The specified values are only of an exemplary nature.

The second preferred embodiment of the invention shown in FIG. 4 uses a dual ortho bifrequency antenna configuration to nullify L1 interference and L2 bypass. In FIG. 4, the same polarimeter and control architecture as shown in Fig. 3 is illustrated. The antenna 225 used for implementation consists of two right-angled microstrip correctors in an orthogonal arrangement so that they receive the L1 and L2 bands with orthogonal linear polarizations, respectively, as shown and described in FIG. 5 or in FIG. 6. In this configuration, the zero output of the second hybrid circuit from the power coupler 9 is combined with an RHCP bypass signal generated for the L2 band (or the output of a parallel zero circuit for the L2 band). The L2 vertical and horizontal components from antennas 61 , 63 are fed to a 90 ° hybrid coupling 227 , whereby the orthogonal signal components are combined, then passed through a bandpass filter 229 at L2 frequency and through a preamplifier 231 to perform the L2 bypass To generate signal via line 233 . The combined signal output, L1 and L2, is fed to the GPS receiver, via a diplexer 235 , as an input in which the interference signal is suppressed.

Of course, other configurations can be implemented in accordance with the present invention, including but not limited to the configurations shown in FIGS. 2A-2F. For example, a configuration similar to FIG. 4 could be provided for L2 band interference suppression with L1 bypass.

Can continue the correction antenna arrays of FIGS. 5 and 6 can be used with any of these configurations of Figs. 2A-2F. The correction antenna configuration from FIG. 5, for example, provides orthogonal vertical and horizontal components for L1 and orthogonal vertical and horizontal components for L2. Each of these components or only a group of L1 or L2 components can only be required, depending on the configuration selected.

Claims (14)

1. System for suppressing interference and interference signals for a receiver of a satellite-based global position detection system (GPS), which can receive GPS transmissions in at least one of the two L-band frequencies L1 and L2

• an antenna system for receiving at least one of the GPS L1 and L2 signals and in-band interference or interference signals, which divides the received signals into two orthogonally polarized antenna output signal components for each of the L1 and L2 frequencies;
• - A polarimeter circuit connected to the antenna system, in which a first variable phase shifter stage receives the two orthogonally polarized antenna output signal components and shifts the phase of the orthogonally polarized antenna output signal components in accordance with first phase shifter control signals from an adaptive phase control means, in which a first hybrid coupler also shifts the phase shifted components stage and phase-shifting components to obtain a pair of first hybrid coupler output signals, and in which a second variable phase shifter stage receives the pair of first hybrid coupler output signals and shifts the phase of the pair of first hybrid coupler output signals in accordance with second phase shifter control signals from the adaptive phase control means, and in which a second hybrid coupler the phase shifted components from the second variable en phase control stage and links the phase-shifted components to form a polarimeter output signal, the adaptive phase control means being connected to the first and second variable phase shifter stages for obtaining the first and second phase shifter control signals in order to repeat the phase shift in accordance with the respective polarimeter output signal until a minimum polarimeter output signal.

2. System according to claim 1, characterized in that the first variable Phase shifter stage has a first pair of variable phase shifters,  one of the two phase shifters of the first pair being one of the two orthogonally polarized antenna output signal components and the other Phase shifter of the first pair the other antenna output signal compo takes up, and that the second variable phase shifter stage a second Has pair of variable phase shifters, one of which is one of the two the first hybrid coupler output signals and the other the other of the first Hybrid coupler output signals. 3. System according to claim 1 or 2, characterized in that resulting from the second phase shift stage resulting phase-shifted components have a phase shift of 90 °. 4. System according to any one of the preceding claims, characterized in that the phase shifted components of the second phase shifter stage in the second hybrid couplers are linked so that they are a 180 ° Show phase difference, the combined resultant being the Output is polarimeter. 5. System according to any one of the preceding claims, characterized in that the first phase shifter control signals the first phase shifter stage control that any phase difference between the orthogonal polarized components is eliminated. 6. System according to any one of the preceding claims, characterized in that the first hybrid coupler the phase-shifted components of the first Phase shift stage combined such that the amplitude difference of the pair of first hybrid coupler output signals is at a minimum. 7. System according to any one of the preceding claims, characterized in that the adaptive phase control means a microprocessor controller which (a) each of the first and second phase shifter stages one Range of phase shift control signals supplied, which incremental  Set phase shifts of the phase shifter stages, (b) the Polarime output signal for each of the incremental phase shift settings lungs and stores, (c) the phase shift settings that lead to a minimum value of the pointer output signal and (d) the Phase shifter sets to those settings where the Minimum value was found. 8. System according to any one of claims 2 to 7, characterized in that a first 90 ° hybrid coupler connected to the first pair of phase shifters and that a second 90 ° hybrid coupler to the second pair of Phase shifter is connected, with the polarimeter output signal on a differential port (A) of the second hybrid coupler is present. 9. System according to any one of the preceding claims, characterized in that the polarimeter output signal to the adaptive phase control means via a Coupling device is supplied, and that the phase of the orthogonal polarized antenna output signal components is shifted as long as until the polarimeter output signal takes a minimum value. 10. System according to any one of the preceding claims, characterized in that the antenna system has a pair of correction antennas, each Correction antenna has a conductive part of rectangular shape with a Length greater than the other length (D2) and to the other Correction antenna adjacent, not overlapping and orthogonal is oriented that the one length (D1) of a correction electrode in the substantially perpendicular to one length (D1) of the other correction electrode extends, and wherein each correction antenna has a pair of output lines each of the orthogonally polarized antenna output signal Components of one of the L1 and L2 frequencies on a first output line from one of the correction antennas and a first output line of the other correction antenna is provided.   11. System according to any one of the preceding claims, characterized in that each of the orthogonally polarized antenna output signal components of the other of the two L1 and L2 frequencies on a second output line one of the correction antennas and on a second output line of the other of the two correction antennas is provided. 12. System according to any one of the preceding claims, characterized in that the conductive part of each correction antenna on a dielectric substrate is arranged above a conductive ground plane, the one length (D1) approximately half the signal wavelength for the L1 frequency in the material of the dielectric substrate and the other length (D2) approximately half a signal wavelength for the L2 frequency in the material of the dielectric substrate corresponds. 13. System according to any one of the preceding claims, characterized in that each correction antenna has a first output line and a second Output line, wherein the first output line one of the Correction antennas the L1 frequency with a polarization P2 and the second Output line of this correction antenna has the L2 frequency with a Orthogonal polarization P1 picks up. 14. System according to any one of the preceding claims, characterized in that the location of the first output line of one of the correction antennas in Intersection of the zero ohm impedance point at the frequency L2 with the 50 Ohm impedance point of frequency L1 and the position of the second Output line of a correction antenna at the intersection of zero ohms Impedance point at frequency L1 and the 50 ohm impedance point at L2.