DE19538575C2 - Inductive proximity sensor - Google Patents

Description

The invention relates to an inductive proximity sensor with a device for Measuring the complex impedance of a lossy coil, the impedance of is dependent on a non-electrical measured variable.

A coil dependent on a non-electrical measured variable - hereinafter referred to as Sen Sorspule or measuring coil called - is used in operational measurement, process monitoring and sensors in a variety of ways as an inductive distance sensor or inductive Proximity switch for distance measurement and as an eddy current sensor for destruction free material testing. The non-electrical parameters can be from Ab was a measurement object - hereinafter referred to as the control flag - or the physika mical properties of the test object, such as electrical conductivity σ, the Per meability µ or derived quantities.

If such a, mostly metallic, object to be measured enters the alternating electromagnetic field the coil of an inductive sensor, the complex impedance of the coil changed by the non-electrical measurands. The non-electrical measurands often affect only one component of the complex at a suitable frequency Impedance, for example, affects the electrical conductivity of the test object Loss resistance of the coil or the permeability of the object to be measured Kitchen sink. However, the coil is always lossy without a measurement object, i. H. an inductive In addition to the basic inductance, the sensor has additional ohmic losses.

The electrical measuring circuits should if possible only the proportion of the impedance or detect a quantity derived from the impedance by the non-electrical Measured variable is dependent. A measuring circuit for evaluating the change in com plex impedance or a derived quantity is inherent in its behavior depending on the coil. The quality Q or the loss factor D and the tempera the behavior of the coil determine the sufficiently high sensitivity Properties of the measurement. In other words, the measurement properties become whole largely determined by the losses of the coil. By suitable choice of measuring circuit is attempted, only the part of the complex impedance dependent on the measured variable or measure a quantity derived from it, for example the effective resistance or the inductance or the quality or the loss factor. There are numerous measuring known circuits to solve this problem, but have few circuits has proven itself in practice. Mostly, the evaluation circuits are Improvement of the measuring properties directly adapted to the sensor coil.

The behavior of electrical and electronic circuits is unique depending on the components used. The quality and the temperature behavior Resistors and capacitors are controlled with modern technologies bar; the properties of modern semiconductor circuits are also characterized by ge suitable choice of circuit can be determined; however, only a few circuits are known that improve the quality and temperature response of a lossy coil.  

The most important non-contact sensors for process control and plant transfer are inductive distance or proximity sensors or proximity switches and Proximity initiators. An inductive proximity sensor contains a coil with a directional high-frequency electromagnetic field. Mostly han is used for this Standard cylindrical single or half-shell cores made of ferromagnetic Ferrite material. This creates a rotationally symmetrical pattern on the exposed legs metric single-shell cores a preferred direction of the electromagnetic high frequency quenzfeld. An electric or magnetic becomes in this directional high-frequency field conductive or ferromagnetic material - a so-called control flag - brought, occurs a damping of the magnetic and electrical components of the Hochfre quenzfeldes and thus a damping of the coil.

A commercially available inductive proximity sensor contains a high-frequency oscillator with an LC resonant circuit. The coil of this LC resonant circuit generates the directed one high frequency electromagnetic field. Due to the damping described above electromagnetic radio frequency field takes the amplitude of the radio frequency Vibrations of the oscillator. The amplitude of the oscillator radio frequency oscillation can act as a measure of the attenuation of the directed electromagnetic high frequency field are used, that is, as a measure of the distance of a control flag. The amplitude of the oscillator high-frequency signal is used to evaluate the damping generally rectified and filtered with a low pass filter. This same directional and filtered radio frequency amplitude signal is a DC voltage or DC signal and can either directly into an analog signal to display the Distance of a control flag or with the help of a subsequent evaluation circuit can be converted into a switching signal, the switching signal at a defined distance of the control flag changes its switching state. Particularly inductive Proximity switches that work according to the function described last serve in numerous embodiments and in large numbers for system control and Plant monitoring. As a circuit for generating the high-frequency vibrations a Meissner oscillator circuit is very often used.

In the case of an inductive proximity sensor, the achievable and usable distance range, ie the distance of the control lug from the "active" surface of the coil, essentially depends on the size and properties of the high-frequency coil. The damping by the control lug changes the quality Q of the coil: As the control flag approaches, the quality Q is reduced from a maximum value Q ₀ in the form of an S curve to a minimum value. The switching point or the measuring range is expediently set in the steepest part of the S curve, generally If you want to enlarge the measuring range, you also have to use the flatter parts of the S-curve. The measurement in a flatter part is associated with a much greater uncertainty, since a certain change in quality that can be evaluated here involves a major change in the distance The influence of the ambient temperature on the relative quality Q / Q _{0 of} a coil shows that the quality increases with The temperature influence on the coil quality considerably limits the usable distance measuring range with inductive proximity sensors, since the change in quality due to the temperature influence in a specified temperature working range can be greater than the change caused by a control flag.

In many applications and in many embodiments, distance measurement is used rich selected so low that no special measures to a Temperaturkom pensation are necessary. In critical applications, temperature is used dependent resistors, for example thermistors or PTC thermistors, around the temperature to compensate for the switching distance. However, this additional effort leads only satisfactory results in a limited temperature range. A it cannot significantly increase the usable distance measuring range can be achieved.

A lossy coil is described by the complex impedance Z. A measuring coil AC voltage U applied to the coil hurries through the coil to flow the measuring coil AC current I by the phase angle ϕ _UI , which is smaller than 90 ° (degrees) by the loss angle δ. The behavior of a lossy sensor coil of an inductive proximity sensor is described by an equivalent circuit in which the field loss resistance R _F by an ohmic parallel resistor in parallel with a reactance X _L , which is formed by a pure inductance L, and in series with this parallel connection, the winding resistance R _CU of the coil are taken into account by an ohmic series resistor, as shown, for example, in German Offenlegungsschrift DE 38 14 131 A1 in FIG. 2a. Represented in this equivalent circuit known per se

• - the field loss resistance R _{F is} the active losses of the alternating electromagnetic field due to eddy currents and magnetic reversal losses in the coil core and in the test object (control flag),
• - The reactance X _L the inductive losses of the measuring coil and the electromagnetic alternating field due to the permeability µ of the test object (control flag) and the dielectric losses due to the capacitance of the measuring coil and
• - The winding resistance R _CU the direct current losses due to the copper resistance of the coil wire and the alternating current losses due to the skin effect in the coil wire.

The quality of a coil is given by the ratio of the reactance Im ( Z ) and the loss resistance Re ( Z ), where Z is the complex impedance (impedance) of the coil.

The influence of the ambient temperature on the impedance Z and on the quality Q of a coil is essentially caused by the temperature dependence of the loss resistances. The temperature coefficient α of the direct current resistance of the copper winding of the coil with approximately 3.95.10 ^-3 / K is known. The temperature responses of the other loss resistors are usually smaller, but their size depends on the type of coil.

The quality and thus the temperature response of a coil is a composite  Size that is strong in different designs and even from specimen to specimen Is subject to fluctuations. Therefore a compensation or a reduction of the temperature response at least separately for each type of coil will.

As a measure of the distance of a measurement object (control flag) from the sensor coil, the measured variables complex impedance Z or, alternatively, quality Q or impedance Z of the measurement coil are measured. The quality is often measured with an LC resonant circuit or the impedance is measured in the non-destructive material test. The disadvantage of these measurements is the dependency on the temperature and the material of the test object and thus the reduced accuracy of the measured variable. In quality measurement with an LC resonant circuit, this resonant circuit is excited to resonate vibrations in an oscillator, the oscillation amplitude of the oscillator being a measure of the quality. The impedance Z can be determined from the simultaneous effective value measurement of the voltage U _eff applied to the measuring coil and the current I _eff flowing through the measuring coil.

The impedance Z contains the active component R (loss resistance) and the frequency-dependent reactive component X _L = ωL. The impedance Z results from the complex impedance Z = R + jωL.

If the effective resistance R is negligible, the reactive resistance X _L results from the current and voltage measurement.

In order to obtain the inductance L from the measurements, the frequency f = ω / 2π of the measuring voltage must also be known. To measure the complex impedance Z that is actually to be detected, two separate measuring processes must be used either

• - The effective resistance R as the real part Re ( Z ) and the reactance X _L as the imaginary part Im ( Z ) of the complex impedance or
• - The impedance Z and the phase angle ϕ _UI between the measuring coil alternating current I flowing through the measuring coil and the measuring coil alternating voltage U applied to the measuring coil are measured. In most applications, only one of the measured variables shown above is measured, and only the one that reacts as selectively as possible to the distance to be measured between the control flag and the measuring coil and which is influenced as little as possible by disturbance variables such as temperature and electromagnetic radiation, so that the Accuracy and resolution is as high as possible.

In the German patent application DE 35 13 403 A1 a method is specified after which the temperature coefficient of the winding copper resistance of the oscillation Circular coil for compensation of the temperature coefficient of the quality of the resonant circuit is used, one proportional to the copper resistance of the resonance circuit coil Voltage is applied to the resonant circuit with a second coil. This tempera However, door compensation can only be implemented under very special conditions. Of the The copper resistance of the second coil must be significantly larger than the copper resistance stood the voice circuit coil, while the inductance of both coils can be the same size got to. In practice, these conditions are very difficult to meet.

In the German patent application DE 38 14 131 A1 a method and a Device specified in which the effective resistance of the sensor coil as a field loss resistance is measured directly with a power measurement. This field loss resists stood the sensor coil, caused by the losses of the electromagnetic Field due to the stray field and due to electromagnetic damping a control flag, depends on the distance when choosing a suitable frequency s the tax flag. When using a four-wire circuit, the measurement is in temperature-independent over a wide temperature range, since the temperature-dependent dependent winding resistance of the coil (also called copper resistance) and the Line resistances of the four supply lines are not included in the measurement. By in-phase multiplication of the coil alternating current and the induced coil AC voltage and subsequent low-pass filtering is the active power in the Coil measured, which is inversely proportional to the field loss resistance.

In a similar way, the phase can be multiplied by 90 ° phase-shifted coil alternating current and the induced coil alternating voltage voltage and subsequent low-pass filtering measure the reactive power in the coil also inversely proportional to the coil reactance or the coil inductance act. For easy separation of the primary and secondary side, i.e. H. of the coil current and the induced coil voltage, the induced coil voltage is measured using a second winding decoupled. In the simplest case, the two windings are wrapped identically and bifilar.

In the German patent application DE 43 28 097 A1 a device for Measuring the impedance of passive sensors (inductive, capacitive and ohmic Sensors) with PLL (phase locked loop) described, wherein in a feedback line a measuring phase shifter as Low-pass or high-pass filter is arranged with the sensor component. The frequencies zanalog signal evaluation of such a phase locked loop has a very high Sensitivity to measurement. Furthermore, the frequency-analog output signal can be integrated further process and convert it into a digital signal.  

In all previously known inductive proximity switches, the switching speed is d. H. the cut-off frequency for switching operations, through low-pass filtering for generation the distance-dependent output signal (usually a DC voltage) and especially by the energy stored in the resonant circuit and in the electromagnetic field determined and thus significantly lowered. The cutoff frequency of the measurable switching device gears is caused by electrical reloading and by inertia of the stationary Ren, alternating electromagnetic field of the sensor coil determined. The frequency behavior Overall, the behavior shows the switching frequency of an inductive proximity switch ten of a low pass filter, d. H. the lowest frequency limit of the measuring chain determines in essentially its entire frequency response. The low pass filter mentioned above serves in addition to the generation of the distance-dependent output signal i. a. also the interferer pressure of internal and external interference signals. The cut-off frequency of the switch is usually gears at around 1 kHz, only with considerable electronic means is a limit to achieve a frequency of 10 kHz, but only with a lower interference suppression or a poorer signal-to-noise ratio.

Furthermore, the temperature response of the impedance of the sensor coil, in particular the winding resistance R _CU , limits the measuring sensitivity here, so that only those measuring methods should be improved that take into account or compensate for the temperature response of the sensor coil.

Furthermore, the analog output signal or the switching point of the material Control flag dependent, whereby the measuring accuracy of the inductive proximity sensor is deteriorating. This dependency is determined either by a material actuator or through complex compensation measures with auxiliary coils on the secondary side a transformer or by a differential method with a mathematically-stale Technical signal processing to compensate for the material actuator eliminated. in the German utility model DE 94 12 765 U1, this problem is addressed by means of a Differential method by measuring the voltage and induced in the sensor coil the resonance frequency of an LC resonant circuit formed with the sensor coil and solved by a mathematical-technical connection of these two signals. Also here is the switching speed of the inductive proximity sensor because of the low pass filtering of the two signals and by the energy stored in the resonant circuit significantly reduced.

It can be seen that correction methods are known for some influencing and disturbance variables are. However, no method and no device is known to date which are based on the Procedure itself the influences and disturbances of the measurement described above reduce or avoid entirely.

The invention is therefore based on the object of the detectable cutoff frequency an inductive proximity sensor or eddy current sensor or the maximum measurable Switching speed of an inductive proximity switch or proximity initiator to increase significantly without deteriorating its measuring properties, so that  the measurement accuracy and susceptibility to interference are maintained or improved.

The invention solves the problem according to claim 1. Thereafter, on a sensor coil of an inductive proximity sensor, the phase shift between applied measuring coil alternating voltage U or induced measuring coil alternating voltage U _ind and flowing measuring coil alternating current I immediately during each half period or a multiple of each half period of Current and voltage measured by electronic means. The solution is based on the knowledge that the complex measuring coil impedance Z or a derived measured variable of the sensor coil, here preferably the phase shift ϕ _UI between current and voltage, can be measured as often and as immediately as possible after a possible change if the measuring coil Impedance Z is determined without a delay by additional filters. Each low-pass and band-pass filter with its cutoff or resonance frequency f ₀ delays the signal between its input and output by the delay time T ₀ = 1 / f ₀ .

Developments of the invention are specified in the subclaims.

The use of digital technology is becoming increasingly prevalent in measurement technology their high signal processing speed up to cut-off frequencies of a few GHz and because of their greater immunity to interference and easier signal processing compared to analog technology. The processing takes place in digital measurement technology Processing of the signals in a value-discrete and time-discrete manner. The necessary quantization of the For the above reasons, measurement signals should occur at the beginning of the electrode, if possible that the entire measuring device with simple and inexpensive digital switching elements and assemblies can be built. In the simplest case, the value of an analog signal to be measured (current or voltage) as a binary signal be given, d. i.e. whether the signal is larger or smaller than a comparison signal. This task can be accomplished for a voltage with a voltage comparator. For example, the value zero can serve as the reference voltage, so that at a pure symmetrical AC voltage with no DC component of zero crossing use as a switching condition for the binary output signal of the voltage comparator can be detected. The switching condition can be finite in the same way Comparative DC voltage are not equal to zero. This is preferred for a AC voltage with a DC voltage component applied. The same procedure as described above can in the same way with an analog AC signal be performed.

In an inductive proximity sensor according to the invention, the measurement of the complex measuring coil impedance Z (hereinafter also referred to as measuring coil alternating current resistance) or alternatively the measurement of the phase shift ϕ _UI at a zero crossing of the measuring coil alternating voltage U and the measuring coil alternating current I or when exceeding or falling below these alternating signals compared to a comparison DC signal. Voltage signals are preferably used so that only the voltage comparators described above can be used. For this purpose, the measuring coil alternating current I with a current-voltage converter, preferably a pure reference impedance Z _ref (hereinafter also referred to as reference alternating current resistance), for example an ohmic resistor, a low-loss capacitor or a low-loss coil, or a mixed comm combination of these components converted into a proportional reference AC voltage U _I. When using a capacitor or a coil in the reference impedance Z _ref an additional phase shift is generated I and the reference alternating voltage U _i between the Meßspulen- _I AC φ, φ, the _UI must be the correct sign in displacement of the phases.

The phase shift ϕ _UI between the measuring coil alternating current and the voltage of the sensor coil results in

and is measured as the time interval T _UI , between the comparator switching point t _{U of} the measuring coil alternating voltage U and the comparator switching point t _{I of} the measuring coil alternating current I , the frequency f or the period T of the alternating current or alternating voltage signal having to be taken into account.

To determine the phase shift ϕ _UI , the time interval T _{UI is} measured based on the period T of the measuring coil alternating signal. By switching the two output signals of the two comparators by means of digital gates, flip-flops, memory elements or flip-flops or combinations of these digital logic elements, a pulse train with a pulse repetition frequency or pulse rate is obtained which is derived from the frequency f of the measuring coil AC voltage and is therefore identical or twice the size. Digital logic elements are also known which contain such comparators as an input stage. In the simplest case, such an analog comparator can be an operational amplifier that is not fed back. The duty cycle of the pulse train thus corresponds to the phase shift ϕ _UI . The measurement of the duty cycle can be converted into an analog or digital signal using analog and digital electrical or electronic means (not described in more detail here) and evaluated mathematically and in terms of circuitry, no low-pass filtering being necessary.

The measurement of the complex measuring coil impedance Z takes place, for example, by means of the above-described measurement of the phase shift ϕ _UI or the measuring coil impedance Z. The measuring coil impedance Z can, for example, by means of an effective value measurement of the measuring coil alternating current I _eff and the measuring coil change voltage U _eff .

So is the measuring coil effective resistance

and the measuring coil reactance

the sensor coil is determined, the calculation of the quantities R and X _L can be carried out by means of a mathematical circuit combination with known electrical engineering means. The complex measuring coil impedance Z of the sensor coil of an inductive proximity sensor is composed, as described above, of the parallel connection of the field loss resistor R _F and the pure reactance resistor X _L and the winding resistor R ₁ as a series resistor. The largest proportion of the active component of the complex measuring coil impedance Z is the strongly temperature-dependent winding resistance R _CU , which is referred to below as R ₁ .

It is therefore the object of the present invention to provide a measuring method and a measuring device in which the temperature dependency of the coil is no longer included in the measurement of the distance of the control flag, so that, for example, in the application as a proximity sensor, the distance or as a proximity switch, the switching distance is very great is measured stably over a wide temperature range. The device according to the invention is characterized in that measuring means are provided to measure the field loss resistance R _F and the pure measuring coil reactance X _L directly. This device according to the invention is based on the knowledge that this field loss resistance R _F and the pure coil reactance X _{L are} dependent on the distance s of the control lug, while the winding resistance R _{1 is} independent of the distance s of the control lug.

The field loss resistance R _F and the pure reactance X _{L of} the sensor coil cannot be measured directly from the measuring coil alternating voltage U and from the measuring coil alternating current I , since the measuring coil alternating voltage U is measured too large by the voltage drop across the winding resistor R ₁ . The invention is therefore based on the further finding that the voltage drop across the parallel circuit of R _F and X _L , the so-called induced measuring coil alternating voltage U _ind , is to be measured. The induced measuring coil alternating voltage U _ind as the induced measuring coil primary alternating voltage U _{1, ind} = U ₁ ) can only be measured by means of an auxiliary coil L ₂ as the induced measuring coil secondary alternating voltage U _{2, ind} = U ₂ . The auxiliary coil L ₂ is mounted directly on the measuring coil L ₁ and directly magnetically coupled to it, so that the coupling factor k is as close as possible to one. Corresponding to the known transmission of alternating voltages by means of a transformer or transformer, the induced measuring coil primary alternating voltage of the primary side U ₁ to the induced measuring coil secondary alternating voltage of the secondary side U ₂ behaves like the number of turns of the primary side N ₁ to the number of turns of the secondary side N ₂ .

The induced measuring coil secondary alternating voltage U ₂ is therefore directly proportional to the induced measuring coil primary alternating voltage U ₁ , which cannot be measured directly, the winding ratio N ₁ / N _{2 being} arbitrary, but must be known. The winding resistance R _{2 of} the secondary auxiliary winding L ₂ does not go into a voltage measurement if this measurement is made with high resistance, ie the measuring current is very small, or the induced measuring coil secondary AC voltage U _{2 is} measured without power; in this case the induced measuring coil secondary alternating voltage U _{2 is} equal to the voltage at the coil connections. The advantage of the powerless measurement of the induced measuring coil secondary AC voltage is that the temperature-dependent winding resistance R ₂ no longer influences the measurement. In a preferred embodiment, the induced measuring coil secondary alternating voltage U _{2 is made the} same size as the induced measuring coil primary alternating voltage U ₁ by making the number of turns on the primary side N ₁ and on the secondary side N _{2 the} same size . This case can be realized in a simple manner by both windings being wound together bifilarly, so that the coupling losses are low and the pure inductance of the coil L ₁ on the primary side becomes almost the same as the inductance of the coil L ₂ on the secondary side. The measurement of the measuring coil alternating current I , which flows for example through the coil L ₁ on the primary side, is preferably carried out by means of a voltage measurement in the manner described above with an ohmic resistor as a current-voltage converter.

In a further embodiment of the invention, the measuring coil forms the inductance of an LC series resonant circuit or LC parallel resonant circuit, which is excited to oscillate at its resonance frequency f ₀ in a known oscillator. The advantage of this excitation of the measuring coil of an inductive proximity sensor is the minimal external energy requirement of an oscillator which is operated in its resonance, ie the power consumption of an oscillator has its minimum when excited in resonance.

In a further embodiment of the invention, in an inductive proximity switch or proximity initiator, a mathematical circuit-technical link is made with digital electronic means that a binary output signal of the switch or initiator is generated in such a way that the state of the binary output signal changes only when the Control flag falls below or exceeds a specified switching distance s ₀ , depending on the design of the switch. The binary output signal of the inductive proximity switch is therefore intended to indicate whether a control flag is located further away from a closer switching distance s ₀ from or closer to the active sensor surface of the measuring coil of the proximity switch.

The devices according to the invention will be explained in more detail with reference to drawings will:

Fig. 1a shows a lossy coil with the impedance Z ;

FIG. 1b shows the equivalent circuit of a lossy coil with inductance L, with the field loss resistance R _F due to the attenuation and losses of the directional electromagnetic high-frequency field and to the wire resistance R _1;

Fig. 2a shows a lossy coil with two magnetically coupled windings;

FIG. 2b shows the equivalent circuit of a lossy coil with two magnetically coupled windings;

Fig. 3 shows the basic method for measuring the phase ϕ _UI regardless of the wire resistance R ₁ ;

Fig. 4 shows the dependence of U _L , I _L and quantities derived therefrom on the distance s of a control vane;

Fig. 5a shows an embodiment of an inductive proximity sensor with a simple measuring coil and two voltage;

FIG. 5b shows the pulse diagram of the inductive proximity sensor according to FIG. 5a;

Fig. 6 shows an extended embodiment of an inductive proximity sensor with a voltage-coupled auxiliary coil according to Figure 2 and excited by means of an oscillator.

FIG. 7 shows an embodiment of an inductive proximity switch for measuring the phase ϕ _UI according to FIG. 3 with a phase shifter and a phase comparator;

FIG. 8 shows an extended exemplary embodiment according to FIG. 7 with a Schmitt trigger output.

An analysis of a lossy coil with a winding according to FIG. 1 shows that the phase shift ϕ _UI in the materials and frequencies in question for an inductive proximity switch is determined by the wire resistance R _{1 of} the lossy coil.

In practice, it can be assumed that the winding or wire resistance R _{1 is} significantly smaller than the field loss resistance R _F , so that the relationship for the phase shift ϕ _{UI is} simplified.

The use of a coil with two magnetically coupled windings according to FIG. 2 allows access to the phase ϕ _UI practically without - as was shown above - the influence of the two measuring coil wire resistors R ₁ and R ₂ . With the coil, which is connected as the first winding with the inductance L ₁ between the terminals A and B, a second winding with the inductance L _{2 is} wound bifilarly, which is led out between the terminals A 'and B'. As a result, both windings have the same inductance L ₁ = L ₂ = L, the combined effect of the mutual inductance

is canceled and the connections A 'and B' allow one of the two wire resistors R ₁ and R ₂ practically uninfluenced measurement. A stranded wire can be used as a winding wire for both windings or solid copper enamelled wire can be used separately for both windings. The connections A and A 'or B and B' denote those connections which have the same voltage polarity when an AC voltage is applied or to which the windings are connected in the same winding direction.

In principle, different windings can also be used, then they are Secondary side parameters considering the turns ratio and the To convert the coupling factor to the primary side. But this bill has none Influence on the phase relationship to be evaluated.

To measure the phase ϕ _UI , an alternating current source with the measuring coil alternating current I is connected to the terminals A and B of the coil according to FIG . In the second winding, the measuring coil secondary AC voltage U _{2 is} coupled, which can be measured directly at the terminals A 'and B' in a powerless, high-resistance voltage measurement with I ₂ = 0. If the search coils I-AC is unknown, the 'present between the terminals A and A and the measuring coils-Wech selstrom I proportional voltage U ₁ = R ₁ can. I (with I ₂ = 0) can be evaluated. If both AC voltages U and U _{i are} measured without power, or if the measuring coil alternating current I is evaluated directly and the measuring coil AC voltage U is measured without power, the following results for the phase:

Thus advantageously neither the amplitude of the excitation current or voltage, nor the temperature-dependent wire resistances in the measurement.

In Fig. 5, an inductive proximity sensor with a simple measuring coil of the com plex measuring coils impedance Z of FIG. 1 is shown. The measurement of the zero crossing of the measuring coil alternating voltage U at the sensor coil or the measurement of the point in time when the first comparison direct voltage U _{V, U is} exceeded and undershot by the measuring coil alternating voltage U is carried out with the voltage comparator K ₁ . At the output of the first comparator K _{1 there} is a pulse train with the pulse repetition frequency f of the measuring coil alternating voltage U , which has "logical one" for the positive half-wave of the measuring coil alternating voltage U or when the comparison direct voltage U _{V, U is} exceeded the negative half-wave of the measuring coil alternating voltage U or when falling below the comparison direct voltage U _{V, U has} "logic zero" or vice versa.

The measurement of the zero crossing of the measuring coil alternating current I as the reference alternating voltage U _I at the reference alternating current resistor Z _ref or the measurement of the point in time at which the second comparative direct voltage U _{V, I is} exceeded and undershot by the reference alternating voltage U _I using the second voltage Comparator K _{2 also} results in a pulse train with the pulse _repetition frequency f, which is however phase-shifted by the phase shift ϕ _UI∞ . With the help of a logical AND gate UG, which logically combines the output signal of the first comparator K ₁ and the inverted output signal of the second comparator K ₂ , a pulse train F with the pulse repetition frequency f or the period T = 1 / f and the pulse width is obtained T _UI , which according to Equation 6 is directly proportional to the phase shift ϕ _UI∞ . With further digital modules, not shown here, such as a clock generator, counter, gate and flip-flops, the phase angle ϕ _UI∞ or the pulse width T _UI can be measured digitally by means of known measuring arrangements.

According to FIG. 4, the phase φ in the absence of the control flag _UI∞ or very large distance is s → ∞ almost exactly 90 degrees. The phase ϕ _UI also decreases as the distance s decreases. When using a proximity switch according to the invention, the phase ϕ _{UI is} compared with a reference value ϕ ₁ , below which a switching process is triggered. If a stable switching point is to be achieved despite a possible drifting frequency f of the excitation, the reference phase ϕ ₁ must be generated independently of the frequency, e.g. B. with a first all-pass. However, if a stable frequency can be assumed, then the reference phase ϕ ₁ can be done with a simpler arrangement, e.g. B. a low-pass filter or a delay element. For example, it is possible to generate the effect of the phase angle ϕ ₁ by means of the reference impedance Z _ref through a phase shift ϕ ₁ between the measuring coil alternating current I and the reference alternating voltage U _I. The field loss resistance R _F and the inductance L of the measuring coil are both different depending on the distance s and material of the control vane, as well as the frequency f of the excitation. If the frequency f is selected correctly, a greatly reduced material dependency can be achieved.

There is also the possibility of using the measuring coil as a frequency-determining part of the excitation, e.g. B. together with a capacitor C in the parallel resonant circuit of an oscillator so as to reduce the energy requirement. Using the oscillation condition of such a parallel resonant circuit, the following then follows for the phase:

With

The most significant advantage of the method according to the invention is the high switching speed that can be achieved thereby. At each zero crossing, the phase difference can be determined, e.g. B. at a frequency f = 50 kHz, a switching process can be triggered approximately every 10 microseconds. In contrast, with conventional inductive proximity sensors which evaluate the amplitude of a parallel resonant circuit, the energy stored in the resonant circuit must first be reduced. The time is constant:

If you take z. B. the numerical values of L = 160 µH and R = 0.3 Ω of a measuring coil typically used in inductive proximity sensors, the time constant τ _{LC is} about 270 µs. The time constant solely by the energy in the resonant circuit is thus many times greater than the achievable switching time of the proximity switch according to the invention.

Due to the high speed and the purely digital output signal, in increased interference suppression by evaluating several in a very simple manner Zero crossings take place.

In FIG. 6, an inductive proximity sensor with a measuring coil L ₁ is shown, the ₂ voltage an auxiliary coil L for coupling out the measured induced measurement coils exchange has U ₁ = U _2. A capacitor C is connected together with the measuring coil L ₁ and a reference resistor Z _ref to the frequency-determining parallel resonant circuit of the oscillator OSZ. The reference resistor Z _ref is used for the current-voltage conversion, since the wire resistor R ₁ which can also be used on the primary side (not shown here in the drawing) often has a value which is too low for reliable evaluation.

A reference phase ϕ ₁ can be generated with an analog all-pass Ph ₁ as the first phase shifter. The signs of the voltage signals thus generated are converted into digital signals with the two analog comparators K ₁ and K ₂ . The two output signals of the two comparators K ₁ and K ₂ are - as described above in the embodiment according to FIG. 5 - evaluated by means of a logical AND gate UG and thus a pulse train with the frequency f = 1 / T is generated.

An exemplary embodiment of a proximity switch according to the invention is shown in FIG. 7. A capacitor C is connected together with the measuring coil L ₁ and a reference AC resistance Z _ref to the frequency-determining parallel resonant circuit of the oscillator OSZ. The reference alternating current resistor Z _ref is used for the current-voltage conversion, since the wire resistor R ₁ which can also be used on the primary side (not shown here in the drawing) often has a value which is too low for reliable evaluation.

The reference phase ϕ ₁ is generated with an analog all-pass Ph ₁ as the first phase shifter. The signs of the voltage signals thus generated are converted into digital signals with the two analog comparators K ₁ and K ₂ . The first comparator K ₁ controls the clock input of the edge-controlled first D memory element or D flip-flop D ₁ . This is used as a digital phase comparator; it stores the state at the time of the clock edge, i.e. the sign of the current, until the next clock edge. This state is led to the output of the switching signal S. The function of a D flip-flop is shown, for example, in E. Schrüfer, Electrical Measurement Technology, 5th edition, Munich, Vienna, Hanser-Verlag, 1992, page 330.

For ease of use, it is desirable that the proximity switch not switch back and forth continuously when the control lug is at the sensing range. The embodiment of an inductive proximity switch with switching hysteresis according to FIG. 8 shows a simple improvement . The additional devices of a second phase shifter Ph ₂ , a second D flip-flop D ₂ and an RS flip-flop RS cause the behavior of a Schmitt trigger. The second phase shifter Ph ₂ delays the sign signal of the current by a small hysteresis phase ϕ ₂ . The resulting signal is fed to the second flip-flop D ₂ , whose function is identical to the first flip-flop D ₁ . The state-controlled flip-flop RS has the effect that only states which are outside the hysteresis range defined by a second phase shifter Ph ₂ cause a change in the switching signal S.

Instead of the gate UG, either a NAND gate or an OR gate or a NOR gate can be used without changing the function and the arrangement of the circuit according to FIGS. 5 and 6.

If either an exclusive-OR gate or antivalence gate or an equivalence gate is used instead of the gate UG in FIGS. 5 and 6, a pulse sequence F with twice the frequency 2 f or with half the period T / 2 = 1 is produced / 2f of the measuring coil alternating current I, in which case the outputs of the two comparators K ₁ and K _{2 are} connected directly to the two inputs of these gates.

Without changing the functions, the two outputs of the two comparators K ₁ and K _{2 can be} interchanged in the circuits according to FIGS. 5 to 8.

Likewise, the first phase shifter Ph _{1 can} also be inserted between the reference alternating current resistance Z _ref and the input of the second comparator K ₂ without the basic behavior of the circuits according to FIGS. 5 to 8 changing.

The output of the second comparator K ₂ may continue in FIG. 8, directly with the D-A-transitions of the two flip-flops D ₁ and D ₂ and also the output of the first comparator K ₁ to the clock input of the second D flip-flop D ₂ and the The output of the second phase shifter Ph _{2 can be} connected to the clock input of the first D flip-flop D ₁ without changing the basic behavior of the circuit.

The rela in the circuits of Figs. 7 and 8 specified memory members hung as flip-flops represent only one embodiment of this invention. The D flip-flop, for example, a clocked RS flip-flop in which the input signal on the set input and the inverted input signal the reset input is routed. Alternatively, the use of transparent or state-controlled D flip-flops is possible. Other versions of digital flip-flops, flip-flops or memory elements can also be used if the functions of the circuits according to the invention presented above are achieved.

Another embodiment of the proximity switch according to the invention is Speed measurement of high-speed shafts and disks as speed sensors. For this purpose, the movement of a cam attached to a rotating shaft or the slot provided there is scanned by the fixed speed sensor.

Claims (29)

1. Inductive proximity sensor for measuring the distance of an electrically conductive or permeable measuring object as a control flag, a measuring coil of the proximity sensor generating an electromagnetic high-frequency field by feeding the measuring coil with a measuring coil alternating current I or with a measuring coil alternating voltage U of frequency f, and wherein this field is influenced by the measurement object, so that the impedance Z of the measuring coil is changed and so that this change in the measuring coil impedance Z serves as a measure of the distance of the control lug and is measured by electronic means, characterized in that

• 1. - that electronic means are available to generate a pulse train whose frequency f is equal to the frequency or twice the frequency of the measuring coil alternating current I or the measuring coil alternating voltage U and whose pulse width T _{UI is} proportional to the phase shift ϕ _UI , which is measured between the measuring coil alternating voltage U and the measuring coil alternating current I directly during each half cycle or a multiple of each half cycle of the current and the voltage,
• 2. - That a current / voltage converter converts the measuring coil alternating current I into a reference alternating voltage U _I proportional to the measuring coil alternating current I and
• 3. - That the pulse width T _{UI is} generated by comparing the measuring coil alternating voltage U and the reference alternating voltage U _I with comparative direct voltages U _{V, U} and U _{V, I} by the time start of the pulse width T _UI when exceeding or falling below the AC voltage U or U _I compared to the comparison DC voltage U _{V, U} or U _{V, I} and the end of time when falling below or exceeding the AC voltage U _I or U compared to the comparison DC voltage U _{V, I} or U _{V, U} are set.

2. Inductive proximity sensor according to claim 1, characterized in

• 1. - that the measuring coil alternating voltage U at the sensor coil of the proximity sensor at the measuring coil impedance Z is generated by the measuring coil alternating current I as a voltage drop,
• 2. - that the reference alternating voltage U _I at a reference alternating current resistor was generated with the impedance Z _ref by the measuring coil alternating current I as a voltage drop,
• 3. - that a first analog comparator (K ₁ ) when the measuring coil AC voltage U is exceeded or undercut compared to the comparison DC voltage U _{V, U} a first DC voltage source, a signal change of its output signal from "logic zero" to "logic one" or generated in reverse,
• 4. - that a second analog comparator (K ₂ ) when exceeding or falling below the reference AC voltage U _I , compared to the comparison DC voltage U _{V, I} a second DC voltage source, a signal change of its output signal from "logic zero" to "logic one""or vice versa and
• 5. - that the output signal of the first comparator (K ₁ ) and the inverted output signal from the second comparator (K ₂ ) or the inverted output signal of the first comparator (K ₁ ) and the non-inverted output signal of the second comparator (K ₁ ) with a gate (UG) is a pulse train F with the pulse width T _UI , the pulse width being directly proportional to the phase shift ϕ _UI between the measuring coil alternating current I and the measuring coil alternating voltage U.

3. Inductive proximity sensor according to claim 2, characterized in that the gate (UG) is an AND gate, so that a pulse train F with the simple frequency f = 1 / T of the measuring coil alternating current I arises. 4. Inductive proximity sensor according to claim 2, characterized in that the gate (UG) is a NAND gate, so that a pulse train F with the simple frequency f = 1 / T of the measuring coil alternating current I arises. 5. Inductive proximity sensor according to claim 2, characterized in that the gate (UG) is an OR gate, so that a pulse train F with the simple frequency f = 1 / T of the measuring coil alternating current I arises. 6. Inductive proximity sensor according to claim 2, characterized in that the gate (UG) is a NOR gate, so that a pulse train F with the simple frequency f = 1 / T of the measuring coil alternating current I arises. 7. Inductive proximity sensor according to claim 2, characterized in that the gate (UG) is an exclusive OR gate or antivalence gate by the outputs of the two comparators (K ₁ ) and (K ₂ ) directly with the two inputs of Gates (UG) are connected so that a pulse train F with twice the frequency 2f of the measuring coil alternating current I is produced. 8. Inductive proximity sensor according to claim 2, characterized in that the gate (UG) is an equivalence gate by the outputs of the two com parators (K ₁ ) and (K ₂ ) ver directly with the two inputs of the gate (UG) be bound so that a pulse train F with twice the frequency 2f of the measuring coil alternating current I is produced. 9. Inductive proximity sensor according to one of claims 3 to 8, characterized in that the reference AC resistance Z _{ref is} an ohmic resistance. 10. Inductive proximity sensor according to one of claims 3 to 8, characterized in that the reference AC resistance Z _{ref is} a low-loss capacitor with the capacitance C _ref , so that the phase shift ϕ _UI by the additional phase shift ϕ ₁ of approximately 90 ° ( Degrees) is reduced. 11. Inductive proximity sensor according to one of claims 3 to 8, characterized in that the reference AC resistance Z _{ref is} a low-loss coil with the inductance L _ref , so that the phase shift ϕ _UI by the additional phase shift ϕ _I of approximately 90 degrees is enlarged. 12. Inductive proximity sensor according to one of claims 3 to 8, characterized in that the reference AC resistance Z _ref consists of a combination of an ohmic resistor, a low-loss capacitor or a low-loss coil, so that the phase shift ϕ _UI by the additional phase shift ϕ _I between the measuring coil alternating current I and the measuring coil alternating voltage U is changed. 13. Inductive proximity sensor according to one of claims 3 to 8, characterized in that the comparison DC voltage U _{V, U is} zero, so that the first comparator (K ₁ ) changes its output signal at a zero crossing of the measuring coil AC voltage U I. 14. Inductive proximity sensor according to one of claims 3 to 8, characterized in that the comparison DC voltage U _{V, I is} zero, so that the second comparator (K ₂ ) changes its output signal at a zero crossing of the reference AC voltage U _I. 15. Inductive proximity sensor according to one of claims 3 to 8, characterized in that an oscillatory resonant circuit is formed from the measuring coil with the measuring coil impedance Z and from the reference AC resistance Z _ref by the series circuit from the measuring coil impedance Z and the reference AC resistance Z _{ref is} supplemented by an AC resistance Z _p connected in parallel. 16. Inductive proximity sensor according to claim 15 and one of claims 9 to 11, characterized in that the AC resistor Z _{p connected in} parallel is a capacitor with the capacitance C. 17. Inductive proximity sensor according to one of claims 3 to 8, characterized in that an oscillatory resonant circuit is formed from the measuring coil with the measuring coil impedance Z and from the reference AC resistance Z _ref by the series circuit from the measuring coil impedance Z and the reference AC resistance Z _{ref is} supplemented by an AC resistance Z _r connected in series. 18. Inductive proximity sensor according to claim 17 and one of claims 9 to 11, characterized in that the series-connected AC resistance Z _{r is} a capacitor with the capacitance C. 19. Inductive proximity sensor according to one of claims 15 to 18, characterized characterized in that the resonant circuit and an electronic amplifier form the components of an oscillatory oscillator by the Reso resonance circuit is the feedback of the oscillator and thus the resonance frequency f of the oscillator. 20. Inductive proximity sensor according to claim 19, characterized in that the Oscillator has a sinusoidal output signal. 21. Inductive proximity sensor according to one of claims 3 to 8, characterized in that

• 1. - That the measuring coil consists of two magnetically coupled windings, the first winding with the connections A and B, with the number of turns N ₁ and with the inductance L ₁ by means of the measuring coil alternating current I generates an electromagnetic high frequency field, and
• 2. - that the measuring coil primary alternating voltage U ₁ induced in the first winding, which is the voltage drop across the inner parallel connection of inductor L ₁ and field loss resistance R _F due to the measuring coil alternating current I , by means of the second winding with the connections A ' and B ', with the number of windings N ₂ and with the inductance L ₂ as the induced measuring coil secondary AC voltage U ₂ , so that the winding resistance R _{1 of} the first winding is not included in the measurement of the measuring coil secondary AC voltage U ₂ comes in.

22. Inductive proximity sensor according to claim 21, characterized in that the induced measuring coil secondary AC voltage U _{2 is} measured at the second winding with high resistance, ie the measuring current I _{2 is} very small and thus almost zero, so that the induced measuring coil secondary AC voltage U _{2 is} almost independent of the winding resistance R _{2 of} the second winding. 23. Inductive proximity sensor according spoke 21 or 22, characterized in that the two windings of the measuring coil are wound bifilarly, so that the number of turns N _{1 is} equal to the number of turns N ₂ , the two windings have the same inductance L ₁ equal to L ₂ and thus the induced measuring coil secondary AC voltage U ₂ on the second winding is almost the same size as the induced induced measuring coil primary AC voltage U _{1 of} the first winding. 24. Inductive proximity sensor according to claim 22 or 23, characterized in that the measuring coil alternating current I is measured as a voltage drop U _i at the winding resistance R _{1 of} the first winding, by measuring the voltage voltage U _i between the connections of the two windings with the same voltage polarity and the two other connections of the two windings are at the same voltage potential. 25. Inductive proximity sensor according to one of claims 3 to 8, characterized in that the measuring coil AC voltage U , which is guided by the measuring coil to the input of the first comparator (K ₁ ), by means of a first phase shifter (Ph ₁ ) by one Reference phase ϕ _{1 is} shifted. 26. Inductive proximity sensor according to one of claims 21 to 24, characterized in that the induced measuring coil secondary AC voltage U ₂ or U ₂ ', from the second winding with the connections A' and B 'to the input of the first comparator (K ₁ ) is carried out by means of a phase shifter (Ph ₁ ) by a reference phase ϕ ₁ . 27. Inductive proximity switch or proximity initiator according to claim 1, characterized in that

• 1. - that the output of the first comparator (K ₁ ) is fed to the clock input of an edge-controlled D memory element or first D flip-flop (D ₁ ),
• 2. - that the output of the second comparator (K ₂ ) is led to the state input or D input of the first D memory element (D ₁ ), so that the D memory element has the function of a digital phase comparator and the output of the D- Memory element leads the switching signal S, and
• 3. - that the switching point or the change in the state of the switching signal S is adjustable by means of the reference phase ϕ _{1 of} the first phase shifter (Ph ₁ ) or by means of the phase of the reference AC resistance Z _ref .

28. Inductive proximity switch or proximity initiator with switching hysteresis according to claim 27, characterized in that

• 1. that either the reset input or the set input of a state-controlled RS flip-flop or RS flip-flop (RS) is connected to the output of the first D flip-flop (D ₁ ),
• 2. that either the set input or the reset input of the state-controlled RS flip-flop (RS) is connected to the output of a second edge-controlled D memory element or D flip-flop (D ₂ ),
• 3. that the two clock inputs of the two D flip-flops (D ₁ ) and (D ₂ ) are connected to the same output of the first comparator (K ₁ ),
• 4. - that the state input or D input of the first D flip-flop (D ₁ ) is connected directly to the output of the second comparator (K ₂ ) and
• 5. - That the state input or D input of the second D flip-flop (D ₂ ) is connected via a second phase shifter (Ph ₂ ) to the output of the second comparator (K ₂ ), the input signal of the D input of the second D -Flipflops (D ₂ ) is shifted by the hysteresis phase ϕ ₂ , so that the behavior of a Schmitt trigger results for the output signal or the switching signal S of the RS flip-flop (RS), ie the output signal S only changes when the change the phase shift ϕ _UI between the two output signals of the comparators (K ₁ and K ₂ ) is greater than the hysteresis phase ϕ ₂ .

29. Inductive proximity holder according to one of claims 1 to 28, characterized characterized in that the inductive proximity sensor or proximity switch is used as a tachometer.