DE112004001976T5 - High linearity Doherty communication amplifier with bias control - Google Patents

High linearity Doherty communication amplifier with bias control

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Publication number
DE112004001976T5
DE112004001976T5 DE112004001976T DE112004001976T DE112004001976T5 DE 112004001976 T5 DE112004001976 T5 DE 112004001976T5 DE 112004001976 T DE112004001976 T DE 112004001976T DE 112004001976 T DE112004001976 T DE 112004001976T DE 112004001976 T5 DE112004001976 T5 DE 112004001976T5
Authority
DE
Germany
Prior art keywords
amplifier
power
voltage control
control signal
output
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
DE112004001976T
Other languages
German (de)
Inventor
Junghyun Kim
Youngwoo Kwon
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Wavics Inc Palo Alto
Wavics Inc
Original Assignee
Wavics Inc Palo Alto
Wavics Inc
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority to US10/690,923 priority Critical
Priority to US10/690,923 priority patent/US7053706B2/en
Application filed by Wavics Inc Palo Alto, Wavics Inc filed Critical Wavics Inc Palo Alto
Priority to PCT/US2004/034797 priority patent/WO2005043747A2/en
Publication of DE112004001976T5 publication Critical patent/DE112004001976T5/en
Ceased legal-status Critical Current

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Classifications

    • HELECTRICITY
    • H03BASIC ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F1/00Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0288Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers using a main and one or several auxiliary peaking amplifiers whereby the load is connected to the main amplifier using an impedance inverter, e.g. Doherty amplifiers

Abstract

A system for biasing a power amplifier, comprising:
a carrier amplifier coupled to an input stage for amplifying an input signal; and
a peaking amplifier coupled to the input stage for amplifying the input signal, the peaking amplifier configured to receive a voltage control signal for biasing the peaking amplifier, the voltage control signal based on power levels of signals transmitted by a remote base station.

Description

  • relative Registrations
  • These Application is a continuation-in-part of the U.S. patent application Serial No. 10 / 432,553, filed May 21, 2003, entitled "Power Amplification Apparatus of Portable Terminal ", which is hereby incorporated by reference is. U.S. Patent Application Serial No. 10 / 432,553 is a National Phase application for and claims the priority from International Application No. PCT / KR02 / 00163 on February 4, 2002, the priority of the Korean utility patent application No. 2002-5924 filed on Feb. 1, 2002, claims both here for everyone Purposes are incorporated by reference.
  • technical area
  • The The present invention relates to a power amplification circuit for use in wireless communication technologies and in particular to a power amplifier circuit at a mobile handset.
  • State of technology
  • There Mobile handsets for Wireless communication services are used, smaller and lighter will also take battery size and performance from. Consequently, the effective speaking time (i.e., transmission time) of mobile computing devices becomes mobile phones and the like (i.e., handsets).
  • at a conventional one Mobile handset, the high frequency (RF) power amplifier consumes the most power consumed, unlike the overall system of the mobile handset. Thus leads the RF power amplifier, which has a low efficiency, usually one Deterioration of the efficiency for the whole system and reduced accordingly the talk time.
  • Out This reason was a lot of effort related to the efficiency of the RF power amplifier the field of power amplification to increase. For a solution Recently, a Doherty-type power amplifier was used as a circuit for Increasing the efficiency of the RF power amplifier introduced. Different as conventional Power amplifier, their efficiency over the low output power range is low, the Doherty-type power amplifier is designed to for optimal efficiency over a broad output power range (eg in low, intermediate and high output power ranges).
  • One ordinary Doherty-type power amplifier design includes both a carrier as well as a top amplifier. The carrier amplifier (i.e. H. Power or main amplifier), which is formed by relatively small transistors, is effective for optimal efficiency up to a certain low Maintain output power level. The peak amplifier (i. H. Supplementary or Auxiliary amplifier) is operative in a cooperative manner with the carrier amplifier to produce a high Efficiency until the power amplifier as Whole produces a maximum output power. If the power amplifier within a low power output range is effective, only the Carrier amplifier effective; the top amplifier, which is biased as a class B or C is not effective. However, if the power amplifier is effective within a high power output range the top amplifier active and can introduce a nonlinearity in the overall power amplifier, since the top amplifier as a highly nonlinear class B or class C amplifier biased is.
  • Theoretically is the one mentioned above Doherty-type power amplifier designed to to be effective while same the linearity specification over one Whole output power range met, and being a high efficiency is maintained. As described above is satisfied, because the Doherty-type power amplifier has a Carrier amplifier and a peak amplifier which are effective with each other, the Doherty-type power amplifier in practice not the linearity specification (eg, in terms of phase or gain characteristics) over the total output power range, with high efficiency is maintained.
  • In summary For example, in the above-mentioned Doherty type power amplifier in FIG the related art difficult, the linearity characteristics of such a Power amplifying device predict what makes it difficult to have such linearity characteristics because the peak amplifier at a relatively constant, low DC level is biased, such. A stream, around the top amplifier as a class B or C amplifier.
  • Short Summary the invention
  • There is a need to overcome the disadvantages of the prior art and to provide at least the advantages described below. In order to solve the above-mentioned problems associated with the prior art, a specific embodiment of the present invention provides a power amplifier in a mobile handset, the efficiency and linearity by applying a voltage control signal to a peak amplifier to bias the peak amplifier, improved. Normally, a baseband modem chipset generates the voltage control signal in accordance with power levels of signals received from a base station. In particular, in a low output power range, a control voltage in a first state is applied to the peak amplifier so that the power amplifier operates in a Doherty mode, and in the high output power range a control voltage in a second state is applied to the peak amplifier to reduce the nonlinearity characteristic of the Power amplifier sufficiently to handle. In one embodiment of the invention, the voltage control signal in the first state is a high voltage state signal, and the voltage control signal in the second state is a low voltage state signal. In another embodiment of the invention, the voltage control signal in the first state is the low voltage state signal, and the voltage control signal in the second state is the high voltage state signal.
  • Of the power amplifier in a mobile handset according to one embodiment The present invention comprises a phase shifter, which with input terminals a carrier amplifier and a peak amplifier coupled to generate a phase difference between carrier amplifier and Peak amplifier input signals to the phase shift at an output of the carrier and peak amplifier to compensate; and an output adjusting unit for transmitting the Output powers from the carrier amplifier and the top amplifier to an output stage. Furthermore includes the top amplifier a voltage control unit configured to receive the voltage control signal to receive and the peak amplifier according to the power levels of signals, that are received from the base station to harness.
  • at an embodiment is the phase shifter z. B. with a 3dB hybrid coupling element implemented for distributing certain input powers to the Carrier amplifier and the top amplifier, to minimize an overlay between the carrier amplifier and the top amplifier and for sending signals in a way that phase one Input power, which is applied to the peak amplifier, essentially by 90 ° with respect to Phase of an input power is applied to the carrier amplifier is.
  • at a further embodiment the phase shifter is a phase difference compensator that intervenes the input stage of the power amplifier and the peak amplifier is connected, to delay the phase of an input signal applied to the peak amplifier is 90 ° relative to the Phase of an input signal applied to the carrier amplifier.
  • The Voltage control unit controls a bias DC current of the peak amplifier via the voltage control signal, such that the power amplifier is operated in a Doherty mode if the power amplifier is within the low output power range is effective. If the power amplifier on the other hand is effective within the high output power range, controls the Voltage control unit the bias DC of the top amplifier via the Voltage control signal such that the power amplifier has non-linearity characteristics Fulfills.
  • Short description the drawings
  • 1 Fig. 10 is a block diagram showing the structure of a power amplifier in a mobile handset according to an embodiment of the present invention;
  • 2 shows an equivalent circuit of a 3dB hybrid coupling element used in the power amplifier of 1 can be used;
  • 3A FIG. 13 is a block diagram of the carrier amplifier incorporated in FIG 1 illustrated in accordance with an embodiment of the invention;
  • 3B FIG. 4 is a block diagram of the input matching unit incorporated in FIG 3A illustrated in accordance with an embodiment of the invention;
  • 3C FIG. 12 is a block diagram of the inter-stage matching unit disclosed in FIG 3A illustrated in accordance with the present invention;
  • 3D FIG. 12 is a block diagram of the first stage amplifier incorporated in FIG 3A illustrated in accordance with an embodiment of the invention;
  • 3E is a block diagram of the second stage amplifier used in 3A illustrated in accordance with an embodiment of the invention;
  • 4A is a block diagram of the peak amplifier used in 1 illustrated in accordance with an embodiment of the invention;
  • 4B FIG. 12 is a block diagram of the second stage amplifier / voltage control unit incorporated in FIG 4A illustrated in accordance with an embodiment of the invention;
  • 4C FIG. 12 is a block diagram of the second stage amplifier / voltage control unit incorporated in FIG 4A is illustrated, according to a further embodiment of the invention;
  • 4D FIG. 12 is a block diagram of the second stage amplifier / voltage control unit incorporated in FIG 4A is illustrated, according to a further embodiment of the invention;
  • 5 FIG. 4 is a block diagram of the exemplary output adaptation unit incorporated in FIG 1 is illustrated;
  • 6 FIG. 12 shows an equivalent circuit of the exemplary output adaptation unit of FIG 5 that is implemented with focus elements;
  • 7 Fig. 12 is a graph illustrating efficiency characteristics in response to a voltage control signal applied to an exemplary peaking amplifier;
  • 8th Fig. 12 is a graph illustrating non-linearity characteristics in response to a voltage control signal applied to an exemplary peaking amplifier;
  • 9 Fig. 12 is a graph illustrating efficiency characteristics according to modes of the power amplifier according to an embodiment of the present invention;
  • 10 Fig. 12 is a graph illustrating nonlinearity characteristics according to modes of the power amplifier according to a specific embodiment of the present invention;
  • 11 Fig. 12 is a graph illustrating gain characteristics according to modes of the power amplifier according to the present invention; and
  • 12 Fig. 10 is a block diagram showing the structure of a power amplifier according to another embodiment of the present invention.
  • Detailed description specific embodiments
  • in the Following is a detailed description with reference to the attached Drawings re an exemplary power amplifier in a mobile handset according to different embodiments of the present invention.
  • 1 illustrates the structure of an exemplary power amplifier in a mobile handset according to a specific embodiment of the present invention. The power amplifier 100 who in 1 is illustrated, has a hybrid coupling element, such. An exemplary 3dB hybrid coupling element 110 , a carrier amplifier 120 , a top amplifier 130 and an output adjusting unit 140 on. The 3dB hybrid coupling element 110 distributes certain input power to the carrier amplifier 120 and the top amplifier 120 , minimizes interference between the carrier amplifier 120 and the top amplifier 130 and sends signals so that the phase of an input power of the peak amplifier 130 by 90 ° (λ / 4) with respect to the phase of an input power of the carrier amplifier 120 is delayed. Accordingly, the 3dB hybrid coupling element compensates 110 a later processing of output signals from the carrier amplifier 120 and the top amplifier 130 through an output adapter 140 by generating a phase delay of 90 ° (λ / 4) at the output matching unit 140 between the phases of output signals from the carrier amplifier 120 and the top amplifier 130 , Thus, the introduction of a phase difference between the phases results in output powers from the carrier amplifier 120 and the top amplifier 130 through the 3dB hybrid coupling element 110 to subsequent processing of the output powers by the output matching unit 140 to compensate for the balancing of the phases of the output powers and an optimal output power signal at an output stage 70 , The 3dB hybrid coupling element 110 will be in connection with 2 discussed in more detail.
  • The carrier amplifier 120 amplifies signals coming from the 3dB hybrid coupling element 110 be received. In an example, the carrier amplifier comprises 120 a transistor smaller than that of a transistor that drives the peak amplifier 130 forms, can be dimensioned. The ratio of these respective transistor sizes partially determines an output power range over which maximum efficiency can be maintained. The higher this ratio, the wider the output power range over which maximum efficiency can be maintained. One skilled in the art should recognize that each amplifier may include one or more transistors or other similar circuit elements. Further, one skilled in the art should recognize that the carrier amplifier 120 and the top amplifier 130 can be implemented in any known semiconductor technologies, such. Si-LDMOS, GaAs-MESFET, GaAs-pHEMT, GaAs-HBT or the like. The carrier amplifier 120 will be described below in connection with the 3A to 3E discussed in more detail.
  • The top amplifier 130 in which it is is another amplifier for amplifying signals coming from the 3dB hybrid coupling element 110 is essentially not operated while low level input signals to the carrier amplifier 120 are created. This is done by applying a voltage control signal Vc to the peak amplifier 130 allows, so that the peak amplifier 130 as a class B or C amplifier is biased when little or no DC current flows.
  • About the low output power range at which the peak amplifier 130 essentially not operated, the carrier amplifier 120 an output impedance having a relatively constant and high value. Because the top amplifier 130 draws no electricity, the power amplifier can 100 obtained an improved efficiency at an output power level that is lower than the highest output power level that the carrier amplifier 120 can generate.
  • The top amplifier 130 is configured to receive the voltage control signal Vc from a baseband modem chipset (not shown) or from a power amplifier RF processing circuitry (not shown). The baseband modem chipset generates the voltage control signal Vc based on power levels of signals received from a base station (not shown). The power amplifier RF processing circuitry processes signals from the baseband modem chipset and is known to those skilled in the art. The top amplifier 130 will be described below in connection with the 4A to 4D discussed in more detail.
  • The output adjustment unit 140 includes a first λ / 4 transformer 143 , The first λ / 4 transformer 143 is effective as an impedance inverter and is used to provide impedance at a carrier amplifier output port 50 to provide an impedance at a peak amplifier output terminal 60 is inverted. A second λ / 4 transformer 145 at the peak amplifier output terminal 60 the top amplifier 130 fits an output impedance of the power amplifier 100 to a characteristic reference impedance, which is normally 50 ohms. The output adjustment unit 140 is below in connection with the 5 to 6 discussed in more detail.
  • 2 shows an equivalent circuit of the 3dB hybrid coupling element 110 according to an embodiment of the present invention. The embodiment of the 3dB hybrid coupling element 110 from 2 has a plurality of lumped elements comprising a capacitor 111 , an inductor 112 , a capacitor 113 , an inductor 114 , an inductor 115 , a capacitor 116 , an inductor 117 and a capacitor 118 include. At an operating frequency of z. B. about 1.8 GHz nominal capacitances of the capacitors 111 . 113 . 116 and 118 some pico-farad (pF), and nominal inductors inductors 112 . 114 . 115 and 117 amount to some Nano-Henry (nH). After signals through an input stage 10 of the 3dB hybrid coupling element 110 which has the signal coupling of about 3 dB or more, these signals are sent to the carrier amplifier input terminal 30 ( 1 ) and to the peak amplifier input terminal 40 ( 1 ) Posted. The signal at the carrier amplifier input terminal 30 and the signal at the peak amplifier input terminal 40 have a phase difference at or about 90 ° (λ / 4 or quarter wave).
  • For example, the 3dB hybrid coupling element 110 be implemented with a transmission line, such. A coupled-line coupling element, a Lange coupling element, a branch line coupling element or other similar coupling circuits known in the art. As another example, the 3dB hybrid coupling element 110 be implemented using a monolithic microwave integrated circuit (MMIC) integrated circuit chip technology, such. GaAs or any other known semiconductor technologies. That is, the exemplary hybrid coupling element 110 can be fabricated as an integrated circuit that can be packaged as a single power amplifier device or a chip. In another example, the 3dB hybrid coupling element 110 be implemented by the low temperature co-fired ceramic (LTCC) method or other similar technologies.
  • 3A is a block diagram of the carrier amplifier 120 who in 1 is illustrated, according to an embodiment of the invention. In the embodiment of the invention of 3A is the carrier amplifier 120 a two-stage amplifier and includes an input matching unit 305 , a first-stage amplifier 310 , an intermediate stage adjusting unit 315 and a second stage amplifier 320 , The input adaptation unit 305 matches an output impedance of the 3dB hybrid coupling element 110 to an input impedance of the carrier amplifier 120 at. Similarly, the interstage adapter fits 315 an output impedance of the first stage amplifier 310 to an input impedance of the second stage amplifier 320 at. The input adaptation unit 305 and the interstage adjusting unit 315 will be described below in connection with the 3B respectively. 3C discussed in more detail.
  • In addition, the carrier amplifier includes 120 conductor lines 325 which are electrically coupled to a DC bias voltage V1 (not shown) and conductor lines 330 electrically coupled to a DC bias voltage V2 (not shown) for biasing the first stage amplifier 310 and the second stage amplifier 320 , In an exemplary embodiment of the invention, 2.8V <V1 <3.0V and 3.2V <V2 <4.2V, although the scope of the invention is different in bias voltages according to operating characteristics of the first stage amplifier 310 and the second stage amplifier 320 covers.
  • 3B Fig. 10 is a block diagram of the input matching unit 305 , in the 3A is illustrated, according to an embodiment of the invention. The input adaptation unit 305 includes an inductor 306 , a capacitor 307 and a capacitor 308 , The inductor 306 couples the 3dB hybrid coupling element 110 ( 1 ) electrically with the capacitor 307 and the capacitor 308 , In addition, the capacitor 307 electrically coupled to ground, and the capacitor 308 is electric with the first stage amplifier 310 ( 3A ) coupled. In one embodiment of the invention, electrical characteristics of the inductor are 306 , the capacitor 307 and the capacitor 308 selected such that an output impedance of the 3dB hybrid coupling element 110 to an input impedance of the carrier amplifier 120 ( 3A ) connected to a port 30 is measured, adjusted. For example, capacitances of the capacitors 307 and 308 nominally some pico-farad, and the inductor 306 has a nominal inductance of some nano-Henry.
  • 3C Fig. 10 is a block diagram of the interstage adjusting unit 315 , in the 3A illustrated in accordance with the present invention. The Interstage Adaptation Unit 315 includes a capacitor 309 , an inductor 311 and a capacitor 312 , The capacitor 309 couples a signal coming from the first stage amplifier 310 ( 3A ) is received electrically with the inductor 311 and the capacitor 312 , In addition, the inductor 311 electrically coupled to ground, and the capacitor 312 is electric with the second stage amplifier 320 ( 3A ) coupled. In one embodiment of the invention, electrical characteristics of the capacitor are 309 , of the inductor 311 and the capacitor 312 selected such that an output impedance of the first stage amplifier 310 ( 3A ) to an input impedance of the second stage amplifier 320 ( 3A ) is adjusted. For example, capacitances of the capacitors 309 and 312 nominally some pico-farad, and the inductor 311 has a nominal inductance of some nano-Henry.
  • 3D is a block diagram of the first stage amplifier 310 who in 3A is illustrated, according to an embodiment of the invention. The first stage amplifier 310 includes a conventional bias unit 1 (CBU1) 335 , a conventional bias unit 2 (CBU2) 340 and a transistor Q11 345 , In the exemplary embodiment of the invention of 3D is the transistor Q11 345 configured as a common emitter npn bipolar transistor. The CBU1 335 includes a resistor 313 , a diode 314 , a diode 316 , a resistance 317 , a capacitor 318 and a transistor Q1A 319 , The CBU2 340 includes a transmission line 321 and a capacitor 322 , As is known to a person skilled in the art, electrical characteristics of the resistor are 313 , the diode 314 , the diode 316 , the resistance 317 , the capacitor 318 and the transistor Q1A 319 , which for descriptive purposes are referred to as first stage base biasing elements, together with bias DC voltages V1 and V2 selected to be a base of the transistor Q11 345 for a normal operating mode. For example, the resistance 313 have a resistance in a range of several hundred ohms to several kilo-ohms, the resistance 317 may have a resistance value in a range of several ohms to several hundreds ohms, and a Q1A: Q11 transistor size ratio may be approximately in a range of 1: 4 to 1:10. In similar ways, electrical characteristics of the transmission line 321 and the capacitor 322 , which are collectively referred to as first stage collector biasing elements together with the bias voltage V2 selected to be a collector of the transistor Q11 345 for a normal operating mode. For example, electrical characteristics of the first stage base biasing elements are selected to be a base-emitter current I BE (not shown) of the transistor Q11 345 and electrical characteristics of the first-stage collector biasing elements are selected to be a collector-emitter voltage V CE (not shown) of the transistor Q11 345 to specify, thereby allowing the transistor Q11 345 in a normal operating mode and with a predefined gain factor.
  • 3E is a block diagram of the second stage amplifier 320 who in 3A is illustrated, according to an embodiment of the invention. The second stage amplifier 320 includes a conventional bias unit 3 (CBU3) 350 and a transistor Q12 355 , The CBU3 350 includes a resistor 323 , a diode 324 , a diode 326 , a resistance 327 , a capacitor 328 and a transistor Q1B 329 , collectively referred to as second stage base biasing elements the. In the embodiment of the invention of 3E is the coupling of the second stage base biasing elements of the CBU3 350 identical to the coupling of the first stage base biasing elements of the CBU1 335 ( 3D ). However, electrical characteristics of the second stage base biasing elements may or may not be identical to electrical characteristics of the first stage base biasing elements. For example, the resistance 313 ( 3D ) and the resistance 323 have different resistance values, and the transistor Q1A 319 ( 3D ) and the transistor Q1B 329 can have different sizes. In operation, electrical characteristics of the resistor 323 , the diode 324 , the diode 326 , the resistance 327 , the capacitor 328 and the transistor Q1B 329 selected together with bias DC voltages V1 and V2 to a base of the transistor Q12 355 for normal mode operation, based on operating characteristics of transistor Q12 355 and specifications of the power amplifier 100 ( 1 ). For example, the resistance 323 have a resistance in a range of several hundred ohms to several kilo-ohms, the resistance 327 may have a resistance in a range of several ohms to several hundred ohms, a Q1B: Q12 transistor size ratio may be in a range of about 1: 4 to 1:10, and a Q11: Q12 transistor size ratio may be approximately in a range of 1: 4 to 1: 8. However, the scope of the present invention covers other transistor size ratios that are within the operating specifications of the carrier amplifier 120 ( 1 ) and the power amplifier 100 ( 1 ) are located. In the exemplary embodiment of the invention of 3E is the transistor Q12 355 configured as a common emitter npn bipolar transistor.
  • 4A is a block diagram of the peak amplifier 130 who in 1 is illustrated, according to an embodiment of the invention. In the embodiment of the invention of 4A is a top amplifier 130 a two-stage amplifier and includes an input matching unit 405 , a first-stage amplifier 410 , an intermediate stage adjusting unit 415 and a second stage amplifier / voltage controller 420 , Various embodiments of the second stage amplifier / voltage control unit 420 are below in connection with the 4B to 4D discussed.
  • In one embodiment of the invention, the input adaptation unit is 405 like the input adaptation unit 305 ( 3B ), wherein electrical characteristics of the inductor 306 ( 3B ), of the capacitor 307 ( 3B ) and the capacitor 308 ( 3B ) are selected such that an output impedance of the 3dB hybrid coupling element 110 ( 1 ) to an input impedance of the peak amplifier 130 at a connection 40 is measured, adjusted. Similarly, the interstage adapter is 415 as the interstage adapter 315 ( 3C ), wherein electrical characteristics of the capacitor 309 ( 3C ), of the inductor 311 ( 3C ) and the capacitor 312 ( 3C ) are selected such that an output impedance of the first stage amplifier 410 to an input impedance of the second stage amplifier / voltage control unit 420 is adjusted. Finally, the first stage amplifier 410 like the first stage amplifier 310 ( 3D ), wherein electrical characteristics of the first stage base biasing elements (ie, resistance 313 , Diode 314 , Diode 316 , Resistance 317 , Capacitor 318 and transistor Q1A 319 ), the first stage collector biasing elements (ie, transmission line 321 and capacitor 322 ) and the transistor Q11 345 ( 3D ) are selected such that the first stage amplifier 410 according to predefined specifications, such as For example, gain, normal mode, and lockout mode specifications.
  • 4B is a block diagram of the second stage amplifier / voltage controller 420 , in the 4A is illustrated, according to an embodiment of the invention. The second stage amplifier / voltage control unit 420 includes a second stage amplifier 445 and a voltage control unit 435 , The second stage amplifier 445 is like the second stage amplifier 320 ( 3E ). For example, the second stage amplifier includes 445 a CBU3 440 and a transistor Q22 450 , The CBU3 440 includes a resistor 423 , a diode 424 , a diode 426 , a resistance 427 , a capacitor 428 and a transistor Q2B 429 , collectively referred to as second stage peak base biasing elements. In operation, electrical characteristics of the second stage peak base bias elements along with DC bias voltages V3 and V4 are selected to be a base of the transistor Q22 450 for normal mode operation, based on operating characteristics of transistor Q22 450 and specifications of the power amplifier 100 ( 1 ). For example, the resistance 423 have a resistance in a range of several hundred ohms to several kilo-ohms, the resistance 427 may have a resistance value in a range of several ohms to several hundreds ohms, a Q2B: Q22 transistor size ratio may be approximately in a range of 1: 4 to 1:10, the DC bias voltage V3 may be in a range of 2, 8V to 3.0V, and the DC bias voltage V4 may be in a range of 3.2V to 4.2V. The second stage amplifier 445 receives a signal from the Zwi rule gradual adjustment unit 415 , amplifies the received signal based on the voltage control signal Vc generated by the voltage control unit 435 is received, and sends the amplified signal to the peak amplifier output terminal 60 ,
  • The voltage control unit 435 receives the voltage control signal Vc (normally in a range of 2.8V to 4.2V) and controls a bias DC current of the second stage amplifier 445 , In the embodiment of the invention of 4B includes the voltage control unit 435 a resistance 431 and a transistor Qc 432 , Usually, the resistance shows 431 a resistance value in a range of several hundred ohms to several kilo-ohms, and a Qc: Q2B transistor size ratio may be approximately in a range of 1: 1 to 1: 8. In operation, a base station (not shown) for receiving, transmitting, and processing RF signals transmits signals in response to RF signals received from the power amplifier 100 to a baseband modem chipset (not shown). The baseband modem chipset processes the signals and generates the voltage control signal Vc. The voltage control unit 435 then receives the voltage control signal Vc from the baseband modem chipset. In a further embodiment of the invention, the power amplifier comprises 100 RF processing circuitry (not shown) for processing the signals received by the baseband modem chipset. In this embodiment, the RF processing circuitry generates the voltage control signal Vc and sends the voltage control signal to the voltage control unit 435 , The RF processing circuitry and baseband modem chipset are known in the art and will not be described in detail.
  • Normally, the baseband modem chipset generates the voltage control signal Vc based on power levels of signals transmitted by the base station and received by the baseband modem chipset. If z. B. the baseband modem chipset upon receiving the signals from the base station determined that the power amplifier 100 in a low power output range, the baseband modem chipset sends a "high" voltage control signal Vc (ie, a high voltage state signal) to the voltage control unit 435 , However, if the baseband modem chipset determines that the power amplifier is in response to receiving the signals from the base station 100 in a high power output range, the baseband modem chipset sends a "low" voltage control signal Vc (ie, low voltage state signal) to the voltage control unit 435 , The scope of the present invention covers a voltage control signal Vc corresponding to any voltage state and any power output range.
  • In operation, when the baseband modem chipset receives a low voltage state control signal Vc to the peak amplifier 130 sends, indicating that the power amplifier 100 is effective in the high power output range, the voltage control unit 435 the low voltage state control signal Vc and provides a bias DC current of the second stage amplifier 445 the top amplifier 130 ( 4A ) via the received low-voltage state control signal Vc. The low-voltage state control signal Vc turns on the transistor Qc 432 increases base-emitter currents (not shown) of transistors Q2B 429 and Q22 450 and spans the top amplifier 130 as a class AB amplifier.
  • However, if the baseband modem chipset is a high voltage state control signal Vc to the peak amplifier 130 sends, indicating that the power amplifier 100 is effective in the low power output range, the voltage control unit receives 435 the high voltage state control signal Vc and provides a bias current of the second stage amplifier 445 the top amplifier 130 via the received high voltage state control signal Vc. The high voltage state control signal Vc turns on the transistor Qc 432 and conducts a base-emitter current of the transistor Q2B 429 to a collector-emitter current of the transistor Qc 432 around. Thus, base-emitter currents of the transistors Q2B 429 and Q22 450 off, and the top amplifier 130 is biased as either a class B or a class C amplifier, depending on a resulting bias state of transistor Q22 450 ,
  • 4C is a block diagram of the second stage amplifier / voltage controller 420 , in the 4A is illustrated, according to a further embodiment of the invention. The second stage amplifier / voltage control unit 420 includes a second stage amplifier 445 and a voltage control unit 455 , The second stage amplifier 445 is identical to the second stage amplifier 445 configured in 4B is illustrated. The voltage control unit 455 includes a resistor 456 , a resistance 457 , a transistor Qc1 458 and a transistor Qc2 459 , In addition, over a line 461 a DC bias voltage V3 to the voltage control unit 455 created. Usually, the resistance shows 456 a resistance in a range of several hundred ohms to several kilo-ohms, the resistance 457 has a resistance value in a range of several ohms to several hundreds Ohms, a Qc2: Qc1 transistor size ratio can be approximately in a range of 1: 1 to 1:10, a transistor size ratio Qc1: Q2B ( 4B ) may be in a range of about 1: 1 to 1: 8, a DC bias voltage V3 may be in a range of 2.8V to 3.0V, a DC bias voltage V4 may be in a range of 3.2 V to 4.2V, and a voltage control signal Vc may be in a range of 2.8V to 4.2V.
  • Input / output characteristics of the voltage control unit 455 are input / output characteristics of the voltage control unit 435 ( 4B ) opposite. That is, a low-voltage state control signal Vc applied to one terminal 61 is received, tension the top amplifier 130 ( 4A ) as either a class B or a class C amplifier, depending on a resulting bias state of transistor Q22 450 ( 4B ), and a high voltage state control signal Vc biases the peak amplifier 130 as a class AB amplifier.
  • 4D is a block diagram of the second stage amplifier / voltage controller 420 , in the 4A is illustrated, according to a further embodiment of the invention. The second stage amplifier / voltage control unit 420 includes a second stage amplifier 445 and a voltage control unit 460 , The second stage amplifier 445 is identical to the second stage amplifier 445 configured in 4B is illustrated. The voltage control unit 460 includes a resistor 462 , a transistor Qc3 463 and a transistor Qc4 464 , In addition, a DC bias voltage V4 via a line 466 to the voltage control unit 460 created. Usually, the resistance shows 462 a Qc3: Qc4 transistor size ratio may be approximately in the range of 1: 1 to 1:10, a transistor size ratio Qc4: Q2B ( 4B ) may be in a range of about 1: 1 to 1: 8, a DC bias voltage V3 may be in a range of 2.8V to 3.0V, a DC bias voltage V4 may be in a range of 3.2 V to 4.2V, and a voltage control signal Vc may be in a range of 2.8V to 4.2V.
  • Input / output characteristics of the voltage control unit 460 are input / output characteristics of the voltage control unit 435 ( 4B ) similar. That is, a low-voltage state control signal Vc biases the peak amplifier 130 as a class AB amplifier, and a high voltage state control signal Vc biases the peak amplifier 130 either as a class B or as a class C amplifier, depending on a resulting bias state of transistor Q22 450 ( 4B ).
  • 5 Fig. 10 is a block diagram of the output matching unit 140 , in the 1 is illustrated. By adjusting α and β (either individually or both) of the first λ / 4 transformer 143 or the second λ / 4 transformer 145 in the output adjustment unit 140 the characteristic impedances of the two λ / 4 transformer lines change. By optimizing α and β, the carrier amplifier can 120 achieve the maximum efficiency at an output power level that is lower than the highest output power level that the carrier amplifier 120 can generate.
  • The first λ / 4 transformer 143 and the second λ / 4 transformer 145 can work with λ / 4 transmission lines (T lines), as it is in 5 is shown, or with concentrated elements 143a . 143b . 143c . 143d , ..., 145a . 145b . 145c . 145d etc., as is in 6 is shown or implemented with similar elements. The output adjustment unit 140 can be used with many different combinations of capacitors and inductors ( 143a . 143b . 143c . 143d , ..., 145a . 145b . 145c . 145d etc.) to provide a specific output impedance at the output stage 70 and a specific impedance at the carrier amplifier output terminal 50 to generate an impedance at a peak amplifier output terminal 60 is inverted. Alternatively, the first λ / 4 transformer 143 and the second λ / 4 transformer 145 implemented either by the LTCC method or a multi-layer method. As another example, the first λ / 4 transformer 143 and the second λ / 4 transformer 145 be formed as a single integrated circuit.
  • 7 FIG. 12 is a graph illustrating efficiency characteristics as shown in FIG. B. be determined by the voltage control signal Vc, the to the peak amplifier 130 ( 1 ) is created. Mode 0 represents the region of amplifier operation in a low output power range (ie, from a minimum output power in dBm to point Q). Mode 1 represents the region of amplifier operation in a high output power range (ie, from point Q to point S and / or T). When a current is increasingly connected to the peak amplifier 130 is applied, an exemplary power amplifier according to an embodiment is first effective, as shown as curve D. Curves C and B represent the efficiency characteristics associated with the exemplary power amplifier when the Be the bias current increases beyond that associated with curve D. Curve A represents the efficiency characteristics of a general power amplifier.
  • When power in the top amplifier 130 begins to flow, the top amplifier starts 130 his operation. This changes the output impedance of the carrier amplifier 120 , reducing the efficiency of the power amplifier 100 is optimized to a certain constant level, as in 7 indicated by D. Accordingly, as it is in 7 is indicated by curve D, the power added efficiency (PAE) the maximum value from the point P (when the peak amplifier 130 begins to work) until either point S, which is the highest allowable output power that meets the given linearity conditions, or point T, which is the saturated output power, as is the same through the power amplifier 100 is produced. Thus, as illustrated, improved efficiency characteristics by an exemplary power amplifier according to an embodiment of the present invention are compared with the efficiency characteristics of a general power amplifier disclosed in US Pat 7 indicated by curve A reached. As described above, this is done by operating the peaking amplifier 130 with class B or C.
  • Through the graph of 8th however, nonlinearity characteristics are illustrated when the voltage control signal Vc is applied to the peak amplifier 130 is created. In this graph is the performance of the power amplifier 100 with respect to the Adjacent Channel Power Ratio (ACPR) when the output power is increased. In this case, it may be difficult to obtain values of overall nonlinearity characteristics (as in 8th indicated by curve D), and thus the nonlinear distortion of the power amplifier becomes 100 undesirable. Accordingly, the ACPR criterion R, which may be required in a specific system, may not be maintained up to the desired output power level associated with point S without violating the ACPR criteria. The ACPR criteria are known and those skilled in the art will recognize that R z. For example, it could represent -42 dBc for a CDMA cellular system or any other value.
  • In other words, it shows how it is in 7 and 8th as compared to general power amplifiers known in the related art and if the peak amplifier 130 at the power amplifier 100 operated with class B or C (ie if the power amplifier 100 operated in a typical Doherty mode), the power amplifier 100 improved efficiency characteristics over conventional power amplifiers, the z. B. be used in wireless communication applications. However, in terms of linearity, the power amplifier may have less predictable values when operating in the high output power range.
  • Therefore, an exemplary power amplifier according to an embodiment of the present invention satisfies high efficiency and linearity requirements in the low output power range, such as low power output. At point Q, where the ACPR criterion R required by the system is met. When operating in the low power mode 0, the criterion R is satisfied even if the voltage control signal Vc applied to the peak amplifier 130 is applied, so that is set the peak amplifier 130 operated with class B or C, where little DC current flows, and thus that of the power amplifier 100 is operated in Doherty mode. On the other hand, the power amplifier 100 In the high output power range during mode 1, excellent linearity is achieved by adjusting the voltage control signal Vc applied to the peak amplifier 130 is created, reach. This linearity can be achieved by increasing the bias DC current to the second stage amplifier 445 the top amplifier 130 can be realized by reducing the voltage control signal Vc to a point where the linearity specification (or linearity level) shown in FIG 8th is denoted by R. In this way, the top amplifier 130 as a class AB amplifier, for example, depending on the mode of operation. This leads to the efficiency and linearity curves B or C in the 7 and 8th ,
  • 9 Fig. 12 is a graph illustrating efficiency characteristics, the modes of the power amplifier 100 ( 1 ) according to an embodiment of the present invention. 10 Fig. 12 is a graph illustrating nonlinearity characteristics, the modes of the power amplifier 100 according to the present invention. When operating the exemplary power amplifier 100 be 10 considered. If the Leistungsver stronger 100 requires an output power level that reaches point Q where mode switching is required, the baseband modem chipset (not shown) sends a low voltage state control signal Vc to the peak amplifier 130 , allowing an increased bias current to the peak amplifier 130 can be created. In this way, the linearity of the power amplifier 100 according to an embodiment of the present invention with little reduction of the effect Grads improved. In one embodiment of the invention, point Q is in a range of 15-19 dBm, but the present invention covers other operating output powers in which the power amplifier 100 Modes changes. The efficiency and linearity curves in mode 1 are those of curves B (FIG. 7 to 8th ) similar. This prevents the criteria R from being violated.
  • 11 Fig. 12 is a graph illustrating gain characteristics, the modes of the power amplifier 100 ( 1 ) according to the present invention. In the present invention, the carrier amplifier 120 and the top amplifier 130 are operated to have the same linear gain characteristics. However, the overall system is not affected, even if the carrier amplifier 120 and the top amplifier 130 are implemented to operate with different linear gain characteristics because two modes can be clearly distinguished and independently operated according to a specific embodiment of the present invention.
  • 12 Fig. 10 is a block diagram showing the structure of a power amplifier in a mobile handset according to another embodiment of the present invention. The power amplifier according to another embodiment of the present invention is substantially the same in structure and operation as the power amplifier 100 who in 1 is shown. Therefore, the same reference numerals refer to the same parts in the power amplifiers according to FIG 1 and 12 , Thus, a detailed description of the power amplifier according to 12 For a professional not necessary, and it is therefore omitted.
  • As it is in 12 1, another exemplary power amplifier according to another embodiment includes a phase difference compensator 180 on top of the 3dB hybrid coupling element 110 from 1 replaced. The phase difference compensator 180 is with the entrance level 10 and the top amplifier 130 coupled so that the input signal to the peak amplifier 130 and to the carrier amplifier 120 is applied, wherein the phase difference compensator 180 has a phase difference of 90 ° (λ / 4).
  • As described above, there would be, as the input signal to the peak amplifier 130 is applied, and the input signal to the carrier amplifier 120 is applied by the operation of the phase difference compensator 180 have a phase difference of 90 ° (λ / 4) when the output powers from the carrier amplifier 120 and the top amplifier 130 in the output adapter 140 unite, no phase difference, and thus the optimum output power can be obtained.
  • If the phase difference compensator 180 instead of the 3dB hybrid coupling element 110 is used, the phase difference compensator 180 be implemented with a simple transmission line. Alternatively, the phase difference compensator 180 with lumped elements, because the simple transmission line can approximate inductance values. In this way, the power amplifier can without a complex 3dB hybrid coupling element 110 or a large-size transmission line outside the amplifier. In addition, since the phase difference compensator 180 can be integrated in a single chip and / or a single integrated circuit, the overall size of the power amplifier 100 be reduced, and the price of the power amplifier 100 can also be reduced.
  • In summary, if a low output power range (mode 0) is transmitted through the power amplifier 100 of the mobile handset is sufficient for proper functioning of a mobile handset / base station pair, as determined by power levels of signals received by the baseband modem chipset, the baseband modem chipset a voltage control signal Vc in a first state to the peak amplifier 130 , such that the power amplifier 100 is operated in Doherty mode (ie, so that the peak amplifier 130 operated as a class B or C amplifier). On the other hand, if a low output power range (mode 0) is transmitted through the power amplifier 100 of the mobile handset is not sufficient for the proper functioning of a mobile handset / base station pair, as determined by the power levels of signals received by the baseband modem chipset, and the base station requires that the power amplifier 100 in the high output power range (mode 1), the baseband modem chipset applies a voltage control signal Vc in a second state to the peak amplifier 130 in such a way that a bias DC current applied to the peak amplifier 130 is applied, and the ACPR is improved to a point R where the non-linearity specification of the power amplifier 100 is satisfied. In one embodiment of the invention, the voltage control signal Vc in the first state is a high voltage state signal, and the voltage control signal Vc in the second state is a low voltage state signal. In a further embodiment of the invention, the voltage control signal Vc in the first state is the Low voltage standstill signal, and the voltage control signal Vc in the second state is the high voltage state signal.
  • Even though several embodiments of the present invention for illustrative purposes will be appreciated by those skilled in the art that various modifications, additions and substitutions possible without departing from the scope and spirit of the invention, as disclosed in the accompanying claims.
  • As As described above, an exemplary power amplifier of the present invention has been described Invention in a mobile handset shown that improved Efficiency and improved linearity by controlling a bias DC current, which connects to a top amplifier of the Handset is applied, via a control signal Vc, the is received from a baseband modem chipset, as more relevant Power level of signals transmitted by the baseband modem chipset be received. For example, in the low output power range, a State of a control signal Vc applied to a peak amplifier is, so selected that the power amplifier of the present invention is operated in Doherty mode, and in the high output power range is the state of the control signal Vc going to the top repeater is created, selected, around the nonlinearity specification of the power amplifier to fulfill.
  • Various Features and aspects of the invention described above can be individually or used together. Furthermore, the invention in a any number of environments and applications that are used over the Described here, without departing from the nature and scope to deviate from the description in a broader sense. The description and the drawings are accordingly to be considered illustrative and not as limiting to be viewed as. The scope of the invention is not limited to the described embodiments limited and should only be attached by the claims be determined.
  • Summary
  • The The present invention relates to a bias control a power amplification circuit a mobile device for improving the efficiency and the linearity characteristics of the power amplifier. In one embodiment the power amplifier improves this Characteristics by receiving a voltage control signal, to bias a supplementary amplifier, so the power amplifier in a low output power range in a Doherty mode and in a high output power range in a non-Doherty mode is. In the non-Doherty mode, the supplemental amplifier will be via the received voltage control signal biased as a class AB amplifier, around the non-linear operating requirements of the power amplifier in to meet the high output power range. The power amplifier generates the voltage control signal based on power levels of signals, received from a remote base station.

Claims (35)

  1. A system for biasing a power amplifier, the having the following features: a carrier amplifier coupled to an input stage is to reinforce an input signal; and a top amplifier that is coupled to the input stage, for amplifying the input signal, wherein the top amplifier is configured to provide a voltage control signal for biasing the peak amplifier receive, wherein the voltage control signal at power levels based on signals sent by a remote base station become.
  2. The system according to claim 1, in which the carrier amplifier is further having the following features: a carrier stage amplifier, the coupled to the input stage; and a carrier second stage amplifier, the with the carrier stage amplifier and a carrier amplifier output port is coupled.
  3. The system according to claim 1, where the peak amplifier further comprises the following features: a top step amplifier that coupled to the input stage; and a top second stage amplifier that with the top step amplifier and a peak amplifier output terminal is coupled; and a voltage control unit associated with the Top second stage amplifier coupled, wherein the voltage control unit is configured, around the top amplifier over that to bias received voltage control signal.
  4. The system according to claim 3, in which the voltage control unit based the peak amplifier on a state of the received voltage control signal as a Class B or class C amplifier biased.
  5. The system according to claim 3, wherein the Voltage control unit biases the peak amplifier based on a state of the received voltage control signal as a class AB amplifier.
  6. The system according to claim 1, where the power amplifier is configured to the voltage control signal in a first state when the power levels of the signals passing through the remote base station are sent, indicate that the power amplifier in a low output power range is effective.
  7. The system according to claim 1, where the power amplifier is configured to the voltage control signal in a second State when the power levels of the signals passing through the remote base station will be sent, indicating that the power amplifier is in a high output power range is effective.
  8. The system according to claim 1, further comprising a 3dB hybrid coupling element which is configured to receive the input signal from the input stage to receive a first input signal to an input of the carrier amplifier and send a second input signal to an input of the peaking amplifier send, with the second input signal at about ninety degrees with respect to the first Input signal is out of phase.
  9. The system according to claim 8, further comprising an output matching unit that configures is to get an output signal from the peak amplifier and an output signal from the carrier amplifier to receive a substantially optimal power amplifier output signal to produce at an output stage.
  10. The system according to claim 9, in which the output matching unit further has the following features having: a first quarter-wavelength transformer, with a carrier amplifier output port is coupled; and a second quarter wavelength transformer, with a peak amplifier output terminal, an output of the first quarter wavelength transformer and the Output stage is coupled.
  11. A method for bias control of a power amplifier, the the following steps: Receiving signals through sending a remote base station; Generating a Voltage control signal based on power levels of the signals; and Biasing a peak amplifier of the power amplifier over the Voltage control signal.
  12. The method of claim 11, wherein the Further generating the step of generating the voltage control signal in a first state when the power levels of the signals show that the power amplifier in a low output power range is effective.
  13. The method of claim 12, wherein the Voltage control signal in the first state, the peak amplifier as biases a class B or Class C amplifier.
  14. The method of claim 11, wherein the Further generating the step of generating the voltage control signal in a second state when the power levels of the signals show that the power amplifier in a high output power range is effective.
  15. The method of claim 14, wherein the Voltage control signal in the second state the peak amplifier as a class AB amplifier biases.
  16. A system for controlling a power amplifier a mobile handset comprising: one Carrier amplifier, the a carrier input port and a carrier output port having; a peak amplifier having a peak input terminal, a peak output terminal and a control terminal for receiving a voltage control signal, wherein the peak amplifier konfigu tion is based on at least one characteristic of the power amplifier to vary on the voltage control signal; a phase shifter, the with the carrier input port and the peak input terminal is coupled to generate a Peak amplifier input signal, the re a carrier amplifier input signal phase delayed is; and an output matching unit connected to the carrier output terminal and the peak output terminal is coupled to send a Carrier output signal and a peak output power signal and for forming a power amplifier output power signal at a power amplifier output stage.
  17. The system of claim 16, further comprising a baseband modem chipset for receiving signals transmitted by a remote base station and for generating the voltage control signal in a first voltage state when power levels of the received signals indicate that the power amplifier is operating in a low power range , and for generating the voltage control signal in a second voltage state when the noise level of the received signals indicate that the power amplifier is operating in a high power range.
  18. The system according to claim 16, in which the phase shifter is a hybrid coupling element for distribution certain input powers to the carrier amplifier and the peak amplifier.
  19. The system according to claim 18, in which the hybrid coupling element is a 3dB hybrid coupling element which is implemented with lumped elements.
  20. The system according to claim 18, in which the hybrid coupling element is replaced by the low temperature cogenerated ceramic (LTCC) Procedure is implemented.
  21. The system according to claim 16, in which the phase shifter is a phase difference compensator.
  22. The system according to claim 21, wherein the phase difference compensator with a transmission line is implemented.
  23. The system according to claim 21, in which the phase difference compensator with concentrated elements is implemented.
  24. The system according to claim 16, in which the output adjusting unit with concentrated elements is implemented.
  25. The system according to claim 16, in which the output matching unit is replaced by a low temperature cogenerated ceramic (LTCC) method is implemented.
  26. The system according to claim 16, in which the at least one characteristic of the power amplifier is linearity.
  27. The system according to claim 17, where the top amplifier further comprises a voltage control unit configured to receive the voltage control signal and a bias current of the peak amplifier so that the power amplifier operates as a Doherty-type amplifier when the voltage control signal is in the first voltage state located, and the top amplifier as a class AB amplifier is operated when the voltage control signal in the second Voltage state is located.
  28. The system according to claim 16, in which the output matching unit further has the following features having: a first transformer having an input with the carrier output port and an output connected to the peak output terminal coupled; and a second transformer, the an input coupled to the output of the first transformer and an output connected to the power amplifier output stage is coupled.
  29. A method of operating a power amplifier a wireless transmitting device in at least two modes, wherein the power amplifier a carrier amplifier and a peak amplifier comprising, the method comprising the steps of: Produce a voltage control signal in a first voltage state when Power level of signals passing through a remote base station sent and received by the power amplifier, that the power amplifier in a low power range is effective; Generating a Voltage control signal in a second voltage state when the Power level of signals sent by the remote base station and through the power amplifier be received, indicate that the power amplifier in a high power range is effective; and Biasing the Top amplifier over that Voltage control signal.
  30. The method of claim 29, wherein the Further biasing the step of biasing the peak amplifier over the Voltage control signal in the first voltage state the power amplifier as a Doherty-type amplifier to operate.
  31. The method of claim 29, wherein the Further biasing the step of biasing the peak amplifier over the Having voltage control signal in the second voltage state, around a nonlinearity characteristic of the power amplifier to improve.
  32. The method of claim 29, wherein the Further biasing the step of biasing the peak amplifier over the Having voltage control signal in the second voltage state, around the top amplifier as a class AB amplifier to operate.
  33. A system for operating a power amplifier in a wireless transmission device in at least two modes, the power amplifier comprising a carrier amplifier and a peak amplifier, the system comprising: means for generating a voltage control signal in a first voltage state when power levels of signals passing through a remote base stations are transmitted and received by the power amplifier, indicating that the power amplifier is operating in a low power range; means for generating a voltage control signal in a second voltage state when the power levels of signals transmitted by the remote base station and received by the power amplifier indicate that the power amplifier is operating in a high power range; and means for biasing the peak amplifier via the voltage control signal.
  34. The system according to claim 33, wherein the means for biasing further comprises means for biasing the peak amplifier to the power amplifier as a Doherty-type amplifier to operate when the voltage control signal in the first Voltage state is located.
  35. The system according to claim 33, wherein the means for biasing further comprises means for biasing the peak amplifier has a nonlinearity characteristic of the power amplifier to improve when the voltage control signal in the second Voltage state is located.
DE112004001976T 2002-02-01 2004-10-19 High linearity Doherty communication amplifier with bias control Ceased DE112004001976T5 (en)

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US10/690,923 2003-10-21
US10/690,923 US7053706B2 (en) 2002-02-01 2003-10-21 High linearity doherty communication amplifier with bias control
PCT/US2004/034797 WO2005043747A2 (en) 2003-10-21 2004-10-19 High linearity doherty communication amplifier with bias control

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE102013220160A1 (en) * 2013-10-05 2015-04-09 Rwth Aachen Sequential broadband Doherty power amplifier with adjustable output line back-off

Families Citing this family (15)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP4927351B2 (en) * 2005-05-27 2012-05-09 ルネサスエレクトロニクス株式会社 Doherty amplifier
JP2008035487A (en) * 2006-06-19 2008-02-14 Renesas Technology Corp Rf power amplifier
JP5049562B2 (en) * 2006-11-17 2012-10-17 株式会社日立国際電気 Power amplifier
JP5217182B2 (en) * 2007-02-22 2013-06-19 富士通株式会社 High frequency amplifier circuit
JP2008258986A (en) * 2007-04-05 2008-10-23 Japan Radio Co Ltd High frequency amplifier circuit
JP5705122B2 (en) * 2009-10-23 2015-04-22 日本碍子株式会社 Doherty amplifier synthesizer
CN102064774B (en) * 2009-11-18 2013-11-06 中兴通讯股份有限公司 Implementation method of power amplifying circuit and power amplifying device
US20120013401A1 (en) * 2010-07-14 2012-01-19 Avago Technologies Wireless Ip (Singapore) Pte. Ltd. Power amplifier with selectable load impedance and method of amplifying a signal with selectable load impedance
CN102158177A (en) * 2011-04-29 2011-08-17 中兴通讯股份有限公司 Doherty power amplifier and power amplifying method
US8829998B2 (en) * 2012-10-23 2014-09-09 Airspan Networks Inc. Doherty power amplifier
US8981852B2 (en) * 2012-11-12 2015-03-17 Avago Technologies General Ip (Singapore) Pte. Ltd. Providing an integrated directional coupler in a power amplifier
CN102983824A (en) * 2012-12-25 2013-03-20 福建邮科通信技术有限公司 Self-adaptive predistortion power amplifier
US9030260B2 (en) * 2013-07-19 2015-05-12 Alcatel Lucent Dual-band high efficiency Doherty amplifiers with hybrid packaged power devices
EP2933918B1 (en) 2014-04-15 2017-11-22 Ampleon Netherlands B.V. Ultra wideband doherty amplifier
CN103945149A (en) * 2014-04-28 2014-07-23 上海东洲罗顿通信股份有限公司 Multiple-complex system of high efficiency digital television transmitter and implementing method thereof

Family Cites Families (13)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4225827A (en) * 1979-02-21 1980-09-30 Harris Corporation Stabilization circuit for transistor RF power amplifiers
US5420541A (en) * 1993-06-04 1995-05-30 Raytheon Company Microwave doherty amplifier
US5739723A (en) * 1995-12-04 1998-04-14 Motorola, Inc. Linear power amplifier using active bias for high efficiency and method thereof
US5757229A (en) * 1996-06-28 1998-05-26 Motorola, Inc. Bias circuit for a power amplifier
JPH118560A (en) * 1997-04-25 1999-01-12 Matsushita Electric Ind Co Ltd Circuit and method for transmission output control
US5880633A (en) * 1997-05-08 1999-03-09 Motorola, Inc. High efficiency power amplifier
US6130579A (en) * 1999-03-29 2000-10-10 Rf Micro Devices, Inc. Feed-forward biasing for RF amplifiers
US6262629B1 (en) * 1999-07-06 2001-07-17 Motorola, Inc. High efficiency power amplifier having reduced output matching networks for use in portable devices
JP4467756B2 (en) * 2000-10-13 2010-05-26 三菱電機株式会社 Doherty amplifier
US6864742B2 (en) * 2001-06-08 2005-03-08 Northrop Grumman Corporation Application of the doherty amplifier as a predistortion circuit for linearizing microwave amplifiers
US20020186079A1 (en) * 2001-06-08 2002-12-12 Kobayashi Kevin W. Asymmetrically biased high linearity balanced amplifier
US6469581B1 (en) * 2001-06-08 2002-10-22 Trw Inc. HEMT-HBT doherty microwave amplifier
KR100553252B1 (en) * 2002-02-01 2006-02-20 아바고테크놀로지스코리아 주식회사 Power Amplification Apparatus of Portable Terminal

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE102013220160A1 (en) * 2013-10-05 2015-04-09 Rwth Aachen Sequential broadband Doherty power amplifier with adjustable output line back-off

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GB2421862A (en) 2006-07-05
GB0605973D0 (en) 2006-05-03
GB2421862B (en) 2007-11-07
CN101164229A (en) 2008-04-16
WO2005043747A2 (en) 2005-05-12
WO2005043747A3 (en) 2005-09-09
JP4860476B2 (en) 2012-01-25
CN101164229B (en) 2010-12-08

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