DE10362081B4 - Detection and correction of a sampling error in a received signal that requires precise block or sampling synchronization, whereby a sampling error is determined from sampled first and second signal sections - Google Patents

Abstract

Device for detecting a sampling error in a sampled received signal has the following characteristics: a sampling arrangement (101) for sampling a received signal that has first and second signal sections. The first and second signal sections have a similar section that is sampled. The sampled signals resulting from sampling of the first and second signal sections have a difference and the sampling error is detected based on the sampled first and second signal sections. An independent claim is made for a method for detecting a sampling error in a sampled received signal.

Description

The The present invention relates to an apparatus and method for determining a sampling error in a sampled received signal and a means for correcting the sampling error.

With increasing transmission rates, achieved through the use of modern digital transmission technologies are, in particular to the receiving side signal processing Components have high requirements for signal processing accuracy posed. In a digital transmission system will be over a transmission channel transmitted Received signal and scanned after filtering to an analog, d. H. continuous-time received signal into a digital, d. H. to transfer discrete-time received signal, so a continuing one digital signal processing, such as digital filtering or digital equalization, possible becomes.

For a correct Signal processing in the receiver, For example, a demodulation of the received signal, but must first a sampling clock are derived from the received signal and it has to be one Transmitter / receiver synchronization carried out become. For example, a transmitted signal has a block structure this must be on Block structure in the receiver be recovered, d. H. for example, it has to be determined at which times the sampling of the received signal carried out must become, so that the receiving end recovered the frame boundaries used in a transmitter can be. For this it is necessary to have a corresponding synchronization of a Block or frame clock to perform (block or frame synchronization). But before that has to be done analog signal (received signal) by means of a scanning process in a discrete-time state are transferred. This is a determination of an optimal sampling time and thus a correct setting of the sample clock is necessary (synchronization the sampling clock). This is understood to mean a digital sampling clock synchronization Correction of the erroneously sampled received signal by one signal state achievable using a correct sampling clock to win (digital synchronization of the sampling clock).

A accurate block or sample clock synchronization is particularly useful OFDM signals of importance (OFDM = orthogonal frequency devision multiplexing, orthogonal frequency division multiplexing). This transmission method enjoys due to the positive transmission characteristics of a increasing interest. This transmission technology is well researched these days and is also considered part of various standards and specifications have been accepted. Examples of standards, in which OFDM are used, for example, the DVB-T standard (standard for digital terrestrial television), WLAN (WLAN = wireless local area network) or the IEEE 802.11a standard. Furthermore it is considered certain that OFDM in future transmission technologies, such as in the transmission systems The fourth generation (4G) is used since the OFDM transmission method provides substantial benefits.

In an OFDM transmission, serial data is divided on the transmitting side into groups of ld (M) bits and converted into complex-valued symbols a _i (n) from an M-fold alphabet of signal-space points. These complex valued symbols are modulated on N subcarriers using an Inverse Discrete Fourier Transform (IDFT). After a parallel-serial conversion, a complex-valued baseband representation of an OFDM symbol with an index i can be described as follows:

To avoid intersymbol interference (ISI), a copy of the last N _g samples is prefixed to each OFDM symbol. Due to the resulting cyclic extension of the OFDM symbol, this interval is termed "cyclic prefix" (CP) 10 this cyclic extension is illustrated by an example of an OFDM symbol, whereby a formation of the cyclic prefix is illustrated. In this case, the last N _g samples, which are a signal section 1101 copied to the beginning of the OFDM symbol, as indicated by the in 10 illustrated arrow is illustrated. The attached N _g samples form the cyclic prefix 1103 such that a resultant OFDM symbol has periodic signal portions. Due to the cyclic expansion, the total symbol duration T _OFDM = (N + N _g ) / f _S , where f _s denotes a sampling frequency, so that the resulting symbol N + N _{g has} samples with an index k. After a transmission-side digital-analogue conversion, a converted signal is converted into the high-frequency range (RF; RF = radio frequency). A spectrum of the signal to be transmitted is limited by means of an analog bandpass filter before transmission takes place over a physical channel. Depending on the characteristics of the transmission channel (RF channel) is a received OFDM signal influenced by a multipath propagation. Due to a maximum channel delay of τ _max , the OFDM symbol is temporally expanded and therefore disturbed by unwanted symbol overlays. If the channel impulse response (CIR) is not longer than the guard interval (CP), the interfering intersymbol interference can be avoided (τ _max ≦ T _g ). The signal transmitted over the transmission channel is then processed in a receiver front end. The subsequent demodulation in the receiver is carried out by performing a signal processing inverse to the transmitter in the receiver.

• – analoge Abtasttaktsynchronisation
• – hybride Abtasttaktsynchronisation
• – digitale Abtastsynchronisation.

The synchronization of the sampling clock can be divided into the following three groups according to the signal processing used in a receiver (analog or digital):

• - analog sampling clock synchronization
• Hybrid sample clock synchronization
• - digital sampling synchronization.

at An analog sampling clock synchronization will be analog only Components used. For a hybrid sample clock synchronization In addition to analog components, digital methods are also used. for example, control a scan. Because of the many The advantages of digital signal processing are the digital synchronization sampling clock as part of digital receiver implementations immense importance for future Applications. Digital synchronization of the sampling clock means but not a direct control of the scan, but rather a control of a correction of the erroneously sampled signal with the help of the obtained results of another procedure, the to detect a scan deviation.

In 11 a basic structure of a conventional digital synchronization of a sampling clock is shown. The structure has an analog signal processing block 1201 (analog signal processing, SV), wherein the analog signal processing block 1201 has an input and an output. With the output of the analog signal processing block 1201 is a scanning device 1203 connected. The scanning device 1203 has an output and another input connected to an output of a block 1205 connected is. The block 1205 In addition to a free-running oscillator may have other components, such as frequency divider and filter. With the output of the scanner 1203 is a correction block 1207 (Correction by digital signal processing) connected. The correction block 1207 also has an output for providing a corrected signal.

The following is the operation of the in 11 illustrated basic structure of a digital synchronization of the sampling clock explained.

• – Detektion der Abtasttaktabweichung (timing error detection, TED)
• – Korrektur der Abtasttaktabweichung (timing error correction, TEC)

In the analog signal processing block 1201 For example, a band limit of an analog received signal takes place. That of the analog signal processing block 1201 output received signal is from the scanning device 1203 sampled. The scan is a free-running scan because the scanner 1203 from the free-running oscillator 1205 which supplies a free-running clock is driven. The term "free-running" means that the oscillator 1205 provides the clock without being coupled, for example, with a transmitter-side clock or synchronized at the receiving end. Due to the free-running sampling, a sampling error occurs in a sampled receive signal at the output of the sampling device 1203 , When the received analog signal x (t) is sampled, with the nominal sampling period designated as T _S , the sampling device provides 1203 a sampled erroneous signal x [k (T _s + .DELTA.T _s )], where .DELTA.T _{S is} the sampling deviation that arises due to the free-running sampling. The erroneously sampled receive signal must therefore be digitally corrected, resulting in the post-correction block 1207 happens. In this case, the erroneously sampled received signal is corrected by a digital signal processing (SV) so that in an ideal case, the sampling error is eliminated, so that at the output of the Kor thkturblocks 1207 a corrected received signal x (kT _s ) is produced in which there is no sampling deviation ΔT _S. The deviations between transmit and receive clock occurring during the sampling of the analog signal with the aid of the free-running clock and the resulting signal modifications are therefore corrected by algorithms of digital signal processing. The digital correction of the effects of a faulty sampling clock can be divided into two function blocks according to the tasks to be realized:

• Detection of the sampling clock deviation (TED)
• Correction of the sampling timing deviation (TEC)

In 12 A synchronization principle with TED and TEC according to the prior art is shown, which can be used in OFDM systems, a typical scenario for an arrangement of the two above-mentioned functional blocks is made clear.

In the 12 The structure shown has a TEC 1301 on that with the output of the scanner 1203 connected is. With an output of the TEC 1301 is an FFT block 1303 connected (FFT = fast Fourier transform, fast Fourier transform). The FFT block 1303 has two outputs, at which one output a receive signal x (n) is output, which is a Fourier transform of a TEC output signal. The other exit is with a TED 1305 connected, the TED 1305 has an output with which the TEC 1301 is connected, so that a feedback structure is formed.

The following is the operation of the in 12 illustrated structure explained.

The sampled with a faulty sampling clock signal is using the TEC 1301 (eg FIR interpolation filter) corrected in the digital domain. Then it gets through the FFT block 1303 transformed into the frequency domain. The spectral evaluation of the signal allows a detection of the sampling deviation in the TED block 1305 , which can be accomplished for example by a phase evaluation of the frequency domain signal x (n). On the basis of the detected sample deviation, in the TED block 1305 generates a parameter (a correction signal) e (i-1) which provides a signal correction by the TEC 1301 controls for a following icon. In other words, in the TEC block 1301 due to the feedback loop does not correct the signal used to determine the sample deviation, but rather a subsequent signal. The sampled signal x [k (T _S + ΔT _S )] is further corrected before the sample deviation in the TED block 1305 is carried out. As already mentioned, the control parameter e (i-1) is generated from the frequency domain signal x (n). Due to the feedback technique used according to the prior art, therefore, the already mentioned symbol delay between TEC arises 1301 and TED 1305 which is indicated by a symbol index i-1.

• – Phasenberechnung über alle Träger nach der FFT 1303
• – Entfernung des Informations-tragenden Teils der Phase
• – Wichtung der Trägerphasen mit einem inversen Trägerindex
• – Mittelung der Phase über alle Träger für ein jeweiliges Empfangssymbol im Frequenzbereich
• – Ermittlung einer Differenz der Phasenmittelwerte aufeinanderfolgender Symbole zur Bestimmung der Abtastfrequenzabweichung

The phase evaluation from the payload data to obtain information about the sampling clock deviation usually involves the following steps:

• - Phase calculation over all carriers after the FFT 1303
• - Removal of the information-carrying part of the phase
• Weighting of carrier phases with an inverse carrier index
• - Averaging the phase over all carriers for a respective receive symbol in the frequency domain
• - Determining a difference of the phase mean values of successive symbols for determining the sampling frequency deviation

Next the described phase evaluation from user data can also used so-called pilot tones, which are transmitted to a clock synchronization with.

A disadvantage of the in 12 As discussed in the prior art, there is a delay due to the feedback loop. Thereby, the sampling clock deviation in an erroneously sampled reception signal is corrected depending on the sampling error determined on the basis of the previous erroneously sampled reception signal. Firstly, it is not possible to accurately compensate for a sampling error in the currently received erroneously sampled received signal, since the sampling error in successive erroneously sampled received signals due to the free running oscillator 1205 may be different, so that the discussed prior art approach always results in an inaccurate correction of the erroneously sampled received signal, even if the sampling deviation between successive symbols changes only slightly. For another, the in 12 The feedback structure shown does not react quickly to rapid changes in sampling deviations, so that accurate detection and correction of the sampling error is not possible.

Another disadvantage of the in 12 The approach described is that for detecting the sampling clock deviation frequency domain signals are used and the correction of the erroneously sampled received signal takes place in the time domain. This mixing of time domain and frequency domain signal processing increases hardware complexity because, for example, the in 12 illustrated TED block 1305 Frequency domain signals receives and beyond the TEC 1301 in which time domain signal processing takes place. In a hardware implementation, if time domain signal processing structures are spatially separated from the frequency domain signal processing structures, such as when the Fourier transform is implemented in an additional processor, additional hardware resources must be applied to the TED 1305 the frequency range provide rich signals available, which in addition to an increase in complexity also increases a control effort.

One Another disadvantage of the prior art approach discussed is therein to see that at a erroneous block synchronization in OFDM signals the following Fourier transform returns a faulty result, resulting in it justified is that one Fourier transform either on infinite episodes or on periodic episodes can be applied. In case of a faulty block synchronization lacks the property of periodicity, so that in the subsequent Fourier transform a Error arises. In this case, the sample deviation can not be accurately determined. The approach according to the prior art is therefore with insufficient Correction results associated, if no optimal block synchronization is present.

Another disadvantage of the in 12 The prior art discussed is that the sampling clock deviation can in principle not be determined with sufficient accuracy, since the phase is averaged over all carriers of an OFDM symbol, so that phase average values are not used to detect the sampling clock deviation, as already mentioned exact phase values of the symbols are used. Thus, the prior art approach can only provide mean information about the sample deviation but not its exact value. If, instead of the phase averaging phase of the aforementioned pilot made, the sampling deviation can still not be calculated accurately and quickly, because the pilot tones are not transmitted often enough due to the bandwidth required for this, a change in the sampling, always with a free-running sampling clock is connected, enough to track closely. On the other hand, transmitting the pilot tones continuously reduces the bandwidth efficiency.

One Another disadvantage of the fed back Approach according to state The technique is that a Correction of the sampling clock deviation take place only after a break-in period can since, for detecting the sampling clock deviation always consecutive Symbols needed and the sampling clock deviation due to the feedback based on possibly erroneously corrected symbols is detected, so that a stationary State where the sampling clock deviation is detected and corrected accurately can not be set immediately. In addition, a reaching of the stationary State not secured, which is given, for example, if the feedback loop becomes unstable. In this case, the sampling clock deviation becomes always erroneously determined, so that the incorrectly sampled symbols can not be corrected correctly, which in turn leads to an erroneous determination of the sample clock deviation leads (hang-up effect). Further leads the feedback Approach to a possible Error propagation.

Of the Article "OFDM timing Synchronization: Possibilities and Limits to the usage of the Cyclic Prefix for Maximum Likelihood Estimation "by D. Matic and others, published in the Proc. IEEE Vehicular Technology Conf., 1999, Vol. 2, pp. 668-672, describes possibilities for correcting the phase of a sampling clock with a non-data-assisted synchronization using a cyclic prefix for OFDM symbol timing in a fully digital implementation. A redundancy of the cyclic prefix, who as an absorber for The channel multipath propagation can also be used for an estimation of a Sample time offsets are used. This method is essential an autocorrelation, which has a fundamental impact on their final precision. As another aspect, the implementation involved pulse shaping taken into account in a time domain. This article shows a study of the original maximum-likelihood method and their possible ones Improvements.

The Publication "ML Estimation of Time and Frequency Offset in OFDM Systems "by J.J. Van de Beek and M. Sandell, released in the IEEE Transactions on Signal Processing, 1997, Vol. 45, no. 7, pp. 1800-1805, describes a combined maximum likelihood symbol time and carrier frequency offset estimator for OFDM systems. Redundant information contained in the cyclic prefix enables the estimate without additional Pilot tones. The frequency estimator can be used in a follow-up mode, and the time estimator can be used in an acquisition mode.

The article "Robust Frequency and Timing Synchronization for OFDM" by TM Schmidl and DC Cox, published in the IEEE Transactions on Communication, 1997, Vol. 12, pp. 1613-1621 describes a method for fast synchronization for an OFDM system using either a continuous transmission or a burst operation over a frequency-selective channel. The presence of a signal can be detected from the reception of only one training sequence consisting of two symbols. The beginning of a frame and the beginning of a symbol can be found, and carrier frequency offsets of many subchannels can be corrected. The algorithms work near an un The lower Cramer-Rao limit for the variance of the estimate in the frequency offset, and inherent averaging over many subcarriers, allows acquisition at very low signal-to-noise ratios.

Of the Article "Low overhead, Low-complexity [Burst] Synchronization for OFDM "by T. M. Schmidl and D. C. Cox, published in the Proceedings of the IEEE International Conference on Converging Technologies for Tomorrow's Applications, 1996, Vol. 3, pp. 1301-1306, describes another Method for fast synchronization for an OFDM system using either a continuous broadcast or a burst over one time variable channel. The named method acquires the signal and provides a channel estimate to receive only one training sequence, consisting of two symbols, out. The same algorithm can also track the signal. The Procedure offers the possibility a symbol timing, a carrier frequency offset and a Estimate sampling frequency offset.

The Object of the present invention is to provide a concept for efficient handling of a sampling error.

These The object is achieved by a device for determining a sampling error according to claim 1, by a method for determining a sampling error according to claim 6 or by a method according to claim 11 solved.

The inventive device for determining a sampling error in a sampled received signal comprises a sampling device for sampling a received signal comprising a first and a second signal portion, wherein the first and the second signal portion at least a same signal component wherein the sampled received signal is a sampled first and second signal portion, each of the Sampling of the first and second signal section is apparent, wherein the sampled signal portion of the first signal section and the sampled Signal portion of the second signal section have a difference, and a sampling error detecting means for detecting the sampling error based on the sampled first and second signal components.

According to one Another aspect of the present invention comprises a device according to the invention for correcting a received signal, a means for correcting the sampling error, which is designed to be based on one of a Abtastfehlererfassungseinrichtung detected sampling errors and a sampled received signal to provide a corrected received signal.

The The present invention further provides a method of determining a sampling error in a sampled received signal, wherein the method comprises the steps of: sampling a received signal, having a first and a second signal portion, wherein the first and the second signal section at least one same Have signal component, wherein the sampled received signal a sampled first and second signal sections, respectively from the sampling of the first and second signal sections, wherein the sampled signal component of the first signal section and the sampled Sig nalanteil the second signal section a Have difference, detecting the sampling error on the basis the sampled first and second signal sections.

According to one In another aspect, the present invention provides a method for correcting a scan error, the method for correcting comprising the steps of: determining the sampling error, correcting the sampling error based on the detected sampling error and the sampled Receive signal, providing a corrected received signal.

According to one In another aspect, the present invention provides a computer program with a program code for carrying out the method according to the invention, when the computer program runs on a computer.

The present invention is based on the finding that a sampling error can be detected by utilizing a structure of a received signal. If, for example, in a transmission system a signal is transmitted at the transmitting end which has periodic signal components, as is the case, for example, in an OFDM transmission system, this periodicity can be used to detect a sampling error which arises due to a sampling clock deviation in the receiver. If the transmission signal has periodic signal sections, for example a first and a second signal section, and if such a signal is transmitted via a real transmission channel, then one has Also signal genes also a first and a second signal section, which correspond respectively to the transmission-side periodic signal sections. If a temporal extent of a channel impulse response (length of the channel impulse response), which characterizes the transmission channel, less than a temporal extent of the respective same signal components, so have the first and the second received signal portion at least a same signal component, wherein the signal component of the first and second signal section may have a different noise process is additively superimposed. If such a received signal is sampled in a receiver, the result is a sampled received signal which analogously has a first and a second signal component, the respective signal components resulting from a sampling of the same signal components of the signal segments of the continuous-time received received signal. If the scanning operation is carried out with the aid of a free-running sampling clock, a sampling error results, which leads to a sampling clock deviation. Because of this sampling error (sampling clock deviation), the sampled signal component of the first signal section of the received signal and the sampled signal component of the second signal section of the received signal now differ from each other. If, for example, the sampling clock deviation is constant, the respective signal components of the first and second signal sections of the received signal are sampled at different times, whereby a difference arises between the sampled signal components. On the basis of this difference, the clock deviation and thus the sampling error can now be detected. For this example, the sampled first and second signal portion of the received signal can be used. Since the sampled signal component of the first signal section and the sampled signal component of the second signal section now differ due to the sampling error, the difference can be determined, for example, from a comparison of the sampled signal components and thus the sampling errors. The scanning error is thus detected according to the invention by utilizing the structure of the received signal, so that the sampling error in the sampled signal having the sampling error can be corrected immediately. Preferably, this is done by means of a forward processing structure which corrects the sampling error and thus the sampling error in the received sampled received signal on the basis of which the sampling error has been determined, thereby achieving sampling clock synchronization.

One Advantage of the present invention is the fact that the sampling error and thus the sample deviation, as already mentioned is, taking advantage of the signal structure of the sampled received signal detected will, so that a forward processing Structure yields no feedback structures having. This results in the detection of the sampling error no delay, as is the case with known prior art methods.

One Another advantage of the present invention is that the sampling error accurately detected is, since according to the invention for detection the sampling error of the difference caused by the sampling the sampled signal components is used. The signal components, which would be the same in an ideal sample clock synchronization can be considered Training symbols are understood in addition to a statistical also enable deterministic sampling error detection, which also leads to that the Sensing error quickly detected can be.

One Another advantage of the present invention is to be seen in that to Detecting the sampling error no Fourier transform is needed since the sampling error is detected on the basis of time domain signals can. The inventive concept leads therefore to a reduction of computational and hardware complexity, since none Mixing of time domain and frequency domain signals takes place as according to the above standing discussed prior art is the case.

One Another advantage of the present invention is that the sampling error based on an OFDM symbol is determined, if the inventive concept for a detection is used by a sampling error in an OFDM system. Consequently For example, the sample deviation may be determined based on a symbol and not from consecutive symbols, for example on the basis of the difference of the phase mean values of successive ones Symbols is determined. Due to this efficient use of the Structure of the OFDM symbols makes the sampling error much faster detected. There are also no pilot tones needed so that one continuous Abtastfehlererfassung is possible. This will further achieved that at a highly variable one Sampling deviation at which the sample deviation at successive Symbols is different, the sampling error basically determined can, as has already been mentioned, the sampling error is detected on the basis of only one symbol.

Another advantage of the present invention is that the sampling error in the received sampled signal on the basis of which the sampling error has been determined can be corrected. This ensures that the sampling error is corrected accurately and quickly, because between the Abtastfehlererfassung and the sample error correction does not give rise to a delay of one or more symbols, as is the case with the prior art discussed above.

One Another advantage of the present invention is to be seen in that at the correction of the sampling error is also a forward processing Structure can be used, eliminating the need for feedback loops become, what an increase stability of the receiving system leads. This further avoids the already mentioned hang-up effects. About that In addition, in the inventive concept no break-in time necessary to correct the received symbols, So that the Correction of the sampling error can be performed immediately and individually for each symbol can.

One Another advantage of the present invention is that due to the forward processing invention Structure in which both the sampling error detection and the Scanning error correction take place, which reduces numerical effort becomes. This is due to the fact that the Abtastfehlererfassung and the sample error correction, for example, to a common one Algorithm be summarized can, if these functional blocks be implemented in a digital signal processing processor. this leads to further to a reduction of the hardware and control effort, because both the error detection and the error correction on a can be implemented anywhere in the system, so that at a system design in this regard no restrictions arise.

One Another advantage of the present invention is to be seen in that both the sample error detection as well as the sample error correction in general can be used in systems where periodic signal sequences are transmitted. This is not only in the ones already mentioned OFDM systems of Case, but basically in systems, for example, on a re-encoding based on the same send sequence being sent multiple times.

One Another advantage of the present invention is that the inventive concept for example, in an OFDM system independent of an OFDM system underlying standard and used without its violation can be. This is also particularly advantageous if, for example An OFDM system consists of components that are different manufacturers produced because of the forward processing structure according to the invention no incompatibilities can arise.

preferred embodiments The present invention will be described below with reference to FIG the enclosed drawings closer explained. Show it:

1 a first embodiment of a device according to the invention for detecting and correcting a sampling error according to the present invention;

2 Representation of a signal having the same signal components;

3 another embodiment of a device for detecting and correcting a sampling error according to the present invention;

4 an embodiment of a signal structure with a cyclic repetition;

5 an example of an inventive indexing of identical signal components in a signal according to 4 ;

6 a further embodiment of an inventive device having an interpolation structure;

7 inventive representation of sampling time differences;

8th an embodiment illustrating an inventive determination of an average over a portion having the smallest standard deviation;

9a inventive determination of an artificially inserted sampling frequency deviation with a channel scenario;

9b inventive determination of an artificially inserted sampling frequency deviation with another channel scenario;

10 schematic representation of an OFDM symbol;

11 principal structure for digital sampling frequency synchronization;

12 conventional structure for correcting a sampling clock deviation.

In 1 A first embodiment of a device for determining a sampling error and for correcting the sampling error is shown.

In the 1 The device shown comprises a scanning device 101 , wherein the scanning device 101 has an input and an output. With the output of the scanner are a Abtastfehlererfassungseinrichtung 103 and a parallel thereto arranged means for correcting a sampling error 105 coupled. The means for correcting a sampling error 105 is further with the Abtastfehlererfassungseinrichtung 103 coupled and further comprises an output for providing a corrected received signal.

The following is the operation of the in 1 explained device explained.

The scanning device 101 receives via the input an analog, ie continuous-time received signal that may have previously been band limited by an analog filter device. The scanning device 101 is designed to convert the time-continuous received signal, hereinafter referred to as receive signal, into a time-discrete receive signal, hereinafter referred to as sampled receive signal, by sampling. The scanning device 101 preferably includes an in 1 for reasons of clarity not shown device for Erzeu conditions of a free-running clock, such as the in 11 illustrated free-running oscillator 1205 ,

In the following it is assumed that the continuous-time received signal, that of the scanning device 101 is supplied, having a first and a second signal portion, wherein the first and the second signal portion have at least a same signal component. This is the case, for example, when the received signal is an OFDM received signal and when the length of the channel impulse response characterizing the transmission channel over which the received signal has been transmitted is, for example, shorter than a guard interval. By sampling the received signal, the sampled received signal having a sampled first and a sampled second signal component, each resulting from the sampling of the first and second signal portion of the received signal. Due to the free-running sampling clock in the scanner 101 There is a sampling error, so that successive samples of the sampled received signal have a sampling clock deviation. Due to the sampling error, the sampled signal components of the first signal section and of the second signal section are now different. This difference depends, in addition to the sampling clock deviation, on the times at which the signal components are sampled, ie how far away the sampled signal components are from one another.

The erroneously sampled received signal becomes the sampling error detecting means 103 and the facility 105 for correcting a sampling error. The sampling error detecting means is configured to detect the sampling error on the basis of the sampled first and second signal portions of the sampled received signal. Here, the two sampled signal components of the sampled received signal are preferably used in order to determine the difference between the two sampled signal components in order to detect the sampling error. On the basis of the detected sampling error, the sampling error detecting means preferably generates a parameter indicative of the sampling error in the sampled received signal and that of the device 105 for correcting a sampling error. On the basis of this parameter, the device can 105 correct the sampling error in the sampled received signal so that the erroneously sampled received signal is converted into a corrected received signal that is present at the output of the device 105 is issued. At the parameter that the sampling error detection means 103 can, for example, be filter coefficients that the device 105 is used to such a filtering of the erroneously sampled received signal, so that after filtering a corrected received signal is formed. In the case of the scanning error detection device 103 In addition, the parameters supplied may be equalizer coefficients of an equalizer, so that the device 105 if executed as an equalizer, corrects the erroneously sampled received signal by equalization, since erroneous signal sampling can also be interpreted as signal distortion.

In the in 1 illustrated embodiment are the Abtastfehlererfassungseinrichtung 103 as well as the decor 105 arranged parallel to illustrate that the Abtastfehlererfassungseinrichtung 103 and the device 105 operate on the same sampled received signal, despite the fact that upon detection of a sampling error inevitably creates a slight time delay necessary to carry out the necessary operations. This time delay can be compensated for example by the fact that the device 105 for example, has a delay element. However, it is also conceivable that the device 105 after the sampling error detection means 103 is arranged, so that there is a forward processing structure in which the of the Abtastfehlererfassungseinrichtung 103 incurred time delay is not problematic.

To clarify the inventive concept of sampling error detection is in 2 a received signal shown, which can be used according to the invention for detecting the sampling error. The received signal has a first signal section 201 and a second signal section 203 on. The signal section 201 comprises a signal component 205 also in the second signal section 203 is included. In case of a faulty sampling of the in 2 shown received signal, the two signal components now differ from each other. According to the invention, however, a detection of the scanning error is not an exact knowledge of the two signal components 205 the received signal necessary. Will that be in 2 shown receive signal "scanned from left to right", ie, the first signal section 201 before the second signal section 203 receive a knowledge of the sample clock deviation in the signal component 201 not necessary since, assuming a constant sampling clock deviation within a received signal, the sampling error starting from the second signal portion 203 relative to the first signal section 201 can be determined. One possible approach to determining the sampling error is described in connection with FIG 3 illustrated embodiment discussed in detail.

3 shows a first embodiment of an inventive device for determining a sampling error in a sampled received signal.

In the 3 The device shown comprises a scanning device 301 for example, like the ones in 1 shown scanning device 101 is constructed. With an output of the scanner 301 is a sampling error detection device (TED) 303 coupled. The scanning error detecting device 303 has a first exit 305 and a second exit 307 on. After the Abtastfehlerer detection device 303 is a facility 309 arranged to correct a sampling error (TEC). The device 309 for correcting a sampling error is with both the first output 305 as well as with the second exit 307 the sampling error detection device 303 coupled. After the setup 309 is optional a facility 311 arranged to perform a Fourier transform associated with an output of the device 309 is coupled to correct a sampling error.

In the 3 illustrated inventive structure has the device 311 for determining the Fourier transformation only to illustrate the inventive concept when used in an OFDM system in which a Fourier transform is performed, as well as the principal differences between the inventive concept and the in 12 illustrated state of the art. It should be noted, however, that the FFT block 311 is not required for a detection of a scanning error according to the invention.

The following will focus on how the in 3 illustrated structure of the invention.

After a sample of the received signal by the scanner 301 For example, the incoming samples x [k (T _S + ΔT _S )] are used to detect the sampling error, which is characterized by sampling clock deviation ΔT _S and which may also be described as a sampling frequency deviation Δf _S , where f _S denotes the sampling frequency , The scanning error detecting device 303 detects the sampling error and generates a control parameter e (i) which is used in the correction of the same samples, on the basis of which the device 303 detects the sampling error. The correction of the samples may, for example, be performed by interpolation when the device 309 for correcting as an FIR interpolation filter. In this case, the parameter e (i) describes, for example, the coefficients of the interpolation filter as a function of the detected sampling error deviation (sampling error).

By virtue of the fact that, according to the invention, a forward-processing structure is now possible, the frequency domain signal x (n) is not influenced by a sampling frequency deviation in an OFDM signal, for example 309 already provides a corrected received signal x (kT _S ). This advantage is included For example, in the in 12 In the prior art, the prior art structure is not shown, since the sampling error is determined only after the Fourier transform, so that, as already mentioned, the frequency domain signals are affected by sampling error deviations.

As related to the in 3 has been discussed, according to the invention is a discrete-time signal, which at the output of the scanning device 301 arises, used to detect a sampling clock deviation. According to the present invention, the sampling clock deviation (sampling error) is performed on the basis of cyclic repeat sequences (first and second signal portions) included in the time-discrete signal. These repetition sequences result from a sampling of the cyclic repetition contained in the continuous-time signal, wherein a content of the sequence to be processed and the corresponding cyclic repetition (same signal component) should be the same, but of course can consist of random values. The prerequisite for utilizing such a signal structure for detecting the sampling error according to the invention is merely a one-time repetition of a sequence, which is the case, for example, with an OFDM signal.

In 4 an example of a signal structure with a cyclic repetition is shown. These are a plurality of OFDM symbols, where, as already described, each OFDM symbol is preceded by N _g samples as a guard interval, which are a copy of the last N _g samples, as represented by the in 1 illustrated arrows is illustrated. Assuming an ideal transmission channel and an ideal sample, the first and second signal portions appearing in a respective OFDM symbol are identical.

In the in 4 In the illustrated embodiment, the first and second portions of the three consecutive OFDM symbols are each sequence 1 (SEQ ID NO. 1) OR SEQUENCE 2 (SEQ ID NO. 2), SEQUENCE 3 (SEQUENCE 3) OR SEQUENCE 4 (SEQUENCE 4) AND SEQUENCE 5 (SEQ ID NO: 5) or SEQ ID NO: 6 (SEQ ID NO: 6). However, with a sampling frequency deviation, ie, Δf _S is not equal to 0, the respective first and second signal portions of the respective sampled OFDM symbol are different from each other, so that they have no identical signal portions after the sampling. If the distances between the respective corresponding samples of the originally identical sequences 1 and 2 are known, then the sampling error, as will be explained below, can be detected.

In practice, this distance is determined by appropriate specifications in the standard underlying the transmission method. Assuming for this distance a value of N sampling periods and denoting the deviation from the nominal sampling period with ΔT _S , we obtain the in 5 illustrated indexing of the samples of both sequences. For a better differentiation of the samples of both sequences, the samples of the respective second sequence (repetition of the first sequence) are marked with an underscore in the further considerations. It can be clearly seen that a time difference of the originally identical samples in the respective sequences is formed by a time difference N (T _S + ΔT _S ). The object of the erfindungsge MAESSEN method for detecting the Abtastfehlers (the detection method) is now to determine .DELTA.T _S. Available for this are the known in a receiver function values of the two sequences. These are no longer identical due to the deviation of the sample clock from the nominal sample clock. This property (difference) may, as already mentioned, be exploited to determine the sampling clock deviation.

The first step is to change the in 5 modeled effect. This means that an approximate formula is determined with the aid of which the illustrated effect can be reproduced. For example, one can use an interpolation polynomial for this, which is a uniquely determined polynomial (n-1) -th degree. This interpolation polynomial can be determined, for example, using the Lagrangian interpolation formula:

In this formula, the values x ₁ ... x _n and y ₁ ... y _{n represent} the values of the first sampled sequence 1 (first signal segment). The values x ₁ ... X _{n characterize} the time interval of the samples of the first sequence, whereas the values y ₁ ... Y _n characterize a possibly complex-valued value of the samples of the sequence 1. The values x and y, on the other hand, represent analog The values of the N sampling periods later sampled sequence 2 are taken into account here. For the values x and y, the or y is the temporal sampling clock deviation (x-axis) or the consequent change in the function values (y-axis). In other words, preferably, any sample of the sequence 2 (second signal portion) that is known is estimated from a plurality of samples taking into account its timing, thereby modeling the effect of sampling clock deviation.

The above-described interpolation formula is preferably used in the sampling error detection device, as described, for example, in US Pat 3 illustrated embodiment, so that the sampling error is a comparison between a sample, the sampled second signal portion (sequence 2, y) and an estimate of the sample based on a plurality of samples, the sampled first signal portion (sequence 1, y ₁ ... y _n ) detected.

6 shows an embodiment of an apparatus having an interpolation structure with which the effect of sampling clock deviation can be modeled. In difference to the in 3 illustrated embodiment includes in 6 illustrated embodiment, an interpolation structure 601 that after the scanning device 301 is arranged. The interpolation structure 601 also has a control input 603 as well as an exit 604 on.

The following is the operation of the in 6 illustrated structure explained.

6 illustrates the effect of the scan deviation on the sequence 1 and sequence 2 (first and second signal section). Since the respective sampling pairs of the first and second signal sections are N samples apart, an amplitude of the samples differs due to a deviation NΔT _S from each other. This effect can be compared to an effect of interpolation, for example, with a slowly varying or a constant sampling frequency deviation Δf _S. For this purpose, the sample taken by the scanner 301 is assumed to be ideal, and the effect of the sample deviation, on the other hand, in the interpolation structure 601 considered. In this case, NΔT _{S determines} a temporal position of the sample x. Thus, assuming that sampling frequency deviations can be corrected by parameter-controlled interpolation, it must be possible to determine the interpolation parameter by calculating interpolation between input and output samples. The above equation (Lagrangian formula) exemplifies such interpolation. If one selects, for example, as the degree of the polynomial-based function n-1, then n samples of the sequence 1 can be used to calculate an interpolated value at any time between the n selected samples. The value y is characterized only by the temporal position x _{i of} the respective sample. Of course, this operation can be inverted. For each value y, it must therefore be possible to calculate n-1 sampling times. On the basis of using a sample from sequence 2 (second signal section), information about a time difference between the first and the second signal section can therefore be obtained. As it is in 6 is illustrated, this difference is determined only by NΔT _S.

To simplify the following calculations, NΔT _{S is} abbreviated to μ. In order to calculate ΔT _S , an ideal sampling clock (eg T _S = 1) may preferably be assumed for the sequence sampled first. Looking now at the Lagrangian formula above, it can be seen that the denominator can be simply calculated, since there is a multiplication of integer values in the denominator. In the counter, however, results in a difference, which will be discussed below.

The subtraction in the counter illustrates this 7 , which illustrates an example of a representation of sample time differences. In order to simplify the further considerations, a cubic polynomial of the third order is assumed (n = 4). This approach also provides a good compromise between accuracy and numeric effort.

In the upper diagram in 7 a scan sequence of the first signal section (sequence 1) is shown. In this case, a standardized time interval, which is illustrated by arrows in each case, is equal to one between successive sampling values. In the lower diagram in 7 on the other hand, a sample value y is shown, which is shifted in time due to the sample clock deviation. In this case, the time interval between the sampling time x and, for example, the sampling time x ₂ in a superposition of the two sequences μ _N , as it is by the arrows are shown. Analogously, the sampling clock deviations between x and x are ₁ μ _N + 1 and so forth. By inserting the in 7 The following formulation can be obtained for the formula shown above:

After being multiplied out, the potentiated values for μ _N ^{3 ... 1 are} excluded. Subsequent conversion of y to the right side of the equation yields the following formula:

Since the values for y _{1 ... n} and y are known (sequence 1 and sequence 2), three different values for μ _N can now be determined with the aid of, for example, a zero-point calculation.

To keep a numerical effort low, the zero computation can be performed, for example, using an efficient eigenvalue computation algorithm. From the three obtained zeros now the relevant zero point must be determined, which indicates the sampling clock deviation. First, the values for μ _N are divided by N. This results in three different values for ΔT _S due to the previously performed substitution (μ _N = NΔT _S ). If the values y _{1 ... n} and y are complex-valued, then the zero values obtained on the basis of the necessarily complex-valued operations are normally complex-valued. An exception must be the relevant value for ΔT _s . Due to the fact that the sought value for ΔT _S must be real, the relevant value for ΔT _S can be determined by a simple search criterion, which determines, for example, the smallest imaginary part. Due to extensive simulations it could be determined, for example, that the imaginary part in this case is almost 0 (10 ^-4 and smaller). The value thus obtained for the sampling clock deviation ΔT _S (= 1 / Δf _S ) can be used subsequent to a control of the correction of the sampled values by means of interpolation methods. Such interpolation methods have been described by FM Gardner in "Interpolation in Digital Modems - Part: Fundamentals," IEEE Transactions on Communications, vol. 41, no. 3, pp. 501-508, March 1993, and by L. Erup et al "Interpolation in Digital Modems Part 2: Implementation and Performance", IEEE Transactions on Communications, vol. 41, no. 6, pp. 998-1008, published June 1993.

According to the invention erroneous sampled received signal, as it has already been mentioned based on the in the erroneously sampled received signal resulting scanning error corrected since according to the invention for this purpose a forward processing Structure is used. For example, assigns the errored one sampled receive signal a sequence of samples, and is the Abtastfehlererfassungseinrichtung invention trained to the sampling error from the sequence of samples the means for correcting a sampling error provides from the erroneous sequence of samples a corrected Sequence of samples. This correction is preferred for each successive received sample sequence performed, resulting in an improvement of the correction accuracy and thus too lowering the bit error probability.

• – durch eine Mehrwegausbreitung kommt es zu frequenzselektiven Effekten
• – diese Effekte sind wiederum zeitvariant

In the above embodiments, it has been assumed that the sequences examined (signal sections) are influenced by the sampling error (sampling clock deviation). In reality, however, the properties of the transmission channel (channel properties) must be taken into account. For a burst-like structured radio transmission, for example, two effects are to be considered:

• - multipath propagation leads to frequency-selective effects
• - These effects are again time-varying

The inventive device for determining the Abtastfehlers and the inventive device for correcting the Abtastfehlers can preferably be used when the Kanalecholaufzeiten are not all that long, that is, for example, are shorter than the Län ge of the first or the second signal portion, which, for example an in-home transmission (indoor transmission) met is. Late received signal components have no effect at all or only insignificantly due to the low transmission power. A delay power density spectrum (delay spread) has an exponentially decaying course in this case. The mean pulse spread Δτ of an indoor scenario, for example, can be up to 250 ns.

The inventive concept can in principle for every sample of the repeat sequence. Has this sequence for example, 16 values, so could 16 independent ones Values for calculate the sampling clock deviation. However, the channel influence must be considered become. If the channel influence as showing a convolution of the transmission data with the channel impulse response, so the samples of the repeat sequence are also sampled outside of the repeated area differently affected.

In order to ensure the least possible interference, the following operation should preferably be maintained for an OFDM transmission for the channel used: τ Max ≤ T GI

The means that one maximum pulse propagation through the channel must be less than that time duration of the repeat sequence (eg cyclic prefix in OFDM). However, this is a general requirement for different applications (eg, OFDM technique to avoid intersymbol interference), This operation is not a serious limitation. However, if the channel impulse response is longer than that used in OFDM Guard interval, so can efficient algorithms are used to perform a pre-equalization in the time domain in the receiver to shorten the channel impulse response. For example, in the paper by H. Schmidt and K.D. Kammeyer, "Impulse truncation for wireless OFDM ", Proceedings of 5-th International OFDM Workshop 2000, Hamburg, September 2000, a simple method proposed to the channel impulse response on the basis of the criterion of the minimum mean square To cut the error. This predistortion can, for example, by a FIR filter can be realized.

Under Channel influence can the determination of those suitable for calculating the clock deviation Values can be achieved in two ways. Is the length of the Channel impulse response known, so you can after the block synchronization Immediately determine which values of the cyclical repetition for the calculation the clock deviation are suitable.

is the length The channel impulse response is not known, but it must be determined in advance become.

According to the invention Length of Channel impulse response from a plurality of successively detected sampling errors certainly. It can which successively recorded Sampling error, for example, from a device for receiving a plurality of sampling errors are received, so that the plurality the received sampling error preferably in a device for Dividing the sampling errors into successive groups of sampling errors is split. For the respective group of sampling errors then becomes a standard deviation certainly. By determining the group of sampling errors that the Having the lowest standard deviation, the length of the Channel impulse response can be determined.

The sampling clock deviation ΔT _S is preferably determined for all values of the cyclical repetition. Subsequently, the N _g values of the clock deviation are divided into different intervals. The interval that has the lowest standard deviation is subsequently processed further. Averaging over this interval yields a reliable sample clock deviation value, which further allows for the suppression of noise effects. As has been described above, the length of the channel impulse response can be determined according to the invention by this procedure.

In 8th The method described above is exemplified by a cyclic repetition of a sequence of 16 samples, as described, for example, in the following: IEEE P802, 11a, Part 11: Wireless LAN Medium Access Control (MAC) and Physical Layer (PHY) specifications: High-speed physical layer in the 5 GHz band ", July 1999 specified clarifies.

Since in-house channels (indoor channels) are very slowly variable, the determination of the averaging-relevant values (interval length) for ΔT _S can only be carried out at very long time intervals, eg. For example, all 100 OFDM symbols. With the aid of oversampling, the process for determining the relevant interval for averaging ΔT _s can be made considerably more precise. A further improvement (reduction of noise effects) of the result for _S .DELTA.T can be preferably achieved by a further averaging successive values for .DELTA.T _S what z. B. durchge by an average of .DELTA.T _S over several OFDM symbols can be led. This averaging can be performed because the variations in the sampling clock deviation due to the oscillator characteristics of the oscillator used in the sampling clock, such as the sampling clock. B. temperature behavior, are slowly changing.

The functionality of the invention according to the apparatus for determining the sampling error or sampling frequency deviation indicative of the sampling error has been verified in several simulations. In this case, a typical WLAN scenario based on OFDM was chosen (N = 64, N _g = 16, T _OFDM = 4 μs, T _g = 0.8 μs). In order to increase the quality of the signal processing, an oversampling by a factor of 2 and 4 was used in the simulations.

In 9a and 9b exemplary simulation results are shown, in each case the Abtastfrequenzabweichung Δ _f [ppm] are plotted on a symbol index. As already mentioned, the quality of detection is most affected by channel influences. In order to construct this influence, two different channel models were used which correspond to a realistic in-house scenario and which are described in the following document: ETSI EP BRAN, J. Medbo, P. Schramm, "Channel Models for HIPERLAN / 2 in Different Indoor Scenarios". , ETSI BRAN 3ERI085B, March 1998.

Additionally, Gaussian distributed white noise (40 dB) was superimposed (AWGN = average white gauss noise). The first channel corresponds to a typical office environment. The upper limit for the delay power density spread (delay spread) is 390 ns. Use of a guard interval with a length of T _g = 800 ns results in uncritical offset detection, which results in 9a is shown. During the simulation, the sampling frequency deviation was changed every 10 OFDM symbols (100, 50, 200, 100, 150 ppm). In the 9b The illustrated curve describes a more critical case. The channel impulse response model used corresponds to a typical large free space environment where there is no NLOS (NLOS). These operations also apply to an office environment with a large delay power density spectrum. The upper limit for the delay power density spectrum (delay spread) is 730 ns. Using a guard interval with a length of T _g = 800 ns thus results in a critical offset detection. For this reason, in the simulations, the channel impulse response was shortened according to the method developed in the specification by Schmidt, KD Kammeyer, "Impulse truncation for wireless OFDM", Proceedings of 5-th International OFDM Workshop 2000, Hamburg, September 2000. The remaining tolerances are small compared with a total detection deviation and can therefore be neglected.

According to the present Invention can Abtasttaktabweichungen preferred purely numerically detected or be detected. The present invention also further allows a pure digital sample clock synchronization. Possible sampling clock deviations are calculated using cyclic repeat sequences. This is a kind of "inverse Interpolation " obtained parameters (sampling clock deviation) is used to control the Correction of the erroneously sampled signal.

One Main feature of the detection method according to the invention exists in the forward processing Structure. Through this forward processing Structure become the stability characteristics the synchronization significantly improved. Furthermore, no Warm-up time necessary and the already mentioned hang-up effects can be avoided become. A significant advantage is due to the low numerical Effort given. Many intermediate results for calculating the clock deviation can during the actual correction of the signal by interpolation be reused. It is possible, that at the calculation according to the invention the clock deviation also assumed an interpolation polynomial can be. Chooses one for one both functional blocks, Detection and correction (TED and TEC) use the same interpolation approach, one can reduce the numerical complexity of the detection process to a few Reduce operations.

in the following is the influence of a non-ideal sampling discussed.

To take advantage of an almost fully digitized receiver, sampling is performed using a free running oscillator. The resulting differences between the sampling frequencies in a transmitter and in a receiver (f _trans + f _rec ) are essentially dependent on the characteristics of the oscillator, as described in C. Muschallik, "Influence of RF Oscillators on OFDM Signal". described IEEE Transactions on Consumer Electronics, Vol 41, No. 4, S 592-603, July 1995. In the following, a slowly varying Abtastfrequenzabweichung is assumed It can be described by a relative value Δ _f....:

Due to two individual sampling frequencies, the above value for Δ _{f must be} doubled, which assumes the worst case. The effect of sampling frequency deviation can only be analyzed in the frequency domain. To simplify the signal calculations, a single carrier signal is assumed, which is described by an exponential function. The signal description in the frequency domain can be obtained by using an N-point discrete Fourier transform (DFT). Now, the resulting spectrum X _i can be calculated for each subcarrier index n, since it is shaped only by a single frequency f _c .

berechnet werden. Um den Einfluß der Abtastfrequenzabweichung Δ_f zu betrachten, muß die die Abtastfrequenzabweichung Δ_f beschreibende Gleichung modifiziert werden. Unter Verwendung des Ausdrucks f_trans = Nf_C/C und der Substitution T_S = T_rec, folgt die Gleichung

Using a geometric series expansion, the summation can be avoided in the above equation. After a separation of phase and magnitude, the spectrum can pass through

be calculated. In order to consider the influence of the Abtastfrequenzabweichung Δ _f must the Abtastfrequenzabweichung Δ _f equation describing be modified. Using the expression _ftrans = Nf _C / C and the substitution T _S = T _rec , the equation follows

Around To derive the real OFDM properties, the equation above must be used completed by a summation which describes the subcarrier influences.

Based on the above equation, all the effects of the non-ideal sample can be calculated. By varying the parameter n, c, and Δ _f can have two different effects can be distinguished in the OFDM signal. The influence of Δ _f to the phase-containing terms indicate a phase rotation that is proportional to the Subträgerindex and the symbol index. For this reason, only the phase of the usable signal components (n = c) has to be considered below. Therefore, for a subcarrier n follows

In addition, the phase is rotated for successive symbols. This rotation can be through φ i (n) = 2πni (1 + Δ f ) be expressed. The second effect can be described by the absolute value of X _i (n).

Two different cases have to be distinguished. For n unlike c, they will interfere considered the signal components. The occurring inter-channel interference (ICI) increases proportionally with the subcarrier index. In addition to this effect, useful spectral components (n = c) are attenuated.

By a processing of non-ideally sampled signals by the Discrete Fourier Transform (DFT) makes the spectrum non-ideal assigned. After a transformation of the orthogonal subcarrier, will the discrete spectrum between the maxima and the equidistant distributed zero crossings placed. To obtain a significant signal to noise ratio (SNR; SNR = signal to noise ratio) Deterioration due to loss to avoid subcarrier orthogonality Preferably, efficient sampling frequency synchronization should be performed.

Depending on the circumstances, the inventive method for determining a sampling error or to correct a sampling error in Hardware or be implemented in software. The implementation can be on a digital storage medium, especially a floppy disk or CD with electronically readable control signals, the so can interact with a programmable computer system that the corresponding procedure is carried out.

Claims (11)

Device for determining a sampling error in a sampled received signal, comprising: a sampling device ( 101 ; 301 ) for sampling a receive signal having a first and a second signal portion, the first and second signal portions having at least a same signal component, the sampled receive signal having a sampled first and second signal portion, each of the first and second samples Signal portion emerges, wherein the sampled signal portion of the first signal portion and the sampled signal portion of the second signal portion have a difference; a scanning error detection device ( 103 ; 303 ) for detecting the sampling error on the basis of the sampled first and second signal sections, the sampling error detection device ( 103 ; 303 ) is adapted to estimate the sampling error by Lagrange interpolation. Apparatus for correcting a received signal, comprising a device for determining a sampling error in a sampled received signal according to claim 1, comprising: a device ( 105 ; 309 ) for correcting the sampling error, which is arranged to provide a corrected received signal based on the detected sampling error and the sampled received signal. Apparatus according to claim 2, in which the sampling error detection means ( 103 ; 303 ) is designed to allow the device ( 105 ; 309 ) for correcting the sampling error to provide a parameter indicative of the sampling error in the sampled received signal, the device ( 105 ; 309 ) for correcting a sampling error to correct the sampled received signal based on the parameter. Apparatus according to claim 3, wherein the erroneous sampled received signal comprises a sequence of samples, the sampling error detection means ( 103 ; 303 ) is arranged to determine the sampling error from the sequence of samples, and wherein the device ( 105 ; 309 ) for correcting a sampling error to provide a corrected sequence of samples from the errored sequence of samples. Device according to a the claims 1 to 4, further comprising: a facility for Receiving a plurality of successively detected sampling errors; a Device for splitting the plurality of received sampling errors in successive group of sampling errors; An institution for determining a standard deviation for the respective group of sampling errors; means for determining a length of a Channel impulse response by determining the group of sampling errors, which has the lowest standard deviation. A method for determining a sampling error in a sampled received signal, comprising the steps of: sampling a received signal having first and second signal portions, the first and second signal portions having at least a same signal component, the sampled one of the sampled signals Receive signal has a sampled first and second signal portion, each resulting from the sampling of the first and second signal portion, wherein the sampled signal portion of the first signal portion and the sampled signal portion of the second signal portion have a difference; Detecting the sampling error based on the sampled first and second signal portions, wherein the sampling error is estimated by a Lagrange interpolation. Method according to claim 6, for detecting the sampling error, a step of comparing between a sample, the sampled second signal portion and an estimate of the sample based on a plurality of samples that the sampled first signal portion has. Method for correcting a received signal, with the following steps: Determining the sampling error according to claim 6 or 7; Correct the scan error based on the detected scan error and the sampled received signal; Deliver a corrected Reception signal. Method according to claim 8, with the following steps: Generating a parameter that indicates the sampling error in the sampled received signal; Correct the sampled received signal based on the parameter. Method according to one the claims 8-9, with the following steps: Receiving a plurality of successively recorded sampling errors; Splitting the plurality of received sampling errors in successive groups of sampling errors; Determine a standard deviation for the respective group of sampling errors; Determining a length of a Channel impulse response by determining the group of sampling errors, which has the lowest standard deviation. Method according to one the claims 6 to 10, which is realized as a computer program.