DE102022205482A1 - Low inductance power converter with a compact commutation cell - Google Patents

Abstract

Power electronic converters are specified. The power electronic converters can be used in energy and propulsion systems for aircraft. In one aspect, a power electronic converter includes a converter commutation cell that includes a power circuit and a gate driver circuit, the power circuit including at least one power semiconductor switching element and at least one capacitor. Each power semiconductor switching element is contained in a power semiconductor prepackage, each prepackage having one or more power semiconductor switching elements embedded in a solid insulating material, each power semiconductor switching element having at least three terminals including a gate terminal. The gate driver circuit is electrically connected to the gate terminal of each of the at least one power semiconductor switching elements and configured to provide switching signals thereto. A peak nominal power of the power electronic converter is greater than 25 kW and a value of a converter parameter α is less than or equal to 5 pHm3, where α is defined as the product of a smallest cuboid volume that encloses the commutation cell and a parasitic inductance of the power circuit of the commutation cell is.

Description

TECHNICAL PART

This disclosure relates to power electronic converters, particularly, but not exclusively, to power electronic converters with power semiconductor prepackages for use in aircraft power and propulsion systems.

BACKGROUND

In aerospace, aircraft and their energy and propulsion systems are increasingly being designed to be electrical. Some of the proposed platforms are purely electric, i.e. i.e. they cover their entire energy and drive requirements completely via batteries or fuel cells. Other proposed platforms are hybrid-electric, and still others are more “electric” in that they derive their propulsion power predominantly or entirely from onboard engines (e.g. gas turbine engines) but have a greater number of electrically powered aircraft and engine systems, subsystems and accessories.

The electrical power systems of these platforms necessarily contain power electronic converters. AC-DC converters (inverters and rectifiers) convert between alternating and direct current, e.g. B. to supply AC power to an electrical machine configured as a motor from a DC power source (e.g. a battery or a DC channel), or to supply DC power from an electrical machine configured as a generator to a DC channel or to provide a rechargeable battery. DC-DC converters can e.g. B. can be used to regulate the DC voltage supplied by a battery to a DC power channel. The electrical power supply systems may also include other power electronic devices, e.g. B. Protection devices such as solid-state power regulators (SSPC) and solid-state circuit breakers, some of which may be integrated into the converters themselves.

So-called power modules or power electronic modules are the dominant state of the art for power electronic converters. In a converter with a power module, the components of the converter circuit - which include power semiconductor devices such as transistors and diodes as well as smoothing intermediate circuit or input capacitors - are mounted on a carrier substrate and electrically connected to one another, usually via wire connections. For example, power module-based converters are widely used in the automotive industry and are used in existing aerospace systems, where they generally provide acceptable performance with an average operating efficiency of around 95%.

The performance of converters based on the power module topology is acceptable for most applications, especially considering their relatively low cost and high availability. However, for aerospace applications it would be advantageous to improve the efficiency and power-to-weight ratio of power electronic converters. Compared to ground-based applications, including automotive applications, aerospace applications are very weight dependent. Particularly in all-electric aircraft applications, even a relatively small increase in converter efficiency could significantly improve aircraft performance and operational range.

Improvements in the performance of converters based on existing power module topologies are expected to be limited to what can be achieved through advances in the underlying semiconductor technologies. This is at least partly due to the inherently high parasitic inductance of the commutation cell of a power module, which is caused in large part by the electrical connections between the components of the commutation cell. Parasitic inductance in the commutation cell is associated with transistor switching losses and turn-off voltage overshoot, which not only limits efficiency and generates heat that must be dissipated, but also limits other performance characteristics such as transistor switching frequency.

SUMMARY

A power electronic converter is disclosed. The power electronic converter comprises a converter commutation cell with a power circuit and a gate driver circuit, wherein the Power circuit comprises at least one power semiconductor switching element and at least one capacitor. Each power semiconductor switching element has at least three terminals including a gate terminal. The gate driver circuit is electrically connected to the gate terminal of each of the at least one power semiconductor switching elements and configured to provide switching signals to this terminal. The power electronic converter may be a power electronic AC-DC converter (ie, an inverter or a rectifier) or a power electronic DC-DC converter.

An electrical power system is also disclosed that includes an electrical machine and an AC-DC power electronic converter. The electrical machine includes one or more windings. The power electronic AC-DC converter includes a commutation cell with a power circuit and a gate driver circuit. The power circuit includes a plurality of power semiconductor switching elements and at least one capacitor. An AC side connection of the power circuit is connected to one or more windings of the electrical machine to supply power to or receive power from the electrical machine. The electric machine can be a motor or a generator, and the AC-DC power electronic converter can be an inverter or a rectifier. The electrical machine can be a motor-generator and the AC-DC converter can be a bidirectional converter that can be operated as an inverter or rectifier.

The following metric prefixes are used here to abbreviate numerical values: Table 1 prefix abbreviation Value exa E ×10 ¹⁸ peta P ×10 ¹⁵ tera T ×10 ¹² giga G ×10 ⁹ mega M ×10 ⁶ kilo k ×10 ³ centi c ×10 ^-2 Thousand m ×10 ^-3 Micro µ ×10 ^-6 nano n ×10 ^-9 pico p ×10 ^-12 femto f ×10 ^-15 atto a ×10 ^-18

Each of the following features can be applied individually or in combination with each other and with the power electronic converter and the electrical power system described above.

Each power semiconductor switching element can be a transistor. Each transistor can be a MOSFET that has at least a gate terminal, a source terminal and a drain terminal. The MOSFETs can be silicon carbide (SiC) MOSFETs. In other examples, the MOSFETs are gallium nitride MOSFETs (GaN).

Each power semiconductor switching element can be included in a power semiconductor prepackage. Each power semiconductor prepackage contains one or more power semiconductor switching elements embedded in a solid insulating material. Each power semiconductor prepackage can contain exactly one power semiconductor switching element.

The power electronic converter can comprise one or more logical power semiconductor switches, each of which contains one or more power semiconductor switching elements connected in parallel. Each power semiconductor logic switch may include one or more power semiconductor prepackages sen, each power semiconductor prepackage containing at least one (and optionally exactly one) power semiconductor switching element. The number of power semiconductor switching elements per logical switch can be greater than or equal to three. The number can range from three to twelve.

The peak rated power of the power electronic converter can be more than 10 kW (corresponding to kVA) and more than 25 kW. The peak rated power may be greater than or equal to 40 kW or greater than or equal to 50 kW. The peak nominal power can be less than or equal to 500 kW. The peak power rating may be less than or equal to 400 kW or less than or equal to 300 kW. The peak power rating can range from 50 kW to 300 kW.

The maximum efficiency of the power electronic converter can be greater than 97%. The maximum efficiency can be greater than 97.5%, greater than 98%, greater than 98.5% or even greater than 99%.

A value of a converter parameter α can be less than or equal to 5 pHm ³ . α is a product of the smallest cuboid volume that encloses the commutation cell and a parasitic inductance of the power circuit of the commutation cell. α can be greater than or equal to 0.3 pHm ³ . α can be less than or equal to 4 pHm ³ . α can satisfy 0.4 pHm ³ ≤ α ≤ 3.5 pHm ³ . α can satisfy 0.5 pHm ³ ≤ α ≤ 2.5 pHm ³ .

The value of α divided by the peak power rating of the power electronic converter may be greater than or equal to 1.5 aHm ³ /W. The value of α divided by the peak value of the rated output power of the power electronic converter can be less than or equal to 100 aHm ³ /W. The value of α divided by the peak nominal output power of the power electronics can be in the range of 2.5 aHm /W ³ to 50 aHm ³ /W.

The product of the parasitic inductance of the commutation cell power circuit and the peak power rating can be in the range of 0.05 mHW to 1.5 mHW. The product can be in the range of 0.1 mHW to 1.2 mHW or in the range of 0.2 mHW to 1.0 mHW.

A value of a parameter β may be greater than or equal to 0.3 PV/s ² . β is a product of a maximum switching frequency of the switching signals and a maximum rate of change of a source-drain voltage of the plurality of power semiconductor switching elements during operation. The value of β can be less than or equal to 10 PV/s ² . The value of β can be greater than or equal to 0.5 PV/s ² . The value of β can satisfy 0.8 PV/s ² ≤ β ≤ 5 PV/s ² , and can satisfy 1.0 PV/s ² ≤ β ≤ 2.5 PV/s ² .

The value of a converter parameter γ can be less than or equal to 150 fFs/W. γ is a total nominal capacity of the at least one capacitor of the power circuit divided by a product of the peak nominal power of the power electronic converter and a maximum switching frequency of the switching signals. The value of γ can be greater than or equal to 1.0 fFs/W. The value of γ can be less than or equal to 100 fFs/W, less than or equal to 75 fFs/W, or less than or equal to 50 fFs/W. The value of γ can be 2.0 fFs/W ≤ γ ≤ 50 fFs/W and 4.0 fFs/W ≤ γ ≤ 25 fFs/W.

The value of a converter parameter δ can be greater than or equal to 0.5 PV/FH. δ is a maximum blocking voltage of the one or more power semiconductor switching elements of the power circuit divided by a product of a parasitic inductance of the power circuit of the commutation cell and a total nominal capacity of the at least one capacitor of the power circuit. The value of δ can be less than or equal to 40 PV/FH. The value of δ can be greater than or equal to 1.5 PV/FH. The value of δ can range from 2.5 PV/FH to 25 PV/FH. It can range from 4.0 PV/FH to 15 PV/FH.

The value of a converter parameter ε can be greater than or equal to 10 ²⁶ V /s ⁴ . ε is equal to: $$\mathrm{\epsilon }=\frac{{\text{f}}_{\text{Max}}\times {\left|\frac{\text{dv}}{\text{German}}\right|}_{\text{Max}}}{\text{L}\times \text{C}}$$

In this equation, f _max is a maximum switching frequency of the switching signals, |dv/dt| _max is a maximum rate of change of a source-drain voltage of the one or more power semiconductor switching elements during operation, L is a parasitic inductance of the power circuit of the commutation cell and C is a total nominal capacity of the at least one capacitor of the power circuit. The value of ε can be less than or equal to 10 ²⁹ V/s ⁴ . The value of ε can be greater than or equal to 5×10 ²⁶ V/s ⁴ . The value of ε can be in the range of 10 ²⁷ V/s ⁴ to 5×10 ²⁸ V/s ⁴ and in the range of 1.5×10 ²⁷ V/s ⁴ to 3×10 ²⁸ V/s ⁴ .

The parasitic inductance of the power circuit of the commutation cell may be less than or equal to 16 nH, less than or equal to 10 nH, less than or equal to 8 nH, less than or equal to 6 nH, less than or equal to 4 nH, less than or equal to 3 nH or even less than or equal to 2 nH. The parasitic inductance can range from 2 nH to 8 nH.

The total rated capacity of the commutation cell power circuit divided by the peak rated power may be less than or equal to 5 nF/W, and may be less than or equal to 3 nF/W. The total rated capacity of the commutation cell power circuit divided by the peak rated power can range from 0.1 nF/W to 2.5 nF/W.

The smallest cuboid volume that encloses the commutation cell can be less than or equal to 1,000 cm ³ . The smallest cuboid volume can be less than or equal to 900 cm ³ , less than or equal to 800 cm ³ , less than or equal to 700 cm ³ or less than or equal to 600 cm ³ . The smallest cuboid volume can be in the range of 100 cm ³ to 800 cm ³ , 100 cm ³ to 700 cm ³ , or 100 cm ³ to 600 cm ³ , 150 cm ³ to 600 cm ³ , or 200 cm ³ to 450 cm ³ .

The maximum rate of change of the source-drain voltage of the one or more power semiconductor switching elements during operation can be greater than or equal to 10 kV/µs. It can be greater than or equal to 15 kV/µs. It can be greater than or equal to 20 kV/µs. It can be in the range 10 kV/µs to 60 kV/µs, in the range 15 kV/µs to 50 kV/µs, in the range 20 kV/µs to 50 kV/µs, in the range 25 kV/µs to 50 kV/µs or in the range 30 kV/µs to 50 kV/µs. It can be in the range of 30 kV/µs to 40 kV/µs.

The maximum switching frequency of the switching signals (f _max ) can be greater than or equal to 10 kHz. It can be greater than or equal to 20 kHz, greater than or equal to 30 kHz, greater than or equal to 40 kHz, or even greater than or equal to 50 kHz. The maximum switching frequency can be less than 100 kHz. It can be in the range of 30 kHz to 70 kHz.

The reverse voltage (i.e., the “source-drain reverse voltage,” sometimes called the “nominal voltage”) of each power semiconductor switching element may be greater than 600 V, greater than 700 V, or greater than 800 V. The reverse voltage can be less than 1,800 V or less than 1,700 V. The reverse voltage can range from 800V to 1,600V, 900V to 1,500V, or 1,000V to 1,400V.

The power electronic converter can further comprise a multilayer planar carrier substrate. The multilayer planar support substrate may define an xy direction parallel to a planar surface of the substrate and a z direction perpendicular to the xy direction. The carrier substrate may include a plurality of electrically conductive layers extending in the xy direction and at least one electrical connection extending in the z direction. The support substrate may have an outer conductive layer on one or both of its opposing planar surfaces.

The multilayer planar substrate may be a rigid printed circuit board (PCB). It can be a flexible circuit board. It can be a ceramic carrier substrate. The multilayer flat substrate can be a structural component of the transducer.

Each power semiconductor prepackage may further comprise at least one electrical connection that extends in the z direction from at least one connection of each of the one or more power semiconductor switching elements through the solid insulating material to an electrical connection side of the power semiconductor prepackage. At least one of the connections of each of the one or more power semiconductor switching elements of the prepackage can be connected to at least one of the conductive layers of the multilayer planar carrier substrate on the electrical connection side of the power semiconductor prepackage.

The electrical connection side of the power semiconductor prepackage may be spaced in the z-direction from the multilayer planar carrier substrate so that a gap (referred to herein as a prepackage gap) is defined between the multilayer planar carrier substrate and the electrical connection side of the prepackage. The size of the prepackage gap in the z direction can be less than or equal to 300µm. The size of the prepackage gap can be less than or equal to 250µm. The size of the prepackage gap can be less than or equal to 200µm. The size of the prepackage gap can be less than or equal to 150µm. The size of the prepackage gap can be greater than or equal to 10µm, greater than or equal to 20µm, greater than or equal to 50µm, or greater than or equal to 80µm. The prepackage gap can be in the range of 20µm to 250µm or in the range of 50µm to 150µm.

A value of a transducer parameter θ may be less than or equal to 300 pm ² /V. θ is a quantity in the z-direction of the prepackage gap divided by a maximum electric field strength in the prepackage gap. Accordingly, the converter parameter θ can be expressed as follows: $$\mathrm{\theta }=\frac{{\text{G}}_1}{{\text{<figure-callout id="1" label="E" filenames="DE102022205482A1\_0053.png" state="{{state}}">E</figure-callout>}}_1}$$

In this equation, G ₁ is the size of the prepackage gap in the z-direction and E ₁ is the maximum electric field strength in the prepackage gap. θ can be greater than or equal to 0.1 pm ² /V. θ can be less than or equal to 250 pm ² /V. θ can be in the range of 2.0 pm ² /V to 20 pm ² /V, or in the range of 3.0 pm ² /V to 10 pm ² /V. θ can be in the range of 0.5 pm ² /V to 100 pm ² /V, or in the range of 2.0 pm ² /V to 50 pm ² /V.

The maximum electric field strength in the prepackage gap may be greater than or equal to 1 kV/mm, greater than or equal to 3 kV/mm, greater than or equal to 5 kV/mm, or greater than or equal to 10 kV/mm. The maximum electric field strength in the prepackage gap may be less than or equal to 50 kV/mm, less than or equal to 40 kV/mm, or less than or equal to 25 kV/mm. The maximum electric field strength in the prepackage gap can be in the range of 5 kV/mm to 40 kV/mm or in the range of 10 kV/mm to 25 kV/mm. The maximum electric field strength in the prepackage gap can be in the range of 3 kV/mm to 25 kV/mm.

The power electronic converter can further comprise a heat sink for dissipating heat from power semiconductor prepackages. The heat sink can be spaced from the multilayer planar carrier substrate in the z direction, so that a gap (referred to here as the heat sink gap) is created between the multilayer planar carrier substrate and the heat sink. The size of the heat sink gap in the z direction between the multilayer planar support substrate and the heat sink may be less than or equal to 10 mm, less than or equal to 5 mm, less than or equal to 3 mm, less than or equal to 2.5 mm, less than or equal to 1 mm or less than or equal to 0.3 mm. The size of the heat sink gap can be greater than or equal to 0.1 mm, greater than or equal to 0.5 mm, or greater than or equal to 1 mm. The size of the heat sink gap can be in the range of 0.5mm to 3mm, in the range of 0.5mm to 2.5mm, in the range of 0.5mm to 2mm, in the range of 1mm to 2mm or in the range of 1.3 mm to 1.7 mm.

A value of a transducer parameter φ can be less than or equal to 20 nm ² /V. φ is a quantity in the z-direction of the heat sink gap divided by the maximum electric field strength in the heat sink gap. Accordingly, the converter parameter φ can be expressed as follows: $$\mathrm{\phi }=\frac{{\text{G}}_2}{{\text{<figure-callout id="2" label="E" filenames="DE102022205482A1\_0053.png" state="{{state}}">E</figure-callout>}}_2}$$

In this equation, G ₂ is the size of the heat sink gap in the z direction and E ₂ is the maximum electric field strength in the heat sink gap. φ can be less than or equal to 15 nm ² /V. φ can be greater than or equal to 0.01 nm ² /V. φ can be in the range of 0.25 nm ² /V to 2.5 nm ² /V, or in the range of 0.5 nm ² /V to 1.5 nm ² /V. φ can be in the range of 0.02 nm ² /V to 10 nm ² /V, or in the range of 0.05 nm ² /V to 5 nm ² /V.

The maximum electric field strength in the heat sink gap can be greater than or equal to 0.1 kV/mm, greater than or equal to 0.2 kV/mm, or greater than or equal to 1 kV/mm. The maximum electric field strength in the heat sink gap can be less than or equal to 20 kV/mm, less than or equal to 15 kV/mm, less than or equal to 10 kV/mm or less than or equal to 5 kV/mm. The maximum electric field strength in the heat sink gap can be in the range of 0.2 kV/mm to 10 kV/mm or in the range of 1 kV/mm to 2 kV/mm or in the range of 1.3 kV/mm to 1.7 kV/mm.

Experts know that the maximum electric field strength in the prepackage gap and/or in the heat sink gap can be determined using a mathematical and/or computational simulation method (e.g. finite element analysis).

The maximum electric field strengths E ₁ , E ₂ are maximum homogeneous field strengths in the respective columns, ie the maximum field strengths can exclude highly localized maxima such as singularities that occur at or near sharp edges. The maximum electric field strengths can be determined at a point or in an area in the respective gap, this point or area being in the xy direction from a singularity, for example an edge or boundary, of the multilayer planar carrier substrate, the power semiconductor prepackage and / or the heat sink is removed. The distance in the xy direction between the named point or area and the singularity, e.g. B. the edge of the multilayer planar carrier substrate, the power semiconductor prepackage and / or the heat sink, can be determined by a number of mesh cells of the simulation method, e.g. B. three mesh cells, be defined. By determining the maximum electric field strength in this point or area, the maximum electric field strength is determined in a region of the electric field that is relatively homogeneous.

In general, the maximum electric field strengths can be determined in a region between two opposing and substantially parallel surface regions.

If a prepackage gap is present, at least a portion of the prepackage gap can be filled with an electrically insulating material. The electrically insulating material, which may be a resin (e.g., a dielectric resin or a polymer resin) or other suitable insulating material, may have a plurality of cavities in the area (volume) it occupies .

A transducer parameter σ can be greater than or equal to 10/mm. σ is defined as an insulation fill factor of the electrically insulating material divided by a maximum cavity size of the plurality of cavities. Accordingly, σ can be expressed as follows: $$\sigma =\frac{F}{R_{max}}$$

In this equation, F is the insulation fill factor and R _max is the maximum cavity size of the plurality of cavities. The insulation fill factor is defined as the cumulative volume of the plurality of cavities (“Void Volume”) subtracted from a volume of electrically insulating material divided by the volume of electrically insulating material. This can be expressed as follows: $$F=\frac{v_{IM}-v_v}{v_{IM}}$$

In this equation, V _IM is the volume of the electrically insulating material and V _V is the cumulative volume of the plurality of cavities. Therefore, the converter parameter σ can also be expressed as follows: $$\sigma =\frac{v_{IM}-v_v}{v_v\times R_{max}}$$

σ can be greater than or equal to 15/mm, or greater than or equal to 18/mm, or greater than or equal to 50/mm, or greater than or equal to 80/mm, or greater than or equal to 80/mm. σ may be less than or equal to 1000/mm, or less than or equal to 800/mm, or less than or equal to 500/mm, or less than or equal to 200/mm, or less than or equal to 150/mm. σ can be in the range of 30/mm to 200/mm or in the range of 50/mm to 150/mm.

The insulation fill factor may be greater than or equal to 90%, greater than or equal to 95%, greater than or equal to 99%, or greater than or equal to 99.99%.

The electrically insulating material can be referred to as the underfill material of the semiconductor prepackage. However, in contrast to some applications of so-called underfill material in the field of electronics, which offer improved mechanical properties (e.g. stiffness), the underfill material according to the present disclosure additionally or alternatively has an electrically insulating function in order to achieve the high to withstand electric fields that arise in the power electronic converters of the present disclosure.

The maximum pore size of the plurality of pores may be less than or equal to 100 μm, less than or equal to 50 μm, less than or equal to 20 μm, less than or equal to 10 μm, less than or equal to 5 μm, or less than or equal to 1 μm .

Those skilled in the art will understand that the term "voids" refers to the inclusion of "foreign" material other than the electrically insulating material in the area occupied by the electrically insulating material. The cavities can be in solid (e.g. particulate), liquid or gaseous form. Examples of voids may include solder material, flux residue, air, washer fluid, and the like. Voids can be unintentionally introduced during the manufacturing process.

The maximum cavity size can be defined as the diameter of an equivalent spherical body that has the same volume as the largest cavity. The largest cavity and/or the maximum cavity size can be determined by methods known to those skilled in the art, e.g. B. by statistical methods to determine the cavity size of a representative number of cavities. This representative number of cavities is smaller than the total number of plurality of cavities in the electrically insulating material.

A converter parameter τ can be less than or equal to 10,000 V. τ is the product of the dielectric strength of the electrically insulating material and the maximum cavity size of the plurality of cavities. Accordingly, τ can be expressed as: $$\tau =D\times R_{max}$$

In this equation, D is the dielectric strength of the electrically insulating material and R _max is the maximum pore size of the plurality of pores. τ can be less than or equal to 1,000 V, less than or equal to 500 V, or less than or equal to 250 V. τ can be greater than or equal to 1V, greater than or equal to 10V, or greater than or equal to 100V. τ can be greater than or equal to 150 V. τ can be in the range of 100V to 300V or in the range of 150V to 250V.

The dielectric strength of the electrically insulating material, D, may be greater than or equal to 1 kV/mm, greater than or equal to 10 kV/mm, or greater than or equal to 15 kV/mm. D may be less than or equal to 250 kV/mm, or less than or equal to 200 kV/mm, or less than or equal to 100 kV/mm, or less than or equal to 50 kV/mm. D can be in the range of 10 kV/mm to 30 kV/mm or in the range of 15 kV/mm to 25 kV/mm.

One or more (e.g., each) power semiconductor prepackage may further include one or more electrically conductive layers. The one or more electrically conductive layers can extend in the xy direction. At least one of the one or more electrically conductive layers may be embedded in the solid electrical insulating material of the power semiconductor prepackage. At least one of the one or more electrically conductive layers can be located on a side of the power semiconductor switching element opposite the electrical connection side of the prepackage and can be connected to at least one of the connections of the power semiconductor switching element. At least one electrical connection can extend in the Z direction from an electrically conductive layer of the prepackage to the electrical connection side of the prepackage.

Each power semiconductor prepackage can also have an electrical. Insulating layer included. The electrical insulating layer can be located on a side of the power semiconductor switching element that is opposite the electrical connection side of the prepackage. The electrical insulating layer can consist of a ceramic material. The electrical insulating layer can be embedded in the solid electrical insulating material of the power semiconductor prepackage. The electrical insulating layer may extend in the xy direction.

The at least one capacitor can be connected to the at least one power electronic switching element via at least one of the conductive layers of the multilayer planar carrier substrate.

The at least one capacitor can be a ceramic capacitor.

The gate driver circuit may be electrically connected to the gate terminal of each power semiconductor switching element through one or more electrical connections extending in the z-direction.

The power electronic converter may also contain a heat sink for dissipating heat from power semiconductor prepackages.

The converter can further have a thermal interface layer (TIL) between a heat dissipation side of the prepackage and the heat sink, the heat dissipation side of the prepackage being opposite the electrical connection side of the prepackage.

The at least one prepackage can be located between the multilayer carrier substrate and the heat sink. The heat sink may include one or more recessed areas that form one or more chambers for receiving the prepackages. Adjacent chambers can be separated by a wall.

The at least one prepackage can be embedded in the multilayer carrier substrate. The heat sink can be arranged adjacent to a heat dissipation side of the prepackage, which is opposite the electrical connection side of the prepackage.

A converter parameter η can be greater than or equal to 100 kW/m ³ K. η is a heat transfer coefficient between the heat dissipation side of the power semiconductor prepackage and a cooling medium of the heat sink divided by the size of the gap in the z direction between the heat dissipation side of the power semiconductor prepackage and the heat sink. Accordingly, η can be expressed as: $$\mathrm{\eta }=\frac{\text{H}}{{\text{G}}_3}$$

In this equation, h is the heat transfer coefficient between the heat dissipation side of the power semiconductor prepackage and the cooling medium of the heat sink, and G ₃ is the size of the gap in the z direction between the heat dissipation side of the power semiconductor prepackage and the heat sink. η may be greater than or equal to 500 kW/m ³ K, greater than or equal to 1 MW/m ³ K, or greater than or equal to 10 MW/m ³ K. η can be less than or equal to 1000 MW/m ³ K. η can be less than or equal to 500 MW/m ³ K, less than or equal to 150 MW/m ³ K or less than or equal to 100 MW/m ³ K. η can be in the range 1 MW/m ³ K to 1000 MW/m ³ K or in the range 10 MW/m ³ K to 100 MW/m ³ K or in the range 20 MW/m ³ K to 50 MW/m ³ K . The converter parameter η can be in the range of 125 kW/m ³ K to 75 MW/m ³ K, or in the range of 30 MW/m ³ K to 45 MW/m ³ K, or in the range of 35 MW/m ³ K up to 40 MW/m ³ K.

The heat transfer coefficient between the heat dissipation side of the power semiconductor prepackage and a cooling medium of the heat sink, h, may be greater than or equal to 0.1 kW/m ² K, greater than or equal to 1 kW/m ² K, or greater than or equal to 5 kW/m ² K . h may be less than or equal to 50 kW/m ² K, less than or equal to 30 kW/m ² K, or less than or equal to 20 kW/m ² K. h can be in the range of 2.5 kW/m ² K to 15 kW/m ² K or in the range of 5 kW/m ² K to 10 kW/m ² K. The size of the gap in the z direction between the heat dissipation side and the heat sink, G ₃ , may be less than or equal to 2 mm, less than or equal to 1 mm, less than or equal to 0.8 mm, or less than or equal to 0.5 mm. G ₃ can be greater than or equal to 0.05 mm, greater than or equal to 0.1 mm, or greater than or equal to 0.2 mm.

The thermal interface layer (TIL) may have thermal conductivity and mechanical compressibility.

A converter parameter Ω can satisfy 0.1 MNK/Wm < Ω < 1 GNK/Wm. Ω is the mechanical compressibility of the thermal interface layer divided by the thermal conductivity of the thermal interface layer. Accordingly, Ω can be expressed as: $$\mathrm{\Omega }=\frac{\text{M}}{\text{k}}$$

In this equation, M is the mechanical compressibility of the thermal interface layer and k is the thermal conductivity of the thermal interface layer. Ω can be greater than or equal to 0.2 MNK/Wm, greater than or equal to 0.4 MNK/Wm, or greater than or equal to 0.6 MNK/Wm. Ω can be less than or equal to 500 MNK/Wm, less than or equal to 100 MNK/Wm, less than or equal to 10 MNK/Wm or less than or equal to 5 MNK/Wm. Ω can be in the range from 0.25 MNK/Wm to 2 MNK/Wm, or in the range from 0.7 MNK/Wm to 1.5 MNK/Wm, or in the range from 0.7 MNK/Wm to 1.5 MNK /Wm, or in the range from 0.8 MNK/Wm to 0.9 MNK/Wm.

The mechanical compressibility of the thermal interface layer, M, may be less than or equal to 3000 MN/m ² (300 MPa), or less than or equal to 100 MN/m ² (10 MPa), or less than or equal to 10 MN/m ² (1 MPa ) be. M can be greater than or equal to 0.5 MN/m ² (0.05 MPa). M can be in the range of 1 MN/m ² (0.1 MPa) to 5 MN/m ² (0.5 MPa) or in the range of 2.5 MN/m ² (0.25 MPa) to 3.5 MN /m ² (0.35 MPa).

The thermal conductivity of the thermal interface layer, k, may be less than or equal to 100 W/mK, or less than or equal to 25 W/mK, or less than or equal to 10 W/mK, or less than or equal to 5 W/mK. k can be greater than or equal to 0.5 W/mK or greater than or equal to 1 W/mK. k can be in the range of 2 W/mK to 10 W/mK or in the range of 3 W/mK to 4 W/mK.

The thickness (z-direction size) of the thermal interface layer may be in the range of 0.05 mm to 3 mm, or in the range of 0.075 mm to 1.5 mm, or in the range of 0.1 mm to 0.75 mm, or in the range range from 0.15 mm to 0.25 mm.

An electrical insulating layer can be arranged between the one or more power semiconductor switching elements of the prepackage and the heat sink. The electrical insulation may be part of the power semiconductor prepackage or located between the prepackage and the heat sink. The electrical insulating layer can be arranged between the prepackage and the thermal interface layer. The electrical insulating layer may be arranged between the thermal interface layer and the heat sink.

A transducer parameter ρ can be greater than or equal to 5 MVW/m ² K. ρ is a product of the thermal conductivity of the thermal interface layer and the electrical breakdown field strength of the electrical insulating layer. Accordingly, ρ can be expressed as: $$\mathrm{\rho }=\text{k}\times {\text{E}}_{\text{Break}}$$

In this equation, k is the thermal conductivity of the thermal interface layer and E _Break is the electrical breakdown field strength (which can also be called dielectric strength) of the electrical insulating layer. ρ may be greater than or equal to 10 MVW/m ² K, greater than or equal to 25 MVW/m ² K, or greater than or equal to 50 MVW/m ² K. ρ can be less than or equal to 25 GVW/m ² K or less than or equal to 5 GVW/m ² K or less than or equal to 500 MVW/m ² K or less than or equal to 250 MVW/m ² K. ρ can be in the range of 25 MVW/m ² K to 5 GVW/m ² K or in the range of 50 MVW/m ² K to 250 MVW/m ² K.

The electric breakdown field strength (dielectric strength) E _Break of the electrical insulating layer can be greater than or equal to 5 kV/mm. E _Break can be less than or equal to 250 kV/mm. E _Break can be in the range of 10 kV/mm to 50 kV/mm or in the range of 10 kV/mm to 100 kV/mm or in the range of 15 kV/mm to 25 kV/mm.

The thickness (z-direction size) of the thermal interface layer may be in the range of 0.05 mm to 3 mm, or in the range of 0.075 mm to 1.5 mm, or in the range of 0.1 mm to 0.75 mm, or in the range range from 0.15 mm to 0.25 mm.

The thickness (z-direction size) of the electrical insulating layer may be in the range of 0.025 mm to 2 mm, or in the range of 0.025 mm to 1 mm, or in the range of 0.05 mm to 1 mm, or in the range of 0.1 mm to 0.8 mm or in the range of 0.2 mm to 0.3 mm.

The thermal interface layer may have a relatively low electrical conductivity (ie, high dielectric strength) to reduce the transducer's dependence on additional electrical insulation measures such as the special electrical insulation layer described above. Therefore, some embodiments may not include an electrical insulating layer. In such embodiments, a transducer parameter λ may be greater than or equal to 1 TW/SK, where λ is defined as a thermal conductivity of the thermal interface layer divided by an electrical conductivity of the thermal interface layer. The converter parameter λ can be expressed as follows: $$\mathrm{\lambda }=\frac{\text{k}}{\text{P}}$$

In this equation, k is the thermal conductivity of the thermal interface layer and P is the electrical conductivity of the thermal interface layer. λ can be less than or equal to 100 TW/SK or less than or equal to 10 TW/SK or less than or equal to 1 TW/SK or less than or equal to 500 TW/SK. λ can be greater than or equal to 10 TW/SK, greater than or equal to 50 TW/SK, greater than or equal to 100 TW/SK or greater than or equal to 200 TW/SK. λ can be in the range of 100 TW/SK to 500 TW/SK or in the range of 300 TW/SK to 400 TW/SK.

The electrical conductivity of the thermal interface layer, P, may be less than or equal to 0.1 pS/m (ie, 1×10 ^-13 S/m). P can be greater than or equal to 1 fS/m (ie 1x10 ^-15 S/m). P can range from 5 fS/m to 50 fS/m or from 7.5 fS/m to 25 fS/m.

The thermal conductivity of the thermal interface layer, k, may be greater than or equal to 0.1 W/mK. k can be less than or equal to 150 W/mK. k can be in the range of 0.5 W/mK to 100 W/mK, or in the range of 1 W/mK to 25 W/mK, or in the range of 2 W/mK to 10 W/mK, or in the range of 2 .5 W/mK to 5 W/mK.

The multilayer planar support substrate may include at least one outer conductive layer and/or at least one inner conductive layer. The at least one inner conductive layer may be thicker than the outer conductive layer (e.g., at least twice or at least three times as thick). The outer conductive layer and the inner conductive layer may be electrically connected by at least one connection (e.g., a plurality of connections) extending in the z-direction through the support substrate. Each power semiconductor prepackage may further include an electrical connection from at least one terminal of the one or more power semiconductor switching elements to the outer conductive layer of the multilayer planar carrier substrate, the electrical connection extending in the z-direction through the solid insulating material.

Each power semiconductor prepackage may further comprise an electrical connection from at least one of its connections to an electrical connection side of the power semiconductor prepackage, the electrical connection extending in the z-direction through the solid insulating material. Electrical connections may extend from each of the connections to the electrical connection side of the power semiconductor prepackage, with each electrical connection extending in the z-direction through the solid insulating material.

At least one of the connections of each of the at least one power semiconductor switching elements can be connected to at least one of the conductive layers of the multilayer planar carrier substrate on the electrical connection side of the power semiconductor prepackage. The at least one of the connections can be a source and/or drain connection of the power semiconductor switching element.

For every power semiconductor prepackage, the electrical connection side of the prepackage can form a flat surface. The prepackage can be surface-mounted on its electrical connection side on a planar surface of the multilayer planar carrier substrate.

For each power semiconductor prepackage, any electrical connection extending from at least one of the terminals of the power semiconductor switching element through the solid insulating material may terminate at the planar surface of the prepackage. Each power semiconductor prepackage can be surface mounted on the surface of the multilayer planar carrier substrate by soldering, sintering, or gluing the terminal electrical terminal to an electrical terminal of the multilayer planar carrier substrate.

The flat surface of the electrical connection side can further have conductive pads (e.g. solder pads) for connecting the connections of the power semiconductor switching element to the surface of the multilayer planar carrier substrate.

The completed electrical connection can be connected (e.g. soldered, sintered or glued) to an outer conductive layer of the multilayer carrier substrate. Through these connections, each power semiconductor prepackage is separated from the planar surface of the multilayer planar Carrier substrate spaced so that a gap (here referred to as a prepackage gap) is created between the multilayer planar carrier substrate and the electrical connection side.

The power electronic converter can be an AC-DC converter (i.e. an inverter or a rectifier). The AC-DC converter can be a multi-phase AC-DC converter. The power circuit can include a phase leg for each of the multiple phases. In this case, the parasitic inductance of the power circuit of the commutation cell corresponds to the parasitic inductance of a phase leg. Those skilled in the art will understand that the parasitic inductance of each phase leg is the same, except for inherent variations due to variations in components and manufacturing.

The AC-DC converter can be a two-stage converter that includes two logical switches per phase. The number of power semiconductor prepackages per logical switch may be greater than or equal to three. In some examples, a multi-phase (e.g., three- or four-phase) AC-DC converter includes more than 50 power semiconductor prepackages.

The power electronic converter can be a DC-DC converter. The power circuit of the commutation cell can also include an inductor.

The at least one capacitor can be a single capacitor, e.g. B. a single “DC-Link” capacitor or “input capacitor”. In other examples, a more complex DC filter with a variety of capacitors may be used.

The electric machine can be of any suitable type and configuration. In a specific example, the electric machine is a transverse flux machine (i.e., a transverse flux machine). The transverse flow electric machine can be air-cooled. The transverse flux electric machine and the power semiconductor prepackages can both be air-cooled by a common cooling system.

An electric propulsion unit (EPU) for an aircraft is also envisaged. The EPU includes an electric motor and an AC-DC power electronic converter according to the AC-DC power electronic converters described above. The AC-DC power electronic converter is designed as an inverter and is arranged so that it supplies a winding of the electric motor with power.

A gas turbine engine is also provided. The gas turbine engine includes a shaft; an electric machine having a rotor mechanically coupled to the shaft; and an AC-DC power electronic converter according to the AC-DC power electronic converters described above. The power electronic AC-DC converter is arranged so that it supplies power to a winding of the electrical machine or receives power from it.

Also provided is an aircraft electrical power system comprising a power electronic converter in accordance with the power electronic converters described above.

An aircraft is also provided that includes the EPU, gas turbine engine, or electrical power system described above. In one group of embodiments, the aircraft is an electric vertical takeoff and landing (eVTOL) aircraft with a plurality of EPUs as set forth above.

One skilled in the art will understand that a feature described in relation to one of the above aspects can also be applied, mutatis mutandis, to any other aspect unless they are mutually exclusive. In addition, unless mutually exclusive, each feature described herein may be applied to any aspect and/or combined with any other feature described herein.

BRIEF DESCRIPTION OF DRAWINGS

• zeigt den Schaltplan der Kommutierungszelle eines einphasigen, zweistufigen AC-DC Wandlers;
• ist ein schematischer Querschnitt des einphasigen, zweistufigen AC-DC-Wandlers aus ;
• ist ein weiterer schematischer Querschnitt des einphasigen, zweistufigen AC-DC Wandlers aus , der weitere Eigenschaften wie z.B. elektrische Felder zeigt;
• Die und zeigen eine schematische Querschnittsansicht und eine Draufsicht des einphasigen, zweistufigen AC-DC-Wandlers aus und veranschaulichen die Berechnung des Volumens der Kommutierungszelle;
• ist ein schematischer Querschnitt eines einzelnen Leistungshalbleiter-Prepackages, der Details zeigt, die in den und nicht sichtbar sind;
• ist ein schematischer Querschnitt des Leistungshalbleiter-Prepackages aus , das zwischen einem mehrschichtigen Trägersubstrat und einem Kühlkörper angeordnet ist;
• ist ein schematischer Querschnitt eines Leistungshalbleiter-Prepackages, das zwischen einem mehrschichtigen Trägersubstrat und einem Kühlkörper angeordnet ist und ein elektrisch isolierendes Material in einem Prepackage-Spalt aufweist;
• veranschaulicht, wie Low- und High-Side-Leistungshalbleiter-Schaltelemente einer Halbbrücke durch ein mehrschichtiges Trägersubstrat elektrisch verbunden werden können;
• zeigt, wie Leistungshalbleiter-Schaltelemente durch ein mehrschichtiges Trägersubstrat elektrisch parallel geschaltet werden können;
• Die und zeigen, wie die einphasigen Anordnungen der auf mehrere Phasen ausgedehnt werden können;
• ist eine weitere schematische Querschnittsdarstellung des leistungselektronischen Wandlers von ;
• ist eine schematische Querschnittsdarstellung einer alternativen Anordnung, bei der Leistungshalbleiter-Prepackages und ein Kühlkörper auf gegenüberliegenden Seiten des mehrlagigen Trägersubstrats angeordnet sind;
• Die und zeigen alternative Anordnungen, bei denen Leistungshalbleiter-Prepackages in ein mehrschichtiges Trägersubstrat eingebettet sind;
• zeigt einen zweistufigen, dreiphasigen Wechselrichter, der einen dreiphasigen Motor versorgt;
• zeigt eine Vielzahl von H-Brücken-Wechselrichter-Schaltungen, die einen Vierphasenmotor versorgen;
• zeigt einen zweistufigen, dreiphasigen Wechselrichter mit einem komplexeren DC-seitigen Filter;
• zeigt einen DC-DC Wandler, der mit den Anschlüssen einer Batterie verbunden ist;
• zeigt einen DC-AC-DC Wandler, der mit den Anschlüssen einer Batterie verbunden ist;
• ist eine schematische Darstellung eines elektrischen Flugzeugtriebwerks und -antriebs;
• ist eine schematische Darstellung eines mehrkanaligen elektrischen Energie- und Antriebssystems für ein Flugzeug mit sechs elektrisch betriebenen Propulsoren;
• ist eine perspektivische Ansicht eines elektrisch senkrecht startenden und landenden (eVTOL) Flugzeugs mit sechs elektrisch angetriebenen Propulsoren in einer eVTOL-Flugkonfiguration;
• ist eine perspektivische Ansicht des eVTOL-Flugzeugs aus in einer Vorwärtsflugkonfiguration;
• zeigt die allgemeine Anordnung eines Mantelstromtriebwerks für ein Flugzeug;
• ist eine schematische Darstellung eines hybridelektrischen Flugzeugantriebssystems; und
• zeigt einen elektrisch betriebenen Antrieb, wie er in einem elektrischen oder hybridelektrischen Antriebssystem verwendet werden kann.

The embodiments will now be described, by way of example only, with reference to the accompanying drawings, which are purely schematic and not to scale, and in which:

• shows the circuit diagram of the commutation cell of a single-phase, two-stage AC-DC converter;
• is a schematic cross section of the single-phase, two-stage AC-DC converter ;
• is another schematic cross section of the single-phase, two-stage AC-DC converter , which shows further properties such as electric fields;
• The and show a schematic cross-sectional view and a top view of the single-phase, two-stage AC-DC converter and illustrate the calculation of the volume of the commutation cell;
• is a schematic cross section of a single power semiconductor prepackage, showing details included in the and are not visible;
• is a schematic cross section of the power semiconductor prepackage , which is arranged between a multilayer carrier substrate and a heat sink;
• is a schematic cross section of a power semiconductor prepackage disposed between a multilayer carrier substrate and a heat sink and having an electrically insulating material in a prepackage gap;
• illustrates how low- and high-side power semiconductor switching elements of a half bridge can be electrically connected through a multilayer carrier substrate;
• shows how power semiconductor switching elements can be electrically connected in parallel through a multilayer carrier substrate;
• The and show how the single-phase arrangements of the can be extended to several phases;
• is another schematic cross-sectional representation of the power electronic converter from ;
• is a schematic cross-sectional representation of an alternative arrangement in which power semiconductor prepackages and a heat sink are arranged on opposite sides of the multilayer carrier substrate;
• The and show alternative arrangements in which power semiconductor prepackages are embedded in a multilayer carrier substrate;
• shows a two-stage, three-phase inverter supplying a three-phase motor;
• shows a variety of H-bridge inverter circuits powering a four-phase motor;
• shows a two-stage, three-phase inverter with a more complex DC-side filter;
• shows a DC-DC converter connected to the terminals of a battery;
• shows a DC-AC-DC converter connected to the terminals of a battery;
• is a schematic representation of an electric aircraft engine and propulsion system;
• is a schematic representation of a multi-channel electrical power and propulsion system for an aircraft with six electrically powered propulsors;
• is a perspective view of an electric vertical takeoff and landing (eVTOL) aircraft with six electrically powered propulsors in an eVTOL flight configuration;
• is a perspective view of the eVTOL aircraft in a forward flight configuration;
• shows the general arrangement of a turbofan engine for an aircraft;
• is a schematic representation of a hybrid-electric aircraft propulsion system; and
• shows an electrically operated drive as it can be used in an electric or hybrid-electric drive system.

DETAILED DESCRIPTION

is a circuit diagram of a power electronic converter 10. For simplicity, one phase of a two-stage AC-DC inverter is shown. However, it should be noted that the This disclosure is not limited to converters of this type of circuit, and additional examples are disclosed.

The power electronic converter 10 has a commutation cell with two parts: a power circuit and a gate driver circuit.

The power circuit has two DC inputs (DC-IN _L and DC-IN _H ) and a single-phase AC output (AC-OUT). A half-bridge circuit consisting of two power semiconductor switching elements 121L, 121H is connected between the DC inputs and the AC output of the power circuit. The letters “L” and “H” denote the low and high voltage sides of the half bridge, which are connected to the low and high voltage DC inputs. The power circuit also includes an intermediate smoothing capacitor 14, which is often referred to as a “DC link capacitor” in connection with an AC-DC converter. The function of the DC link capacitor is known to those skilled in the art.

Each power semiconductor switching element 121L, 121H has three terminals. In the case of a MOSFET, the connections are called source (S), drain (D) and gate (G). The current flowing from the DC inputs to the AC output flows between the source (S) and drain (D), while the voltage and current at the gate (G) control whether the path between the source (S) and drain (D) is conductive or not . The power semiconductor switching elements are preferably MOSFETs, in particular MOSFETs based on silicon carbide (SiC), although other semiconductor technologies such as gallium nitride (GaN) can also be used. As those skilled in the art know, the use of MOSFETs eliminates the need for discrete, parallel-connected diodes due to the inherent “body diode” nature of a MOSFET.

The gate driver circuit 13L, 13H is electrically connected to the gate terminal of the MOSFET 121L, 121H and configured to provide switching signals thereto to control the conduction of the MOSFETs, i.e. H. to control whether current can flow between the source and drain or whether current flow is blocked. The gate driver circuit effectively acts as an amplifier of signals received from a control device (not shown), e.g. B. a digital control device that works with signals of lower voltages, e.g. B. 3 V to 5 V, works and delivers this. In this example, the gate driver circuit is also connected to the drains of the MOSFETs 121L, 121H, although those skilled in the art will appreciate that this is not necessarily the case and that the gate and drains may be isolated from each other.

The power circuit further includes an inductor L _P. It should be understood that the inductance L _P is not a discrete component of the power circuit, but represents the combined parasitic inductance of the power circuit. Parasitic inductance is the inherent inductance of components and the connections between components that is not intentionally introduced into the circuit. The gate driver circuits 13L, 13H also include inductors L _G ; These also represent the parasitic inductances of the gate driver circuits 13L, 13H and are not discrete components.

Parasitic inductance is a notable problem in power electronic converters because it causes a loss mechanism: switching losses. The higher the parasitic inductance, the higher the switching losses. The level of switching losses also increases with the operating voltage of the power semiconductor switches 121L, 121H and with the switching frequency of the power semiconductor switches 121L, 121H. This means that high parasitic inductance also limits the system designer's ability to choose higher values for the converter's operating voltage and maximum switching frequency, as these must be kept lower to keep switching losses at a tolerable level. These are undesirable restrictions. Using a lower voltage requires a higher current to achieve the same power (P = I × V), which increases the resistance losses (i.e. I ² R losses) in the power circuit and e.g. B. increased in the windings of the electrical machine to which the converter is connected. Using a lower switching frequency limits the quality of the output voltage/current waveform, resulting in undesirable effects such as torque ripple in the rotor of an electrical machine connected to power electronics.

The present disclosure provides power electronic converters with commutation cells with reduced parasitic inductances. This not only reduces switching losses, but also allows the use of higher operating voltages, higher switching frequencies and higher voltage and current ramp rates when switching. Overall, this significantly increases the operating efficiency of the converters compared to state-of-the-art power electronic converters.

Table 2 contains exemplary values for the parasitic inductance of the power circuit and the maximum operating efficiency of AC-DC converters in the power range 50-400kW. The stated powers are peak nominal powers, ie the highest electrical power that can be regulated by the converter. It differs from the continuous rated power, which, for example, B. depends on the ambient conditions during operation and the capabilities of the converter's cooling system. Table 2 Peak rated power (kW) Parasitic inductance of the power circuit, L _P (nH) Peak operating efficiency Example of the latest technology Example of this disclosure Example of the latest technology Current disclosure 50 40 8th 95% to 96% 97% to > 99% 100 20 4 95% to 96% 97% to > 99% 150 15 3 95% to 96% 97% to > 99% 200 10 2 95% to 96% 97% to > 99% 400 5 1 95% to 96% 97% to > 99%

Only the parasitic inductance L _P of the power circuit and not the entire commutation cell is given. This is because the gate driver circuit is electrically decoupled from the power circuit and therefore the parasitic inductances of the two circuits are not combined. However, it should be noted that the tight integration of the gate driver circuit into the commutation cell according to the present disclosure results in a lower value of L _G.

As can be seen from Table 2, the parasitic inductance value of power electronic converters according to the present disclosure can be approximately five times lower than the comparable prior art. Together with other measures disclosed herein, this results in operating efficiencies of up to 99% and greater, compared to values of 95-96% commonly achieved in prior art power electronic modules.

It can also be seen from Table 2 that the parasitic inductance of the power circuit generally decreases as the rated power increases. This is because the peak power rating is generally increased by parallelizing the power semiconductors of the power circuit, i.e. i.e., at higher performances z. B. each low-side MOSFET 121L of a certain phase is realized by several MOSFETs connected in parallel. This parallel connection of the components has the additional effect of reducing the parasitic inductance of the power circuit. In this way, the parasitic inductance of the power circuit can be reduced to an arbitrarily low value, but the additional components significantly increase the weight and volume of the converter and reduce the power density.

L_P = Parasitärinduktivität des Leistungskreises
Vol. = Kleinstes quaderförmiges Volumen, das die Kommutierungszelle umschließtThus, a power electronic converter can be characterized by a converter inductance volume parameter α, which is defined as the product of the parasitic inductance of the power circuit and the volume of the commutation cell: $$\mathrm{\alpha }={\text{L}}_{\text{P}}\times \text{Vol}\text{.}$$

L _P = parasitic inductance of the power circuit
Vol. = Smallest cuboid volume that encloses the commutation cell

As already mentioned, the volume is defined as the smallest cuboid volume that encloses the entire commutation cell, that is, the combination of power circuit and gate driver circuit. An example of the volume of the commutation cell is in the and and is described below.

Table 3 shows exemplary values of α for power electronic converters according to the present disclosure. The values of α are characteristically lower than in the prior art and are associated with a combination of high efficiency and high power density. The values of α are given in pHm ³ (i.e. pico-Hm ³ or ×10 ^-12 Hm ³ ). Table 3 Peak rated power (kW) α = L _P × Vol. (pHm) ³ example 1 Example 2 Example 3 50 0.60 1.70 4.80 100 0.42 1.20 3.40 150 0.37 1.04 2.94 200 0.30 0.85 2.40 400 0.21 0.60 1.70

Example values for the parasitic inductance of the power circuit are listed in Table 2 and are preferably less than or equal to 16 nH for converters with peak powers in the range of 25-500 kW. The volume of the commutation cells generally increases with the rated power, and values of less than 1,000 cm ³ are preferred for converters with a power of up to 500 kW. The volume of the commutation cells is generally greater than or equal to 100 cm ³ , with volumes of 150 to 600 cm ³ representing a good balance between power density and relatively easy heat dissipation.

For reference, Table 4 contains the values of α _p , that is, the power normalized value of α, and the values of the product of the power and parasitic inductance of the power circuit. Table 4 Peak rated power (kW) αP ₌ α/P (aHm ³ /W) L *P _P (mHW) example 1 Example 2 example 1 Example 2 50 12 96 0.2 0.8 100 4.2 34 0.2 0.8 150 2.5 20 0.2 0.8 200 1.5 12 0.2 0.8 400 0.7 5.7 0.2 0.8

In the components of the commutation cell are arranged on a carrier substrate 11. The carrier substrate 11 is a multi-layer carrier substrate 11, e.g. B. a rigid printed circuit board (PCB) that has alternating insulating and conductive layers extending in the xy plane. The components of the commutation cell, including the power circuit and the gate driver circuit, are mounted on the multilayer planar support substrate 11 and electrically connected through the multilayer support substrate 11. The electrical connections are made through a combination of the conductive layers of the multilayer planar support substrate 11 and connections extending through the support substrate 11 in a z-direction defined perpendicular to the xy-direction of the planar substrate and its conductive layers. The connections extending in the z direction can be, for example, conductive vias or filled holes (e.g. laser micro-vias with diameters of the order of approximately 100 μm) which have both an x and/or y -Component as well as a z-component (ie, the connections can run parallel to the z-direction or form an angle with it).

In It is also shown schematically that the power semiconductor switching elements 121L, 121H of the power circuit are each contained in a power semiconductor prepackage 12L, 12H. The person skilled in the art understands this to mean that the power semiconductors (e.g. MOSFETs) are embedded in a solid insulating material that electrically isolates the power semiconductors and their connections from their surroundings. The Prepacka ges and their connections to each other and to other components of the commutation cell are described in more detail below.

is a schematic cross section of a power electronic converter 10 according to the present disclosure. For the sake of clarity and simplicity, one phase of a two-stage AC-DC converter is also shown here. The x-direction and the z-direction are indicated. The y-direction extends in the plane of the page.

The converter 10 includes a multilayer carrier substrate 11, power semiconductor prepackages 12, a gate driver circuit 13, a DC link capacitor 14, direct current inputs (DC-IN) and an alternating current output (AC-OUT). In An integrated heat sink 15 is also shown, which is connected to a cooling side (here also referred to as the heat dissipation side) of the prepackages 12 via a thermal interface layer 16.

The multilayer planar support substrate 11 has opposing first and second planar surfaces 111a and 111b that define an xy direction and a z direction perpendicular to the xy direction. The multilayer substrate 11 contains alternating layers of insulating and electrically conductive material. The electrically conductive layers 112 are typically made of copper, but can also be formed of any other suitable conductive material such as silver, gold or aluminum. The insulating layers can consist of the base material of the carrier substrate 11.

The multilayer carrier substrate 11 may preferably be a rigid circuit board, wherein the base material and the conductive layers may consist of a glass fabric impregnated with resin, as is common in circuit board manufacturing. However, the multi-layer carrier substrate 11 can also have a different shape, e.g. B. a ceramic-based carrier substrate or a flexible circuit board with a flexible polymer film base. The use of a rigid material is generally preferred, also to allow the support substrate 11 to function effectively as a structural component of the transducer 10.

The number of layers in the multilayer carrier substrate 11 can vary depending on the application and depends in part on the specifics of the power circuit, e.g. B. on the number of phases in an AC-DC converter and the number of power semiconductors connected in parallel in each logical switch. In a specific example, there are sixteen layers, including eight insulating layers and eight conductive layers 112.

Shortly closed 4B : In some examples, at least a portion of one of the planar surfaces 111b of the support substrate 11 carries a metal layer 1121 that defines one or more areas for electrical connections on the surface 111b of the substrate 11. Again referring to 2A , additionally or alternatively to the outer layer, electrical connections in the form of, for example, vias or filled holes 114 extend in the z-direction through the carrier substrate 11 and end at the planar surface 111b. The points at which the electrical connections 114 end define electrical connections on the planar surface 111b of the carrier substrate 11.

Each prepackage 12 includes a power semiconductor switching element 121 embedded in a solid insulating material 122. Embedding the power semiconductor switching elements 121 in solid insulating material means that the semiconductor chips and the connections are not surrounded by air gaps, which reduces the risk of electrical breakdowns even at high converter voltages. This allows higher voltages to be used and/or the power semiconductors 121 and other components to be closer together, which increases the power density of the converter 10.

Table 5 contains example values for the maximum reverse voltage (ie, the source-drain reverse voltage or “nominal voltage”) of the power semiconductor switching elements 121 according to the present disclosure. Table 5 Peak rated power (kW) Power semiconductor reverse voltage (V) example 1 Example 2 Example 3 50 1200 800 1600 100 1200 800 1600 150 1200 800 1600 200 1200 800 1600 400 1200 800 1600

As can be seen, the source-drain blocking voltages used in accordance with the present disclosure are high. In general, the reverse voltage can range from about 600 V to 1,800 V, with values greater than or equal to 800 V being preferred to limit peak current and reduce conduction losses. State-of-the-art power electronic converters typically have much lower reverse voltages, with voltages of even 600 V being rare. It can also be seen that the reverse voltage does not increase with the peak power of the converter. This is because most or all of the increase in peak power is achieved through parallelization in the power circuit. In other examples, a slightly higher reverse voltage may be used for converters with higher power ratings (e.g. over 200 kW).

In general, each power semiconductor switching element 121 has at least three terminals, including a gate terminal (G) for switching the conduction state of the switching element 121. In some embodiments, the power semiconductor switching element 121 may have more than three terminals, for example when one or more terminals for measurements are provided (e.g. a Kelvin connection) or if additional shorted connections are present. In preferred embodiments, the power semiconductor switching elements 121 are MOSFETs, with the connections referred to as source (S), gate (G), and drain (D). In principle, however, other material-semiconductor switching elements (e.g. IGBTs) can also be used instead of MOSFETs. The solid insulating material 122 can be any suitable insulating material, e.g. E.g. FR4.

As follows with reference to the 4A-4B As will be described in more detail, electrical connections extend from the connections of the power semiconductor switching elements 121 through the solid insulating material 122 and end at an electrical connection side/surface 123a of the prepackage 12. The electrical connection side 123a of the prepackage 12 thus forms a substantially flat surface exposed electrical connection points.

The electrical connection side of the prepackages 123a faces one of the planar surfaces 111b of the carrier substrate 11. The prepackages 12 are surface mounted at their flat electrical connection sides 123a on the planar surface 111b, e.g. B. by soldering, sintering or gluing (e.g. sinter bonding using a mixture of adhesive and sintering paste) of the electrical connection points of the prepackages 12 with the electrical connection points or the area(s) of the planar surface of the carrier substrate 11. shows schematically the soldered/sintered/glued connections 113.

The thickness, in particular the size in the z-direction, of the connections 113, which form a prepackage gap between the opposing surfaces 123a, 111b of the prepackages 12 and the carrier substrate 11, is small. For example, the thickness and the gap, measured parallel to the z-direction, can be less than 500 μm, for example between 20 μm and 250 μm. In a particular embodiment, the gap is approximately 100 μm.

Terminating the electrical connections from the chip connections to a flat surface 123a of the prepackage 12 and surface mounting to make the further electrical connections through the circuit board reduces the overall size of the converter 10 in the z direction, thereby reducing the size and weight of the converter 10 . In addition, the surface mounting of the prepackages 12 reduces the effects of “open-loop” effects in the electrical connections between the power semiconductors 121, the gate driver circuit 13 and the DC link capacitor 14. This allows the parasitic inductance of the commutation cell to be significantly reduced, which reduces switching losses and, for example, B. enables the use of a higher switching frequency.

The gate driver circuit 13 is electrically connected to the gate terminals of the power semiconductor switching elements 121 and configured to supply switching signals to these terminals. In the illustrated embodiment, the gate driver circuit 13 is mounted on the first planar surface 111a of the carrier substrate 11, opposite the second planar surface 111b facing the power semiconductor prepackages 12. In other embodiments, the prepackages 12 and the gate driver circuit 13 may be on the same side of the carrier substrate 11, e.g. B. on the second page 111b. In the illustrated embodiment, the electrical connection 114 between the gate driver circuit 13 and the gate terminals of the power semiconductor switching elements 121 extends in the z direction through the carrier substrate to the surface 111b of the carrier substrate 11. The further path then leads via a solder connection 113 and then via the electrical connection, which leads in the z direction through the solid insulating material 122 of the prepackages 12 to the gate terminal. In other embodiments, the connection between the gate driver circuit 13 and the gate terminals may be made through one or more conductive layers 112 of the carrier substrate 11.

Table 6 contains exemplary values for the switching frequency of the power semiconductor switching elements of a power electronic converter according to the present disclosure. Table 6 also contains maximum absolute values for the rate of change of the source-drain voltage (measured in units of kilovolts per microsecond) of the power semiconductor switching elements 121 during a switching cycle. The values given are maximum values of the switching frequency that can occur during operation. Table 6 Peak rated power (kW) f _max (kHz) |dV/dt| _max (kV/µs) example 1 Example 2 example 1 Example 2 50 50 80 30 45 100 50 80 30 45 150 50 80 30 45 200 50 80 30 45 400 50 80 30 45

In many converter arrangements, the switching frequency of each power semiconductor switching element 121 is the same and the same voltage value is used, so that the above-mentioned maximum values are the same for each individual power semiconductor switching element 121. However, it is known that some converter architectures, e.g. B. multi-level converter architectures (e.g. modular multi-level converter architectures), which are connected to several mains voltage levels, use different voltages and / or switching frequencies for different power semiconductor switching elements 121. In these cases, the values given correspond to the maximum value of each individual power semiconductor switching element 121 in the converter.

Converters according to the present disclosure use maximum frequencies greater than 10 kHz, although switching frequencies greater than 30 kHz are generally preferred, which, as discussed below, may facilitate a reduction in the required capacitance in the power circuit. Unlike many prior art systems where the parasitic inductance of the converter commutation cell limits the maximum usable frequency, the low parasitic inductance of the converter commutation cell may mean that other system limitations limit the maximum frequency. For example, the insulation of the windings of an electrical machine that is connected to an AC-DC converter can dictate a maximum desirable switching frequency. Generally, a switching frequency of 100 kHz or less is used.

The use of a high switching frequency and a high source-drain block voltage results in a particularly high value for the maximum rate of change of the source-drain voltage during operation. The rapid switching between the on and off states of the power semiconductor switching elements results in clean switching operations and improved output waveforms, which limits the harmonic content of the waveforms and e.g. B. improves the efficiency of the converter and reduces the torque ripple of the electrical machine.

f_max = Maximale Frequenz der Schaltsignale
|_max = maximale Änderungsrate der Source-Drain-Spannung während des BetriebsA power electronic converter can be characterized by a converter switching parameter β, which is defined as the product of the maximum switching frequency of the switching signals and the maximum rate of change of the source-drain voltage of the power semiconductor switching elements during operation. $$\mathrm{\beta }={\text{f}}_{\text{Max}}\times {\left|\text{dv/german}\right|}_{\text{Max}}$$

f _max = Maximum frequency of the switching signals
| _max = maximum rate of change of source-drain voltage during operation

Table 7 shows exemplary values of β for power electronic converters according to the present disclosure. The values of β are characteristically higher than the prior art and are associated with a combination of high efficiency, high power density and high quality (e.g. highly sinusoidal AC) output waveforms. The values of β are given in PV/s ² (i.e. peta-V/s ² or ×10 ¹⁵ V/s ² ). Table 7 Peak rated power (kW) β = f _max × |dv/dt| _max (PV/s) ² example 1 Example 2 Example 3 50 0.5 1.5 5.0 100 0.5 1.5 5.0 150 0.5 1.5 5.0 200 0.5 1.5 5.0 400 0.5 1.5 5.0

In general, converters according to the present disclosure have a value of β that is greater than or equal to 0.3 PV/s ² . The values of β are preferably below about 10 PV/s ² to avoid problems such as: B. to mitigate insulation breakthroughs. Values in the range of 0.8 PV/s ² ≤ β ≤ 5 PV/s ² and especially 1.0 PV/s ² ≤ β ≤ 2.5 PV/s ² can provide a good balance between the competing effects.

Back to : The intermediate capacitor 14, which is part of the power circuit of the commutation cell, is electrically connected to the power semiconductor switching elements 121 via one or more conductive layers 112 of the multilayer carrier substrate 11. In the illustrated embodiment, the capacitor 14 is mounted on the first planar surface 111a of the carrier substrate 11, which faces the second planar surface 111b, which faces the power semiconductor prepackages 12. In other embodiments, the prepackages 12 and the capacitor 14 can be on the same side of the carrier substrate 11, e.g. B. on the second page 111b. Electrical connections between the capacitor 14 and the conductive layers 112 of the multilayer carrier substrate 11 can be made in various ways. In one example, the capacitor 14 is surface mounted in a manner similar to the prepackages 12, that is, by soldering, sintering, or bonding electrical contacts of the capacitor 14 to an electrical connection portion of the surface 111a of the substrate 11. The electrical connection may be external conductive layer on the surface 111a of the substrate (similar to that in 4B shown layer 1121) or an exposed end of one or more vias or filled holes 114, which extends in the z-direction from the surface 111a of the substrate 11 to an inner conductive layer 112 of the substrate 11. As an alternative to SMT (Surface Mount Technology), the connection between the capacitor 12 and the multilayer substrate 11 can also be made using THT (Through Hole Technology), both methods being known to those skilled in the art.

In modern converters, the capacitors that are an integral part of most AC-DC and DC-DC converter architectures add significant size and weight. This is a particular problem in the context of aerospace applications that are both size and weight sensitive and require relatively high power outputs (e.g. compared to electric vehicles and household appliances), requiring a higher overall power circuit capacity rating. However, by reducing the parasitic inductance, a lower capacity per unit of power can be achieved. This is partly because the low parasitic inductance allows for a high switching frequency. Increasing the switching frequency can reduce the required capacity of the power circuit.

Table 8 contains exemplary values for the total nominal capacity of the power circuit. The values of the total nominal capacity normalized to the peak power of the converter are also included. This shows that the values of the total nominal capacity and the standardized capacity are low compared to state-of-the-art power electronic converters. Table 8 Peak rated power (kW) Total nominal capacity C (µF) C/P (nF/W) example 1 Example 2 example 1 Example 2 50 15 40 0.3 0.8 100 30 80 0.3 0.8 150 45 120 0.3 0.8 200 60 160 0.3 0.8 400 120 320 0.3 0.8

In general, the values of the total nominal capacitance normalized to the peak nominal power are less than 5 nF/W, with values of less than 1 nF/W being preferred in order to achieve a small size and low weight of the converter. The use of low capacitance values can also enable the use of light weight capacitor technologies, particularly ceramic capacitors, resulting in further weight reduction. Capacity is known to vary somewhat depending on operating conditions, so rated capacity values are usually given in the literature. Table 8 shows the capacitances measured at nominal conditions of 25°C (298 K) and 1,000 V DC, which is typical for capacitance measurements.

The term “total nominal capacity of the power circuit” means the total capacity of all capacitors in the power circuit. In the simplest cases it can be a single capacitor. The single-phase two-stage AC-DC converter circuit of for example, contains a single intermediate circuit capacitor, as does the three-phase two-stage AC-DC converter circuit of . In other examples, there may be multiple capacitors: There may be a separate DC link capacitor for each phase, as in the H-bridge circuit in and the three-phase two-level circuit in the case is. When the power semiconductors in the power circuit are parallelized, each power semiconductor switching element can be connected to its own capacitor, or several power semiconductor switching elements 121 connected in parallel can be connected to a single larger capacitor.

C = Gesamtnennkapazität des Leistungskreises der Kommutierungszelle
P = Spitzen-Nennleistung des leistungselektronischen Wandlers
f_max = Maximale Frequenz der SchaltsignaleA power electronic converter can be characterized by a frequency-capacity converter parameter γ, which is defined as the total rated capacity of the power circuit divided by the product of the peak rated power of the power electronic converter and the maximum switching frequency of the gate switching signals: $$\mathrm{\gamma }=\frac{\text{C}}{\text{P}\times {\text{f}}_{\text{Max}}}$$

C = total nominal capacity of the power circuit of the commutation cell
P = peak rated power of the power electronic converter
f _max = Maximum frequency of the switching signals

Table 9 shows exemplary values of γ for power electronic converters according to the present disclosure. The values of γ are characteristically lower than in the prior art and are associated with a combination of high efficiency and high power density. The values of γ are given in fFs/W (i.e. femto-Fs/W or ×10 ^-15 Fs/W). Table 9 Peak rated power, P (kW) γ = C / (P × f) _max (fFs/W) example 1 Example 2 Example 3 50 2.5 10 100 100 2.5 10 100 150 2.5 10 100 200 2.5 10 100 400 2.5 10 100

In general, power electronic converters according to the present disclosure have values of γ less than or equal to 150 fFs/W. γ can be greater than about 1.0 fFs/W, with a lower limit provided to avoid problems associated with e.g. B. Insulation breakdown is limited at high switching frequencies. The values of γ may preferably be in the range of 4.0 fFs/W ≤ γ ≤ 25 fFs/W, whereby a good balance between high power density and reliable operation can be achieved.

V_block = Maximale Source-Drain-Sperrspannung von Leistungshalbleitern
L = Parasitärinduktivität des Leistungskreises
C = Gesamtnennkapazität des Leistungskreises der KommutierungszelleA power electronic converter can also be characterized by a converter frequency-capacity parameter δ, which is defined as the maximum source-drain blocking voltage of the power semiconductor switching elements of the power circuit divided by the product of the parasitic inductance of the power circuit and the total nominal capacity of the power circuit: $$\mathrm{\delta }=\frac{{\text{v}}_{\text{block}}}{\text{L}\times \text{C}}$$

V _block = Maximum source-drain blocking voltage of power semiconductors
L = parasitic inductance of the power circuit
C = total nominal capacity of the power circuit of the commutation cell

(PV/FH) Beispiel 1 Beispiel 2 Beispiel 3 50 1.0 6.0 30 100 1.0 6.0 30 150 1.0 6.0 30 200 1.0 6.0 30 400 1.0 6.0 30 Table 10 shows exemplary values of δ for power electronic converters according to the present disclosure. The values of δ are characteristically higher than in the prior art and are associated with a combination of high efficiency and high power density. The values of δ are given in PV/FH (i.e. peta-V/FH or ×10 ^-5 V/FH). Table 10 Peak rated power (kW) $\mathrm{\delta }=\frac{{\text{v}}_{\text{block}}}{\text{L}\times \text{C}}$

(PV/FH) example 1 Example 2 Example 3 50 1.0 6.0 30 100 1.0 6.0 30 150 1.0 6.0 30 200 1.0 6.0 30 400 1.0 6.0 30

In general, power electronic converters according to the present disclosure have values of δ greater than 0.5 PV/FH. The value of δ can be less than or equal to 40 PV/s ² . The value of δ can be greater than or equal to 1.5 PV/s ² . The value of γ can range from 2.5 PV/s ² to 25 PV/s ² . It can range from 4.0 PV/s ² to 15 PV/s ² .

f_max = Maximale Frequenz der Schaltsignale
/_max = maximale Änderungsrate der Source-Drain-Spannung während des Betriebs L = Parasitärinduktivität des Leistungskreises
C = Gesamtnennkapazität des Leistungskreises der KommutierungszelleA power electronic converter can also be characterized by a frequency-capacity converter parameter ε, which is defined as follows: $$\mathrm{\epsilon }=\frac{{\text{f}}_{\text{Max}}\times {\left|\frac{\text{dv}}{\text{German}}\right|}_{\text{Max}}}{\text{L}\times \text{C}}$$

f _max = Maximum frequency of the switching signals
/ _max = maximum rate of change of source-drain voltage during operation L = parasitic inductance of the power circuit
C = total nominal capacity of the power circuit of the commutation cell

(×10 ²⁷ V/s) ⁴ Beispiel 1 Beispiel 2 Beispiel 3 50 1.9 7.5 30 100 1.9 7.5 30 150 1.9 7.5 30 200 1.9 7.5 30 400 1.9 7.5 30 Table 11 shows exemplary values of ε for power electronic converters according to the present disclosure. The values of ε are characteristically lower than the prior art and are associated with a combination of high efficiency, high power density and high quality output waveforms. The values of ε are given in units of ×10 ²⁷ V/s ⁴ . Table 11 Peak rated power (kW) $\mathrm{\epsilon }=\frac{{\text{f}}_{\text{Max}}\times {\left|\frac{\text{dv}}{\text{German}}\right|}_{\text{Max}}}{\text{L}\times \text{C}}$

(×10 ²⁷ V/s) ⁴ example 1 Example 2 Example 3 50 1.9 7.5 30 100 1.9 7.5 30 150 1.9 7.5 30 200 1.9 7.5 30 400 1.9 7.5 30

In general, power electronic converters according to the present disclosure have values of ε less than or equal to 10 ²⁹ V/s ⁴ . The value of ε is generally greater than 5×10 ²⁶ V/s ⁴ , since lower values e.g. B. may be associated with insulation failure, although this depends to some extent on the application requirements (e.g. whether high quality insulation can be provided in an electrical machine). The values of ε may preferably be in the range of 1.5x10 ²⁷ V/s ⁴ to 3x10 ²⁸ PV/s ² as this represents a good balance between power density, efficiency and reliability.

Back to : The components of the converter 10, in particular the power semiconductor switching elements 121, generate heat that must be dissipated. Heat dissipation can be a particularly important aspect in power electronic converters according to the present disclosure due to their compactness and high power density. For this purpose, the converter 10 includes an integrated heat sink 15, which is in close thermal contact with the power semiconductor prepackages 12. To ensure efficient heat dissipation, a fixed path is provided between a heat dissipation side 123b of the prepackages (ie the bottom of the prepackage 12, as in shown, opposite the electrical connection side 123a), so that the heat sink 15 dissipates the heat from the prepackages 12 by thermal conduction. In the example shown, several prepackages 12 of the converter 10, and optionally all prepackages 12 of the converter, share a common heat sink 15.

The heat sink 15 itself can be designed in any suitable shape. He can e.g. B. made of aluminum or another heat-conducting material and cooled by a coolant flow, which can be a gas (e.g. air) or a liquid (e.g. water or oil). In some embodiments, a surface of the heat sink 15 opposite the prepackages 12 can be subjected to an impinging coolant flow in order to increase the heat transfer coefficient between the prepackages 12 and the cooling medium of the heat sink 15.

In order to ensure efficient heat dissipation by conduction, it is necessary that a high quality and uniform thermal interface exists between the heat dissipation side 123b of each of the prepackages 12 and the heat sink 15. This can be a challenge because although the heat dissipation side 123b of the prepackages 12 can be designed flat, there are certain manufacturing and assembly tolerances. For example, the thickness of the electrical contacts 113 between and within the prepackages 12 may vary slightly, which may lead to an inclination of the prepackages and/or to uneven distances between the heat dissipation side 123b and the heat sink 15. Another example is that the multilayer carrier substrate 11 can bend or locally deform. In view of this, the converter 10 is also shown as having a thermal interface layer (TIL) 16 between the heat dissipation side 123b of the prepackages 12 and the heat sink 15. The TIL 16 is intended to ensure a high quality thermal interface between the prepackages 12 and the heat sink 15. The TIL accommodates tolerance issues while providing a thermally conductive path in the z direction between the heat dissipation side 123b of each of the prepackages 12 and the heat sink 15. In some examples, the TIL 16 may also have a high thermal conductivity in the xy plane to dissipate heat to distribute over the surface of the heat sink 15. This can be particularly useful when a single TIL 16 serves multiple prepackages 12.

The TIL 16 can take one of several different forms, including solids (e.g., a layer of solder, a foil or film), semi-solids (e.g., a paste), or a liquid. In one group of examples, the TIL 16 is a layer of solder (e.g., indium-tin solder). In this case, to ensure good quality of the solder connection, each prepackage 12 can have its own TIL 16 and not just a single TIL. In another group of examples the TIL is a foil, e.g. B. Indium-tin or graphene foil, which is both thermally conductive and flexible. The use of a TIL with a certain compressibility can be advantageous in order to compensate for manufacturing tolerances as well as to separate the heat sink 15 from the prepackages 12, e.g. B. in the event of vibrations. The TIL 16 preferably has a thermal conductivity of at least 1 W/mK, and more preferably at least 2.5 W/mK. The thickness of the TIL 16 is generally less than a few mm, preferably less than 1 mm, and in a group of examples has a thickness between 100 μm and 500 μm.

h = Wärmeübergangskoeffizient vom Prepackage zum Kühlmedium
G₃ = Abstand in z-Richtung zwischen der Wärmeabfuhrseite der Prepackage und dem KühlkörperA power electronic converter can be characterized by a heat transfer converter parameter η, which is defined as follows: $$\mathrm{\eta }=\frac{\text{H}}{{\text{G}}_3}$$

h = heat transfer coefficient from the prepackage to the cooling medium
G ₃ = Distance in the z direction between the heat dissipation side of the prepackage and the heat sink

(MW/m ³ K) Beispiel 1 Beispiel 2 Beispiel 3 50 0.125 37.5 150 100 0.125 37.5 150 150 0.125 37.5 150 200 0.125 37.5 150 400 0.125 37.5 150 The gap G ₂ is in labeled. Table 12 shows exemplary values of η for power electronic converters according to the present disclosure. The values of η are characteristically high and are associated with a combination of compactness, high efficiency and high power density. The values of η are given in units of MW/m ³ K. Table 12 Peak rated power (kW) $\mathrm{\eta }=\frac{\text{H}}{{\text{G}}_3}$

(MW/ ^m3K ) example 1 Example 2 Example 3 50 0.125 37.5 150 100 0.125 37.5 150 150 0.125 37.5 150 200 0.125 37.5 150 400 0.125 37.5 150

In general, power electronic converters according to the present disclosure have η values greater than or equal to 100 kW/m ³ K (0.1 MW/m ³ K). However, values of η greater than or equal to 10 MW/m ³ K are generally preferred.

Table 13 shows exemplary values for the heat transfer coefficient h and the size of the gap G ₃ between the heat dissipation side of the power semiconductor prepackage and the heat sink. Table 13 Peak rated power (kW) H G3 _{_} (kW/m ² K) (mm) example 1 Example 2 Example 3 example 1 Example 2 Example 3 50 0.1 7.5 30 0.8 0.2 0.05 100 0.1 7.5 30 0.8 0.2 0.05 150 0.1 7.5 30 0.8 0.2 0.05 200 0.1 7.5 30 0.8 0.2 0.05 400 0.1 7.5 30 0.8 0.2 0.05

M = mechanische Kompressibilität der thermischen Schnittstellenschicht
k = Wärmeleitfähigkeit der thermischen SchnittstellenschichtA power electronic converter can also be characterized by a thermal interface parameter Ω, which is defined as follows: $$\mathrm{\Omega }=\frac{\text{M}}{\text{K}}$$

M = mechanical compressibility of the thermal interface layer
k = thermal conductivity of the thermal interface layer

(MNK/Wm) Beispiel 1 Beispiel 2 Beispiel 3 50 0 0.86 10³ 100 0 0.86 10³ 150 0 0.86 10³ 200 0 0.86 10³ 400 0 0.86 10³ Table 14 shows exemplary values for the thermal interface parameter Ω. The values of Ω are given in units of MNK/Wm (i.e. Mega-NK/Wm, equal to 10 ⁶ NK/Wm). Table 14 Peak rated power (kW) $\mathrm{\Omega }=\frac{\text{M}}{\text{k}}$

(MNK/Wm) example 1 Example 2 Example 3 50 0 0.86 10 ³ 100 0 0.86 10 ³ 150 0 0.86 10 ³ 200 0 0.86 10 ³ 400 0 0.86 10 ³

Transducers of the present disclosure preferably have Ω values corresponding to 0.1 MNK/Wm < Ω < 1 GNK/Wm, with the range 0.25 MNK/Wm < Ω < 2 MNK/Wm being preferred. Thermal interface parameters in this range can provide a good combination of heat transfer and mechanical properties.

Table 15 contains exemplary values for the mechanical compressibility M of the thermal interface layer and for the thermal conductivity k of the thermal interface layer. Table 15 Peak rated power (kW) M k (MN/m) ² (W/mK) Example 1 Example 2 Example 3 Example 1 Example 2 Example 3 50 0 3 100 1 3.5 90 100 0 3 100 1 3.5 90 150 0 3 100 1 3.5 90 200 0 3 100 1 3.5 90 400 0 3 100 1 3.5 90

is another representation of the converter 10 and illustrates the electric fields that develop between the components of the transducer 10.

The in Prepackage gap designated G1 is visible between the second surface 111b of the multilayer carrier substrate 11 and the electrical connection side/surface 123a of the prepackages 12. The prepackage gap G1 has a size that can be measured in the z direction. A first electric field 50 is built up in the prepackage gap G1, which results from the potential difference (voltage) between the second surface 111b of the multilayer carrier substrate 11 and the electrical connection surface 123a of the prepackages 12.

G₁ = Größe des Prepackage-Spalts in z-Richtung
E₁ = Maximale elektrische Feldstärke im Prepackage-SpaltA power electronic converter can be characterized by a converter parameter θ, which is defined as the size of the prepackage gap in the z-direction divided by a maximum electric field strength in the prepackage gap (referred to here as the first maximum electric field strength): $$\mathrm{\theta }=\frac{{\text{G}}_1}{{\text{<figure-callout id="1" label="E" filenames="DE102022205482A1\_0053.png" state="{{state}}">E</figure-callout>}}_1}$$

G ₁ = size of the prepackage gap in the z direction
E ₁ = Maximum electric field strength in the prepackage gap

Beispiel 1 Beispiel 2 Beispiel 3 50 0.5 6.25 250 100 0.5 6.25 250 150 0.5 6.25 250 200 0.5 6.25 250 400 0.5 6.25 250 Table 16 shows exemplary values for the converter parameter θ for power electronic converters according to the present disclosure. The values of θ are characteristically lower than in the prior art and are associated with a high power density. The values of θ are given in units of pm ² /V (pico-m ² /V, or ×10 ^-12 m ² /V). Table 16 Peak rated power (kW) $\mathrm{\theta }=\frac{{\text{G}}_1}{{\text{E}}_1}\left({\text{<figure-callout id="2" label="pm" filenames="DE102022205482A1\_0053.png" state="{{state}}">pm</figure-callout>}}^2\text{/V}\right)$

example 1 Example 2 Example 3 50 0.5 6.25 250 100 0.5 6.25 250 150 0.5 6.25 250 200 0.5 6.25 250 400 0.5 6.25 250

In general, power electronic converters according to the present disclosure have values of θ less than or equal to 300 pm ² /V. The value of θ is generally greater than or equal to 0.1 pm ² /V, as lower values e.g. B. may be associated with a greater risk of electrical shock. The values of θ may preferably be in the range of 2.0 pm ² /V to 50 pm ² /V as this represents a good balance between power density and reliability.

The person skilled in the art knows that the first maximum electric field strength is a maximum homogeneous electric field strength. In other words, it is the maximum field strength that is determined at a location that is sufficiently far away from sharp edges and/or obstacles in the gap Singularities of the electric field or other strongly localized maxima can lead. For example, shows a first region 52 in the prepackage gap in which the first maximum electric field strength can be determined because it is spaced in the xy direction from each edge or boundary of the multilayer planar carrier substrate or the power semiconductor prepackage. Here it is shown that the first region 52 is spaced in the x direction by a first distance 54 from a prepackage edge 123e. Similarly, the first region 52 is spaced from another prepackage edge 123e in the y direction (not visible here since the y direction is perpendicular to the drawing plane). Such an offset of the first region 52 in the xy direction towards the inside of the prepackage gap ensures that the maximum field strength is determined over a suitably homogeneous region of the electric field. If the maximum electric field strength is determined through modeling (e.g. finite element analysis, FEA), the maximum electric field strength can be determined at a location that is at least three mesh cells away from a singularity in the model (e.g. peaks , sharp edges, triple points).

Table 17 contains exemplary values for the size G ₁ of the prepackage gap in the z direction as well as for the maximum electric field strength E ₁ in the prepackage gap. Table 17 Peak rated power (kW) G ₁ (µm) E ₁ (kV/mm) example 1 Example 2 Example 3 example 1 Example 2 Example 3 50 20 100 250 1 16 40 100 20 100 250 1 16 40 150 20 100 250 1 16 40 200 20 100 250 1 16 40 400 20 100 250 1 16 40

The second surface 111b of the multilayer carrier substrate 11 and a heat sink 15 are arranged essentially parallel to one another in the z direction, so that a heat sink gap is created, which is in is labeled G2. A second electric field 56 is built up in the heat sink gap G2, which results from the potential difference (voltage) between the second surface 111b of the multilayer carrier substrate 11 and the heat sink.

G₂ = Größe des Kühlkörper-Spalts in z-Richtung
E₂ = maximale elektrische Feldstärke im Kühlkörper-SpaltA power electronic converter can be characterized by a converter parameter φ, which is defined as the size of the heat sink gap in the z direction divided by a maximum electric field strength in the heat sink gap (referred to here as the second maximum electric field strength): $$\mathrm{\phi }=\frac{{\text{G}}_2}{{\text{<figure-callout id="2" label="E" filenames="DE102022205482A1\_0053.png" state="{{state}}">E</figure-callout>}}_2}$$

G ₂ = size of the heat sink gap in the z direction
E ₂ = maximum electric field strength in the heat sink gap

(nm ² N) Beispiel 1 Beispiel 2 Beispiel 3 50 0.05 1.00 15 100 0.05 1.00 15 150 0.05 1.00 15 200 0.05 1.00 15 400 0.05 1.00 15 Table 18 shows exemplary values for the converter parameter φ. The values of φ are characteristically lower than in the prior art and are associated with a high power density. The values of φ are given in units of nm ² /V (nano-m ² /V, or x10 ^-9 m ² /V). Table 18 Peak rated power (kW) $\mathrm{\phi }=\frac{{\text{G}}_2}{{\text{<figure-callout id="2" label="E" filenames="DE102022205482A1\_0053.png" state="{{state}}">E</figure-callout>}}_2}$

( ^nm2N ) example 1 Example 2 Example 3 50 0.05 1.00 15 100 0.05 1.00 15 150 0.05 1.00 15 200 0.05 1.00 15 400 0.05 1.00 15

In general, power electronic converters within the meaning of the present disclosure have values of φ less than or equal to 20 nm ² /V. The value of φ is generally greater than or equal to 0.01 nm ² /V, as lower values e.g. B. may be associated with a greater risk of electrical shock. The values of φ may preferably be in the range of 0.05 nm ² /V to 5 nm ² /V as this represents a good balance between power density and reliability.

As with the first maximum electric field strength E _1, the second maximum electric field strength E ₂ is also a maximum homogeneous electric field strength, so that singularities and other highly localized maxima are excluded. An example is in 2 B a second area 58 for determining a second maximum electric field strength of the second electric field 56 is shown. The second region 58 is located within the heat sink gap G2 and is spaced from all obstacles and edges within the heat sink gap G2, such as those connected to the prepackages 12. If the maximum electric field strength is determined through modeling (e.g. finite element analysis, FEA), the maximum electric field strength can be determined at a location that is at least three mesh cells away from a singularity in the model.

Table 19 contains exemplary values for the size G ₂ of the heat sink gap between the multilayer planar carrier substrate 11 and the heat sink 15 in the z-direction and the maximum electric field strength E ₂ in the heat sink gap. Table 19 Peak rated power (kW) G2 ₍ mm) E2 ₍ kV/mm) example 1 Example 2 Example 3 example 1 Example 2 Example 3 50 0.5 1.5 3.0 0.2 1.5 10 100 0.5 1.5 3.0 0.2 1.5 10 150 0.5 1.5 3.0 0.2 1.5 10 200 0.5 1.5 3.0 0.2 1.5 10 400 0.5 1.5 3.0 0.2 1.5 10

The and show the smallest cuboid volume that encloses the commutation cell of a power electronic converter. Here too, the example of a single-phase two-stage converter is chosen to make the explanation easier. Reference is made to equation 1, which defines the inductance volume parameter α in the form of the smallest cuboid volume that encloses the commutation cell.

it's also in illustrated cross-sectional view. The x and z directions are indicated. The dashed box labeled A _xz is a cross section in the xz plane through the smallest volume that encloses the commutation cell of the converter 10. The cross-sectional area A _xz includes both the power circuit (ie the power semiconductors 121 formed in the prepackages 12, the capacitor 14, the DC inputs and the AC outputs and the connections between them) as well as the gate driver circuit 13 and its connections to the power semiconductors 121. The area A _xz can be the product of the size of the housing (L _x ) measured in the x direction and the size of the housing (L _z ) measured in the z direction be calculated.

is a top view of the converter . The x and z directions are indicated. The dashed box labeled A _xy is a cross section in the xy plane through the smallest cuboid volume that encloses the commutation cell of the converter 10. The cross-sectional area A _xy includes both the power circuit (ie, the power semiconductors 121, the capacitor, the DC inputs and the AC outputs, and the connections therebetween) as well as the gate driver circuit 13 and its connections to the power semiconductors. The area A _xz can be calculated as the product of the size of the housing (L _x ) measured in the x direction and the size of the housing (L _y ) measured in the y direction.

It is easy to see that the smallest cuboid volume that encloses the commutation cell can be calculated as the product of the three dimensions L _x , L _x and L _z .

It is noted that all components of the commutation cell in 3B are visible, although some of the in the cross-sectional view of 3A Components shown are located on the underside of the carrier substrate 11. It should be noted that this is for illustrative purposes only and that in practice not all components of the converter 10 are 3A-3B are visible in a single top view. It should also be noted that the xy position of the gate driver circuit 13 in 3B has been moved for better illustration (ie so that it does not overlap with the xy positions of the prepackages 12, as in 3A the case is). However, flexibility in the xy position of the commutation cell components is an advantageous feature of the transducers according to the present disclosure; this is partly because changes in xy positions do not significantly change the parasitic inductance of the connections between components.

It is also noted that in only two power semiconductor prepackages 12L and 12H can be seen while it is in There are six prepackages 12L _1-3 and 12H _1-3 . This serves to illustrate the parallelization of the power semiconductor switching elements 121. In particular, It can be seen that the low side of the half bridge includes three power semiconductor prepackages, 12L ₁ , 12L ₂ and 12L ₃ , which contain three corresponding power semiconductor switching elements connected in parallel. Likewise, the high side of the half bridge includes three power semiconductor prepackages, 12H _{1 ,} 12H ₂ and 12H ₃ , which contain three corresponding power semiconductor switching elements connected in parallel. In , in which the y-direction is obscured, only one of the three prepackages 12L, 12H on the low and high sides is visible.

shows a single power semiconductor prepackage 12 and shows details that are in the and are not visible.

The power semiconductor switching element 121, which in this example is a MOSFET in the form of a semiconductor chip, is made of solid insulating material 122, e.g. B. FR4, embedded. The electrical connections 124, 125i, which can be filled holes, plated-through holes or the like, extend in the z direction from the connections of the semiconductor switching element 121 to the electrical connection side 123a of the prepackage 12, where they terminate in a flat surface 123a. Although vertical connections 124, 125i are shown, the connections could also have a component in the xy plane.

The MOSFET 121 has at least three terminals, namely the source, drain and gate terminals. A first electrical connection 124 extends from the source terminal to the flat electrical connection surface 123a. In this example, the drain and gate terminals are electrically connected by an electrically conductive metallization layer 125ii on the bottom of the MOSFET chip 121. A second electrical connection 125i extends from the conductive layer 125ii to the flat electrical connection surface 123a. In other examples, the gate and drain terminals are not connected to each other and the terminals 125i, 125ii are related, for example. B. only on the drain connection, with the gate terminal being served by a separate connection from the gate terminal to the flat electrical connection surface 123a.

The illustrated prepackage 12 also includes an optional electrical insulating layer (EIL) 126. The purpose of the EIL 126, which in this example is a layer of ceramic material, is to electrically isolate the MOSFET 121 and its terminals from the heat sink 15 isolate, the is arranged on the underside of the prepackage 12 (see . In other embodiments, the EIL 126 may be omitted from the prepackage 12, for example if a separate EIL is provided between the bottom of the prepackage 12 and the heat sink 15 or if the TIL 16 can provide sufficient electrical insulation. If present, the EIL 126 preferably has a thickness of less than 5 mm, with thicknesses of less than 1 mm being preferred to keep the transducer compact while still achieving the isolation function. Lower thicknesses, e.g. B. less than 0.1 mm can be used if e.g. B. an organic substrate such as IMS is used.

The prepackage 12 shown also includes an optional metal layer 127 on the bottom of the EIL 126. The metal layer 127 improves thermal conduction between the prepackage 12 and the TIL 16. The metal layer 127 can also form a suitable material interface between the bottom of the prepackage and the TIL 16 . If the TIL 16 is, for example, a solder layer, it may be necessary for the underside 123b of the prepackage 12 to carry a material suitable for a solder connection. The metal layer 127 can be omitted if e.g. B. a different TIL is used as solder.

Prepackage 12 shows , which is embedded between a multilayer carrier substrate 11 and a heat sink 15. It further illustrates how the electrical connections 124, 125ii, which extend from the connections of the power semiconductor 121 to the electrical connection surface 123a, can be connected to the conductive elements of the multilayer carrier substrate 11 and thus to other components of the commutation cell.

In the example shown, a planar surface 111b of the multilayer planar carrier substrate 11 includes one or more regions that carry an outer conductive layer 1121. These areas 1121 enable the formation of soldered or sintered connections 113 for connecting the substrate 11 to the distal ends of the electrical connections 124, 125i of the prepackage 12. In the example shown, in addition to the distal ends of the electrical connections 124, 125i, there are also metallization areas 1241 (e.g B. conductive contact surfaces such as solder pads) are provided to facilitate the formation of soldered, sintered or adhesive connections. In other examples, these may be omitted and solder connections 113 may be made directly to the exposed distal ends of terminals 124, 125ii.

Since the connections of the power semiconductor switching elements 121 are now connected to the conductive areas 1121 of the carrier substrate 11, the connections to the other components of the commutation cell are made by a connection to the conductive areas 1121. These connections can be achieved through a combination of conductive layers 112 of the carrier substrate 11 (see 2A) and electrical connections 114 are formed, which extend in the z direction from the conductive areas 1121 through the carrier substrate. For example, layer 1121, which is soldered to terminal 124, which is connected to the source terminal of MOSFET 121, may be connected to one or more (e.g., many) connections 114a extending in the z direction inner conductive layer 112 extend. The inner conductive layer 112 can then be connected to a connection of the intermediate circuit capacitor 14. As another example, layer 1121, which is soldered to terminal 125i, which is connected to the gate terminal of MOSFET 121, may be connected to one or more terminals 114b extending in the z-direction throughout substrate 11 the connections of the gate driver circuit 13 extend.

k = Wärmeleitfähigkeit der thermischen Schnittstellenschicht (TIL)
E_Break = elektrische Durchschlagsfeldstärke der elektrischen Isolierschicht (EIL)The power electronic converters described here can be characterized by a converter parameter ρ, which is defined as follows: $$\mathrm{\rho }=\text{k}\times {\text{E}}_{\text{Break}}$$

k = thermal conductivity of the thermal interface layer (TIL)
E _Break = electric breakdown field strength of the electrical insulating layer (EIL)

Table 20 shows exemplary values for the converter parameter ρ. The converters described here can have characteristically high values for ρ, which can be associated with a combination of good heat dissipation from the prepackages and good resistance to electrical breakdown. The values of ρ are given in units of MVW/m ² K (mega-VW/m ₂ K, or ×10 ⁶ VW/m ² K). Table 20 Peak rated power (kW) ρ = k × E _Break (MVW/m ² K) example 1 Example 2 Example 3 Example 4 50 10 70 4.5×10 ³ 22.5×10 ³ 100 10 70 4.5×10 ³ 22.5×10 ³ 150 10 70 4.5×10 ³ 22.5×10 ³ 200 10 70 4.5×10 ³ 22.5×10 ³ 400 10 70 4.5×10 ³ 22.5×10 ³

In general, power electronic converters according to the present disclosure have values of ρ greater than or equal to 5 MVW/m ² K, with values greater than 20 MVW/m ² K being preferred.

Table 21 contains exemplary values for the electric breakdown field strength (which is also referred to as dielectric strength in the literature), E _Break , the electrical insulating layer (EIL). Example values for the thermal conductivity k of the thermal interface layer can be found in Table 15. Table 21 Peak rated power (kW) E _Break (kV/mm) example 1 Example 2 Example 3 Example 4 50 10 20 50 250 100 10 20 50 250 150 10 20 50 250 200 10 20 50 250 400 10 20 50 250

Higher values for E _Break , e.g. B. in Example 4 apply to EIL containing organic materials, while the lower values, e.g. B. in Examples 1, 2 and 3, EIL that contain inorganic materials.

As already mentioned, the EIL 126 is optional. In alternative embodiments, the EIL 126 is omitted and a TIL 16 with suitable electrical insulation properties is provided. So the TIL 16 can provide electrical insulation between the prepackages and the heatsink as well as a good thermal path between the prepackages and the heatsink.

k = Wärmeleitfähigkeit der thermischen Schnittstellenschicht (TIL)
P = elektrische Leitfähigkeit der TILSuch embodiments can be characterized by a TIL parameter λ, which is defined as the thermal conductivity of the TIL divided by the electrical conductivity of the TIL: $$\mathrm{\lambda }=\frac{\text{k}}{\text{P}}$$

k = thermal conductivity of the thermal interface layer (TIL)
P = electrical conductivity of the TIL

Beispiel 1 Beispiel 2 Beispiel 3 50 1 350 90×10³ 100 1 350 90×10³ 150 1 350 90×10³ 200 1 350 90×10³ 400 1 350 90×10³ Table 21 shows example values for the parameter λ. The values of λ are characteristically high. The values of λ are given in units of TW/SK (Tera-W/SK, or ×10 ¹² W/SK). Table 22 Peak rated power (kW) $\mathrm{\lambda }=\frac{\text{k}}{\text{P}}\left(\text{TW/SK}\right)$

example 1 Example 2 Example 3 50 1 350 90×10 ³ 100 1 350 90×10 ³ 150 1 350 90×10 ³ 200 1 350 90×10 ³ 400 1 350 90×10 ³

In general, power electronic converters according to the present disclosure have values of λ greater than or equal to 1 TW/SK, with values greater than 100 TW/SK being preferred.

Table 23 contains exemplary values for the electrical conductivity P of the thermal interface layer. Table 23 Peak rated power (kW) P (S/m) example 1 Example 2 Example 3 50 1×10 ^-15 1×10 ^-14 1×10 ^-13 100 1×10 ^-15 1×10 ^-14 1×10 ^-13 150 1×10 ^-15 1×10 ^-14 1×10 ^-13 200 1×10 ^-15 1×10 ^-14 1×10 ^-13 400 1×10 ^-15 1×10 ^-14 1×10 ^-13

shows a further arrangement of a power electronic converter 10, in which at least part of the prepackage gap G1 is covered with electrically insulating material 60, e.g. B. a resin is filled. The electrically insulating material 60 is arranged in and optionally around the prepackage gap G1. “Um” in this context means that the electrically insulating material 60 extends beyond the prepackage 12 in the xy direction. For example, the material 60 may additionally cover the metal layers 1121 extending from the terminals 114a to the terminals 113.

With the help of the electrically insulating material 60, both a creepage distance 62 and an air gap distance 64 can be reduced and kept physically small. With the help of the electrically insulating material 60, small distances between the components of the power electronic converter 10, for example between the metal layers 1121 and the connections 114a, can be achieved without there being any significant risk of adverse electrical effects such as sparking or leakage currents. This is particularly advantageous considering the high voltages used in the transducers of the present disclosure and the excessive potential differences e.g. B. between the metal layers 1121 and the surfaces of the prepackage 12 and the heat sink 15 can lead.

F = Isolationsfüllfaktor
R_max = Maximale Hohlraumgröße in der IsolierungIn general, the electrically insulating material 60 used contains voids (e.g., air) in its volume due to imperfections in the manufacturing process. A power electronic converter that uses electrically insulating material in a prepackage gap can be characterized by a converter parameter σ, which is defined as the insulation fill factor divided by a maximum size of the cavities: $$\sigma =\frac{F}{R_{max}}$$

F = insulation filling factor
R _max = Maximum cavity size in the insulation

V_IM = Volumen des elektrisch isolierenden Materials (einschließlich Hohlräume)
V_V = Kumuliertes Volumen der HohlräumeIn this equation, F is the insulation fill factor and R _max is the maximum cavity size of the plurality of cavities. The insulation fill factor is defined as the cumulative volume of the plurality of cavities (the “Void Volume”) subtracted from a volume of electrically insulating material divided by the volume of electrically insulating material. This can be expressed as follows: $$\text{F}=\frac{{\text{v}}_{\text{IN THE}}-{\text{v}}_{\text{v}}}{{\text{v}}_{\text{IN THE}}}$$

V _IM = volume of electrically insulating material (including voids)
V _V = Cumulative volume of voids

Therefore, the converter parameter σ can also be expressed as follows: $$\sigma =\frac{{\text{v}}_{\text{IN THE}}-{\text{v}}_{\text{v}}}{{\text{v}}_{\text{v}}\times {\text{R}}_{\text{Max}}}$$

(1/mm) Beispiel 1 Beispiel 2 Beispiel 3 50 18 100 1000 100 18 100 1000 150 18 100 1000 200 18 100 1000 400 18 100 1000 Table 24 shows exemplary values for the transducer parameter σ, expressed in the unit 1/mm. Table 24 Peak rated power (kW) F $\mathrm{\sigma }=\frac{\text{F}}{{\text{R}}_{\text{Max}}}$

(1 mm) example 1 Example 2 Example 3 50 18 100 1000 100 18 100 1000 150 18 100 1000 200 18 100 1000 400 18 100 1000

Transducers according to the present disclosure preferably have σ values greater than or equal to 10/mm to ensure good electrical insulation properties. However, values greater than or equal to 50/mm may be preferred, particularly at higher operating voltages.

Table 26 shows exemplary values for the insulation filling factor F and the maximum cavity size R _max Table 25 Peak rated power (kW) F (%) R _max (µm) example 1 Example 2 Example 3 example 1 Example 2 Example 3 50 90 99 99.99 1 10 50 100 90 99 99.99 1 10 50 150 90 99 99.99 1 10 50 200 90 99 99.99 1 10 50 400 90 99 99.99 1 10 50

The values of R _max can be determined by an equivalent spherical method in which void size measurements are made for a representative sample of the electrically insulating material and a maximum void size is statistically estimated, assuming that the voids are spherical and the measured sizes are diameters of balls are.

D = Durchschlagfestigkeit des elektrisch isolierenden Materials
R_max = Maximale Hohlraumgröße in der IsolierungA power electronic converter that uses electrically insulating material in a prepackage gap may also be characterized by a converter parameter τ, which is defined as the product of the dielectric strength of the electrically insulating material and the maximum cavity size: $$\mathrm{\tau }=\text{D}\times {\text{R}}_{\text{Max}}$$

D = dielectric strength of the electrically insulating material
R _max = Maximum cavity size in the insulation

Table 26 shows exemplary values for the converter parameter τ, expressed in units of volts. Table 26 Peak rated power (kW) τ = D × R _max (V) example 1 Example 2 Example 3 50 1 200 10,000 100 1 200 10,000 150 1 200 10,000 200 1 200 10,000 400 1 200 10,000

Converters according to the present disclosure preferably have τ values of at most 1,000 V to ensure good electrical insulation properties. However, values less than or equal to 100 V may be preferred, particularly at higher operating voltages.

Table 27 contains exemplary values for the dielectric strength D of the electrically insulating material. Table 27 Peak rated power (kW) D (kV/mm) example 1 Example 2 Example 3 50 1 20 200 100 1 20 200 150 1 20 200 200 1 20 200 400 1 20 200

shows how low-side and high-side power semiconductor switching elements 121L, 121H of the prepackages 12L, 12H can be electrically connected using the multilayer carrier substrate 11. The x and z directions are indicated.

As in 4B A planar surface 111b of the substrate 11 has regions 1121i, 1121ii, 1121iii which carry an outer conductive layer. These areas 1121i-iii facilitate connection to the connections of the power semiconductors 121L, 121H by soldering or sintering. The multilayer substrate 11 is further shown having an inner conductive layer 112a electrically connected to one of the outer layer regions 1121ii via a series of connections 114a extending through the substrate 11 in the z-direction.

The source terminal (S) of the high-side power semiconductor switching element 121H is electrically connected to the high-side direct current (DC+) input via a terminal (e.g., a soldered, sintered, or bonded connection) to the third outer layer region 1121iii . The drain terminal (D) of the low-side power semiconductor switching element 121L is electrically connected to the low-side direct current (DC-) input via a terminal to the first outer layer region 1121i. The drain connection (D) of the High-side power semiconductor switching element 121H and the source terminal (S) of low-side power semiconductor switching element 121L are electrically connected to each other and to the inner conductive layer 112 of the substrate through connections to the second outer layer region 1121ii.

The inner layer 112 is thicker in the z direction than the outer layer regions 1121i-1121iii. This greater thickness reduces the resistance and thus increases the current carrying capacity of the inner conductive layer 112. This reflects the fact that in this example the inner conductive layer 112 carries a high current while the outer layer regions 1121i-iii act as electrical contact rather than as Paths for power transmission between components are used. The thin outer layer regions 1121i-1121iii may have a thickness of less than 100 µm (e.g. 50 µm), while the thicker inner layer 114 may have a thickness of greater than 100 µm (e.g. 100-400 µm). . A variety of vias (e.g. five, ten or more) can be used to connect a thin outer contact area and the thick inner layer. By increasing the number of vias for an electrical path, the current carrying capacity can be increased accordingly.

shows how power semiconductor switching elements P1 and P2 can be connected in parallel. P1 and P2 can, for example, be two low-side power semiconductors in one phase of a two-stage AC-DC converter. The y and z directions are for comparison with specified.

As in 5A A planar surface 111b of the substrate 11 has regions 1121iv, 1121v, 1121vi which carry an outer conductive layer. These areas 1121iv, 1121v, 1121vi facilitate the connection to the connections of the power semiconductors P1 and P2, e.g. B. by soldering, sintering or gluing. The multilayer substrate 11 also contains two inner conductive layers 112b, 112c. Each of the inner layers 112b, 112c is electrically connected to one of the outer layer regions 1121v, 1121vi through a corresponding set of connections 114b, 114c extending through the substrate 11 in the z-direction. As in 5A the inner layers 112b, 112c are thicker than the outer layer regions 1121iv-vi.

The source terminal (S) of the first power semiconductor switching element P1 is electrically connected to the AC side of the converter 10 via a terminal to the first outer layer region 1121iv. The source terminal (S) of the second power semiconductor switching element P2 is electrically connected to the AC side of the converter 10 via a terminal to the third outer layer region 1121vi, which is connected to the inner conductive layer 112c of the drive via the terminal 114c. The drain terminal (D) of the first power semiconductor switching element P1 and the drain terminal (D) of the second power semiconductor switching element P2 are connected to each other and to the inner conductive layer 112b of the substrate 11 through soldered, sintered or bonded connections to the second outer one Layer area 1121 v electrically connected.

• - Die verschiedenen Innenschichten 112a-c so dargestellt sind, dass sie die gleiche Dicke haben und sich in der gleichen Tiefe durch das Substrat 11 erstrecken. Dies muss nicht der Fall sein und wird im Allgemeinen auch nicht der Fall sein. Die inneren Schichten können unterschiedliche Dicken haben und in z-Richtung und/oder x-y-Richtung versetzt sein.
• - Die Außenschichten 1121 i-vi und die Innenschichten 112a-c gleich dick oder unterschiedlich dick sein können. Die geeigneten Dicken hängen bis zu einem gewissen Grad von den Anforderungen der Anwendung ab (z. B. Leistung). Es ist im Allgemeinen einfacher, dicke Innenschichten 112a-c z. B. durch bekannte Leiterplattenherstellungsverfahren herzustellen als dicke Außenschichten, die z. B. durch Abscheidung erzeugt werden können.

It goes without saying that the connection arrangements of the 5A-5B are just an example and that the connections can be made in different ways, with different combinations of outer layers, inner layers and z-direction connections. It should further be understood that:

• - The various inner layers 112a-c are shown as having the same thickness and extending through the substrate 11 to the same depth. This does not have to be the case and generally will not be the case. The inner layers can have different thicknesses and be offset in the z direction and/or xy direction.
• - The outer layers 1121i-vi and the inner layers 112a-c can be the same thickness or different thicknesses. The appropriate thicknesses depend to some extent on the requirements of the application (e.g. performance). It is generally easier to use thick inner layers 112a-c, e.g. B. produced by known circuit board manufacturing processes as thick outer layers, e.g. B. can be generated by deposition.

In each of the examples described above, the AC-DC converter 10 has only a single phase. However, this is for purposes of illustration and explanation only, and AC-DC converters according to the present disclosure can and generally will have multiple phases. To this end, they illustrate , how the concepts described above can be extended to multiple phases. The show a two-stage, three-phase AC-DC converter in which the low side and the high side of each phase comprise several (in this case eight) power semiconductor switching elements connected in parallel.

is a schematic cross section of the transducer 10. The x-direction and the z-direction are indicated. The converter 10 has three phases, designated U, V and W. Each phase has its own set of power semiconductor prepackages: the first phase U has prepackages 12U-L and 12U-H; the second phase V has prepackages 12V-L and 12V-H; and the third phase W has prepackages 12W-L and 12W-H. The prepackages 12 of each phase are surface mounted and electrically connected to a common multilayer planar support substrate 11, which may be a printed circuit board as before. The heat dissipation sides of the prepackages 12 face a heat sink 15, with thermal interface layers 16U, 16V, 16W being located between the undersides of the prepackages 12U-W and the heat sink 15.

For better illustration, the other components of the commutation cell, namely the gate driver circuit 13, the capacitor(s) 14 and the electrical connections between these components are not shown. However, it is clear that these components and their compounds are essentially as described above with reference to 1-5 are described.

In this example there is a common heatsink 15 powering the entire converter 10, but it goes without saying that there could instead be a separate heatsink for each phase U, V, W, similar to that in shown arrangement for a single phase. In addition, in this example, the prepackages sit in a recess 151 in the heat sink 15. The use of a recess can advantageously reduce the thickness of the converter in the z direction and also create a secondary heat conduction path between the prepackages 12 and the heat sink 15, which passes through the substrate 11 runs. In addition, provided the recess is suitably sealed, a liquid or gaseous cooling medium can be passed through the recess to directly cool the prepackages 12.

In the example shown, the heat sink 15 and the substrate 11 are fastened and pressed together using fasteners 17. This is not strictly necessary, but the use of fasteners to press the substrate 11 and prepackages 12 to the heat sink 15 may be preferable to ensure a better thermal interface to the heat sink 15.

The heat sink has barrier walls 152 that divide the recess 151 into three chambers (one for each phase) to provide isolation between phases. This can be useful for error limitation, but can also be omitted in other examples. The barrier walls 152 may also not be integrated into the heat sink 15, although integrated barrier walls may improve the quality of the secondary heat conduction path between the prepackages 12 and the heat sink 15. If the barrier walls 152 are omitted, a TIL 16 can be used that extends over the prepackages of several (e.g. all) phases U, V, W of the converter 10.

is a schematic top view of the converter 10 . Only the prepackages 12 are shown, and as in 3B the prepackages 12 are shown, although they are located on the underside of the substrate and are covered by the heat sink 15. Since the y-direction is no longer obscured, it can be seen that the low side and the high side of each phase U, V, W comprises eight power semiconductor switching elements, each of which is housed in a power semiconductor prepackage. For example, the eight prepackages 12U-L ₁ to 12U-L ₈ are labeled.

Experts will understand that the example of the 6A-B can be expanded to any number of phases and to any number of power semiconductor switching elements connected in parallel. It also becomes clear that it can be extended to DC/DC converter circuits, which may include only a single power semiconductor switching element or a set of power semiconductor switching elements connected in parallel.

In general, any power electronic converter can be understood as consisting of one or more “logic switches,” each of which includes one or more power semiconductor switching elements connected in parallel. In the case of a two-stage AC-DC converter, there are two logical switches per phase (one low-side and one high-side switch). In a DC-DC converter there can also only be one logical switch (see e.g. , while other DC-DC converters may include multiple logical switches (see e.g. , which has four logical switches; two per side of the transformer 250'). Converters according to the present disclosure may include any number of power semiconductors (and thus prepackages) per logic switch, with 3 to 10 prepackages per logic switch being typical.

It is worth considering how a change in the number of phases affects the definition of the volume of the commutation cell and the parasitic inductance of the power circuit. Each phase forms part of the commutation cell. The smallest cuboid volume that encloses the commutation cell therefore encloses each phase of the converter. However, each phase circuit is substantially independent of the other phase circuits, with its switching and conduction between its DC and AC sides being independent of the other phase circuits. Therefore, from the parasitic inductance point of view, a multi-phase power circuit can be considered as equivalent to several independent single-phase power circuits, and the parasitic inductance of the power circuit is therefore equal to the parasitic inductance of one of the phases. The parasitic inductance of each phase is the same (apart from small unavoidable variations due, for example, to the manufacturing tolerance of the components and the quality of the electrical contacts), so it does not matter which phase is chosen. In principle it is possible to design a converter where each phase has a different parasitic inductance, but this would be undesirable.

10 fFs/W 10 fFs/W $\mathrm{\delta }=\frac{{\text{V}}_{\text{block}}}{\text{L}\times \text{C}}$

6 PV/FH 7 PV/FH $\mathrm{\epsilon }=\frac{{\text{f}}_{\text{max}}\times {\left|\frac{\text{dv}}{\text{dt}}\right|}_{\text{max}}}{\text{L}\times \text{C}}$

7,5 ×10²⁷ V/s⁴ 8,8 ×10²⁷ V/s⁴ $\mathrm{\theta }=\frac{{\text{G}}_1}{{\text{E}}_1}$

18.25 Uhr /V² 18.25 Uhr /V² $\mathrm{\phi }=\frac{{\text{G}}_2}{{\text{E}}_2}$

1 nm /V² 1 nm /V² $\mathrm{\sigma }=\frac{\text{F}}{{\text{R}}_{\text{max}}}$

80/mm 100/mm τ = K × R _max 20 V 20 V ρ = k × E _Break 25 GV/m K² 25 GV/m K² $\mathrm{\eta }=\frac{\text{h}}{{\text{G}}_3}$

0,3 MW/m K³ 0,7 MW/m K³ $\mathrm{\Omega }=\frac{\text{M}}{\text{K}}$

0,8 MNK/Wm 0,8 MNK/Wm Table 28 contains, as a specific example, the specifications of two converters according to the present disclosure. Both are two-stage, three-phase AC-DC converters, but this should not be seen as a limitation. Table 28 Example 1 (100 kW, 2-stage, 3-phase AC-DC) Example 2 (200 kW, 2-stage, 3-phase AC-DC) Substrate type Rigid multi-layer PCB Rigid multi-layer PCB Power semiconductor type SiC MOSFET SiC MOSFET Prepackage type FR4 insulation with integrated ceramic EIL FR4 insulation with integrated ceramic EIL Heat sink type Air-cooled aluminum Liquid-cooled aluminum TIL type Indium tin solder Indium tin foil Top performance 100kW 200kW MOSFET source-drain reverse voltage 1,200V 1,400V Rated peak current 200A 350A Maximum switching frequency 50kHz 50KHz Maximum source-drain voltage ramp rate [dv/dt] 30 kV/µs 35 kV/µs Parasitic inductance of the power circuit [L] 4 nH 2 nH Total capacity of the power circuit [C, @298K, 1,000V DC] 50µF 100µF Volume of the commutation cell [smallest cuboid] ^{300cm3 424cm3} Number of prepackages per logical switch 6 12 Total number of prepackages 36 72 Gap between substrate and prepackages 100 µm 120µm Gap between substrate and heat sink 1.5mm 1.6mm TIL strength 200 µm 150 µm EIL thickness 0.25mm 0.25mm TIL thermal conductivity 3.5W/mK 2.5W/mK Efficiency 99% 99% α = L × Vol 1.2 ^pHm3 0.85 ^pHm3 β = f _max × |dv/dt| _Max 1.5PV/s ² 1.75 PV/s ² $\mathrm{\gamma }=\frac{\text{C}}{\text{P}\times {\text{f}}_{\text{Max}}}$

10 fFs/W 10 fFs/W $\mathrm{\delta }=\frac{{\text{v}}_{\text{block}}}{\text{L}\times \text{C}}$

6 PV/FH 7 PV/FH $\mathrm{\epsilon }=\frac{{\text{f}}_{\text{Max}}\times {\left|\frac{\text{dv}}{\text{German}}\right|}_{\text{Max}}}{\text{L}\times \text{C}}$

7.5×10 ^27V /s ⁴ 8.8×10 ^27V /s ⁴ $\mathrm{\theta }=\frac{{\text{G}}_1}{{\text{<figure-callout id="1" label="E" filenames="DE102022205482A1\_0053.png" state="{{state}}">E</figure-callout>}}_1}$

6.25 p.m. /V ² 6.25 p.m. /V ² $\mathrm{\phi }=\frac{{\text{G}}_2}{{\text{<figure-callout id="2" label="E" filenames="DE102022205482A1\_0053.png" state="{{state}}">E</figure-callout>}}_2}$

1nm/ ^V2 1nm/ ^V2 $\mathrm{\sigma }=\frac{\text{F}}{{\text{R}}_{\text{Max}}}$

80/mm 100/mm τ = K × R _max 20V 20V ρ = k × E _break 25 GV/m ^K2 25 GV/m ^K2 $\mathrm{\eta }=\frac{\text{H}}{{\text{G}}_3}$

0.3 MW/m ^K3 0.7 MW/m ^K3 $\mathrm{\Omega }=\frac{\text{M}}{\text{K}}$

0.8 MNK/Wm 0.8 MNK/Wm

The 7A-7B and 8A-8B show several alternative ways in which the power semiconductor prepackages 12 can be arranged and connected in relation to the multilayer carrier substrate 11. In any case, the gate driver circuit 13 and capacitor(s) 14 are omitted, but it will be understood that these could be incorporated into the alternative arrangements substantially as described above.

shows the general transducer assembly 10 described above with reference to and described in Table 28. Each power semiconductor prepackage 12 is surface mounted on the underside of the multilayer carrier substrate 11, with the electrical connections between the two taking the form of soldered, sintered or adhesive connections 113. The other side of the prepackages 12 faces a common heat sink 15, with a TIL 16 forming the mechanical and thermal interface between the prepackage 12 and the heat sink 15.

shows an alternative converter arrangement 10 ', in which the prepackages 12 and the heat sink 15 are mounted on opposite sides of the carrier substrate 11. As before, the prepackages 12 are surface-mounted on the substrate 11 using soldered or sintered connections 113. The heat sink 15 is mechanically and thermally connected to the substrate 11 via a TIL 16. The length and thermal resistance of the thermal path between the prepackages 12 and the heat sink 15 are increased by mounting the prepackages 12 and the heat sink 15 on opposite sides of the substrate 11. For this reason, the arrangement in the one in be preferred. However, the addition of thermally conductive elements 115 integrated into the supporting substrate can provide a useful and sufficient reduction in thermal resistance. The heat-conducting elements could be, for example: B. be copper vias.

The 8A and 8B show further alternative transducer arrangements 10'', 10''', in which the prepackages 12 are embedded in the multilayer carrier substrate 11. Embedding the prepackages 12 can offer several advantages. First, this eliminates some air gaps in the transducer that might otherwise be susceptible to electrical breakdowns due to the high voltages and electrical fields that may be used in transducers of the present disclosure. Secondly, it is possible to make the converter somewhat more compact in the z direction, as some gaps are eliminated. Thirdly, fewer connections need to be made between the prepackages and the conductive elements of the substrates 11, e.g. B. soldered or sintered connections, which somewhat reduces the complexity of production and eliminates a source of errors. The heat dissipation from the arrangements 10'', 10''' of 8A and 8B however, is more difficult.

differs from in that the prepackages 12 are completely embedded in the substrate 11 and are therefore completely surrounded by insulating material. In the underside of the prepackage 12 is exposed and flush with one of the planar surfaces of the substrate 11. The complete embedding of the prepackages 12 in the substrate 11 leads to a further improvement in the electrical insulation of the power semiconductors 121, but also increases the thermal resistance between the prepackages 12 and the heat sink 15 further. In Heat-conducting elements 115 are integrated into the substrate 11 to reduce the thermal resistance of the heat-conducting path.

In each of the examples described above, the power electronic converters 10, 10', 10'', 10''' have taken the form of a two-stage AC-DC converter. Those skilled in the art will understand that the concepts described can also be applied to other types of power electronic converters, including alternative AC-DC converter topologies (including multi-level converter topologies) and DC-DC converters. The show various AC-DC and power electronic DC-DC converter circuits that can be used in accordance with the embodiments described above.

shows a three-phase electrical machine 140 connected to a two-stage, three-phase AC-DC converter 100. The electric machine 140 is configured as a motor and the converter 100 as an inverter, but it will be understood that the arrangement of a generator and rectifier would be very similar.

In this example, one end of each phase winding 140u-w of the motor 140 is connected to a common point (the star or Y point), although a delta connection of the windings 140u-w could also be used. The other end of each winding is connected to a corresponding phase leg 110u-w of the inverter 100 at a phase connection point. Each phase branch 110u-w is also connected to a high and a low DC input DC-H, DC-L, which are connected, for example, to a DC bus such as the DC bus 330 in 12A can be connected. Each phase leg includes a high and a low side transistor 112H, 112L and associated parallel diodes 112H-d, 112L-d connected between the high and low side DC inputs DC-H, DC-L.

The inverter 100 further includes a smoothing DC link capacitor 114 connected between the high and low DC inputs DC-H, DC-L, and a gate driver circuit 113 connected to the gate terminals of the transistors 112H, 112L and is configured to deliver switching signals to it. The gate driver circuit 113 generally receives low power control signals from a controller (not shown) and amplifies the low power signals to provide the gate terminals with switching signals used to control the on/off state of the transistors 112H, 112L of the phase legs 110u-w are suitable.

In this example, the transistors 112L, 112H are MOSFETs (e.g., silicon carbide (SiC) MOSFETs). As those skilled in the art know, the parallel diodes 112H-d, 112L-d, which are associated with the MOSFETs 133, 134, are not discrete components, but rather the so-called “body diodes” of the MOSFETs, i.e. H. about the diodes inherent in the MOSFETs. In other examples, diodes 112H-d, 112L-d may be discrete devices separate from transistors 112H, 112L.

During operation, the inverter 100 is supplied with direct electrical voltage via the electrical connections DC-H, DC-L. The gate terminals of the transistors 112H, 112L receive switching signals from the gate driver circuit 113. As will be apparent to those skilled in the art, the switching signals switch the transistors 112L, 112H of each phase leg 110u-w between conducting and non-conducting ("on" and "off") ) States, whereby the current from the upper and lower branches of each phase leg 110u-w is switched into the respective phase winding 140u-w of the motor 140. The times and duration of the switching are controlled so that the phase windings 140u-w of the motor 140 are supplied with alternating electrical voltage via the AC phase connection points.

shows another inverter circuit 100'. In this example, a four-phase electric motor 140' is supplied with electrical energy from a two-stage direct current supply via an AC-DC power electronic converter 100', which includes four independent H-bridge circuits. For the sake of clarity, Only two of the four H-bridge circuits 110u', 110v' are shown, which are connected to the corresponding two of the four phases 140u', 140v' of the motor 140'.

Each H-bridge circuit (e.g., H-bridge circuit 110u') includes four transistors 112L-1', 112H-1', 112L-2', 112H-2' and associated parallel diodes arranged in an H-bridge configuration between the high and low DC terminals DC-H, DC-L and one of the phase windings 140u' of the motor 140'. A DC link capacitor 113 is also connected between the DC terminals DC-H, DC-L. During operation, the DC terminals supply DC electrical voltage to the H-bridge circuit 110u', and the gate driver circuit 113 supplies switching signals to the gate terminals of the transistors 112L-1', 112H-1', 112L-2', 112H-2' . Switching the transistors between their conducting and non-conducting states causes a reversal of the DC power supply to AC power supply to power the phase windings 140u' of the motor 140'.

shows a circuit of the inverter 100" with a DC-side filter that is more complex than the individual DC link capacitor of the converters 100, 100' of . The inverter 100" is like the inverter 100 in a two-stage three-phase inverter. However, it contains a DC filter with three capacitors 114-1, 114-2, 114-3 and four inductors 119-1, 119-2, 119-3, 119-4, which are connected between the DC connections DC-L, DC-H and the phase legs 11 0u-w of the converter are connected. Any suitable DC side filter circuit may be used in accordance with the embodiments described herein.

shows a boost type DC-DC converter 200. Buck type and buck-boost type DC converters are also known.

The DC-DC converter 200 is on one of its sides with a battery 220 and on the other side, for example, with a DC power channel such as the DC power channel 330 of 11A or the DC power channel 530 of 14A tied together.

The DC-DC converter 200 contains a transistor 212, the gate terminal (g) of which is connected to a gate driver circuit 213. As in the previous examples, transistor 212 is a MOSFET, and the parallel diode associated with transistor 212 may be an additional discrete diode to the MOSFET's own body diode. The converter circuit 200 further includes a diode 218, a smoothing capacitor 214 (which can also be referred to as an input capacitor in the context of a DC-DC converter) and an inductor 219.

In operation, the DC-DC converter circuit 200 receives direct current, either from the terminals of the battery 220 or from the DC terminals DC-L, DC-H. The gate driver circuit 213 supplies the gate terminal (g) of the transistor with switching signals to control the on/off state of the transistor 212 to effect the desired voltage increase or decrease.

shows another type of DC-DC converter 200'. The DC-DC converter 200' is of the DC-AC-DC type and includes back-to-back inverter 210' and rectifier 230' stages with an intermediate transformer 250'. In this example, both the inverter 210' and the rectifier 230' are of the H-bridge type, although other inverter and rectifier circuits could also be used. Both have associated DC-side filters 214', 234'.

During operation, the DC-DC converter 200' receives direct current from a direct current source, for example the energy storage system 230 or via the DC connections DC-H, DC-L. The gate terminals of the transistors of the H-bridge circuits 210', 230' receive switching signals from the gate driver circuit 213, which causes inversion of the direct current and rectification of the alternating current to supply power to or current from the windings of the transformer 250' to receive them. The direct current that comes from one of the first and second H-bridge circuits 210', 230' is outputted to either the battery 220 or a DC network via the DC connections DC-L, DC-H.

The show exemplary energy and drive systems for aircraft 300, 400, 500 with power electronic converters. The power electronic converters can advantageously be of the efficient, power-dense types described here.

is a schematic representation of an all-electric aircraft propulsion system 300. The system 300 includes an electrical energy storage unit 320, in this case a high-voltage battery, which supplies a DC channel 330 with DC electrical voltage. The DC channel 330 supplies electrical energy to electrical loads, including an electric motor 340 that drives a propulsor 350 (e.g., a propeller or a fan). An inverter 310 converts the direct current from the direct current channel 330 into alternating current to supply the windings of the motor 340. In some embodiments, the inverter 310 may be integrated with the motor 340 (e.g., sharing a common housing structure with it) and/or the motor 340 may be integrated into the propulsor 350.

Although not shown, the electric drive system 300 may optionally include a DC-DC converter connected between the terminals of the battery pack 320 and the DC channel 330 to regulate the voltage on the DC channel. For example, the connection voltage of the battery pack 320 tends to decrease, e.g. B. by a factor of up to about two when the battery pack 320 is discharged from a maximum state of charge to a lower state of charge. For example, the voltage may drop from a maximum voltage level of 900 V to approximately 450 V over the course of the discharge. A DC-DC converter can therefore be used to increase the terminal voltage and maintain a constant voltage on the DC power channel 330. Other arrangements may eliminate the DC-DC converter and compensate for the voltage drop and associated power loss by increasing the current delivered to the loads (e.g., motor 340).

This in Electric power and propulsion system 300 shown includes only a single energy channel (or energy “trace”) and a single propulsor. In practice, a power and propulsion system may include multiple channels and have a more complex configuration. Shows for illustrative purposes an exemplary electric vertical takeoff and landing (eVTOL) aircraft 360 with a distributed propulsion system, in this case including six propulsors 350a-f. In this arrangement, each propulsor 350a-f can be assigned its own power channel, e.g. B. six power channels of the in In another example, one or more of the propulsors 350a-f may share a power channel such that there are fewer than six power channels.

As in shown, four of the propulsors 350a-d are connected to the wings 361 of the aircraft 360, while the remaining two propulsors 350e-f are connected to the rear flight surfaces 362. The wings 361 and the aft flight surfaces 362 tilt between a VTOL configuration (in shown), in which the propulsors 350a-f provide thrust for lift, and a forward flight configuration (in shown), in which at least some of the propulsors (the rear propulsors 350e-f in Deliver forward thrust. It is understood that a different number of propulsors (e.g. four or eight) is also possible, as are other eVTOL configurations (e.g. multicopter designs and variations of the tiltrotor design shown are also known).

shows how six electrically driven propulsors (e.g. the six propulsors 350a-f of the ) can be arranged in an electrical power supply system 300. The power system 300 includes three independent power channels 300a, 300b, 300c, each of which is connected to two of the six propulsors 350a-f. In this particular example, each of the first two power channels 300a, 300b is associated with a front propeller 350b, 350c and a rear propeller 350e, 350f, while the third channel 300c is associated with a front left propeller 350a and a front right propeller 350d. Each propulsor 350a-f is assigned a corresponding motor 340a-f and a corresponding inverter 340a-f. Other power system configurations are possible and will be apparent to those skilled in the art. For example, the system 300 could include connections between the power channels 300a-c and the use of multiple redundant motor windings to enable power sharing between some or all of the power channels 300a-c and increase fault tolerance and power availability.

The use of a distributed drive system as described in the and is shown can be in terms of improved flight characteristics, lower aerodynamic noise and lower air resistance can be very desirable. However, this generally also increases the number of power electronic converters in the electrical power supply system. When using the system in For example, the platform could include nine independent power electronic converters (six inverters and three DC converters), which could result in significant additional weight to the platform and additional losses in the power system. The improvements in efficiency and power density of the converters disclosed here can therefore be particularly advantageous for platforms of this type.

shows a general arrangement of an engine 400 for an aircraft. The engine 400 is designed as a turbofan engine and therefore includes a ducted fan 401 which receives intake air A and generates two pressurized air streams: a bypass stream B which flows axially through a bypass duct 402 and a core stream C which flows in a nuclear gas turbine enters.

The core gas turbine includes, in axial series, a low-pressure compressor 403, a high-pressure compressor 404, a combustion chamber 405, a high-pressure turbine 406 and a low-pressure turbine 407.

In operation, the core stream C is compressed by the low pressure compressor 403 and then passed into the high pressure compressor 404 where further compression takes place. The compressed air exiting the high pressure compressor 404 is directed into the combustion chamber 405 where it is mixed with fuel and the mixture is burned. The resulting hot combustion products expand through the high-pressure turbine 406, driving the low-pressure turbine 407, before being exhausted and providing a small portion of the total thrust.

The high-pressure turbine 406 drives the high-pressure compressor 404 via a connecting shaft. The low-pressure turbine 407 drives the low-pressure compressor 403 via a further connecting shaft. Together, the high-pressure compressor 404, the high-pressure turbine 406 and the associated connecting shaft form part of the high-pressure shaft of the engine 400. Similarly, the low-pressure compressor 403, the low-pressure turbine 407 and the associated connecting shaft form part of the low-pressure shaft of the engine 400. Such terms are known to those skilled in the art . Experts also know that the engine pictured has two shafts, while other gas turbine engines have a different number of shafts, e.g. B. three waves.

The fan 401 is driven by the low-pressure turbine 407 via a reduction gear in the form of a planetary planetary gear 408. In this configuration, the low-pressure turbine 407 is connected to a sun gear of the gear 408. The sun gear is in engagement with a plurality of planet gears arranged in a rotating carrier, which in turn are in engagement with a static ring gear. The rotating carrier drives the fan 401 via a fan shaft 410. It will be appreciated that in alternative embodiments, a star-shaped planetary gear (where the planet carrier is static and the ring gear rotates and provides the output) may be used instead, and that the gear 408 may even be omitted entirely so that the fan 401 is provided directly from the low-pressure turbine 407 is driven.

It is increasingly desirable to enable greater levels of electrical functionality on the airframe and engine. For this purpose, the engine includes 400 in one or more rotating electrical machines, generally capable of operating as both a motor and a generator. The number and arrangement of rotating electrical machines depends to a certain extent on the desired functionality. Some embodiments of the engine 400 include a single rotating electric machine 420 driven by the high pressure shaft, e.g. B. by a core-mounted auxiliary drive 421 of conventional configuration. Such a configuration facilitates the generation of electrical power for the engine and aircraft as well as driving the high pressure shaft to facilitate engine starting in place of an air turbine starter. Other embodiments, including those in shown include both a first rotating electrical machine 420 coupled to the high pressure shaft and a second rotating electrical machine 430 coupled to the low pressure shaft. In addition to generating electrical energy and starting the engine 400, the fact that both the first and second rotating machines 420, 430 are connected by power electronics can facilitate the transfer of mechanical energy between the high and low pressure shafts Improve operability, fuel consumption, etc.

As mentioned above, in the first rotating electric machine 420 is driven by the high pressure shaft through a core-mounted auxiliary drive 421 of conventional configuration. In alternative embodiments, the first electric machine 420 may be mounted coaxially with the turbomachines in the engine 400. For example, the first electric machine 420 may be mounted axially in line with the channel between the low and high pressure compressors 403 and 403. In the second electric machine 430 is mounted coaxially with the turbomachine in the tail cone 409 of the engine 400 and is coupled to the low-pressure turbine 407. In alternative embodiments, the second rotating electric machine 430 may be disposed axially in line with the low pressure compressor 403, which may adopt a paddle disk or paddle drum configuration to accommodate the second rotating electric machine 430. Of course, those skilled in the art know that any other suitable position can be chosen for the first and (if any) second electrical machine.

The first and second electrical machines 420, 430 are connected to power electronics. The removal of energy from the electrical machines or the supply of energy to them is carried out by power electronic converters 440. In the present embodiment, the power electronic converters 440 are attached to the fan housing 411 of the engine 400, but it goes without saying that they are also on can be attached elsewhere, e.g. B. on the core of the gas turbine or in the vehicle to which the engine 400 is attached.

In the present example, the power electronic converter 440 and the first and second electric machines 420 and 430 are controlled by an electronic engine controller (EEC) 450. In the present embodiment, the EEC 450 is a fully digital engine control unit (FADEC), the configuration of which is known to those skilled in the art is understandable. It therefore controls all aspects of the engine 400, i.e. H. both the core gas turbine and the first and second electric machines 420 and 430. In this way, the EEC 450 can respond holistically to both the thrust demand and the demand for electrical power.

The one or more rotating electrical machines 420, 430 and the power electronic converters 440 may be configured to deliver or receive electrical power to one, two, or more DC buses or power channels. The DC channels enable the distribution of electrical power to other electrical consumers in the engine and to electrical consumers on the airframe.

Those skilled in the art will understand that the gas turbine engine 400 described above may be considered a more "electric" gas turbine engine because the electric machines 420, 430 play a larger role than in conventional gas turbines.

In an exemplary energy and propulsion system 500 of a hybrid electric aircraft is shown. The system 500 includes a generator set 501, which includes an engine 560 and an electrical generator 540a, and a battery pack 520. Both the generator 501 and the battery pack 520 are used as energy sources for driving an engine-driven propulsor 502, which is in is shown as an example.

The illustrated drive system 500 also includes a rectifier 510a, a DC distributor 530, an inverter 510b and a DC converter 510c. It will be appreciated that a propulsion system 500 may include more than one generator 501 and/or one or more propulsors 502, although in this example a generator 501 and a propulsor 502 are shown.

A shaft or shaft of the engine 560 is coupled to and rotates a shaft of the generator 540a, thereby generating alternating current. The rectifier 510a, located opposite the generator 540a, converts the alternating current into direct current, which is fed to various electrical systems and loads via the direct current distribution bus 530. These electrical systems include non-driving consumers (in not shown) and the motor 540b, which drives the fan 550 of the propulsor 502 via the inverter 510b.

The battery pack 520, which may consist of a series of battery modules connected in series and/or parallel, is connected to the DC distribution bus 530 via the DC-DC converter 510c. The DC-DC converter 510c converts between a terminal voltage of the battery pack 520 and a voltage of the DC distribution bus 530. In this way, the battery pack 520 can replace or supplement the power provided by the generator 501 (by discharging and thereby damaging the DC distribution bus 530 feeds) or is charged with the power of the generator 501 (by being fed from the DC distribution bus 530).

As in shown, the propulsor 502 in this example has the shape of a ducted fan with an electrical machine 540b. The fan 550 is enclosed in a fan channel 551 defined in a nacelle 552 and attached to a core nacelle 553. The fan 550 is driven by the electric machine 540b via a drive shaft 554, both of which can also be viewed as components of the propulsor 502. In this embodiment, a gear 555 is provided between the electric machine 540b and the drive shaft 554.

The electrical machine 540b is powered via the DC bus 530 from a power source, e.g. B. the generator set 501 and / or the battery 520, supplied with electrical energy. The electric machine 540b of the propulsor 502 and also the electric machine 540a of the generator set 501 may be of any suitable type, for example of the permanent magnet synchronous type.

The inverter 510b can be integrated into the electrical machine 540b (e.g. share a common housing structure with it) and thus form part of the propulsor 502. Likewise, the rectifier 510a can be integrated into the electrical machine 540a (e.g., share a common housing structure with it). The DC-DC converter 510c could in turn be integrated into the energy storage system 520.

Experts will recognize that the drive system is 500 14A-B is a series hybrid. Other hybrid electric propulsion systems are of the parallel type, while still others are of the turboelectric type or have features of more than one type. The configuration of the “more electric” engine 400 in can be considered similar to a parallel hybrid system, with the main difference being the role of the electric machines. Thus, the electrical machines of a "more electric" engine are generally used only in engine operation to start the engine and improve the operability of the engine, while the electrical machines of a parallel hybrid propulsion system are used to drive the shafts to provide the power of the engine The propulsion thrust generated by the turbo machine can be sensibly increased.

Experts also know that the in The hybrid architecture shown is just an example, and other architectures are known and will occur to those skilled in the art.

Various examples have been described, each with different combinations of features. Those skilled in the art will appreciate that each of the features may be used alone or in combination with other features unless they are clearly mutually exclusive, and the invention extends to all combinations and subcombinations of one or more features described herein.

It should also be noted that although the embodiments have been described with reference to an aircraft and turbofan engines, the principles of the electrical systems described may also be applied to other equipment, e.g. B. on aircraft with turboprop and open rotor engines, in marine environments such as on a naval ship and on other transport applications including trains.

Claims (20)

Power electronic converter, comprising: a converter commutation cell with a power circuit and a gate driver circuit, the power circuit having at least one power semiconductor switching element and at least one capacitor, wherein: each power semiconductor switching element is included in a power semiconductor prepackage, each prepackage being a or comprises a plurality of power semiconductor switching elements embedded in a solid insulating material, each power semiconductor switching element having at least three terminals including a gate terminal; the gate driver circuit is electrically connected to the gate terminal of each of the at least one power semiconductor switching elements and configured to provide switching signals; and a peak rated power of the power electronic converter is greater than 25 kW and a value of one Converter parameter α is less than or equal to 5 pHm ³ , where α is defined as the product of a smallest cuboid volume that includes the commutation cell and a parasitic inductance of the power circuit of the commutation cell. Power electronic converter according to Claim 1 , where the value of α is greater than or equal to 0.3 pHm ³ . Power electronic converter according to Claim 1 or Claim 2 , where the value of α is less than or equal to 4 pHm ³ . Power electronic converter according to one of the preceding claims, wherein the value of α satisfies 0.4 pHm ³ ≤ σ ≤ 3.5 pHm ³ . Power electronic converter according to one of the preceding claims, wherein the value of α satisfies 0.5 pHm ³ ≤ α ≤ 2.5 pHm ³ . Power electronic converter according to one of the preceding claims, wherein the product of the parasitic inductance of the power circuit of the commutation cell and the peak nominal power is in the range of 0.05 mHW to 1.5 mHW. Power electronic converter according to one of the preceding claims, wherein the value of σ divided by the peak nominal power of the power electronic converter is in the range of 1.5 aHm ³ /W to 100 aHm ³ /W. Power electronic converter according to one of the preceding claims, in which a total nominal capacity of the power circuit of the commutation cell divided by the peak nominal power is less than or equal to 5 nF/W. Power electronic converter according to one of the preceding claims, further comprising: a multilayer planar carrier substrate defining an xy direction parallel to a planar surface of the substrate and a z direction perpendicular to the xy direction, the carrier substrate having a plurality of electrically conductive layers extending in the xy direction, and at least one electrical connection that extends in the z direction, where: each power semiconductor prepackage further has an electrical connection from at least one of its terminals to an electrical connection side of the power semiconductor prepackage, the electrical connection extending in the z-direction through the solid insulating material; and at least one of the connections of each of the at least one power semiconductor switching elements is connected to at least one of the conductive layers of the multilayer planar carrier substrate on the electrical connection side of the power semiconductor prepackage. Power electronic converter according to Claim 9 , wherein for each power semiconductor prepackage, the electrical connection side of the prepackage forms a flat surface and the prepackage is surface-mounted on its electrical connection side on a flat surface of the multilayer planar carrier substrate. Power electronic converter according to Claim 10 , wherein for each power semiconductor prepackage, each electrical connection extending from the at least one of the terminals of the power semiconductor switching element through the solid insulating material terminates at the flat surface of the prepackage. Power electronic converter according to Claim 11 , wherein each power semiconductor prepackage is surface mounted on the surface of the multilayer planar carrier substrate by soldering, sintering or gluing the terminating electrical terminal to an electrical connection of the multilayer planar carrier substrate. Power electronic converter according to Claim 12 , wherein the soldered, sintered or bonded connections of each power semiconductor prepackage are spaced from the planar surface of the multilayer planar carrier substrate to define a gap, the size of the gap, measured in the z-direction, being less than or equal to 300µm. Power electronic converter according to one of the Claims 9 until 13 , wherein the multilayer planar substrate: a rigid printed circuit board (PCB); a flexible circuit board; or is a ceramic carrier substrate. Power electronic converter according to one of the preceding claims, which is an AC-DC converter. Power electronic converter according to one of the Claims 1 until 14 , which is a DC-DC converter. Electric drive unit (EPU) for an aircraft, the EPU being an electric motor and the power electronic AC-DC converter Claim 15 has, wherein the power electronic AC-DC converter is configured as an inverter and arranged so that it provides power to a winding of the electric motor. Gas turbine engine, comprising a shaft; an electric machine having a rotor mechanically coupled to the shaft; and the power electronic AC-DC converter Claim 15 , wherein the power electronic AC-DC converter is arranged so that it provides power to a winding of the electrical machine or receives power from it. Power supply system for an aircraft, comprising a power electronic converter according to one of Claims 1 until 16 . Airplane that the EPU is following Claim 17 , the gas turbine engine Claim 18 or the power system Claim 19 having.

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