DE102021208278A1 - Power converter circuit for generating a potential-separated DC voltage - Google Patents

Abstract

Converter circuit (10) for generating a potential-separated DC voltage (uout), having: a switchable input converter (30), a switchable oscillating circuit converter (40), an m-phase transformer (50) and a rectifier bridge (60); wherein the input converter (30) is configured to be connected to an n-phase power grid (20) and to generate a pulse-shaped output current (IDC) from n input currents (i1; i2; i3) at an input converter switching frequency f0; wherein the tank circuit converter (40) has m phases and is configured to convert the pulsed output current (IDC) of the input converter (30) in each of the m phases at a tank circuit converter switching frequency f0/m into a tank circuit current (ia; ib; ic); wherein the m resonant circuit currents (ia; ib; ic) in the m phases are each shifted in phase angle to one another; each of the m phases of the tank circuit converter (40) having a tank circuit capacitor (Cp) and forming a parallel tank circuit with a main inductance (Lh) of the m-th phase of the transformer (50); wherein in each case the tank circuit capacitor (Cp) of the m-th phase of the tank circuit converter (40) is configured to provide a magnetizing current for the m-th phase of the transformer (50), and the transformer (50) is configured to provide m secondary-side output voltages (uab; ubc; uca) to generate isolated from the primary side of the transformer (50); wherein the m phases of the transformer (50) are connected to the rectifier bridge (60) on the secondary side and the rectifier bridge (60) is configured to rectify the m secondary-side output voltages (uab; ubc; uca) of the transformer (50) and at one output the electrically isolated to generate DC voltage (uout); and wherein n is a positive integer and m is a positive integer greater than or equal to 2.

Description

The invention relates to a converter circuit and a converter circuit system for generating a potential-separated direct voltage and its use, as well as a method for generating a potential-separated direct voltage by a converter circuit.

Mobile electrical applications, especially in vehicle technology, require ever more powerful traction batteries and accumulators in order to have sufficient electrical energy supplied over a longer period of time. For charging such high-performance traction batteries with direct current, because of the high protection requirements for such charging stations, electrical isolation is required between the AC mains supply and the direct voltage to be applied to a traction battery. In the prior art, this electrical isolation occurs either on the network side by means of transformers or within a power converter by using a high-frequency clocked converter. These converters in the power converters usually have a single-phase design and are fed from an upstream DC link. For high charging capacities over 100 kW, charging stations with a line-side transformer or a power converter with a DC link and converter are very expensive. A major reason for this is the cost of the coiled goods and the capacitors for intermediate energy storage, which are required for such electrical circuits of the charging stations. In this case, suitable mains transformers cannot be manufactured automatically and, in addition, high production costs arise due to the high material costs for iron cores and winding goods. When converters are used in this power range, the switching frequencies to be used are comparatively low and therefore require manual and therefore cost-intensive wiring of the converter circuits. However, if the field of electromobility is to be used extensively in private use in addition to the commercial environment, powerful fast charging devices must become significantly cheaper.

It is therefore an object of the present invention to provide a power converter circuit and a method for generating a potential-separated DC voltage which causes lower costs for the winding goods and energy storage devices to be used and enables largely automated production.

This problem is solved by the subjects of the independent claims. Advantageous embodiments are the subject matter of the dependent claims.

One aspect of solving the problem relates to a converter circuit for generating a potential-separated DC voltage, having: a switchable input converter, a switchable oscillating circuit converter, an m-phase transformer and a rectifier bridge; wherein the input converter is configured to be connected to an n-phase power grid and to generate a pulse-shaped output current from n input currents at an input converter switching frequency f ₀ ; wherein the tank converter has m phases and is configured to convert the pulsed output current of the input converter into a tank current in each of the m phases at a tank converter switching frequency f ₀ /m; wherein the m resonant circuit currents in the m phases are each shifted in phase angle to one another; each of the m phases of the oscillating circuit converter having an oscillating circuit capacitor and forming a respective parallel oscillating circuit with a main inductance of the m-th phase of the transformer; wherein each tank circuit capacitor of the m-th phase of the tank circuit converter is configured to provide a magnetizing current for the m-th phase of the transformer and the transformer is configured to generate m secondary-side output voltages isolated from the primary side of the transformer; wherein the m phases of the transformer are connected to the rectifier bridge on the secondary side and the rectifier bridge is configured to rectify the m secondary-side output voltages of the transformer and to provide the isolated DC voltage at an output; and wherein n is a positive integer and m is a positive integer greater than or equal to 2.

The power converter circuit offers the advantage that higher switching frequencies are possible compared to conventional power converters with a DC link and converter. This is accompanied by a reduction in the size of the windings used for input inductances and transformers. The use of large and therefore expensive capacitors, such as electrolytic capacitors for intermediate storage of energy in the DC voltage intermediate circuit, can be dispensed with entirely. The smaller coils and capacitors that can be used for the converter circuit can also be integrated more easily in an automated production process for the converter circuit and, in addition to the lower material costs, also make the production process cheaper.

In this case, the power converter circuit does not require a direct voltage intermediate circuit, but rather, viewed from the input side in the direction of the output, has an input power converter, a m-phase resonant circuit converter, an m-phase transformer and a rectifier bridge and the converter circuit can be connected and/or connected to an n-phase power supply system on the input side and to a rechargeable battery on the output side.

The input converter and the m-phase oscillating circuit converter are each configured to be switchable. This means that the input converter and the m-phase resonant circuit converter each have at least one switching device, which provide an input signal in a switched-on state for a switch-on period T _On at their output and which do not provide the input signal in a switched-off state for a period T _Off at their output make available. A periodic, hard switching on and off of the at least one switching device with a predetermined input converter switching frequency f ₀ can thus be used to generate an output signal with a pulsed profile from a continuous input signal. In this case, hard switching means very fast switching of the switching device in order to obtain an output signal with a very steep edge profile during the switching process.

An m-phase transformer is always described below as a transformer configuration in which a primary side of the transformer has m phases, each with a primary winding, and a secondary side of the transformer has m phases, each with a secondary winding. In this case, the primary winding of the m-th phase of the primary side is always magnetically coupled to the secondary winding of the m-th phase of the secondary side. In other words, the transformer has m primary windings and m secondary windings. Here, the m-phase transformer may preferably have a common core for all m phases to be used more effectively, and the core may preferably be made of ferrite or amorphous metals. However, the m-phase transformer can also be configured by connecting m single-phase transformers together. The transformer core can also be made of any soft magnetic material. The primary side is isolated from the secondary side of the m-phase transformer. The use of the m-phase transformer in the converter circuit thus causes potential isolation between the input and the output of the converter circuit.

The power grid to which the converter circuit can be connected and/or to which the converter circuit can be connected can be a 1-phase power grid or a multi-phase power grid with any number n of phases. In this case, the converter circuit can preferably be connected on the input side to an n-phase AC network, the n network currents preferably being sinusoidal. Another AC shape such as a triangular shape is also possible. However, the converter circuit can also be connected to an n-phase direct current network and/or to n batteries.

The power converter circuit for a multi-phase AC network can preferably be connected to a symmetrical AC network, since the input power converter can then also be constructed symmetrically and the circuit design is simplified. The input power converter can have n input phases, with each input phase having its own connection terminal. In particular, the number of input phases of the input power converter and the number of phases of the AC network can be identical. However, the number n of input phases of the input converter can also be larger or smaller than the number of phases of the power grid, in which case not all phases of the power grid can be connected with a smaller number of input phases or not all input phases with one phase with a larger number of input phases of the power grid can be connected. In such a configuration, only the phases of the input converter that are actually connected are processed further. The flexibility in the design of the power converter circuit allows it to be used in a wide variety of applications that are not limited to a specific type of power grid. When connecting the power converter circuit to the power grid, it should be noted that an input inductance can be or is connected between each phase of the power grid and each input, ie on each input phase of the input power converter. The size of the input inductances depends on an input converter switching frequency f ₀ of the switching devices used in the input converter. In this case, the input converter switching frequencies for an input converter without a subsequent DC link can be configured higher than for an input converter with a subsequent DC link, since charging and discharging a large capacitance in the DC link can be omitted. This means that smaller and less expensive input inductances can also be selected for input converters without a subsequent DC voltage intermediate circuit.

During operation, the input converter switches its switching devices hard at an input converter switching frequency f ₀ and generates an input current for each input phase n, with the input currents at the output of the input converter are combined and/or added to form a pulsed output current. When connecting to an n-phase AC network with n sinusoidal network currents, a sinusoidal input current is generated for each input phase. The pulse-shaped output current of the input converter is not fed into a DC voltage intermediate circuit as in conventional converters, but is converted by the m-phase oscillating circuit converter in its m phases into m oscillating circuit currents. The phase angles of the m resonant circuit currents in the m phases of the resonant circuit converter are shifted relative to one another. For this purpose, all switching devices of the resonant circuit converter are switched with a resonant circuit converter switching frequency f ₀ /m, with all switching devices of an m-th phase being switched at the same time. However, the switching times of the m phases differ and are shifted by a phase angle. In this case, the number of phases m of the resonant circuit converter is at least two and preferably an odd number of phases.

In the oscillating circuit converter, an oscillating circuit consisting of an oscillating circuit capacitor and a main inductance of the transformer is configured per phase, the main inductance being at least part of a primary-side winding in one phase of the m-phase transformer. In this case, the parallel resonant circuits are preferably operated at a resonant frequency or close to the resonant frequency. The oscillating circuit capacitors and also the main inductances can also be configured differently from one another, depending on the requirements placed on the power converter circuit. In general, resonant circuit capacitors can be configured on the primary and secondary side. In this case, the resonant circuit capacitors can be distributed centrally on the primary side or distributed on the primary side and the secondary side. The total capacitance of the oscillating circuits results from the entirety of the oscillating circuit capacitors. For the transmission from the primary side of the transformer to its secondary side, a magnetizing current is supplied per phase from the respective oscillating circuit capacitor into the winding transmitting on the primary side. The output voltages of the m-phase transformer, which are transmitted in isolated form to the secondary side of the transformer, are then rectified by the rectifier bridge and combined to form a common, isolated direct voltage. An accumulator to be charged can then be and/or connected to this DC voltage in order to be charged.

In a preferred embodiment of the converter circuit, the converter circuit can be connected to a 3-phase AC network and/or the resonant circuit converter and the transformer have three phases. Optionally, the main inductances of the transformer are configured in a star or delta arrangement.

The 3-phase alternating current network to which the converter circuit can be connected can preferably be a symmetrical three-phase network with a network frequency of 50Hz or 60Hz, which is available almost everywhere in both commercial and private environments. The phase angles of the three alternating voltages and currents of the three-phase system each differ by 120°. A three-phase configuration of the tank circuit converter and the transformer also simplifies the circuit design of the converter circuit. Here, the primary-side windings of the 3-phase transformer can be configured in a delta or delta arrangement, in a star arrangement or in a zigzag arrangement. In addition, the secondary side windings of the 3-phase transformer can be configured in a delta or delta arrangement or in a star arrangement. Preferably, both the primary and secondary windings of the 3-phase transformer are configured the same and both sides of the transformer have the delta or delta configuration, or both sides of the transformer have the star configuration. However, mixed arrangements on the primary side and the secondary side of the 3-phase transformer are also possible. The main inductances of the three phases can each make up the entire winding or just part of it.

In a further preferred embodiment of the converter circuit, the converter circuit has a protective diode between the input converter and the m phases of the oscillating circuit converter and/or the pulsed output current of the input converter is configured to have a current intensity of 0 A when the m phases of the oscillating circuit converter are respectively switched . For configurations in which the current intensity can deviate from 0A when switching the m phases of the oscillating circuit converter, a configuration with protective diodes should preferably be used to protect the converter circuit.

In a further preferred embodiment of the converter circuit, the m main inductances of the m-phase transformer can each be adjusted via an air gap. In this way, the secondary-side output voltages of the m-phase transformer and the potentialge easily control isolated DC voltage at the output of the power converter circuit.

In another preferred embodiment of the power converter circuit, the rectifier bridge is configured as a midpoint circuit or as a B6 rectifier bridge. In this case, the ripple of the DC voltage at the output of the rectifier bridge can be further reduced by a smoothing capacitor. When the power converter circuit is configured as a midpoint circuit, there is only one diode in the current path during operation, as a result of which the losses at the diodes can be reduced. Therefore, the midpoint circuit is preferred for smaller output voltages, such as 48V. The diodes of the midpoint circuit have twice the blocking voltage of the diodes of the B6 rectifier bridge.

In a further preferred embodiment of the power converter circuit, the input power converter has a control circuit and the control circuit is configured to generate pulse patterns for switching the input power converter, the pulse patterns controlling the input power converter in such a way that the reactive power of the input power converter is minimized. The use of a reactive power-optimized control circuit can thus also increase the efficiency of the converter circuit.

In a further preferred embodiment of the converter circuit, the input converter switching frequency f ₀ of the converter circuit can be adjusted before or during operation to optimize overvoltages and efficiency or to compensate for parameter deviations. In this case, the input power converter switching frequency f ₀ can preferably be adjusted via the control circuit.

A further aspect of solving the problem relates to a converter circuit system for generating a potential-separated DC voltage, having: a plurality j of converter circuits as described above, the plurality j of converter circuits being arranged in parallel with one another, and j being a positive integer greater than 1.

On the input side, the individual power converter circuits can be connected in parallel. In a single-phase supply on the mains side, the converter circuits can also be arranged in a series connection to reduce the voltage stress on the semiconductors and/or to reduce the distortions. On the secondary side of the converter circuits, there is always the possibility of parallel connection, series connection or a combination of both variants in order to minimize the distortion of the output current. When the converter circuits are connected in parallel, there is no power transmission via the resonant circuit capacitors and the power transmission also takes place in parallel, with the individual converter circuits needing to transmit less power than without the parallel connection. Therefore, in turn, the components of the individual power converter circuits can be made smaller. This in turn facilitates automated production of the entire power converter circuit system.

When a plurality of converter circuits are connected in series, the power transfer across each of the converter circuits is much higher than when the converter circuits are connected in parallel.

In a preferred embodiment of the converter circuit system, the plurality j of converter circuits are configured in such a way that the pulse patterns of the input converters of the parallel-connected converter circuits are shifted by a phase angle 2π/j to one another with the same input converter switching frequency f ₀ in relation to their period duration T ₀ . In this case, a phase angle shift is not limited to 2π/j, but can also have a different phase angle, which preferably improves and/or optimizes the transformation ratio at the transformers. With this configuration, the ripple of the input current and the output current of the converter circuit system is reduced. The permissible ripple or current ripple of the output current is standardized here. The ripple can be kept below the standardized value by a sufficiently high number of phase-shifted power converter circuits arranged in parallel. With a sufficiently high number of phase-shifted converter circuits arranged in parallel, the input inductances can also be further reduced and the production of the converter circuit system can in turn be automated more easily. Such a parallel connection is only possible due to the electrical isolation in the output stage.

A further aspect of solving the problem relates to the use of a converter circuit or a converter circuit system as described above for generating a potential-separated DC voltage.

A further aspect of solving the problem relates to a method for generating a potential-separated DC voltage using a converter circuit, the converter circuit having an m-phase oscillating circuit converter with an oscillating circuit capacitor per phase and an m- phase transformer, wherein the tank circuit capacitor of the m th phase of the tank circuit converter forms a parallel tank circuit with a magnet inductance of the m th phase of the transformer, respectively, and the method comprises: receiving n input currents; generating a pulsed current at a switching frequency f ₀ from the n input currents; Converting the pulsed current into m resonant circuit currents, each with a switching frequency f ₀ /m, the phase angle of the resonant circuit currents being shifted relative to one another; providing a respective magnetizing current for the mth phase of the transformer through the resonant circuit capacitor of the mth phase of the resonant circuit converter; generating m secondary-side output voltages which are electrically isolated from the primary side of the transformer; rectifying the m secondary output voltages of the transformer and generating the isolated DC voltage; and wherein n is a positive integer and m is a positive integer greater than or equal to 2.

The statements made above or below regarding the embodiments of the respective other aspects also apply to the above-mentioned aspects and in particular to embodiments that are preferred in this regard.

In the following, individual embodiments for solving the problem are described by way of example with reference to the figures. Some of the individual embodiments described have features that are not absolutely necessary to implement the claimed subject matter, but which provide desired properties in certain applications. Thus, embodiments that do not have all the features of the embodiments described below are also to be regarded as being disclosed as falling under the technical teaching described. Furthermore, in order to avoid unnecessary repetition, certain features are only mentioned in relation to individual embodiments described below. It is pointed out that the individual embodiments should therefore not only be considered individually, but should also be viewed as a whole. Based on this synopsis, the person skilled in the art will recognize that individual embodiments can also be modified by incorporating individual or multiple features of other embodiments. It is pointed out that a systematic combination of the individual embodiments with individual or multiple features that are described in relation to other embodiments can be desirable and useful and should therefore be considered and should also be regarded as covered by the description.

character list

• 1 FIG. 12 shows a schematic representation of a power converter circuit according to a preferred embodiment of the present invention.
• 2 shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention without protective diodes.
• 3 shows a time course of input currents, an output current and an output voltage in an input power converter according to a preferred embodiment of the present invention.
• 4 shows a detailed time profile of an output current in an input converter and of oscillating circuit currents in an oscillating circuit converter according to the preferred embodiments of the present invention from FIGS 1 and 2 .
• 5 FIG. 12 shows a detailed time course of a tank circuit current and associated voltages of a tank circuit converter according to the preferred embodiment of the present invention 1 ,
• 6 FIG. 12 shows a detailed time course of a resonant circuit current and associated voltages of a resonant circuit converter according to the further preferred embodiment of the present invention without protective diodes 2 .
• 7 shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention without protective diodes and an alternative primary-side transformer circuit.
• 8th shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention without protective diodes and an alternative transformer circuit in a delta arrangement.
• 9 shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention without protective diodes and an alternative rectifier bridge circuit.
• 10 shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention without protective diodes and an alternative rectifier bridge with center point circuit.
• 11 12 shows a schematic representation of a power converter circuit according to a further preferred embodiment of the present invention with an alternative transformer and rectifier bridge circuit.
• 12 FIG. 12 shows a block diagram of a control circuit of an input converter according to a preferred embodiment of the present invention.
• 13 shows a time course of the input voltages and the input currents, as well as the active and reactive power consumed in the event of a setpoint jump in the mains current, according to a preferred embodiment of the present invention.
• 14 shows a schematic representation of a power converter circuit system with a plurality of parallel-connected power converter circuits according to a preferred embodiment of the present invention.
• 15 FIG. 1 shows a time profile of input currents in a converter circuit system with a different number of parallel-connected converter circuits, each with a temporal phase offset, according to a preferred embodiment of the present invention.
• 16 shows a time course of output currents in a converter circuit system with a different number of parallel-connected converter circuits, each with a temporal phase offset, according to a preferred embodiment of the present invention.

Detailed description of the figures

The 1 FIG. 1 shows a schematic representation of a power converter circuit 10 according to a preferred embodiment of the present invention. The converter circuit has an input converter 30, a resonant circuit converter 40, a transformer 50 and a rectifier bridge 60, viewed from the input side in the direction of the output. The power converter circuit 10 can be connected and/or connected to an AC network 20 on the input side and to a rechargeable battery 70 to be charged on the output side. It should be noted here that the transformer 50 in FIGS 1 , 2 , 7 until 10 and 14 is represented in an idealized manner, i.e. without coil resistances, hysteresis losses in the core and leakage inductances.

The AC mains 20 in 1 is shown as a 3-phase, symmetrical three-phase network, but can also be a 1-phase AC network or a multi-phase AC network with any number of phases. In order to be able to connect the AC network 20 to the input converter 30 on the input side, an input inductance L _E is preferably interposed between each phase of the AC network 20 and each input of the input converter 30 . In other words, an input inductance L _E can be connected to each input, ie to each input phase of the input converter 30, via which the input converter can be connected to the AC network. The size of the input inductances L _E depends on a switching frequency f ₀ of the switching devices used in the input converter 30 and can be designed to be correspondingly smaller for a higher input converter switching frequency f ₀ . The input converter 30 itself has, following its inputs, two symmetrically arranged switching devices for each phase, whose input converter switching frequency f ₀ is controlled via a control input by a control circuit 80 (in 1 not shown) can be controlled. The switching devices are shown here as MOSFETs, but they can also be replaced by other conventional field effect transistors or bipolar transistors. The symmetrically arranged switching devices form a positive and a negative branch for each phase, and the outputs of the positive branch of all phases are combined and/or added in a common positive output. The outputs of the negative branch of all phases are also combined and/or added in a common negative output. During operation, the input converter 30 switches the switching devices hard, or with a very steep edge, at an input converter switching frequency f ₀ and generates a sinusoidal input current i ₁ , i ₂ and i ₃ for each phase at an input voltage u ₁ , u ₂ and u ₃ . During operation, a pulsed output voltage U _DC , which is approximately a DC voltage, is present between the common positive and negative outputs of the input converter 30 . The combined and/or added output current I _DC also has a pulsed profile.

The pulse-shaped output current I _DC is not fed into a DC voltage intermediate circuit, as is the case with conventional converters, but serves as an input current for the oscillating circuit converter 40. The oscillating circuit converter 40 in 1 has three phases, but can also be divided into two or more phases. The oscillating circuit converter 40 has two symmetrically arranged switches for each phase directions, each with an upstream protective diode 42, in which one switching device is connected to the positive branch of the output of the input converter 30 and the other switching device is connected to the negative branch of the output of the input converter 30. The switching devices are in this case with a resonant circuit converter switching frequency f ₀ / 3 and thereby generate a resonant circuit current i _a , i _b and i _c per phase at a respective voltage of u _a , u _b and u _c between the three phases and the negative branch of the input converter 30. This occurs when the switching devices of the resonant circuit converter are switched 40 no switching losses, since the switching devices are each switched in a currentless state of the pulsed output current I _DC of the input converter 30. For this reason, the protective diodes 42 are not absolutely necessary if the pulsed output current I _DC is regulated in a stable manner. In the oscillating circuit converter 40, an oscillating circuit consisting of an oscillating circuit capacitor C _P and a main inductance L _H is configured per phase, the main inductance L _H being part of a primary-side winding in one phase of a 3-phase transformer. Here is in 1 the main inductance L _H per phase is arranged in parallel with a second primary-side inductance and the 3-phase transformer is constructed in a star shape on the primary and secondary sides. The voltage between the star point of the transformer and the negative branch of the input converter 30 is designated by the voltage u ₀ - . In 1 only transfers the second primary-side inductance to the secondary side of the transformer 50. For the transfer from the primary side of the transformer 50 to its secondary side, a magnetizing current is supplied by the respective resonant circuit capacitor C _P per phase. For each phase, this results from the difference between the resonant circuit currents i _a , _{ib and ic and currents i ha , i hb and i hc through the main} inductances L _H . In order to fix the resonant circuits in the respective phases, the main inductances L _H can be configured to be adjustable via an air gap.

The output voltages u _ab , u _bc and u _ca of the 3-phase transformer, which are transmitted to the secondary side of the transformer with potential isolation, are rectified by the rectifier bridge 60 and an output current i _out is provided at an output voltage u _out at the output of the converter circuit 10 . Here is in 1 the rectifier bridge 60 is configured in a B6 configuration and the output voltage is further smoothed at the output by a smoothing capacitor C _G .

The 2 shows a schematic representation of a converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42. The configuration of the converter circuit 10 is identical to the configuration in FIG 1 .

The 3 shows a time profile of input currents i ₁ , i ₂ and i ₃ , an output current I _DC and an output voltage U _DC in an input converter 30 according to a preferred embodiment of the present invention. The behavior of the converter circuits 10 from the 1 and 2 in the further description of the figures always with a mains frequency of 500 Hz, with an input inductance L _E of 200 µH, with an input converter switching frequency f ₀ of the input converter 30 of 83 kHz, with a capacitance of the resonant circuit capacitor C _P of 188 nF, in each case a main inductance L _H of 169uH and a voltage at the accumulator 70 of 800V simulated. Here point for the 1 and 2 the phase-shifted, sinusoidal input currents i ₁ , i ₂ and i ₃ , due to the hard switching of the switching devices of the input converter 30, develop a jagged course with the switching frequency of 83 kHz of the input converter 30. The pulse-shaped output voltage U _DC has an approximately DC-shaped profile and the pulse-shaped output current I _DC is composed of the combination and/or addition of the input currents i ₁ , i ₂ and i ₃ . The invention is not limited to the above frequencies, voltages and currents. Rather, these examples merely serve to illustrate the inventive principle.

The 4 shows a detailed time profile of an output current I _DC in an input converter 30 and of resonant circuit currents i _a , i _b and i _c in a resonant circuit converter 40 according to the preferred embodiments of the present invention from FIGS 1 and 2 . Here, the output current I _DC of the input converter 30 compared to 3 shown with a higher resolution over time in order to clarify the pulse-shaped course. The resonant circuit currents i _a , i _b and i _c of the resonant circuit converter 40 result from the output current I _DC of the input converter 30 by phase-shifted switching of the switching devices of the resonant circuit converter 40 at a switching frequency of f ₀ /3. The oscillating circuit currents i _a , i _b and i _c have a symmetrical, pulsed profile with negative and positive amplitude values.

The 5 FIG. 12 shows a detailed time course of a resonant circuit current i _a and associated voltages of a resonant circuit converter 40 according to the preferred embodiment of the present invention 1 with protective diodes 42. The resonant circuit current i _a is compared to 4 again with a higher temporal resolution. In addition, the resulting negative voltages u _a -, u _b - and u _c - of the resonant circuit converter 40, the secondary-side output voltages u _ab , u _bc and u _ca at the transformer 50 and the voltage u ₀ - between the star point of the transformer 50 and the negative branch of the input converter 30 is shown.

The 6 shows a detailed time course of a resonant circuit current i _a and associated voltages of a resonant circuit converter 40 according to the further preferred embodiment of the present invention without protective diodes 2 . In this case, the omission of the protective diodes 42 in 2 a smoothing of stresses, as compared to 5 is visible.

The 7 shows a schematic representation of a power converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative primary-side transformer circuit 50. Here, the embodiment is based in 7 on the embodiment 2 . In contrast to the primary of the transformer 50 in 2 Here, no second primary-side inductance is configured parallel to the main inductance L _H per phase, but rather the main inductance L _H is also used for transmission in the transformer 50 to the secondary side.

The 8th shows a schematic representation of a power converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative transformer circuit 50 in a delta arrangement. Here, the embodiment is based in 8th on the embodiment 7 . Unlike the transformer 50 in 7 with a star configuration the main inductances L _H are in 8th the main inductances L _H configured in a triangle arrangement. In this case, the resulting capacitance of the resonant circuit capacitors C _P facing the input converter 30 is greater in the delta arrangement than in the star connection. Among other things, this leads to lower overvoltages in the converter circuit 10.

The 9 shows a schematic representation of a power converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative rectifier bridge circuit 60. Here, the embodiment is based in 9 on the embodiment 7 . Unlike the rectifier bridge 60 in 7 with a B6 configuration, in 9 a rectifier bridge is shown with additional secondary-side resonant circuit capacitors C _P . These oscillating circuit capacitors C _P together with the main inductance of the transformer 50 form an oscillating circuit on the secondary side. The division of the oscillating circuit capacitors C _P serves to protect individual components in the converter circuit 10. The total capacitance of the oscillating circuits in the converter circuit 10 results from the entirety of the oscillating circuit capacitors C _P .

The 10 FIG. 12 shows a schematic representation of a power converter circuit 10 according to a further preferred embodiment of the present invention without protective diodes 42 and an alternative rectifier bridge 60 with neutral point connection. Here, the embodiment is based in 10 on the embodiment 7 . Unlike the rectifier bridge 60 in 7 with a B6 configuration, in 10 a rectifier bridge with a midpoint circuit is shown. In this case, the output voltage u _out is tapped off from midpoints of the secondary main inductances to one side of the main inductances and thus has half the output voltage u _out in the case of a B6 configuration of the rectifier bridge 60 .

The 11 shows a schematic representation of a power converter circuit 10 according to a further preferred embodiment of the present invention with an alternative transformer circuit 50 and rectifier bridge circuit 60. Here, the embodiment is based in 11 on the embodiment 1 , but represents the transformer circuit 50 as an equivalent circuit diagram of a real transformer 50 with leakage inductances. A distributed arrangement of the oscillating circuit capacitors C _P can better integrate parasitic capacitances and/or inductances into the circuit concept, depending on the configuration. In this case, the specific configuration depends on the structure of the overall arrangement.

The 12 FIG. 8 shows a block diagram of a control circuit 80 of an input power converter 30 according to a preferred embodiment of the present invention. In this case, the reactive power consumed by the input converter 30 is to be regulated to zero with the control circuit 80 .

For this purpose, the input voltages u ₁ , u ₂ and u ₃ and the input currents i ₁ , i ₂ and i ₃ are converted in a space vector converter 82 into mains voltage and current space vectors. The reciprocal amount of û _N is calculated in a reciprocal converter 84 from the αβ components u _Nα and u _Nβ of the network voltage space vector. This reciprocal absolute value is then multiplied by the ab components of the mains voltage in two first multipliers 86, resulting in αβ components arise with the amplitude 1 one. These αβ components of the voltage are then multiplied by a current reference value î _Nsetpoint of the superimposed control in two second multipliers 88, thereby forming the reference values i _Nαsetpoint and i _Nβsetpoint of the αβ mains currents. These current setpoints i _Nαsoll and i _Nβsoll are then phase-shifted in a current setpoint phase shifting unit 90, which will be discussed again below. These phase-shifted desired current values are compared with the actual current values i _Nα and i _Nβ in a comparator 92 and supplied to αβ current controllers 94 . In order to improve the dynamics, at the output of the αβ current controller 94 a disturbance variable 96 is carried out with the αβ components of the mains voltage. Desired control voltages u _SRαsoll and u _SRβsoll are generated as an output signal and made available to the input converter 30 for pulse pattern generation. A reactive power control takes place in the core element of the control circuit 80 . For this purpose, the reactive power consumed is calculated from the αβ components of the network variables in a reactive power calculation unit 98 and compared with the reactive power value zero. In this case, a reactive power arises when there is a phase shift between the input voltages u ₁ , u ₂ and u ₃ and the input currents i ₁ , i ₂ and i ₃ . Due to the control of alternating variables, there is always a phase shift between the actual current value and the current setpoint. A reactive power controller 100 generates a phase angle φ which shifts the phase of the target value of the grid currents in such a way that no reactive power is consumed. Due to the possibility of using very high switching frequencies, the control circuit 80 has high control dynamics with a very stable control behavior at the same time. In the simulation data shown here, a switching frequency f ₀ of the input converter 30 of 83 kHz is used, for example, which allows the use of SIC components with a simplified structure of the converter circuit 10 .

The 13 shows a time course of the input voltages u ₁ , u ₂ and u ₃ and the input currents i ₁ , i ₂ and i ₃ , as well as the consumed active power p(t) and reactive power q(t) in the event of a setpoint jump in the mains current according to a preferred embodiment of the present Invention. It is clear here that the control circuit 80 controls the input converter 30 of the converter circuit in such a way that the reactive power q(t) consumed is again at zero shortly after the setpoint jump. For better representation, for the simulation in 13 a mains frequency of the feeding mains of 500Hz is selected.

The 14 1 shows a schematic representation of a power converter circuit system 1 with a plurality j of parallel-connected power converter circuits 10 according to a preferred embodiment of the present invention. Here, the embodiments of the power converter circuit 10 described above are configured in parallel between the AC grid 20 and the accumulator 70 . This parallel connection is possible due to the potential isolation in the output stage of the individual power converter circuits 10 . The pulse patterns of the input converters 30 of the parallel-connected converter circuits 10 are shifted with respect to their period T ₀ by a phase angle 2π/j relative to one another with the same input converter switching frequency f ₀ . With the phase-shifted switching sequence of the parallel power converter circuits 10, the ripple of the input currents i ₁ , i ₂ and i ₃ can also be reduced with the same size of the input inductances L _E . If the ripple is optimized, it is possible to further reduce the size of the input inductances L _E .

The 15 shows a time course of input currents i ₁ in a converter circuit system 1 with a different number of parallel-connected converter circuits 10, each with a time phase offset according to a preferred embodiment of the present invention. Here shows the 15a the parallel time profile of three individual input currents i ₁ from three parallel-connected converter circuits 10 in a diagram, the input currents i ₁ of the parallel converter circuits 10 being generated by phase-shifted switching of the input converter 30. The 15b now shows the combined input current i ₁ of the current circuit system 1 consisting of the three added input currents i ₁ of the parallel, phase-shifted connected input converters 30, the amplitude of which was divided by the number three of the parallel converter circuits 10. The time offset of the phases results in T ₀ /3. 15c shows the combined input current i ₁ of a current circuit system 1 consisting of six parallel-connected power converter circuits 10. The time offset of the phases results in T ₀ /6. In this case, compared to 15b It is clear that the ripple of the combined input current i ₁ can be reduced by a higher number of converter circuits 10 connected in parallel.

The 16 shows a time course of output currents i _out in a converter circuit system 1 with a different number of parallel-connected converter circuits 10, each with a time phase offset according to a preferred embodiment of the present invention dung. Here shows the 16a , comparable to 15a , the parallel time course of three individual output currents i _out of three parallel-connected power converter circuits 10 in a diagram. 16b then shows the addition of the three individual output currents i _out 16a . In 16c is, in turn, comparable to 15c , the addition of six individual output currents i _out of a current circuit system 1 consisting of six converter circuits 10 connected in parallel can be seen. Here, too, the ripple of the combined output current i _out is reduced by a higher number of converter circuits 10 connected in parallel.

Reference List

1

power converter circuit system

10

converter circuit

20

AC grid

30

input power converter

40

resonant circuit converter

42

protection diodes

50

transformer

60

rectifier bridge

70

accumulator

80

control circuit

82

space vector converter

84

reciprocal converter

86

first multipliers

88

second multipliers

90

Current command phase shift unit

92

comparator

94

αβ current controller

96

Disturbance control

98

reactive power calculation unit

100

reactive power controller

LE

Input inductance of the input power converter

CP

resonant circuit capacitor

cg

smoothing capacitor

Lh

main inductance of the transformer

i1..n

Input current of the nth phase of an AC network

u1..n

Input voltage of the nth phase of an AC network

I.D.C

Pulsed output current of the input converter

U.D.C

pulsed output voltage of the input converter

ia,b,c

Resonant circuit currents in the m-phases of the resonant circuit converter

yeah, hb, hc

Currents through the main inductances in the m-phases

etc.,b,c

Voltages between the mth phase of the resonant circuit converter and the negative branch of the input converter

uab,bc,approx

Secondary-side output voltages at the transformer

u0-

Voltage between the neutral point of the transformer and the negative branch of the input converter

iout

Output current of the power converter circuit

out

Output voltage of the power converter circuit

uNα,β

αβ components of the input voltage space vectors

uNα,βset

Target values of the αβ components of the input voltage space vector

iNα,β

αβ components of the input current space vectors

iNα,βset

Target values of the αβ components of the input current space vector

U.N

reciprocal of the αβ components of the input voltage space vector

îNset

current setpoint

uSRα,βset

Reference control voltage for pulse pattern generation in the input converter

φ

phase angle

Q

reactive power

p(t)

active power consumed

q(t)

absorbed reactive power

n

Number of input phases of the input converter

m

Number of phases of the resonant circuit converter and the transformer

j

Number of converter circuits connected in parallel

f0

Input converter switching frequency

T0

Period duration at an input converter switching frequency f ₀

f0/m

Oscillating circuit converter switching frequency

Claims (10)

Converter circuit (10) for generating a potential-separated DC voltage (u _out ), having: a switchable input converter (30); a switchable resonant circuit converter (40); an m-phase transformer (50); and a rectifier bridge (60); wherein the input converter (30) is configured to be connected to an n-phase power grid (20) and to generate a pulsed output current (I _DC ) from n input currents (i ₁ ; i ₂ ; i ₃ ) at an input converter switching frequency f ₀ ; wherein the tank circuit converter (40) has m phases and is configured to convert the pulsed output current (I _DC ) of the input converter (30) in each of the m phases at a tank circuit converter switching frequency f ₀ /m into a tank circuit current (i _a ; i _b ; i _c ) to convert; wherein the m resonant circuit currents (i _a ; i _b ; i _c ) in the m phases are each shifted in phase angle to one another; each of the m phases of the oscillating circuit converter (40) having an oscillating circuit capacitor (C _p ) and forming a respective parallel oscillating circuit with a main inductance (L _h ) of the m-th phase of the transformer (50); wherein in each case the resonant circuit capacitor (C _p ) of the mth phase of the resonant circuit power converter (40) is configured to provide a magnetizing current for the mth phase of the transformer (50) and the transformer (50) is configured to output m secondary-side output voltages (u _ab ; to generate u _bc ; u _ca ) isolated from the primary side of the transformer (50); wherein the m phases of the transformer (50) are connected to the rectifier bridge (60) on the secondary side and the rectifier bridge (60) is configured to rectify the m secondary-side output voltages (u _ab ; u _bc ; u _ca ) of the transformer (50) and at one output to generate the isolated DC voltage (u _out ); and wherein n is a positive integer and m is a positive integer greater than or equal to 2. Converter circuit (10) after claim 1 , wherein the converter circuit (10) can be connected to a 3-phase AC network (20); and/or wherein the tank circuit converter (40) and the transformer (50) have three phases and optionally the magnetizing inductances (L _h ) of the transformer (50) are configured in a star or delta arrangement. Converter circuit (10) after claim 1 or 2 , wherein the converter circuit (10) between the input converter (30) and the m phases of the resonant circuit converter (40) each having a protective diode (42); and/or wherein the pulsed output current (I _DC ) of the input converter (30) is configured to have a current intensity of 0 A when the m phases of the oscillating circuit converter (40) are each switched. Converter circuit (10) according to one of Claims 1 until 3 , wherein the m main inductances (L _h ) of the m-phase transformer (50) can each be adjusted via an air gap; and/or wherein the rectifier bridge (60) is configured as a midpoint circuit or as a B6 rectifier bridge. Converter circuit (10) according to one of Claims 1 until 4 , wherein the input power converter (30) has a control circuit (80), and the control circuit (80) is configured to generate pulse patterns for switching the input power converter (30); and wherein the pulse patterns regulate the input converter (30) in such a way that the reactive power (Q) of the input converter (30) is minimized. Converter circuit (10) according to one of Claims 1 until 5 , wherein the input power converter switching frequency f ₀ of the power converter circuit (10) can be adjusted before or during operation to optimize overvoltages and efficiency or to compensate for parameter deviations. Converter circuit system (1) for generating a potential-separated DC voltage (u _out ), comprising: a plurality j of converter circuits (10) according to one of Claims 1 until 6 ; wherein the j plurality of power converter circuits (10) are arranged in parallel with one another, and wherein j is a positive integer greater than 1. Converter circuit system (1) according to claim 7 , wherein the plurality j of converter circuits (10) are configured in such a way that the pulse patterns of the input converters (30) of the parallel-connected converter circuits (10) are shifted by a phase angle 2π/j to one another with the same input converter switching frequency f ₀ in relation to their period T ₀ . Use of a converter circuit (10) or a converter circuit system (1) according to one of Claims 1 until 8th for generating a potential-separated DC voltage (u _out ). Method for generating a potential-separated DC voltage (u _out ) by a converter circuit (10), the converter circuit (10) having an m-phase oscillating circuit converter (40) with an oscillating circuit capacitor (C _p ) per phase and an m-phase transformer (50). , wherein the tank circuit capacitor (C _p ) of the mth phase of the tank circuit converter (40) forms a respective parallel tank circuit with a magnetizing inductance (L _h ) of the mth phase of the transformer (50), and the method comprises: receiving n input currents (i ₁ ; i ₂ ; i ₃ ); generating a pulsed current (I _DC ) at a switching frequency f ₀ from the n input currents (i ₁ ; i ₂ ; i ₃ ); Converting the pulsed current (I _DC ) into m resonant circuit currents (i _a ; i _b ; i _c ) each at a switching frequency f ₀ /m, the resonant circuit currents (i _a ; i _b ; i _c ) being shifted in their phase angle with respect to one another; Providing in each case a magnetizing current for the mth phase of the transformer (50) through the resonant circuit capacitor (C _p ) of the mth phase of the resonant circuit converter (40); generating m secondary-side output voltages (u _ab ; u _bc ; u _ca ), which are electrically isolated from the primary side of the transformer (50); rectifying the m secondary-side output voltages (u _ab ; u _bc ; u _ca ) of the transformer (50); and generating the isolated DC voltage (u _out ); and wherein n is a positive integer and m is a positive integer greater than or equal to 2.

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