DE102020215040A1 - LiDAR sensor system - Google Patents

Abstract

The present invention relates to a LiDAR sensor system (1) having a transmission unit (3) with a laser source (2), a phase modulator (7) for modulating a phase of the light from the laser source (2) and transmission optics (5) for emitting the the phase modulator (7) modulated light in an environment (10) of the LiDAR sensor system (1), a receiving unit (4) with receiving optics (6) for receiving light reflected on an object (11) of the environment (10) and with an evaluation unit (8) for evaluating the light received by the receiving optics (6), the transmission unit (3) being designed to emit a plurality of transmission sequences (100) of the light, each transmission sequence (100) having a signal phase-modulated by the phase modulator (7). is, and wherein the evaluation unit (8) is designed to remove an influence of the phase-modulated signal on a received light signal by exponentiation of the received light signal in order to thus from the received light signal a Dop pler frequency (fD) to determine and based on the Doppler frequency (fD) and the received light signal to determine a distance to the object (11).

Description

State of the art

The present invention relates to a LiDAR sensor system. The LiDAR sensor system can be used in particular in a vehicle. The LiDAR sensor system enables reliable Doppler compensation.

LiDAR sensor systems are known from the prior art. These mostly work according to the "direct detection" principle, in which an intensity of a reflected light is detected in order to infer an object in the environment. In addition, LiDAR systems are known that have coherent receivers. Such systems emit frequency-modulated signals in order to be able to detect a distance from the detected object on the one hand and a speed of the detected object on the other. The underlying method "frequency modulated continuous wave (FMCW)" is already known from radar technology. As an alternative to frequency modulation, phase codes can also be used, in which the phase of the emitted light is modulated. Such a method is, for example, from WO 2018/144853 A1 known.

Disclosure of Invention

The LiDAR system according to the invention enables reliable Doppler compensation for phase-modulated light signals. This makes it possible, in particular, to compensate for the large frequency shifts that are possible with the LiDAR measurement principle due to the Doppler effect, in order to continue to enable a correlation between the transmitted signal and the received signal, which in turn enables the light propagation time and thus the distance between the LiDAR system and the detected object to be determined allows. The structure of the LiDAR sensor system according to the invention remains slim, in particular a speed estimation can be carried out with little hardware effort. In addition, the use of phase codes enables improved parallelizability of the LiDAR sensor system, the possibility of detecting multiple targets and performing a reliable and unambiguous distance estimation.

The LiDAR sensor system has a transmission unit and a reception unit. The transmission unit in turn has a laser source, a phase modulator and transmission optics. The phase modulator serves to modulate a phase of the light from the laser source. The transmission optics are used to emit the light modulated by the phase modulator into an area surrounding the LiDAR sensor system. The light emitted by the transmitter unit into the environment thus has a phase code that is introduced by the phase modulator.

The receiving unit has receiving optics and an evaluation unit. The receiving optics are used to receive light from the environment. In particular, it is thus possible to receive reflected light that was emitted by the transmission unit and reflected on an object in the vicinity. The evaluation unit is used to evaluate the light received by the receiving optics, with the received light representing in particular a light signal.

The transmission unit is preferably also designed to emit a plurality of transmission sequences of the light. Each transmission sequence is a signal phase-modulated by the phase modulator and therefore has said phase code.

If a transmission sequence is sent out by the transmission unit and reflected on an object in the vicinity, then this reflection can be received by the reception unit and thus evaluated by the evaluation unit. The evaluation unit is designed to minimize an influence of the phase-modulated signal on the received light signal by increasing the power of the received light signal. The exponentiation is, for example, a squaring of the received light signal. In particular, an optimal exponentiation can be determined for each type of phase code used in the transmission sequence, which results in the influence of this phase code on the light signal being minimized, in particular disappearing.

As already described above, the received light signal is time-shifted, with the distance to the object reflecting the light being reflected in the time shift, and also frequency-shifted due to the Doppler effect. An FFT analysis of the received signal would lead to a difficult or impossible to detect Doppler frequency due to the spectrum of the phase code.

The evaluation unit is thus designed to determine the Doppler frequency by raising the received signal to a higher power and thereby eliminating the influence of the phase code on the received signal. As soon as the Doppler frequency is known, it is possible for the evaluation unit to carry out Doppler compensation easily and with little effort, and thus to determine a time shift and thus a distance to the object based on the Doppler frequency and the received light signal.

The dependent claims relate to preferred developments of the invention.

Provision is preferably made for the evaluation unit to be a simple demodulator. Alternatively or additionally, the evaluation unit is a complex demodulator. Even the use of a simple demodulator enables a reliable distance estimation and at least an absolute estimation of the speed of the detected object in the area. If a complex demodulator is used, it is also possible to estimate the speed of the object with the correct sign.

The evaluation unit particularly advantageously has a photodetector and evaluation logic. The evaluation logic includes, in particular, an analog/digital converter and a digital signal processor.

Each transmission sequence is preferably a unique code. The unique code has a unique autocorrelation function. In this way, a time shift in the transmission sequence can be identified on the basis of the cross-correlation of each received light signal with the transmission sequence originally transmitted. This time shift can in turn be used to determine the distance to the object in the area from which the transmission sequence was reflected. The time shift of the transmission sequence can be determined easily and with little effort using the Doppler compensation described above. The exponentiation of the signal provided for the Doppler compensation is particularly dependent on the significance of the phase code. If, for example, a phase code with a value of 2 is used, the exponentiation is advantageously a squaring.

The codes are particularly advantageously biphase codes, in particular Baker codes or maximum sequences, also known as maximum length sequences (MLS), or Gold codes or Kasami codes, or polyphase codes. All of these codes have a unique autocorrelation function. Said codes are therefore advantageously suitable for use as a transmission sequence.

Furthermore, it is particularly advantageously provided that the code has a duration of between 3 μs and 20 μs. In this way, in particular, a reliable Doppler estimation is made possible with a short measurement duration. The frequency resolution of the Doppler estimation depends on the transmission time of the code and can be brought within an optimal range with the specified values. This enables an accurate and reliable measurement with the LiDAR sensor system.

The evaluation unit is advantageously designed to raise the received light signal to a power by that factor which corresponds to a number of phase states of the code. The number of phase states of the code used for phase modulation is also considered the weight of the code. Thus, in the case of higher-order codes, a higher-order exponentiation is carried out. In this way, there is great flexibility when using the phase codes of the phase modulator, it also being ensured that the Doppler frequency can be determined simply and reliably. Thus, the Doppler compensation is made possible without having to restrict the phase codes to be used to a large extent.

The transmission unit is advantageously designed to transmit two transmission sequences with the same code but with a phase offset of 90°. In this case, the transmission unit is designed in particular to transmit the two transmission sequences directly one after the other or to transmit the two transmission sequences in an interleaved manner. In this case, the receiving unit, in particular the evaluation unit, is designed to separate the two transmission sequences again when the reflected light is received, in order to process them separately from one another. Due to the phase shift of 90°, an IQ demodulation can be carried out, with only a real demodulator having to be provided as the necessary hardware. The IQ demodulation thus takes place sequentially with the correspondingly received transmission sequences. This enables the provision of the functionality of a complex demodulator, ie, esp performs the signed Doppler estimation and thus the signed velocity estimation of the object in the environment without having to provide the actual hardware for a complex demodulator. This reduces the cost of manufacturing and assembling the LiDAR sensor system.

The transmission unit preferably has a number of parallel phase modulators in order to send out a number of transmission sequences in parallel. In particular, separate transmission optics are provided for each phase modulator. The transmission sequences transmitted in parallel are preferably coded orthogonally to one another by the respective phase modulators. It is thus possible to differentiate between each transmission sequence transmitted in parallel and the corresponding other transmission sequences transmitted in parallel. A parallel LiDAR sensor system is thus achieved. For this purpose, either a single receiving optics and a single evaluation unit is advantageously provided, with alternatively a plurality of receiving optics and evaluation units being present in order to thus also provide parallelization in the receiving unit.

The invention also relates to a vehicle. The vehicle has a LiDAR sensor system as previously described. This ensures safe and reliable detection of objects in the area surrounding the vehicle, with the costs for the vehicle being minimized due to the structure of the LiDAR sensor system described above.

character list

• 1 eine schematische Abbildung eines Fahrzeugs gemäß einem Ausführungsbeispiel der Erfindung,
• 2 eine schematische Ansicht einer ersten Alternative des LiDAR-Sensorsystems des Fahrzeugs gemäß dem Ausführungsbeispiel der Erfindung,
• 3 eine schematische Ansicht einer zweiten Alternative des LiDAR-Sensorsystems des Fahrzeugs gemäß dem Ausführungsbeispiel der Erfindung,
• 4 eine schematische Abbildung der Phasencodes zweier sequentiell ausgesandten Sendesequenzen mit 90° Phasenversatz,
• 5 eine schematische Abbildung der Phasencodes zweier verschachtelt ausgesandten Sendesequenzen mit 90° Phasenversatz,
• 6 eine schematische Ansicht einer dritten Alternative des LiDAR-Sensorsystems des Fahrzeugs gemäß dem Ausführungsbeispiel der Erfindung, und
• 7 eine schematische Ansicht eines Signalverarbeitungsplans zur Dopplerschätzung bei Verwendung des LiDAR-Sensorsystems gemäß der dritten Alternative.

Exemplary embodiments of the invention are described in detail below with reference to the accompanying drawings. In the drawing is:

• 1 a schematic illustration of a vehicle according to an embodiment of the invention,
• 2 a schematic view of a first alternative of the LiDAR sensor system of the vehicle according to the embodiment of the invention,
• 3 a schematic view of a second alternative of the LiDAR sensor system of the vehicle according to the embodiment of the invention,
• 4 a schematic representation of the phase codes of two sequentially transmitted transmission sequences with a 90° phase offset,
• 5 a schematic representation of the phase codes of two interleaved transmitted transmission sequences with a 90° phase offset,
• 6 a schematic view of a third alternative of the LiDAR sensor system of the vehicle according to the embodiment of the invention, and
• 7 a schematic view of a signal processing plan for Doppler estimation when using the LiDAR sensor system according to the third alternative.

Embodiments of the invention

1 FIG. 1 schematically shows a vehicle 9 according to an exemplary embodiment of the invention. The vehicle 9 in turn has a LiDAR sensor system 1 . Objects 11 in an environment 10 of the LiDAR sensor system 1 and thus of the vehicle 9 can be detected with the LiDAR sensor system 1 .

The LiDAR sensor system 1 has a transmission unit 3 and a reception unit 4 . The transmitter unit 3 is used to emit a light signal which is reflected by the object 11 so that this reflected signal can be received by the receiver unit 4 . Based on a time shift between the transmitted signal and the received signal, conclusions can be drawn about the light propagation time and thus about the distance between LiDAR sensor system 1 and object 11 . The occurring Doppler effect also enables the speed at which the object 11 is moving to be estimated.

In principle, it is provided that the LiDAR sensor system 1 works according to coherent phase code approaches, so that the phase of an emitted laser light is varied. A first alternative for the implementation of such a LiDAR sensor system 1 is in 2 shown. As described above, the LiDAR sensor system 1 according to the first alternative has the transmission unit 3 and the reception unit 4 . The transmission unit 3 comprises a laser source 2, a phase modulator 7 and a transmission optics 5. The reception unit 4 has a reception optics 6 and an evaluation unit 8.

In order to emit a light signal by means of the transmitter unit 3, it is initially provided that laser light is emitted by means of the laser source 2. This light is guided by a splitter 12 both to the receiving unit 4 and to the phase modulator 7 . The phase modulator 7 serves to modulate the phase of the light from the laser source 2 in order to thus generate a phase code. The light modulated in this way is then emitted by the transmission optics 5 into the environment 10 .

The emitted light is reflected by an object 11 in the surroundings 10 and thus reaches the receiving optics 6 of the receiving unit 4. From there, the received light signal reaches a coupler 13, which couples in the light from the laser source 2 branched off via the splitter 12. Finally, the received signal can be evaluated via the evaluation unit 8 . The evaluation unit 8 has a photodetector 8a and evaluation logic with an analog/digital converter 8b and a digital signal processor 8c.

To evaluate the received light signal, the received light signal is correlated with the phase code that was applied by the phase modulator 7 . For this purpose, it is provided that the phase code applied by the phase modulator 7 has a unique autocorrelation function. For example, the applied phase code is a biphase code, in particular a Baker code or maximum sequences or Gold codes or Kasami codes. Alternatively, polyphase codes can be used. A time shift of the received light signal with respect to the emitted light signal can thus be determined on the basis of the correlation. This time shift is characteristic of the light propagation time between the LiDAR sensor system 1 and the object 11. A distance d between the LiDAR sensor system 1 and the object 11 can thus be determined on the basis of the time shift.

However, the received light signal is not only time-shifted, but also Doppler-shifted in its frequency. The evaluation unit 8 is therefore designed to first carry out a Doppler estimation in order to then achieve Doppler compensation. For this purpose it is provided that the evaluation unit 8 is designed to minimize the influence of the phase code applied by the phase modulator 7 by raising the received signal to a power. The exponentiation occurs as a function of the phase code of the phase modulator 7 used. For example, in a biphase code, i.e. in a code with a value of 2, in which two different phase states can occur, the received signal is squared. In the case of higher-value codes, a higher exponentiation takes place, so that in the case of an n-value code, an exponentiation by n takes place. The influence of the phase code on the received signal is minimized by the exponentiation, as a result of which a reliable Doppler estimation can be carried out. This enables Doppler compensation, which then allows the time shift of the received signal to be determined. The evaluation unit is therefore designed to determine the time shift and thus the distance d of the object from the LiDAR sensor system 1 after the Doppler compensation. The movement speed v of the object 11 in the surroundings 10 can also be determined, at least in terms of amount, on the basis of the Doppler frequency.

k:

Digitale Laufvariable k ∈ ℕ0

Ts

Abtastintervall; entspricht invertiert der Abtastzeit: fs = 1/Ts

Ä:

Signalamplitude, beinhaltet Dämpfung über Signalpfad, Verstärkung im Mischprozess und durch Verstärker, etc.

ωD:

Kreisfrequenz der Doppler-Frequenz:

ω0τ:

Phasenverschiebung resultierend aus Time of Flight bezogen auf Kreisträgerfrequenz

n(k):

Zusammengefasstes diskretisiertes Rauschen (Schrotrauschen, thermisches Rauschen, Phasenrauschen)

φ(kTs- τ):

Zeitverschobene Codesequenz

When using the LiDAR sensor system 1 as in 4 shown with a real demodulator, this results in the following received signal C _RX : $$C_{RX}\left(k\right)={\widehat{A}}_c\cdot \text{cos}\left({\omega }_{D}k{T}_s+{\omega }_{0}T-\Phi \left(k{T}_s-\tau \right)\right)+n_c\left(k\right)$$

c:

Digital running variable k ∈ ℕ0

ts

sampling interval; corresponds inverted to the sampling time: fs = 1/Ts

Ä:

Signal amplitude, includes attenuation over the signal path, amplification in the mixing process and through amplifiers, etc.

ωD:

Angular frequency of the Doppler frequency:

ω0τ:

Phase shift resulting from time of flight related to circular carrier frequency

n(k):

Summarized discretized noise (shot noise, thermal noise, phase noise)

φ(kTs- τ):

Time-shifted code sequence

1. I. $\frac{A_c}{2}cos\left(2{\omega }_{D}k{T}_s+2{\omega }_0\tau -2\varphi \left(k{T}_s-\tau \right)\right):$
   
   Das Quadrieren erzeugt einen Kosinus-Term, der das Argument des ursprünglichen Kosinus Terms, multipliziert mit zwei, enthält. Da ein biphasiger Code verwendet wird, gilt 2Φ =konstant → Effektiv verschwindet also der Code aus diesem Term. Es bleibt ein Kosinus-Term, welcher mit der zweifachen Dopplerfrequenz f_D oszilliert. Dies ist der Nutzterm, welcher zur Doppler-Schätzung verwendet wird!
2. II. $\frac{{\overline{A}}_c}{2}:$
   
   Der zweite Term der ausmultiplizierten Klammer ist ein DC Anteil, welcher keine Information enthält. Dies bedeutet weiterhin, dass nur die Hälfte der Signalenergie in den Nutzterm geht.
3. III. $n_c^2\left(k\right):$
   
   Das Rauschen wird ebenso quadriert, was zu der bereits angesprochenen Performance-Degradation beim reellen Demodulator im Vergleich zum komplexen Demodulator führt. Starkes Rauschen wird also noch mehr verstärkt; das SNR sinkt.
4. IV. $2n_c\left(k\right)\cdot {\widehat{A}}_c\text{cos}\left({\omega }_{D}k{T}_s+{\omega }_0\tau -\varphi \left(k{T}_s-\tau \right)\right):$
   
   Kreuzterm zwischen ursprünglichem Signal und Rauschen, effektiv ein Rauschterm, welcher etwa die doppelte ,Stärke' besitzt wie das ursprüngliche Rauschen im nicht quadrierten Signal.

However, the received signal C _RX is not only time-shifted, from which the distance d can be derived, but also Doppler-shifted due to the reflection on the object 11 . The Doppler frequency f _D cannot be detected from this signal C _RX , or at least only with a great deal of computing effort. If an FFT analysis of the signal C _RX were carried out, a bandwidth B _C of the phase code would appear as additional noise and thus make it more difficult to detect the Doppler frequency f _D . As previously described, the signal C _RX is therefore exponentiated in order to enable the determination of the Doppler frequency f _D . The example used is based on biphase codes, which means that the phase codes have a value of 2. In this case, raising to the power is squaring. The squared signal is thus obtained as follows: $$\begin{array}{l}{c}_{RX}^2\left(k\right)=\frac{{\widehat{A}}_{\mathrm{<figure-callout\; id="2"\; label="c"\; filenames="DE102020215040A1\_0020.png,DE102020215040A1\_0021.png"\; state="\{\{state\}\}">c</figure-callout>}}}{2}\cdot \left[\text{cos}\left(2{\omega }_{D}k{T}_s+2{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0\tau -2\varphi \left(k{T}_s-\tau \right)\right)+1\right]+n_c^2\left(k\right)+2n_c\left(k\right)\\ \cdot {\widehat{A}}_c\text{cos}\left({\omega }_{D}k{T}_s+{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0\tau -\varphi \left(k{T}_s-\tau \right)\right)\end{array}$$

1. I $\frac{A_c}{2}cOs\left(2{\omega }_{D}k{T}_s+2{\omega }_0\tau -2\varphi \left(k{T}_s-\tau \right)\right):$
   
   The squaring produces a cosine term containing the argument of the original cosine term multiplied by two. Since a biphasic code is used, 2Φ = constant applies → the code effectively disappears from this term. A cosine term remains, which oscillates with twice the Doppler frequency f _D . This is the useful term that is used for Doppler estimation!
2. II. $\frac{{\overline{A}}_c}{2}:$
   
   The second term of the multiplied out bracket is a DC component which contains no information. This also means that only half of the signal energy goes into the useful term.
3. III. $n_c^2\left(k\right):$
   
   The noise is also squared, which leads to the aforementioned performance degradation in the real demodulator compared to the complex demodulator. Strong noise is thus amplified even more; the SNR decreases.
4. IV $2n_c\left(k\right)\cdot {\widehat{A}}_c\text{cos}\left({\omega }_{D}k{T}_s+{\omega }_0\tau -\varphi \left(k{T}_s-\tau \right)\right):$
   
   Cross term between original signal and noise, effectively a noise term which is about twice the 'strength' of the original noise in the unsquared signal.

Thus, as described above, the signal component I. enables the Doppler frequency f _D to be determined, with said determination being able to be carried out reliably and with little computational effort. If the Doppler frequency f _D is known, the evaluation unit 8 can also determine the distance d. This can also be done reliably as soon as the Doppler frequency f _D is known. The received signal C _RX is advantageously correlated with two Doppler-compensated templates. Alternatively, correlation with just one template is also possible. The variant with two code templates is shown below as an example:

The first code template is: $$c_{TC}\left(k\right)=\text{cos}\left({\varpi }_{D}k{T}_s-\varphi \left(k{T}_s\right)\right)$$

However, since the phase term ω ₀ τ is unknown, a cross-correlation is also performed with the sine version: $$c_{TC}\left(k\right)=\text{sin}\left({\varpi }_{D}k{T}_s-\varphi \left(k{T}_s\right)\right)$$

ω̃ _D describes the previously estimated Doppler frequency. The correlation can now be described mathematically for both templates as follows, where c _T stands for the cosine or the sine version of the template: $$R_{sc}\left(\mathrm{and}\right)={\displaystyle {\sum }_{k=0}^{N_{CS}}{c}_{RX}}\left(k\right)c_T\left(k-\mathrm{and}\right)$$

Here, Ncs is the length of the code sequence and results from N _CS = T _CS · ƒ _s , where Tcs is the total duration of the code sequence. The two correlation functions for the different template functions can then be summarized as follows: $$R_{SCGes}\left(\mathrm{and}\right)=\left|R_{SCsin}\left(\mathrm{and}\right)\right|+\left|R_{SCcOs}\left(\mathrm{and}\right)\right|$$

The frequency resolution Δƒ of the Doppler estimation results from the transmission duration of the phase-modulated signal. The frequency resolution Δƒ corresponds approximately to the reciprocal of this transmission time. This results in: $$\mathrm{\Delta }ƒ\approx 1\text{/}{T}_{CW}$$

The distance resolution Δd depends on the bandwidth Bc of the phase code and is approximately: $$\mathrm{\Delta }\mathrm{i.e}\approx {\mathrm{<figure-callout\; id="0"\; label="c"\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">c</figure-callout>}}_0\text{/}\left(2B_C\right)$$

In particular, transmission unit 3 is designed to transmit transmission sequences 100 (cf. 4 and 5 ) to emit the light signals with the introduced by the phase modulator 7 phase code. Such transmission sequences are preferably between 3 μs and 20 μs long. In this way, an optimal frequency resolution Δƒ for Doppler estimation can be achieved with a short measurement duration. A typical bandwidth of the phase code is in particular in the three-digit MHz range up to a range of a few GHz. This in turn enables an optimal range resolution Δd.

3 shows schematically a second variant of the LiDAR sensor system 1 of the vehicle 9 according to the exemplary embodiment of the invention. The transmission unit 3 is analogous to the first variant as previously described. There is a difference in the receiving unit 4, which is in 3 second variant shown has a complex demodulator 18, 19. This is particularly advantageous in the case of higher-order phase codes. For this purpose, a first evaluation unit 18 is provided for the I path and a second evaluation unit 19 for the Q path. The first evaluation unit 18 and the second evaluation unit 19 each have their own photodetector 8a and their own evaluation logic with an analog/digital converter 8b and a digital signal processor 8c.

In addition, the receiving unit 4 has a first additional splitter 15, which splits the reference signal from the splitter 12 of the transmitting unit 3 again in order to transmit it to the respective coupler 13a, 13b of the corresponding paths of the complex demodulator 18, 19. It is provided that a phase offset of 90° is introduced for the Q path. A second additional splitter 14 is also present, which splits the light received from the receiving optics 6 into the two paths of the complex demodulator 18, 19.

$$\text{Q}-\text{Anteil}:c_Q\left(k\right)={\widehat{A}}_Q\cdot \text{sin}\left({\omega }_{D}k{T}_s+{\omega }_0\tau -\varphi \left(k{T}_s-\tau \right)\right)+n_Q\left(k\right)$$

If a complex demodulator is used, then two baseband signals are present. These baseband signals are an I component and a Q component, which can be described as follows: $$\text{I}-\text{Portion}:c_I\left(k\right)={\widehat{A}}_I\cdot \text{cos}\left({\omega }_{D}k{T}_s+{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0\tau -\varphi \left(k{T}_s-\tau \right)\right)+n_I\left(k\right)$$

$$\text{Q}-\text{Portion}:c_Q\left(k\right)={\widehat{A}}_Q\cdot \text{sin}\left({\omega }_{D}k{T}_s+{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0\tau -\varphi \left(k{T}_s-\tau \right)\right)+n_Q\left(k\right)$$

The two real signals can be interpreted as a complex baseband signal (z=I+jQ), for this we assume that  ₁ =  _Q =  and define (n _I (k) + jn _Q (k)) = n $$c_c\left(k\right)=\widehat{A}{e}^{j\left({\omega }_{D}k{T}_s+{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0\tau -\varphi (k{T}_s-\tau \right)}\underset{\_}{n}$$

1. I. $\widehat{A}{e}^{2{\displaystyle \int \left({\omega }_{D}k{T}_s+{\omega }_0\tau -\varphi (k{T}_s-\tau \right)}}:$
   
   Dies ist das Nutzsignal. Wie beim reellen Demodualtor in 2 gezeigt und zuvor beschrieben,verschwindet der Code aufgrund der Multiplikation mit 2; man erhält wiederum eine Oszillation auf der zweifachen Dopplerfrequenz f_D. Des Weiteren bleibt die volle Signalenergie im Nutzsignal.
2. II. $2\underset{\_}{n}\cdot \widehat{A}{e}^{j\left({\omega }_{D}k{T}_s+{\omega }_0\tau -\varphi (k{T}_s-\tau \right)}:$
   
   Mischterm zwischen Signal und Rauschen. Genau wie beim reellen Demodulator nimmt die Bedeutung dieses Terms im Vergleich zum quadratischen Rauschterm mit geringer werdendem Input-SNR ab.
3. III. ${\underset{\_}{n}}^2=n_I^2-n_Q^2+2j{n}_I{n}_Q:$
   
   Quadriertes komplexen Rauschen. Zu der zuvor beschriebenen Variante des LiDAR-Sensorsystems 1 mit reellem Demodulator werden hier die beiden quadratischen Rauschanteile voneinander abgezogen, Je nach Korrelationsgrad erhält man somit eine Rauschunterdrückung, die DC Rauschterme werden in jedem Fall stark gedämpft. Simulationen legen nahe, dass immer eine gewisse Unterdrückung der quadratischen Teile stattfindet. Der Mischterm bleibt jedoch in jedem Fall erhalten, beeinflusst die Performance allerdings weniger dramatisch.

If this signal is now squared, one obtains: $$c_c^2\left(k\right)=\widehat{A}{e}^{2j\left({\omega }_{D}k{T}_s+{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0\tau -\varphi (k{T}_s-\tau \right)}{\underset{\_}{n}}^2+2\underset{\_}{n}\cdot \widehat{A}{e}^{{\displaystyle \int \left({\omega }_{D}k{T}_s+{\mathrm{<figure-callout\; id="0"\; label="\omega "\; filenames="DE102020215040A1\_0022.png"\; state="\{\{state\}\}">\omega </figure-callout>}}_0\tau -\varphi (k{T}_s-\tau \right)}}$$

1. I $\widehat{A}{e}^{2{\displaystyle \int \left({\omega }_{D}k{T}_s+{\omega }_0\tau -\varphi (k{T}_s-\tau \right)}}:$
   
   This is the useful signal. As with the real demutualtor in 2 shown and previously described, the code vanishes due to multiplication by 2; an oscillation at twice the Doppler frequency f _D is again obtained. Furthermore, the full signal energy remains in the useful signal.
2. II. $2\underset{\_}{n}\cdot \widehat{A}{e}^{j\left({\omega }_{D}k{T}_s+{\omega }_0\tau -\varphi (k{T}_s-\tau \right)}:$
   
   Mixed term between signal and noise. As with the real demodulator, the importance of this term decreases compared to the square-law noise term as the input SNR decreases.
3. III. ${\underset{\_}{n}}^2=n_I^2-n_Q^2+2j{n}_I{n}_Q:$
   
   Squared complex noise. In the previously described variant of the LiDAR sensor system 1 with a real demodulator, the two quadratic noise components are subtracted from one another here. Depending on the degree of correlation, noise suppression is thus obtained, the DC noise terms are strongly damped in any case. Simulations suggest that there is always some suppression of the quadratic parts. However, the mixed term is retained in any case, but affects the performance less dramatically.

The performance with such an IQ demodulator is increased compared to the previously described variant with a real demodulator. This follows due to the DC and improved noise rejection in the quadratic components of the signal. In addition, the complex demodulator or IQ demodulator enables a Doppler estimation with the correct sign and thus the determination of the speed v of the object 11 not only in terms of absolute value, but also with the correct sign.

After the Doppler estimation f _D has been carried out, the Doppler compensation and the determination of the distance d to the object 11 take place in the same way as previously described.

In 3 the variant of the LiDAR sensor system 1 with a complex demodulator is shown. As previously described, such a complex demodulator has advantages over a real demodulator. However, the hardware complexity is also increased. A further variant is therefore provided, in which sequential IQ demodulation is carried out with a real demodulator. This is in the 4 and 5 shown schematically.

As described above, the transmission unit 3 is designed to transmit multiple transmission sequences 100 . For this variant, it is provided that two transmission sequences 100a and 100b are sent, the second transmission sequence 100b having the same phase code as the first transmission sequence 100a, but a phase offset of 90°. The first transmission sequence 100a and the second transmission sequence 100b can optionally be transmitted sequentially, as in 4 shown, or can be sent nested or mixed or interleaved, as in 5 shown. Because of the time offset between the first transmission sequence 100a and the second transmission sequence 100b, the corresponding components can also be identified and separated from the signal reflected and received by the reception unit 4. This enables complex demodulation. In particular, the first transmission sequence 100a can be interpreted and further processed as an I part and the second transmission sequence 100b as a Q part. This enables IQ demodulation, with only a real demodulator being sufficient as the receiving unit 4 . This enables optimized demodulation while at the same time minimizing hardware costs. As an alternative to the time component as described above, a bandwidth can also be used in order to send out the first transmission sequence 100a and the second transmission sequence 100b. In this case, the transmission sequence 100a, 100b is sent out twice as fast as described above, but with half the chip length.

The signal processing described above, in particular the exponentiation of the received signal, is advantageously carried out in the digital domain.

As previously described, the digital signal processor 8c is present for this purpose. Alternatively, the exponentiation can also be implemented in hardware.

One advantage of the LiDAR sensor system 1 is that multiple targets 11 can be detected. If a number of targets 11 were to be hit by a transmission sequence 100, then a number of Doppler frequencies f _D can be determined. The individual distances d of the different objects 11 can then be determined iteratively by compensated correlation as previously described. The LiDAR sensor system 1 enables thus a reliable differentiation of several objects 11, the individual distances of these objects 11 being able to be reliably determined by the LiDAR sensor system 1.

6 finally shows another variant of the LiDAR sensor system 1, in which case the same receiving unit 4 as in 2 is used. The transmission unit 3 differs in that parallelization takes place. Such parallelization makes the LiDAR sensor system 1 particularly suitable for use in vehicles. Several transmission pixels are processed with the same reception channel, which saves costs.

The transmission unit 3 has a plurality of phase modulators 7a, 7b, 7c, each of the phase modulators 7a, 7b, 7c introducing a different code into the laser light of the laser source 2 by phase modulation. All of these codes are orthogonal to each other, so that a correlation of the different codes does not lead to a peak but only to noise. The signals encoded by the phase modulators 7a, 7b, 7c can be sent out via their own transmission optics 5a, 5b, 5c.

The transmission unit 3 can thus transmit several transmission sequences 100 at the same time, with 11 for example, the possibility of parallel transmission of three transmission sequences 100 is given. All of these transmission sequences 100 differ in the phase code applied by the respective phase modulator 7a, 7b, 7c, since the coding of the transmission sequences 100 is orthogonal to the other transmission sequences 100 transmitted in parallel.

In 6 is, as previously described, shown by way of example that all of these transmitted transmission sequences 100 are received and evaluated via a common receiving optics 6 and a common evaluation unit 8 . Alternatively, a large number of receiving optics 6 and/or evaluation units 8 can be provided here, so that each transmission sequence 100 generated in parallel can also be processed via its own receiving channel, ie via its own receiving optics 6 and its own evaluation unit 8 .

With such an embodiment of the LiDAR sensor system 1, a number of Doppler frequencies f _D can in turn be determined, with a Doppler frequency f _D being present for each of the parallel transmission sequences 100 and for each object 11 . At least one Doppler frequency f _D is assigned to each transmitted transmission sequence 100 if the corresponding transmission sequences 100 were reflected on an object 11 . In order to determine the associated distances of the corresponding transmission channels, a compensation of all codes of the phase modulators 7a, 7b, 7c used for the transmission sequences 100 can be used as the simplest variant with each detected Doppler frequency f _D . These compensated codes are then correlated with the received signals. If the Doppler frequency f _D and the code of the respective phase modulator 7a, 7b, 7c lead to a peak in the correlogram, the frequency and channel as well as the distance are defined via the peak position in the correlogram. Such a trial and error procedure is easy to implement.

In 7 a flow chart is shown of how such a method is to be carried out. In this case, K represents the number of detected Doppler frequencies f _D and N represents the number of transmission channels or codes, ie the number of frequency modulators 7a, 7b, 7c. At the start 300 of the method, run variables n and k are also each set to 1, with run variable n being a reference for the transmission channels or codes and k being a reference for the detected Doppler frequencies f _D . First there is a generation step 301, in which a code template with the Doppler frequency k and the phase code n is created. A correlation step 302 of the previously generated template with the received second section 102 then takes place. The further procedure is determined via a peak detection query 303 . If a peak was discovered, a storage step 304 takes place, in which the result found is stored. In addition, the control variable n is set to 1. The run variable k is then incremented and checked via a Doppler number query 305 . If the run variable k is greater than the number K, the process is complete and comes to its end 310. If this is not the case, however, further Doppler frequencies f _D must be checked, which is why the process returns to the generation step with the run variable k now incremented 301 runs.

However, if query 303 determines that no peak was detected, run variable n is incremented and a code number query 306 takes place. If run variable n is greater than number N, storage step 304 described above is carried out again and the procedure continues as previously described. If, on the other hand, the run variable n is not greater than the number N, the generation step 301 is carried out again, this time with an incremented run variable n.

The LiDAR sensor system 1 thus allows a safe and reliable determination of the distance and speed, at least one amount of the speed, of an object 11 in the vicinity of the LiDAR sensor system 1. Both the possibility of parallelization and the possibility of detecting multiple objects are possible 11 given. In addition, a construction of the LiDAR sensor system 1 is made possible in a simple and cost-effective manner. The LiDAR sensor system 1 is therefore ideally suited for use in vehicles due to its low production costs.

QUOTES INCLUDED IN DESCRIPTION

This list of the documents cited by the applicant was generated automatically and is included solely for the better information of the reader. The list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Patent Literature Cited

• WO 2018/144853 A1 [0002]

Claims (10)

LiDAR sensor system (1) having • a transmission unit (3) with a laser source (2), a phase modulator (7) for modulating a phase of the light from the laser source (2) and transmission optics (5) for emitting the light emitted by the phase modulator (7 ) modulated light in an area (10) of the LiDAR sensor system (1), • a receiving unit (4) with receiving optics (6) for receiving light reflected on an object (11) of the area (10) and with an evaluation unit ( 8) for evaluating the light received by the receiving optics (6), • the transmission unit (3) being designed to emit a plurality of transmission sequences (100) of the light, • each transmission sequence (100) being a phase-modulated signal by the phase modulator (7). , and • wherein the evaluation unit (8) is designed to minimize an influence of the phase-modulated signal on a received light signal by exponentiating the received light signal in order to thus determine a Doppler frequency (f _D ) from the received light signal stuff and based on the Doppler frequency (f _D ) and the received light signal to determine a distance to the object (11). LiDAR sensor system (1) according to claim 1 , characterized in that the evaluation unit (8) is a simple demodulator or has a complex demodulator (18, 19). LiDAR sensor system (1) according to one of the preceding claims, characterized in that the evaluation unit (8) has a photodetector (8A) and evaluation logic, in particular an analog/digital converter (8B) and a digital signal processor (8C). LiDAR sensor system (1) according to one of the preceding claims, characterized in that the transmission sequence (100) is a unique code, the codes having a unique autocorrelation function. LiDAR sensor system (1) according to claim 4 , characterized in that the codes are biphase codes, in particular Baker codes or maximal sequences or Gold codes or Kasami codes, or polyphase codes. LiDAR sensor system (1) according to claim 4 or 5 , characterized in that the code has a duration between 3 µs and 20 µs. LiDAR sensor system (1) according to one of Claims 4 until 6 , characterized in that the evaluation unit (8) is designed to raise the received light signal to a power by that factor which corresponds to a number of phase states of the code. LiDAR sensor system (1) according to one of Claims 4 until 7 , characterized in that the transmission unit (3) is designed to transmit two transmission sequences (100) with the same code but with a phase offset of 90°, the two transmission sequences (100) directly following one another or being interleaved. LiDAR sensor system (1) according to one of the preceding claims, characterized in that the transmission unit (3) has a plurality of parallel phase modulators (7A, 7B, 7C) in order to emit a plurality of transmission sequences (100) in parallel, with transmission sequences (100) transmitted in parallel the respective phase modulators (7A, 7B, 7C) are encoded orthogonally to each other. Vehicle (9) having a LiDAR sensor system (1) according to one of the preceding claims.

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