DE102020134241A1 - Quasi-resonant (QR) flyback converter with zero-voltage detection and fast signal response - Google Patents

Abstract

The present invention relates to a quasi-resonant (QR) flyback converter comprising a transformer circuit (112) having a primary winding (112-1) receiving an input voltage from the AC input (102), a secondary winding (112-2) receiving so is configured to provide an output voltage to the DC output (104), and includes an auxiliary winding (112-3) connected to a primary side zero crossing detection circuit (116-1) for detecting at least one zero voltage condition in the auxiliary winding (112- 3) and to record the time for it. A primary-side control circuit (132) that controls a switching element (126) so that the switching element sequentially turns ON and OFF the input voltage across the primary winding in a current cycle. A secondary-side control circuit (134) detects the level of the output voltage and/or the level of an output current supplied to the DC output and sends at least a second response signal to the primary-side control circuit if a change in the detected level of the output voltage and/or the detected level of the output current is a predetermined one exceeds threshold; wherein, in the event that the primary-side control circuit receives the at least one second response signal from the secondary-side control circuit, it is configured so that for a later cycle, the OFF duration until the input voltage is switched on again, regardless of the OFF duration of the current Cycle, to a preconfigured length of time in accordance with the received at least one second response signal.

Description

The present invention relates to a quasi-resonant QR flyback converter. More particularly, the invention relates to a QR flyback converter having a secondary-side transient load detection circuit that transmits a fast response signal to the primary-side control circuit that controls the QR flyback converter.

In the past, quasi-resonant (QR) flyback converters have been widely discussed due to their numerous advantages, e.g U.S. 2010/0219802 A1 . A simplified equivalent circuit diagram of such an example QR flyback converter is shown in 4a shown. Furthermore, the functioning of the exemplary QR flyback converter is shown in 4b shown.

In relation to 4a For example, the example QR flyback converter includes a QR regulator that controls a current in a transformer circuit T having a primary winding Lm, a secondary winding, and an auxiliary winding. For exemplary purposes, a turns ratio of N:1 between the primary winding Lm and the secondary winding is considered.

The primary winding Lm of the transformer circuit T in the exemplary QR flyback converter is supplied with an input voltage from a voltage supply Vin. The primary winding Lm is also connected to a switching element S1 (e.g. a MOSFET) which switches the current flowing in the primary winding. The switching element S1 is controlled by a QR controller according to a pulse width modulated (PWM) control signal with an adjustable period.

The example QR flyback converter also includes a stray inductance L_stray, which corresponds to the stray component of the primary inductance Lm, and a parasitic capacitance Cp, which corresponds to the parasitic effects on the drain of the switching element S1. Both the stray inductance L_stray and the parasitic capacitance Cp are responsible for parasitic effects during the operation of the exemplary QR flyback converter.

Also, in the exemplary QR flyback converter, the secondary winding of transformer circuit T provides a voltage to diode D1 that (after rectification) is stored on capacitor C1 to provide an output voltage V_OUT to a load. In addition, the auxiliary winding of the transformer circuit T provides a degaussing signal DEM which is received by the QR controller.

Regarding 4b the operation of the exemplary QR flyback converter is explained in terms of the changes over time in the voltage level across the switching element S1 (e.g. drain-source voltage Vds for a MOSFET).

At time t0, the QR controller controls the switching element S1 to assume an ON state. Accordingly, the input current in the primary winding Lm increases, and the transformer circuit T stores the energy in the form of the induced magnetic field.

Then, at time t1, the QR controller controls the switching element S1 to be in the OFF state. Accordingly, the energy stored in the primary winding Lm is transferred to the secondary winding of the transformer circuit T due to the inductive coupling between the windings.

At the same time, a reverse current is induced in the primary winding Lm of the transformer circuit T, so that the switching element S1 is subjected to a higher voltage (e.g. drain-source voltage for a MOSFET) than before. The magnitude of the peak voltage is determined by the leakage inductance L_stray, the voltage Vin and the reflected voltage Vr in the transformer circuit T. The voltage Vr is equal to N*V_OUT (at a turns ratio of N:1).

More precisely, the peak voltage across the switching element S1 (e.g. drain-source voltage for a MOSFET) oscillates due to the stray inductance L_stray, which opposes the parasitic capacitance Cp.

Then, at time t2, the inductive spike has died down and the voltage level across switching element S1 returns to voltage Vin plus the reflected voltage Vr, which corresponds to V1. In other words, at this point in time t2, the stray inductance L_Streu is (completely) demagnetized.

Subsequently, at time t3, the output current flowing through diode D1 falls to zero and in response the voltage level across switching element S1 begins to decrease. This voltage level at the switching element S1 falls between the times t3 and t4 to the level of the voltage Vin and even further to V2 due to resonance effects. At time t4, the demagnetization of the primary winding Lm ends.

The drop in the voltage level results from resonance effects, ie from energy oscillating in a resonance circuit made up of the primary winding Lm and the parasitic capacitance Cp, with the resonance period 2*tV being determined by Lm and Cp. As in 4b As shown, the oscillations may produce one or more troughs, with a first trough occurring at time t4 and a second trough occurring at time t5.

At one of these valleys, the QR controller starts again to control the switching element to assume an ON state. Since the voltage level across the switching element S1 (e.g. the drain-source voltage Vds for a MOSFET) is zero or a local minimum in the valleys, the switching efficiency improves at this time. For example, the first valley can be chosen because it represents the smallest local minimum.

However, since the first valley also implies the shortest possible cycle time, this mode of operation of the QR flyback converter dictates the transfer of the maximum amount of energy over time, regardless of whether it is required by the secondary connected load or not.

Advances have been made in the past to more flexibly control the operation of conventional QR flyback converters.

First, traditional QR flyback converters were adapted to include a voltage detector that senses the voltage Vin. Since the input voltage Vin determines the current in the primary winding Lm of the transformer circuit T, it also determines the energy that is stored in the primary winding Lm and subsequently transferred to the secondary winding of the transformer circuit T. The determined level of the voltage Vin can be used by the QR controller to select the appropriate operating mode.

Secondly, conventional QR flyback converters have been adapted to additionally include a feedback circuit which, on the secondary side of the transformer circuit T, generates, for example, the voltage V_OUT across a storage capacitor (see C1 in 4A ) detected. Accordingly, a (first) response signal that indicates the magnitude of the voltage V_OUT allows the QR flyback converter to adapt to the load.

According to the invention, each switching cycle first includes an ON period (referred to as tON) and then an OFF period (referred to as t_OFF).

Thus, when switching from OFF to ON at time t0, QR flyback converter operation transitions from a previous switching cycle (referred to as t_cycle -1) to another (subsequent) current cycle (referred to as t_cycle). With the switch from OFF to ON at time t5, the operation of the QR flyback converter transitions from the other cycle (referred to as t_ cycle) to yet another (subsequent) switching cycle (referred to as t_cycle +1).

Both advances have been shown to improve the flexibility of QR flyback converter operation, but are rather slow compared to fast load transients resulting from sudden load changes at the output of the traditional QR flyback converter.

In this context, the object of the invention is to enable a faster response in the operation of the QR flyback converter by additionally providing the primary-side controller of the QR flyback converter with a fast (i.e. second) response signal.

• eine Transformatorschaltung mit einer Primärwicklung, die so konfiguriert ist, dass sie eine Eingangsspannung von dem Wechselstromeingang empfängt, einer Sekundärwicklung, die so konfiguriert ist, dass sie eine Ausgangsspannung an den Gleichstromausgang liefert, und einer Hilfswicklung, die mit einer primärseitigen Nulldurchgangs-Detektionsschaltung verbunden ist, um mindestens einen Nullspannungszustand in der Hilfswicklung und deren Zeitpunkt zu detektieren; und
• eine primärseitige Steuerschaltung, die so konfiguriert ist, dass sie ein Schaltelement so steuert, dass das Schaltelement in einem aktuellen Zyklus die Eingangsspannung an der Primärwicklung nacheinander EIN- und AUS-schaltet, wobei die primärseitige Steuerschaltung ferner so konfiguriert ist, dass sie Folgendes umfasst
• ein Zeitpunkt (t0) zum Einschalten der Eingangsspannung ist der Zeitpunkt, an dem die Energieübertragung in den Transformator für den aktuellen Zyklus beginnt,
• ein Zeitpunkt zum Ausschalten der Eingangsspannung so eingestellt wird, dass ein in der Primärwicklung fließender Strom einen Spitzenstrompegel nicht überschreitet,
• ein Zeitpunkt für das erneute Einschalten der Eingangsspannung für den nächsten Zyklus synchron mit dem Zeitpunkt des erfassten mindestens einen Nullspannungszustands eingestellt wird, und
• wobei zusätzlich
• die AUS-Dauer bis zum erneuten Einschalten der Eingangsspannung eingestellt wird, indem die AUS-Dauer des Zyklus eines vorkonfigurierten Zeitintervalls entsprechend einem von einem sekundärseitigen Spannungsdetektor übertragenen ersten Antwortsignal variiert wird;
• ferner umfassend:
  • eine sekundärseitige Steuerschaltung, die so konfiguriert ist, dass sie den Pegel der Ausgangsspannung und/oder den Pegel eines dem Gleichstromausgang zugeführten Ausgangsstroms erfasst, und die so konfiguriert ist, dass sie mindestens ein zweites Antwortsignal an die primärseitige Steuerschaltung sendet, falls eine Änderung des erfassten Pegels der Ausgangsspannung und/oder des erfassten Pegels des Ausgangsstroms einen vorbestimmten Schwellenwert überschreitet; und
  • wobei für den Fall, dass die primärseitige Steuerschaltung das mindestens eine zweite Antwortsignal von der sekundärseitigen Steuerschaltung empfängt, diese so konfiguriert ist, dass sie für einen späteren Schaltzyklus die AUS-Dauer bis zum erneuten Einschalten der Eingangsspannung unabhängig von der AUS-Dauer des aktuellen Zyklus auf eine vorkonfigurierte Zeitdauer in Übereinstimmung mit dem empfangenen mindestens einen zweiten Antwortsignal einstellt.

According to a first aspect of the invention, a quasi-resonant (QR) flyback converter is proposed, which has an AC input and a DC output, and which comprises:

• a transformer circuit having a primary winding configured to receive an input voltage from the AC input, a secondary winding configured to provide an output voltage to the DC output, and an auxiliary winding connected to a primary side zero crossing detection circuit connected to detect at least one zero voltage condition in the auxiliary winding and the timing thereof; and
• a primary-side control circuit configured to control a switching element such that the switching element sequentially turns ON and OFF the input voltage to the primary winding in a current cycle, the primary-side control circuit being further configured to include the following
• a time (t0) to turn on the input voltage is the time when energy transfer into the transformer for the current cycle starts,
• a timing for switching off the input voltage is set so that a current flowing in the primary winding does not exceed a peak current level,
• a timing for turning the input voltage back on for the next cycle is set synchronously with the timing of the detected at least one zero voltage condition, and
• where additional
• the OFF duration until the input voltage is switched on again is adjusted by varying the OFF duration of the cycle of a preconfigured time interval according to a first response signal transmitted from a secondary-side voltage detector;
• also including:
  • a secondary-side control circuit configured to sense the level of the output voltage and/or the level of an output current supplied to the DC output, and configured to send at least a second response signal to the primary-side control circuit if there is a change in the sensed level of the output voltage and/or the sensed level of the output current exceeds a predetermined threshold; and
  • in the event that the primary-side control circuit receives the at least one second response signal from the secondary-side control circuit, this is configured so that for a later switching cycle, the OFF duration until the input voltage is switched on again, independently of the OFF duration of the current cycle to a pre-configured length of time in accordance with the received at least one second response signal.

It should be noted here that the setting of the OFF duration does not necessarily take place within an operating cycle of the primary-side control circuit (referred to as the control cycle), but only for the following control cycle. The later switching cycle, in which the OFF duration until the input voltage is switched on again, independently of the OFF duration of the current cycle, is set to a preconfigured time duration according to the received at least one second response signal, is therefore, for example, a switching cycle that is ten switching cycles after the switching cycle lies in which the second response signal is received.

According to an advantageous embodiment, the QR flyback converter further comprises a primary-side rectifier circuit connected between the AC input and the primary winding, and another secondary-side rectifier circuit connected between the secondary winding and the DC output, and/or a power factor correction (PFC) stage.

According to a further advantageous embodiment, the QR flyback converter further comprises a primary-side electrolytic capacitor configured to buffer the input voltage and a further secondary-side electrolytic capacitor configured to buffer the output voltage.

The at least one second response signal can advantageously be transmitted from the secondary-side control circuit to the primary-side control circuit via an optocoupler or a Y-capacitor. This ensures safe galvanic isolation of the primary and secondary sides.

Additionally, the primary side control circuitry may be configured to determine the peak current level for the following cycle depending on the level of the input voltage received from the AC input of the current cycle. Normally the peak current level remains a constant predetermined value. However, the level of the input voltage can be added as an offset to a shunt voltage.

Then the resulting peak voltage in each switching cycle is also dependent on the actually applied input voltage.

According to an advantageous embodiment, the primary-side control circuit is configured such that it synchronously adjusts the point in time for switching on the input voltage in a cycle so that it corresponds to the point in time of one of the at least one zero-voltage states detected by the zero-crossing detection circuit in the current cycle.

According to a further advantageous embodiment, the preconfigured time interval in the primary-side control circuit is configured according to a difference in the times between two successively detected zero-voltage states, and/or the preconfigured time period is configured in the primary-side control circuit according to the relative point in time of a predetermined one of the at least one detected zero-voltage states.

In particular, the pre-configured length of time in the primary-side control circuit can be configured depending on a level of the input voltage of the AC input.

Advantageously, the primary-side control circuit and optionally the secondary-side control circuit is a microcontroller.

According to a further aspect of the present invention, the primary-side control circuit is designed in such a way that, upon receipt of the at least one second response signal, it interrupts the control process for setting the switch-off duration for a subsequent cycle.

According to a further advantageous embodiment of the present invention, the primary-side control circuit is designed such that when a first of the at least one second response signal is received, the OFF duration of the subsequent cycle is set according to a first preconfigured time interval, and when a second of the at least one second response signal is received, the OFF duration of the subsequent cycle is set according to a second preconfigured time interval, the first preconfigured time interval being shorter than the second preconfigured time interval.

Advantageously, the first preconfigured amount of time may be configured in the primary side control circuit according to the relative timing of a first or second of the at least one sensed zero voltage condition within the previous cycle; and the second preconfigured amount of time is configured in the primary side control circuit according to the relative timing of a twentieth or thirtieth of the at least one sensed zero voltage conditions within the previous cycle.

According to a further advantageous aspect, the secondary-side control circuit is also designed to transmit a data signal to the primary-side control circuit.

In particular, the data can be transmitted from the secondary-side control circuit to the primary-side control circuit via an optocoupler or a Y-capacitor.

The data signal advantageously includes the level of the output voltage and the level of an output current which is detected by the secondary-side control circuit.

• 1 zeigt eine schematische Darstellung eines quasiresonanten (QR) Sperrwandlers 100 gemäß einer beispielhaften Ausführungsform der Erfindung;
• 2a und 2b zeigen ein Signaldiagramm des Betriebs des quasiresonanten (QR) Sperrwandlers 100 gemäß einer beispielhaften Ausführungsform der Erfindung;
• 3 zeigt ein schematisches Diagramm eines quasiresonanten (QR) Sperrwandlers 200 gemäß einer weiteren beispielhaften Ausführungsform der Erfindung; und
• 4a und 4b zeigen ein schematisches Diagramm und ein entsprechendes Signaldiagramm des Betriebs eines herkömmlichen QR-Sperrwandlers.

Moreover, several aspects of the embodiments - individually or in different combinations - can form solutions according to the present invention. Other features and advantages will be apparent from the following more detailed description of various embodiments of the invention as illustrated in the accompanying drawings, in which like references refer to like elements, and in which:

• 1 12 shows a schematic representation of a quasi-resonant (QR) flyback converter 100 according to an exemplary embodiment of the invention;
• 2a and 2 B 12 shows a signal diagram of the operation of the quasi-resonant (QR) flyback converter 100 according to an exemplary embodiment of the invention;
• 3 12 shows a schematic diagram of a quasi-resonant (QR) flyback converter 200 according to another exemplary embodiment of the invention; and
• 4a and 4b 12 shows a schematic diagram and corresponding signal diagram of the operation of a conventional QR flyback converter.

In the following a detailed description of different embodiments of the QR flyback converter according to the invention is given. In particular, all embodiments offer the advantage of quickly responding to load transients resulting from sudden changes in the load on the QR flyback converter output.

According to the invention, the transformer circuit of the QR flyback converter comprises a primary winding (or several windings) and a secondary winding (or several windings) with a predefined turns ratio Ns:Np. The primary and secondary windings are galvanically isolated from each other. The transfer of energy occurs from the primary to the secondary winding of the transformer circuit and vice versa through an inductive (or magnetic) coupling between the two.

Within the scope of the invention, reference is made to the primary-side and secondary-side circuits of the QR flyback converter. This term is used to denote the circuits placed either on the primary or on the secondary side of the isolation (galvanic) boundary of the transformer circuit (shown as a dashed line in the figures).

According to the invention, the transformer circuit also comprises an auxiliary winding (or several windings) with a predetermined turns ratio Na:Ns (or Na:Np) with respect to the secondary winding (or to the primary winding). These windings are also galvanically isolated from each other. In this respect, at least part of the energy, but not the main part of the energy, is absorbed by inductive coupling from the secondary winding (or the primary winding) into the auxiliary winding.

In the context of the invention, the QR flyback converter operates in cycles, each cycle including an OFF duration and an ON duration. In this regard, the time at which the QR flyback converter controls switching from ON to OFF corresponds to the beginning of a cycle. Accordingly, the timing at which the QR flyback converter controls switching from OFF to ON corresponds to an intermediate timing of the same cycle.

The following description refers to switching cycles (i.e. switching cycles of the switching element). In particular, reference is made to a previous cycle, a current cycle, and a subsequent cycle. Although these cycles can be contiguous, a subsequent cycle can also be non-consecutive.

For example, the previous cycle may refer to a cycle with cycle number N, the current cycle to a cycle with cycle number N+i (where i is an integer and greater than 0, e.g. 15) and the following cycle a cycle with cycle number N+j (where j is another integer greater than 0, e.g. 30).

Whether or not the previous cycle, the current cycle and the following cycle are contiguous depends mainly on the processing speed of the primary-side control circuit (and/or the processing speed of the secondary-side control circuit), which will be described in more detail below. From this it can be seen that the specific implementation of the primary or secondary-side control circuits affects the control behavior, e.g. B. with fast load transients can delay.

It is important that such a delay in the control behavior has no influence on the switching speed of the switching element, i. H. the speed at which the control circuit controls the switching element.

Additionally, the present disclosure relates to control cycles. A control cycle is a recurring sequence of work steps of a microcontroller, sometimes also referred to as the machine cycle of the microcontroller.

In an example implementation, the primary-side control circuitry includes separate (eg, dedicated) circuitry configured to control the switching element. This separate circuit includes, among other things, a voltage comparator (eg in the form of an operational amplifier), the voltage input outputs to the control circuit, eg the peak current value discussed above, with a reference voltage value and controls the switching element on this basis.

• In 1 ist eine beispielhafte Ausführungsform des quasiresonanten QR-Sperrwandlers 100 in Form eines schematischen Diagramms dargestellt. Dieses schematische Diagramm zeigt die Funktionsschaltungen des QR-Sperrwandlers als abstrakte Blöcke mit dazwischenliegenden Verbindungen, es ist also kein Schaltplan. Diese Darstellungsform wurde gewählt, um die Wechselwirkungen zwischen den Funktionsschaltungen besser sichtbar zu machen.

Referring now to an exemplary embodiment of a QR flyback converter 100:

• In 1 1, an exemplary embodiment of the quasi-resonant QR flyback converter 100 is illustrated in schematic diagram form. This schematic diagram shows the functional circuits of the QR flyback converter as abstract blocks with interconnections in between, so it is not a schematic. This form of representation was chosen to make the interactions between the functional circuits more visible.

The QR flyback converter 100 according to the exemplary embodiment includes an AC input 102 and a DC output 104. With the AC input 102, the QR flyback converter 100 can be connected to a mains supply. In addition, the QR flyback converter 100 with the DC output 104 can drive a DC load, such as. B. drive an industrial device in the field of industrial automation.

The AC input 102 is configured to receive an input voltage, for example in the range of 0-264 VAC, preferably one of 85 VAC, 120 VAC, 230 VAC, and 264 VAC input voltages. The DC output 104 is configured to provide an output voltage in the range of 0-110 volts DC, preferably one of 5 volts DC, 15 volts DC, 24 volts DC, and 110 volts DC.

The QR flyback converter 100 also includes a transformer circuit 112 with a fixed turns ratio between the primary and secondary windings (i.e., Ns:Np). Thus, without switching on and off (i.e. switching element 126), the level of the output voltage would be the product of the level of the input voltage times the turns ratio Ns:Np. However, by switching the switching element 126, which is also contained in the QR flyback converter 100, the level of the output voltage can be set variably.

In other words, despite the fixed Ns:Np turns ratio, QR flyback converter 100 is configured to provide variable output voltage levels. This is accomplished by the QR flyback converter 100 controlling the switching element 126 to transfer a variable amount of energy from the primary winding 112 - 1 to the secondary winding 112 - 2 of the transformer circuit 112 .

• In dem QR-Sperrwandler 100 ist der Wechselstromeingang 102 mit einer EMI-Filterschaltung 106 verbunden, die so konfiguriert ist, dass sie elektromagnetische Störungen auf der Eingangswechselspannung reduziert. Verschiedene Implementierungen der EMI-Filterschaltung 106 sind in der Technik bekannt, einschließlich richtig abgestimmter analoger Filterschaltungen, die die Störungen im niedrigen Band reduzieren, und Abschirmungsschaltungen, die dieselben Störungen im hohen Band reduzieren.

QR flyback converter 100 is described in more detail below:

• In the QR flyback converter 100, the AC input 102 is connected to an EMI filter circuit 106 configured to reduce electromagnetic interference on the AC input voltage. Various implementations of EMI filter circuit 106 are known in the art, including properly tuned analog filter circuits that reduce the low band noise and shielding circuits that reduce the same high band noise.

The EMI filter circuit 106 is connected to a rectifier circuit 108 which rectifies the AC input voltage (which has been filtered) into a constant input voltage. Various implementations of the rectifier circuit 108 are known in the art, including a full-wave rectifier illustrated in the figures by way of example. For reasons of clarity, the rectifier circuit 108 can also be referred to as a primary-side rectifier circuit 108 .

Alternatively, the rectifier circuit 108 can be connected directly to the AC input 102 (i.e., without the EMI filter circuit 106). Such a configuration might be advisable for a QR flyback converter when a lower part count has to be achieved or when production size and/or production cost constraints have to be met.

The rectifier circuit 108 is connected to an electrolytic capacitor 110 (abbreviated to elcap in the figures) which serves to buffer the (constant) input voltage which is then used in the transformer circuit 112 . Alternatively, the electrolytic capacitor 110 can also be replaced by a ceramic capacitor or another capacitor provided a decoupling effect is achieved. For reasons of clarity, the electrolytic capacitor 110 can also be referred to as the primary-side electrolytic capacitor 110 .

The electrolytic capacitor 110 is connected to a transformer circuit 112 which converts the energy of the (rectified) input voltage (which has also been decoupled) from the rectifier circuit 108 into energy which enables the DC output 104 to provide a (constant) output voltage for driving a load .

In addition, a power factor correction (PFC) stage can also be provided, as is known in the art. With power factor correction, the input current of offline power supplies is shaped in such a way that the real power available from the grid is maximized. The choice of power factor correction solutions ranges from passive circuits to a variety of active circuits. Depending on the power level and other specifics of the application, the appropriate solution will vary. Advances in discrete semiconductors in recent years and the availability of lower-cost control ICs have made active PFC solutions more suitable for a wider range of applications. However, any suitable PFC solution can be combined with the principles of the present disclosure.

Specifically, the transformer circuit 112 includes the primary winding(s) 112-1 connected to the rectifier circuit 108 with the electrolytic capacitor 110 therebetween. Accordingly, the primary winding 112-1 is configured to receive an input voltage (i.e., a constant input voltage). This input voltage is ultimately provided by the AC input 102, albeit as an AC input voltage, and has subsequently been rectified.

The transformer circuit 112 also includes a secondary winding (or windings) 112-2 connected to a further rectifier circuit 114 which provides a constant output voltage to the DC output 104. Accordingly, the secondary winding 112-2 is configured to ultimately provide an output voltage, albeit as an AC output voltage that is subsequently rectified. For the sake of clarity, the other rectifier circuit 114 can also be referred to as the secondary-side rectifier circuit 114 .

Further, the transformer circuit 112 includes an auxiliary winding (or windings) 112-3 connected to a zero crossing detection circuit 116-1.

This zero crossing detection circuit 116-1 is configured to detect at least one zero voltage condition (or valley) of the voltage provided by the auxiliary winding 112-3. In other words, it detects a condition where the auxiliary winding is in a demagnetized state (i.e., no current flows in the auxiliary winding 112-3). In addition, the zero crossing detection circuit 116-1 detects the time (t_Tal) of the at least one zero voltage condition (or valley).

For the sake of clarity, it should be noted that the zero voltage state can also be referred to as valley and the time as t_valley. Accordingly, in view of this terminology, it can be said that the turn-on of the input voltage to the primary winding 112-1 is controlled to occur at a time corresponding to the time of a trough, specifically synchronous with the time (t_Tal) of a trough previous cycle. By changing the number of valleys at which the input voltage turns on, the turn-off time t_OFF can be changed.

In an exemplary implementation, the zero crossing detection circuit 116-1 is configured to detect each zero voltage condition and their respective times in a cycle. In an alternative implementation, the zero crossing detection circuit 116-1 is configured to detect only the first zero voltage state and its timing during an OFF state of a switching element 126 in a cycle.

As already discussed in connection with the prior art, for the quasi-resonant operation of a QR flyback converter it is only necessary to detect the first zero voltage state (or valley) and its timing in each cycle, since the subsequent zero voltage states (or valleys) follow in a constant interval 2*tV, which corresponds to the resonant frequency caused by the inductance L (see L_stray in 4a ) of the primary winding 112-2 and the parasitic capacitance (see Cp in 4a ) of the switching element 126 is determined.

It is worth noting that the interval 2*tV is constant only under ideal conditions. In practice, it can be observed that the interval 2*tV slowly decreases with decreasing amplitude (e.g. the vibration frequency increases). However, this decrease is slow and negligible between successive valleys.

Consequently, even if the zero-crossing detection circuit 116 - 1 detects only the first valley, the timings of all subsequent OFF-state valleys can be derived based on the internal parameters of the QR flyback converter 100 . Alternatively, the zero crossing detection circuit 116-1 detects up to 20 zero voltage states and their respective times in one cycle.

In addition, the auxiliary winding 112-3 of the transformer circuit 112 is connected to an overvoltage protection circuit 116-2.

This overvoltage protection circuit 116-2 is configured to compare the level of the voltage across the auxiliary winding 112-3 with a predetermined threshold. When this threshold is exceeded, the overvoltage protection circuit 116-2 sends an overvoltage detection signal to the connected control circuit 132. For example, the control circuit 132 may (immediately) terminate all operations of the QR flyback converter 100 upon receipt of the overvoltage protection signal. However, since this process is not the focus here, a detailed description has been omitted for the sake of brevity.

The other rectifier circuit 114 is connected to another electrolytic capacitor 118 (abbreviated to elcap in the figures) that is configured to buffer the (constant) output voltage that is ultimately fed to the DC output 102 . Alternatively, the electrolytic capacitor 118 can also be replaced by a ceramic capacitor or another capacitor, provided that a decoupling effect is achieved. For the sake of clarity, the other electrolytic capacitor 118 can also be referred to as the secondary-side electrolytic capacitor 118 .

More specifically, the (constant) output voltage from the other rectifier circuit 114 is fed to the DC output 102 via the other electrolytic capacitor 118 , a filter circuit 120 and a shunt resistor 122 .

The filter circuit 120 is configured as a low-pass filter for the output voltage used as a (first) response signal for operating the QR flyback converter 100 .

In an example implementation, the filter circuit 120 is a resistor-capacitor, RC, low-pass filter circuit or an inductor-capacitor, LC, low-pass filter circuit configured to attenuate high-frequency transients in the output voltage. The cutoff frequency of the low-pass filter circuit can be configured to suppress switching transients in the output voltage that result from the input voltage across the primary winding 112 - 1 of the transformer circuit 112 continuously switching off and on. For this purpose, the cut-off frequency z. B. be 100 Hz.

For the sake of clarity, it should be emphasized that this filter circuit 120 is a cause of a slow response when the QR converter 100 operates only on the basis of the (first) response signal. On the one hand, the filter circuit 120 must provide a long averaging window for the (oscillating) output voltage in order to achieve good filtering results, on the other hand, the filtering introduces a significant delay in the first response signal, which the embodiments attempt to solve.

The filter circuit 120 is connected to a voltage detector 124 which detects the level of the output voltage for voltage regulation purposes (referred to as the V regulation circuit 124 in the figures). Specifically, the voltage detector 124 is configured to sense (or adjust) the level of the low-pass filtered output voltage and transmit a first response signal corresponding to the sensed level to the primary side of the QR flyback converter 100 .

The voltage detector 124 is another reason for a slow response when the QR flyback converter 100 operates solely on the (first) response signal. Voltage detector 124 is typically set to operate at a slow regulation rate to avoid oscillatory regulation of QR flyback converter 100, particularly in situations where QR flyback converter 100 is designed with a high output capacitance.

In an example implementation, the voltage detector 124 is an operational amplifier that amplifies/attenuates the output voltage (after filtering) to convert it into the first response signal that corresponds to the level of the output voltage.

Furthermore, the filter circuit 120 is also connected to the shunt resistor 122, which enables the detection of the output current, which corresponds at least in part to the current induced in the secondary winding 112-2 of the transformer circuit 112 and (subsequently) flows to the DC output 104, in particular the output current flowing to a load to be connected to QR flyback converter 100 .

It is clear that the placement of the shunt resistor 122 between the filter circuit 120 and the DC output 104 is only one of several possibilities. For example, shunt resistor 122 can be connected not only in the forward (plus) path to DC output 104, but also in the reverse (negative) path from DC output 104 to secondary winding 112-2. In both paths, the shunt resistor 122 enables the output current induced in the secondary winding 112-2 to be sensed.

In the transformer circuit 112, the primary winding 112 - 1 is also connected to the switching element 126 . The switching element 126 is configured to sequentially switch the input voltage to the primary winding 112-1 OFF and ON. In other words: the switching element 126 switches the current flowing in the primary winding 112-1. In an example implementation, the switching element 126 is a MOSFET, an IGBT, a high voltage NPN transistor, or other power switching field effect transistor (FET).

When switching element 126 is in an ON state, current from AC input 102 flows through EMI filter circuit 106, rectifier circuit 108, charges or bypasses electrolytic capacitor 110, and flows into primary winding 112-1. From this primary winding 112-1, the same current flows through the switching element 126 in the ON state (i.e. in the conducting state) via a further shunt resistor 128 to ground.

In addition, when switching element 126 is in the OFF state, current can flow from AC input 102 through EMI filter circuit 106 and rectifier circuit 108 to charge electrolytic capacitor 110 . However, current cannot flow into primary winding 112-1 because it cannot exit primary winding 112-1 when switching element 126 is in the OFF state (i.e., non-conductive state).

Accordingly, the switching element 126 turns ON or OFF the input voltage provided by the AC input 102 in the primary winding 112-1, thereby allowing or blocking current flow in the primary winding 112-1. In short, the switching element 126 switches ON and OFF the input voltage to the primary winding 112-1.

Furthermore, a snubber circuit 130 is connected in parallel to the primary winding 112-1 between the node connecting the primary winding 112-1 to the switching element 126 and the node connecting the electrolytic capacitor 110 to the primary winding 112-1. The snubber circuit 130 is configured to prevent voltage spikes when the switching element 126 is turned off. In addition, the snubber circuit 130 improves electromagnetic compatibility. In an exemplary implementation, the snubber circuit 130 is a series combination of a diode and a capacitor, both connected in parallel with a resistor, with the diode drawing current from the primary winding 112-1 into the electrolytic capacitor 110 when switching from an ON to an OFF state directs.

Finally, the QR flyback converter 100 includes a control circuit 132 on the primary side and another control circuit 134 on the secondary side. Both the primary-side control circuit 132 and the secondary-side control circuit 134 are advantageously embodied as microcontrollers (abbreviated to μC in the figures). Due to the isolation limit, both microcontrollers are connected to separate linear voltage regulators 136 and 138, each providing a supply voltage (e.g. +5 volts).

Advantageously, the use of microcontrollers with low operating speeds allows for a significant reduction in manufacturing cost of the QR flyback converter 100 without compromising the fast response to load transients resulting from changes in the loading of the QR flyback converter output, as will become apparent below.

• Die Steuerschaltung 132 ist mit dem Elektrolytkondensator 110 verbunden. Damit ist die Steuerschaltung 132 in der Lage, einen Pegel der Eingangsspannung zu erfassen. Der gleiche Eingangsspannungspegel wird (nacheinander) an die Primärwicklung 112-1 angelegt, nämlich beim Einschalten des Schaltelements 126. Insbesondere erfasst die Steuerschaltung 132 die gleichgerichtete Eingangsspannung, die von der Gleichrichterschaltung 108 ausgegeben wird, und nicht die Eingangswechselspannung, die vom Wechselstromeingang 102 empfangen wird.

The control circuit 132 will now be discussed in more detail:

• The control circuit 132 is connected to the electrolytic capacitor 110 . With this, the control circuit 132 is able to detect a level of the input voltage. The same input voltage level is (sequentially) applied to the primary winding 112-1, namely when the switching element 126 turns on. In particular, the control circuit 132 detects the rectified input voltage that is output from the rectifier circuit 108, and not the AC input voltage that is received from the AC input 102 .

In addition, the control circuit 132 is also connected to the shunt resistor 128 . Thus, the control circuit 132 is able to detect a current flowing through the switching element 126 (and the primary winding 112-1) when the switching element 126 is turned on. In other words, the control circuit 132 senses the current flowing through the primary winding 112-1 and the switching element 126 when it is in the ON state (i.e., conductive state).

Sensing the input voltage level and/or sensing the current flowing in the primary winding 112 - 1 allows the control circuit 132 to determine the amount of energy being transferred from the primary winding 112 - 1 to the secondary winding 112 - 2 of the transformer circuit 112 . This has the beneficial effect that QR flyback converter 100 can convert different (i.e., varying) input voltages to an equal or different output voltage.

Based on the sensed current, the control circuit 132 may limit the amount of energy to be transferred in the transformer circuit 112 by continuously checking that the current does not exceed a peak current level. To do this, the control circuit 132 repeatedly compares the current detected via the shunt resistor 128 with a predefined peak current level and controls the switching off of the switching element 126 before the peak current level is exceeded. The amount of energy transmitted is controlled by shifting the first response signal added to the shunt signal.

Specifically, the control circuit 132 may be configured to determine the peak current level for the subsequent cycle depending on the magnitude of the input voltage received from the AC input of the current cycle. This allows the control circuit 132 to determine a peak voltage level required to transfer an equal amount of energy to the secondary winding 112-2 over varying input voltages.

In particular, the control circuit 132 is configured to determine the peak current level for the following cycle depending on the magnitude of the voltage across the electrolytic capacitor 110 . This voltage across the electrolytic capacitor 110 corresponds to the input voltage. Importantly, the peak current level determined by the control circuit 132 prevents the primary winding 112-1 from being driven into saturation.

For example, if the control circuit 132 detects a lower level of the input voltage, the control circuit 132 can adjust the t_ON time of the switch to a longer ON time according to the feedback signal offset added to the shunt signal, so that the same amount of energy is transferred from the primary winding 112-1 to the secondary winding 112 -2 is transmitted. In another example, when the control circuit 132 detects an increased input voltage level, it may adjust the switch tON time to a shorter ON time, again allowing the same amount of energy to be delivered over a period of multiple switching cycles (e.g., at least 10 cycles) is transmitted. The peak current level always remains the same and is used by the microcontroller for the comparison.

More specifically, when the control circuit 132 controls the switching element 126 to switch to the ON state, a current begins to flow through the primary winding 112-1, the switching element 126 and the shunt resistor 128 and increases to a peak current level.

Once the control circuit 132 detects a current corresponding to the peak current using the shunt resistor 128, the control circuit 132 controls the switching element to an OFF state. In other words, the determined peak current determines the point in time at which the control circuit 132 switches off the input voltage at the primary winding.

More specifically, the control circuit 132 controls the ON state of the switching element 126 based on the predetermined peak current level and the sensed current flowing through the shunt resistor 128 flows. The control circuit 132 includes, for example, a comparator (e.g., in the form of an operational amplifier) that compares the predetermined peak current level to the sensed current.

When the switching element 126 turns ON, no current is yet flowing through the shunt resistor 128. The comparison of the detected current to the predetermined peak current level therefore produces a positive output. In this case, the control circuit 132 drives the switching element 126 to remain in the ON state.

After the switching element 128 is turned ON, a slowly increasing current begins to flow through the primary winding 112-1 and thus through the shunt resistor 128 as well. As long as the comparison of the sensed current to the predetermined peak current level produces a positive output signal, the control circuit 132 controls the switching element 126 to remain in the ON state.

At a certain point in time, the slowly increasing current flowing through the primary winding 112-1 and the shunt resistor 128 exceeds the predetermined peak current level. Then, the comparison result becomes a negative output, and the control circuit 132 controls the switching element 126 to turn to the OFF state. Thus, by comparing the current through the shunt resistor 128 to a predetermined peak current level, the control circuit 132 controls the duration of the ON state.

The above description refers to the simplified case that (only) the current flowing through the shunt resistor 128 is detected by the control circuit 132 . For QR flyback converter 100, the current sensed by control circuit 132 corresponds to the current through shunt resistor 128 and has an offset superimposed thereon. The shift corresponds to an amplified version of the first response signal.

The shunt resistor 128 is connected to a feedback circuit 142 for this purpose. The feedback circuit 142 provides the amplified version of a first response signal (discussed further below). This output signal is combined (e.g., superimposed) with the signal corresponding to the current flowing through the shunt resistor 128 . In other words, the control circuit 132 detects a combination of the current flowing through the shunt resistor 128 shifted by the first response signal.

The first response signal can preferably assume values between 0V and 5V. This voltage is adjusted to the shunt signal via a voltage divider. The higher the offset, the more the turn-on time is reduced, and the lower the offset, the longer the turn-on time of the switch. if e.g. For example, when the first response signal is high, the current flowing through the shunt resistor 128 is increased by a large offset due to the interference. The control circuit 132 detects a current in excess of the predetermined peak current level much sooner. Therefore, the control circuit 132 drives the switching element 126 to switch from the ON state to the OFF state much earlier. The ON duration becomes shorter.

In another example, when the first response signal has a low value, the sensed current flowing through the shunt resistor 128 is reduced by a low offset due to the interference. The control circuit 132 does not detect a current in excess of the predetermined peak current level until much later. Therefore, the control circuit 132 controls the switching element 126 to switch from ON to OFF state much later. The ON duration becomes longer. In other words, the first response signal is the offset superimposed on the shunt signal and causes the turn-off threshold "peak current level" to be reached earlier, so that the switch 126 turns off earlier. The detection of the switch-off condition always follows the relationship below: $$\text{V}\_\text{IP}=\text{V}\_\text{shunt}\left[\text{f}\left(\text{I}\_112-1\right)\right]+\text{V}\_\text{shift}\left[\text{f}\left(\text{R}\ddot{\text{and}}\text{feedback}142\right)\right]>\text{peak current threshold}$$

Importantly, the filter circuit 120 and voltage detector 124 are slow and provide only a relatively small offset in the form of the first response signal. In addition, the control circuits require several control cycles to reset the time values when necessary. Therefore, this control does not allow sufficiently fast control response to load transients.

The control circuit 132 is also connected to a gate driver circuit 140 . The control circuit 132 is thus able to control the switching element 126 . This is necessary when the tax Circuit 132 itself is unable to (directly) supply switching element 126 with a suitable range of control voltages. In an alternative configuration, the control circuit 132 is configured to drive the switching element 126 directly.

In addition, the control circuit 132 is connected to the zero crossing detection circuit 116-1. Thus, the control circuit 132 is able to drive the switching element 126 in one cycle so that the timing for turning on the input voltage is adjusted synchronously with the timing of the at least one zero-crossing state (or valley) detected by the zero-crossing detection circuit 116- 1 is detected.

In other words, the control circuit 132 is configured to drive the switching element 126 in one switching cycle to switch the input voltage from OFF to ON (i.e. define the OFF duration) when a zero voltage (or valley) condition is present. namely, when the primary winding 112-1 is in a demagnetized state. In other words, the timing for turning on the input voltage is determined in synchronism with the troughs detected by the zero-cross detection circuit 116-1 in the cycle. By changing the number of valleys at which the input voltage turns on, the turn-off time t_OFF can be changed. Such a change in the number of valleys can be made by the control circuit 132 during a control cycle for one of the following control cycles.

In the context of the invention, the term "synchronous" is to be understood in such a way that it refers to a (specific) point in time that corresponds to the point in time (see t_Tal in 4b ) corresponds to one of the at least one zero-stress states or valleys occurring in the cycle (see t_cycle in 4b ) can be detected by the zero crossing detection circuit 116-1.

For example, if the zero crossing detection circuit 116-1 detects only a single zero voltage condition or valley in the cycle, then the term "synchronous" is clearly understood to refer to that (specific) point in time. For example, if the zero crossing detection circuit 116-1 detects multiple, for example 10, zero voltage states in the cycle, then the term "synchronous" can be understood as referring to each of these (specific) points in time.

This improves the switching efficiency of the QR flyback converter 100 compared to switching at various times with no zero voltage (or valley) condition.

If the primary winding 112-1 is in a demagnetized state at the time of switching and the zero crossing detection circuit 116-1 detects a zero voltage condition, the switching element 126 will experience a minimum voltage level reflected from the secondary winding 112-2 to the primary winding 112-1 becomes. Thus, when switching element 126 turns on, only the minimum voltage level is discharged to ground, resulting in improved switching efficiency.

When the control circuit 132 is configured to switch the switching element 126 synchronously with the detected zero voltage condition, this is commonly understood as quasi-resonant operation, i. H. as specifying the operation of the QR flyback converter 100. In other words, the term quasi-resonant flyback converter, QR, already—inherently—dictates the flyback converter of the present exemplary embodiment that it supports quasi-resonant operation.

Again, for the sake of clarity, quasi-resonant operation of a flyback converter does not dictate the exact time (i.e., valley) at which the switching element 126 is controlled to switch from OFF to ON. In other words, since there are numerous repetitive zero voltage states (or troughs) for switching the switching element 126 at the various instants of the zero voltage states, the precise instant in the control circuit 132 must be further determined.

To this end, the control circuit 132 receives a first response signal (referred to as feedback in the figures) which is detected on the secondary side of the transformer circuit 112 . Depending on this first response signal, the control circuit 132 determines the OFF duration of the input voltage at the switching element 126, i. H. the duration that no current flows through the primary winding 112-1, thereby determining the amount of energy transferred to the secondary winding 112-2.

Furthermore, the control circuit 132 controls the duration of the switch-off, i.e. until it is switched on, synchronously with the time (t Tal in 4b ) of the at least one zero voltage state (or valley) detected by zero crossing detection circuit 116-1. In other words: In addition to the first response signal, the control circuit 132 controls the switching element 126 such that the QR flyback converter is always in quasi-resonant operation, ie switching only takes place when the primary winding 112-1 is in a demagnetized state .

This is achieved by the control circuit 132 controlling the switching element 126 to switch from OFF to ON (definition of OFF duration) by varying the OFF duration in increments or decrements of a pre-configured time interval, e.g. B. in increments or decrements of 2*tV. This variation in the OFF duration is determined by the control circuit 132 in response to the first response signal. For example, the input voltage turn-on time can be changed from turn-on at the first valley to turn-on at the third valley. Several control cycles are required for such a change in the switch-on time, since the values stored in the respective time units must be set to the desired new values.

For example, if the control circuit 132 receives a first response signal indicative of a reduction in the output voltage level, then the control circuit 132 determines that the OFF duration should be increased by the preconfigured time interval, e.g. 2*tV, while the control circuit 132, upon receiving a first response signal indicative of an increase in the output voltage level, determines that the OFF duration is increased by the (same) preconfigured time interval, e.g. B. 2*tV, is increased.

Following this example, the control circuit 132 may decrease the OFF duration as follows. Assuming the OFF duration has been determined to correspond to the tenth zero voltage state or valley, the control circuit 132 may decrease the OFF duration to correspond to the ninth zero voltage state or valley. This reduction corresponds to the pre-configured time interval, e.g. B. 2*tV. Changing the point in time for switching the input voltage on again takes several control cycles, so that the rapid response signal according to the invention is required.

Also in this example, the control circuit 132 can increase the off-time as follows. For example, assuming the OFF duration was determined to correspond to the fifteenth zero voltage state or valley, the control circuit 132 may increase the OFF duration to correspond to the sixteenth zero voltage state or valley. This increase in turn corresponds to the pre-configured time interval, e.g. B. 2*tV.

The time interval (e.g. 2*tV in 4b ) can be preconfigured to match the calculated resonant frequency caused by the inductance L (see L_stray in 4a ) of the primary winding 112-2 and the parasitic capacitance (see Cp in 4a ) of the switching element 126 is determined. Alternatively, the time interval can be preconfigured to correspond to the time duration (time difference) between the points in time (t_Tal) of a certain number, e.g. B. corresponds to two (or more), consecutive zero voltage conditions detected by the zero crossing detection circuit 116-1.

Coming back to the example above, the time interval is preconfigured to correspond to the time difference between the instants (t_Tal) of (precisely) two consecutive zero voltage states, namely the time interval 2*tV. Then the control circuit 132 increases/decreases the OFF duration by adding/subtracting the time difference between the instants of the (precisely) two consecutive zero voltage states.

In an alternative implementation, control circuit 132 includes a counter circuit that counts the zero voltage states (or valleys) detected by zero crossing detection circuit 116-1 for each cycle and compares the total number of zero voltage states counted to the specified target number of zero voltage states. When the control circuit 132 determines that the target number of zero voltage states has been reached, it controls the switching element 126 to change from the OFF state to the ON state (OFF duration).

In this implementation, the control circuit 132 increases/decreases the OFF duration by varying the target number of zero voltage states. When the control circuit 132 receives a first response signal indicative of a decrease in the output voltage level, the control circuit 132 decreases the OFF duration by decreasing the target number of zero voltage states according to the pre-configured number. The control circuit 132 receives a first response signal indicating an increase in output voltage level, then the control circuit 132 increases the OFF duration by increasing the target number of zero voltage states according to the preconfigured number.

Since the time interval (e.g. 2*tV in 4b ) is preconfigured in accordance with the calculated or detected instants of the zero-voltage states, it is consequently ensured that the control circuit 132 turns on the input voltage consistently (only) synchronously at the different instants of the zero-voltage states, even if the OFF-duration compared to the previous cycle according to the first Response signal is varied.

Specifically, the control circuit 132 receives the first response signal from the voltage detector 124 on the secondary side of the transformer circuit 112. The voltage detector 124 detects the level of the output voltage on the secondary side and then forwards a corresponding first response signal via an optocoupler 144 to the primary side of the transformer circuit 112.

The optocoupler 144 serves to maintain the isolation (galvanic) boundary between the primary and secondary sides of the transformer circuit 112. However, this is just one of several alternative implementations for propagating the first response signal across the isolation (galvanic) boundary. The first response signal can also be transmitted via a Y-capacitor, for example.

The first response signal is received by the control circuit 132 on the primary side. In particular, the first response signal is first received by a feedback circuit 142 . The first response signal can have a voltage level between 0V and 5V due to the supply voltage (usually +5V) of the optocoupler 144 . The feedback circuit 142 bypasses a voltage divider to generate an offset signal for the shunt voltage, as will become apparent as the description proceeds. Optionally, the feedback circuit 142 may also amplify and/or filter the first response signal of the optocoupler 144 and then provide the conditioned signal to the control circuit 132, thereby reducing oscillations and unwanted transient signals. This can ensure that the first response signal has a level range that is suitable for detection by the control circuit 132 .

In addition, the feedback circuit 142 is connected to the shunt resistor 138 . With this connection, the feedback circuit 142 provides the (optionally conditioned) version of the first response signal as an offset upon sensing the current flowing through the shunt resistor 128 . In other words, the first response signal shifts the current to be sensed by the control circuit 132, thereby adjusting the ON duration, namely when the control circuit 132 controls the switching element 126 from ON to OFF.

However, the first response signal is slow (e.g., due to the first filter circuit 120 and due to processing in the voltage detector 124) and is unable to reflect fast load transients at the DC output 104. Accordingly, with this first response signal, the control circuit 132 can only imprecisely (i.e. with a large time delay) determine the point in time at which the input voltage at the primary winding 112-1 is switched on. In addition, the operations within the microcontroller that forms the control circuit 132 are also slower than the switching cycles. As already mentioned, changing the moment at which the input voltage is switched on requires several control cycles.

In order to more accurately (e.g., without delay) determine when to turn on the input voltage, the control circuit 132 receives a rapid (second) response signal from the other control circuit 134 on the secondary side. The fast (second) response signal allows faster control than with the first response signal alone.

For the sake of clarity, it should be emphasized in this connection that the second response signal, which is transmitted from the other control circuit 134 to the control circuit 132, cannot be regarded as a substitute for the first response signal in the present embodiment. Rather, this embodiment is based on the understanding that the second response signal complements the first response signal in certain situations, namely in the case of fast load transients.

• Die weitere Steuerschaltung 134 ist auf der Sekundärseite der Transformatorschaltung 112 vorgesehen und ist dazu ausgebildet, die Steuerschaltung 132 auf der Primärseite zu unterstützen. Insbesondere unterstützt die Steuerschaltung 134 der Sekundärseite die Steuerschaltung 132 beim Auftreten schneller Lasttransienten. Zu diesem Zweck ist die Steuerschaltung 134 kommunikativ mit der Steuerschaltung 132 gekoppelt und stellt diesem ein zweites Antwortsignal zur Verfügung.

Now to the control circuit 134 in detail:

• The further control circuit 134 is provided on the secondary side of the transformer circuit 112 and is configured to support the control circuit 132 on the primary side. Especially secondary side control circuit 134 assists control circuit 132 in the event of fast load transients. For this purpose, the control circuit 134 is communicatively coupled to the control circuit 132 and provides it with a second response signal.

In particular, the control circuit 134 is connected to the control circuit 132 via Y-capacitors 146, 148, with each of the Y-capacitors 146, 148 forwarding a separate one of the at least one second response signal from the control circuit 134 to the control circuit 132. In an alternative implementation, the connection can also be realized via separate optocouplers or other means as long as the (galvanic) isolation limit is maintained.

Specifically, the secondary side control circuit 134 is connected to the DC output 104 and configured to sense a level of the output voltage provided to the DC output 104 by the secondary winding 112 - 2 . The sensed output voltage level corresponds to that at the DC output 104. In addition, the control circuit 134 is connected to the shunt resistor 122 and is configured to sense a level of output current induced in the secondary winding 112-2 and thus also to the DC output 104 is delivered. The sensed level of the output current corresponds to that at the DC output 104.

Based on the sensed output voltage and current level, the control circuit 134 is configured to detect fast load transients. The control circuit 134 is configured to determine when the change in sensed level of the output voltage and output current exceeds (i.e., is greater than) a predetermined threshold.

For example, in the case of a rapid increase in load, the sensed value of the output current changes (i.e. increases) so that the change may exceed the predetermined threshold. Also, in this example, the sensed level of the output voltage changes (i.e., decreases) simultaneously with the output current, so the change may also exceed the predetermined threshold.

In another example, the sensed output current level changes (i.e. decreases) in the event of a rapid decrease in load and may thus exceed the predetermined threshold. In this example, the detected level of the output voltage changes (i.e. increases) simultaneously with the output current, so that the change can also exceed the predetermined threshold.

Accordingly, by detecting the output current level and the output voltage level in combination and comparing the change in the detected levels with a predetermined threshold value, the control circuit 134 can detect different types (e.g. rising and falling) of fast load transients and thus the control circuit 132 on the primary side of the Transformer circuit 112 provide valuable support.

For example, the different types of fast load transients can be identified by comparing the change in sensed levels to a look-up table with predetermined thresholds to determine the appropriate response to the fast load transient.

In any case, the control circuit 134 on the secondary side sends at least a second response signal to the control circuit 132 when the sensed level of the output voltage and/or the change in the sensed levels of the output voltage and/or the output current exceeds the predetermined threshold.

In an example implementation, the control circuit 134 sends a second response signal (designated FR_High in the figures) when the sensed level of the output current increases and the change (rate of change) thereof exceeds a predetermined threshold, and/or sends the same second response signal (designated FR_High denoted) when the sensed level of the output voltage falls and the change (rate of change) thereof exceeds another predetermined threshold.

In another example implementation, control circuit 134 sends another second response signal (referred to as FR_Tief in the figures) when the sensed level of the output current decreases and its change (rate of change) exceeds another predetermined threshold, and/or sends the same second response signal (as FR_Tief) when the level detected is the Off output voltage increases and its change (rate of change) exceeds a still further predetermined threshold. It should be noted that the current responds faster to a change in the load than the output voltage, which rises with a delay due to the output capacitor that is usually present. The output current is therefore advantageously monitored. Of course, the output voltage can also be monitored.

In the above, the determination of whether or not the change (rate of change) exceeds the threshold value is described as separate operations, namely for the detected level of the output voltage and for the detected level of the output current. However, they can also be carried out in combination.

For example, if the control circuit 134 determines that a sensed level of the output voltage or output current exceeds the threshold, it may also determine whether or not the other sensed level also exceeds the threshold. Depending on these conditional determination operations, the control circuit 134 can then be configured whether or not the corresponding one of the at least one second response signal is transmitted.

For example, the sensed output voltage level may reflect a load change (e.g., due to electrolytic capacitor 118) more slowly, but may still help the control circuit 134 determine whether a change in the sensed output current level actually justifies the control circuit 134 applying the appropriate des sends at least one second response signal or not.

The control circuit 134 on the secondary side thus sends two different second response signals to the control circuit 132 indicative of a fast load transient, based on which the control circuit 132 can control the switching element 126 to transfer more or less energy from the primary winding 112-1 to the secondary winding 112-2 of the transformer circuit 112 transmits.

In an example implementation, each of the two different second response signals is defined to correspond to a rising edge between two separate signal levels. A first level indicates that the change in the respectively detected level of the output voltage and/or the output current does not exceed the predetermined threshold (the signal is inactive), while the transition to a second level (rising edge) indicates that the change in the respectively detected level of the output voltage and/or output current exceeds the predetermined threshold (the signal is referred to as FR_High or FR_Low).

In an alternative exemplary implementation, the two different second response signals have more signal levels, which also indicate by how much the change in the respectively detected level of the output voltage and/or the output current exceeds the predetermined threshold value.

Depending on the at least one second response signal received, the primary-side control circuit 132 is set up to set the duration of the switching-off of the switching element 126 to a preconfigured point in time for the subsequent cycle.

In particular, this point in time is preconfigured in the control circuit 132 to be independent of the duration of the input voltage turn-off during the previous cycle. For example, the preconfigured time may set the duration of turning off the input voltage corresponding to the (absolute) time of a particular (e.g., first/second) zero voltage state (or first/second valley) detected by the zero crossing detection circuit 116-1 .

A fast load transient occurs independent of any cycle boundaries. Thus, the detection process by the control circuit 134 on the secondary side and the transmission to the control circuit 132 on the primary side is not necessarily completed before the control circuit 132 drives the switching element 126 in the current cycle. In other words, even if the fast load transient occurs in the previous cycle, a corresponding response of the primary control circuit 132 in the subsequent cycle cannot be guaranteed. Therefore, the control circuit 132 sets the duration of turning off the input voltage only for the subsequent cycle.

Such an embodiment of the control circuit 132 ensures that the switching element 126 is always actuated after the at least one second response signal has been received, specifically regardless of the current cycle. The second response signal affects (only) the OFF duration of the subsequent cycle. The second response signal can thus also be used during the starting operation. In particular, a pre-configured period of time may be used by the control circuit 132 regardless of whether or not the same control circuit 132 controlled the switching element 126 immediately before. Because the microcontroller is slower, multiple switching cycles are always performed with the same valley settings. The switch-on and switch-off times can only be changed slightly within a processor (control) cycle due to the signal V_feedback.

In an example implementation, the time duration in the control circuit 132 for the first of the at least one second response signal (referred to as FR_High) is preconfigured to correspond to a time of a first/second zero voltage state (or a first/second valley). This pre-configured amount of time may then be used by the control circuit 132 in response to receipt of the first of the at least one second response signal (referred to as FR_High) indicative of a fast load transient increase of the connected load.

In another example implementation, the time duration in control circuit 132 for the second of the at least one second response signal (referred to as FR_Low) is preconfigured to correspond to a twentieth/thirtieth zero voltage state (or twentieth/thirtieth valley) point in time. This preconfigured amount of time may then be used by the control circuit 132 in response to receipt of the second of the at least one second response signals (referred to as FR_Tief) indicative of a rapid load transient decrease in the connected load.

In another exemplary implementation, the control circuitry 132 is configured, upon receipt of the at least one second response signal, to interrupt any control operation in progress to extend the OFF duration, i. H. until the input voltage is switched on, for the next cycle. This interrupt processing in the control circuit 132 can ensure a quick response to the at least one second response signal (e.g., FR_High and FR_Low). In other words, upon receipt of the at least one second response signal, the control circuit 132 enters the interrupt processing to adjust the OFF-duration of the switching element 126 in the subsequent cycle, thereby avoiding a delay due to different control operations by the same control circuit 132.

• Unter Bezugnahme auf die 2a und 2b sollen die Vorteile des QR-Sperrwandlers 100 gemäß der beispielhaften Ausführungsform näher erläutert werden, und zwar im Zusammenhang mit einem beispielhaften Hochlastzustand, wie er in 2a dargestellt ist, und einem beispielhaften Niedriglastzustand, wie er in 2b dargestellt ist.

Now for an example operation of the QR flyback converter 100:

• Referring to the 2a and 2 B 1, the advantages of the QR flyback converter 100 according to the example embodiment will be discussed in more detail in the context of an example high load condition as illustrated in FIG 2a is shown, and an exemplary low load condition as shown in FIG 2 B is shown.

In both figures, the operation of the QR flyback converter 100 is shown in the form of a signal diagram over time, with the relevant signals influencing it being identified. Listed below each other are the drain-source voltage at switch 126 (labeled U_Drain_Switch), the first response signal (labeled Feedback) as received by control circuit 132 on the primary side, a second response signal (labeled FR_High in 2a and referred to as FR_Tief in 2 B ) as passed from the control circuit 134 on the secondary side to the control circuit 132 on the primary side, and the output current (designated I_OUT as also sensed by the control circuit 134 on the secondary side.

In 2a For example, the QR flyback converter 100 is initially open circuit and the output voltage (denoted as V_OUT) is held at a constant level. The drain-source voltage across switch 126 (referred to as U_Drain_Switch) is the voltage of electrolytic capacitor 110 when it is not switched. At this moment (standby) the output current I_AUS is equal to zero. Typically, this process can be rooted in a QR flyback converter 100 that is in an idle state, e.g. when it has its AC input 102 connected to an external power supply but not its DC output 104 connected to an external load.

Then the QR flyback converter 100 is connected to a load. At time t1, QR flyback converter 100 is still idling, but is undergoing a transition from a no-load state to a high-load state. This is evidenced by the output current ramping from a low to a high value. In other words, the QR flyback converter 100 has to deal with a fast load transient in this example. The time t1 is in 2a shown as an example. The exact The position of the time t1 depends on the cycle time of the secondary-side control circuit (e.g. in the range of 20 μs). Since the secondary-side control circuit has to compare two measured values, the point in time t1 is always somewhere in the rising area of the current curve C4.

The transition to the high-load state primarily affects the output current, which can lead to a breakdown in the output voltage at the electrolytic capacitor 118 .

Since the control circuit 134 on the secondary side continuously detects the level of the output voltage and/or the output current, it also detects an increasing level of the output current at time t1. Incidentally, this point in time t1 is only one of many recording points in time on which the following explanations focus.

Then, the control circuit 134 determines that the change in the level of the output current detected at time t1 exceeds a predetermined threshold. As already mentioned, the threshold value can be predetermined in the control circuit 134 in order to be able to determine an increase and/or a decrease in the load condition.

In response to this determination, the control circuit 134 on the secondary side sends at least a second response signal to the control circuit 132 on the primary side. For the increased load condition, the control circuit 134 sends a corresponding signal labeled FR_High to the control circuit 132. This at least one second response signal is transmitted from the control circuit 134 to the control circuit 132 via the Y capacitor 148.

At time t2, the control circuit 132 receives the at least one second response signal. In this example, the second response signal has the form of a pulse with a high signal level, a steep rising edge and a steep falling edge. This allows the control circuit 132 to easily recognize the time t2. It can be easily distinguished from the signal transients (noise) of the second response signal after time t3. These signal transients result from the continuous operation of the switching element 126, but have no influence on the function of the QR flyback converter 100.

In response to receiving the at least one second response signal at time t2, the control circuit 132 suspends its processing to extend the OFF duration for a subsequent cycle, i. H. the duration until the input voltage is switched on, to a preconfigured period of time. The preconfigured duration is independent of the OFF duration of the current cycle. In other words, even if the control circuit 132 has not yet started operation, it determines the OFF duration for the subsequent cycle according to a preconfigured value.

In this particular example, the OFF duration is adjusted by the control circuit 132 to correspond to the timing of the second zero voltage condition (or second valley). This can be seen in the figure during the subsequent cycle between times t3 and t4. In this subsequent cycle, the control circuit 132 actually controls the switching element 126 to switch from OFF to ON at a time corresponding to the detected time of the second zero voltage state (or valley).

In addition, the peak current level is defined by a reference voltage value, e.g. B. is set by hardware. The internal comparator of the primary-side control circuit 132 compares the voltage across the shunt 128 with the reference voltage value. If the shunt voltage exceeds the reference voltage value, the input voltage is switched off. The duration of the ON time t_AN thus results from the switch-off condition specified by the comparator.

In addition, a maximum duty cycle can be set. This maximum duty cycle can be viewed as a protective measure. If the controller does not correctly detect the peak current Ipeak, the input voltage will still be switched off after the maximum ON time has elapsed. This means that the device is not damaged even in the event of an error. However, the maximum switch-on time is set in such a way that, under normal conditions, the switch-off criterion I_peak is met first.

Normally the peak current level remains a constant predetermined value. However, the input voltage level can be added as an offset to a shunt voltage. Then the resulting peak voltage in each switching cycle is also dependent on the actually applied input voltage.

The control circuit 132 then controls the switching element 126 to switch from OFF to ON at any instant since each turn-on instant is synchronous. This switching ON occurs at the start after the control circuit 132 has set the OFF duration and determined the peak current level as previously described. After these two steps, it can be ensured that the control circuit 132 can continuously control the switching element 126 with the correct timing.

Since the control circuit 132 starts up, it does not have to meet the synchronous switching requirement. There are simply no times when the at least one zero voltage condition was detected in the previous cycle after which the turn-on time can be scheduled. The situation is different when the control circuit 132 is not in the starting phase but is in continuous operation.

Further, the control circuit 132 controls the turning off of the input voltage so that the current flowing in the primary winding 112-1 does not exceed the peak current level. As previously mentioned, the detected peak current level, along with the sensed current through shunt resistor 128 shifted by the first response signal, allows control circuit 132 to control the ON duration of switching element 126.

In this example, the control circuit 132 detects a current superimposed with a lower offset corresponding to a reduced first response signal. Due to the still falling feedback voltage, the number of valleys is reduced again by 1 about 90 µs after t4 in order to further increase the energy to be transmitted. At the same time, the decreasing displacement voltage at the shunt resulting from V_Feedback ensures that the ON time is also increased and thus the transmitted energy per switching cycle increases. As a result, the control circuit 132 controls the switching element 126 to have a longer ON duration in the current cycle than in the previous cycle, i.e. the cycle between times t3 and t4.

Upon turn-off at time t3, the current flowing in the primary winding 112-1 ceases and a limited amount of energy is transferred to the secondary winding 112-2 of the transformer circuit 112. Accordingly, after time t3, an increase in the drain-source voltage across switching element 126 (referred to as U_Drain_Switch) can be observed.

However, since only a finite amount of energy is being transferred, the drain-to-source voltage across switching element 126 (referred to as U_Drain_Switch) subsequently decreases, and first and second instants of a zero-voltage state (or first and second valleys) can be derived from the zero-crossing detection circuit 116-1.

Since, as stated, the turn-off time before the input voltage on the primary winding 112-1 is switched on again was determined by the control circuit 132 in the previous cycle, it is already configured in such a way that the switches are switched on synchronously, i. H. at the time of the second zero voltage state (or second valley).

Thus, at time t4, the control circuit 132 controls the timing of turning on the input voltage synchronously with the timing of the at least one zero voltage (or valley) condition. At time t5, switch 126 is turned off again.

In summary, the QR flyback converter 100 transfers a high amount of energy from the AC input 102 to the DC output 104 due to the longer ON duration caused by the decreasing displacement value on the shunt 128 and the shorter OFF duration. As can be seen from curve C1, 90 µs after time t5 the number of valleys before the switch is turned on again decreases again. This shows how long it takes for the microcontroller 132 to achieve a change in timing to turn the switch back on using only the V_feedback dependency. In contrast, the at least one second control circuit 134 allows the QR flyback converter 100 to respond quickly to load transients.

In 2 B At first, QR flyback converter 100 is in steady state operation, and the drain-to-source voltage across switching element 126 (called U_Drain_Switch) continuously oscillates between a high and a low voltage level. With this oscillating drain-source voltage, the QR flyback converter 100 generates a sufficiently high output current (referred to as I_OUT) to be required by a load connected to the DC output 104 .

In this example, QR flyback converter 100 initially operates in a high load condition, but then transitions to operation in a low load condition, as will become clear below. This is reflected in the output current dropping from a high current level to a low current level. In other words, the QR flyback converter has to cope with a fast load transient in this example as well.

The transition to the low-load state primarily affects the output current, which can also affect the output voltage.

Since the control circuit 134 on the secondary side continuously detects the level of the output voltage and/or the output current, it also detects a falling level of the output current at time t1. Incidentally, this point in time t1 is only one of many recording points in time on which the following explanations focus.

Then the control circuit 134 determines that the change in the level of the output current determined at time t1 exceeds a predetermined threshold value. As already mentioned, the threshold value can be predetermined in the control circuit 134 in order to be able to determine an increase and/or a decrease in the load condition.

In response to this determination, the control circuit 134 on the secondary side sends at least a second response signal to the control circuit 132 on the primary side. For the reduced load condition, the control circuit 134 sends a corresponding signal labeled FR_Tief to the control circuit 132. This at least one second response signal is transmitted from the control circuit 134 to the control circuit 132 via the Y capacitor 148.

At time t2, the control circuit 132 receives the at least one second response signal. In this example, the second response signal has the form of a pulse with a high signal level, a steep rising edge and a steep falling edge. This allows the control circuit 132 to easily recognize the time t2. It can be easily distinguished from the signal transients (noise) of the second response signal before and after time t2. These signal transients result from the continuous operation of the switching element 126, but do not affect the function of the QR flyback converter 100.

In response to receiving the at least one second response signal at time t2, the control circuit 132 suspends its processing to extend the off duration for a subsequent cycle, i. H. the duration of the switch-on of the input voltage, to a different pre-configured period of time. The pre-configured duration is independent of the switch-off duration of the current cycle. In other words, even if the control circuit 132 controls the OFF duration of the current cycle in increments or decrements of a preconfigured time interval, e.g. B. 2 * tV according to the first response signal transmitted from the voltage detector 124, it determines the OFF duration for the subsequent cycle as a pre-configured value.

In this particular example, the OFF duration is adjusted by the control circuit 132 to correspond to the time of the twentieth of the detected zero voltage conditions (or twentieth valley). This can be seen in the figure during the subsequent cycle between times t3 and t4. In this subsequent cycle, the control circuit 132 actually controls the switching element 126 to switch from OFF to ON at a time corresponding to the twentieth detected time of the twentieth zero voltage state (or twentieth valley).

According to the present invention, the peak current signal I_peak is only affected by the offset caused by the feedback signal. This displacement is small at high loads, so that as much energy as possible can be transferred. After load shedding, the feedback signal changes slowly and without the present invention, too much energy would be transferred for many switching cycles until the peak current is reduced to the point where the output voltage no longer increases. According to the invention, the second response signal (FR_Tief) provides the control circuit 132 with the information that the output power has been significantly reduced and therefore less energy has to be transmitted. The control circuit 132 responds as quickly as possible by increasing the duration of the OFF time. In this way, the transmitted energy can be reduced immediately while the charged energy remains the same.

Then, the control circuit 132 controls the switching element 126 to turn from OFF to ON at a timing that is a variation of the OFF duration of the previous cycle. The preset length of time only takes effect in the following cycle.

Thus, the control circuit 132 schedules the turn-on time synchronously with the time at which at least one zero voltage condition was detected in the previous cycle. In particular, the control circuit 132 adjusts the OFF-duration to turn-on by varying the OFF-duration of the previous cycle in increments or decrements of the preconfigured time interval 2*tV according to the first voltage detector 126 response signal.

Further, the control circuit 132 controls the turning off of the input voltage so that the switch 126 is turned off when the current flowing in the primary winding 112-1 reaches the peak current level. Reaching the peak current level is the turn-off condition. In particular, as previously mentioned, the predetermined peak current level, along with the sensed current through the shunt resistor 128 shifted by the first response signal, allows the control circuit 132 to control the ON duration of the switching element 126.

In this example, the control circuit 132 detects a current superimposed with an increasing offset corresponding to an increased first response signal. The increasing offset level results in an increase in the input voltage at the I_peak terminal of the control circuit 132 and hence a shorter ON time of the switch 126. Thus, a higher offset value results in a lower peak voltage through the primary winding 112-1.

Upon turn-off at time t3, the current flowing through the primary winding 112-1 ceases and a limited amount of energy is transferred to the secondary winding 112-2 of the transformer circuit 112.

In addition, the control circuit 132 has transitioned to turning on the switch 126 only at the twentieth valley, such that a first through twentieth instant of a zero voltage condition (or a first through the twentieth valley) is detected by the zero-crossing detection circuit 116-1 before turn-on occurs .

Since, as stated, the OFF duration before the input voltage at the primary winding 112-1 is switched on again was set by the control circuit 132 in the previous cycle, it is already configured in such a way that the switching element 126 is synchronous, i. H. turns ON at the time of the twentieth zero voltage state (or twentieth valley).

Thus, at time t4, the control circuit 132 controls the timing of turning on the input voltage synchronously with the timing of the at least one zero voltage (or valley) condition.

In summary, QR flyback converter 100 transfers a small amount of energy from AC input 102 to DC output 104 due to the shorter ON duration (caused by the first response signal) and longer OFF duration (caused by control circuit 132). Thus, the at least one second control circuit enables QR flyback converter 100 to respond quickly to load transients.

• Unter Bezugnahme auf 3 ist eine weitere beispielhafte Ausführungsform des quasiresonanten QR-Sperrwandlers 200 in Form einer schematischen Darstellung gezeigt.
• Auch dieses schematische Diagramm zeigt die Funktionsschaltungen des QR-Sperrwandlers als abstrakte Blöcke mit dazwischenliegenden Verbindungen, es ist also kein Schaltplan.

Now for another exemplary embodiment of a QR flyback converter:

• With reference to 3 Another exemplary embodiment of the quasi-resonant QR flyback converter 200 is shown in schematic diagram form.
• Again, this schematic diagram shows the functional circuits of the QR flyback converter as abstract blocks with interconnections in between, so it's not a schematic.

This form of representation was chosen to make the interaction between the functional circuits more visible.

The QR flyback converter 200 of the other exemplary embodiment is very similar to the QR flyback converter 100, and description of the same functional circuits or their interconnection has been omitted for the sake of brevity. In addition, the QR flyback converter 200 differs from the QR flyback converter 100 in that the primary-side control circuit 132 and the secondary side control circuit 134 exchange not only the second response signal, but also a data signal including the detected output voltage level and / or the detected output current level, the temperature level of individual secondary-side circuits, z. the electrolytic capacitor 118, and/or control information such as the threshold levels to be used by the secondary side control circuit 134.

To this end, the control circuit 134 and the control circuit 132 are coupled to enable data communication, optionally bi-directional data communication. In this example embodiment, the bi-directional communication may be via a universal asynchronous receiver-transmitter (UART) included in both control circuitry 132 and control circuitry 134 .

More specifically, the control circuit 134 is configured to transmit the data signal including the detected output voltage level and/or the detected output current level to the control circuit 132 on the primary side. In other words, the second response signal is a data signal (not a control signal) containing at least one of the detected output voltage level and the detected output current level as data. In this regard, the control circuit 132 is able to receive the data signal including the sensed output voltage level and/or the sensed output current level with a short transmission delay that is substantially shorter than the time delay in the first response signal.

In particular, the control circuit 134 is connected via an optocoupler 146 to the control circuit 132, which forwards the data signal in one direction and optionally a corresponding request in the other direction. In an alternative implementation, the connection can also be realized via separate Y-capacitors for the receiving and/or transmitting processes or in some other way, as long as the (galvanic) isolation limit is observed.

With the data signal containing the detected output voltage level and/or the detected output current level, the control circuit 132 is able to more precisely adjust the turn-off duration for each cycle since it obtains the exact levels of the load transients with a response time that depends only on the transmission delay depends. Again, the control cycle can set the off-time before the input voltage turns on, regardless of the off-time of the cycle, to a pre-configured time interval that matches the received data signal.

QUOTES INCLUDED IN DESCRIPTION

This list of documents cited by the applicant was generated automatically and is included solely for the better information of the reader. The list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Patent Literature Cited

• US 2010/0219802 A1 [0002]

Claims (15)

A quasi-resonant (QR) flyback converter (100) having an AC input (102) and a DC output (104), comprising: a transformer circuit (112) having a primary winding (112-1) configured to receive an input voltage from the AC input (102), a secondary winding (112-2) configured to provide an output voltage to the DC output (104), and an auxiliary winding (112-3) connected to a primary side zero crossing detection circuit (116- 1) is connected to at least one zero voltage condition in the auxiliary winding (112-3) and its time (t Tal - 6b ) capture; and a primary-side control circuit (132) configured to control a switching element (126) such that the switching element (126) in a current cycle (t_cycle - 6b ) the input voltage at the primary winding (112-1) switches ON and OFF sequentially, where: - a time (t0) of switching on the input voltage is the time at which the energy transfer into the transformer for the current cycle (t_cycle - 6b ) begins, - a point in time (t1 - 6b ) for switching off the input voltage is set so that a current flowing in the primary winding (112-1) does not exceed a peak current level, - a point in time (t5 - 6b ) to switch the input voltage on again for the next cycle (t_cycle +1 - 6b ) synchronous with the time (t Tal - 6b ) of the detected at least one zero-voltage state is set, and in addition - the OFF duration (t_OFF - 6b ) until the input voltage is switched on again by varying the OFF duration of the cycle (t_cycle - 6b ) in increments/decrements of a preconfigured time interval (2*tV - 6b ) is adjusted according to a first response signal transmitted from a secondary side voltage detector (124); further comprising: a secondary side control circuit (134) configured to sense the level of the output voltage and/or the level of an output current supplied to the DC output (104), and configured to transmit at least a second response signal to the primary side control circuitry (132) transmits when the change in the sensed output voltage level and/or the sensed output current level exceeds a predetermined threshold; and wherein in the event that the primary-side control circuit (132) receives the at least one second response signal from the secondary-side control circuit (134), the secondary-side control circuit (134) is configured to measure the OFF duration (t_OFF - 4b ) until the input voltage is switched on again, regardless of the OFF duration of the current cycle (t_cycle - 4b ) to a preconfigured time period (t4; t5 - 6b ) in accordance with the received at least one second response signal. QR flyback converter claim 1 , further comprising a primary-side rectifier circuit (108) connected between the AC input (102) and the primary winding (112-1), and a further secondary-side rectifier circuit (114) connected between the secondary winding (112-2) and the DC output ( 104) and/or a power factor correction stage (PFC). QR flyback converter claim 1 or 2 , further comprising a primary side electrolytic capacitor (110) configured to buffer the input voltage and a further secondary side electrolytic capacitor (118) configured to buffer the output voltage. QR flyback converter according to one of Claims 1 until 3 , wherein the at least one second response signal is transmitted from the secondary-side control circuit (134) via an optocoupler (146) or a Y-capacitor (146; 248) to the primary-side control circuit (132). QR flyback converter according to one of Claims 1 until 4 , wherein the primary side control circuit (132) is configured to control the peak current level for the subsequent cycle (t_cycle +1 - 4b ) in accordance with the level of current cycle (t_cycle - 4b ) received input voltage. QR flyback converter according to one of Claims 1 until 5 , wherein the primary-side control circuit (132) is configured such that it adjusts the time (t5) for the synchronous switch-on of the input voltage in one cycle so that it corresponds to: the time (t_Tal - 4b ) one of the at least one in the current cycle (t_cycle - 4b ) zero voltage conditions detected by the zero crossing detection circuit (116-1). QR flyback converter according to one of Claims 1 until 6 , where: the preconfigured time interval (2*tV - 4b ) in the primary-side control circuit (132) corresponding to a difference in time (t Tal - 4b ) is configured between two subsequently detected zero voltage conditions, and/or the pre-configured time period (t4; t5 - 4b ) in the primary-side control circuit (132) corresponding to the relative point in time (t Tal - 4b ) of a predetermined one of the at least one detected zero-voltage conditions. QR flyback converter claim 7 , wherein the pre-configured length of time is configured in the primary-side control circuit (132) according to a level of the input voltage from the AC input (102). QR flyback converter according to one of Claims 1 until 8th , wherein the primary-side control circuit (132) and optionally the secondary-side control circuit (134) is a microcontroller. QR flyback converter according to one of Claims 1 until 9 , wherein the primary-side control circuit (132) is configured such that, in the event of receipt of the at least one second response signal, it interrupts the control process for setting the OFF duration for a subsequent cycle. QR flyback converter according to one of Claims 1 until 10 , wherein the primary-side control circuit (132) is designed such that, - if a first of the at least one second response signal is received, the OFF duration of the subsequent cycle is adjusted according to a first preconfigured time interval, and - if a second of the at least one second response signal setting the OFF duration of the subsequent cycle to a second preconfigured time interval, the first preconfigured time interval being shorter than the second preconfigured time interval. QR flyback converter claim 11 , wherein the first preconfigured period of time (t4; t5) in the primary-side control circuit (132) according to the relative timing (t Tal - 4b ) a first or second of the at least one detected zero voltage conditions is configured within the previous cycle; and the second preconfigured period of time (t4; t5 - 4b ) in the primary-side control circuit (132) corresponding to the relative point in time (t Tal - 4b ) of a twentieth or thirtieth of the at least one sensed zero voltage conditions within the previous cycle. QR flyback converter according to one of Claims 1 until 12 wherein the secondary side control circuit (134) is further configured to transmit a data signal to the primary side control circuit (132). QR flyback converter Claim 13 , wherein the data is transmitted from the secondary-side control circuit (134) to the primary-side control circuit (132) via an optocoupler (250) or a Y-capacitor. QR flyback converter Claim 13 or 14 , wherein the data signal comprises the level of the output voltage and the level of an output current detected by the secondary-side control circuit (134).

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